Control circuit for high-frequency power amplification circuit, and high-frequency power amplification circuit

Abstract

According to the present invention, the waveform of a high-frequency signal transmitted through either an input-side or an output-side transmission line of a power amplifier that amplifies the high-frequency signal is picked up by a detector. A detection circuit detects the waveform picked up by the detector and outputs an amplitude-dependent physical quantity which is dependent on the amplitude of the waveform. An integration circuit time-integrates the amplitude-dependent physical quantity and outputs an integration-result physical quantity according to an integration value. An attenuation circuit attenuates the high-frequency signal transmitted through the output-side transmission line of the power amplifier on the basis of the integration-result physical quantity.

Description

Control circuit for high-frequency power amplification circuit and high-frequency power amplification circuit

[0001] The present invention relates to a control circuit for a high-frequency power amplifier circuit and to a high-frequency power amplifier circuit.

[0002] A high-frequency power transmission circuit is known that can protect power amplification elements from damage when load characteristics change and reflected power increases (Patent Document 1). In this circuit, the peak value of reflected power is monitored by a power detection circuit, and if the peak power of the reflected wave increases above a certain value, a signal attenuation circuit is inserted into the transmission line. With this configuration, the reflected peak power acting on the power amplification element can be attenuated and protected from damage.

[0003] Japanese Patent Publication No. 2006-166153

[0004] Even if the peak power of the traveling or reflected wave does not exceed a predetermined threshold, elements that can be destroyed by an increase in average power may be inserted into the transmission line. Protection by detecting peak power is insufficient to protect such elements. The object of the present invention is to provide a control circuit and a high-frequency power amplifier circuit that can suppress the destruction of elements due to an increase in average power.

[0005] According to one aspect of the present invention, a control circuit is provided comprising: a detector for detecting the waveform of a high-frequency signal transmitted through one of the transmission lines on the input and output sides of a power amplifier that amplifies a high-frequency signal; a detection circuit for detecting the waveform detected by the detector and outputting an amplitude-dependent physical quantity that depends on the amplitude of the waveform; an integration circuit for integrating the amplitude-dependent physical quantity over time and outputting an integrated physical quantity corresponding to the integration value; and an attenuation circuit for attenuating the high-frequency signal transmitted through the transmission line on the output side of the power amplifier based on the integrated physical quantity.

[0006] According to another aspect of the present invention, a high-frequency power amplification circuit is provided, comprising the control circuit, the power amplifier, and a filter inserted in the transmission line on the output side of the power amplifier.

[0007] By integrating amplitude-dependent physical quantities over time, an integrated physical quantity equivalent to the average power of a high-frequency signal over a certain period can be obtained. By controlling the attenuation circuit based on this integrated physical quantity, it becomes possible to suppress the destruction of elements due to an increase in average power.

[0008] Figure 1 is a block diagram of a high-frequency power amplifier circuit according to the first embodiment. Figure 2 shows the integrated physical quantity V. ３ and control signal V ４ This graph shows an example of the relationship. Figure 3 is an equivalent circuit diagram showing one configuration example of the detection circuit 20. Figure 4A is an equivalent circuit diagram showing another configuration example of the detection circuit 20, and Figure 4B is the voltage waveform V applied to the gate of transistor 20E. １ and current I ＲＥＣThis is a graph showing the relationship. Figure 5A is an equivalent circuit diagram showing one configuration example of the integrating circuit 30, and Figure 5B is an equivalent circuit diagram showing another configuration example of the integrating circuit 30. Figure 6A is an equivalent circuit diagram showing one configuration example of the threshold calculation circuit 41, and Figure 6B is an equivalent circuit diagram showing another configuration example of the voltage-to-current conversion circuit 41A of the threshold calculation circuit 41. Figure 7 is a block diagram of the high-frequency power amplifier circuit according to the second embodiment. Figure 8A is an equivalent circuit diagram showing one configuration example of the integrating circuit 30, Figure 8B is an equivalent circuit diagram showing an example of a variable resistance element 30A, and Figure 8C is an equivalent circuit diagram showing an example of a variable capacitance capacitor 30B. Figure 9A is an equivalent circuit diagram showing another configuration example of the integrating circuit 30, Figure 9B is an equivalent circuit diagram showing an example of a feedback resistance element 30H, and Figure 9C is an equivalent circuit diagram showing an example of a feedback capacitor 30G. Figure 10 is a block diagram and equivalent circuit diagram of the time constant adjustment circuit 50, the temperature sensor 60, and the integrating circuit 30. Figure 11 is the equivalent circuit diagram of the first adder circuit 52. Figure 12 is the block diagram and equivalent circuit diagram of the time constant adjustment circuit 50 and the integrating circuit 30. Figure 13 is the block diagram of the high-frequency power amplifier circuit according to the third embodiment. Figure 14 is the equivalent circuit diagram of the heat generation simulation circuit 70. Figure 15 is the block diagram of the high-frequency power amplifier circuit according to the fourth embodiment. Figure 16 is the block diagram of the high-frequency power amplifier circuit according to the fifth embodiment. Figure 17 is the block diagram of the high-frequency power amplifier circuit according to the sixth embodiment. Figure 18 shows the integrated result physical quantity V in the high-frequency power amplifier circuit according to the seventh embodiment. ３ and control signal V ４ This graph shows the relationship. Figure 19 is an equivalent circuit diagram of the threshold calculation circuit 41 used in the high-frequency power amplifier circuit according to the seventh embodiment.

[0009] [First Embodiment] A high-frequency power amplifier circuit and its control circuit according to the first embodiment will be described with reference to the drawings from Figure 1 to Figure 6B.

[0010] Figure 1 is a block diagram of a high-frequency power amplifier circuit according to the first embodiment. (High-frequency signal RF) ＩＮA high-frequency signal transmitted through a transmission line on the output side of a power amplifier 80 that amplifies is input to a filter selection switch 81. The filter selection switch 81 selects one filter 82 from a plurality of filters 82. The high-frequency signal input to the filter selection switch 81 is input to the one filter 82 selected by the filter selection switch 81.

[0011] The high-frequency signal that has passed through the filter 82 selected by the antenna switch 83 is output from an antenna terminal 84. A high-frequency signal RF output from the antenna terminal 84 ＯＵＴ is supplied to an antenna and radiated from the antenna. As the filter 82, for example, a surface acoustic wave (SAW) filter using an elastic material, a bulk acoustic wave (BAW) filter, or the like is used.

[0012] A detector 10 is coupled to a transmission line on the output side of the power amplifier 80. The detector 10 detects the waveform of a traveling wave of the high-frequency signal transmitted through the transmission line. As the detector 10, for example, a directional coupler is used. Furthermore, a first transistor 42 is inserted in series into the transmission line on the output side of the power amplifier 80. The position where the first transistor 42 is inserted is on the downstream side of the traveling wave relative to the position where the detector 10 is coupled. That is, the detector 10 is coupled to the transmission line between the output node of the power amplifier 80 and the first transistor 42.

[0013] The waveform of the high-frequency signal detected by the detector 10 is input to a detection circuit 20. The waveform of the high-frequency signal input to the detection circuit 20 is a voltage waveform. This voltage waveform is denoted as V １ . The detection circuit 20 receives the input voltage waveform V １ and outputs an amplitude-dependent physical quantity V that depends on the amplitude of ２ . The amplitude-dependent physical quantity V ２ is, for example, a voltage, and the amplitude of the voltage waveform V １ and the amplitude-dependent physical quantity V ２ have a positive correlation. Therefore, as the power of the high-frequency signal increases, the amplitude-dependent physical quantity V ２ also increases. The amplitude-dependent physical quantity V ２ is input to an integration circuit 30.

[0014] The integrating circuit 30 receives the amplitude-dependent physical quantity V input from the detection circuit 20. ２ The integral is performed over time, and the resulting physical quantity V is obtained based on the value of the integral. ３ Outputs the integrated physical quantity V. ３ For example, voltage. The resulting physical quantity V ３ The signal is input to the damping circuit 40. The damping circuit 40 receives the result of the integration, the physical quantity V. ３ Based on this, the transmission line on the output side of the power amplifier 80 is attenuated to transmit high-frequency signals.

[0015] The attenuation circuit 40 includes a threshold calculation circuit 41 and a first transistor 42. For example, an NMOSFET is used as the first transistor 42, with the source of the first transistor 42 connected to the output node of the power amplifier 80 and the drain connected to the filter selection switch 81. The attenuation circuit 40 controls the integral result physical quantity V ３ Based on this, the transmission line on the output side of the power amplifier 80 is attenuated to transmit high-frequency signals.

[0016] The threshold calculation circuit 41 calculates the integral result physical quantity V ３ Based on this, control signal V controls the conduction state of the first transistor 42. ４ Outputs control signal V. ４ This is applied to the gate of the first transistor 42. That is, the control signal V ４ This is a voltage signal applied to the gate of the first transistor 42. When the gate voltage of the first transistor 42 is changed, the resistance between the source and drain of the first transistor 42 changes. Increasing the resistance between the source and drain of the first transistor 42 increases the attenuation of the high-frequency signal transmitted through the transmission line.

[0017] Figure 2 shows the resulting physical quantity V. ３ and control signal V ４ This graph shows an example of the relationship. The horizontal axis represents the integrated physical quantity V. ３ This graph represents the magnitude (voltage value) of the signal, with the vertical axis representing the control signal V. ４ This represents the magnitude (voltage value). The integral result is the physical quantity V. ３ The limit start threshold V ３ＴＨWithin the following range, the threshold calculation circuit 41 controls the signal V to turn on the first transistor 42. ４ It outputs [this value]. Therefore, the attenuation circuit 40 does not attenuate the high-frequency signal transmitted through the transmission line.

[0018] Integration result physical quantity V ３ The limit start threshold V ３ＴＨ When it exceeds this value, the threshold calculation circuit 41 calculates the integral result physical quantity V ３ As the control signal V increases ４ The control signal V is gradually reduced. ４ When V decreases, the resistance of the first transistor 42 increases, and the high-frequency signal transmitted through the transmission line is attenuated. That is, the integral physical quantity V ３ The limit start threshold V ３ＴＨ Beyond the range, the damping circuit 40 controls the resulting physical quantity V. ３ As the value increases, the attenuation of high-frequency signals gradually increases.

[0019] Next, the specific configuration of the detection circuit 20 will be described with reference to Figures 3, 4A, and 4B.

[0020] Figure 3 is an equivalent circuit diagram showing one example configuration of the detection circuit 20. A diode 20A is connected between the input node and the output node of the detection circuit 20. For example, the diode 20A is connected so that the direction from the input node to the output node is forward. A capacitor 20B is connected between the output node and the reference potential (ground potential). The diode 20A extracts the positive side of the voltage waveform, and the capacitor 20B removes the fundamental wave component. As a result, the voltage waveform V is extracted from the output node. １ V is an amplitude-dependent physical quantity that depends on the amplitude of the amplitude. ２ The following will be output.

[0021] Figure 4A is an equivalent circuit diagram showing another configuration example of the detection circuit 20. Voltage waveform V １ However, this is input to the gate of the NMOSFET transistor 20E via the DC-cut capacitor 20C. The bias circuit 20F applies a bias voltage to the gate of the transistor 20E via the resistor element 20D.

[0022] The power supply voltage is applied to the drain of transistor 20E, and its source is connected to a reference potential via resistor element 20G. Voltage waveform V １ Current I is supplied to the resistive element 20G according to the magnitude of the voltage. ＲＥＣ A current flows. A voltage equivalent to the voltage drop across the resistor 20G is generated at the source of the transistor 20E. This voltage is filtered by the fundamental wave component removal filter 20H, and the amplitude-dependent physical quantity V is reduced. ２ It will be output as follows.

[0023] Figure 4B shows the voltage waveform V applied to the gate of transistor 20E. １ and current I ＲＥＣ This graph shows the relationship between the voltage waveform V. The horizontal axis represents the voltage waveform V. １ This graph shows the magnitude of the voltage, with the vertical axis representing the current I. ＲＥＣ This represents the voltage waveform V. １ As the voltage increases, the current I ＲＥＣ Although it also increases, the relationship between the two is nonlinear. Voltage waveform V applied to the gate of transistor 20E １ This is the bias voltage V １０ It changes periodically around this point.

[0024] Voltage waveform V with small amplitude １Ｓ When this is input, the effect of nonlinearity is almost invisible. Therefore, the voltage waveform V １Ｓ The current I that flows when the input is received １Ｓ The average value I ＡＶＥＳ The voltage waveform V １Ｓ The average value (i.e., bias voltage V) １０ ) Corresponding current I ＲＥＣ０ It becomes equal to.

[0025] Large amplitude voltage waveform V １Ｌ When this is input, the effects of nonlinearity appear, and the voltage waveform V １Ｌ Current I when it touches in the positive direction １Ｌ The current I when the amplitude is in the negative direction １Ｌ It becomes larger than the amplitude. Therefore, current I １Ｌ The average value I ＡＶＥＬ However, the voltage waveform V has a small amplitude. １Ｓ Current I when input １Ｓ The average value IＡＶＥＳ It will get bigger.

[0026] The amplitude-dependent physical quantity V output through the fundamental wave component removal filter 20H ２ The voltage value is equal to the current I １Ｓ , I １Ｌ Average value I ＡＶＥＳ , I ＡＶＥＬ It depends on the voltage waveform V. １ As the amplitude increases, the amplitude-dependent physical quantity V ２ It gets bigger.

[0027] Next, the specific configuration of the integrating circuit 30 (Figure 1) will be described with reference to Figures 5A and 5B.

[0028] Figure 5A is an equivalent circuit diagram showing one example configuration of the integrating circuit 30. A π-type RC low-pass filter composed of capacitors 30B and 30C and a resistor element 30A can be used as the integrating circuit 30. Alternatively, a T-type or L-type RC low-pass filter may be used instead of the π-type RC low-pass filter.

[0029] Figure 5B is an equivalent circuit diagram showing another configuration example of the integrating circuit 30. The integrating circuit 30 uses a circuit including an operational amplifier 30D. Amplitude-dependent physical quantity V ２ The signal is input to the inverting input node of the operational amplifier 30D via the resistor element 30E. The non-inverting input node of the operational amplifier 30D is connected to a reference potential via the pull-down resistor element 30F. A feedback resistor element 30H and a feedback capacitor 30G are connected between the inverting input node and the output node of the operational amplifier 30D. The integrated physical quantity V is input from the output node of the operational amplifier 30D. ３ The following will be output.

[0030] Next, the specific configuration of the threshold calculation circuit 41 will be described with reference to Figures 6A and 6B.

[0031] Figure 6A is an equivalent circuit diagram showing one example configuration of the threshold calculation circuit 41. The integral result physical quantity V, which is a voltage value, is transmitted from the integrating circuit 30 to the voltage-to-current conversion circuit 41A. ３ The input is V. The voltage-current conversion circuit 41A converts the integral result physical quantity V. ３ current I ３is converted into. The configuration of the voltage-current conversion circuit 41A will be described below.

[0032] An integration result physical quantity V is input to the inverting input node of the operational amplifier 41AA ３ . Gates of two PMOSFETs of the current mirror circuit 41AB formed of PMOSFETs are connected to the output node of the operational amplifier 41AA, and sources thereof are connected to a power supply voltage. A drain of the PMOSFET on the reference current side is connected to a reference potential via the resistance element 41AC. A voltage corresponding to the voltage drop across the resistance element 41AC is input to the non-inverting input node of the operational amplifier 41AA.

[0033] An output current of the current mirror circuit 41AB constitutes a reference current for a current mirror circuit 41AD formed of NMOSFETs. An output current of the current mirror circuit 41AD is a current I output from the voltage-current conversion circuit 41A ３ .

[0034] The current I output from the voltage-current conversion circuit 41A ３ constitutes a reference current for a current mirror circuit 41B formed of PMOSFETs. An output current of the current mirror circuit 41B is equal to the current I ３ . From the current I ３ , the extraction circuit 41C extracts a current corresponding to a threshold determination current I ３ＴＨ .

[0035] The extraction circuit 41C includes a threshold determination current source 41D that outputs the threshold determination current I ３ＴＨ , and an NMOS current mirror circuit that copies and outputs the threshold determination current I ３ＴＨ . A current corresponding to the output current of this NMOS current mirror circuit is extracted from the current I ３ .

[0036] A remaining current I obtained by extracting the threshold determination current I ３ from the current I ３ＴＨ is I ３ -I ３ＴＨ , which constitutes a reference current for a current mirror circuit 41E formed of NMOSFETs. A mirror ratio of the current mirror circuit 41E is n, and an output current of the current mirror circuit 41E is a current I ３-I ３ＴＨ It becomes n times that. Current I ３ -I ３ＴＨ The current that flows is n times the original current. The voltage, which is lower than the power supply voltage by the voltage drop across the pull-up resistor element 41F, is the control signal V. ４ It will be output as follows.

[0037] current I ３ The threshold determination current I ３ＴＨ In the following cases, no reference current flows through the current mirror circuit 41E. Therefore, the control signal V ４ The voltage becomes equal to the power supply voltage. This state corresponds to the integrated physical quantity V shown in Figure 2. ３ The limit start threshold V ３ＴＨ This corresponds to the following range: Limit start threshold V ３ＴＨ The threshold determination current I ３ＴＨ It is determined by [something].

[0038] current I ３ The threshold determination current I ３ＴＨ When it exceeds this value, current flows through the current mirror circuit 41E, causing the control signal V ４ The voltage drops below the power supply voltage. Current I ３ As the control signal V increases, ４ The decrease in the integral result physical quantity V shown in Figure 2 becomes larger. ３ The limit start threshold V ３ＴＨ This corresponds to the range exceeding the control signal V. ４ The slope of the decrease is determined by the mirror ratio n of the current mirror circuit 41E.

[0039] Figure 6B is an equivalent circuit diagram showing another configuration example of the voltage-to-current conversion circuit 41A. The non-inverting input node of the operational amplifier 41AE receives the integrated physical quantity V from the integrating circuit 30. ３The following is input. The gates of the two NMOSFETs of the current mirror circuit 41AG, which consists of NMOSFETs, are connected to the output node of the operational amplifier 41AE, and the source is connected to the reference potential. The drain of the NMOSFET on the reference current side is connected to the power supply voltage via the resistor element 41AF. A voltage equivalent to the voltage drop across the resistor element 41AF is input to the inverting input node of the operational amplifier 41AE. The output current of the current mirror circuit 41AG is the current I output from the voltage-to-current conversion circuit 41A. ３ This is the result.

[0040] Next, the excellent effects of the first embodiment will be described. When the power of the high-frequency signal going from the power amplifier 80 (Figure 1) to the filter selection switch 81 (Figure 1) increases, the voltage waveform V １ The amplitude increases. As a result, the amplitude-dependent physical quantity V output from the detection circuit 20 becomes larger. ２ The value increases. When the average power of a high-frequency signal over a certain time interval increases, the integrated physical quantity V output from the integrating circuit 30 increases. ３ The value becomes larger. The integration time constant of the integration circuit 30 is set according to the time width over which the average power of the high-frequency signal should be calculated.

[0041] Average power of a high-frequency signal (integral result physical quantity V) ３ ) is the limit start threshold V shown in Figure 2. ３ＴＨ When the power exceeds a certain level, the gate voltage of the first transistor 42 (Figure 1) decreases, and the resistance between the source and drain of the first transistor 42 increases. As a result, the high-frequency signal is attenuated, and the power of the high-frequency signal input to the filter 82 via the filter selection switch 81 is kept low. This suppresses the breakdown of the filter 82 caused by an increase in the average power of the high-frequency signal.

[0042] Filters made of elastic materials are susceptible to damage due to increases in the average power of high-frequency signals. In a method that attenuates high-frequency signals according to their peak power, the filter cannot be adequately protected if the peak power does not exceed a predetermined threshold, but the average power exceeds the threshold that would damage the filter. In the first embodiment, since the high-frequency signal is attenuated based on the average power of the high-frequency signal, filter damage caused by an increase in the average power of the high-frequency signal can be suppressed.

[0043] The average power of the high-frequency signal is the limiting threshold V shown in Figure 2. ３ＴＨ As long as the power does not exceed the equivalent power, the first transistor 42 (Figure 1) is turned on, so the impact on high-frequency signal transmission due to the addition of the attenuation circuit 40 (Figure 1) is minimal.

[0044] Furthermore, in the first embodiment, the average power of the high-frequency signal is the limiting start threshold V shown in Figure 2. ３ＴＨ If the power exceeds a certain level, the first transistor 42 (Figure 1) is gradually switched from the ON state to the OFF state. As a result, communication is not completely interrupted, and communication can continue.

[0045] Furthermore, in the first embodiment, the threshold determination current I output by the threshold determination current source 41D (Figure 6A) ３ＴＨ By adjusting the limit start threshold V ３ＴＨ (Figure 2) can be adjusted. Also, by adjusting the mirror ratio n of the current mirror circuit 41E (Figure 6A), the resulting physical quantity V can be adjusted. ３ The limit start threshold V ３ＴＨ Beyond the range, the resulting physical quantity V ３ Control signal V when it becomes large ４ The slope of the decrease can be adjusted.

[0046] [Second Embodiment] Next, a high-frequency power amplifier circuit and control circuit according to the second embodiment will be described with reference to Figures 7 to 12. Hereinafter, the configuration common to the high-frequency power amplifier circuit and control circuit according to the first embodiment, described with reference to Figures 1 to 6B, will be omitted from the explanation.

[0047] Figure 7 is a block diagram of a high-frequency power amplifier circuit according to the second embodiment. In the first embodiment (Figure 1), the time constant of the integrating circuit 30 is fixed. In contrast, in the second embodiment, the time constant of the integrating circuit 30 is variable. A time constant adjustment signal is input to the time constant adjustment circuit 50, and the time constant adjustment circuit 50 changes the time constant of the integrating circuit 30 based on the input time constant adjustment signal.

[0048] The temperature sensor 60 measures the ambient temperature. Based on the temperature measurement, the temperature sensor 60 outputs a temperature-dependent time constant adjustment signal. The processor 61 controls the filter selection switch 81 and the antenna switch 83 based on the frequency band information of the high-frequency signal to be amplified, and also outputs a band-dependent time constant adjustment signal. The time constant adjustment signals output from the temperature sensor 60 and the processor 61 are input to the time constant adjustment circuit 50. The time constant adjustment circuit 50 changes the time constant of the integrating circuit 30 based on the temperature-dependent time constant adjustment signal and the band-dependent time constant adjustment signal.

[0049] Figure 8A is an equivalent circuit diagram showing one configuration example of the integrating circuit 30. In the integrating circuit according to the first embodiment (Figure 5A), the resistance value of the resistor element 30A and the capacitances of capacitors 30B and 30C are fixed values. In contrast, in the second embodiment, the resistance value of the resistor element 30A and the capacitances of capacitors 30B and 30C in the integrating circuit 30 (Figure 8A) are variable. Changing the resistance value and capacitance changes the time constant of the integrating circuit 30.

[0050] Figure 8B is an equivalent circuit diagram showing an example of a variable resistance element 30A. An NMOSFET can be used as the resistance element 30A. The resistance value can be changed by changing the gate voltage of the NMOSFET.

[0051] Figure 8C is an equivalent circuit diagram showing an example of a variable capacitance capacitor 30B. The configuration of the other capacitor 30C is the same as that of capacitor 30B. Capacitor 30B has a configuration in which a fixed capacitance capacitor 30BA and a varicap 30BB are connected in series. The cathode of varicap 30BB is connected to capacitor 30BA. By changing the voltage applied to the cathode of varicap 30BB, the capacitance of capacitor 30B can be changed.

[0052] Figure 9A is an equivalent circuit diagram showing another configuration example of the integrating circuit 30. In the integrating circuit according to the first embodiment (Figure 5B), the capacitance of the feedback capacitor 30G and the resistance of the feedback resistor 30H are fixed values. In contrast, in the second embodiment, the capacitance of the feedback capacitor 30G and the resistance of the feedback resistor 30H in the integrating circuit 30 (Figure 9A) are variable. Changing the resistance and capacitance changes the time constant of the integrating circuit 30.

[0053] Figure 9B is an equivalent circuit diagram showing an example of a feedback resistor element 30H. Three circuit elements, each consisting of a fixed-resistance resistor element and a switching element connected in parallel, are connected in series. For example, an NMOSFET is used as the switching element. By controlling the switching element on and off, the resistance value of the feedback resistor element 30H can be changed.

[0054] Figure 9C is an equivalent circuit diagram showing an example of a feedback capacitor 30G. Two circuit elements, each consisting of a capacitor with fixed capacitance and a switching element connected in series, are connected in parallel. For example, an NMOSFET is used as the switching element. By controlling the switching element on and off, the capacitance of the feedback capacitor 30G can be changed.

[0055] The integrating circuit 30 shown in Figure 8A changes the time constant by changing the applied voltage in an analog manner. In contrast, the integrating circuit 30 shown in Figure 9A changes the time constant by controlling the on / off state of the switching element with a digital signal.

[0056] Next, referring to Figures 10 and 11, the configuration of the time constant adjustment circuit 50 when the integrating circuit 30 shown in Figures 8A, 8B, and 8C is adopted will be described.

[0057] Figure 10 shows the block diagram and equivalent circuit diagram of the time constant adjustment circuit 50, temperature sensor 60, and integration circuit 30. The temperature sensor 60 includes a constant current source 60A and a temperature-dependent diode 60B. The forward resistance of the temperature-dependent diode 60B changes depending on the ambient temperature. The output voltage (temperature-dependent time constant adjustment voltage V) changes according to this change in resistance. Ｔ The frequency band information changes. A digital signal identifying the frequency band information is input from the processor 61 to the AD converter 51. The AD converter 51 converts this digital signal into an analog signal and outputs a voltage (band-dependent time constant adjustment voltage V) corresponding to the frequency band information. Ｐ ) Output.

[0058] The first adding circuit 52 controls the first fixed voltage V Ａ and temperature-dependent time constant adjustment voltage V Ｔ and band-dependent time constant adjustment voltage V Ｐ The first summing voltage V is obtained by adding the two together. Ａ +V Ｔ +V Ｐ The second adder circuit 53 outputs the second fixed voltage V. Ｂ and temperature-dependent time constant adjustment voltage V Ｔ and band-dependent time constant adjustment voltage V Ｐ The second summing voltage V is obtained by adding the two together. Ｂ +V Ｔ +V Ｐ Outputs.

[0059] The integrating circuit 30 includes a resistive element 30A made of NMOSFETs, and capacitors 30B and 30C. The first summing voltage V is applied to the cathodes of the varicaps of capacitors 30B and 30C. Ａ +V Ｔ +V Ｐ A second summing voltage V is applied. Ｂ +V Ｔ +V Ｐ It is applied.

[0060] Figure 11 is an equivalent circuit diagram of the first adder circuit 52. The other second adder circuit 53 also has the same circuit configuration as the first adder circuit 52.

[0061] First fixed voltage V Ａ and temperature-dependent time constant adjustment voltage V Ｔ These are applied to the non-inverting input node of the operational amplifier 52A via resistors R1 and R2, respectively. The voltage obtained by dividing the voltage at the output node of the operational amplifier 52A and the reference potential by resistors R3 and R4 is applied to the inverting input node of the operational amplifier 52A. With this configuration, the voltage at the output node of the operational amplifier 52A is V Ａ +V Ｔ It becomes equal to.

[0062] Voltage V Ａ +V Ｔ and band-dependent time constant adjustment voltage V Ｐ These are applied to the non-inverting input node of the operational amplifier 52B via resistors R5 and R6, respectively. The voltage obtained by dividing the voltage at the output node of the operational amplifier 52B and the reference potential by resistors R7 and R8 is applied to the inverting input node of the operational amplifier 52B. With this configuration, the voltage at the output node of the operational amplifier 52B is V Ａ +V Ｔ +V Ｐ It becomes equal to.

[0063] Next, referring to Figure 12, the configuration of the time constant adjustment circuit 50 when the integrating circuit 30 shown in Figures 9A, 9B, and 9C is adopted will be described.

[0064] Figure 12 shows the block diagram and equivalent circuit diagram of the time constant adjustment circuit 50 and the integrating circuit 30. The integrating circuit 30 has a configuration similar to the π-type RC integrating circuit shown in Figure 8A. The resistive element 30A of the integrating circuit 30 includes a circuit element in which a resistive element and a switching element are connected in parallel, similar to the feedback resistive element 30H shown in Figure 9B. In Figure 9B, three circuit elements are connected in series, but in the example shown in Figure 12, two circuit elements are connected in series. The capacitors 30B and 30C have a configuration similar to the feedback capacitor 30G shown in Figure 9C, in which two circuit elements in which a capacitor and a switching element are connected in series are connected in parallel.

[0065] The time constant adjustment circuit 50 outputs a 2-bit digital signal that controls the switching elements of the resistor 30A, capacitor 30B, and capacitor 30C based on the frequency band information provided by the processor 61. This digital signal can take on one of four values: 00, 01, 10, or 11.

[0066] Two bits of digital information are supplied to the two switching elements of the resistor element 30A, thereby controlling the on / off state of the two switching elements. One bit of information is inverted by the knot circuit 30K and then supplied to one of the switching elements of capacitors 30B and 30C, while the other bit of information is inverted by the knot circuit 30L and then supplied to the other switching element of capacitors 30B and 30C.

[0067] Based on frequency band information provided by the processor 61, the time constant adjustment circuit 50 determines the value of the digital signal and controls the on / off state of the switching element, thereby changing the time constant of the integrating circuit 30.

[0068] The temperature-dependent time constant adjustment voltage V is output from the temperature sensor 60 shown in Figure 10. Ｔ The signal is converted using AD conversion, and a temperature-dependent time constant adjustment voltage V is applied. Ｔ The configuration may also be used to determine the value of the digital signal, taking this into consideration.

[0069] Next, the excellent effects of the second embodiment will be described. If the time constant of the integrating circuit 30 becomes too long, the response characteristics of the attenuation circuit 40 (Figure 7) will deteriorate. If the time constant becomes too short, the voltage waveform V １ The peak value of the integral result physical quantity V ３ This is largely reflected in the integral physical quantity V. ３ As a result, the peak power is reflected more heavily than the average power of the high-frequency signal output from the power amplifier 80. Therefore, attenuation control based on average power is no longer performed.

[0070] In the second embodiment, the time constant can be appropriately adjusted based on the ambient temperature and the frequency band of the high-frequency signal to be amplified. This enhances the effect of protecting the filter 82. For example, the integrating circuit 30 can be controlled so that the time constant increases as the frequency increases. Also, since the breakdown resistance of the filter 82 decreases as the ambient temperature increases, the integrating circuit 30 can be controlled so that the time constant decreases.

[0071] Next, a modification of the second embodiment will be described. In the second embodiment, a 2-bit digital signal was used to control the integrating circuit 30, but a digital signal with 3 or more bits may also be used. The number of circuit elements constituting the resistor 30A, capacitor 30B, and 30C should be increased according to the number of bits. Increasing the number of bits in the digital signal allows for finer control of the time constant.

[0072] In the second embodiment, the temperature sensor 60 (Figure 10) is configured to measure the ambient temperature, but it may also be configured to measure the temperature of the filter 82. For example, the temperature-dependent diode 60B (Figure 10) of the temperature sensor 60 may be placed in a location affected by the temperature change of the filter 82. In this way, the time constant can be appropriately adjusted in accordance with the temperature change of the filter 82.

[0073] [Third Embodiment] Next, a high-frequency power amplifier circuit and control circuit according to the third embodiment will be described with reference to Figures 13 and 14. Hereinafter, the description of components common to the high-frequency power amplifier circuit and control circuit according to the second embodiment, which were described with reference to Figures 7 to 12, will be omitted.

[0074] Figure 13 is a block diagram of a high-frequency power amplifier circuit according to the third embodiment. In the second embodiment (Figure 7), the ambient temperature is measured by a temperature sensor 60, and the time constant of the integrating circuit 30 is adjusted based on the temperature measurement result. In contrast, in the third embodiment, instead of measuring the actual temperature, the heat generation simulation circuit 70 simulates heat generation based on the power of the high-frequency signal output from the power amplifier 80, and the time constant adjustment signal V is generated based on the estimated temperature due to the simulated heat generation. ＴＳ This is output to the integrating circuit 30.

[0075] The heat generation simulation circuit 70 receives an amplitude-dependent physical quantity V from the detection circuit 20. ２ Heat generation is electrically simulated based on this. Furthermore, a time constant adjustment signal V is generated based on the temperature estimate obtained by simulating heat generation. ＴＳ This generates and provides it to the time constant adjustment circuit 50.

[0076] Figure 14 is an equivalent circuit diagram of the heat generation simulation circuit 70. The heat generation simulation circuit 70 includes a voltage-to-current conversion circuit 71, a current mirror circuit 72, a temperature simulation current source 73, and a temperature simulation capacitor 74. The voltage-to-current conversion circuit 71 receives an amplitude-dependent physical quantity V from the detection circuit 20. ２ Converts to an electric current, and amplitude-dependent current I ２ The temperature simulation current source 73 outputs the temperature simulation threshold current I. ＴＴＨ Outputs the following: Temperature simulation threshold current I ＴＴＨ This constitutes the reference current of the current mirror circuit 72. The output current of the current mirror circuit 72 is the temperature simulation threshold current I ＴＴＨ It is equal to.

[0077] amplitude dependent current I ２ The threshold current I used for temperature simulation ＴＴＨ When it is greater than, amplitude-dependent current I ２ From temperature simulation threshold current I ＴＴＨ The remaining current I after it has been removed ２ -I ＴＴＨ The temperature simulation capacitor 74 is charged by this. Current I flows into the temperature simulation capacitor 74. ２ -I ＴＴＨ Accordingly, the voltage across the terminals of the temperature simulation capacitor 74 increases. Amplitude-dependent physical quantity V ２ As it increases, the current I ２ -I ＴＴＨ The time constant adjustment signal V also increases. ＴＳ The rate of increase becomes faster. Note that the time constant adjustment signal V ＴＳ The upper limit is restricted by the power supply voltage of the voltage-current conversion circuit 71.

[0078] Amplitude-dependent physical quantity V ２This reflects the power of the high-frequency signal output from the power amplifier 80 (Figure 13). As the power of the high-frequency signal increases, the amount of heat generated also increases, so the time constant adjustment signal V ＴＳ This can be considered a simulation of a temperature rise due to heat generation. Furthermore, the longer the period of high-frequency signal power, the greater the temperature rise. This is because the charging time of the temperature simulation capacitor 74 continues for a longer period, resulting in a time constant adjustment signal V ＴＳ This corresponds to a larger increase in the value.

[0079] amplitude dependent current I ２ The threshold current I used for temperature simulation ＴＴＨ When it is smaller, current I ２ -I ＴＴＨ This becomes negative. This means that the temperature simulation capacitor 74 is discharging. The discharge of the temperature simulation capacitor 74 occurs when the power of the high-frequency signal is equal to the temperature simulation threshold current I ＴＴＨ This simulates a decrease in temperature due to natural heat dissipation when the power is below a certain level. Note that the time constant adjustment signal V ＴＳ The lower limit is restricted by the reference potential.

[0080] The time constant adjustment circuit 50 receives the time constant adjustment signal V ＴＳ The time constant of the integrating circuit 30 (Figure 13) is changed based on this. For example, the time constant adjustment signal V ＴＳ When the voltage increases, control is performed to shorten the time constant.

[0081] Next, the excellent effects of the third embodiment will be described. When the power of the high-frequency signal increases and the temperature of the filter 82 (Figure 13) rises, the filter 82 becomes more susceptible to damage. In the third embodiment, the rise and fall in temperature of the filter 82 in response to the power of the high-frequency signal are simulated using an electrical circuit. When the simulated temperature of the filter 82 rises, the time constant of the integrating circuit 30 is shortened, thereby speeding up the response of the filter protection. As a result, the filter 82 can be appropriately protected according to its temperature.

[0082] Conversely, if the temperature of the filter 82 is assumed to be sufficiently low, the time constant of the integrating circuit 30 can be increased to prevent excessive attenuation of high-frequency signals when the output of the power amplifier 80 temporarily increases.

[0083] Next, a high-frequency power amplification circuit and control circuit according to a modification of the third embodiment will be described. In this modification, a resistive element is connected in parallel with the temperature simulation capacitor 74 instead of the current mirror circuit 72. This resistive element will be called a discharge resistor. An amplitude-dependent current I is supplied to the discharge resistor. ２ The voltage drop when the current flows is the time constant adjustment signal V ＴＳ When it is greater than this, the temperature simulation capacitor 74 is charged. Conversely, an amplitude-dependent current I is drawn to the discharge resistance element. ２ The voltage drop when the current flows is the time constant adjustment signal V ＴＳ When the value is smaller, the temperature simulation capacitor 74 is discharged. In this way, the charging and discharging of the temperature simulation capacitor 74 simulates a rise and fall in temperature.

[0084] [Fourth Embodiment] Next, a high-frequency power amplifier circuit and control circuit according to the fourth embodiment will be described with reference to Figure 15. Hereinafter, the configuration common to the high-frequency power amplifier circuit and control circuit according to the first embodiment, described with reference to Figures 1 to 6B, will be omitted from the explanation.

[0085] Figure 15 is a block diagram of a high-frequency power amplifier circuit according to the fourth embodiment. In the first embodiment (Figure 1), the detector 10 is coupled to the transmission line between the power amplifier 80 and the first transistor 42. In contrast, in the fourth embodiment, the detector 10 is coupled to the transmission line between the first transistor 42 and the filter selection switch 81. Therefore, the detector 10 detects the voltage waveform V of the high-frequency signal that is attenuated by the attenuation circuit 40 before being input to the filter 82. １ It detects this voltage waveform V １ An amplitude-dependent physical quantity V obtained by detection. ２ This reflects the input power to filter 82.

[0086] Next, the excellent effects of the fourth embodiment will be described. In the fourth embodiment, the attenuation of the high-frequency signal can be controlled based on the input power of the filter 82. While the control of the control circuit in the first embodiment is feedforward control, the control of the control circuit in the fourth embodiment is feedback control. The filter 82 can also be protected by performing feedback control as in the fourth embodiment.

[0087] Next, a modification of the fourth embodiment will be described. As a modification of the fourth embodiment, the detector 10 may be coupled to the transmission line between the antenna switch 83 and the antenna terminal 84.

[0088] Alternatively, the detector 10 may be coupled to the transmission line on the input side of the power amplifier 80. In this case, the control by the control circuit will be feedforward control, similar to that in the first embodiment.

[0089] [Fifth Embodiment] Next, a high-frequency power amplifier circuit and control circuit according to the fifth embodiment will be described with reference to Figure 16. Hereinafter, the configuration common to the high-frequency power amplifier circuit and control circuit according to the first embodiment, described with reference to Figures 1 to 6B, will be omitted from the explanation.

[0090] Figure 16 is a block diagram of a high-frequency power amplifier circuit according to the fifth embodiment. In the first embodiment (Figure 1), a directional coupler is used as the detector 10, and the voltage waveform V based on the traveling wave of the transmission line on the output side of the power amplifier 80 １ The voltage waveform V appearing in the transmission line is detected. In contrast, in the fifth embodiment, the detector 10 is constructed by wiring directly connected to the transmission line. １ This is then input directly into the detection circuit 20.

[0091] Voltage waveform V input to detection circuit 20 １ This is the sum of the voltage waveforms of the forward wave and the reflected wave. Therefore, the attenuation circuit 40 is controlled based on the power obtained by adding the power of the forward wave and the power of the reflected wave.

[0092] Next, the excellent effects of the fifth embodiment will be described. In the fifth embodiment, the sum of the power of the traveling wave and the power of the reflected wave is the limiting start threshold V ３ＴＨ When the signal exceeds (Figure 2), control is performed to attenuate the high-frequency signal. This protects the filter 82. In addition, in the fifth embodiment, a directional coupler is not used, so the circuit can be simplified.

[0093] Next, a modification of the fifth embodiment will be described. In the fifth embodiment, the detector 10 is connected to the transmission line between the power amplifier 80 and the first transistor 42, but it may be connected to other locations. For example, it may be connected to the input terminal of the filter 82, the output terminal of the filter 82, or to the transmission line between the antenna switch 83 and the antenna terminal 84.

[0094] Alternatively, the detector 10 may be connected to the transmission line on the input side of the power amplifier 80. Since impedance matching is achieved on the input side of the detector 10, reflected waves can be ignored. Therefore, the attenuation circuit 40 can be controlled based on the power of the forward wave.

[0095] [Sixth Embodiment] Next, a high-frequency power amplifier circuit and control circuit according to the sixth embodiment will be described with reference to Figure 17. Hereinafter, the configuration common to the high-frequency power amplifier circuit and control circuit according to the first embodiment, described with reference to Figures 1 to 6B, will be omitted from the explanation.

[0096] Figure 17 is a block diagram of a high-frequency power amplifier circuit according to the sixth embodiment. In the first embodiment (Figure 1), the attenuation circuit 40 includes a first transistor 42 inserted in series with the transmission line. In contrast, in the sixth embodiment, the attenuation circuit 40 includes a second transistor 43 connected between the transmission line and the reference potential, instead of the first transistor 42. For example, an NMOSFET is used as the second transistor 43.

[0097] Control signal V applied to the gate of the second transistor 43 ４ When the voltage is set below the ON voltage and the second transistor 43 is turned off, the high-frequency signal is not attenuated. Control signal V ４When the voltage is increased, the second transistor 43 conducts, and the resistance between the source and drain gradually decreases. As the resistance between the source and drain decreases, the attenuation of the high-frequency signal increases.

[0098] Next, the excellent effects of the sixth embodiment will be described. In the sixth embodiment, as in the first embodiment, the filter 82 can be protected by attenuating the high-frequency signal.

[0099] [Seventh Embodiment] Next, the high-frequency power amplifier circuit and control circuit according to the seventh embodiment will be described with reference to Figures 18 and 19. Hereinafter, the configuration common to the high-frequency power amplifier circuit and control circuit according to the first embodiment, described with reference to Figures 1 to 6B, will be omitted from the explanation.

[0100] Figure 18 shows the integrated physical quantity V in the high-frequency power amplifier circuit according to the seventh embodiment. ３ and control signal V ４ This is a graph showing the relationship. The threshold calculation circuit 41 (Figure 6A) used in the high-frequency power amplifier circuit according to the first embodiment (Figure 1) is, as shown in Figure 2, the integral result physical quantity V ３ The limit start threshold V ３ＴＨ Within the above range, the resulting physical quantity V ３ As the control signal V increases ４ The control signal V is gradually reduced. ４ It has two values: high level and low level.

[0101] Integration result physical quantity V ３ is the first threshold B ３ＴＨＨ As it increases, the threshold calculation circuit 41 (Figure 1) controls the control signal V ４ By transitioning to a low level, the first transistor 42 is turned off. As a result, the high-frequency signal output from the power amplifier 80 is no longer transmitted to the filter 82. Integral result physical quantity V ３ The first threshold V ３ＴＨＨ From a larger range, the first threshold V ３ＴＨＨ A smaller second threshold V ３ＴＨＬ As it becomes smaller, the control signal V ４By transitioning to a high level, the first transistor 42 is turned on. As a result, the high-frequency signal output from the power amplifier 80 is transmitted to the filter 82.

[0102] Figure 19 is an equivalent circuit diagram of the threshold calculation circuit 41 used in the high-frequency power amplifier circuit according to the seventh embodiment. A hysteresis-equipped comparator is used as the threshold calculation circuit 41. The integrated result physical quantity V is connected to the inverting input node of the comparator 41G. ３ The following is input: The non-inverting input node of comparator 41G is connected to the power supply voltage via resistor element 41H and to a reference potential via resistor element 41J. The output node of comparator 41G is connected to the power supply voltage via pull-up resistor element 41L and to the non-inverting input node via resistor element 41K. By adjusting the resistance values ​​of resistor elements 41H, 41J, and 41K, the first threshold voltage V ３ＴＨＨ and the second threshold V ３ＴＨＬ It can be adjusted.

[0103] Next, the excellent effects of the seventh embodiment will be described. In the seventh embodiment, when the output power of the power amplifier 80 becomes excessively large, the first transistor 42 is turned off to block the input of the high-frequency signal to the filter 82, thereby protecting the filter 82. In addition, in the seventh embodiment, the threshold calculation circuit 41 can be implemented with a comparator with hysteresis, so the circuit can be simplified compared to the threshold calculation circuit of the first embodiment (Figure 6A).

[0104] The embodiments described above are illustrative, and it goes without saying that partial substitution or combination of the configurations shown in different embodiments is possible. Similar effects and benefits from similar configurations in multiple embodiments will not be mentioned sequentially for each embodiment. Furthermore, the present invention is not limited to the embodiments described above. For example, it will be obvious to those skilled in the art that various modifications, improvements, and combinations are possible.

[0105] 10 Detector 20 Detection circuit 20A Diode 20B Capacitor 20C DC cut capacitor 20D Resistor 20E Transistor 20F Bias circuit 20G Resistor 20H Fundamental wave component removal filter 30 Integrating circuit 30A Resistor 30B, 30BA Capacitor 30BB Varicap 30C Capacitor 30D Op-amp 30E Resistor 30F Pull-down resistor 30G Feedback capacitor 30H Feedback resistor 30K, 30L NOT circuit 40 Attenuation circuit 41 Threshold calculation circuit 41A Voltage-to-current conversion circuit 41AA Op-amp 41AB Current mirror circuit 41AC Resistor 41AD Current mirror circuit 41AE Op-amp 41AF Pull-up resistor 41AG Current mirror circuit 41B Current mirror circuit 41C Pull-out circuit 41D Threshold determination current source 41E Current mirror circuit 41F Pull-up resistor element 41G Comparator 41H, 41J, 41K Resistor element 41L Pull-up resistor element 42 First transistor 43 Second transistor 50 Time constant adjustment circuit 51 AD converter 52 First adder circuit 52A, 52B Operational amplifier 53 Second adder circuit 60 Temperature sensor 60A Constant current source 60B Temperature-dependent diode 61 Processor 70 Heat simulation circuit 71 Voltage-current conversion circuit 72 Current mirror circuit 73 Current source for temperature simulation 74 Capacitor for temperature simulation 80 Power amplifier 81 Filter selection switch 82 Filter 83 Antenna switch 84 Antenna terminal

Claims

1. A control circuit comprising: a detector for detecting the waveform of a high-frequency signal transmitted through one of the transmission lines on the input or output side of a power amplifier that amplifies a high-frequency signal; a detection circuit for detecting the waveform detected by the detector and outputting an amplitude-dependent physical quantity that depends on the amplitude of the waveform; an integration circuit for integrating the amplitude-dependent physical quantity over time and outputting an integrated physical quantity corresponding to the integration value; and an attenuation circuit for attenuating the high-frequency signal transmitted through the transmission line on the output side of the power amplifier based on the integrated physical quantity.

2. The control circuit according to claim 1, wherein the attenuation circuit does not attenuate the high-frequency signal when the integrated physical quantity is smaller than the limiting start threshold, and when the integrated physical quantity exceeds the limiting start threshold, the high-frequency signal is gradually attenuated as the integrated physical quantity increases.

3. The control circuit according to claim 2, wherein the attenuation circuit includes a first transistor inserted in series with the transmission line on the output side of the power amplifier, and a threshold calculation circuit that turns on the first transistor when the integrated physical quantity is smaller than the limit start threshold, and controls the first transistor to turn off in proportion to the increase in the integrated physical quantity when the integrated physical quantity exceeds the limit start threshold.

4. The control circuit according to claim 3, wherein the threshold calculation circuit includes a voltage-current conversion circuit that outputs a current corresponding to the integral result physical quantity, an extraction circuit configured to extract a current corresponding to the limit start threshold from the current output from the voltage-current conversion circuit, and a current mirror circuit configured so that the remaining current extracted by the extraction circuit from the current output from the voltage-current conversion circuit flows as a reference current, and the threshold calculation circuit outputs a control signal that controls the conduction of the first transistor based on the output current of the current mirror circuit.

5. The control circuit according to claim 3, wherein the threshold calculation circuit is configured to turn off the first transistor when the integral result physical quantity becomes greater than a first threshold, and to turn on the first transistor when the integral result physical quantity falls from a range greater than the first threshold to a second threshold smaller than the first threshold.

6. The control circuit according to any one of claims 1 to 4, wherein the integrating circuit has a configuration that allows the time constant to be changed, and further comprises a time constant adjustment circuit that changes the time constant of the integrating circuit based on an input time constant adjustment signal.

7. The control circuit according to claim 6, further comprising a temperature sensor that measures temperature and generates the time constant adjustment signal based on the measured temperature.

8. The control circuit according to claim 6, further comprising a heat generation simulation circuit that generates the time constant adjustment signal based on the amplitude-dependent physical quantity, the heat generation simulation circuit comprising a temperature simulation current source that generates a temperature simulation threshold current, and a temperature simulation capacitor configured to be charged by the remaining current obtained by subtracting the temperature simulation threshold current from the current corresponding to the amplitude-dependent physical quantity when the current corresponding to the amplitude-dependent physical quantity is greater than the temperature simulation threshold current, and to be discharged by a current corresponding to the difference between the current corresponding to the amplitude-dependent physical quantity and the temperature simulation threshold current when the current corresponding to the amplitude-dependent physical quantity is less than the temperature simulation threshold current, wherein the control circuit generates the time constant adjustment signal based on the voltage generated in the temperature simulation capacitor.

9. The control circuit according to any one of claims 6 to 8, wherein the integrating circuit includes a resistor element whose resistance value is variable and which affects the time constant, and a capacitor whose capacitance is variable.

10. The control circuit according to claim 9, wherein the time constant adjustment circuit includes a first adding circuit that adds a time constant adjustment voltage for adjusting the time constant and a first fixed voltage, and a second adding circuit that adds the time constant adjustment voltage and a second fixed voltage, and is configured to change the capacitance of the capacitor based on the first adding voltage obtained by the first adding circuit, and to change the resistance value of the resistive element based on the second adding voltage obtained by the second adding circuit.

11. The control circuit according to any one of claims 3 to 5, wherein the detector detects the waveform of a high-frequency signal transmitted in the transmission line between the output node of the power amplifier and the location where the first transistor is connected.

12. The control circuit according to any one of claims 3 to 5, wherein the detector detects the waveform of the high-frequency signal after it has passed through the first transistor in the transmission line on the output side of the power amplifier.

13. A high-frequency power amplification circuit comprising a control circuit according to any one of claims 1 to 12, a power amplifier, and a filter inserted in the transmission line on the output side of the power amplifier.

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