Power conversion apparatus

Abstract

The present invention addresses the problem of reducing current distortion. In a power conversion apparatus (100), a control device (50) generates, in the n-th control cycle, a plurality of first PWM signals (SU1, SV1, SW1), a plurality of second PWM signals (SU2, SV2, SW2), and a plurality of control signals (SU6, SU7, SV6, SV7, SW6, SW7) for the (n+1)-th control cycle. Here, n is a natural number. The control device (50) stops the operation of a power conversion circuit (11) when an input current value detected by a current detection unit (16) exceeds a current protection threshold value. The control device (50) increases the current protection threshold value for the (n+1)-th control cycle when the control device (50) determines, at the n-th control cycle, that two-phase resonant currents corresponding to two switching circuits (10) among a plurality of switching circuits (10) simultaneously flow through a resonant inductor (L1) in the (n+1)-th control cycle.

Description

Power converter

[0001] This disclosure relates to a power conversion device, and more particularly to a power conversion device capable of converting DC power to AC power.

[0002] The power conversion device disclosed in Patent Document 1 includes a first DC terminal and a second DC terminal, a power conversion circuit, a plurality of AC terminals, a plurality of switches, a plurality of resonance capacitors, a resonance inductor, a capacitor, and a control device. The power conversion circuit includes a plurality of first switching elements and a plurality of second switching elements. In the power conversion circuit, a plurality of switching circuits formed by connecting the plurality of first switching elements and the plurality of second switching elements in one-to-one series are connected in parallel to each other. In the power conversion circuit, the plurality of first switching elements are connected to the first DC terminal, and the plurality of second switching elements are connected to the second DC terminal. The plurality of AC terminals correspond to the plurality of switching circuits in one-to-one correspondence. Each of the plurality of AC terminals is connected to a connection point between the first switching element and the second switching element in the corresponding switching circuit among the plurality of switching circuits. The plurality of switches correspond to the plurality of switching circuits in one-to-one correspondence. Each of the plurality of switches has a first end connected to a connection point between the first switching element and the second switching element in the corresponding switching circuit among the plurality of switching circuits, and a second end commonly connected to a common connection point. The plurality of resonance capacitors correspond to the plurality of switches in one-to-one correspondence. Each of the plurality of resonance capacitors is connected between the first end of the corresponding switch among the plurality of switches and the second DC terminal. The resonance inductor has a first end and a second end. In the resonance inductor, the first end of the resonance inductor is connected to the common connection point. The capacitor is connected between the second end of the resonance inductor and the second DC terminal. The control device controls the plurality of first switching elements, the plurality of second switching elements, and the plurality of switches. When the control device determines that two-phase resonance currents corresponding to two switching circuits among the plurality of switching circuits simultaneously flow through the resonance inductor, the control device performs control (shift control) to shift the on-periods of the first switching element and the second switching element in one of the two switching circuits, respectively.

[0003] In the power conversion device disclosed in Patent Document 1, soft switching can be performed more reliably.

[0004] In the power conversion device disclosed in Patent Document 1, the current distortion of the load current may increase due to the difference in line voltage generated by shifting the ON periods of the first switching element and the second switching element in a single switching circuit.

[0005] International Publication No. 2023 / 074636

[0006] The purpose of this disclosure is to provide a power conversion device capable of reducing current distortion.

[0007] A power conversion device according to one embodiment of the present disclosure comprises a first DC terminal, a second DC terminal, a power conversion circuit, a plurality of AC terminals, a plurality of switches, a plurality of resonant capacitors, a resonant inductor, a regenerative capacitor, a current detection unit, and a control device. The power conversion circuit has a plurality of first switching elements and a plurality of second switching elements. In the power conversion circuit, a plurality of switching circuits, each in which the plurality of first switching elements and the plurality of second switching elements are connected in series in a one-to-one relationship, are connected in parallel to each other. In the power conversion circuit, the plurality of first switching elements are connected to the first DC terminal, and the plurality of second switching elements are connected to the second DC terminal. The plurality of AC terminals correspond one-to-one to the plurality of switching circuits. Each of the plurality of AC terminals is connected to the connection point of the first switching element and the second switching element in the corresponding switching circuit among the plurality of switching circuits. The plurality of switches correspond one-to-one to the plurality of switching circuits. Each of the plurality of switches has a first end and a second end. Each of the plurality of switches has its first end connected to the connection point of the first switching element and the second switching element in the corresponding switching circuit among the plurality of switching circuits, and its second end is commonly connected to a common connection point. The plurality of resonant capacitors correspond one-to-one with the plurality of switches. Each of the plurality of resonant capacitors is connected between the first end and the second DC terminal of the corresponding switch among the plurality of switches. The resonant inductor has a third end and a fourth end. In the resonant inductor, the third end of the resonant inductor is connected to the common connection point. The regenerative capacitor has a fifth end and a sixth end. In the regenerative capacitor, the fifth end is connected to the second DC terminal, and the sixth end is connected to the fourth end of the resonant inductor. The current detection unit detects the input current flowing from the first DC terminal to the power conversion circuit.The control device generates a plurality of first PWM signals to control the plurality of first switching elements, a plurality of second PWM signals to control the plurality of second switching elements, and a plurality of control signals to control the plurality of switches, based on a plurality of duty cycle command values ​​of a plurality of phases corresponding one-to-one to the plurality of switching circuits. The control device generates the plurality of first PWM signals, the plurality of second PWM signals, and the plurality of control signals for the (n+1)th control cycle in the nth control cycle, where n is a natural number. If the input current detected by the current detection unit exceeds the current protection threshold, the control device stops the operation of the power conversion circuit. If the control device determines in the nth control cycle that a two-phase resonant current corresponding to two of the plurality of switching circuits flows simultaneously through the resonant inductor in the (n+1)th control cycle, it increases the current protection threshold for the (n+1)th control cycle.

[0008] Figure 1 is a circuit diagram of a system equipped with a power converter according to Embodiment 1. Figure 2 is an explanatory diagram of the relationship between the duty cycle command value and the load current used in the control device of the power converter. Figure 3 is an explanatory diagram of the operation of the power converter. Figure 4 is an explanatory diagram of the operation of the power converter when the U-phase load current > 0 and the U-phase resonant capacitor is being charged. Figure 5 is an explanatory diagram of the operation of the power converter when the V-phase load current > 0 and the V-phase resonant capacitor is being charged. Figure 6 is an explanatory diagram of the operation of the power converter when the W-phase load current > 0 and the W-phase resonant capacitor is being charged. Figure 7 is an explanatory diagram of the operation of the power converter when the U-phase load current > 0 and the U-phase resonant capacitor is being discharged. Figure 8 is an explanatory diagram of the operation of the power converter when the U-phase load current < 0 and the U-phase resonant capacitor is being discharged. Figure 9 is an explanatory diagram of the operation of the power converter described above when the load current of the U phase is < 0 and the resonant capacitor of the U phase is being charged. Figure 10 is a flowchart explaining the operation of the power converter described above. Figure 11 is an explanatory diagram of the operation of the power converter according to Embodiment 2. Figure 12 is another explanatory diagram of the operation of the power converter described above.

[0009] (Embodiment 1) Below, the power conversion device 100 according to Embodiment 1 will be described with reference to Figures 1 to 10.

[0010] (1) Overall Configuration of the Power Converter The power converter 100 includes, for example, a first DC terminal 31, a second DC terminal 32, and a plurality (for example, three) AC terminals 41, as shown in Figure 1. A DC power supply E1 is connected between the first DC terminal 31 and the second DC terminal 32 of the power converter 100, and an AC load RA1 is connected to the plurality of AC terminals 41. The AC load RA1 is, for example, a three-phase servo motor. The power converter 100 converts the DC output from the DC power supply E1 into AC power and outputs it to the AC load RA1. The DC power supply E1 includes, for example, an AC-DC converter, a DC-DC converter, a solar cell, or a fuel cell. In the power converter 100, if the plurality of AC terminals 41 are three AC terminals 41, the AC power is, for example, three-phase AC power having U-phase, V-phase, and W-phase.

[0011] The power converter 100 comprises a power conversion circuit 11, a plurality (for example, three) of switches 8, a plurality (for example, three) of resonant capacitors 9, a regenerative capacitor 15, a resonant inductor L1, a current detection unit 16, and a control device 50. The power converter 100 further comprises a protection circuit 17 and a capacitor C10. Each of the plurality of switches 8 is, for example, a bidirectional switch.

[0012] The power conversion circuit 11 has a plurality (for example, three) of first switching elements 1 and a plurality (for example, three) of second switching elements 2. In the power conversion circuit 11, a plurality (for example, three) of switching circuits 10, each consisting of a plurality of first switching elements 1 and a plurality of second switching elements 2 connected in a one-to-one series relationship, are connected in parallel to each other. In the power conversion circuit 11, the plurality of first switching elements 1 are connected to a first DC terminal 31, and the plurality of second switching elements 2 are connected to a second DC terminal 32. The plurality of AC terminals 41 correspond one-to-one to the plurality of switching circuits 10. Each of the plurality of AC terminals 41 is connected to the connection point 3 of the first switching element 1 and the second switching element 2 in the corresponding switching circuit 10 among the plurality of switching circuits 10. The plurality of switches 8 correspond one-to-one to the plurality of switching circuits 10. Each of the plurality of switches 8 has a first end 81 and a second end 82. Each of the multiple switches 8 has its first end 81 connected to the connection point 3 of the first switching element 1 and the second switching element 2 in the corresponding switching circuit 10 among the multiple switching circuits 10. The multiple resonant capacitors 9 correspond one-to-one with the multiple switches 8. Each of the multiple resonant capacitors 9 is connected between the first end 81 and the second DC terminal 32 of the corresponding switch 8 among the multiple switches 8. The resonant inductor L1 has a third end and a fourth end. In the resonant inductor L1, the third end of the resonant inductor L1 is connected to the common connection point 25. The regenerative capacitor 15 has a fifth end 153 and a sixth end 154. In the regenerative capacitor 15, the fifth end 153 is connected to the second DC terminal 32, and the sixth end 154 is connected to the fourth end of the resonant inductor L1. Therefore, the sixth end 154 of the regenerative capacitor 15 is connected to the common connection point 25 via the resonant inductor L1. The current detection unit 16 detects the input current Iin flowing from the first DC terminal 31 to the power conversion circuit 11.The control device 50 generates a plurality of first PWM (Pulse Width Modulation) signals SU1, SV1, SW1 for controlling a plurality of first switching elements 1, a plurality of second PWM signals SU2, SV2, SW2 for controlling a plurality of second switching elements 2, and a plurality of control signals SU6, SU7, SV6, SV7, SW6, SW7 for controlling a plurality of switches 8.

[0013] (2) Details of the power converter In the following description, for the sake of convenience, the switching circuits 10 corresponding to the U-phase, V-phase, and W-phase may be referred to as switching circuit 10U, switching circuit 10V, and switching circuit 10W, respectively. Also, in the following description, the first switching element 1 and the second switching element 2 of switching circuit 10U may be referred to as the first switching element 1U and the second switching element 2U. Also, in the following description, the first switching element 1 and the second switching element 2 of switching circuit 10V may be referred to as the first switching element 1V and the second switching element 2V. Also, in the following description, the first switching element 1 and the second switching element 2 of switching circuit 10W may be referred to as the first switching element 1W and the second switching element 2W. Furthermore, in the following, the connection point 3 between the first switching element 1U and the second switching element 2U may be referred to as connection point 3U, the connection point 3 between the first switching element 1V and the second switching element 2V may be referred to as connection point 3V, and the connection point 3 between the first switching element 1W and the second switching element 2W may be referred to as connection point 3W. Furthermore, in the following, the AC terminal 41 connected to connection point 3U may be referred to as AC terminal 41U, the AC terminal 41 connected to connection point 3V may be referred to as AC terminal 41V, and the AC terminal 41 connected to connection point 3W may be referred to as AC terminal 41W. Furthermore, in the following, the resonant capacitor 9 connected in parallel to the second switching element 2U may be referred to as resonant capacitor 9U, the resonant capacitor 9 connected in parallel to the second switching element 2V may be referred to as resonant capacitor 9V, and the resonant capacitor 9 connected in parallel to the second switching element 2W may be referred to as resonant capacitor 9W. Furthermore, in the following, the switch 8 connected to connection point 3U may be referred to as switch 8U, the switch 8 connected to connection point 3V as switch 8V, and the switch 8 connected to connection point 3W as switch 8W.

[0014] The power converter 100 has the high-potential output terminal (positive terminal) of the DC power supply E1 connected to the first DC terminal 31, and the low-potential output terminal (negative terminal) of the DC power supply E1 connected to the second DC terminal 32. In addition, the power converter 100 has, for example, three AC terminals 41U, 41V, and 41W connected to the U-phase terminal, V-phase terminal, and W-phase terminal of the AC load RA1, respectively.

[0015] In the power conversion circuit 11, each of the multiple (e.g., three) first switching elements 1 and each of the multiple (e.g., three) second switching elements 2 has a control terminal, a first main terminal, and a second main terminal. The control terminals of the multiple first switching elements 1 and the multiple second switching elements 2 are connected to the control device 50. In each of the multiple switching circuits 10 of the power conversion device 100, the first main terminal of the first switching element 1 is connected to the first DC terminal 31, the second main terminal of the first switching element 1 is connected to the first main terminal of the second switching element 2, and the second main terminal of the second switching element 2 is connected to the second DC terminal 32. In each of the multiple switching circuits 10, the first switching element 1 is a high-side switching element (P-side switching element), and the second switching element 2 is a low-side switching element (N-side switching element). Each of the multiple first switching elements 1 and the multiple second switching elements 2 is, for example, an IGBT (Insulated Gate Bipolar Transistor). Therefore, the control terminal, first main terminal, and second main terminal of each of the multiple first switching elements 1 and the multiple second switching elements 2 are the gate terminal, collector terminal, and emitter terminal, respectively.

[0016] The power conversion circuit 11 further includes a plurality of first diodes 4 connected in antiparallel in a one-to-one relationship to a plurality of first switching elements 1 (three), and a plurality of second diodes 5 connected in antiparallel in a one-to-one relationship to a plurality of second switching elements 2 (three). In each of the plurality of first diodes 4, the anode of the first diode 4 is connected to the second main terminal (emitter terminal) of the first switching element 1 corresponding to the first diode 4, and the cathode of the first diode 4 is connected to the first main terminal (collector terminal) of the first switching element 1 corresponding to the first diode 4. In each of the plurality of second diodes 5, the anode of the second diode 5 is connected to the second main terminal (emitter terminal) of the second switching element 2 corresponding to the second diode 5, and the cathode of the second diode 5 is connected to the first main terminal (collector terminal) of the second switching element 2 corresponding to the second diode 5.

[0017] The connection point 3U between the first switching element 1U and the second switching element 2U is connected via AC terminal 41U to, for example, the U-phase terminal of the AC load RA1. The connection point 3V between the first switching element 1V and the second switching element 2V is connected via AC terminal 41V to, for example, the V-phase terminal of the AC load RA1. The connection point 3W between the first switching element 1W and the second switching element 2W is connected via AC terminal 41W to, for example, the W-phase terminal of the AC load RA1.

[0018] Multiple resonant capacitors 9 correspond one-to-one with multiple switches 8. Each of the multiple resonant capacitors 9 is connected between the first terminal 81 and the second DC terminal 32 of the corresponding switch 8 among the multiple switches 8. The power converter 100 has multiple resonant circuits. The multiple resonant circuits include a resonant circuit having a resonant capacitor 9U and a resonant inductor L1, a resonant circuit having a resonant capacitor 9V and a resonant inductor L1, and a resonant circuit having a resonant capacitor 9W and a resonant inductor L1. The resonant inductor L1 is common to all of the multiple resonant circuits.

[0019] Each of the multiple switches 8 has a third switching element 6 and a fourth switching element 7. Each of the multiple (three in the example of Figure 1) third switching elements 6 and each of the multiple (three in the example of Figure 1) fourth switching elements 7 has a control terminal, a first main terminal and a second main terminal. The control terminals of the multiple third switching elements 6 and the multiple fourth switching elements 7 are connected to the control device 50. Each of the multiple third switching elements 6 and the multiple fourth switching elements 7 is, for example, an IGBT. Therefore, the control terminal, first main terminal and second main terminal of each of the multiple third switching elements 6 and the multiple fourth switching elements 7 are the gate terminal, collector terminal and emitter terminal, respectively. In each of the multiple switches 8, the third switching element 6 and the fourth switching element 7 are connected in reverse series. In each of the multiple switches 8, the first main terminal (collector terminal) of the third switching element 6 is connected to the first main terminal (collector terminal) of the fourth switching element 7. In each of the multiple switches 8, the second main terminal (emitter terminal) of the third switching element 6 is connected to the connection point 3 of the corresponding switching circuit 10 among the multiple switching circuits 10, and the second main terminal (emitter terminal) of the fourth switching element 7 is connected to the resonant inductor L1. Furthermore, each of the multiple switches 8 has a diode 61 connected in antiparallel to the third switching element 6 and a diode 71 connected in antiparallel to the fourth switching element 7.

[0020] In the power converter 100, switch 8U is connected to the connection point 3U between the first switching element 1U and the second switching element 2U. Switch 8V is connected to the connection point 3V between the first switching element 1V and the second switching element 2V. Switch 8W is connected to the connection point 3W between the first switching element 1W and the second switching element 2W. For convenience of explanation, the third switching element 6 and the fourth switching element 7 of switch 8U may be referred to as the third switching element 6U and the fourth switching element 7U, respectively; the third switching element 6 and the fourth switching element 7 of switch 8V may be referred to as the third switching element 6V and the fourth switching element 7V, respectively; and the third switching element 6 and the fourth switching element 7 of switch 8W may be referred to as the third switching element 6W and the fourth switching element 7W, respectively.

[0021] Multiple switches 8 are controlled by the control device 50. In other words, the third switching element 6U, the fourth switching element 7U, the third switching element 6V, the fourth switching element 7V, the third switching element 6W, and the fourth switching element 7W are controlled by the control device 50.

[0022] The resonant inductor L1 has a third terminal and a fourth terminal. In the resonant inductor L1, the third terminal is connected to the common connection point 25. The fourth terminal of the resonant inductor L1 is connected to the sixth terminal 154 of the regenerative capacitor 15.

[0023] The regenerative capacitor 15 is connected between the fourth terminal of the resonant inductor L1 and the second DC terminal 32. The regenerative capacitor 15 is, for example, a film capacitor.

[0024] The protection circuit 17 includes a third diode 13 and a fourth diode 14. The third diode 13 is connected between the common connection point 25 and the first DC terminal 31. The anode of the third diode 13 is connected to the common connection point 25. Also, the cathode of the third diode 13 is connected to the first DC terminal 31. The fourth diode 14 is connected between the common connection point 25 and the second DC terminal 32. The anode of the fourth diode 14 is connected to the second DC terminal 32. Also, the cathode of the fourth diode 14 is connected to the common connection point 25. Therefore, the fourth diode 14 is connected in series with the third diode 13.

[0025] Capacitor C10 is connected between the first DC terminal 31 and the second DC terminal 32, and is connected in parallel to the power conversion circuit 11. Capacitor C10 is, for example, an electrolytic capacitor.

[0026] As described above, the current detection unit 16 detects the input current Iin flowing from the first DC terminal 31 to the power conversion circuit 11. More specifically, the current detection unit 16 detects the input current Iin flowing from the DC power supply E1 to the power conversion circuit 11 via the first DC terminal 31. The current detection unit 16 is a current detection resistor, but it is not limited to a current detection resistor; for example, a current transformer may also be used.

[0027] The control device 50 controls a plurality of first switching elements 1, a plurality of second switching elements 2, and a plurality of switches 8. The execution entity of the control device 50 includes a computer system. The computer system has one or more computers. The computer system mainly consists of a processor and memory as hardware. The function of the control device 50 as the execution entity in this disclosure is realized by the processor executing a program recorded in the memory of the computer system. The program may be pre-recorded in the memory of the computer system, or it may be provided via a telecommunications line, or it may be recorded and provided on a non-temporary recording medium such as a memory card, optical disk, or hard disk drive (magnetic disk) that can be read by the computer system. The processor of the computer system is composed of one or more electronic circuits including a semiconductor integrated circuit (IC) or a large-scale integrated circuit (LSI). The plurality of electronic circuits may be aggregated on a single chip or distributed across multiple chips. The plurality of chips may be aggregated on a single device or distributed across multiple devices.

[0028] The control device 50 outputs first PWM signals SU1, SV1, and SW1 to control the on / off state of each of the multiple first switching elements 1U, 1V, and 1W. Each of the first PWM signals SU1, SV1, and SW1 is a PWM signal whose potential level changes between, for example, a first potential level (hereinafter also referred to as a low level) and a second potential level (hereinafter also referred to as a high level) that is higher than the first potential level. The first switching elements 1U, 1V, and 1W are turned on when the first PWM signals SU1, SV1, and SW1 are at a high level and turned off when they are at a low level, respectively. The control device 50 also outputs second PWM signals SU2, SV2, and SW2 to control the on / off state of each of the multiple second switching elements 2U, 2V, and 2W. Each of the second PWM signals SU2, SV2, and SW2 is a PWM signal whose potential level changes between, for example, a first potential level (hereinafter also called the low level) and a second potential level (hereinafter also called the high level) that is higher than the first potential level. The second switching elements 2U, 2V, and 2W are turned on when the second PWM signals SU2, SV2, and SW2 are at a high level and turned off when they are at a low level, respectively.

[0029] The control device 50 generates a plurality of first PWM signals SU1, SV1, SW1 and a plurality of second PWM signals SU2, SV2, SW2 based on the three-phase duty command values ​​du, dv, and dw (see Figure 2) of the three-phase modulation scheme. Figure 2 shows a U-phase modulated wave with the U-phase duty command value du as an instantaneous value, a V-phase modulated wave with the V-phase duty command value dv as an instantaneous value, and a W-phase modulated wave with the W-phase duty command value dw as an instantaneous value. The U-phase modulated wave, V-phase modulated wave, and W-phase modulated wave are sinusoidal and have a phase difference of 120° from each other. The control device 50 generates the three-phase modulated duty command values ​​du, dv, and dw based on information regarding the state of the AC load RA1. When the AC load RA1 is a three-phase servo motor, information regarding the state of the AC load RA1 includes, for example, the detected values ​​from a plurality (three) of current sensors 18a, 18b, and 18c that detect the output currents (hereinafter also referred to as load currents) iU, iV, and iW flowing through the U-phase, V-phase, and W-phase terminals of the AC load RA1, respectively. Each of the three current sensors 18a, 18b, and 18c is a current detection resistor, but it is not limited to current detection resistors; for example, a current transformer may also be used.

[0030] The control device 50 generates, for example, a plurality of first PWM signals SU1, SV1, SW1, a plurality of second PWM signals SU2, SV2, SW2, and a plurality of control signals SU6, SU7, SV6, SV7, SW6, SW7, using the three-phase duty command values ​​du, dv, dw (see Figure 2) and the triangular wave carrier signal CA1 for each cycle of the carrier signal CA1 (see Figure 3). In this embodiment, the control device 50 generates the plurality of first PWM signals SU1, SV1, SW1, a plurality of second PWM signals SU2, SV2, SW2, and a plurality of control signals SU6, SU7, SV6, SV7, SW6, SW7 for the (n+1)th control cycle in the nth control cycle, where n is a natural number. In this embodiment, the control cycle is the same as one cycle of the carrier signal CA1. In Figure 3, the nth control period is labeled "the nth period," and the (n+1)th control period is labeled "the (n+1)th period." Note that one period of the carrier signal CA1 is shorter than one period of each of the U-phase modulated wave, V-phase modulated wave, and W-phase modulated wave.

[0031] The control device 50 generates a first PWM signal SU1 and a second PWM signal SU2 to be supplied to the first switching element 1U and the second switching element 2U, respectively, based on the carrier signal CA1 and the duty cycle command value du. The control device 50 also generates a first PWM signal SV1 and a second PWM signal SV2 to be supplied to the first switching element 1V and the second switching element 2V, respectively, based on the carrier signal CA1 and the duty cycle command value dv of the V phase. The control device 50 also generates a first PWM signal SW1 and a second PWM signal SW2 to be supplied to the first switching element 1W and the second switching element 2W, respectively, based on the carrier signal CA1 and the duty cycle command value dw of the W phase. In this embodiment, the control device 50 sets the maximum value of the carrier signal CA1 to 1 and the minimum value to 0. Furthermore, the control device 50 sets the maximum value (corresponding to 100%) of each of the three-phase duty command values ​​du, dv, and dw to 1, and the minimum value (corresponding to 0%) to 0. In this embodiment, when the duty command value du of the U phase is greater than 0.5, the polarity of the U phase load current iU is positive, and when the duty command value du of the U phase is less than 0.5, the polarity of the U phase load current iU is negative. Also in this embodiment, when the duty command value dv of the V phase is greater than 0.5, the polarity of the V phase load current iV is positive, and when the duty command value dv of the V phase is less than 0.5, the polarity of the V phase load current iV is negative. Also in this embodiment, when the duty command value dw of the W phase is greater than 0.5, the polarity of the W phase load current iW is positive, and when the duty command value dw of the W phase is less than 0.5, the polarity of the W phase load current iW is negative. Furthermore, the control device 50 may set the maximum and minimum values ​​of the carrier signal CA1 to values ​​other than 1 and 0, respectively, and the maximum and minimum values ​​of the three-phase duty command values ​​du, dv, and dw to values ​​other than 1 and 0, respectively. In this case, if the duty command value du of the U phase is greater than {(maximum value of duty command value du) - (minimum value of duty command value du)} / 2, the polarity of the U phase load current iU is positive, and if it is less than {(maximum value of duty command value du) - (minimum value of duty command value du)} / 2, the polarity of the U phase load current iU is negative.Furthermore, if the duty cycle command value dv of the V phase is greater than {(maximum value of duty cycle command value dv) - (minimum value of duty cycle command value dv)} / 2, the polarity of the V phase load current iV is positive, and if it is less than {(maximum value of duty cycle command value dv) - (minimum value of duty cycle command value dv)} / 2, the polarity of the V phase load current iV is negative. Also, if the duty cycle command value dw of the W phase is greater than {(maximum value of duty cycle command value dw) - (minimum value of duty cycle command value dw)} / 2, the polarity of the W phase load current iW is positive, and if it is less than {(maximum value of duty cycle command value dw) - (minimum value of duty cycle command value dw)} / 2, the polarity of the W phase load current iW is negative.

[0032] The duty cycles of the first PWM signal SU1 and the second PWM signal SU2, generated by the control device 50, change based on the duty cycle command value du. The control device 50 generates the first PWM signal SU1 by comparing the duty cycle command value du with the carrier signal CA1. More specifically, the control device 50 compares the duty cycle command value du with the carrier signal CA1 and generates the first PWM signal SU1 which is high level when the duty cycle command value du is greater than the carrier signal CA1 and low level when the duty cycle command value du is less than or equal to the carrier signal CA1. The control device 50 also generates the second PWM signal SU2 by inverting the first PWM signal SU1. Furthermore, the control device 50 sets a dead time period Td (see Figures 3 and 4) between the high-level period of the first PWM signal SU1 and the high-level period of the second PWM signal SU2 so that the ON period of the first switching element 1U and the ON period of the second switching element 2U do not overlap.

[0033] The duty cycles of the first PWM signal SV1 and the second PWM signal SV2, both generated by the control device 50, change based on the duty cycle command value dv. The control device 50 generates the first PWM signal SV1 by comparing the duty cycle command value dv with the carrier signal CA1. More specifically, the control device 50 compares the duty cycle command value dv with the carrier signal CA1 and generates the first PWM signal SV1 which is high level during periods when the duty cycle command value dv is greater than the carrier signal CA1, and low level during periods when the duty cycle command value dv is less than or equal to the carrier signal CA1. The control device 50 also generates the second PWM signal SV2 by inverting the first PWM signal SV1. Furthermore, the control device 50 sets a dead time period Td (see Figures 3 and 5) between the high-level period of the first PWM signal SV1 and the high-level period of the second PWM signal SV2 so that the ON period of the first switching element 1V and the ON period of the second switching element 2V do not overlap.

[0034] The duty cycles of the first PWM signal SW1 and the second PWM signal SW2, both generated by the control device 50, change based on the duty cycle command value dw. The control device 50 generates the first PWM signal SW1 by comparing the duty cycle command value dw with the carrier signal CA1. More specifically, the control device 50 compares the duty cycle command value dw with the carrier signal CA1 and generates the first PWM signal SW1 which is high level when the duty cycle command value dw is greater than the carrier signal CA1 and low level when the duty cycle command value dw is less than or equal to the carrier signal CA1. The control device 50 also generates the second PWM signal SW2 by inverting the first PWM signal SW1. Furthermore, the control device 50 sets a dead time period Td (see Figure 6) between the high-level period of the first PWM signal SW1 and the high-level period of the second PWM signal SW2 so that the ON period of the first switching element 1W and the ON period of the second switching element 2W do not overlap.

[0035] The control device 50 sets a dead time period Td between the high-level period of the first PWM signal to the first switching element 1 and the high-level period of the second PWM signal to the second switching element 2 for each of the plurality of switching circuits 10, and sets the high-level period of the control signal to each of the plurality of switches 8 based on the dead time period Td for the corresponding switching circuit 10 among the plurality of switching circuits 10.

[0036] Multiple switches 8, resonant inductors L1, multiple resonant capacitors 9, and regenerative capacitors 15 are provided to perform zero-voltage soft switching of multiple first switching elements 1 and multiple second switching elements 2.

[0037] In the power conversion device 100, the control device 50 controls not only the multiple first switching elements 1 and the multiple second switching elements 2 of the power conversion circuit 11, but also the multiple switches 8.

[0038] The control device 50 generates control signals SU6, SU7, SV6, SV7, SW6, and SW7 to control the on / off state of the third switching element 6U, the fourth switching element 7U, the third switching element 6V, the fourth switching element 7V, the third switching element 6W, and the fourth switching element 7W, respectively, and outputs them to the gate terminals of the third switching element 6U, the fourth switching element 7U, the third switching element 6V, the fourth switching element 7V, the third switching element 6W, and the fourth switching element 7W.

[0039] When the third switching element 6U is ON and the fourth switching element 7U is OFF, switch 8U can allow the charging current flowing through the path of regenerative capacitor 15 - resonant inductor L1 - switch 8U - resonant capacitor 9U to pass through. The charging current is the current that charges the resonant capacitor 9U. When the third switching element 6U is OFF and the fourth switching element 7U is ON, switch 8U can allow the discharge current flowing through the path of resonant capacitor 9U - switch 8U - resonant inductor L1 - regenerative capacitor 15 to pass through. The discharge current is the current that discharges the charge from the resonant capacitor 9U.

[0040] Switch 8V allows the charging current flowing through the path of regenerative capacitor 15 - resonant inductor L1 - switch 8V - resonant capacitor 9V to pass through when the third switching element 6V is ON and the fourth switching element 7V is OFF. The charging current is the current that charges the resonant capacitor 9V. Switch 8V allows the discharge current flowing through the path of resonant capacitor 9V - switch 8V - resonant inductor L1 - regenerative capacitor 15 to pass through when the third switching element 6V is OFF and the fourth switching element 7V is ON. The discharge current is the current that discharges the charge from the resonant capacitor 9V.

[0041] Switch 8W can pass the charging current flowing through the path of regenerative capacitor 15 - resonant inductor L1 - switch 8W - resonant capacitor 9W when the third switching element 6W is ON and the fourth switching element 7W is OFF. The charging current is the current that charges the resonant capacitor 9W. Switch 8W can pass the discharge current flowing through the path of resonant capacitor 9W - switch 8W - resonant inductor L1 - regenerative capacitor 15 when the third switching element 6W is OFF and the fourth switching element 7W is ON. The discharge current is the current that discharges the charge from the resonant capacitor 9W.

[0042] (3) Operation of the power converter In the following, the current iL1 flowing through the resonant inductor L1 will be described as positive when flowing in the direction of the arrow in Figure 1, and negative when flowing in the opposite direction to the arrow in Figure 1. Furthermore, the load currents iU, iV, and iW flowing through the U-phase, V-phase, and W-phase terminals of the AC load RA1 will be described as positive when flowing in the direction of the arrow in Figure 1, and negative when flowing in the opposite direction to the arrow in Figure 1. Furthermore, the currents i9U, i9V, and i9W flowing through the resonant capacitors 9U, 9V, and 9W will be described as positive when flowing in the direction of the arrow in Figure 1, and negative when flowing in the opposite direction to the arrow in Figure 1. Therefore, in the discharge operation where the resonant capacitors 9U, 9V, and 9W are discharged, the polarity of the currents i9U, i9V, and i9W is positive, and in the charging operation where the resonant capacitors 9U, 9V, and 9W are charged, the polarity of the currents i9U, i9V, and i9W is negative. Furthermore, in the following explanation, the voltage value of the DC power supply E1 will be Vd (see Figures 4 to 9).

[0043] In the power converter 100, for example, the third switching element 6U of the switch 8U may change from a state where a current iL1 flows in the resonant inductor L1 with positive polarity when the switch 8U is ON to a state where the third switching element 6U of the switch 8U is OFF. In this case, the current iL1 flowing through the resonant inductor L1 is regenerated to the power converter circuit 11 via the third diode 13 until the energy of the resonant inductor L1 is consumed and the current iL1 becomes zero. Also, in the power converter 100, for example, the fourth switching element 7U of the switch 8U may change from a state where a current iL1 flows in the resonant inductor L1 with negative polarity when the switch 8U is ON to a state where the fourth switching element 7U is OFF. In this case, the current iL1 flowing through the resonant inductor L1 flows through the path of the fourth diode 14 - resonant inductor L1 - regenerative capacitor 15 until the energy of the resonant inductor L1 is consumed and the current iL1 becomes zero.

[0044] Furthermore, in the power conversion apparatus 100, for example, there is a case where the third switching element 6V of the switch 8V changes from an on state, in which the current iL1 flows through the resonance inductor L1 with a positive polarity, to an off state. In this case, the current iL1 flowing through the resonance inductor L1 is regenerated to the power conversion circuit 11 via the third diode 13 until the energy of the resonance inductor L1 is consumed and the current iL1 becomes zero. Furthermore, in the power conversion apparatus 100, for example, there is a case where the fourth switching element 7V of the switch 8V turns off from an on state, in which the current iL1 flows through the resonance inductor L1 with a negative polarity. In this case, the current iL1 flowing through the resonance inductor L1 flows along a path of the fourth diode 14 - the resonance inductor L1 - the regeneration capacitor 15 until the energy of the resonance inductor L1 is consumed and the current iL1 becomes zero.

[0045] Furthermore, in the power conversion apparatus 100, for example, there is a case where the third switching element 6W of the switch 8W turns off from an on state, in which the current iL1 flows through the resonance inductor L1 with a positive polarity. In this case, the current iL1 flowing through the resonance inductor L1 is regenerated to the power conversion circuit 11 via the third diode 13 until the energy of the resonance inductor L1 is consumed and the current iL1 becomes zero. Furthermore, in the power conversion apparatus 100, for example, there is a case where the fourth switching element 7W of the switch 8W turns off from an on state, in which the current iL1 flows through the resonance inductor L1 with a negative polarity. In this case, the current iL1 flowing through the resonance inductor L1 flows along a path of the fourth diode 14 - the resonance inductor L1 - the regeneration capacitor 15 until the energy of the resonance inductor L1 is consumed and the current iL1 becomes zero.

[0046] The control device 50 sets a dead time period Td (see Figures 4 to 9) between the high-level period of the first PWM signals SU1, SV1, SW1 to the first switching elements 1U, 1V, 1W and the high-level period of the second PWM signals SU2, SV2, SW2 to the second switching elements 2U, 2V, 2W for each of the multiple switching circuits 10. The control device 50 also sets the high-level period of the control signal to each of the multiple switches 8 based on the dead time period Td for the corresponding switching circuit 10 among the multiple switching circuits 10. Here, the control device 50 sets the length of the high-level period of the control signal to each of the multiple switches 8 as, for example, the sum of the length of the first period and the length of the second period. The length of the first period is an integer (N) multiple of the resonant half-period determined by the capacitance of the resonant capacitor 9 corresponding to the switch 8 and the inductance of the resonant inductor L1. If the resonant period is Tres, the length of the first period is N × (Tres / 2). It is desirable that the end of the first period, that is, the end of an integer multiple of the resonant half-period, coincides with the end of the dead time period Td for the switching circuit 10 corresponding to the switch 8. For example, the example in Figure 7 is when N = 1, and the capacitance of the resonant capacitor 9 and the inductance of the resonant inductor L1 are selected so that Tres / 2 is shorter than the dead time period Td. The length of the second period is, as an example, an additional time (additional time Tau in the example in Figure 7) determined by the voltage V15 of the regenerative capacitor 15, the inductance of the resonant inductor L1, and the load current value. The length of the first period described above is an ideal design example, and may be 90% to 110% of the length N × (Tres / 2). Furthermore, the length of the second period described above is an ideal design example, and may be 90% to 110% of the additional time determined by the voltage of the regenerative capacitor 15, the inductance of the resonant inductor L1, and the load current value.

[0047] Hereinafter, the basic operation of zero-voltage soft switching for each of the plurality of first switching elements 1 and the plurality of second switching elements 2 will be described with reference to FIGS. 4 to 9. The basic operation is an operation in which resonance currents passing through two or more of the plurality of switches 8 do not simultaneously flow through the resonance inductor L1. After describing the basic operation, an operation when the control device 50 determines that two-phase resonance currents flow simultaneously will be described.

[0048] (3.1) Basic Operation In zero-voltage soft switching of the first switching element 1, it is necessary to set the voltage across the first switching element 1 to zero immediately before turn-on of the first switching element 1 that is the target of zero-voltage soft switching. Also, in zero-voltage soft switching of the second switching element 2, it is necessary to set the voltage across the second switching element 2 to zero immediately before turn-on of the second switching element 2 that is the target of zero-voltage soft switching. Hereinafter, the switching element (the first switching element 1 or the second switching element 2) that is a target of zero-voltage soft switching is also referred to as a target switching element.

[0049] The basic operation of the control device 50 differs depending on the difference between the polarity (positive / negative) of a load current flowing through an AC terminal 41 connected to the target switching element, and the operation (charging operation / discharging operation) of a resonance capacitor 9 connected in series or parallel to the target switching element. The load current has a positive polarity when flowing from the AC terminal 41 toward the AC load RA1, and has a negative polarity when flowing from the AC load RA1 toward the AC terminal 41. During the charging operation of the resonance capacitor 9, the voltage across the resonance capacitor 9 increases. Also, during the discharging operation of the resonance capacitor 9, the voltage across the resonance capacitor 9 decreases. The voltage across each of the plurality of second switching elements 2 is the same as the voltage across the resonance capacitor 9 connected in parallel to the second switching element 2.

[0050] (3.1.1) When the load current > 0, the operation control device 50 for zero-voltage soft switching of the first switching element turns on the third switching element 6 corresponding to the first switching element 1 when the target of zero-voltage soft switching is the first switching element 1 (hereinafter also referred to as the target first switching element 1) and the polarity of the load current flowing through the AC terminal 41 connected to the target first switching element 1 is positive. As a result, the control device 50 causes the resonant inductor L1 and the resonant capacitor 9 connected to the target first switching element 1 to resonate, charging the resonant capacitor 9 from the regenerative capacitor 15, and setting the voltage across the target first switching element 1 to zero. As a result, the power converter 100 can achieve zero-voltage soft switching of the target first switching element 1.

[0051] Figure 4 shows the first PWM signal SU1, the second PWM signal SU2, the control signal SU6, the load current iU, the current iL1, the voltage V1u across the first switching element 1U, and the voltage V2u across the second switching element 2U, in the case where the target first switching element is the first switching element 1U of the switching circuit 10U. Figure 4 also shows the additional time Tau set for the control signal SU6 in the control device 50. The additional time Tau will be described later.

[0052] Figure 5 also illustrates the case where the target first switching element is the first switching element 1V of the switching circuit 10V, showing the first PWM signal SV1, the second PWM signal SV2, the control signal SV6, the load current iV, the current iL1, the voltage V1v across the first switching element 1V, and the voltage V2v across the second switching element 2V. Figure 5 also shows the additional time Tav set for the control signal SV6 in the control device 50. The additional time Tav will be described later.

[0053] Figure 6 shows the first PWM signal SW1, the second PWM signal SW2, the control signal SW6, the load current iW, the current iL1, the voltage V1w across the first switching element 1W, and the voltage V2w across the second switching element 2W, in the case where the target first switching element is the first switching element 1W of the switching circuit 10W. Figure 6 also shows the additional time Taw set for the control signal SW6 in the control device 50. The additional time Taw will be described later.

[0054] The additional time Tau described above is set to achieve zero-voltage soft switching of the target switching element (first switching element 1U) regardless of the value of the load current iU, by making the high-level period of the control signal SU6 longer than the resonant half-period (Tres / 2), as shown in Figure 4. The length of the additional time Tau is set based on the value of the load current iU. The end of the high-level period of the control signal SU6 can be the same as or later than the end of the dead time period Td t3. Figure 4 shows an example where the end of the high-level period of the control signal SU6 is set later than the end of the dead time period Td t3. In the switching circuit 10U, the voltage V2u across the second switching element 2U becomes Vd at the end of the dead time period Td t3, and the voltage V1u across the first switching element 1U becomes zero at the end of the dead time period Td t3. In the example shown in Figure 4, the current iL1 flowing through the resonant inductor L1 begins at time t1, when the high-level period of the control signal SU6 starts, and becomes zero at time t4, after an additional time Tau has elapsed from time t3, when the dead time period Td ends. With respect to the current iL1, from time t2, when the half-cycle of resonance begins, iL1 ≥ iU, and the current iL1 in the shaded region of the fourth current waveform from the top in Figure 4 flows into the resonant capacitor 9U, causing LC resonance. After time t3, when the dead time period Td ends, the current iL1 is regenerated to the power conversion circuit 11 via the third diode 13 directly connected to the resonant inductor L1.

[0055] As described above, the control device 50 determines the additional time Tau based on the load current iU such that iL1 = iU at time t2 when the resonant half-cycle begins, in order to start LC resonance at time t2 when the resonant half-cycle begins and end the resonant half-cycle at time t3 when the dead time period Td ends. More specifically, the control device 50 determines the additional time Tau by calculating Tau = iU × (L / V15) using, for example, the detection result of the load current iU by the current sensor 18a (analog detection value) or its signal processing value (A / D converted digital detection value), or an estimated value of the load current iU, the inductance L of the resonant inductor L1 which is stored in advance, and the detection result of the voltage V15 of the regenerative capacitor 15 (potential V15 at the sixth terminal 154 of the regenerative capacitor 15). The detected load current iU or its signal processing value at this time is the detected load current iU at the period in which additional time Tau is added to Tres / 2 in the carrier signal CA1, or at the timing closest to that period. The detected load current iU used in the above calculation in the control device 50 is the detected load current iU (analog value) by the current sensor 18a or the detected value (digital value) after A / D conversion of the detected load current (analog value) by the current sensor 18a by the second A / D conversion unit 54 described later. The estimated value of the load current iU at this time is the value estimated from the load current iU at the period in which additional time Tau is added to Tres / 2 in the carrier signal CA1. The resonant half-period in the basic operation is half of the resonant period, which is the reciprocal of the resonant frequency of the resonant circuit including the resonant inductor L1 and one resonant capacitor 9. Therefore, if the inductance of the resonant inductor L1 is L and the capacitance of the resonant capacitor 9 is C, then the resonant half-period is π × (L・C) １／２ That is the case.

[0056] The additional time Tav mentioned above is set to achieve zero-voltage soft switching of the target switching element (first switching element 1V) regardless of the load current iv, by making the high-level period of the control signal SV6 longer than the resonant half-period (Tres / 2), as shown in Figure 5. The length of the additional time Tav is set based on the value of the load current iv. The end of the high-level period of the control signal SV6 can be the same as or later than the end of the dead time period Td t7. Figure 5 shows an example where the end of the high-level period of the control signal SV6 is set later than the end of the dead time period Td t7. In the switching circuit 10V, the voltage V2v across the second switching element 2V becomes Vd at the end of the dead time period Td t7, and the voltage V1v across the first switching element 1V becomes zero at the end of the dead time period Td t7. In the example shown in Figure 5, the current iL1 flowing through the resonant inductor L1 begins at time t5, when the high-level period of the control signal SV6 starts, and becomes zero at time t8, after the additional time Tav has elapsed from time t7, when the dead time period Td ends. With respect to the current iL1, from time t6, when the half-cycle of resonance begins, iL1 ≥ iV, and the current iL1 in the shaded region of the fourth current waveform from the top in Figure 5 flows into the resonant capacitor 9V, causing LC resonance. After time t7, when the dead time period Td ends, the current iL1 is regenerated to the power conversion circuit 11 via the third diode 13 directly connected to the resonant inductor L1.

[0057] As described above, the control device 50 determines the additional time Tav based on the load current iv such that iL1 = iv at the time t6 when the resonant half-cycle begins, in order to start LC resonance at time t6 when the resonant half-cycle begins and end the resonant half-cycle at time t7 when the dead time period Td ends. More specifically, the control device 50 determines the additional time Tav by calculating Tav = iv × (L / V15) using, for example, the detection result of the load current iv by the current sensor 18b (analog detection value) or its signal processing value (A / D converted digital detection value), or an estimated value of the load current iv, the inductance L of the resonant inductor L1 stored in advance, and the detection result of the voltage V15 of the regenerative capacitor 15. The detection result of the load current iv at this time or its signal processing value is the detection value of the load current iv at the time when the additional time Tav is added to Tres / 2 in the carrier signal CA1, or at the timing closest to that time. In the control device 50, the detected value of the load current iV used in the above calculation is either the detected value of the load current iV by the current sensor 18b (analog value) or the detected value (digital value) after A / D conversion of the detected value of the load current iV by the current sensor 18b (analog value) by the second A / D conversion unit 54 described later. Furthermore, the estimated value of the load current iV at this time is the value estimated from the load current iV at the period in which the additional time Tav is added to Tres / 2 in the carrier signal CA1.

[0058] The additional time Taw described above is set to achieve zero-voltage soft switching of the target switching element (first switching element 1W) regardless of the value of the load current iW, by making the high-level period of the control signal SW6 longer than the resonant half-period (Tres / 2), as shown in Figure 6. The length of the additional time Taw is set based on the value of the load current iW. The end of the high-level period of the control signal SW6 can be the same as or later than the end of the dead time period Td t11. Figure 6 shows an example where the end of the high-level period of the control signal SW6 is set later than the end of the dead time period Td t11. In the switching circuit 10W, the voltage V2w across the second switching element 2W becomes Vd at the end of the dead time period Td t11, and the voltage V1w across the first switching element 1W becomes zero at the end of the dead time period Td t11. In the example shown in Figure 6, the current iL1 flowing through the resonant inductor L1 begins at time t9, when the high-level period of the control signal SW6 starts, and becomes zero at time t12, after the additional time Taw has elapsed from time t11, when the dead time period Td ends. With respect to the current iL1, from time t10, when the half-cycle of resonance begins, iL1 ≥ iW, and the current iL1 in the shaded region of the fourth current waveform from the top in Figure 6 flows into the resonant capacitor 9W, causing LC resonance. After time t11, when the dead time period Td ends, the current iL1 is regenerated to the power conversion circuit 11 via the third diode 13 directly connected to the resonant inductor L1.

[0059] As described above, the control device 50 starts LC resonance at time t10 when the resonant half-cycle begins and ends the resonant half-cycle at time t11 when the dead time period Td ends, by determining the additional time Taw based on the load current iW such that iL1 = iW at time t10 when the resonant half-cycle begins. More specifically, the control device 50 determines the additional time Taw by calculating Taw = iW × (L / V15) using, for example, the detection result of the load current iW by the current sensor 18c (analog detection value) or its signal processing value (A / D converted digital detection value), or an estimated value of the load current iW, the inductance L of the resonant inductor L1 stored in advance, and the detection result of the voltage V15 of the regenerative capacitor 15. The detection result of the load current iW at this time or its signal processing value is the detection value of the load current iW at the time when the additional time Taw is added to Tres / 2 in the carrier signal CA1, or at the timing closest to that time. In the control device 50, the detected value of the load current iW used in the above calculation is either the detected value of the load current iW by the current sensor 18c (analog value) or the detected value (digital value) after A / D conversion of the detected value of the load current iW by the current sensor 18c (analog value) by the second A / D conversion unit 54 described later. Furthermore, the estimated value of the load current iW at this time is the estimated value of the load current iW at the carrier period in which the additional time Taw is added, etc.

[0060] (3.1.2) The control device 50 for zero-voltage soft switching of the second switching element when the load current > 0 compares the current value of the load current with the first current threshold when the target of zero-voltage soft switching is the second switching element 2 (hereinafter also referred to as the target second switching element 2), and the polarity of the load current (load current iU, load current iV, or load current iW) flowing through the AC terminal 41 connected to the target second switching element 2 is positive. As an example of the design of the operation of the control device 50, if the current value of the load current is greater than the first current threshold, the control device 50 does not turn on the switch 8 corresponding to the target second switching element 2, and if the current value of the load current is less than the first current threshold, it turns on the switch 8 corresponding to the target second switching element 2 during the dead time period Td. Here, as in section (3.1.1), the resonant half-period is, for example, shorter than the length of the dead time period Td. In the power converter 100, when the load current value is greater than the first current threshold, the control device 50 can discharge the resonant capacitor 9U connected in parallel to the target second switching element 2 using the load current iU without turning on the switch 8 corresponding to the target second switching element 2. As a result, the power converter 100 can achieve zero-voltage soft switching of the target second switching element 2.

[0061] Figure 7 shows the first PWM signal SU1, the second PWM signal SU2, the control signal SU7, the load current iU, the current i9U, and the voltage V2u across the second switching element 2U in the case where the target second switching element 2 is the second switching element 2U of the switching circuit 10U, and the current value of the load current iU is greater than the first current threshold. Also in Figure 7, the additional time Tau set in the control device 50 for the control signal SU7 of the fourth switching element 7U of the switch 8U.

[0062] The control device 50 does not provide a high-level period for the control signal SU7 when the load current iU is greater than the first current threshold. In this case, in the power converter 100, current i9U starts flowing from the resonant capacitor 9U at time t22 when the resonant half-cycle begins, and the current i9U drops to zero before time t23 when the dead time period Td ends, and the voltage V2u across the second switching element 2U becomes zero before time t23 when the dead time period Td ends. As a result, in the power converter 100, when the second PWM signal SU2 changes from a low level to a high level at time t23 when the dead time period Td ends, the second switching element 2U is soft-switched to zero voltage.

[0063] When the load current iU is less than the first current threshold, the control device 50 provides a high-level period for the control signal SU7, for example, as shown by the dashed line in Figure 7. The start time of the high-level period for the control signal SU7 is, for example, the same as the start time t22 of the resonant half-cycle. The end time of the high-level period for the control signal SU7 is the same as the end time t23 of the dead time period Td. As a result, in the power converter 100, the voltage V2u across the second switching element 2U becomes zero before the end time t23 of the dead time period Td. Therefore, in the power converter 100, when the second PWM signal SU2 changes from a low level to a high level at the end time t23 of the dead time period Td, the second switching element 2U is soft-switched to zero voltage. The start time of the high-level period for the control signal SU7 may be a time t21 that is earlier by an additional time Tau than the start time t22 of the resonant half-cycle. The end of the high-level period of the control signal SU7 may be at a time t24 that is later than the end of the dead time period Td (t23) by an additional time Tau. The additional time before and after the period overlapping with the dead time period Td in the high-level period of the control signal SU7 is not limited to the additional time Tau, but may be any other set time. Furthermore, the relationship between the first current threshold of the load current and the method for determining the high-level period of the control signal to switch 8 described above is an ideal design example and is not limited to the above design example. For example, the control device 50 may set the high-level period of the control signal for switch 8 so that switch 8 is turned on during the dead time period Td, even if the current value of the load current is greater than the first current threshold. Also, the control device 50 does not have to turn on switch 8 during the dead time period Td even if the current value of the load current is less than the first current threshold. Furthermore, the control device 50 may set the high-level period of the control signal for switch 8 so that switch 8 is always turned on during the dead time period Td, regardless of the first current threshold. Furthermore, the control device 50 may keep the switch 8 in the off state at all times, regardless of the first current threshold.

[0064] (3.1.3) The operation control device 50 for zero-voltage soft switching of the second switching element when the load current is less than 0 turns on the fourth switching element 7 corresponding to the second switching element 2 when the polarity of the load current (load current iU, load current iV, or load current iW) flowing through the AC terminal 41 connected to the target second switching element 2 is negative. As a result, the control device 50 causes the resonant capacitor 9 and the resonant inductor L1 connected to the target second switching element 2 to resonate, causing the resonant capacitor 9 to discharge and making the voltage across the target second switching element 2 zero. As a result, the power converter 100 can achieve zero-voltage soft switching of the target second switching element 2.

[0065] Figure 8 shows the first PWM signal SU1, the second PWM signal SU2, the control signal SU7, the load current iU, the current iL1, and the voltage V2u across the second switching element 2U, in the case where the target second switching element 2 is the second switching element 2U of the switching circuit 10U. Figure 8 also shows the additional time Tau that the control device 50 sets for the control signal SU7 of the fourth switching element 7U of the switch 8U. The end time of the high-level period of the control signal SU7 can be the same as or later than the end time t33 of the dead time period Td. Figure 8 shows an example where the end time of the high-level period of the control signal SU7 is set to be the same as the end time t33 of the dead time period Td. The control device 50 sets the high-level period of the control signal SU7 to Tau + (Tres / 2). In the switching circuit 10U, the voltage V2u across the second switching element 2U becomes zero at time t33, when the dead time period Td ends. In the example in Figure 8, the current iL1 flowing through the resonant inductor L1 begins to flow from time t31, when the high-level period of the control signal SU7 begins, and becomes zero at time t34, when an additional time Tau has elapsed from time t33, when the dead time period Td ends. With respect to the current iL1, iL1 ≤ iU occurs from time t32, when the dead time period Td begins, causing LC resonance to occur and a resonant current (discharge current of the resonant capacitor 9U) to flow from the resonant capacitor 9U towards the resonant inductor L1. After time t33, when the dead time period Td ends, the current iL1 is regenerated to the power conversion circuit 11 via the fourth diode 14, which is directly connected to the resonant inductor L1.

[0066] The control device 50 determines the additional time Tau based on the load current iU so that iL1 = iU at the time t32 when the resonant half-cycle begins, in order to start LC resonance at time t32 when the dead time period Td ends at time t33. More specifically, the control device 50 determines the additional time Tau by calculating Tau = |iU| × (L / V15) using, for example, the detection result of the output current iU by the current sensor 18a or its signal processing value, or an estimated value of the load current iU, the inductance L of the resonant inductor L1 which is stored in advance, and the detection result of the voltage V15 of the regenerative capacitor 15. The load current value at this time (the detection result of the load current iU or its signal processing value) is the detected value of the load current iU at the period in which the additional time Tau is added to Tres / 2, or at the timing closest to that period. Furthermore, the estimated value of the load current iU at this time is obtained by estimating the load current iU over a period in which an additional time Tau is added to Tres / 2, etc.

[0067] (3.1.4) The control device 50 for zero-voltage soft switching of the first switching element when the load current < 0 compares the current value of the load current (load current iU, load current iV, or load current iW) flowing through the AC terminal 41 connected to the target first switching element 1 with a second current threshold when the polarity of the load current is negative. The second current threshold is a negative value, and the absolute value of the second current threshold is, for example, the same as the value of the first current threshold. As an example of the design of the operation of the control device 50, if the current value of the load current is less than the second current threshold, the control device 50 does not turn on the switch 8 corresponding to the target first switching element 1, and if the current value of the load current is greater than the second current threshold, it turns on the switch 8 corresponding to the target first switching element 1 during the dead time period Td. Here, as in section (3.1.1), the resonant half-period is, for example, shorter than the length of the dead time period Td. When the load current is less than the second current threshold, the power converter 100 can charge the resonant capacitor 9U connected in series with the target first switching element 1 using the load current without the control device 50 turning on the switch 8 corresponding to the target first switching element 1. This enables the power converter 100 to achieve zero-voltage soft switching of the target first switching element 1.

[0068] Figure 9 shows the first PWM signal SU1, the second PWM signal SU2, the control signal SU6, the load current iU, the current i9U, the voltage V1u across the first switching element 1U, and the voltage V2u across the second switching element 2U when the target first switching element 1 is the first switching element 1U of the switching circuit 10U, and the current value of the load current is greater than the second current threshold (in other words, when the absolute value of the load current is less than the absolute value of the second current threshold).

[0069] The control device 50 does not provide a high-level period for the control signal SU6 when the load current value is smaller than the second current threshold (in other words, when the absolute value of the load current is larger than the absolute value of the second current threshold). In this case, in the power converter 100, after the dead time period Td has started, current i9U begins to flow through the resonant capacitor 9U at time t41, when the resonant half-cycle begins. As a result, in the power converter 100, the resonant capacitor 9U is charged and the voltage V2u across the second switching element 2U increases, the current i9U becomes zero before time t42, when the dead time period Td ends, and the voltage V1u across the first switching element 1U becomes zero before time t42, when the dead time period Td ends. As a result, in the power converter 100, when the first PWM signal SU1 changes from a low level to a high level at time t42, when the dead time period Td ends, the first switching element 1U is soft-switched at zero voltage.

[0070] The control device 50, when the load current value is greater than the second current threshold (in other words, when the absolute value of the load current is less than the absolute value of the second current threshold), sets a high-level period for the control signal SU6, as shown by the dashed line in Figure 9. The start time of the high-level period for the control signal SU6 is the same as the start time t41 of the resonant half-cycle. The end time of the high-level period for the control signal SU6 is the same as the end time t42 of the dead time period Td. As a result, in the power converter 100, the voltage V1u across the first switching element 1U becomes zero before the end time t42 of the dead time period Td. Therefore, in the power converter 100, when the first PWM signal SU1 changes from a low level to a high level at the end time t42 of the dead time period Td, the first switching element 1U is soft-switched to zero voltage. Furthermore, the relationship between the second current threshold of the load current and the method for determining the high-level period of the control signal to switch 8, as described above, is an ideal design example and is not limited to the above design example. For example, the control device 50 may set the high-level period of the control signal to switch 8 so that switch 8 turns on during the dead time period Td, even when the load current value is smaller than the second current threshold. Also, the control device 50 does not have to turn on switch 8 during the dead time period Td even when the load current value is larger than the second current threshold. Also, the control device 50 may set the high-level period of the control signal to switch 8 so that switch 8 is always turned on during the dead time period Td, regardless of the second current threshold. Also, the control device 50 may keep switch 8 always in the off state, regardless of the second current threshold. Also, the control device 50 does not have to make the high-level period of the control signal to switch 8 coincide with the dead time period Td as described above. For example, the high-level period of the control signal to switch 8 may be designed to be other than the length of the dead time period Td, depending on the design time of the resonant half-period.

[0071] (3.2) The current protection operation control device 50 stops the operation of the power conversion circuit 11 when the input current detected value by the current detection unit 16 exceeds the current protection threshold Ith.

[0072] The control device 50 includes, for example, a first A / D conversion unit 53 that performs A / D conversion on the input current detection value (analog value) of the current detection unit 16. By comparing the input current detection value (digital value) after A / D conversion with the digital value of the current protection threshold Ith, the control device 50 determines whether the input current detection value from the current detection unit 16 exceeds the current protection threshold Ith.

[0073] In the control device 50, the current protection threshold Ith during basic operation is defined as the first current protection threshold I1 (see Figure 3). The first current protection threshold I1 is, for example, 5 [A].

[0074] The control device 50 stops the operation of the power conversion circuit 11 when it determines that the input current detection value exceeds the current protection threshold Ith. When the control device 50 "stops the operation of the power conversion circuit 11", it means that the control device 50 sets each of the first PWM signals SU1, SV1, SW1 and the second PWM signals SU2, SV2, SW2 to a low level, or stops outputting the first PWM signals SU1, SV1, SW1 and the second PWM signals SU2, SV2, SW2.

[0075] (3.3) In the operating power converter 100 when two-phase resonant currents flow simultaneously, the duty cycle command values ​​of two of the three-phase duty cycle command values ​​du, dv, and dw approach each other at electrical angles of 60°, and during one cycle of the carrier signal CA1, the time difference between the start of the ON period of each of the two-phase switches 8 becomes shorter, and there is a possibility that the resonant currents flowing through the resonant capacitors 9 of the two switching circuits 10 corresponding to the two-phase duty cycle command values ​​may flow simultaneously in the resonant inductor L1. In region A1 of Figure 2, the duty cycle command values ​​du and dv are each 0.75 or close to 0.75. In region A2 of Figure 2, the duty cycle command values ​​dv and dw are each 0.25 or close to 0.25.

[0076] For example, in the example shown in Figure 2, if the duty cycle command values ​​du and dv are both 0.75 or close to 0.75, the time difference between the start of the high-level period of the control signal SU6 and the start of the high-level period of the control signal SV6 becomes shorter, and it is possible that the resonant current passing through the resonant capacitor 9U (hereinafter also referred to as the "U-phase resonant current") and the resonant current passing through the resonant capacitor 9V (hereinafter also referred to as the "V-phase resonant current") will flow through the resonant inductor L1 simultaneously. In this case, with respect to the current iL1 flowing through the resonant inductor L1, the polarity of the U-phase resonant current and the V-phase resonant current is positive.

[0077] Furthermore, in the example shown in Figure 2, if the duty cycle command values ​​dv and dw are both 0.75 or close to 0.75, the time difference between the start of the high-level period of the control signal SV6 and the start of the high-level period of the control signal SW6 becomes shorter, and it is possible that the resonant current of the V phase and the resonant current passing through the resonant capacitor 9W (hereinafter also referred to as the "W phase resonant current") will flow through the resonant inductor L1 simultaneously. In this case, with respect to the current iL1 flowing through the resonant inductor L1, the polarity of the V phase resonant current and the W phase resonant current is positive.

[0078] Furthermore, in the example shown in Figure 2, if the duty cycle command values ​​dw and du are both 0.75 or close to 0.75, the time difference between the start of the high-level period of the control signal SW6 and the start of the high-level period of the control signal SU6 becomes shorter, and it is possible that the resonant current of the W phase and the resonant current of the U phase will flow simultaneously through the resonant inductor L1. In this case, with respect to the current iL1 flowing through the resonant inductor L1, the polarity of the resonant current of the W phase and the resonant current of the U phase is positive.

[0079] Furthermore, in the example shown in Figure 2, if the duty cycle command values ​​du and dv are both 0.25 or close to 0.25, the time difference between the start of the high-level period of the control signal SU7 and the start of the high-level period of the control signal SV7 becomes shorter, and it is possible that the resonant current of the U-phase and the resonant current of the V-phase will flow simultaneously through the resonant inductor L1. In this case, with respect to the current iL1 flowing through the resonant inductor L1, the polarity of the resonant current of the U-phase and the resonant current of the V-phase is negative.

[0080] Furthermore, in the example shown in Figure 2, if the duty cycle command values ​​dv and dw are both 0.25 or close to 0.25, the time difference between the start of the high-level period of the control signal SV7 and the start of the high-level period of the control signal SW7 becomes shorter, and it is possible that the resonant current of the V-phase and the resonant current of the W-phase will flow simultaneously through the resonant inductor L1. In this case, with respect to the current iL1 flowing through the resonant inductor L1, the polarity of the resonant current of the V-phase and the resonant current of the W-phase is negative.

[0081] Furthermore, in the example shown in Figure 2, if the duty cycle command values ​​dw and du are both 0.25 or close to 0.25, the time difference between the start of the high-level period of the control signal SW7 and the start of the high-level period of the control signal SU7 becomes shorter, and it is possible that the resonant current passing through the resonant capacitor 9W and the resonant current passing through the resonant capacitor 9U will flow through the resonant inductor L1 simultaneously. In this case, with respect to the current iL1 flowing through the resonant inductor L1, the polarity of the resonant current passing through the resonant capacitor 9W and the resonant current passing through the resonant capacitor 9U is negative.

[0082] In the power converter 100, when two-phase current flows simultaneously through the resonant inductor L1, current flows simultaneously through each of the two resonant capacitors 9. In the power converter 100, when two-phase resonant current flows simultaneously through the resonant inductor L1, the resonant frequency of the resonant circuit including the resonant inductor L1 changes compared to when one-phase current flows through the resonant inductor L1. As a result, the resonant capacitors 9 are not sufficiently charged, and for example, as shown in Figure 3, the input current Iin may temporarily become large.

[0083] The control device 50 determines, at each control cycle, whether two-phase resonant currents corresponding to two of the multiple switching circuits 10 will flow simultaneously through the resonant inductor L1 in the next control cycle. If it determines that two-phase resonant currents will flow simultaneously, it increases the current protection threshold Ith. In this embodiment, the control cycle is the same as one cycle of the carrier signal CA1.

[0084] "When it is determined that two-phase resonant currents flow simultaneously" means when the control device 50 has previously estimated that resonant currents corresponding to each of the two switching circuits 10 will flow simultaneously through the resonant inductor L1. For the sake of explanation, below, "two-phase resonant currents corresponding to each of the two switching circuits 10 flow simultaneously through the resonant inductor L1" may also be referred to as "two-phase resonant currents flowing simultaneously through the resonant inductor L1."

[0085] The control device 50 first determines whether the resonant currents of two of the U-phase, V-phase, and W-phase overlap. If it determines in advance that the resonant currents of two phases flow simultaneously through the resonant inductor L1, it sets the current protection threshold Ith to a second current protection threshold I2 (see Figure 3), which is greater than the first current protection threshold I1 (see Figure 3). In this embodiment, the first current protection threshold I1 is set to 5 [A] and the second current protection threshold I2 is set to 8 [A], but the first current protection threshold I1 is not limited to 5 [A], and the second current protection threshold is not limited to 8 [A]. Furthermore, if the control device 50 determines that the resonant currents of two phases do not flow simultaneously after setting the current protection threshold Ith to the second current protection threshold I2, it sets the current protection threshold Ith to the first current protection threshold I1. In the example shown in Figure 3, if the control device 50 determines in the nth control period (where n is a natural number) that the resonant current of the U phase and the resonant current of the V phase flow simultaneously in the (n+1)th control period, it sets the current protection threshold Ith in the (n+1)th control period as the second current protection threshold I2. If the control device 50 determines in the (n+1)th control period that the resonant current of the U phase and the resonant current of the V phase do not flow simultaneously in the (n+2)th control period, it sets the current protection threshold Ith in the (n+2)th control period as the first current protection threshold I1.

[0086] Furthermore, if the control device 50 determines that two-phase resonant currents corresponding to two of the multiple switching circuits 10 are flowing simultaneously through the resonant inductor L1, it invalidates the detected values ​​of the two-phase load currents corresponding to the two switching circuits 10 (invalidates the detected values ​​of the two-phase load currents). The control device 50 includes a second A / D conversion unit 54 that performs A / D conversion on the detected values ​​(analog values) of each current sensor 18a, 18b, and 18c. "Invalidating the detected values ​​of the two-phase load currents" means that the second A / D conversion unit 54 does not perform A / D conversion on the detected values ​​(analog values) of the two-phase load currents, or that the second A / D conversion unit 54 performs A / D conversion on the two-phase load currents and invalidates the detected values ​​(digital values) after A / D conversion. In the example in Figure 3, if it is determined that the U-phase resonant current and the V-phase resonant current flow simultaneously in the (n+1)th control cycle during the nth (n is a natural number) control cycle, the detected values ​​of the two-phase load currents iU and iV are invalidated. If it is determined that the U-phase resonant current and the V-phase resonant current do not flow simultaneously in the (n+2)th control cycle during the (n+1)th control cycle, the detected values ​​of the two-phase load currents iU and iV are validated.

[0087] In this embodiment, the control device 50 determines that the resonant currents of two phases overlap when the difference between the duty cycle command values ​​of any two of the multiple phases (du, dv, dw) is less than a predetermined threshold Δdth. For example, the control device 50 compares the predetermined duty cycle command values ​​(du, dv, dw) of multiple phases. Here, the control device 50 calculates the difference Δd (Δduv = |du - dv|) between the duty cycle command value (du) of the U phase and the duty cycle command value (dv) of the V phase, and determines that the resonant currents of the U phase and the V phase overlap when Δduv is less than a predetermined threshold Δdth. In this embodiment, the predetermined threshold Δdth is set such that when Δduv is less than a predetermined threshold Δdth, the resonant current of the U phase and the resonant current of the V phase overlap in the resonant inductor L1, and when Δduv is greater than or equal to the predetermined threshold Δdth, the resonant current of the U phase and the resonant current of the V phase do not overlap in the resonant inductor L1. Furthermore, the control device 50 calculates the difference Δd (Δdvw = |dv - dw|) between the duty cycle command value dv of the V phase and the duty cycle command value dw of the W phase, and determines that the resonant current of the V phase and the resonant current of the W phase overlap when Δdvw is less than the predetermined threshold Δdth. In this embodiment, the predetermined threshold Δdth is set such that when Δdvw is less than a predetermined threshold Δdth, the resonant current of the V phase and the resonant current of the W phase overlap in the resonant inductor L1, and when Δdvw is greater than or equal to the predetermined threshold Δdth, the resonant current of the V phase and the resonant current of the W phase do not overlap in the resonant inductor L1. Furthermore, the control device 50 calculates the difference Δd (Δdwu = |dw - du|) between the duty cycle command value dw of the W phase and the duty cycle command value du of the U phase, and determines that the resonant current of the W phase and the resonant current of the U phase overlap when Δdwu is less than the predetermined threshold Δdth. In this embodiment, the case where the W-phase resonant current and the U-phase resonant current overlap in the resonant inductor L1 when Δdwu is less than a predetermined threshold Δdth is included, and the predetermined threshold Δdth is set so that the W-phase resonant current and the U-phase resonant current do not overlap in the resonant inductor L1 when Δdwu is equal to or greater than the predetermined threshold Δdth.

[0088] The operation example of the control device 50 will be explained below based on the flowchart shown in Figure 10. Figure 10 is a flowchart of the main function related to current control in the overall operation flowchart of the control device 50 (not shown).

[0089] The control device 50 sets the current protection threshold Ith to the first current protection threshold I1 (step S11).

[0090] After step S11, the control device 50 determines whether the resonant current of the U phase and the resonant current of the V phase overlap (flow simultaneously) (step S12).

[0091] If the control device 50 determines in step S12 that the resonant current of the U phase and the resonant current of the V phase overlap (step S12: Yes), it increases the current protection threshold Ith for the next control cycle (one cycle of the carrier signal CA1) from the first current protection threshold I1 to the second current protection threshold I2 (step S13).

[0092] In this embodiment, after step S13, the control device 50 invalidates the detected values ​​of the two-phase load current (step S14), and returns to the flowchart of the overall operation of the control device 50.

[0093] If the control device 50 determines in step S12 that the resonant current of the U phase and the resonant current of the V phase do not overlap (step S12: No), it determines whether the resonant current of the V phase and the resonant current of the W phase overlap (flow simultaneously) (step S15).

[0094] If the control device 50 determines in step S15 that the resonant current of the V phase and the resonant current of the W phase overlap (step S15: Yes), it increases the current protection threshold Ith in the next control cycle (one cycle of the carrier signal CA1) from the first current protection threshold I1 to the second current protection threshold I2 (step S13).

[0095] In this embodiment, after step S13, the control device 50 invalidates the detected values ​​of the two-phase load current (step S14), and returns to the flowchart of the overall operation of the control device 50.

[0096] If the control device 50 determines in step S15 that the resonant current of the V phase and the resonant current of the W phase do not overlap (step S15: No), it determines whether the resonant current of the W phase and the resonant current of the U phase overlap (flow simultaneously) (step S16).

[0097] If the control device 50 determines in step S16 that the resonant current of the W phase and the resonant current of the U phase overlap (step S16: Yes), it increases the current protection threshold Ith in the next control cycle (one cycle of the carrier signal CA1) from the first current protection threshold I1 to the second current protection threshold I2 (step S13).

[0098] In this embodiment, after step S13, the control device 50 invalidates the detected values ​​of the two-phase load current (step S14), and returns to the flowchart of the overall operation of the control device 50.

[0099] If the control device 50 determines in step S16 that the resonant current of the W phase and the resonant current of the U phase do not overlap (step S16: No), it returns to the flowchart of the overall operation of the control device 50.

[0100] Figure 3 illustrates the timing chart for two control cycles, showing the state of the first PWM signal SU1, the second PWM signal SU2, the first PWM signal SV1, the second PWM signal SV2, the control signals SU6, SU7, SV6, SV7, the current iL1, the current protection threshold Ith, the input current Iin, and the load current detection state. Figure 3 shows an example where, in the (n-1)th control cycle, it is determined that the two-phase resonant currents do not overlap in the nth control cycle, and an example where, in the nth control cycle, it is determined that the two-phase resonant currents overlap in the (n+1)th control cycle. In Figure 3, the U-phase resonant current and the V-phase resonant current, which partially overlap each other in the (n+1)th control cycle, are shown separately. In Figure 3, the nth control cycle is labeled "nth cycle," and the (n+1)th control cycle is labeled "(n+1)th cycle." In Figure 3, the "Normal Period" for "Overlap Judgment" means that when it is determined that the two-phase resonant currents do not overlap in the (n-1)th control cycle, it is determined that the two-phase resonant currents do not overlap in the nth control cycle. The "Overlap Judgment Period" for "Overlap Judgment" means that when it is determined that the two-phase resonant currents overlap in the nth control cycle, it is determined that the two-phase resonant currents overlap in the (n+1)th control cycle. The "Enabled" status for "Load Current Detection State" means that when it is determined in the (n-1)th control cycle that the U-phase resonant current and the V-phase resonant current do not flow simultaneously in the nth control cycle, the detected values ​​of the two-phase load currents iU and iV are enabled in the nth control cycle. The "Disabled" status means that when it is determined in the nth control cycle that the U-phase resonant current and the V-phase resonant current flow simultaneously in the (n+1)th control cycle, the detected values ​​of the two-phase load currents iU and iV are disabled in the (n+1)th control cycle.

[0101] (4) Advantages In the power conversion device 100 according to Embodiment 1, the current detection unit 16 detects the input current Iin flowing to the power conversion circuit 11. The control device 50 generates a plurality of first PWM signals SU1, SV1, SW1 for controlling a plurality of first switching elements 1, a plurality of second PWM signals SU2, SV2, SW2 for controlling a plurality of second switching elements 2, and a plurality of control signals SU6, SU7, SV6, SV7, SW6, SW7 for controlling a plurality of switches 8, based on a plurality of duty cycle command values ​​of a plurality of phases corresponding one-to-one to a plurality of switching circuits 10. The control device 50 generates a plurality of first PWM signals SU1, SV1, SW1 and a plurality of second PWM signals SU2, SV2, SW2 and a plurality of control signals SU6, SU7, SV6, SV7, SW6, SW7 for the (n+1)th control cycle in the nth control cycle. Here, n is a natural number. The control device 50 stops the operation of the power conversion circuit 11 if the input current detected by the current detection unit 16 exceeds the current protection threshold Ith. If the control device 50 determines during the nth control cycle that two-phase resonant currents corresponding to two of the multiple switching circuits 10 flow simultaneously through the resonant inductor L1 during the (n+1)th control cycle, it increases the current protection threshold Ith during the (n+1)th control cycle.

[0102] The above configuration makes it possible to reduce current distortion. More specifically, with the above configuration, if it is determined during the nth control cycle that two-phase resonant currents corresponding to two of the multiple switching circuits 10 flow simultaneously through the resonant inductor L1 during the (n+1)th control cycle, the current protection threshold Ith during the (n+1)th control cycle is increased, but the on-periods of the first and second switching elements in one of the switching circuits are not shifted, thus making it possible to reduce current distortion while preventing the operation of the power conversion circuit 11 from stopping. Furthermore, with the above configuration, since the shift control in the control device of the power conversion device disclosed in Patent Document 1 is not performed, the processing time for each control cycle in the control device 50 can be reduced. "Shift control" means a control in which the control device shifts the on-periods of the first and second switching elements in one of the two switching circuits.

[0103] Furthermore, in the power conversion device 100 according to Embodiment 1, if the control device 50 determines that two-phase resonant currents corresponding to two of the multiple switching circuits 10 are flowing simultaneously through the resonant inductor L1, it invalidates the detected values ​​of the two-phase load currents corresponding to the two switching circuits 10.

[0104] With the above configuration, it becomes possible to further reduce the processing time for each control cycle in the control device 50.

[0105] (Embodiment 2) Hereinafter, the power conversion device 100 according to Embodiment 2 will be described based on Figures 1, 11, and 12.

[0106] (1) The circuit configuration of the power converter 100 according to Embodiment 2 is the same as the circuit configuration of the power converter 100 according to Embodiment 1 (see Figure 1), so the illustration and description are omitted.

[0107] The power converter 100 according to Embodiment 2 differs from the power converter 100 according to Embodiment 1 in that the length of the resonant half-period (Tres / 2) in the control device 50 is set to be the same as the length of the dead time period Td. Furthermore, the power converter 100 according to Embodiment 2 differs from the power converter 100 according to Embodiment 1 in that the end of the high-level period of the control signal is made to coincide with the end of the dead time period Td.

[0108] Figure 11 shows the timing chart of the first PWM signal SU1, second PWM signal SU2, first PWM signal SV1, second PWM signal SV2, control signal SU6, control signal SV6, U-phase load current iU, V-phase load current iV, and current iL1 when the control device 50 determines that the resonant current of the U-phase and the V-phase resonant current overlap when the polarity of the U-phase load current iU and the V-phase load current iV are both positive.

[0109] In this embodiment, similar to Embodiment 1, if the control device 50 determines in the nth control cycle that the resonant current of the U phase and the resonant current of the V phase will overlap in the next (n+1)th control cycle, it sets the current protection threshold Ith in the (n+1)th control cycle to the second current protection threshold I2.

[0110] Figure 12 shows the timing chart of the first PWM signal SU1, second PWM signal SU2, first PWM signal SV1, second PWM signal SV2, control signal SU7, control signal SV7, U-phase load current iU, V-phase load current iV, and current iL1 when the control device 50 determines that the U-phase resonant current and the V-phase resonant current overlap when the polarity of the U-phase load current iU and the V-phase load current iV are negative.

[0111] In this embodiment, similar to Embodiment 1, if the control device 50 determines in the nth control cycle that the resonant current of the U phase and the resonant current of the V phase will overlap in the next (n+1)th control cycle, it sets the current protection threshold Ith in the (n+1)th control cycle to the second current protection threshold I2.

[0112] (2) Advantages The power converter 100 according to Embodiment 2 has the same advantages as the power converter 100 according to Embodiment 1.

[0113] (Other Modifications) Embodiments 1 and 2 described above are merely one of many embodiments of the present disclosure. Embodiments 1 and 2 described above can be modified in various ways depending on the design, etc., as long as the objectives of the present disclosure are achieved.

[0114] The carrier signal CA1 is used to generate multiple first PWM signals SU1, SV1, SW1 and multiple second PWM signals SU2, SV2, SW2. However, it is not limited to a triangular waveform; for example, a sawtooth waveform carrier signal may also be used.

[0115] In the power converter 100, the time difference between the start times of the high-level periods of the two first PWM signals corresponding to the first switching elements 1 of each of the two-phase switching circuits 10 may be used as a criterion for determining whether or not two-phase resonant currents flow simultaneously. For example, the control device 50 may determine (estimate) that if the time difference ΔTuv between the start time of the high-level period of the first PWM signal SU1 and the start time of the high-level period of the first PWM signal SV1 is greater than or equal to a threshold, the resonant currents of the U-phase and V-phase do not overlap in the resonant inductor L1, and if the time difference ΔTuv is less than the threshold, the resonant currents of the U-phase and V-phase overlap. The threshold may be set to, for example, the value of Tres / 2. The threshold is not limited to the value of Tres / 2. For example, considering errors in the value of Tres / 2, the threshold may be set to a value even larger than Tres / 2. Furthermore, the method for calculating the time difference used by the control device 50 to determine whether or not two-phase resonant currents flow simultaneously is not limited to the above example. For example, the control device 50 may use the time difference between the end times of the high-level periods of the two second PWM signals as the time difference used to determine whether or not two-phase resonant currents flow simultaneously. For example, the time difference ΔTuv used to determine whether or not the U-phase resonant current and the V-phase resonant current flow simultaneously may be the time difference between the end of the high-level period of the second PWM signal SU2 and the end of the high-level period of the second PWM signal SV2.

[0116] Furthermore, in the power conversion device 100 of Embodiments 1 and 2, the control device 50 may be configured not to perform the operation in step S14.

[0117] Furthermore, each of the multiple first switching elements 1 and the multiple second switching elements 2 is not limited to IGBTs, but may be, for example, a MOSFET. In this case, each of the multiple first diodes 4 may be replaced with a parasitic diode of the MOSFET constituting the corresponding first switching element 1 among the multiple first switching elements 1. Similarly, each of the multiple second diodes 5 may be replaced with a parasitic diode of the MOSFET constituting the corresponding second switching element 2 among the multiple second switching elements 2. The MOSFETs are, for example, Si-based MOSFETs or SiC-based MOSFETs. Each of the multiple first switching elements 1 and the multiple second switching elements 2 may be, for example, a bipolar transistor or a GaN-based GIT (Gate Injection Transistor).

[0118] Furthermore, in the power conversion device 100, if the capacitance of each of the multiple resonant capacitors 9 is relatively small, instead of externally attaching the multiple resonant capacitors 9, the parasitic capacitance between the ends of the multiple second switching elements 2 may serve as the multiple resonant capacitors 9.

[0119] Furthermore, the length of the resonant half-period is not limited to being shorter than the length of the dead time period Td; it may also be set to be the same length as the resonant half-period.

[0120] The dead time period Td may be set by a dead time generation circuit, such as a gate driver IC (Integrated Circuit), which is provided separately from the control device 50. Alternatively, the control device 50 may include a gate driver IC, and the dead time generation circuit of the gate driver IC may set the dead time period Td.

[0121] Regarding the setting methods for each of the multiple additional times Tau, Tav, and Taw described in the "(3.1) Basic Operation" section of Embodiment 1, the calculation formulas are ideal design examples and are not limited to cases where calculations are always performed using these formulas. In some cases, each of the multiple additional times Tau, Tav, and Taw may be set to 0 or another fixed time. Furthermore, values ​​obtained by calculations using other calculation formulas may be used as long as the purpose of each of the multiple additional times Tau, Tav, and Taw is achieved. For example, in the basic operation, the additional time Tau is determined by the calculation Tau = iU × (L / V15), but the additional time Tau may be set to 0 without performing the calculation Tau = iU × (L / V15), the additional time Tau may be set to between 0 and iU × (L / V15), it may be set to a constant additional time, it may be calculated using another calculation formula, or a combination of these may be used to set them. The settings for additional time Tav and additional time Taw are the same as for additional time Tau.

[0122] In each of the multiple switches 8, the second main terminal (emitter terminal) of the third switching element 6 is connected to the second main terminal (emitter terminal) of the fourth switching element 7, the first main terminal (collector terminal) of the third switching element 6 is connected to the connection point 3 of the corresponding switching circuit 10 among the multiple switching circuits 10, and the first main terminal (collector terminal) of the fourth switching element 7 is connected to the common connection point 25. In this case, each of the multiple switches 8 further has a diode antiparallel connected to the third switching element 6 and a diode antiparallel connected to the fourth switching element 7.

[0123] Furthermore, each of the third switching element 6 and the fourth switching element 7 may be replaced with a MOSFET or a bipolar transistor.

[0124] In each of the multiple switches 8, the third switching element 6 and the fourth switching element 7 may be connected in antiparallel. Furthermore, each of the multiple switches 8 may further include a diode connected in antiparallel to the third switching element 6 and a diode connected in antiparallel to the fourth switching element 7.

[0125] Furthermore, in each of the multiple switches 8, the third switching element 6 and the fourth switching element 7 may be MOSFETs.

[0126] Each of the multiple switches 8 may be a dual-gate type GaN-based GIT having a first source terminal, a first gate terminal, a second gate terminal, and a second source terminal. In this case, a control signal SU6 is applied between the first gate terminal and the first source terminal of the dual-gate type GaN-based GIT constituting switch 8U, and a control signal SU7 is applied between the second gate terminal and the second source terminal. Similarly, a control signal SV6 is applied between the first gate terminal and the first source terminal of the dual-gate type GaN-based GIT constituting switch 8V, and a control signal SV7 is applied between the second gate terminal and the second source terminal. Furthermore, a control signal SW6 is applied between the first gate terminal and the first source terminal of the dual-gate type GaN-based GIT constituting switch 8W, and a control signal SW7 is applied between the second gate terminal and the second source terminal.

[0127] In each of the power converter 100 of Embodiment 1 and the power converter 100 of Embodiment 2, a capacitor may be further provided, connected between the fourth end of the resonant inductor L1 and the first DC terminal 31. In this case, in each of the power converter 100 of Embodiment 1 and the power converter 100 of Embodiment 2, a series circuit of the capacitor and the regenerative capacitor 15 is connected between the first DC terminal 31 and the second DC terminal 32.

[0128] (Aspects) The following aspects are disclosed herein.

[0129] The power conversion device (100) according to the first embodiment includes a first DC terminal (31), a second DC terminal (32), a power conversion circuit (11), a plurality of AC terminals (41), a plurality of switches (8), a plurality of resonant capacitors (9), a resonant inductor (L1), a regenerative capacitor (15), a current detection unit (16), and a control device (50). The power conversion circuit (11) has a plurality of first switching elements (1) and a plurality of second switching elements (2). In the power conversion circuit (11), a plurality of switching circuits (10), each in which a plurality of first switching elements (1) and a plurality of second switching elements (2) are connected in series one to one, are connected in parallel to each other. In the power conversion circuit (11), a plurality of first switching elements (1) are connected to the first DC terminal (31), and a plurality of second switching elements (2) are connected to the second DC terminal (32). Multiple AC terminals (41) correspond one-to-one with multiple switching circuits (10). Each of the multiple AC terminals (41) is connected to the connection point (3) of the first switching element (1) and the second switching element (2) in the corresponding switching circuit (10) among the multiple switching circuits (10). Multiple switches (8) correspond one-to-one with multiple switching circuits (10). Each of the multiple switches (8) has a first end (81) and a second end (82). Each of the multiple switches (8) has its first end (81) connected to the connection point (3) of the first switching element (1) and the second switching element (2) in the corresponding switching circuit (10) among the multiple switching circuits (10), and its second end (82) is commonly connected to a common connection point (25). Multiple resonant capacitors (9) correspond one-to-one with multiple switches (8). Each of the multiple resonant capacitors (9) is connected between the first terminal (81) and the second DC terminal (32) of the corresponding switch (8) among the multiple switches (8). The resonant inductor (L1) has a third terminal and a fourth terminal. In the resonant inductor (L1), the third terminal of the resonant inductor (L1) is connected to a common connection point (25). The regenerative capacitor (15) has a fifth terminal (153) and a sixth terminal (154).In the regenerative capacitor (15), the fifth terminal (153) is connected to the second DC terminal (32), and the sixth terminal (154) is connected to the fourth terminal of the resonant inductor (L1). The current detection unit (16) detects the input current (Iin) flowing from the first DC terminal (31) to the power conversion circuit (11). The control device (50) generates multiple first PWM signals (SU1, SV1, SW1) to control multiple first switching elements (1), multiple second PWM signals (SU2, SV2, SW2) to control multiple second switching elements (2), and multiple control signals (SU6, SU7, SV6, SV7, SW6, SW7) to control multiple switches (8), based on multi-phase duty command values ​​(du, dv, dw) that correspond one-to-one to multiple switching circuits (10). The control device (50) generates multiple first PWM signals (SU1, SV1, SW1), multiple second PWM signals (SU2, SV2, SW2), and multiple control signals (SU6, SU7, SV6, SV7, SW6, SW7) in the nth control cycle, where n is a natural number. The control device (50) stops the operation of the power conversion circuit (11) if the input current detected by the current detection unit (16) exceeds the current protection threshold (Ith). If the control device (50) determines in the nth control cycle that two-phase resonant currents corresponding to two of the multiple switching circuits (10) flow simultaneously through the resonant inductor (L1) in the (n+1)th control cycle, it increases the current protection threshold (Ith) in the (n+1)th control cycle.

[0130] According to this embodiment, zero-voltage soft switching is possible, and current distortion can be reduced.

[0131] In the power converter (100) according to the second embodiment, in the first embodiment, if the control device (50) determines during the nth control cycle that two-phase resonant currents corresponding to two of the multiple switching circuits (10) flow simultaneously through the resonant inductor (L1) during the (n+1)th control cycle, it invalidates the detected values ​​of the two-phase load currents corresponding to the two switching circuits (10) during the (n+1)th control cycle.

[0132] According to this embodiment, it is possible to further reduce the processing time for each control cycle in the control device (50).

[0133] 1 First switching element 2 Second switching element 3 Connection point 8 Switch 9 Resonant capacitor 10 Switching circuit 11 Power conversion circuit 15 Regenerative capacitor 153 Fifth terminal 154 Sixth terminal 16 Current detection unit 31 First DC terminal 32 Second DC terminal 41 AC terminal 50 Control device 100, 100A Power converter CA1 Carrier signal du, dv, dw Duty shift command value iU, iv, iW Output current (load current) L1 Resonant inductor RA1 AC load SU1, SV1, SW1 First PWM signal SU2, SV2, SW2 Second PWM signal SU6, SU7, SV6, SV7, SW6, SW7 Control signal

Claims

1. A power conversion circuit comprising: a first DC terminal; a second DC terminal; a plurality of switching circuits having a plurality of first switching elements and a plurality of second switching elements, wherein the plurality of first switching elements and the plurality of second switching elements are connected in series one to one, and the plurality of first switching elements are connected to the first DC terminal and the plurality of second switching elements are connected to the second DC terminal; a plurality of AC terminals corresponding one to one of the plurality of switching circuits, each connected to the connection point of the first switching element and the second switching element in the corresponding switching circuit; a plurality of switches corresponding one to one of the plurality of switching circuits, each having a first end and a second end, the first end being connected to the connection point of the first switching element and the second switching element in the corresponding switching circuit and the second end being commonly connected to a common connection point; a plurality of resonant capacitors corresponding one to one of the plurality of switches, each connected between the first end and the second DC terminal of the corresponding switch; and a resonant inductor having a third end and a fourth end, the third end being connected to the common connection point. A regenerative capacitor having a fifth and a sixth terminal, the fifth terminal of which is connected to the second DC terminal and the sixth terminal of which is connected to the fourth terminal of the resonant inductor; a current detection unit for detecting the input current flowing from the first DC terminal to the power conversion circuit; a control device for generating a plurality of first PWM signals for controlling the plurality of first switching elements, a plurality of second PWM signals for controlling the plurality of second switching elements, and a plurality of control signals for controlling the plurality of switches, based on a plurality of duty cycle command values ​​of a plurality of phases corresponding one-to-one to the plurality of switching circuits, wherein the control device generates the plurality of first PWM signals, the plurality of second PWM signals and the plurality of control signals for the (n+1)th control cycle in the nth control cycle, where n is a natural number, and stops the operation of the power conversion circuit when the input current detected value by the current detection unit exceeds the current protection threshold.A power converter that, when it is determined during the nth control cycle that two-phase resonant currents corresponding to two of the plurality of switching circuits flow simultaneously through the resonant inductor during the (n+1)th control cycle, increases the current protection threshold during the (n+1)th control cycle.

2. The power conversion device according to claim 1, wherein the control device determines during the nth control cycle that two-phase resonant currents corresponding to two of the plurality of switching circuits flow simultaneously through the resonant inductor during the (n+1)th control cycle, and invalidates the detected values ​​of the two-phase load currents corresponding to the two switching circuits during the (n+1)th control cycle.

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