Ground wire energy taking device and method based on virtual impedance

Abstract

The invention discloses a ground wire energy taking device and method based on virtual impedance, the ground wire energy taking device comprises a primary side energy taking module installed on a power transmission line ground wire and a secondary side energy taking module arranged in mutual inductance with the primary side energy taking module, and the output end of the secondary side energy taking module is connected with a voltage type PWM rectifier; the voltage type PWM rectifier comprises power switch tubes Q1 to Q4, power switch tube body diodes D1 to D4, an inductor L and a filter capacitor Cf. The method has the advantages that the problems of CT parameter fluctuation, incomplete constant value capacitive element compensation and output power reduction caused by uncertain air gap width during mounting of a magnetic core opening and closing type structure of the ground wire energy taking device can be solved, and the energy taking device can realize maximum power output in a resonance state.

Description

Technical Field

[0001] The present invention relates to the technical field of power grids, and in particular to a ground wire energy harvesting device and energy harvesting method based on virtual impedance. Background Art

[0002] A stable, safe and reliable power grid is inseparable from the normal operation of online monitoring equipment. With the construction of smart grids and the access of distributed energy to the grid, a large number of condition monitoring devices have been applied in the grid, but the operating environment of the condition monitoring equipment has brought huge challenges to its own power supply. In the context of accelerating the construction of smart grids, the power supply technology of grid condition monitoring equipment needs to meet the following requirements: 1) It has an output power that matches the load and can meet the energy consumption requirements of the condition monitoring device; 2) It has high power supply reliability, strong environmental adaptability, and good robustness, and can provide reliable and stable power supply for the monitoring device under harsh environmental conditions; 3) It has a wide range of application scenarios and is easy to install, and can be used in different scenarios to power distributed sensor networks; 4) It has good economic benefits and can achieve long-term maintenance-free requirements.

[0003] Compared with transmission power supply, photovoltaic power supply and battery power supply, inductive power supply technology has the advantages of high reliability, good economy and wide application scenarios. However, the inductive power supply based on overhead transmission lines is not suitable for most online monitoring equipment. As a relatively ideal online power supply method, the research on optical power ground wire (OPGW) of overhead transmission lines has gradually increased in recent years.

[0004] The key technical difficulty of ground wire inductive energy harvesting is how to obtain sufficient power. However, in ground wire inductive energy harvesting, the air gap width of the open-closed structure of the ground wire energy harvesting device is uncertain during installation, resulting in fluctuations in the parameters of the current transformer (CT). After compensation by fixed capacitive elements, the device usually operates in a non-resonant state, and the picked-up power is reduced. Summary of the invention

[0005] In view of the above-mentioned deficiencies of the prior art, the technical problem to be solved by the present invention is: how to provide a ground wire energy harvesting device and energy harvesting method that can compensate for the reactive power generated by the detuning of the device due to the uncertain air gap width of the device's magnetic core, so that the energy harvesting device is in a resonant state, which is beneficial to improving the energy harvesting power.

[0006] In order to solve the above technical problems, the present invention adopts the following technical solutions:

[0007] A ground wire energy acquisition device based on virtual impedance, characterized in that it includes a primary side energy acquisition module for installation on the ground wire of a power transmission line, a secondary side energy acquisition module mutually inductively arranged with the primary side energy acquisition module, and the output end of the secondary side energy acquisition module is connected to a voltage type PWM rectifier; the voltage type PWM rectifier includes power switch tubes Q1~Q4, power switch tube body diodes D1~D4, an inductor L and a filter capacitor C f .

[0008] Furthermore, the secondary side energy acquisition module includes at least two current transformers arranged in series.

[0009] Furthermore, the AC side voltage u of the voltage-type PWM rectifier ab The direct axis and quadrature axis components in the dq coordinate system satisfy:

[0010]

[0011] In the formula, u ind 、u inq are the direct axis and quadrature axis components of the AC side voltage in the dq coordinate system respectively; i ind 、i inq are the direct-axis and quadrature-axis components of the AC current in the dq coordinate system, L is the inductance of the PWM rectifier, Ts is the switching period, i ind *、i inq *Reference values ​​of rectifier active current and reactive current.

[0012] Furthermore, the active current inner loop reference value i ind * satisfy:

[0013]

[0014] Where I1 is the effective value of the ground wire current, and N is the number of turns on the secondary side of the CT.

[0015] A method for obtaining energy from a ground wire energy-taking device based on virtual impedance, characterized in that the ground wire energy-taking device based on virtual impedance as described above is first obtained, a primary side energy-taking module is installed on the ground wire of the transmission line, and before obtaining energy, a current source i1 is collected and converted to obtain an active current inner loop reference value i ind * , collect DC output voltage u o After phase shift control, the reactive current inner loop reference value i is obtained inq * , modulate the voltage-type PWM rectifier to convert the DC output voltage u o The reactive current inner loop reference value i under the maximum value state inq *As a correction value; when taking energy, the correction value is used to modulate the voltage-type PWM rectifier.

[0016] Furthermore, the active current inner loop reference value i ind * satisfy:

[0017]

[0018] Where I1 is the effective value of the ground wire current, and N is the number of turns on the secondary side of the CT.

[0019] Compared with the prior art, the present invention has the following advantages:

[0020] 1. Aiming at the problem that the air gap width of the open-and-closed structure of the ground wire energy harvesting device is uncertain during installation, resulting in CT parameter fluctuation, incomplete compensation of fixed-value capacitive components and reduced output power, the energy harvesting device can achieve maximum power output under resonant state.

[0021] 2. The use of multiple current transformers connected in series can effectively solve the problem of insufficient output power of a single current transformer.

[0022] 3. After the ground wire energy harvesting device is installed, only one correction is required to keep the subsequent energy harvesting process in a resonant state. BRIEF DESCRIPTION OF THE DRAWINGS

[0023] Figure 1 This is the equivalent circuit diagram of the energy harvesting device.

[0024] Figure 2 To simplify the equivalent circuit diagram.

[0025] Figure 3 This is the circuit diagram of the PWM rectifier.

[0026] Figure 4 It is the vector relationship diagram of the AC side of the PWM rectifier.

[0027] Figures 5 to 10 This is the loop of the current in different switching modes in the unipolar modulation method.

[0028] Fig.11 Schematic diagram of coordinate transformation.

[0029] Fig.12 It is the control strategy of the energy acquisition device.

[0030] Fig.13 This is the dq current decoupling control block diagram.

[0031] Fig.14 This is the simulation waveform diagram of the energy harvesting device in the resonant state.

[0032] Fig.15This is a simulation waveform diagram of the energy harvesting device in the over-compensation (capacitive) state.

[0033] Fig.16 This is a simulation waveform diagram of the energy harvesting device in the under-compensated (inductive) state.

[0034] Fig.17 This is the simulation waveform of the energy harvesting device when the source side current changes.

[0035] Fig.18 is the coil inductance under different core air gaps. DETAILED DESCRIPTION

[0036] The present invention is further described in detail below in conjunction with embodiments.

[0037] A ground wire energy harvesting device based on virtual impedance uses a mutual inductance model to convert the energy harvesting module into a circuit model. Figure 1 As shown, the virtual impedance is a voltage-type PWM rectifier, and the processing module and monitoring equipment are equivalent to a resistor R L The front end is the equivalent circuit of the energy extraction module, u1 and i1 are the ground wire induced potential and induced current, R1 is the ground wire equivalent resistance, L1 and L 21 M1 and M2 are the self-inductance of the primary and secondary coils of the energy transformer and the mutual inductance of the primary and secondary coils, respectively. C1 is the secondary side compensation capacitor. 21 ~i 25 is the current flowing through the secondary coil of the energy extraction circuit, i 21 ′~i 25 ′ is the current flowing through the parallel compensation capacitor of the secondary coil of the energy extraction circuit, u in 、i in is the voltage and current of VSR port, L is the inductance of PWM rectifier, Q1~Q4 are power switch tubes, D1~D4 are the body diodes of power switch tubes, C f is the filter capacitor, R L It is the equivalent resistance of other processing modules and monitoring equipment.

[0038] For the energy taking device, two CTs are connected in series for energy taking. The mutual inductor can be regarded as a circular core transformer, and the ground wire of the overhead transmission line is the primary coil. Since the leakage reactance and internal resistance of the capacitor of a single CT are much smaller than the excitation impedance and load impedance, they can be ignored in the analysis. Set R m ′, L m ′ is the magnetic loss resistance and excitation inductance of a single CT converted to the secondary side. Assuming that the parameters of the two CTs are completely consistent, the equivalent circuit diagram of the energy harvesting device converted to the secondary side of the CT is as follows: Figure 2 shown.

[0039] The primary input current is affected by the grid load and has nothing to do with the energy extraction device. Therefore, it can be regarded as a constant current source. i1′ is the current value converted from the primary current to the secondary side. R e ′, L e ′ are the converted values ​​of equivalent magnetic loss resistance and equivalent excitation inductance, C e is the secondary side equivalent compensation capacitor, C0, L0, R L is the equivalent capacitance, inductance and resistance of the VSR module. p 、i q They are respectively the active current and reactive current flowing into the VSR. Controlling the two can change the VSR equivalent impedance. q ≠0, the equivalent impedance is capacitive or inductive (the switch is thrown at contacts a and c); adjust i q =0, the equivalent impedance is resistive (the switch is thrown at contact b). The parameters have the following relationship:

[0040]

[0041] Where N is the number of turns on the secondary side of the CT.

[0042] From the equivalent circuit, we can see that when the equivalent compensation capacitor C e and equivalent excitation inductance L e 'When parallel resonance occurs, the equivalent impedance Z f maximum.

[0043]

[0044] From the above formula, we can see that when the circuit is in the resonant state, the equivalent compensation capacitor C e The values ​​are:

[0045]

[0046] So the value of compensation capacitor C1 is:

[0047]

[0048] In practical engineering applications, the uncertainty of the core air gap width leads to the excitation inductance L m ' Fluctuation, that is, Figure 2 The equivalent excitation inductance L e ' is uncertain, through the equivalent capacitance C e After compensation, the energy-taking device is usually capacitive or inductive to the outside, and the VSR must be controlled to work in an inductive or capacitive state to correct the state of the energy-taking device. If the energy-taking device is resistive to the outside, the VSR can work in a resistive state.

[0049] Therefore, when the compensation capacitor C e , Excitation inductance L eWhen the inductance L0 or capacitance C0 equivalent to VSR is in resonance, the circuit will present a pure resistive characteristic. Since the ground wire current is not affected by the energy extraction device, the circuit can be equivalent to the relationship between the current source with internal resistance and the external load. According to the maximum power transfer law, when the equivalent load R is adjusted L and magnetic loss resistance R e ′ are equal, the maximum output active power P can be obtained max :

[0050]

[0051] Further decomposition to obtain:

[0052]

[0053] At the same time, it can be seen from the above formula that the output power of using two CTs in series to obtain energy is twice that of a single CT. That is, using multiple CTs in series to obtain energy can effectively solve the problem of insufficient output power of a single CT.

[0054] To illustrate that the change in coil inductance of the energy harvesting module is caused by the core air gap, under the same test environment, with the same core material and coil turns, the core air gap size is slightly changed. The inductance values ​​measured by the RLC bridge when the core air gap size is slightly changed are 23.1427mH, 32.5787mH and 28.5623mH respectively. Fig.18 shown.

[0055] Obviously, due to the difference in the air gap of the magnetic core, the inductance value changes accordingly. Combined with formula (4), it can be further explained that the constant compensation capacitor cannot completely compensate for the reactive power generated by the excitation inductance in actual engineering applications.

[0056] To facilitate analysis, the operating principle and process of PWM rectifier are further explained. Figure 3 This is the PWM rectifier circuit diagram. As can be seen from the figure, PWM consists of three parts: the front-stage AC side circuit, the bridge circuit composed of the middle MOSFET, and the rear-stage DC side circuit.

[0057] Figure 3 in, u in 、i in are the voltage and current of the AC power supply, u o 、i o are the DC side voltage and current respectively.

[0058] When the loss is ignored, the power balance relationship between the AC and DC sides is:

[0059] u in i in =u oi o (7)

[0060] It can be seen from the above formula (7) that the PWM rectifier can control the output of the DC side through the AC side, and at the same time, it can also control the voltage and current output of the AC side through the DC side in the opposite direction.

[0061] The following analyzes its four-quadrant operation principle. Figure 4 This is the vector relationship diagram of the AC side of the PWM rectifier. By controlling the AC side voltage, it can be controlled to operate in the four-quadrant range.

[0062] in, is the AC power supply voltage, is the AC power supply current, is the modulation voltage on the AC side, is the inductor voltage on the AC side.

[0063] Due to the existence of the inductor L on the AC side, it has a filtering effect, ignoring the high-frequency harmonic components on the AC side, only considering the fundamental component, and ignoring the effect of the resistance in the line. Assume that the electromotive force of the power supply under test on the AC side is unchanged, you can control The amplitude and phase of

[0064] The phase relationship between them.

[0065] like Figure 4 As shown in (a), the voltage vector When the end point runs to point A, the AC power supply voltage leads the power supply current by 90°, and it presents a pure inductive characteristic; Figure 4 As shown in (b), the voltage vector When the end point runs to point B, the AC side power supply voltage and power supply current are in phase, and it presents a forward pure resistive characteristic; Figure 4 As shown in (c), the voltage vector When the endpoint runs to point C, the AC power supply voltage lags behind the power supply current by 90°, and at this time it presents a pure capacitive characteristic; Figure 4 As shown in (d), the voltage vector When the endpoint runs to point D, the AC side power supply voltage and power supply current are in opposite phases, and at this time it presents a negative pure resistive characteristic.

[0066] The basic rules of operation of the four quadrants are:

[0067] (1) When When operating in the AB section, energy flows in the forward direction, and the system absorbs active power and inductive reactive power from the AC side power supply and sends it to the DC side, which is in the rectification state. When the endpoint is at point B, the AC side power supply voltage and power supply current are in phase, and the system is in a unity power factor rectification state.

[0068] (2) When When the BC section is in operation, the energy flows forward, and the system absorbs active power and capacitive reactive power from the AC side power supply and sends it to the DC side, which is in the rectification state. When the endpoint is at point C, the AC side power supply voltage lags the power supply current by 90°, and the system only absorbs capacitive reactive power from the AC side power supply.

[0069] (3) When When the CD segment is running, the energy flows in the reverse direction, and the system transmits active and capacitive reactive power from the DC side to the AC side, and is in an active inverter state. When the endpoint is at point D, the AC side power supply voltage and power supply current are in opposite phases, and the system is in a unity factor active inverter state.

[0070] (4) When When the DA section is running, the energy flows in the reverse direction, and the system transmits active and inductive reactive power from the DC side to the AC side, and is in an active inverter state. When the endpoint is at point A, the AC side power supply voltage leads the power supply current by 90°, and the system only absorbs inductive reactive power from the AC side power supply.

[0071] The unipolar modulation method and the bipolar modulation method are the most commonly used modulation methods for PWM rectifier control. The following will explain the working mode of the PWM rectifier circuit when the unipolar modulation method is used. When the unipolar modulation method is used, the AC side voltage u ab in u o , 0, -u o , 0 switch back and forth, so the PWM rectifier circuit has four switching modes, see Table 1 for details. Among them, mode 3 and mode 4 are called "zero" mode. Figure 5 It is the current loop of PWM rectifier in different modes.

[0072] Table 1 Unipolar PWM modulation working mode

[0073]

[0074] The operating mode of the PWM rectifier (virtual impedance) is described below.

[0075] Working mode 1: Figure 5 As shown, the circuit operates in mode 2, and the AC current i in In the positive half cycle, the current passes through the inductor L, the power switch tube Q2, and the capacitor C in turn. f , power switch tube Q3, back to the power supply u in The current iin Continue to increase, the inductor voltage is greater than 0, its energy storage increases, and the capacitor C f And power supply in To charge the inductor L, the capacitor C f For R L powered by;

[0076] Working mode 2: Figure 6 As shown, the circuit operates in mode 4, and the AC current i in In the positive half cycle, the current passes through the inductor L, the power switch tube Q2, the body diode D4, and returns to the power supply u in The current i in Continue to increase, the inductor voltage is greater than 0, its energy storage increases, the power supply u in To charge the inductor L, the capacitor C f For R L powered by;

[0077] Working mode 3: Figure 7 As shown, the circuit operates in mode 1, and the AC current i in In the positive half cycle, the current passes through the inductor L, the body diode D1, and the capacitor C in turn. f , body diode D4, back to the power supply u in The current i in decreases, the inductor voltage is less than 0, its energy storage decreases, and the power supply u in and the inductor L together to the capacitor C f Charge and give R L powered by;

[0078] Working mode 4: Figure 8 As shown, the circuit operates in mode 1, and the AC current i in In the negative half cycle, the current passes through the power supply u in , power switch tube Q4, capacitor C f , power switch tube Q1, inductor L. In this stage, the current i in Increase, the inductor voltage is less than 0, its energy storage increases, and the capacitor C f And power supply in To charge the inductor L, the capacitor C f For R L powered by;

[0079] Working mode 5: Fig. 9 As shown, the circuit works in mode 3, and the AC current i in In the negative half cycle, the current passes through the power supply u in , body diode D3, power switch tube Q1, inductor L. In this stage, the current i in Continue to increase, the inductor voltage is less than 0, its energy storage increases, the power supply u inTo charge the inductor L, the capacitor C f For R L powered by;

[0080] Working mode 6: Fig.10 As shown, the circuit operates in mode 2, and the AC current i in In the negative half cycle, the current flows through the power supply u in , body diode D3, capacitor C f , body diode D2, through inductor L. In this stage, the current i in decreases, the inductor voltage is greater than 0, its energy storage decreases, and the power supply u in and the inductor L together to the capacitor C f Charge and give R L powered by.

[0081] PWM rectifier mathematical model: Figure 3 In the figure, ignoring the parasitic internal resistance of the inductor, Kirchhoff's voltage law and Kirchhoff's current law are applied to the AC side and DC side circuits respectively to obtain:

[0082]

[0083] For single-phase PWM rectifiers and other AC / DC hybrid systems, in order to simplify the controller design, the AC quantity in the stationary coordinate system is converted into the DC quantity in the dq coordinate system by introducing coordinate transformation. The coordinate transformation requires two orthogonal AC components. Obviously, the single-phase system does not meet the above requirements. Since the coordinate transformation is only a mathematical transformation and has no practical significance, the dq coordinate transformation can be realized by artificially constructing an AC quantity with a phase difference of 90° with the single-phase system. The schematic diagram is shown in the figure. Fig.11 shown.

[0084] In fact, as long as the actual AC quantity is delayed by 90° in a single-phase system, an orthogonal component can be virtualized. On this basis, the orthogonal components u can be virtualized by delaying them by 90°. inm ,i inm , then:

[0085]

[0086] Among them, u ind 、u inq are the direct axis and quadrature axis components of the AC side voltage in the dq coordinate system respectively; i ind 、i inq are the direct axis and quadrature axis components of the AC side current in the dq coordinate system; u abd 、u abq are the direct axis and quadrature axis components of the rectifier bridge AC side dq coordinate system respectively; u ab 、u abmThey are the AC side voltage of the rectifier bridge and its virtual orthogonal quantity. The coordinate transformation matrix T is:

[0087]

[0088] Substituting equations (10), (11) and (13) into equation (8), we can get the mathematical model of the single-phase PWM rectifier in the dq coordinate system. Simplified, we can get the grid-side voltage u in Voltage component u in the two-phase rotating dq coordinate system ind 、u inq They are:

[0089]

[0090] Then the rectifier input voltage u ab The voltage component u in the dq coordinate system abd ,u abq They are:

[0091]

[0092] Control strategy (outer loop control): In dq coordinates, the rectifier active power and reactive power can be expressed as:

[0093]

[0094] Combining formula (16), it can be seen that by adjusting the port voltage u in With the port current i in These two physical quantities can be used to adjust the active power and reactive power of the VSR. Since the source of the CT is a current source, the voltage at its port is affected by the load characteristics, that is, by adjusting i ind 、i inq That is, u can be indirectly controlled ind The size of ind Indirectly set to a fixed value, you can adjust i inq To indirectly control u in , thereby achieving the regulation of active power and reactive power (i.e. VSR state).

[0095] Therefore, this paper adopts the outer loop control of indirectly setting the d-axis current and changing the q-axis current through phase shift control to correct the state of the energy harvesting device so that it is in resonance and outputs maximum power. When the energy harvesting device is in overcompensation (capacitive), the VSR is characterized as inductive to correct the state of the energy harvesting device to resonance; when the energy harvesting device is in undercompensation (inductive), the VSR module is characterized as capacitive to correct the state of the energy harvesting device to resonance. Fig.12 The figure shows the outer loop control strategy of the energy taking device.

[0096] According to the conversion relationship between α-β and dq, iind 、i inq with i p 、i q The following relationship exists:

[0097]

[0098] To ensure that the energy-taking device can take maximum energy when the ground current is I1, the active current i can be determined based on I1 and N. ind * The size is:

[0099]

[0100] Inner loop control: From the above analysis, we can know that control i ind ,i inq The load active power and reactive power can be controlled. The inner loop current control method will be described below. According to the predictive current control principle, let:

[0101]

[0102] Where Ts is the switching period, i in * is the input current reference value.

[0103] Therefore, formula (15) can be further simplified as:

[0104]

[0105] According to formula (20), the dq current decoupling control block diagram can be obtained as follows: Fig.13 shown.

[0106] i ind *、i inq * is the reference value of the rectifier active current and reactive current, combined with formula 18 and Fig.12 The set value can be determined.

[0107] Common modulation methods for single-phase PWM rectifiers include unipolar modulation and bipolar modulation. The equivalent switching frequency of unipolar PWM modulation is twice that of bipolar PWM modulation. The sine control effect controlled by unipolar PWM modulation strategy is better and there is no need to consider the dead zone problem of zero crossing. While reducing switching losses, the THD of the AC side voltage of the single-phase rectifier bridge can be made smaller, thereby making the fluctuation of the input current smaller. Therefore, this embodiment adopts unipolar PWM modulation.

[0108] Circuit simulation analysis: A circuit simulation model is built based on Matlab / Simulink software to verify the correctness of the above theory. The model mainly simulates and analyzes the following two points: 1) When the energy-taking device does not reach the resonant state, the VSR adjusts its own load properties to correct the device state, so that the system is in resonance and the output power of the device is increased; 2) When the VSR module adjusts its own load properties to correct the system to be in a resonant state, the system state is not affected by the current fluctuation of the OPGW on the source side, and the device only needs to be corrected once. The simulation parameters are shown in Table 2.

[0109] Table 2 Simulation parameters of energy extraction device

[0110]

[0111] In order to verify that VSR can change its own properties according to the actual state of the energy harvesting device to correct the system to reach a resonant state, a simulation analysis is performed.

[0112] Fig.14 The following is a simulation waveform of the energy harvesting device in the resonant state (the CT primary voltage is amplified 5 times). In the simulation, the VSR virtual impedance control is set to start at 1s. Fig.14 (a) It can be seen that the CT primary side voltage and current are in phase 1s ago, indicating that the reactive power generated by the compensation capacitor just compensates for the reactive power of the excitation inductance, that is, the energy harvesting device works in a resonant state. Fig.14 (b) It can be seen that after 1s, the port voltage and current of VSR are in phase and work in a resistive state.

[0113] Fig.15 The following is a simulation waveform of the energy-taking device in the over-compensation (capacitive) state (the CT primary voltage is amplified 5 times). In the simulation, the VSR virtual impedance control is set to start at 1s. Fig.15 (a) It can be seen that the voltage phase of the primary side of the CT lags behind the current phase 1s ago, and the device is in a capacitive state, indicating that the reactive power generated by the compensation capacitor is greater than the reactive power generated by the excitation inductance. Fig.15 (b) It can be seen that after 1s, the voltage phase at the VSR port leads the current phase, and the VSR is in an inductive state. At this time, the voltage and current on the primary side of the CT are in phase, and the system is corrected to a resonant state. In summary, when the energy harvesting device is in a capacitive state, the VSR can work in an inductive state to trim the system to a resonant state and increase the energy harvesting power of the device.

[0114] Fig.16 The following is a simulation waveform of the energy-taking device under the under-compensation (inductive) state (the CT primary voltage is amplified 5 times). In the simulation, the VSR virtual impedance control is set to start at 1s. Fig.16(a) It can be seen that the voltage phase of the primary side of the CT is ahead of the current phase 1s ago, and the device is in an inductive state, which means that the reactive power generated by the compensation capacitor is less than the reactive power generated by the excitation inductor. Fig.16 (b) It can be seen that after 1s, the voltage phase of the VSR port lags behind the current phase, and the VSR is in a capacitive state. At this time, the voltage and current on the primary side of the CT are in phase, and the system is corrected to a resonant state. In summary, when the energy harvesting device is in an inductive state, the VSR can work in a capacitive state to trim the system to a resonant state while increasing the energy harvesting power of the device.

[0115] In order to verify that after the VSR corrects the energy harvesting device to the resonant state, the device only needs to be corrected once and is not affected by the current fluctuation on the source side, the current change condition on the source side is simulated.

[0116] Fig.17 The following is a simulation waveform of the source side current when the energy device is under-compensated (inductive) and the VSR is corrected in a steady state (the CT primary side voltage is magnified 5 times). In the simulation, the source side current suddenly changes from 10A to 7.5A at 2s. Fig.17 (a) and 17(b) show that at 2s, the source-side current changes suddenly, the primary-side voltage and current of the energy-taking device CT remain in phase, and the voltage phase at the VSR port lags behind the current phase, indicating that the VSR provides capacitive reactive power for the device and the system is in a resonant state.

[0117] The ground wire energy harvesting device based on virtual impedance control of this embodiment has the characteristics of simple circuit structure, simple method, flexible control and high power supply reliability. The energy harvesting device of this embodiment can well enable the energy harvesting device to achieve maximum power output in a resonant state, overcoming the limitations of traditional solutions.

[0118] The above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions and improvements made within the spirit and principles of the present invention should be included in the protection scope of the present invention.

Claims

1. A ground wire energy extraction device based on virtual impedance, characterized in that: The invention comprises a primary side energy acquisition module for installation on the ground wire of the power transmission line, a secondary side energy acquisition module mutually inductively arranged with the primary side energy acquisition module, and the output end of the secondary side energy acquisition module is connected with a voltage type PWM rectifier; the voltage type PWM rectifier comprises power switch tubes Q1-Q4, power switch tube body diodes D1-D4, an inductor L and a filter capacitor C f .

2. The ground wire energy harvesting device based on virtual impedance according to claim 1, characterized in that: The secondary side energy acquisition module includes at least two current transformers arranged in series.

3. The ground wire energy harvesting device based on virtual impedance according to claim 1 or 2, characterized in that: The AC side voltage u of the voltage type PWM rectifier ab The direct axis and quadrature axis components in the dq coordinate system satisfy: In the formula, u ind 、u inq are the direct axis and quadrature axis components of the AC side voltage in the dq coordinate system respectively; i ind 、i inq are the direct-axis and quadrature-axis components of the AC current in the dq coordinate system, L is the inductance of the PWM rectifier, Ts is the switching period, i ind *、i inq *Reference values ​​of rectifier active current and reactive current.

4. The ground wire energy harvesting device based on virtual impedance according to claim 3, characterized in that: The active current inner loop reference value i ind * satisfy: Where I1 is the effective value of the ground wire current, and N is the number of turns on the secondary side of the CT.

5. A method for obtaining energy from a ground wire energy obtaining device based on virtual impedance, characterized in that: First, obtain the ground wire energy harvesting device based on virtual impedance as described in claim 1 or 2, install the primary side energy harvesting module on the ground wire of the transmission line, and before harvesting energy, collect the current source i1 for conversion processing to obtain the active current inner loop reference value i ind * , collect DC output voltage u o After phase shift control, the reactive current inner loop reference value i is obtained inq * , modulate the voltage-type PWM rectifier to convert the DC output voltage u o The reactive current inner loop reference value i under the maximum value state inq * As a correction value; when taking energy, the correction value is used to modulate the voltage-type PWM rectifier.

6. The energy harvesting method of the ground wire energy harvesting device based on virtual impedance according to claim 5, characterized in that: The active current inner loop reference value i ind * satisfy: Where I1 is the effective value of the ground wire current, and N is the number of turns on the secondary side of the CT.

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