Optical phase finite-difference based radio frequency frequency measurement system and method

Abstract

This invention relates to a radio frequency (RF) frequency measurement system and method based on optical phase finite difference, belonging to the field of microwave photonics measurement. It includes a laser source, an optical single-sideband generator, a fixed optical delay unit, an optical angle diversity receiver, an analog-to-digital converter (ADC) array, a digital signal processing module, and a real-time laser frequency monitoring module. This method modulates a rapidly changing RF signal onto an optical carrier and achieves optical single-sideband modulation, converting the RF frequency change into a laser frequency change. After optical power beam splitting and a fixed time delay, self-coherent detection is performed in the optical angle diversity receiver. The baseband I / Q signals are obtained in the digital signal processing module via the ADC array. The I / Q signal phase is extracted in real time. Using the time-domain laser phase finite difference method, it is proven that the I / Q signal phase is the difference between the instantaneous phase of the laser, i.e., an estimate of the instantaneous laser frequency. This allows the phase flow to be converted into the trajectory of the instantaneous RF frequency change.

Description

Technical Field

[0001] This invention belongs to the field of microwave photonics measurement technology, and relates to a radio frequency measurement system and method based on optical phase finite difference. Background Technology

[0002] Instantaneous frequency measurement (IFM) technology plays a crucial role in radar countermeasures, spectrum monitoring, and microwave photonic frequency shift keying (FSK) communication systems. With the widespread application of modern complex modulation signals (such as linear, nonlinear frequency modulation, and frequency hopping), high-performance requirements for IFM systems are emerging, demanding multi-dimensional coordination: high precision (error better than MHz), high speed (response speed down to sub-nanosecond levels), wide range (covering tens of GHz frequency bands), and dynamic continuous tracking capability for multiple modulation formats. Traditional RF, especially millimeter-wave and even higher frequency IFM, is limited by the bandwidth and loss bottlenecks of electronic devices. While existing microwave photonic IFM schemes have made progress in low-speed, wide-range frequency measurement by utilizing the high bandwidth and low loss characteristics of photonic methods, they still face significant challenges in achieving the aforementioned comprehensive performance.

[0003] While current mainstream schemes based on optical filter frequency-power nonlinear mapping can support wideband measurements, their core mechanisms have inherent limitations: nonlinear response leads to uneven measurement sensitivity within the frequency band and complex calibration; the power detection mechanism is susceptible to link noise; more importantly, the optical filter-based method itself suffers from a conflict between measurement range and measurement accuracy. Alternative schemes based on frequency-time / space mapping suffer from either limited corresponding speeds or complex system structures and high costs. Theoretically more promising frequency-phase linear mapping paths, while possessing inherent linear advantages and continuous tracking potential, are limited by core bottlenecks such as high-precision phase extraction and wideband phase unwrapping. None of the above methods are strictly instantaneous frequency measurements, making it difficult for existing technologies to achieve breakthroughs in high-precision, high-speed, and wide-range real-time dynamic frequency tracking.

[0004] Therefore, there is an urgent need to develop a new IFM method that can break through the limitations of the existing non-instantaneous mapping mechanism based on optical filters, fundamentally change the frequency-intensity mapping mechanism, solve the problem of instantaneous frequency acquisition and processing, and thus achieve high-precision, high-speed, wide-range, multi-mode continuous tracking in wideband, high-dynamic scenarios. Summary of the Invention

[0005] In view of this, the purpose of the present invention is to provide a radio frequency measurement system and method based on optical phase finite difference, which realizes broadband, high speed, high precision, high linearity and universality of instantaneous frequency measurement.

[0006] To achieve the above objectives, the present invention provides the following technical solution: On one hand, the present invention provides a radio frequency measurement system based on optical phase finite difference, including a continuous laser source, an optical single-sideband generator, an optical power beam splitter, a fixed optical delay unit, an optical angle diversity receiver, a digital signal processing module, and a laser frequency real-time monitoring module. The continuous laser source is used to emit a laser carrier wave; The optical single-sideband generator is used to modulate the radio frequency signal under test onto the laser carrier and realize optical single-sideband modulation, converting the change of radio frequency frequency into the frequency change of continuous laser source. The optical power beam splitter is used to split the modulated optical signal into two branches, with the lower branch connected to the fixed optical delay unit for delaying the optical signal in the lower branch. The optical angle diversity receiver is used to receive the optical signals from the upper and lower branches, perform self-coherent detection, generate in-phase component I-channel electrical signals and quadrature component Q-channel electrical signals, and then perform analog-to-digital conversion on the electrical signals to the digital domain. The digital signal processing module is used to perform digital signal processing on the signal converted to the digital domain and calculate the instantaneous frequency of the radio frequency signal under test. The laser frequency real-time monitoring module is used to monitor the frequency drift of the laser, and its output is connected to the digital signal processing module for compensation and correction.

[0007] Furthermore, the optical single-sideband generator includes a laser input channel, a first radio frequency (RF) signal input channel, a second RF signal input channel, an optical IQ modulator, and an optical sideband signal output terminal. The laser input channel is used to connect to a continuous laser source to provide an optical carrier. The first RF signal input channel is equipped with an electrical amplifier to amplify the RF signal under test and input it to the I port of the optical IQ modulator. The second RF signal input channel is equipped with an electrical amplifier to amplify the RF signal under test after a 90-degree phase shift and input it to the Q port of the optical IQ modulator. The optical IQ modulator operates in a carrier-suppressed single-sideband modulation mode to modulate two orthogonal RF signals onto the optical carrier, outputting a single-sideband signal with only one sideband retained, the center carrier, and the other sideband suppressed. The optical sideband signal output terminal is used to output the optical single-sideband signal to an optical power beam splitter.

[0008] Furthermore, the optical single-sideband generator includes a laser input channel, an RF signal input channel, an optical MZ modulator, an optical bandpass filter, and an optical sideband signal output terminal. The laser input channel is used to connect a continuous laser source to provide an optical carrier. The RF signal input channel is equipped with an electrical amplifier to amplify the RF signal under test and input it to the optical MZ modulator. The optical MZ modulator operates in a conventional double-sideband modulation mode to modulate the RF signal onto the optical carrier, outputting an optical signal containing the optical carrier and its two sidebands. The optical bandpass filter is used to spectrally shape the modulated optical signal, filtering out one sideband and the optical carrier component, retaining only the target single-sideband. The optical sideband signal output terminal is used to output the optical single-sideband signal to an optical power beam splitter.

[0009] Furthermore, the optical angle diversity receiver adopts a 0-degree and 180-degree diversity structure, including a photodetector and an analog-to-digital converter; the photodetector is used to receive optical signals from the 0-degree and 180-degree phase diversity channels, and outputs analog electrical signals through the photodetector; the analog-to-digital converter is used to convert the analog electrical signals output by the photodetector into digital signals, and the digital signals are orthogonalized in the digital domain after Hilbert transformation to recover the complete in-phase and quadrature component information.

[0010] Furthermore, the optical angle diversity receiver adopts a four-phase diversity structure of 0 degrees, 90 degrees, 180 degrees, and 270 degrees, including a photodetector array and an analog-to-digital converter array. The photodetector array contains four photodetectors, which respectively receive optical signals of 0 degrees, 180 degrees, 90 degrees, and 270 degrees of phase diversity, and output four analog electrical signals after photoelectric conversion. Among them, the two analog electrical signals corresponding to 0 degrees and 180 degrees are subtracted by a subtractor to obtain the in-phase component signal with the DC component eliminated; the two analog electrical signals corresponding to 90 degrees and 270 degrees are subtracted by a subtractor to obtain the quadrature component signal with the DC component eliminated. The analog-to-digital converter array contains two analog-to-digital converters, which are used to synchronously convert the above-mentioned in-phase component signal and quadrature component signal into two digital signals. These two digital signals directly constitute complete in-phase component and quadrature component information without the need for subsequent digital orthogonalization processing.

[0011] Furthermore, the optical angle diversity receiver adopts a three-phase diversity structure of 0 degrees, 120 degrees, and 240 degrees, including three photodetectors and three analog-to-digital converters; the first photodetector is used to receive the optical signal of 0-degree phase diversity and output one analog electrical signal; the second photodetector is used to receive the optical signal of 120-degree phase diversity and output one analog electrical signal; the third photodetector is used to receive the optical signal of 240-degree phase diversity and output one analog electrical signal; the three analog-to-digital converters are used to synchronously convert the three analog electrical signals into three digital signals.

[0012] Furthermore, the digital signal processing process of the digital signal processing module includes optical angle diversity receiver IQ imbalance compensation, complex domain signal reconstruction, digital filtering, phase calculation, phase unwrapping, and finally, based on the linear mapping relationship between the I / Q signal phase, the laser instantaneous frequency, and the radio frequency instantaneous frequency, the phase stream is converted into the trajectory of the instantaneous radio frequency change, and the instantaneous frequency is calculated.

[0013] Furthermore, the laser frequency real-time monitoring module adopts a multi-phase diversity structure, including a balanced detector and an analog-to-digital converter; the balanced detector is used to receive the optical signals of each phase diversity and output multiple analog electrical signals; the analog-to-digital converter is used to convert the analog electrical signals into digital signals, process the digital signals, extract the laser frequency drift, and perform correction.

[0014] On the other hand, the present invention provides a radio frequency measurement method based on optical delay self-coherence and phase difference, comprising the following steps: S1: The radio frequency signal to be tested is modulated onto the laser carrier using an optical single-sideband generator, and optical single-sideband modulation is achieved, converting the change of radio frequency into the frequency change of a continuous laser source. S2: The modulated optical signal is split into two branches using an optical power beam splitter, with the optical signal in the lower branch being processed by an optical delay line. S3: The upper and lower branch optical signals are respectively input to the optical angle diversity receiver to generate the in-phase component I-channel electrical signal and the quadrature component Q-channel electrical signal, and then the electrical signals are converted from analog to digital to the digital domain. S4: Perform digital signal processing on the signal converted to the digital domain, utilizing the linear mapping relationship between the frequency change Δf of the RF signal and the phase change ΔΦ of the demodulated baseband signal. The instantaneous frequency of the radio frequency signal under test is calculated, where the calibration coefficient is... k =1 / (2πτ), where τ is the introduced fixed optical delay.

[0015] Furthermore, in step S4, before calculating the instantaneous frequency of the RF signal under test, the delay of the lower branch is calibrated. The calibration process includes the following steps: S41: Input a calibration radio frequency signal whose frequency varies linearly with time; S42: Acquire and process the two baseband quadrature electrical signals output for the calibration radio frequency signal, perform digital filtering and receiver I / Q amplitude equalization on the signals, and then calculate the continuous instantaneous calibration phase. S43: Compare and perform linear regression analysis on the known frequency change sequence of the calibration radio frequency signal with the calculated continuous instantaneous calibration phase sequence; S44: Through the linear regression analysis, a mapping relationship between a radio frequency signal phase value and a laser frequency value is determined, thereby completing the system calibration; S45: Repeat calibration steps S42-S44 to obtain a series of calibration coefficient samples. Calculate the arithmetic mean of the sample set to determine the final optimized calibration coefficients.

[0016] The beneficial effects of this invention are as follows: (1) Based on the principle of linear time-varying mapping, this invention can measure various complex dynamic signals such as linear chirp, quadratic linear frequency modulation, triangular wave frequency modulation and frequency hopping in real time without distortion, and solves the distortion or failure problem of traditional schemes in dynamic frequency measurement.

[0017] (2) This invention achieves frequency calculation through high-precision phase measurement. Within a 32 GHz instantaneous bandwidth, the accuracy of linear frequency and triangular frequency is better than 3 MHz, the accuracy of sinusoidal frequency is less than 10 MHz, the frequency hopping accuracy is less than 50 MHz, and the response time reaches the nanosecond level, thus achieving a unity of high precision, large bandwidth and fast response.

[0018] (3) The core advantage of this invention is that its key mapping relationship is determined only by a fixed physical time delay τ. System calibration only requires a single injection of a dynamic chirped signal, followed by repeated measurements and statistical averaging of this signal, thereby achieving high-precision calibration quickly while ensuring high reliability. By flexibly adjusting the fiber length to control τ, system performance can be easily configured to adapt to different application scenarios.

[0019] (4) The system is built on mature optical communication devices, with a simple structure, no need for high-speed and complex circuits, inherent advantages in electromagnetic interference resistance, high reliability, and easy integration and application.

[0020] Other advantages, objectives, and features of the invention will be set forth in part in the description which follows, and in part will be apparent to those skilled in the art from the following examination, or may be learned from practice of the invention. The objectives and other advantages of the invention can be realized and obtained through the following description. Attached Figure Description

[0021] To make the objectives, technical solutions, and advantages of the present invention clearer, the preferred embodiments of the present invention will be described in detail below with reference to the accompanying drawings, wherein: Figure 1 This invention presents a system block diagram of instantaneous frequency measurement based on time-delayed self-coherence and phase difference, as well as an optical single-sideband generator, an optical angle diversity receiver, and a laser frequency real-time monitoring module. Figure 2 This is a flowchart of the system calibration process of the present invention; Figure 3 This is a flowchart of the signal processing of the present invention; Figure 4 This is a block diagram of an embodiment of an IQ modulator SSB and a coherent receiver; Figure 5 Block diagram of an embodiment for MZM suppressed carrier double-sideband modulation + receiver processing of upper and lower sidebands respectively; Figure 6 A structural block diagram of an embodiment of microwave photonic FSK transmission; Figure 7 This is a block diagram of an FMCW radar signal ranging implementation example; Figure 8 This is the spectrum after IQ modulation according to the present invention; Figure 9 This is a graph showing the correspondence between the actual frequency and the measured phase after system calibration. Figure 10 The diagram shows a comparison of the actual and measured frequencies of the five signals corresponding to this invention, as well as an error analysis result diagram. Detailed Implementation

[0022] The following specific examples illustrate the implementation of the present invention. Those skilled in the art can easily understand other advantages and effects of the present invention from the content disclosed in this specification. The present invention can also be implemented or applied through other different specific embodiments, and various details in this specification can be modified or changed based on different viewpoints and applications without departing from the spirit of the present invention. It should be noted that the illustrations provided in the following embodiments are only schematic representations of the basic concept of the present invention. Unless otherwise specified, the following embodiments and features can be combined with each other.

[0023] It should be noted that the illustrations provided in the following embodiments are only schematic representations of the basic concept of the present invention. Therefore, the drawings only show the components related to the present invention and are not drawn according to the actual number, shape and size of the components in the actual implementation. In the actual implementation, the form, quantity and proportion of each component can be arbitrarily changed, and the layout of the components may also be more complex.

[0024] In the following description, numerous details are explored to provide a more thorough explanation of embodiments of the invention. However, it will be apparent to those skilled in the art that embodiments of the invention may be practiced without these specific details. In other embodiments, well-known structures and devices are shown in block diagram form rather than in detail to avoid obscuring embodiments of the invention.

[0025] Example 1: This invention provides an instantaneous frequency measurement system and method based on time-delayed self-coherence and phase difference, such as... Figure 1 As shown, it includes an optical single-sideband generator, an optical angle diversity receiver, and a laser frequency real-time monitoring module, wherein: An optical single-sideband generator modulates the radio frequency signal under test onto a continuous laser source to generate an optical sideband signal suitable for subsequent time-delayed self-coherent processing. Two optional implementation schemes are available: The first scheme employs an optical IQ modulator structure, including a laser input, a first RF signal input channel, a second RF signal input channel, an optical IQ modulator, and an optical sideband signal output terminal. Specifically: the laser input is used to connect to a continuous laser source to provide an optical carrier; the first RF signal input channel is equipped with an amplifier to amplify the RF signal under test before inputting it to the I port of the optical IQ modulator; the second RF signal input channel is also equipped with an amplifier to amplify the RF signal under test after a 90-degree phase shift before inputting it to the Q port of the optical IQ modulator; the optical IQ modulator operates in a carrier-suppressed single-sideband modulation mode, used to modulate two orthogonal RF signals onto the optical carrier, outputting a single-sideband signal with only one sideband, the center carrier, and the other sideband suppressed; the optical sideband signal output terminal outputs this optical signal to subsequent modules.

[0026] The second approach employs an optical MZ modulator combined with an optical bandpass filter structure, including a laser input, an RF signal input channel, an optical MZ modulator, an optical bandpass filter, and an optical sideband signal output terminal. Specifically: the laser input is used to connect a continuous laser source to provide an optical carrier; the RF signal input channel is equipped with an amplifier to amplify the RF signal under test before inputting it to the optical MZ modulator; the optical MZ modulator operates in a conventional double-sideband modulation mode to modulate the RF signal onto the optical carrier, outputting an optical signal containing the optical carrier and its two sidebands; the optical bandpass filter performs spectral shaping on the modulated optical signal, filtering out one sideband and the optical carrier component, retaining only the target single-sideband; the optical sideband signal output terminal outputs a clean optical single-sideband signal to subsequent modules.

[0027] An optical angle diversity receiver is used to perform time-delayed self-coherent processing on the optical signal output from an optical single-sideband generator, generating multiple electrical signals with specific phase relationships. Three optional implementation schemes are available: The first scheme employs a 0-degree and 180-degree diversity structure, including a photodetector and an analog-to-digital converter. Specifically, the photodetector receives optical signals from both 0-degree and 180-degree phase diversity paths and outputs an analog electrical signal. The analog-to-digital converter converts the analog electrical signal output from the photodetector into a digital signal. This digital signal undergoes Hilbert transform and is then orthogonalized in the digital domain to recover the complete in-phase and quadrature component information.

[0028] The second scheme employs a four-phase diversity structure with 0°, 90°, 180°, and 270° phase diversity, comprising a photodetector array and an analog-to-digital converter array. The photodetector array contains four photodetectors that receive optical signals from the 0°, 180°, 90°, and 270° phase diversity phases, respectively. After photoelectric conversion, four analog electrical signals are output. The analog electrical signals corresponding to 0° and 180° are subtracted by a subtractor to obtain the in-phase component signal with DC component eliminated. The analog electrical signals corresponding to 90° and 270° are subtracted by a subtractor to obtain the quadrature component signal with DC component eliminated. The analog-to-digital converter array contains two analog-to-digital converters to synchronously convert the in-phase and quadrature component signals into two digital signals. These two digital signals directly constitute complete in-phase and quadrature component information without subsequent digital orthogonalization processing.

[0029] The third scheme employs a three-phase diversity structure with 0°, 120°, and 240° phases, comprising three photodetectors and three analog-to-digital converters. Specifically: the first photodetector receives the optical signal from the 0° phase diversity and outputs one analog electrical signal; the second photodetector receives the optical signal from the 120° phase diversity and outputs one analog electrical signal; the third photodetector receives the optical signal from the 240° phase diversity and outputs one analog electrical signal; and the three analog-to-digital converters synchronously convert the three analog electrical signals into three digital signals.

[0030] The laser frequency real-time monitoring module is used to monitor the laser's frequency drift and perform real-time correction on instantaneous frequency measurement results, including three optional implementation schemes: The first scheme employs a 0-degree and 180-degree diversity structure, including a balanced detector and an analog-to-digital converter (ADC). The balanced detector receives optical signals from the 0-degree and 180-degree phase diversity and outputs an analog electrical signal. The ADC converts the analog electrical signal into a digital signal, processes the digital signal, extracts the laser frequency drift, and corrects it.

[0031] The second scheme employs a four-phase diversity structure with 0°, 90°, 180°, and 270° phase diversity, comprising four photodetectors, two subtractors, and two analog-to-digital converters. The four photodetectors receive optical signals from the 0°, 180°, 90°, and 270° phase diversity phases, respectively, and output four analog electrical signals after photoelectric conversion. The two analog electrical signals corresponding to 0° and 180° are subtracted by the first subtractor to obtain in-phase component signals with DC component eliminated. The other two analog electrical signals corresponding to 90° and 270° are subtracted by the second subtractor to obtain quadrature component signals with DC component eliminated. The two analog-to-digital converters synchronously convert the in-phase and quadrature component signals into two digital signals, respectively. After joint processing of the two digital signals, the laser frequency drift is extracted and corrected.

[0032] The third scheme employs a three-phase diversity structure with 0°, 120°, and 240° phases, comprising three balanced detectors and three analog-to-digital converters (ADCs). The three balanced detectors receive optical signals from 0°, 120°, and 240° phase diversity, respectively, and output three analog electrical signals. The three ADCs synchronously convert the three analog electrical signals into three digital signals, and after joint processing of the three digital signals, the laser frequency drift is extracted and corrected.

[0033] The following explanation, combining the principles of interference and signal models, illustrates the frequency-phase linear mapping relationship established in this invention, to clarify its theoretical correctness and the source of its high-precision implementation. The optical single-sideband generator is illustrated by Scheme 1, and the optical angle diversity receiver is illustrated by Scheme 2.

[0034] The output light field of a laser can be expressed as: (1) In the formula, A0 is the light field intensity, and fc is the center frequency. The phase is generated by the laser frequency jitter. The complex baseband of the RF signal under test generated by AWG is expressed as: (2) in For the RF signal amplitude, its real and imaginary parts correspond to the two driving voltages of the IQ modulator, respectively: (3) (4) Instantaneous phase With instantaneous frequency satisfy: (5) Here That is, the time-varying instantaneous frequency to be measured.

[0035] The single-sideband optical signal obtained after IQ modulation is expressed as: (6) Where η is the modulation efficiency factor. Then, it passes through a 1×2 beam splitter, splitting into a reference branch and a delay branch. The delay branch introduces a fixed delay τ through a delay fiber. The two optical signals can be represented as follows: (7) (8) The two optical signals are then injected into an optical angle diversity receiver, whose internal 90° optical mixer outputs four signals with a fixed phase relationship: (9) The incident light power is then converted into a corresponding photocurrent by a photodetector through the photoelectric effect. The relationship between its output current and the input light field can be expressed as: (10) Where R is the detector responsivity (A / W). Substituting E1(t) into (10), we obtain the current corresponding to E1(t): (11) Similarly, we can obtain The DC term is eliminated by differential detection using balanced photoelectric sensing. Then, the output in-phase (I) branch and quadrature (Q) branch are obtained: (12) (13) The formula , Substituting (12) and (13), we obtain the final output in-phase and quadrature electrical signals: (14) (15) Thus, the expression for the IQ signal is obtained: (16) in j is the complex unit. It has the following relationship with the transmission frequency. (17) The last term represents the initial phase of the IQ signal at the receiving end. Since this paper focuses on dynamic frequencies, the last term will not be discussed further. According to Equation 16, the final frequency of the IQ signal is the rate of change of the RF signal frequency multiplied by the delay time. When we obtain the mapping coefficient between the actual frequency and the IQ phase... k Then, the actual frequency change can be calculated using the IQ phase change, thus achieving frequency tracking. Theoretically... Unit: Hz / rad. In practical systems, a known frequency is input to the system and the corresponding phase change is measured. ΔΦ Obtained by linear fitting k Finally, the instantaneous frequency estimation formula is: (18) in Let be the phase of the IQ signal. This formula establishes a direct mapping from the interference phase to the radio frequency, providing a mathematical basis for real-time frequency tracking.

[0036] Although the system principle establishes the mapping relationship between frequency and phase, the theoretical calculation of its proportionality coefficient k = 1 / (2πτ) depends on the precise value of the fiber delay. In practical systems, the effective refractive index of the fiber, the modulator group delay, and the non-ideals of the receiving channel can cause deviations between the actual delay and the theoretical value, thus introducing systematic measurement errors. Therefore, experimental calibration is necessary to obtain the accurate actual mapping coefficient k. Please refer to... Figure 2 The specific calibration process includes the following steps: S1: Input a calibration radio frequency signal whose frequency changes linearly with time into the system; S2: Acquire and process the two baseband quadrature electrical signals output by the system for the calibration radio frequency signal, perform digital filtering, resampling, I / Q amplitude equalization on the signals, and then calculate the continuous instantaneous calibration phase; S3: Compare and perform linear regression analysis on the known frequency change sequence of the calibration radio frequency signal with the calculated continuous instantaneous calibration phase sequence; S4: Through the linear regression analysis, a proportionality coefficient is determined to map the phase value to the frequency value, thereby completing the system calibration; S5: Repeat calibration steps 2-4 to obtain a series of calibration coefficient samples. Calculate the arithmetic mean of the sample set to determine the final optimized calibration coefficients.

[0037] Example 2: See Figure 3 The signal processing flowchart shown is the core computational architecture of the instantaneous frequency measurement system in one embodiment of the present invention. This flowchart defines a complete digital signal processing link from raw data acquisition to instantaneous frequency output: First, the two baseband quadrature electrical signals (I / Q) output by the coherent receiver are synchronously acquired and digitized; then, an analytical signal is constructed through complex synthesis, and its instantaneous phase is calculated; subsequently, a phase unwrapping algorithm is used to eliminate 2π ambiguity, obtaining a continuous phase trajectory; finally, the instantaneous frequency value of the input RF signal is directly calculated by time differentiation of the unwrapped phase and multiplication by a calibration coefficient (k=1 / (2πτ)). This processing flow realizes fully digital computation from raw signal to frequency information, ensuring the measurement accuracy and real-time performance of the system under high dynamic conditions.

[0038] Please see Figure 4This embodiment provides an instantaneous frequency measurement device based on an IQ modulator and coherent reception, including a laser, an optical IQ modulator, an electrical amplifier, a 1x2 optical power beamsplitter, an optical delay line, a 90° optical mixer, a photodetector module, an analog-to-digital converter, and a digital signal processing module. The optical carrier generated by the laser is injected into the optical IQ modulator. The radio frequency signal under test is amplified and applied to the I and Q paths of the optical IQ modulator. The optical IQ modulator operates in single-sideband mode, performing single-sideband modulation on the radio frequency signal under test and outputting an optical single-sideband signal. The optical single-sideband signal is split into two paths by the 1x2 optical power beamsplitter: one path is directly output as signal light, and the other path is output as local oscillator light after passing through the optical delay line. The two optical signals enter the 90° optical mixer for coherent mixing, generating four phase-orthogonal signals of 0°, 90°, 180°, and 270° through its internal structure, achieving phase diversity reception. The four signals are sent to the digital signal processing module after photoelectric detection and analog-to-digital conversion to calculate the instantaneous frequency of the radio frequency signal under test.

[0039] Please see Figure 5 This embodiment provides an instantaneous frequency measurement device based on suppressed carrier double-sideband modulation and diversity reception, including a laser, an optical suppressed carrier double-sideband generator, a 1x2 optical power beamsplitter, an optical delay line, an optical filter beamsplitter, an optical angle diversity receiver array, an analog-to-digital converter array, and a digital signal processing module. The optical carrier generated by the laser is injected into the optical suppressed carrier double-sideband generator. The radio frequency signal under test drives the optical suppressed carrier double-sideband generator, generating a suppressed carrier double-sideband modulated signal at its output. This signal contains two components: an upper sideband and a lower sideband. The suppressed carrier double-sideband signal is split into two paths by the first 1x2 optical power beamsplitter: one path is directly output, and the other is output after passing through the optical delay line. Both paths enter their respective optical filter beamsplitters. Each optical filter beamsplitter separates the upper and lower sideband components of the input signal and outputs them to the upper sideband optical angle diversity receiver and the lower sideband optical angle diversity receiver, respectively. The upper and lower sideband signals are coherently received by optical angle diversity receivers, generating multiple phase-orthogonal signals. These signals are then synchronously sampled by an analog-to-digital converter array and sent to the digital signal processing module. The digital signal processing module subtracts the received results from the upper and lower sidebands to eliminate laser jitter common-mode noise and calculates the instantaneous frequency of the radio frequency signal under test.

[0040] Please see Figure 6This embodiment provides an FSK signal transmission and demodulation device based on microwave photonics technology, including a laser, an optical IQ modulator, an electrical amplifier, a 1x2 optical power beamsplitter, an optical delay line, a photodetector module, an analog-to-digital converter, a digital signal processing module, and a laser frequency real-time monitoring module. The optical carrier generated by the laser is injected into the optical IQ modulator. The radio frequency FSK signal is amplified by the electrical amplifier and applied to the I and Q paths of the optical IQ modulator. The optical IQ modulator operates in single-sideband mode, performing single-sideband modulation on the radio frequency FSK signal and outputting an optical single-sideband signal. The optical single-sideband signal is split into two paths by the 1x2 optical power beamsplitter: one path is directly output to the photodetector module as signal light, and the other path is output to the photodetector module as local oscillator light after passing through the optical delay line. The two optical signals are coherently received in the photodetector module, which generates four phase-orthogonal signals (0°, 90°, 180°, and 270°) through its internal structure, achieving phase diversity reception. The four signals are synchronously sampled by the analog-to-digital converter and then sent to the digital signal processing module. The digital signal processing module performs FSK demodulation on the received signal to recover the original transmitted data. The laser frequency real-time monitoring module monitors laser frequency drift, and its output is connected to the digital signal processing module for compensation and correction.

[0041] Please see Figure 7 This embodiment provides an FMCW radar ranging device based on microwave photonics technology, including a laser, an optical IQ modulator, an electrical amplifier, a 1x2 optical power beamsplitter, an optical delay line, a photoelectric detection module, an analog-to-digital converter, an FMCW radar signal processing module, and a laser frequency real-time monitoring module. The optical carrier generated by the laser is injected into the optical IQ modulator. The FMCW radar signal is amplified and applied to the I and Q paths of the optical IQ modulator. The optical IQ modulator operates in single-sideband mode, outputting an optical single-sideband signal. The optical single-sideband signal is split into two paths by the 1x2 optical power beamsplitter: one path is directly output as signal light, carrying target echo information; the other path is output as local oscillator light after passing through the optical delay line, serving as a transmission reference signal. The two optical signals are received by phase diversity reception at 0°, 90°, 180°, and 270° and then photoelectrically detected. The output signals are converted from analog to digital and sent to the FMCW radar signal processing module. The FMCW radar signal processing module calculates the difference frequency between the transmitted and echo signals, thereby estimating the target distance. The laser frequency real-time monitoring module is used to monitor laser frequency drift and perform compensation and correction.

[0042] Please see Figure 8 In this embodiment, a tunable external cavity laser with a center wavelength of 1550 nm and an output power of 15.5 dBm is first used as a continuous wave source. Its output optical carrier is fed into the optical input terminal of a dual-polarization IQ modulator. To verify the performance of the system of the present invention, a radio frequency signal with a frequency of 10 GHz is input to the optical modulation unit. Figure 8The optical signal spectrum after modulation by the IQ modulator is shown. Measurements and analysis revealed that the carrier rejection ratio (CSRR) of the modulated signal reached 27.82 dB, and the sideband rejection ratio (SRR) reached 44.06 dB. Based on these measurements, the calculated optical sideband modulation efficiency was 97.54%. This result demonstrates that the optical modulation unit effectively achieves high-performance single-sideband modulation with suppressed carrier, laying the foundation for subsequent high-precision frequency measurements.

[0043] Figure 9 The diagram shows the linear fit results between a set of known RF signal frequencies and the corresponding phase values ​​measured by the system during system calibration. After one linear fit calculation, the obtained frequency-phase calibration coefficient (i.e., the slope of the fitted line) is k = 1.3993 × 10⁻⁶. 7 Hz / rad. To improve the accuracy and reliability of the coefficients, the calibration process was repeated multiple times and data was collected. The final average value of the calibration coefficients was determined to be 1.3992 × 10⁻⁶. 7 Hz / rad, or 13.992 MHz / rad. The linear fitting results (correlation coefficient R² approaches 1) and the high consistency of the coefficients fully demonstrate that the time-delay interferometric structure proposed in this invention can achieve a highly linear and stable mapping relationship between frequency and phase, thereby ensuring the accuracy of instantaneous frequency measurement.

[0044] according to Figure 10 Figures ae (i) and (ii) show the actual and measured frequencies of the system for five typical frequency modulation signals, respectively. The results demonstrate that the system has excellent measurement performance. For linear and quadratic chirped signals, the frequency range was set to 7.5–9.5 GHz, with a total chirp duration of 5.68 μs. The triangular wave frequency-modulated signal adopted a symmetrical linear sweep frequency method: the frequency linearly increased from 7.5 GHz to 9.5 GHz with a duration of T / 2; then linearly decreased from 9.5 GHz to 7.5 GHz with the same duration of T / 2, and the period T = 5.68 μs. The frequency change of the sinusoidal frequency-modulated signal was centered at 8.5 GHz, exhibiting a complete sinusoidal periodic change: that is, increasing from 8.5 GHz to 9.5 GHz, then decreasing to 7.5 GHz, and finally returning to 8.5 GHz. In the frequency hopping test, the carrier frequencies were sequentially set to 8.2 GHz, 11.1 GHz, and 9.5 GHz, with a dwell time of approximately 1.85 μs at each frequency, to evaluate the system's response characteristics to frequency switching. Error statistical analysis showed that within a 32 GHz instantaneous bandwidth, Figure 10Figure (iii) of section ae shows the distribution histogram of the corresponding frequency estimation errors, with root mean square errors of approximately 1.2526 MHz, 1.9244 MHz, 2.0913 MHz, 7.7335 MHz, and 43.3581 MHz, respectively. The system achieves a root mean square error better than 3 MHz for tracking continuous frequency modulated signals (linear, quadratic linear, and triangular waves), a tracking accuracy better than 10 MHz for sinusoidal signals, and a tracking accuracy better than 50 MHz for frequency hopping signals. This verifies that the system possesses high-precision and high-dynamic continuous frequency tracking capability for signals with different modulation formats over a wide bandwidth.

[0045] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention and are not intended to limit it. Although the present invention has been described in detail with reference to preferred embodiments, those skilled in the art should understand that modifications or equivalent substitutions can be made to the technical solutions of the present invention without departing from the spirit and scope of the present invention, and all such modifications or substitutions should be covered within the scope of the claims of the present invention.

Claims

1. An optical phase finite-difference based radio frequency frequency measurement system, characterized by: It includes a continuous laser source, an optical single-sideband generator, an optical power beam splitter, a fixed optical delay unit, an optical angle diversity receiver, a digital signal processing module, and a laser frequency real-time monitoring module. The continuous laser source is used to emit a laser carrier wave; The optical single-sideband generator is used to modulate the radio frequency signal under test onto the laser carrier and realize optical single-sideband modulation, converting the change of radio frequency frequency into the frequency change of continuous laser source. The optical power beam splitter is used to split the modulated optical signal into two branches, with the lower branch connected to the fixed optical delay unit for delaying the optical signal in the lower branch. The optical angle diversity receiver is used to receive the optical signals from the upper and lower branches, perform self-coherent detection, generate in-phase component I-channel electrical signals and quadrature component Q-channel electrical signals, and then perform analog-to-digital conversion on the electrical signals to the digital domain. The digital signal processing module is used to perform digital signal processing on the signal converted to the digital domain and calculate the instantaneous frequency of the radio frequency signal under test. The laser frequency real-time monitoring module is used to monitor the frequency drift of the laser, and its output is connected to the digital signal processing module for compensation and correction.

2. The optical phase finite-difference based radio frequency frequency measurement system of claim 1, wherein: The optical single-sideband generator includes a laser input channel, a first radio frequency (RF) signal input channel, a second RF signal input channel, an optical IQ modulator, and an optical sideband signal output terminal. The laser input channel is used to connect to a continuous laser source to provide an optical carrier. The first RF signal input channel is equipped with an electrical amplifier to amplify the RF signal under test and input it to the I port of the optical IQ modulator. The second RF signal input channel is equipped with an electrical amplifier to amplify the RF signal under test after a 90-degree phase shift and input it to the Q port of the optical IQ modulator. The optical IQ modulator operates in a carrier-suppressed single-sideband modulation mode to modulate two orthogonal RF signals onto the optical carrier, outputting a single-sideband signal with only one sideband retained, the center carrier, and the other sideband suppressed. The optical sideband signal output terminal is used to output the optical single-sideband signal to an optical power beam splitter.

3. The optical phase finite-difference based radio frequency frequency measurement system of claim 1, wherein: The optical single-sideband generator includes a laser input channel, an RF signal input channel, an optical MZ modulator, an optical bandpass filter, and an optical sideband signal output terminal. The laser input channel is used to connect a continuous laser source to provide an optical carrier. The RF signal input channel is equipped with an electrical amplifier to amplify the RF signal under test and input it to the optical MZ modulator. The optical MZ modulator operates in a conventional double-sideband modulation mode to modulate the RF signal onto the optical carrier, outputting an optical signal containing the optical carrier and its two sidebands. The optical bandpass filter is used to spectrally shape the modulated optical signal, filtering out one sideband and the optical carrier component, retaining only the target single-sideband. The optical sideband signal output terminal is used to output the optical single-sideband signal to an optical power beam splitter.

4. The optical phase finite-difference based radio frequency frequency measurement system of claim 1, wherein: The optical angle diversity receiver adopts a 0-degree and 180-degree diversity structure, including a photodetector and an analog-to-digital converter; The photodetector is used to receive optical signals with phase diversity at 0 degrees and 180 degrees, and outputs analog electrical signals through photodetection; the analog-to-digital converter is used to convert the analog electrical signals output by the photodetector into digital signals. After the digital signals are transformed by Hilbert, they are orthogonalized in the digital domain to recover the complete in-phase and quadrature component information.

5. The optical phase finite-difference based radio frequency frequency measurement system of claim 1, wherein: The optical angle diversity receiver adopts a four-phase diversity structure of 0 degrees, 90 degrees, 180 degrees and 270 degrees, including a photodetector array and an analog-to-digital converter array; The photodetector array comprises four photodetectors, which receive optical signals with phase diversity at 0 degrees, 180 degrees, 90 degrees, and 270 degrees, respectively. After photoelectric conversion, four analog electrical signals are output. Among them, the two analog electrical signals corresponding to 0 degrees and 180 degrees are subtracted by a subtractor to obtain the in-phase component signal with the DC component eliminated; the two analog electrical signals corresponding to 90 degrees and 270 degrees are subtracted by a subtractor to obtain the quadrature component signal with the DC component eliminated. The analog-to-digital converter array comprises two analog-to-digital converters, which are used to synchronously convert the above in-phase component signal and quadrature component signal into two digital signals. These two digital signals directly constitute complete in-phase component and quadrature component information without the need for subsequent digital orthogonalization processing.

6. The optical phase finite-difference based radio frequency frequency measurement system of claim 1, wherein: The optical angle diversity receiver adopts a three-phase diversity structure of 0 degrees, 120 degrees and 240 degrees, including three photodetectors and three analog-to-digital converters; the first photodetector is used to receive the optical signal of 0-degree phase diversity and output one analog electrical signal. The second photodetector is used to receive optical signals with 120-degree phase diversity and output one analog electrical signal; the third photodetector is used to receive optical signals with 240-degree phase diversity and output one analog electrical signal; the three analog-to-digital converters are used to synchronously convert the three analog electrical signals into three digital signals.

7. The optical phase finite-difference based radio frequency frequency measurement system of claim 1, wherein: The digital signal processing module includes optical angle diversity receiver IQ imbalance compensation, complex domain signal reconstruction, digital filtering, phase calculation, phase unwrapping, and finally, based on the linear mapping relationship between the I / Q signal phase, the laser instantaneous frequency, and the radio frequency instantaneous frequency, the phase stream is converted into the trajectory of the instantaneous radio frequency change, and the instantaneous frequency is calculated.

8. The optical phase finite-difference based radio frequency frequency measurement system of claim 1, wherein: The laser frequency real-time monitoring module adopts a multi-phase diversity structure, including a balanced detector and an analog-to-digital converter. The balanced detector is used to receive the optical signals of each phase diversity and output multiple analog electrical signals. The analog-to-digital converter is used to convert the analog electrical signals into digital signals, process the digital signals, extract the laser frequency drift, and perform correction.

9. A method of radio frequency frequency measurement based on optical phase finite-difference, characterized in that: Includes the following steps: S1: The radio frequency signal to be tested is modulated onto the laser carrier using an optical single-sideband generator, and optical single-sideband modulation is achieved, converting the change of radio frequency into the frequency change of a continuous laser source. S2: The modulated optical signal is split into two branches using an optical power beam splitter, with the optical signal in the lower branch being processed by an optical delay line. S3: The upper and lower branch optical signals are respectively input to the optical angle diversity receiver to generate the in-phase component I-channel electrical signal and the quadrature component Q-channel electrical signal, and then the electrical signals are converted from analog to digital to the digital domain. S4: digital signal processing is performed on the signal converted to the digital domain, and a linear mapping relationship between the frequency change Δf of the radio frequency signal and the phase change ΔΦ of the demodulated baseband signal is utilized , the instantaneous frequency of the radio frequency signal to be measured is calculated, wherein the calibration coefficient k =1 / (2πτ), τ is the introduced fixed optical time delay.

10. The optical phase finite-difference based radio frequency frequency measurement method of claim 9, wherein: In step S4, before calculating the instantaneous frequency of the RF signal under test, the delay of the lower branch is calibrated. The calibration process includes the following steps: S41: Input a calibration radio frequency signal whose frequency varies linearly with time; S42: Acquire and process the two baseband quadrature electrical signals output for the calibration radio frequency signal, perform digital filtering and receiver I / Q amplitude equalization on the signals, and then calculate the continuous instantaneous calibration phase. S43: Compare and perform linear regression analysis on the known frequency change sequence of the calibration radio frequency signal with the calculated continuous instantaneous calibration phase sequence; S44: Through the linear regression analysis, a mapping relationship between a radio frequency signal phase value and a laser frequency value is determined, thereby completing the system calibration; S45: Repeat calibration steps S42-S44 to obtain a series of calibration coefficient samples. Calculate the arithmetic mean of the sample set to determine the final optimized calibration coefficients.

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