Ac motor control device and electric vehicle

Abstract

To provide an AC motor control device capable of eliminating divergence between a current (or magnetic flux) and its command value when a torque command is changed in voltage phase control.SOLUTION: An AC motor control device 100 includes: rectangular wave voltage phase control units 8, 13 for outputting a first gate signal for controlling a voltage phase angle θv so that torque or a current of an AC motor 1 coincides with a torque command value T* or current command values Id*, Iq*; a magnetic flux command calculation unit 15 for calculating magnetic flux command values φd* and φq* from the current command values Id*, Iq*; and a low pass filter 19 that has a magnetic flux command angle calculation unit 41b for calculating a flux command angle ∠φ* of the magnetic flux command values φd*, φq*, performs delay processing corresponding to a response of the rectangular wave voltage phase control unit 8 to the magnetic flux command angle ∠φ*, and outputs magnetic flux filter command values φdf*, φqf*.SELECTED DRAWING: Figure 1

Description

[Technical field]

[0001] The present invention relates to a control device for driving an AC motor and an electric vehicle. [Background technology]

[0002] To improve the voltage utilization rate, pulse-saving control such as square wave drive is used. In order to realize a voltage close to the inverter output voltage limit, Patent Document 1 discloses voltage phase control that controls the voltage phase angle so that the torque matches a command value. Furthermore, Patent Document 1 proposes a means for switching between voltage phase control and two-axis vector control based on the required voltage amplitude calculated from the speed / torque command, current command value, and voltage phase. Patent Document 2 discloses an apparatus including a voltage calculation unit that calculates a magnetic flux command value from a current command value, estimates a magnetic flux value from a current detection value, and creates a voltage command value so that the magnetic flux command value and the magnetic flux value match, and a damping ratio control unit that creates a correction amount for the voltage command value based on the vibration component of the magnetic flux value so that the vibration component is attenuated. In Patent Documents 1 and 2, the current command value and magnetic flux command value used for control are generally subjected to delay processing using a low-pass filter in consideration of the control delay of the current (or magnetic flux) control. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] Patent No. 4939127 [Patent Document 2] Patent Publication No. 2022-144060 Summary of the Invention [Problem to be solved by the invention]

[0004] In voltage phase control, the voltage is controlled to be constant, so when delay processing is performed on the dq-axis current (or magnetic flux) command value described in the above prior art document, taking into account the control delay, a deviation occurs between the actual current (or magnetic flux) and the trajectory of the current command value after delay processing in voltage phase control. For example, FIG. 13 shows a current trajectory 205 and a current command value trajectory 203 when the torque changes stepwise from positive to negative. Since the voltage is basically constant during voltage phase control, it operates on a constant voltage ellipse 201. On the other hand, FIG. 14 shows a magnetic flux trajectory 215 and a magnetic flux command value trajectory 213 when the torque changes stepwise from positive to negative. In the case of magnetic flux, it also operates on a constant voltage ellipse.

[0005] In this way, in both documents, when the torque changes, the current (or magnetic flux) deviates from the command value, and problems arise such as damping ratio control, which is based on the assumption that the current and magnetic flux match the command values, not operating stably, and when switching to two-axis control, the deviation causes the control after the switch to become unstable.

[0006] In consideration of this problem, an object of the present patent is to provide an AC motor control device and an electric vehicle that can eliminate the deviation between the current (or magnetic flux) and its command value when the torque command changes in voltage phase control. [Means for solving the problem]

[0007] In order to achieve the above object, the present invention is configured as follows.

[0008] The AC motor control device includes a rectangular wave voltage phase control unit that outputs a first gate signal that controls a voltage phase angle so that the torque or current of an AC motor coincides with a torque command value or a current command value, a flux command calculation unit that calculates a flux command value from the current command value, and a low pass filter that has a flux command angle calculation unit that calculates a flux command angle of the flux command value, performs delay processing on the flux command angle equivalent to the response of the rectangular wave voltage phase control unit, and outputs a flux filter command value. Effect of the Invention

[0009] The present invention makes it possible to provide an AC motor control device and an electric vehicle that can eliminate the deviation between the current (or magnetic flux) and its command value when the torque command changes in voltage phase control.

[0010] In voltage phase control, the current or magnetic flux will not deviate from the command value, and the control operation using the current command value (or magnetic flux command value) can be stabilized. In addition, torque fluctuations when switching to two-axis vector control can be suppressed. This is expected to improve the ride comfort at high speeds for electric vehicles. [Brief description of the drawings]

[0011] [Figure 1] 1 is a block diagram showing an overall configuration of an AC motor control device according to a first embodiment. [Diagram 2] 2 is an example of a block diagram of a torque / command value calculation unit in the first embodiment. FIG. [Diagram 3] 2 is an example of a block diagram of a voltage phase control unit in the first embodiment. FIG. [Figure 4] 3 is a block diagram of a damping ratio control unit according to the first embodiment. FIG. [Diagram 5] FIG. 2 is an example of a block diagram of a square wave generating unit in the first embodiment. [Figure 6A] FIG. 2 is a block diagram of an example of an LPF in the first embodiment. [Figure 6B] FIG. 4 is a block diagram of another example of the LPF in the first embodiment. [Figure 7] This shows the relationship between the voltage vector and the magnetic flux vector in voltage phase control. [Figure 8] FIG. 11 is a block diagram showing an overall configuration of a motor control device according to a second embodiment. [Figure 9] FIG. 11 is a block diagram of an example of a PWM mode switching unit in the second embodiment. [Figure 10] FIG. 11 is a block diagram of an example of an LPF in the second embodiment. [Figure 11]FIG. 13 is a block diagram of an example of a PWM mode switching unit in a modified example of the second embodiment. [Figure 12] FIG. 11 is a configuration diagram of an electric vehicle according to a third embodiment. [Figure 13] 1A and 1B are diagrams illustrating a current command value and a current locus in voltage phase control. [Figure 14] 11A and 11B are diagrams illustrating a magnetic flux command value and a trajectory of magnetic flux in voltage phase control. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0012] Hereinafter, an embodiment of the present invention will be described with reference to the accompanying drawings.

[0013] Although the following description focuses on a permanent magnet synchronous motor (PMSM), the present invention is not limited to permanent magnet synchronous motors, and similar effects can be obtained with any synchronous machine, such as a synchronous reluctance motor, a permanent magnet synchronous generator, or a wound-rotor synchronous machine.

[0014] Furthermore, the semiconductor switching elements of the inverter device are IGBTs, but the effects of the present invention are not limited to this, and may be MOSFETs or other power semiconductor elements. EXAMPLES

[0015] Example 1 The configuration of the first embodiment will be described.

[0016] FIG. 1 is a block diagram showing an overall configuration of an AC motor control device 100 according to a first embodiment of the present invention.

[0017] The overall configuration of the AC motor control device 100 of the first embodiment will be described with reference to FIG.

[0018] In Fig. 3, a power conversion unit 2 converts DC power from a DC voltage source 9 (e.g., a battery) into AC power in accordance with a gate signal (rectangular wave pulse signal) described later, to drive a PMSM (permanent magnet synchronous motor) 1. A phase current detection means 3 is made up of a Hall CT (Current Transformer) or the like, and detects current waveforms Iuc, Ivc, Iwc of three phases, U phase, V phase, and W phase, flowing from the power conversion unit 2 to the PMSM 1. A magnetic pole position detection means 4 is made up of a resolver or the like, and detects the magnetic pole position of the PMSM 1 to obtain magnetic pole position information θ * The frequency calculation unit 5 outputs the magnetic pole position information θ * From this, for example, the speed information ω1 * The coordinate conversion unit 7 converts the current waveforms Iuc, Ivc, and Iwc detected by the phase current detection unit 3 into magnetic pole position information θ * The coordinates are converted and the d-axis and q-axis current detection values ​​Idc and Iqc are output.

[0019] The current command value generator 17 generates a torque command value T * , speed information ω1 * Based on the DC voltage Vdc detected by the DC voltage detector 6, the dq-axis current command value Id * , Iq * The torque command value calculation unit 12 generates the dq-axis current command value Id * , Iq * Based on the d-axis current detection values ​​Idc and Iqc, the torque command value T ** In FIG. 2, the dq-axis magnetic flux command calculation unit 21 calculates the dq-axis current command value Id * , Iq * For example, the dq axis magnetic flux command φd * , φq * Based on the following equation (1), the torque command value T ** Calculate the following.

[0020]

number

[0021] Similarly, a dq-axis magnetic flux calculation unit 22 calculates dq-axis magnetic fluxes φd, φq from the dq-axis current detection values ​​Idc, Iqc by means similar to that of the dq-axis magnetic flux command calculation unit 21, and calculates torque T based on the following equation (2) using multipliers 24, 26, a subtractor 28, and an amplifier 30.

[0022]

number

[0023] The voltage phase control unit 13 determines whether the torque T is a torque command value T ** For example, as shown in Fig. 3, the torque T and the torque command value T ** The difference between these is calculated by a subtractor 71, passed through a PI controller 73 (or an I controller), and subjected to limit processing by a limiter 75 to output a voltage phase angle θv.

[0024] In FIG. 1, a magnetic flux estimation unit 14 uses d-axis and q-axis currents Idc and Iqc to estimate d-axis and q-axis magnetic flux estimated values ​​φdc and φqc by referring to, for example, a look-up table. A magnetic flux command calculation unit 15 calculates a d-axis and q-axis current command value Id * , Iq * Using this, for example, a look-up table is referenced to determine the dq-axis magnetic flux command value φd * , φq * The LPF 19 outputs the dq-axis magnetic flux command value φd * , φq * The dq-axis flux filter command value φdf taking into account the control delay in voltage phase control using * , φqf * The LPF19 outputs the magnetic flux command angle ∠φ * A delay process equivalent to the response of the voltage phase control unit 13 is performed on the LPF 19. This is the key point of the present invention, so the details of the LPF 19 will be described later. The damping ratio control unit 20 is a controller similar to the controller described in Patent Document 2, and is configured, for example, as shown in FIG. 4. As shown in FIG. 4, the damping ratio control unit 20 calculates a q-axis magnetic flux estimated value φqc and a q-axis magnetic flux filter command value φqf *Difference between (φqc-φqf * ) is calculated by an adder / subtractor 53. The vibration component of the difference calculation value is extracted by a high-pass filter 55 (the transfer function of which is shown in FIG. 4). In addition, the d-axis magnetic flux estimated value φdc and the d-axis magnetic flux filter command value φdf * Difference between (φdc-φdf * ) is calculated by an adder / subtractor 54. The vibration component of the difference calculation value is extracted by a high-pass filter 56 (the transfer function of which is shown in FIG. 4).

[0025] That is, the vibration components of the q-axis magnetic flux and the d-axis magnetic flux are extracted by the high-pass filters 55 and 56, respectively. Furthermore, the vibration component of the q-axis magnetic flux extracted by the high-pass filter 55 is multiplied by φqf * is multiplied by the multiplier 57. Also, the vibration component of the d-axis magnetic flux extracted by the high-pass filter 56 is multiplied by φdf * is multiplied by multiplier 58. The product by multiplier 57 and the product by multiplier 58 are added by adder 59. The sum by adder 59 corresponds to the inner product of the magnetic flux command vector and the magnetic flux vibration component vector.

[0026] In addition, φqf * and φdf * is input to the multipliers 57 and 58 as well as to the square sum calculator 60. The square sum calculator 60 calculates φqf * Squared and φdf * The sum of the squares calculated by the square sum calculator 60 is ((φqf * ) 2 +(φdf * ) 2 ) and the sum by adder 59 are input to divider 61. Divider 61 executes division ((sum)÷(sum of squares)) by dividing the square sum calculated by square sum calculator 60 by the sum by adder 59. Here, the division value by divider 61 is the value of the component of the magnetic flux oscillation component in the amplitude direction of the magnetic flux command ((the inner product of the magnetic flux command vector and the magnetic flux oscillation component vector)÷(the magnitude of the magnetic flux command vector (=(φqf * ) 2 +(φdf * )2 )) 1 / 2 )) is converted into the phase correction amount of the voltage command (provisional correction amount before gain multiplication) ((amplitude direction component of magnetic flux command) ÷ (magnitude of magnetic flux command (=(φqf * ) 2 +(φdf * ) 2 )) 1 / 2 The divided value by the divider 61 is multiplied by a gain (2ζ) by the proportional multiplier 62. As a result, the stabilization voltage command phase correction amount θd * is created.

[0027] In FIG. 1, a subtractor 11 subtracts a stabilization voltage command phase correction amount θd from a voltage phase angle θv. * and outputs the voltage phase angle θv2 to the square wave generating unit 8.

[0028] As shown in FIG. 5, the square wave generating unit 8 receives magnetic pole position information θ * A voltage phase signal is generated by adding voltage phase angle θv2 and π / 2 to this, and a remainder is calculated by dividing this by 2π in a remainder calculation unit 87. π is subtracted in a subtractor 93, and a rectangular wave pulse signal Su is calculated according to the sign determined by a sign determination unit 96.

[0029] Similarly, the adder 83, the remainder calculation unit 89, the subtractor 94, and the sign determiner 97 add 4π / 3 to the voltage phase signal to calculate a square wave pulse signal Sv. Also, the adder 85, the remainder calculation unit 91, the subtractor 95, and the sign determiner 98 add 2π / 3 to the voltage phase signal to calculate a square wave pulse signal Sw. A gate signal 1 (first gate signal) is generated from the square wave pulse signals Su, Sv, and Sw taking into account the dead time, and is output.

[0030] Voltage phase control section 13 and rectangular wave generating section 8 constitute a rectangular wave voltage phase control section.

[0031] Next, the LPF 19, which is the key feature of the present invention, will be described in detail.

[0032] FIG. 6A is a diagram showing an example of the LPF 19.

[0033] 6A, the flux command amplitude angle calculator 41 includes a flux command amplitude calculator 41a and a flux command angle calculator 41b. The flux command amplitude calculator 41a calculates a dq-axis flux command value φd * , φq * from the magnetic flux command amplitude |φ * |(=((φq * ) 2 +(φd * ) 2 )) 1 / 2 The magnetic flux command angle calculation unit 41b calculates and outputs the magnetic flux command angle ∠φ * (=tan ―1 (φq * / φd * )) and output it.

[0034] The delay calculation unit 43 calculates the magnetic flux command amplitude |φ * The flux command filter amplitude |φf * The delay calculation unit 45 outputs the magnetic flux command angle ∠φ * The flux command filter angle ∠φf is calculated by taking into account the control delay of the voltage phase control. * The delay calculation unit 43 and the delay calculation unit 45 are, for example, a first-order delay with the reciprocal of the cutoff angular frequency ωc of the voltage phase control system as a time constant.

[0035] The dq-axis flux filter calculation unit (flux filter calculation unit) 47 is a flux command filter amplitude |φf * | and flux command filter angle ∠φf * to dq-axis magnetic flux filter command value φdf * , φqf * Output.

[0036] FIG. 6B is a diagram showing another example of the LPF 19. In FIG.

[0037] The difference between the example shown in FIG. 6A and the example shown in FIG. 6B is that in FIG. 6B, the delay calculation unit 43 is omitted, and the magnetic flux command amplitude |φ output from the magnetic flux command amplitude angle calculation unit 41 is *is output to the dq-axis magnetic flux filter calculation unit 47. In other words, since the magnetic flux amplitude is basically constant, the present invention is valid even if the delay processing 43 is omitted.

[0038] The principle will be explained below.

[0039] Voltage phase control is generally used in the medium to high speed range, so if the primary resistance component is ignored, the voltage V and magnetic flux φ will be orthogonal as shown in Fig. 7. Also, the amplitude of magnetic flux φ is the voltage V divided by the speed, and since the change in speed is generally sufficiently slow compared to the change in torque, the speed can be considered constant when the torque changes. Therefore, even if the torque command value changes, the amplitude of the magnetic flux command value can be considered to be approximately constant, and the equal voltage ellipse 211 shown in Fig. 14 becomes a circle.

[0040] Therefore, if the magnetic flux command value is converted into an amplitude and an angle and delay processing is performed as shown in Fig. 6A and Fig. 6B in the first embodiment, the magnetic flux command value will also pass through the equal voltage ellipse 211 in the same manner as the magnetic flux locus 215 in Fig. 14. Therefore, no deviation occurs between the magnetic flux and the locus of the magnetic flux command value. This enables the damping ratio control unit 20 to appropriately control the damping ratio in the high speed range.

[0041] Since the present embodiment 1 is configured as described above, it is possible to provide an AC motor control device that can eliminate the deviation between the current (or magnetic flux) and its command value when the torque command changes in voltage phase control.

[0042] In the first embodiment, an example in which a rectangular wave pulse signal is output is shown, but the same effect can be obtained when, for example, the voltage approaches the output limit as in three-pulse control and the torque is controlled by voltage phase control. In the first embodiment, in order to stably operate the damping ratio control using the flux command value, the dq-axis flux filter command value is calculated by converting the dq-axis flux command value into an amplitude and an angle, and then performing delay processing, but when it is desired to match the trajectory of the current and the current command value in voltage phase control, the dq-axis current filter command value may be calculated from the dq-axis flux filter command value calculated in the first embodiment, for example, using a lookup table.

[0043] Example 2 Next, a second embodiment of the present invention will be described.

[0044] FIG. 8 is a block diagram showing an overall configuration of an AC motor control device 101 according to a second embodiment of the present invention.

[0045] The difference in configuration between the first and second embodiments will be described.

[0046] In the second embodiment, vector control of two axes is added, and the PWM mode switching unit 18 switches between square wave control and PWM control. The current control unit 16 performs PI control so that the currents on the d and q axes match, and outputs the d and q axis voltage command values ​​Vd * , Vq * The coordinate conversion unit 7B outputs the dq-axis voltage command value Vd * , Vq * The magnetic pole position θ * and the stabilization voltage command phase correction amount θd * Based on the three-phase voltage command value Vu * , Vv * , Vw * The PWM control unit (overmodulation control unit) 10 generates a pulse by comparing it with a triangular wave, for example, and outputs a gate signal 2 (second gate signal).

[0047] The PWM mode switching unit 18 controls the dq-axis magnetic flux filter command value φdf * , φqf * and the torque command value T * Based on this, the PWM mode is switched from asynchronous PWM mode to square wave mode (i.e., gate signal 2 corresponding to the asynchronous PWM mode and gate signal 1 corresponding to the square wave mode are switched, and output to power converter 2 which converts the power supplied to PMSM 1, which is an AC motor).

[0048] An example of the configuration of the PWM mode switching unit 18 is shown in FIG.

[0049] In FIG. 9, the flux filter command angle calculation unit 121 calculates the dq-axis flux filter command value φdf* , φqf * from the flux filter command angle ∠φf * (=tan ―1 (φqf * / φdf * The switching angle calculation 123 calculates and outputs the magnetic flux switching angle ∠φcp1 according to the torque command value T*. * , ∠φcp2 * For example, a lookup table is used to calculate the flux angle at MTPA (Maximum Torque Per Ampere) as the flux switching angle ∠φcp1 * The magnetic flux switching angle is ∠φcp1 * The angle obtained by adding the hysteresis to the above is the magnetic flux switching angle ∠φcp2 * The output is as follows:

[0050] The switching decision 125 is the magnetic flux filter command angle ∠φf * and the magnetic flux switching angle ∠φcp1 * and ∠φcp2 * Compared with the flux filter command angle ∠φf * is the magnetic flux switching angle ∠φcp2 * When this is reached, switch from asynchronous PWM mode to square wave mode and set the flux filter command angle ∠φf * is the magnetic flux switching angle ∠φcp1 * When the PWM mode is the square wave mode, the mode is switched to the asynchronous PWM mode. The gate signal selection 127 outputs gate signal 1 when the PWM mode is the square wave mode, and outputs gate signal 2 when the PWM mode is the asynchronous PWM mode.

[0051] The LPF 19B switches its operation depending on the PWM mode. An example of the configuration of the LPF 19B is shown in Figure 10. The differences from the LPF 19 shown in Figure 6A will be described below.

[0052] 10, the LPF 19B includes a first low-pass filter 50 and a second low-pass filter 51, and a mode changeover switch 49 switches between the output of the first low-pass filter 50 and the output of the second low-pass filter 51.

[0053] The first low-pass filter 50 has a configuration similar to that of the LPF 19 shown in FIG. 6A.

[0054] The delay calculation unit 42 of the second low-pass filter 51 takes into consideration the control delay on the d-axis side in the current control unit 16, and for example, sets a first-order delay with the reciprocal of the cutoff angular frequency ωcd of the d-axis current control system as a time constant. The delay calculation unit 44 takes into consideration the control delay on the q-axis side in the current control unit 16, and for example, sets a first-order delay with the reciprocal of the cutoff angular frequency ωcq of the q-axis current control system as a time constant. In other words, the second low-pass filter 51 calculates the magnetic flux command value φd * , φq * A first-order delay process corresponding to the response of the overmodulation control unit 10 is then performed.

[0055] Switch 49 is φdf2 when the PWM mode is asynchronous PWM mode. * , φqf2 * In square wave mode, φdf1 is output. * , φqf1 * Output.

[0056] In the second embodiment, in addition to the same effects as those in the first embodiment, the following effects can also be obtained.

[0057] In the second embodiment, the difference between the magnetic flux and the magnetic flux filter command value is small even during a transient response when the torque command value changes, so switching can be performed at an appropriate timing. This makes it possible to prevent torque shock and control instability during switching, as well as hunting in the PWM mode.

[0058] Furthermore, since LPF 19B can take into consideration that the trajectory of the magnetic flux changes depending on the selected control, even in a situation where the torque changes, the damping ratio control unit 20 can achieve stable operation with a small deviation between the magnetic flux and the magnetic flux filter command value regardless of the PWM mode.

[0059] (Modification of the second embodiment) In the modified example of the second embodiment, the PWM mode switching unit 18 is different from that in the second embodiment. The PWM mode switching unit 18 in the modified example is shown in FIG.

[0060] In FIG. 11, the d-axis MTPA magnetic flux calculation unit 122 calculates the torque command value T * Based on this, the d-axis magnetic flux command value φdMTPA in MTPA * The switching decision 124 calculates the d-axis magnetic flux command value φdMTPA in MTPA. * and d-axis magnetic flux filter command value φdf * The difference between these two is taken, and when the difference is equal to or greater than a set value 1, taking hysteresis into consideration, the mode is switched from asynchronous PWM mode to square wave mode, and when the difference is equal to or less than a set value 2, the mode is switched from square wave mode to asynchronous PWM mode (in other words, gate signal 2 corresponding to asynchronous PWM mode and gate signal 1 corresponding to square wave mode are switched, and the signal is output to power converter 2, which converts the power to be supplied to PMSM 1, an AC motor).

[0061] In this modified example, the switching timing can be appropriately set, as in Example 2. Additionally, even when using the present invention to perform switching based on a required voltage amplitude as in Patent Document 1, for example, the magnetic flux filter command value obtained in the present invention (or the current filter command value calculated from the magnetic flux filter command value) is used to calculate the required voltage amplitude, so that the switching timing can be appropriately set when the torque command value changes.

[0062] Example 3 Next, a description will be given of a third embodiment. The third embodiment is an example in which the first or second embodiment is applied to a motor control device 300 for an electric vehicle 308.

[0063] FIG. 12 is a configuration diagram of an electric car 308 in the third embodiment of the present invention.

[0064] As shown in the first and second embodiments, the AC motor control device 300 controls the power supplied from the power conversion unit (inverter) 2 to the PMSM 1. A DC voltage source (e.g., a battery) 9 supplies power to the inverter 2. The PMSM 1 is connected to a transmission 301. The transmission 301 supplies power to wheels 307 connected to a drive shaft 305 via a differential gear 303. Note that a configuration may be adopted in which the transmission 305 is not provided and the PMSM 1 is directly connected to the differential gear 303, or a configuration in which the PMSM 1 and the inverter 2 are applied to the front wheels and the rear wheels, respectively.

[0065] When electric vehicle 308 is an automobile, torque fluctuations are directly linked to the ride comfort, so requirements for torque fluctuations are stricter than in other applications, and this is an application in which the effects of the present invention are more likely to appear.

[0066] Similarly, torque fluctuations in railway vehicles are directly linked to the ride comfort, so this is an application in which the effects of the present invention are likely to be seen.

[0067] By applying the present invention to the motor control device 300 of the electric vehicle 308, it is possible to improve the ride comfort at high speeds in electric vehicles such as automobiles and railway cars. [Explanation of symbols]

[0068] 1···PMSM, 2···power conversion section, 3···current detection means, 4···magnetic pole position detector, 5···frequency calculation section, 6···DC voltage detection section, 7, 7B···coordinate conversion section, 8···rectangular wave generation section, 9···DC voltage source (battery), 10···PWM control section (overmodulation control section), 11···subtractor, 12···torque / command value calculation section, 13···rectangular wave voltage phase control section, 14···magnetic flux estimation section, 15···magnetic flux command calculation section, 16···current control section, 17···current command value generation section, 18···PWM mode switching section, 19·· 20···LPF, 19B···LPF (embodiment 2), 20···damping ratio control section, 21···dq-axis magnetic flux command calculation section, 22···dq-axis magnetic flux calculation section, 23···multiplier, 24···multiplier, 25, 26···multiplier, 27, 28···subtractor, 29, 30···amplifier, 41···magnetic flux command amplitude angle calculation section, 41a···magnetic flux command amplitude calculation section, 41b···magnetic flux command angle calculation section 41b, 42, 43, 44, 45···delay calculation section, 47···dq-axis magnetic flux filter calculation section (magnetic flux filter calculation section), 49···mode changeover switch, 50 1. First low-pass filter, 51, second low-pass filter, 53, 54, subtractor, 55, 56, high-pass filter, 57, 58, multiplier, 59, adder, 60, sum-of-squares calculator, 61, divider, 62, proportional unit, 71, subtractor, 73, PI controller, 75, limiter, 81, 83, 85, adder, 87, 89, 91, remainder calculator, 93, 94, 95, subtractor, 96, 97, 98, sign detector, 100, 101, 300, AC motor control device, 121, ·Flux filter command angle calculation unit, 122··d-axis MTPA flux calculation unit, 123···switching angle calculation unit, 124, 125···switching decision unit, 127···gate signal selection unit, 141···voltage vector, 143···flux vector, 201, 211···constant voltage ellipse, 203···current command locus, 205···current locus, 213···flux command locus, 215···flux locus, 301···transmission, 303···differential gear, 305···drive shaft, 307···wheel, 308···electric vehicle

Claims

1. A rectangular wave voltage phase control unit that outputs a first gate signal that controls the voltage phase angle so that the torque or current of an AC motor matches the torque command value or current command value, A magnetic flux command calculation unit that calculates a magnetic flux command value from the current command value, A low-pass filter has a magnetic flux command angle calculation unit that calculates the magnetic flux command angle of the magnetic flux command value, performs a delay processing on the magnetic flux command angle corresponding to the response of the rectangular wave voltage phase control unit, and outputs a magnetic flux filter command value. An AC motor control device characterized by comprising the following features.

2. In the AC motor control device according to claim 1, The AC motor control device is characterized in that the low-pass filter has a magnetic flux amplitude calculation unit that calculates the magnetic flux command amplitude of the magnetic flux command value, and performs a delay processing on the magnetic flux command amplitude that corresponds to the response of the rectangular wave voltage phase control unit.

3. In the AC motor control device according to claim 2, The AC motor control device is characterized in that the low-pass filter has a magnetic flux filter calculation unit that calculates a magnetic flux command filter command value from the magnetic flux command amplitude and the magnetic flux command angle after the delay processing.

4. In the AC motor control device according to claim 1, The rectangular wave voltage phase control unit has a rectangular wave generation unit that generates a rectangular wave pulse signal based on the voltage phase angle, An AC motor control device further comprising a power conversion unit that converts DC power to AC power based on the aforementioned rectangular wave pulse signal, and characterized in that the AC motor is driven by the AC power.

5. In the AC motor control device according to claim 1, An overmodulation control unit controls the voltage amplitude and voltage phase so that the torque or current of the AC motor matches the torque command value or the current command value, and outputs a second gate signal. A mode switching determination unit switches the first gate signal and the second gate signal based on the magnetic flux filter command value output from the low-pass filter and the torque command value, and outputs to a power converter that converts the power supplied to the AC motor. An AC motor control device characterized by comprising the following features.

6. In the AC motor control device according to claim 1, An overmodulation control unit controls the voltage amplitude and voltage phase so that the torque or current of the AC motor matches the torque command value or the current command value, and outputs a second gate signal. A mode switching determination unit switches the first gate signal and the second gate signal based on the difference between the d-axis magnetic flux filter command value output from the low-pass filter and the d-axis magnetic flux command value in the MPTA calculated based on the torque command value, and outputs to a power converter that converts the power supplied to the AC motor. An AC motor control device characterized by comprising the following features.

7. In the AC motor control device according to claim 5, The aforementioned low-pass filter is A first low-pass filter performs a delay process on the magnetic flux command angle corresponding to the response of the rectangular wave voltage phase control unit, A second low-pass filter performs a first-order delay process on the magnetic flux command value, which is the output of the magnetic flux command calculation unit, corresponding to the response of the overmodulation control unit. A mode switching switch for switching between the first low-pass filter and the second low-pass filter, It has, The aforementioned mode selector switch is When the rectangular wave voltage phase control unit is operating, it outputs the magnetic flux filter command value output from the first low-pass filter. When the overmodulation control unit is operating, it outputs the magnetic flux filter command value output from the second low-pass filter. An AC motor control device characterized by the following features.

8. An electric vehicle characterized by comprising an AC motor control device according to any one of claims 1 to 7. Calculate, The temperature of the steam discharged from the compressor is compared with the saturation temperature. The amount of water separated from the steam drawn into the compressor and supplied to the compressor is controlled so that the temperature of the steam discharged from the compressor reaches the saturation temperature. Steam generation method.

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