Hybrid rectifier bridge LLC resonant converter with wide output voltage range and control method thereof

Abstract

The invention discloses a hybrid rectifier bridge LLC resonant converter with a wide output voltage range and a control method thereof, and belongs to the technical field of power generation, power transformation or power distribution. The converter comprises a primary-side inversion network, a resonance network, a high-frequency transformer bank and a secondary-side hybrid rectification network with a bidirectional switch, and ultra-wide output voltage gain is achieved. According to the control method, the primary side adopts phase shift control, and the secondary side adaptively switches the working modes of the full-bridge rectification network and the voltage doubling rectification network through a superposition type asymmetric PWM (Pulse Width Modulation) and phase shift control mode, so that a half-cycle soft switching is provided for the bidirectional switch while the voltage regulation range is greatly widened, and the voltage regulation efficiency is improved. And meanwhile, the loss of the diode is greatly reduced, and the optimization between low switching loss and low conduction loss is realized. According to the invention, free switching between a high-voltage mode and a low-voltage mode can be realized under the condition that the original advantages of LLC are kept, the control method is simple and effective, and improvement of conversion efficiency can be ensured while an ultra-wide output voltage range is realized.

Description

Technical Field

[0001] The present invention discloses a hybrid rectifier bridge LLC resonant converter with a wide output voltage range and a control method thereof, relating to power electronics technology, specifically to DC-DC power conversion and converter control technology, and belonging to the technical field of power generation, power transformation or power distribution. Background Art

[0002] In the fields of new energy vehicles, spacecraft, new energy power generation, data centers, etc., since the terminal voltage of the load has a wide range of characteristics, its power supply system needs to achieve power conversion in a wide voltage range while ensuring high conversion efficiency in the entire range.

[0003] Power converters are core components of power supply systems. LLC resonant converters are popular due to their small size, high efficiency, and high power density. Phase-shift control is widely used to broaden the output voltage gain range. However, when the phase-shift angle is too large, one of the switches in the LLC converter's primary circuit loses its soft-switching characteristics, resulting in a loss of efficiency. Adding a pair of bidirectional switches to modify the secondary-side structure has become a promising solution. This allows the secondary-side variable-structure rectifier bridge to be configured in various modes, including bridge rectification and voltage doubler rectification, covering a wider gain range with multiple gain modes. However, the addition of additional bidirectional switches increases both switching and conduction losses, significantly impacting converter efficiency. To reduce losses, existing research has proposed adding a single switching transistor to reduce losses. However, in high-voltage mode, this method can only operate in voltage-doubling rectification mode, losing the flexible switching advantage of the hybrid rectifier bridge. This can improve converter efficiency but cannot meet the requirements of widening the output voltage range. To reduce conduction losses, existing research has proposed a PWM control method to reduce conduction losses. However, because symmetrical PWM control prioritizes output voltage control, it can only moderately reduce diode conduction losses. At the same time, hard switching still introduces switching losses, which means that while widening the output voltage range, it also reduces converter efficiency. Therefore, in the application of secondary-side variable-structure rectifier bridges, it is challenging to propose an adaptive, efficient, reliable, and simple control method. Summary of the Invention

[0004] The purpose of the present invention is to address the shortcomings of the above-mentioned background technology and propose a hybrid rectifier bridge LLC resonant converter with a wide output voltage range and a control method thereof, so as to solve the technical problem of the contradiction between wide gain and high efficiency of the existing LLC converter, and achieve the invention purpose of obtaining an ultra-wide output voltage range while simplifying control and improving efficiency.

[0005] The present invention adopts the following technical solutions to achieve the above-mentioned purpose:

[0006] A hybrid rectifier bridge LLC resonant converter with a wide output voltage range, comprising: a primary-side half-bridge inverter network, a resonant network, a high-frequency transformer group, and a secondary-side hybrid rectifier network;

[0007] A primary-side half-bridge inverter network includes a first half-bridge inverter unit and a second half-bridge inverter unit, wherein the first half-bridge inverter unit and the second half-bridge inverter unit are respectively connected in parallel with the input DC power supply, the first half-bridge inverter unit is composed of a first switching tube and a second switching tube connected in series, and the second half-bridge inverter unit is composed of a third switching tube and a fourth switching tube connected in series;

[0008] The resonant network includes a first resonant tank and a second resonant tank, wherein the first resonant tank is connected to the output end of the first half-bridge inverter unit, and the second resonant tank is connected to the output end of the second half-bridge inverter unit;

[0009] A high-frequency transformer group includes an inverting high-frequency transformer and a positive-phase high-frequency transformer, wherein the primary winding of the inverting high-frequency transformer is connected in parallel with the excitation inductance in the first resonant slot, the primary winding of the positive-phase high-frequency transformer is connected in parallel with the excitation inductance in the second resonant slot, and the secondary winding of the inverting high-frequency transformer is connected in positive series with the secondary winding of the positive-phase high-frequency transformer;

[0010] The secondary-side hybrid rectifier network includes a full-bridge rectifier network and a pair of bidirectional switches. The pair of bidirectional switch tubes include a fifth switch tube and a sixth switch tube connected with a common source. The positive polarity end of the secondary winding of the inverting high-frequency transformer and the secondary winding of the positive-phase high-frequency transformer after being connected in series in a forward direction is connected to the midpoint of the first bridge arm of the full-bridge rectifier network. The negative polarity end of the secondary winding of the inverting high-frequency transformer and the secondary winding of the positive-phase high-frequency transformer after being connected in series in a forward direction and the drain of the fifth switch tube are connected to the midpoint of the second bridge arm of the full-bridge rectifier network. The drain of the sixth switch tube is connected to the filter output unit.

[0011] As a further optimization scheme for a hybrid rectifier bridge LLC resonant converter with a wide output voltage range, in the first half-bridge inverter unit, the drain of the first switching tube is connected to the positive electrode of the input DC power supply, the drain of the second switching tube is connected to the source of the first switching tube, the source of the second switching tube is connected to the negative electrode of the input DC power supply, and the drain and source of the second switching tube constitute the output end of the first half-bridge inverter unit; in the second half-bridge inverter unit, the drain of the third switching tube is connected to the positive electrode of the input DC power supply, the drain of the fourth switching tube is connected to the source of the third switching tube, the source of the fourth switching tube is connected to the negative electrode of the input DC power supply, and the drain and source of the fourth switching tube constitute the output end of the second half-bridge inverter unit.

[0012] As a further optimization scheme for a hybrid rectifier bridge LLC resonant converter with a wide output voltage range, the first resonant tank includes: a first resonant inductor, a first excitation inductor, and a first resonant capacitor. One end of the first resonant inductor is connected to the source of the first switching tube and the drain of the second switching tube, the other end of the first resonant inductor is connected to one end of the first excitation inductor, the other end of the first excitation inductor is connected to the positive electrode of the first resonant capacitor, the negative electrode of the first resonant capacitor is connected to the source of the second switching tube, and the primary excitation inductor of the inverting high-frequency transformer serves as the first excitation inductor. The second resonant tank includes: a second resonant inductor, a second excitation inductor, and a second resonant capacitor. One end of the second resonant inductor is connected to the source of the third switching tube and the drain of the fourth switching tube, the other end of the second resonant inductor is connected to one end of the second excitation inductor, the other end of the second excitation inductor is connected to the positive electrode of the second resonant capacitor, and the negative electrode of the second resonant capacitor is connected to the source of the fourth switching tube. The primary excitation inductor of the positive high-frequency transformer serves as the second excitation inductor.

[0013] As a further optimization scheme for the hybrid rectifier bridge LLC resonant converter with a wide output voltage range, the first end of the primary winding of the inverting high-frequency transformer is connected to the other end of the first resonant inductor and one end of the first excitation inductor, and the second end of the primary winding of the inverting high-frequency transformer is connected to the positive electrode of the first resonant capacitor; the first end of the primary winding of the positive-phase high-frequency transformer is connected to the other end of the second resonant inductor and one end of the second excitation inductor, and the second end of the primary winding of the positive-phase high-frequency transformer is connected to the positive electrode of the second resonant capacitor.

[0014] As a further optimization scheme for a hybrid rectifier bridge LLC resonant converter with a wide output voltage range, a full-bridge rectifier network includes: first to fourth diodes, the cathode of the first diode is connected to the cathode of the third diode, the cathode of the second diode is connected to the anode of the first diode to form the midpoint of the first bridge arm of the full-bridge rectifier network, the cathode of the fourth diode is connected to the anode of the third diode to form the midpoint of the second bridge arm of the full-bridge rectifier network, and the anode of the second diode is connected to the anode of the fourth diode; the first end of the secondary winding of the inverting high-frequency transformer is connected to the midpoint of the first bridge arm of the full-bridge rectifier network as a positive polarity end, the first end of the secondary winding of the positive-phase high-frequency transformer is connected to the second end of the secondary winding of the inverting high-frequency transformer, and the second end of the secondary winding of the positive-phase high-frequency transformer is connected to the midpoint of the second bridge arm of the full-bridge rectifier network as a negative polarity end, the first end of the secondary winding of the inverting high-frequency transformer and the second end of the primary winding of the inverting high-frequency transformer are the same-named terminals, and the first end of the secondary winding of the positive-phase high-frequency transformer and the first end of the primary winding of the positive-phase high-frequency transformer are the same-named terminals.

[0015] Control method of hybrid rectifier bridge LLC resonant converter with wide output voltage range,

[0016] A hybrid rectifier bridge LLC resonant converter with a wide output voltage range and its control method include two operating modes, each using a different control method. In low-voltage mode, primary-side phase-shift control is adopted; in high-voltage mode, the primary side adopts phase-shift control, and the secondary side adopts superimposed asymmetric PWM and phase-shift control strategies. This can achieve ultra-wide output voltage regulation while improving conversion efficiency. The specific steps are as follows:

[0017] Step 1: The control system controls the sampling chip to collect the output voltage and secondary-side current of the converter, and determines whether the expected voltage belongs to the high-voltage mode or the low-voltage mode. If the expected voltage is within the output voltage adjustment range of the high-voltage mode, the process jumps to step 3; if the expected voltage is within the output voltage adjustment range of the low-voltage mode, the process jumps to step 2.

[0018] Step 2: The difference between the converter output voltage collected in step 1 and the desired voltage is adjusted through a voltage regulator to obtain a required phase shift angle α, and the control signals of the first switch tube and the second switch tube are phase-shifted to obtain control signals of the third switch tube and the fourth switch tube. The required first switch tube control signal, second switch tube control signal, third switch tube control signal, and fourth switch tube control signal are output to adjust the converter output voltage. The fifth switch tube and the sixth switch tube are not controlled. The converter operating mode is similar to that of a phase-shifted full-bridge, and the normalized voltage gain is 0-1.

[0019] Step 3: Entering the high-voltage mode, performing a 180° phase shift adjustment on the control signals of the first switch tube and the second switch tube to obtain the control signals of the third switch tube and the fourth switch tube, and outputting the required first switch tube control signal, second switch tube control signal, third switch tube control signal, and fourth switch tube control signal;

[0020] Step 4: Send the difference between the converter output voltage collected in step 1 and the desired voltage to the voltage regulator for adjustment, and output the first duty cycle D y1 The first PWM wave control waveform of the fifth switch tube;

[0021] Step 5: Similar to step 4, the first duty cycle is obtained as D y1 The PWM wave is then phase-shifted by 180° to obtain the first PWM wave control waveform of the sixth switch tube. The fifth and sixth switch tubes driven by the first PWM wave of the fifth and sixth switch tubes are turned on, and the natural conduction of their body diodes can achieve wide output voltage regulation.

[0022] Step 6, according to the formula Get Δt1 through D y1 、D y2 The relationship between Δt1 and the second duty cycle is calculated as D y2The PWM wave of the fifth switch is phase-shifted by 180° to obtain the second PWM wave of the fifth switch. The current that originally flows through the body diode of the fifth switch is changed to flow through the switch channel, reducing the diode conduction loss. At the same time, the time difference Δt1 provides the condition for the fifth switch to achieve zero voltage turn-on.

[0023] Step 7: Similarly, obtain the second duty cycle D y2 The PWM wave of the sixth switch tube is obtained to obtain the second PWM wave of the sixth switch tube, so as to reduce the diode conduction loss of the sixth switch tube and realize zero voltage turn-on of the sixth switch tube, thereby reducing the conduction loss and switching loss;

[0024] Step 8: The secondary current i collected in step 1 is sec The zero-crossing time t is obtained by inputting the current zero-crossing detection circuit. zero , thus obtaining the angle θ of the current zero crossing point lagging behind the primary side voltage, which is used for phase shift control correction;

[0025] Step 9: superimpose the first PWM wave and the second PWM wave control signals of the fifth switch tube, and then phase-shift the obtained signal by an angle θ to obtain a control signal of the fifth switch tube in the final high-voltage mode;

[0026] Step 10: Similarly, after superimposing the first PWM wave and the second PWM wave control signals of the sixth switch tube, the obtained signal is phase-shifted by an angle θ to obtain the final sixth switch tube control signal; at this point, the control signals under all high-voltage mode adjustments are obtained.

[0027] The present invention adopts the above technical solution to solve the technical problem of the contradiction between wide gain and high efficiency of the existing LLC converter, and has the following beneficial effects:

[0028] (1) The hybrid rectifier bridge LLC resonant converter with a wide output voltage range proposed in the present invention has a converter topology with the primary side in parallel and the secondary side in series. The energy generated by the primary-side parallel circuit structure is transmitted to the secondary-side series structure through a positive-phase high-frequency transformer and a negative-phase high-frequency transformer, thereby achieving an ultra-wide output voltage gain.

[0029] (2) Compared with the existing high-efficiency LLC resonant converter technology, the hybrid rectifier bridge LLC resonant converter control method with a wide output voltage range proposed in the present invention combines the primary side phase-shift control strategy and the secondary side superimposed asymmetric PWM and phase-shift control strategy. Under the premise of ultra-wide output voltage gain, it can reduce the conduction time of the diode and realize soft switching of half a cycle of the bidirectional switch, thereby achieving an optimal balance between reducing switching loss and conduction loss, suppressing voltage spikes, and making the control simple and convenient.

[0030] (3) Compared with the existing wide-gain LLC resonant converter technology, the hybrid rectifier bridge LLC resonant converter control method with a wide output voltage range proposed in the present invention does not require frequency modulation and the addition of too many additional magnetic components, has fewer topological devices, and a simple structure; at the same time, the converter always operates at the resonant point, and the magnetic components are easy to design; the voltage gain can be greatly widened by only using software to implement the switching of the two working modes, and the implementation method is simple and reliable. BRIEF DESCRIPTION OF THE DRAWINGS

[0031] Figure 1 This is a topological diagram of the hybrid rectifier bridge LLC resonant converter with a wide output voltage range proposed by the present invention.

[0032] FIG2(a) and FIG2(b) are switching timing waveform diagrams of the converter proposed in the present invention in low-voltage mode and high-voltage mode.

[0033] Figure 3 It is the key waveform diagram under the superimposed asymmetric PWM and phase shift control of the present invention.

[0034] Figure 4 It is an equivalent circuit diagram under superimposed asymmetric PWM and phase shift control of the present invention.

[0035] FIG5(a) is a graph showing the relationship between the voltage gain and the phase shift angle α of the converter proposed in the present invention when it is in low-voltage mode, and FIG5(b) is a graph showing the relationship between the voltage gain and the duty cycle D when the converter proposed in the present invention is in high-voltage mode. y1 relationship curve diagram.

[0036] FIG6(a) shows the control signal S1 of the first switch tube, the control signal S5 of the fifth switch tube, and the control signal S6 of the first switch tube, as well as their voltage waveforms v under the working conditions of an output voltage of 300V and an output power of 3000W in one embodiment of the present invention. ds6 6(b) is an embodiment of the present invention, under the working conditions of output voltage of 300V and output power of 3000W, the control signal S1 of the first switch tube, the control signal S5 of the fifth switch tube and the secondary side voltage V sec And the resonant current i Lr1 , Excitation current waveform i Lm1 .

[0037] FIG7(a) shows the control signal S1 of the first switch tube, the control signal S5 of the fifth switch tube, and the control signal S6 of the first switch tube, as well as their voltage waveforms v under the working conditions of an output voltage of 500V and an output power of 3000W in one embodiment of the present invention. ds67(b) is a graph showing the control signal S1 of the first switch tube, the control signal S5 of the fifth switch tube and the secondary side voltage V under the working conditions of the output voltage of 500V and the output power of 3000W in one embodiment of the present invention. sec And the resonant current i Lr1 , Excitation current waveform i Lm1 .

[0038] Description of the symbols in the figure: V in , input DC power supply, Q1, first switch tube, Q2, second switch tube, L r1 , the first resonant inductor, C r1 , the first resonant capacitor, L m1 , the first excitation inductor, Q3, the third switch tube, Q4, the fourth switch tube, L r2 , the second resonant inductor, C r2 , the second resonant capacitor, L m2 , second excitation inductor, T1, reverse phase high frequency transformer, T2, positive phase high frequency transformer, D1~D4, first to fourth diodes, Q5, fifth switch tube, Q6, sixth switch tube, C o1 , the first filter capacitor, C o2 , the second filter capacitor, R o , load resistance, V o , output DC voltage. DETAILED DESCRIPTION

[0039] The technical solution of the present invention will be described in detail below with reference to the accompanying drawings.

[0040] like Figure 1 The figure shows the topology of the hybrid rectifier bridge LLC resonant converter with a wide output voltage range proposed by the present invention, which includes a primary side half-bridge inverter network, a resonant network, a high-frequency transformer group, and a secondary side hybrid rectifier network. The primary side half-bridge inverter network consists of two half-bridge inverter units that share a secondary side hybrid rectifier network. Each half-bridge inverter unit is connected to the input DC power supply V in In parallel, the output end of each half-bridge inverter unit is connected to a resonant tank respectively. Each resonant tank is composed of a resonant inductor, an excitation inductor, and a resonant capacitor in series. The primary excitation inductor of the reverse high-frequency transformer and the primary excitation inductor of the positive high-frequency transformer respectively participate in the work of the resonant tank as excitation inductors. Due to the different winding methods of the two transformers, the energy polarities transmitted to the secondary side by the two resonant tanks through the transformer are different; the secondary side hybrid rectification network is composed of a full-bridge rectifier network and a pair of common-source back-to-back bidirectional switches in series. The secondary winding of the reverse high-frequency transformer and the secondary winding of the positive high-frequency transformer are connected in series and connected between the middle of the bridge arm of the full-bridge rectifier network; the special structure of the original parallel and anti-series can achieve ultra-wide range voltage output in low voltage mode.

[0041] Figure 1 The specific connection relationship of the converter topology shown is as follows: the complementary conductive first switch tube Q1 and the second switch tube Q2 are connected in series to form a first half-bridge inverter network, and the drain of the first switch tube Q1 is connected to the input DC power supply V in The positive electrode is connected to the source of the second switch tube Q2 and the input DC power supply V in Negative connection, first resonant inductor L r1 One end is connected to the source of the first switch tube Q1 and the drain of the second switch tube Q2, and the first resonant inductor L r1 The other end is connected to the first excitation inductor L m1 One end, the first excitation inductor L m1 The other end is connected to the first resonant capacitor C r1 Positive electrode, first resonant capacitor C r1 The negative electrode is connected to the source of the second switch tube Q2, and the drain of the second switch tube Q2 is connected to the first resonant inductor L r1 The connection point between one end and the source of the first switch tube Q1 is a1, and the source of the second switch tube Q2 and the first resonant capacitor C r1 The connection point of the negative electrode is b1; the complementary conductive third switch tube Q3 and the fourth switch tube Q4 are connected in series to form a second half-bridge inverter network, and the third switch tube Q3 is connected to the input DC power supply V in The positive electrode is connected to the source of the fourth switch tube Q4 and the input DC power supply V in Negative connection, second resonant inductor L r2 One end is connected to the source of the third switch tube Q3 and the drain of the fourth switch tube Q4, and the second resonant inductor L r2 The other end is connected to the second excitation inductor L m2 One end, the second excitation inductor L m2 The other end is connected to the second resonant capacitor C r2 Positive pole, second resonant capacitor C r2 The negative electrode is connected to the source of the fourth switch tube Q4, and the drain of the fourth switch tube Q4 is connected to the second resonant inductor L r2 The connection point between one end and the source of the third switch tube Q3 is a2, and the source of the fourth switch tube Q4 and the second resonant capacitor C r2 The connection point of the negative pole is b2; the primary winding of the inverting high-frequency transformer T1 and the first excitation inductor L m1 In parallel, the primary winding of the positive phase high frequency transformer T2 and the second excitation inductor L m2 In parallel, the first excitation inductor L m1 It is realized by the primary excitation inductance of the inverting high-frequency transformer, and the second excitation inductance L m2It is realized by the primary excitation inductance of the positive high-frequency transformer; the full-bridge rectifier network is composed of the first to fourth diodes D1 to D4, the first diode D1 and the second diode D2 are connected in series to form a rectifier bridge arm, the third diode D3 and the fourth diode D4 are connected in series to form another rectifier bridge arm, the first end of the secondary winding of the inverting high-frequency transformer T1 is connected to the anode of the first diode D1 and the cathode of the second diode D2, the first end of the secondary winding of the inverting high-frequency transformer T1 is connected to the first resonant capacitor C r1 The second ends of the positive pole of the inverting high-frequency transformer T1 primary winding are the same-name ends, the first end of the positive phase high-frequency transformer T2 secondary winding is connected to the second end of the inverting high-frequency transformer T1 secondary winding, and the first end of the positive phase high-frequency transformer T2 secondary winding is connected to the second resonant inductor L r2 The first ends of the primary windings of the positive-phase high-frequency transformer T2 are connected to each other as the same-named ends, and the first end of the secondary winding of the positive-phase high-frequency transformer T2 is connected to the anode of the third diode D3 and the cathode of the fourth diode D4; a pair of common-source back-to-back bidirectional switches is composed of a fifth switch tube Q5 and a sixth switch tube Q6 connected in series, the drain of the fifth switch tube Q5 is connected to the anode of the third diode D3 and the cathode of the fourth diode D4, the source of the fifth switch tube Q5 is connected to the source of the sixth switch tube Q6, and the drain of the sixth switch tube Q6 is connected to the first filter capacitor C o1 First pole, second filter capacitor C o2 One pole is connected, the first filter capacitor C o1 The other pole, the second filter capacitor C o2 The other pole is connected in parallel with a load resistor R o The two resonant tanks have the same parameters, namely L r1 =L r2 ,C r1 =C r2 ,L m1 =L m2 .

[0042] against Figure 1 In the converter shown, by adjusting the duty cycle and phase of a pair of variable structure rectifier bidirectional switches on the secondary side, the conduction loss can be reduced while achieving an ultra-wide range voltage output in the high-voltage mode.

[0043] Figures 2(a) and 2(b) show the switching timing waveforms of the converter in the low-voltage and high-voltage operating modes. The low-voltage mode includes one degree of freedom: the phase difference between the turn-on moments of the switches Q1 and Q3, defined here as the phase shift angle α, to achieve smooth control over the voltage range of 0 to 1. The high-voltage mode includes three degrees of freedom: the duty cycle D of the control signals S5' and S6'. y1 , the duty cycle D of the control signals S5" and S6" y2 , and the phase shift angle θ between the turn-off time of the switches Q5 and Q6 and the turn-off time of the switch Q1, that is, the secondary side current isec The angle of the primary side voltage after the zero crossing point, the primary side voltage is the voltage between points a1 and b1, θ = ω0·Δt2. Δt1 needs to meet the critical conditions of soft switching of switch tubes Q5 and Q6, that is, Δt2 is equal to the secondary side current i after the primary side switch is switched sec The time it takes to go to zero naturally. By adjusting D y1 To control the output voltage, control D y2 To optimize conduction and switching losses, the phase-shift angle θ is used for phase correction, thereby suppressing secondary-side voltage spikes. Because the control signal for switch Q5 is formed by the superposition of S5' and S5", and the control signal for switch Q6 is formed by the superposition of S6' and S6", and because S5' and S5" have different on-duty cycles, and S6' and S6" have different on-duty cycles, combined with phase-shift control correction, the control method in high-voltage mode is called superimposed asymmetric PWM and phase-shift control.

[0044] Figure 3 The key waveforms under the superimposed asymmetric PWM and phase-shift control of the present invention are shown, showing the control waveforms and important voltage and current waveforms of each switch tube. Switch tubes Q1 and Q2 are complementary and turned on during the dead zone, and switch tubes Q3 and Q4 are complementary and turned on during the dead zone. The control signals of Q1 and Q2 are phase-shifted by 180° to obtain the control signals of Q3 and Q4. The two half-bridge networks transfer energy to the secondary side through the resonant network in an interleaved manner. The control signals of switch tubes Q5 and Q6 are consistent with those in Figure 2(b). For the convenience of discussion, the variables are unified into angles for explanation. In π(1-D y1 ) angle, the resonant current is equal to the excitation current and the two components continue to resonate. The current transmitted to the secondary side is naturally reversed. The switch tubes Q5 and Q6 are not turned on, the bidirectional switch is turned off, and the hybrid rectifier bridge working mode is adaptively changed to full-bridge rectification. V sec =-V o In the angle range of ω0·Δt1, the switch tube Q5 is turned on and the switch tube Q6 is still in the off state. Since the current i sec The negative current discharges the junction capacitance of the switch tube Q6 and flows through the body diode of the switch tube Q6, forming a voltage doubling network. The working mode of the hybrid rectifier bridge changes to voltage doubling rectification. V sec =-1 / 2V o In πD y2 In this angle, the switch tubes Q5 and Q6 are both in the on state, and the hybrid rectifier bridge is still working in the voltage doubler rectification mode. sec =-1 / 2V o By using phase shift control correction, the turn-off time of the switch tubes Q5 and Q6 is aligned with the secondary side current i secThe commutation time is the same, effectively suppressing the secondary side voltage spike. sec It can be seen from the waveform that the output voltage can be adjusted by changing the duty cycle and phase shift angle of the superimposed asymmetric PWM wave on the secondary side, and then changing the time when the hybrid rectifier network changes mode.

[0045] Figure 4 The figure shows the control circuit of the phase-shift control and superimposed asymmetric PWM and phase-shift control strategy of the present invention, and its control mode is divided into two modes: high-voltage mode and low-voltage mode.

[0046] When a given voltage V o_ref When the voltage is less than 300V, the converter will be in low voltage mode and adopt phase shift control strategy for the primary side, which is simple to control. At this time, the output voltage V o With a given voltage V o_ref The difference is sent to the voltage regulator, and its output value is passed through the carrier phase shift controller to obtain the phase shift angle α, that is, the carrier V saw2 Phase lags behind the basic digital carrier V saw At angle α, the converter switches Q3 and Q4 lag behind switches Q1 and Q2, so that part of the power is not transferred to the secondary side, thereby achieving ultra-wide output voltage regulation.

[0047] When a given voltage V o_ref When the voltage is not less than 300V, the converter will be in high-voltage mode, using a phase-shift control strategy for the primary side and a superimposed asymmetric PWM and phase-shift control strategy for the secondary side. The phase-shift control strategy adopted for the primary side in high-voltage mode is as follows: the switch tubes Q3 and Q4 lag behind the upper switch tubes Q1 and Q2 by a fixed angle of 180°, and the two switch tubes alternately provide energy to the secondary side. Combined with Figure 2(a) and Figure 2(b), under the superimposed asymmetric PWM and phase-shift control strategy, the moment when the secondary side current reaches zero is the turn-off moment of Q5 and Q6, so a zero-crossing detection circuit can be added to control the turn-off of Q5 and Q6, which can suppress the voltage spike of the secondary side voltage. The turn-on moment of Q5 and Q6 is determined according to the output voltage V o To adjust the output voltage V o With a given voltage V o_ref The difference is fed into the voltage regulator and compared with the carrier V saw After comparison, the duty cycle is D y1The control signals S5' and S6' are controlled by the control signals S5' and S6', and the control signal S6' lags behind the control signal S5' by 180 degrees. Under the control of the control signals S5' and S6', Q5 and Q6 can realize the adaptive working mode switching between the secondary side full-bridge rectification and the voltage doubling rectification. By controlling the conduction of Q5 and Q6 by S5' and S6' and the natural freewheeling conduction of the body diodes of Q5 and Q6, the ultra-wide range output voltage regulation can be achieved. Since the positive current on the secondary side flows through the channel of Q6 and the body diode of Q5, and the negative current flows through the channel of Q5 and the body diode of Q6, the diode conduction brings about a large conduction loss; after the discharge time Δt1 of the junction capacitance of Q5 and Q6 is sent to the multiplier and combined with the duty cycle D y1 Subtracting the duty cycle to get D y2 The PWM wave S6” is phase-shifted by 180° to obtain S5”, which is used to replace the natural freewheeling conduction function of the original body diode, reducing the conduction loss. At the same time, the time difference of Δt1 reduces the switching loss. The secondary side current i collected in step 1 is converted to sec The zero-crossing time t is obtained by inputting the current zero-crossing detection circuit. zero , thereby obtaining the angle θ at which the current zero-crossing point lags behind the primary-side voltage, which is used for phase-shift control correction. The asymmetric PWM waves S5' and S5" are superimposed and then subjected to phase-shift control correction with an angle of θ to obtain the final control signal for switch tube Q5. The final control signal for switch tube Q6 is obtained by superimposing S6' and S6" and then subjected to phase-shift control correction with an angle of θ. While achieving a wide range of output voltage changes, it also reduces conduction loss and switching loss and suppresses voltage spikes.

[0048] When the converter is in low-voltage mode, the relationship between the phase shift angle α and the voltage gain G satisfies the following equation:

[0049]

[0050] Where Q is the quality factor;

[0051] When the converter is in high voltage mode, it adopts superimposed asymmetric PWM plus phase shift control mode, and the duty cycle D y1 、D y2 The relationship with Δt1 is:

[0052]

[0053] Among them, C eq is the equivalent junction capacitance of the switch tube, and ω0 is the resonant angular frequency.

[0054] The turn-off time of S5' and S6" is the same and consistent with the turn-off time of S1. The turn-off time of S6' and S5" is the same and consistent with the turn-on time of S1. The zero-crossing detection circuit is used to obtain the secondary side current zero-crossing time t zero , calculate t zeroThe difference Δt2 between the turn-off moment or the turn-on moment of S1 and the phase shift angle θ is θ=ω0·Δt2.

[0055] The control method proposed in this invention combines phase-shift control with superimposed asymmetric PWM summed phase-shift control. The primary side controls the phase-shift angle, while the secondary side controls the duty cycle and phase of the fifth and sixth switches Q5 and Q6, achieving high-efficiency voltage regulation. When the converter operates in low-voltage regulation mode, the control signals for Q1 and Q2 are phase-shifted to obtain the control signals for Q3 and Q4. The fifth and sixth switches Q5 and Q6 are not controlled. This operation is similar to that of a phase-shifted full-bridge converter, with a normalized voltage gain of 0 to 1. When the converter operates in high-voltage regulation mode, the control signals for Q1 and Q2 are phase-shifted by 180° to obtain the control signals for Q3 and Q4. The duty cycles of the fifth and sixth switches Q5 and Q6 are then adjusted and phase-corrected. When the duty cycle is 0, the normalized gain is 1, and when the duty cycle is 1, the normalized gain is 2. The control method proposed in the present invention can suppress voltage spikes and achieve continuous adjustment of normalized gain within the range of 1 to 2 by adjusting the duty cycle and phase of the superimposed asymmetric PWM without adjusting the frequency.

[0056] FIG5(a) shows the relationship curve between the voltage gain and the phase shift angle α of the converter proposed in the present invention in the low-voltage mode, and FIG5(b) shows the relationship curve between the voltage gain and the duty cycle D of the converter proposed in the present invention in the high-voltage mode. y1 In low-voltage mode, after writing out the time-domain equation, it is found that it is consistent with the time-domain equation of the full-bridge LLC controlled by phase shift, which can be used for plotting analysis to obtain Figure 5(a); in high-voltage mode, according to Figure 3 The key waveforms and equivalent circuits are listed in the time domain, and solved and plotted using MATLAB software to obtain Figure 5(b). The resulting curve shows weak load dependence, which is more conducive to stable voltage range regulation. Switching between low-voltage and high-voltage modes significantly broadens the output voltage range.

[0057] The following is a simulation example of the present invention. The main parameters of the converter are as follows: input voltage 380V, output voltage 5-500V, and transformer ratio 38:30.

[0058] FIG6(a) shows the control signal S1 of the first switch tube, the control signal S5 of the fifth switch tube, and the control signal S6 of the first switch tube, as well as their voltage waveforms v under the working conditions of an output voltage of 300V and an output power of 3000W in one embodiment of the present invention. ds6 , at this time the duty cycle D y1The sixth switch tube Q6 realizes half-cycle soft switching; FIG6( b ) shows the control signal S1 of the first switch tube, the control signal S5 of the fifth switch tube and the secondary side voltage V under the working conditions of the output voltage of 300V and the output power of 3000W in one embodiment of the present invention. sec And the resonant current i Lr1 , Excitation current waveform i Lm1 , the phase-shift control-corrected Δt2 realizes the secondary side voltage spike suppression, and the waveform is completely consistent with the theoretical analysis.

[0059] FIG7(a) shows the control signal S1 of the first switch tube, the control signal S5 of the fifth switch tube, and the control signal S6 of the first switch tube, as well as their voltage waveforms v under the working conditions of an output voltage of 500V and an output power of 3000W in one embodiment of the present invention. ds6 , at this time the duty cycle D y1 The voltage gain curve is consistent with the analysis of the voltage gain curve. The sixth switch Q6 realizes half-cycle soft switching and optimizes the conduction loss. FIG7( b ) shows the control signal S1 of the first switch, the control signal S5 of the fifth switch and the secondary voltage V under the working conditions of the output voltage of 500 V and the output power of 3000 W in one embodiment of the present invention. sec And the resonant current i Lr1 , Excitation current waveform i Lm1 , the phase-shift control-corrected Δt2 realizes the secondary side voltage spike suppression, and the waveform is completely consistent with the theoretical analysis.

[0060] The above embodiments are only for illustrating the technical idea of ​​the present invention and cannot be used to limit the protection scope of the present invention. Any changes made on the basis of the technical solution in accordance with the technical idea proposed by the present invention shall fall within the protection scope of the present invention.

Claims

1. Hybrid rectifier bridge LLC resonant converter with wide output voltage range, characterized in that: include: A primary side half-bridge inverter network includes a first half-bridge inverter unit and a second half-bridge inverter unit, wherein the first half-bridge inverter unit and the second half-bridge inverter unit are respectively connected in parallel with an input DC power supply, wherein the first half-bridge inverter unit is composed of a first switch tube and a second switch tube connected in series, and the second half-bridge inverter unit is composed of a third switch tube and a fourth switch tube connected in series; A resonant network, comprising a first resonant tank and a second resonant tank, wherein the first resonant tank is connected to an output end of a first half-bridge inverter unit, and the second resonant tank is connected to an output end of a second half-bridge inverter unit; A high-frequency transformer group, comprising an inverting high-frequency transformer and a positive-phase high-frequency transformer, wherein the primary winding of the inverting high-frequency transformer is connected in parallel with the excitation inductance in the first resonant slot, the primary winding of the positive-phase high-frequency transformer is connected in parallel with the excitation inductance in the second resonant slot, and the secondary winding of the inverting high-frequency transformer is connected in positive series with the secondary winding of the positive-phase high-frequency transformer; and, The secondary side hybrid rectification network includes a full-bridge rectification network and a pair of bidirectional switches, wherein the pair of bidirectional switch tubes include a fifth switch tube and a sixth switch tube connected with a common source, the positive polarity end of the secondary winding of the inverting high-frequency transformer and the secondary winding of the positive-phase high-frequency transformer after being connected in series in a forward manner is connected to the midpoint of the first bridge arm of the full-bridge rectification network, the negative polarity end of the secondary winding of the inverting high-frequency transformer and the secondary winding of the positive-phase high-frequency transformer after being connected in series in a forward manner and the drain of the fifth switch tube are connected to the midpoint of the second bridge arm of the full-bridge rectification network, and the drain of the sixth switch tube is connected to the filter output unit.

2. The hybrid rectifier bridge LLC resonant converter with a wide output voltage range according to claim 1, characterized in that: In the first half-bridge inverter unit, the drain of the first switch tube is connected to the positive electrode of the input DC power supply, the drain of the second switch tube is connected to the source of the first switch tube, the source of the second switch tube is connected to the negative electrode of the input DC power supply, and the drain and source of the second switch tube constitute the output end of the first half-bridge inverter unit; In the second half-bridge inverter unit, the drain of the third switch tube is connected to the positive electrode of the input DC power supply, the drain of the fourth switch tube is connected to the source of the third switch tube, the source of the fourth switch tube is connected to the negative electrode of the input DC power supply, and the drain and source of the fourth switch tube constitute the output end of the second half-bridge inverter unit.

3. The hybrid rectifier bridge LLC resonant converter with a wide output voltage range according to claim 2, characterized in that: The first resonant tank comprises: a first resonant inductor, a first excitation inductor and a first resonant capacitor, one end of the first resonant inductor is connected to the source of the first switch tube and the drain of the second switch tube, the other end of the first resonant inductor is connected to one end of the first excitation inductor, the other end of the first excitation inductor is connected to the positive electrode of the first resonant capacitor, the negative electrode of the first resonant capacitor is connected to the source of the second switch tube, and the primary excitation inductor of the inverting high-frequency transformer serves as the first excitation inductor; The second resonant tank includes: a second resonant inductor, a second excitation inductor and a second resonant capacitor. One end of the second resonant inductor is connected to the source of the third switch tube and the drain of the fourth switch tube. The other end of the second resonant inductor is connected to one end of the second excitation inductor. The other end of the second excitation inductor is connected to the positive electrode of the second resonant capacitor. The negative electrode of the second resonant capacitor is connected to the source of the fourth switch tube. The primary excitation inductor of the positive phase high-frequency transformer serves as the second excitation inductor.

4. The hybrid rectifier bridge LLC resonant converter with a wide output voltage range according to claim 3, characterized in that: The first end of the primary winding of the inverting high-frequency transformer is connected to the other end of the first resonant inductor and one end of the first excitation inductor, and the second end of the primary winding of the inverting high-frequency transformer is connected to the positive electrode of the first resonant capacitor; The first end of the primary winding of the positive phase high frequency transformer is connected to the other end of the second resonant inductor and one end of the second excitation inductor, and the second end of the primary winding of the positive phase high frequency transformer is connected to the positive electrode of the second resonant capacitor.

5. The hybrid rectifier bridge LLC resonant converter with a wide output voltage range according to claim 4, characterized in that: The full-bridge rectifier network comprises: first to fourth diodes, wherein the cathode of the first diode is connected to the cathode of the third diode, the cathode of the second diode is connected to the anode of the first diode to form a midpoint of a first bridge arm of the full-bridge rectifier network, the cathode of the fourth diode is connected to the anode of the third diode to form a midpoint of a second bridge arm of the full-bridge rectifier network, and the anode of the second diode is connected to the anode of the fourth diode; The first end of the secondary winding of the inverting high-frequency transformer is connected to the midpoint of the first bridge arm of the full-bridge rectifier network as a positive polarity end, the first end of the secondary winding of the positive-phase high-frequency transformer is connected to the second end of the secondary winding of the inverting high-frequency transformer, the second end of the secondary winding of the positive-phase high-frequency transformer is connected to the midpoint of the second bridge arm of the full-bridge rectifier network as a negative polarity end, the first end of the secondary winding of the inverting high-frequency transformer and the second end of the primary winding of the inverting high-frequency transformer are the same-name ends, and the first end of the secondary winding of the positive-phase high-frequency transformer and the first end of the primary winding of the positive-phase high-frequency transformer are the same-name ends.

6. The control method of the hybrid rectifier bridge LLC resonant converter with a wide output voltage range according to any one of claims 1 to 5, characterized in that: When the converter operates in low voltage mode, a phase shift control strategy is adopted on the primary side; When the converter operates in a high-voltage mode, a phase-shift control strategy is adopted on the primary side, and a superimposed asymmetric PWM and phase-shift control strategy is adopted on the secondary side.

7. The control method according to claim 6, characterized in that: When the converter operates in low-voltage mode, a phase-shift control strategy is adopted on the primary side, specifically: voltage regulation is performed on the difference between the converter output voltage sampling value and the expected voltage to obtain a phase shift angle α, and control signals of the first switch tube and the second switch tube are respectively phase-shifted by a phase shift angle of α to obtain control signals of the third switch tube and the fourth switch tube.

8. The control method according to claim 7, characterized in that: When the converter operates in the high-voltage mode, a phase shift control strategy is adopted for the primary side, specifically: the control signals of the first switch tube and the second switch tube are respectively adjusted by 180° phase shift to obtain the control signals of the third switch tube and the fourth switch tube.

9. The control method according to claim 8, characterized in that: When the converter operates in high voltage mode, a superimposed asymmetric PWM and phase shift control strategy is adopted on the secondary side, specifically: After voltage regulation is performed on the difference between the converter output voltage sampling value and the expected voltage, a first duty cycle D is obtained. y1 The PWM wave, the first duty cycle is D y1 The PWM wave of the fifth switch tube is the first PWM wave of the fifth switch tube, and the first PWM wave of the fifth switch tube is adjusted by 180° phase shift to obtain the first PWM wave of the sixth switch tube; According to the critical condition of the soft switching of the fifth switch tube and the sixth switch tube and the first duty cycle D y1 , get the second duty cycle as D y2 The PWM wave, the second duty cycle is D y2 The PWM wave is the second PWM wave of the fifth switch tube, and the second PWM wave of the fifth switch tube is phase-shifted by 180° to obtain the second PWM wave of the sixth switch tube; According to the zero-crossing time of the secondary-side current of the converter, the angle θ of the zero-crossing point of the secondary-side current lagging behind the primary-side voltage is calculated; According to the angle θ at which the zero-crossing point of the secondary current lags behind the primary voltage, a phase shift control correction is performed on the waveform obtained by superimposing the first PWM wave and the second PWM wave of the fifth switch tube to obtain a control signal of the fifth switch tube; According to the angle θ at which the zero-crossing point of the secondary current lags behind the primary voltage, a waveform obtained by superimposing the first PWM wave and the second PWM wave of the sixth switch tube is corrected by phase shift control to obtain a control signal of the sixth switch tube.

10. The control method according to claim 9, characterized in that: The phase shift angle α and the voltage gain G satisfy the relationship The critical condition for the soft switching of the fifth switch tube and the sixth switch tube is The second duty cycle D y2 for The angle θ at which the secondary current zero-crossing point lags behind the primary voltage is θ=ω0·Δt2, where Q is the quality factor, i sec is the secondary current, Δt1 is the discharge time of the junction capacitance of the fifth switch tube and the sixth switch tube, C eq is the equivalent junction capacitance of the switch tube, V sec is the secondary voltage, V O is the converter output voltage, ω0 is the resonant angular frequency, and Δt2 is the time it takes for the secondary current to naturally reach zero after the primary switch tube is switched.

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