Power conversion device and motor control system

Abstract

The invention provides a power conversion device and a motor control system which can realize stable speed control even if the speed response is improved. The present invention is provided with a power converter (2) that converts DC power into AC power on the basis of AC voltage command values (vu *, vv *, vw *) in stationary coordinates and outputs the AC power to an induction motor, and a controller (CT) that calculates the voltage command value in stationary coordinates by vector control. A controller CT calculates a slip frequency command value [omega] s * * by adding a steady-state value of a slip frequency determined by a d-axis current (id), a q-axis current (iq) and a secondary time constant (T2) to a transient value determined by a differential operation using the d-axis current (id) and the q-axis current (iq).

Description

Technical Field

[0001] The present invention relates to a power conversion device and a motor control system, for example, to a control technology of an induction motor. Background Art

[0002] Patent document 1 (Japanese Patent Publication No. 2018-182989) shows a method of converting the coordinates of the command value or the detection value when driving the induction motor by speed sensorless vector control into the value on the control axis (mt axis) with the direction of the primary current and the direction 90 degrees behind it as the rotating coordinate system, and calculating the speed estimation value of the induction motor based on the converted value. In addition, in formula (6) in Patent document 1, the current command value I of the d-axis and q-axis is used. d * ,I q * , use the following formula (1) to calculate the slip frequency command value ω s * In formula (1), T2 is the quadratic time constant, T ACR It is the time constant equivalent to the delay of current control.

[0003] [Mathematical formula 1]

[0004]

[0005] Prior art literature

[0006] Patent Literature

[0007] Patent Document 1: Japanese Patent Application Publication No. 2018-182989 Summary of the invention

[0008] Problems to be solved by the invention

[0009] In the method of Patent Document 1, the speed of the induction motor is estimated using the mt coordinate which is different from the dq coordinate. When such a method is used, the torque shortage caused by the change of the primary resistance value R1 on the stator side of the induction motor according to the winding temperature can be suppressed, and the error (R1 * -R1) low sensitivity motor control. In addition, in Patent Document 1, the speed control calculation unit uses the proportional gain (Kp ASR ) and integral gain (Ki ASR ) to calculate the q-axis current command value (I q * ). The proportional gain and integral gain determine the response performance of the speed control system.

[0010] On the other hand, when controlling an induction motor, it is required to quickly adjust the speed of the induction motor without error with respect to the speed command value. That is, high-precision and high-response speed control is required. However, in the method of Patent Document 1, when the response of the speed control system is made higher than the value equivalent to the inverse of the secondary time constant in the induction motor, the speed control becomes unstable due to the change of the magnetic flux in the mt coordinate, and there is a risk of vibration in the motor speed and torque.

[0011] The present invention has been made in view of the above circumstances, and one object of the present invention is to provide a power conversion device and a motor control system that can realize stable speed control even when the speed response is improved.

[0012] The above and other objects and novel features of the present invention will be explained by the description of this specification and the accompanying drawings.

[0013] Technical solutions to solve problems

[0014] The outline of typical embodiments of the invention disclosed in this application will be briefly described as follows.

[0015] A power conversion device according to one embodiment includes a power converter that converts DC power into AC power based on a voltage command value in a stationary coordinate and outputs the converted power to an induction motor, and a controller that calculates the voltage command value in the stationary coordinate by vector control. The controller calculates the slip frequency command value by adding a steady-state value of a slip frequency determined by a d-axis current, a q-axis current and a quadratic time constant to a transient value determined by a differential operation using the d-axis current and the q-axis current.

[0016] Effects of the Invention

[0017] The effects obtained by the representative embodiments of the invention disclosed in this application will be briefly described. Even if the speed response is improved, stable speed control can be achieved. BRIEF DESCRIPTION OF THE DRAWINGS

[0018] Figure 1 This is a block diagram showing a configuration example of a motor control system according to the first embodiment.

[0019] Figure 2 Yes Description Figure 1 Vector diagram of the dq coordinates in as a control coordinate.

[0020] Figure 3 Yes Description Figure 1 Vector diagram of mt coordinates as another control coordinate.

[0021] Figure 4 Yes means Figure 1A block diagram showing a detailed configuration example of the frequency estimation and phase calculation unit in FIG.

[0022] Figure 5 This is a diagram showing an example of the step response characteristics of the speed when the slip frequency command value is calculated using the calculation formula shown in Patent Document 1.

[0023] Figure 6 This is a diagram showing an example of the step response characteristics of the speed when the slip frequency command value is calculated using the calculation formula shown in Patent Document 1.

[0024] Figure 7 This is a diagram showing an example of the step response characteristics of the speed when the slip frequency command value is calculated using the calculation formula of the first embodiment.

[0025] Figure 8 This is a diagram showing an example of the step response characteristics of the speed when the slip frequency command value is calculated using the calculation formula of the first embodiment.

[0026] Fig. 9 This is a diagram for explaining an example of a method for observing the slip frequency in an induction motor.

[0027] Fig.10 Yes means Figure 1 and Figure 4 A block diagram showing an example configuration of a power conversion device shown in FIG.

[0028] Fig.11 This is a diagram showing a power conversion device according to Embodiment 2. Figure 1 A block diagram showing a detailed configuration example of the frequency estimation and phase calculation unit in FIG.

[0029] Fig.12 This is a diagram showing a power conversion device according to Embodiment 3. Figure 1 A block diagram showing a detailed configuration example of the frequency estimation and phase calculation unit in FIG.

[0030] Fig.13 This is a diagram showing a power conversion device according to a fourth embodiment. Figure 1 A block diagram showing a detailed configuration example of the frequency estimation and phase calculation unit in FIG.

[0031] Fig.14 This is a block diagram showing a configuration example of a motor control system according to the fifth embodiment.

[0032] Fig.15 Yes means Fig.14 A block diagram of a detailed configuration example of the frequency detection and phase calculation unit in FIG.

[0033] Fig.16 This is a block diagram showing a configuration example of a motor control system according to a sixth embodiment.

[0034] Fig.17 This is a block diagram showing a configuration example of a motor control system according to a seventh embodiment. DETAILED DESCRIPTION

[0035] Hereinafter, embodiments of the present invention will be described in detail based on the drawings. In addition, in all the drawings for describing the embodiments, the same reference numerals are attached to the same components in principle, and their repeated descriptions are omitted.

[0036] (Implementation Method 1)

[0037] <Overview of Motor Control System>

[0038] Figure 1 This is a block diagram showing a configuration example of a motor control system according to the first embodiment. Figure 1 The motor control system shown includes an induction motor 1 and a power conversion device 20 that controls the rotation of the induction motor 1 by outputting AC power to the induction motor 1. The power conversion device 20 controls the rotation of the induction motor 1 using vector control, here speed sensorless vector control. First, use Figure 2 and Figure 3 right Figure 1 This section describes the control axes used in vector control.

[0039] Figure 2 Yes Description Figure 1 Vector diagram of the dq coordinates in as a control coordinate. Figure 2 In FIG. 1 , the d-axis is an axis indicating the direction of the magnetic flux of the induction motor 1, and the q-axis is an axis indicating a direction orthogonal to the d-axis, specifically, an axis indicating a direction leading the d-axis by 90° (π / 2). The power conversion device 20 appropriately controls the d-axis current i flowing on the d-axis. d and the q-axis current i flowing on the q-axis q , to control the primary current i1 flowing in the stator of the induction motor 1.

[0040] The primary current i1 is composed of the d-axis current i d and q-axis current i q That is, the value of the primary current i1 is determined by the vector sum of the d-axis current i d The value of the q-axis current i q The value of is determined by equation (2). In addition, the d-axis current i d The phase angle θ between the phase of the primary current i1 φ It is determined by formula (3).

[0041] [Mathematical formula 2]

[0042]

[0043] [Mathematical formula 3]

[0044]

[0045] Figure 3 Yes Description Figure 1 The motor control system of the first embodiment performs vector control using mt coordinates in addition to the general dq coordinates, similarly to the case of Patent Document 1. Figure 3 In the equation, the t-axis is the axis indicating the direction of the primary current i1, and the m-axis is the axis indicating the direction orthogonal to the t-axis, specifically, the direction lagging behind the t-axis by 90° (π / 2). The value of the primary current i1 is calculated using the m-axis current i m The value of the t-axis current i t The value of is determined by equation (4). In equation (4), the m-axis current i m The value of is zero, the t-axis current i t The value of is equal to the value of the primary current i1. In addition, the phase angle θ between the m-axis and the d-axis mt is "θ mt =(π / 2)-θ φ ”.

[0046] [Formula 4]

[0047]

[0048] return Figure 1 , the induction motor 1 and the current due to the magnetic flux axis component, that is, the d-axis current i d The magnetic flux (φ 2d ), and the torque axis component current orthogonal to the magnetic flux axis, i.e., the q-axis current i q The mt coordinate is mainly used when estimating the rotation speed of the induction motor 1. The power conversion device 20 includes a power converter 2 connected to a DC power supply 3, a current detector 4, and a voltage command value v for the power converter 2 calculated by vector control. u * 、v v * 、v w * Controller CT.

[0049] The power converter 2 is based on the voltage command value v of the three-phase AC u * 、v v * 、v w *The DC voltage from the DC power supply 3 is converted into a three-phase AC voltage and output to the induction motor 1. The output amplitude and output frequency of the three-phase AC voltage are based on the voltage command value v u * 、v v * 、v w * The power converter 2 is implemented by, for example, a three-phase inverter circuit including switching elements such as IGBT (Insulated Gate Bipolar Transistor) and MOSFET (Metal Oxide Semiconductor Field Effect Transistor).

[0050] The current detector 4 detects the three-phase AC current i flowing in the induction motor 1. u 、i v 、i w , output three-phase current detection value i uc 、i vc 、i wc In addition, the current detector 4 can also detect the phase current of two phases, for example, the U phase and the W phase, among the three phases in the induction motor 1, according to the AC condition, that is, "i u +i v +i w =0" is represented by "i v =-(i u +i w )” calculates the line current of the V phase. The following describes the details of the controller CT.

[0051] The coordinate conversion unit 5 uses the phase estimated value θ dc The three-phase current detection value i under the stationary coordinate uc 、i vc 、i wc The current detection values ​​i of the two phases, i.e., the d-axis and q-axis, are transformed into rotating coordinates. dc 、i qc And output. Phase estimated value θ dc The phase angle θ is an estimated value between the stationary coordinate and the magnetic flux axis (d-axis) rotating with respect to the stationary coordinate. mtc The current detection values ​​i of the d-axis and q-axis dc 、i qc Transformed into the current detection value i of the m-axis and t-axis mc 、i tc And output. Phase estimation angle θ mtc is the phase angle θ between the m-axis and the d-axis mt The estimated value of .

[0052] The frequency estimation and phase calculation unit 7 uses the d-axis and q-axis current command values ​​i d * 、i q * , m-axis and t-axis current detection values ​​i mc 、i tc , voltage command value v of m-axis mc ** , d-axis secondary magnetic flux command value φ 2d * , and the circuit parameters (R1, R2', Lσ, M, L2) of the induction motor 1. Then, the frequency estimation and phase calculation unit 7 outputs the speed estimation value ω of the induction motor 1. r ^、Slip frequency command valueω s ** , output frequency command value ω1 * and the phase estimated value θ dc , as the result of the operation.

[0053] The reference phase calculation unit 8 uses the slip frequency command value ω s ** , use formula (5) to calculate and output the phase angle θ between the m-axis and the d-axis mt The estimated angle θ mtc In the formula (5), T2 is the secondary time constant of the induction motor 1, and is a value determined by the resistance component (R2) and the self-inductance component (L2) of the rotor.

[0054] [Mathematical formula 5]

[0055]

[0056] The field current / magnetic flux setting unit 9 sets and outputs a positive field current command value, that is, a d-axis current command value i d * , the secondary magnetic flux command value φ of the positive polarity d-axis 2d * , and the secondary magnetic flux command value φ of the q-axis is "0" 2q * The speed control calculation unit 10 calculates the speed command value ω for the induction motor 1 based on the speed command value ω. r * The estimated speed value ω of the induction motor 1 calculated by the frequency estimation and phase calculation unit 7 r ^ Deviation "ω r * -ω r ^", proportional integral (PI) control is used to calculate and output the current command value i of the q axis q * .

[0057] The coordinate conversion unit 11 estimates the angle θ using the phase mtc The current command values ​​i of the d-axis and q-axis are d * 、i q * Converted into the current command value i of the m-axis and t-axis m * 、i t * The coordinate conversion unit 12 uses the phase to estimate the angle θ mtc The secondary magnetic flux command values ​​φ of the d-axis and q-axis 2d * ,φ 2q * Converted into the flux command value φ of the m-axis and t-axis m * ,φ t * And output.

[0058] The mt-axis vector control calculation unit 13 calculates the magnetic flux command values ​​φ of the m-axis and t-axis based on the circuit parameters (R1, R2', Lσ, M, L2) of the induction motor 1. m * ,φ t * , current command value i m * 、i t * , current detection value i mc 、i tc , estimated rotor speed value ω r ^, and the output frequency command value ω1 to the stator * Then, the mt-axis vector control operation unit 13 outputs the voltage command values ​​v of the m-axis and the t-axis. mc ** 、v tc ** , as the result of the operation.

[0059] The coordinate conversion unit 14 estimates the angle θ using the phase mtc The voltage command values ​​v of the m-axis and t-axis are mc ** 、v tc ** Converted into voltage command values ​​v of d-axis and q-axis dc ** 、v qc ** The coordinate conversion unit 15 uses the phase estimated value θ dc The voltage command values ​​v of the d-axis and q-axis in the rotating coordinate system are dc ** 、vqc ** Transformed into the voltage command value v of the three-phase AC in the stationary coordinate u * 、v v * 、v w * And output.

[0060] Next, for Figure 1 The basic operation of the motor control system shown in the figure using speed sensorless vector control is described. First, the exciting current / magnetic flux setting unit 9 outputs a secondary magnetic flux value φ for generating a d-axis in the induction motor 1. 2d The required d-axis current command value i d * In addition, the speed control calculation unit 10 sets the speed estimated value ω r ^Following speed command valueω r * The current command value i of the q-axis is calculated according to formula (6): q * In formula (6), Kp_ASR is the proportional gain of speed control, Ki_ASR is the integral gain of speed control, and s is a complex number.

[0061] [Mathematical formula 6]

[0062]

[0063] The mt-axis vector control calculation unit 13 uses the current command values ​​i for the d-axis and q-axis. d * 、i q * The current command values ​​i of the m-axis and t-axis obtained by coordinate transformation are m * 、i t * , circuit parameters of induction motor 1 (R1, R2', Lσ, M, L2), flux command values ​​φ of the m-axis and t-axis m * ,φ t * , speed estimation value ω r ^, and the output frequency command value ω1 * , calculation formula (7). Thus, the mt-axis vector control calculation unit 13 calculates the voltage reference value v of the m-axis and the t-axis mc *、v mt *.

[0064] [Formula 7]

[0065]

[0066] In formula (7), R1 * It is the primary resistance value, that is, the resistance value of the stator. * It is the secondary resistance value converted to the primary side, that is, the rotor resistance value converted to the stator side. * is the leakage inductance value. M * is the mutual inductance value. L2 * is the self-inductance of the secondary side, i.e. the rotor. ACR is the time constant corresponding to the delay of the current control, and s is a complex number.

[0067] In addition, the mt-axis vector control calculation unit 13 makes the m-axis current detection value i mc Following current command value i m * The voltage correction value Δv of the m-axis is calculated by performing PI control according to formula (8). m * Similarly, the mt-axis vector control calculation unit 13 makes the t-axis current detection value i tc Following current command value i t * The voltage correction value Δv of the t-axis is calculated by performing PI control according to formula (8). t * In formula (8), Kp_m is the proportional gain of the current control of the m-axis, Ki_m is the integral gain of the current control of the m-axis, Kp_t is the proportional gain of the current control of the t-axis, and Ki_t is the integral gain of the current control of the t-axis.

[0068] [Mathematical formula 8]

[0069]

[0070] Then, the mt-axis vector control calculation unit 13 calculates the voltage reference values ​​v of the m-axis and t-axis calculated in equation (7) as shown in equation (9). mc * 、v mt * Add the voltage correction value Δv of the m-axis and t-axis calculated in equation (8) m * , Δv t * , to calculate the voltage command value v of the m-axis and t-axis mc ** 、v tc ** The voltage command value v of the m-axis and t-axis mc ** 、v tc **After the coordinate transformation to the d-axis and q-axis and the coordinate transformation to the u-axis, v-axis, and w-axis, it is transformed into the voltage command value v of the three-phase AC voltage. u * 、v v * 、v w * .

[0071] [Mathematical formula 9]

[0072]

[0073] <Details of frequency estimation and phase calculation unit>

[0074] Figure 4 Yes means Figure 1 A block diagram of a detailed configuration example of the frequency estimation and phase calculation unit 7 in FIG. Figure 4 The frequency estimation and phase calculation unit 7a shown includes a low-pass filter (LPF) 7a1, a slip command calculation unit 7a2, a frequency estimation calculation unit 7a3, an adding unit 7a4, and a phase estimation calculation unit 7a5.

[0075] The low-pass filter (LPF) 7a1 is shown in equation (10) by using a time constant T ACR The first-order lag transfer function "1 / (1+T ACR ·s)” makes the current command value of the q axis i q * Delay, to calculate the delayed current command value i of the q axis q * td . Time constant T ACR As described in equation (7), the value corresponding to the delay of the current control is determined. q * td By making the q-axis current command value i q * Delay time constant T ACR , and the analog ground represents the current detection value i of the q axis qc .

[0076] [Formula 10]

[0077]

[0078] The slip command calculation unit 7a2 uses the delayed current command value i of the q-axis as shown in equation (11). q * td , d-axis current command value i d *The slip frequency command value ω is calculated by using the secondary time constant T2 of the induction motor 1. s ** The slip frequency command value ω shown in formula (11) s ** The slip frequency command value ω shown in equation (1) s * Added "d / dt(tan -1 (i q * td / i d * ))” item.

[0079] [Mathematical formula 11]

[0080]

[0081] Here, the first term “i q * td / (T2i d * )”, that is, the slip frequency command value ω shown in formula (1) s * represents the steady-state value of the slip frequency command. On the other hand, the second term "d / dt(tan -1 (i q * td / i d * ))” indicates the instantaneous value of the slip frequency command, indicating Figure 2 and Figure 3 The d-axis current I d The phase angle θ between the primary current i1 φ The time differential value of .

[0082] The frequency estimation operation unit 7a3 uses the secondary magnetic flux command value φ of the d-axis as shown in equation (12). 2d * , voltage command value v of m-axis mc ** and the current detection value i mc , t-axis current detection value i tc , slip frequency command value ω s ** , output frequency command value ω1 * 、Phase estimation angle θ mtc , and the circuit parameters of the induction motor 1, to calculate the estimated speed value ω of the induction motor 1 r The circuit parameters of the induction motor 1 include R1 described in equation (7).* 、R2' * , Lσ * 、M * 、L2 * In addition, T obs It is the time constant of the filter included in the disturbance observer constituting the frequency estimation operation unit 7a3.

[0083] [Mathematical formula 12]

[0084]

[0085] Here, in equation (12), the frequency estimation operation unit 7a3 calculates the speed estimation value ω based on the command value and the detection value in the mt coordinate. r In this case, "(R1 * +R2' * )i mc ", the current detection value i mc Depend on Figure 3 It can be seen that is approximately zero. On the other hand, the speed estimation value ω is calculated based on the command value and the detection value in the dq coordinates r ^, replace the current detection value i mc The current detection value used (i dc ) is not approximately zero. Thus, by using mt coordinates instead of dq coordinates, for the primary resistance value R1 * The error sensitivity is low, and the speed estimation value ω can be calculated with high accuracy. r ^.

[0086] The adding unit 7a4 calculates the estimated speed value ω as shown in equation (13). r ^ and slip frequency command value ω s ** Add to calculate the output frequency command value ω1 * The phase estimation operation unit 7a5 calculates the output frequency command value ω1 as shown in equation (14). * The phase estimation value θ of the magnetic flux axis of the induction motor 1 is calculated by integrating dc The phase of the magnetic flux axis is estimated as θ dc The reference phase is used to perform sensorless control.

[0087] [Mathematical formula 13]

[0088] ω1 * =ω r ^+ω s ** …(13)

[0089] [Formula 14]

[0090]

[0091] <Verification results of speed control characteristics>

[0092] Figure 5 and Figure 6 This is a diagram showing an example of the step response characteristics of the speed when the slip frequency command value is calculated using the calculation formula shown in Patent Document 1. Figure 5 and Figure 6 In FIG. 1 , the characteristic when the formula (1) is used in the calculation of the slip frequency command value is shown. Figure 5 The response angular frequency ω of the speed control system is shown in ASR The characteristic is set to "1 / T2" (rad / s), which is the inverse of the secondary time constant T2 of the induction motor 1. The response angular frequency ω of the speed control system ASR It is determined by the values ​​of the proportional gain Kp_ASR and the integral gain Ki_ASR of the speed control shown in equation (6).

[0093] like Figure 5 As shown, at time point A, the speed command value ω r * When the speed changes by 3 Hz in a step-like manner, the speed command value ω r * The speed ω of the induction motor 1 changes accordingly. r Also changes. It can be seen that the speed ω r It reaches the maximum at time B, decreases temporarily at time C, and increases to time D. That is, it can be known that the speed ω of the induction motor 1 is r Vibration occurred.

[0094] On the other hand, Figure 6 The response angular frequency ω of the speed control system is shown in ASR Set to Figure 5 The characteristics of the case of a higher "5 / T2" (rad / s). By increasing the response angular frequency ω ASR , for the speed command value ω r * The change of can increase the speed of induction motor 1ω r However, Figure 6 The speed ω r and Figure 5 Compared with the case of the maximum speed, the vibration is larger from the point E when the maximum speed is reached. Then, the vibration continues until Figure 5 The situation is longer at point F.

[0095] Thus, when the method of Patent Document 1 is used, it is difficult to improve the response of the speed control system from the viewpoint of vibration, etc. The reason for this is that the magnetic flux in the dq coordinates and the slip frequency ω s In contrast, the magnetic flux in the mt coordinate is constant regardless of s Specifically, the magnetic flux φ on the d-axis 2d It is "M×i d ”, magnetic flux φ on the q axis 2q is zero. On the other hand, the magnetic flux φ on the m-axis m It is "ω s ×T2×φ t ”, magnetic flux φ on t axis t It is "M×i t -ω s ×T2×φ m ”.

[0096] That is, the magnetic flux φ in the mt coordinate m ,φ t , in detail, the secondary magnetic flux and the slip frequency ω s Accordingly, the steady state is reached after a change in time corresponding to the quadratic time constant T2. Therefore, in particular, the response angular frequency ω of the speed control system is made ASR When the degree is higher than "1 / T2", the speed control becomes unstable in the transient state when the secondary magnetic flux changes in the time equivalent to the secondary time constant T2, and the following occurs: Figure 6 Vibration shown.

[0097] Therefore, the slip command calculation unit 7a2 does not use the steady-state value "i q * td / (T2i d * )”, but uses the formula (1) composed of steady-state value and transient value “d / dt(tan -1 (i q * td / i d * ))” to calculate the slip frequency command value ω s ** That is, the slip command calculation unit 7a2 uses the delayed current command value i following the q-axis in the transient state. q * td and the current command value i of the d-axis d * The transient value of the change, here is the phase angle θ φ The differential value of is used to correct the slip frequency command value ω s** .

[0098] Figure 7 and Figure 8 : is a diagram showing an example of the step response characteristics of the speed when the slip frequency command value is calculated using the calculation formula of the first embodiment. Figure 7 and Figure 8 In FIG. 1 , the characteristic when the formula (11) is used in the calculation of the slip frequency command value is shown. Figure 7 In, with Figure 6 The case also shows that the response angular frequency ω ASR The characteristic when set to "5 / T2" (rad / s) is Figure 8 It is shown that further increasing the response angular frequency ω ASR , Characteristics when set to "10 / T2" (rad / s).

[0099] Depend on Figure 6 The characteristics shown are Figure 7 As can be seen from the comparison of the characteristics shown in FIG. 1 , the speed ω of the induction motor 1 can be suppressed or prevented by using the method of the first embodiment. r Vibration. To further increase the response angular frequency ω ASR In the case of Figure 8 As shown, the speed ω of the induction motor 1 can be suppressed or prevented. r Thus, by using the method of the first embodiment, even if the speed response is improved, the speed ω of the induction motor 1 can be increased. r Stable speed control is achieved without vibration.

[0100] <Method of observing slip frequency>

[0101] Fig. 9 This is a diagram for explaining an example of a method for observing the slip frequency in an induction motor. Fig. 9 In the power conversion device 20 driving the induction motor 1, a voltage detector 21 and a current detector 22 are installed, and an encoder 23 is installed on the shaft of the induction motor 1. In addition, a computer having a dq axis voltage / current calculation unit 24 and a slip calculation unit 25 is provided.

[0102] As a first step, the voltage detection value v of the three-phase AC detected by the voltage detector 21 is input to the dq axis voltage / current calculation unit 24. uc 、v vc 、v wc , the three-phase AC current detection value i detected by the current detector 22 uc 、i vc 、i wc, and the position θ detected by the encoder 23. The dq axis voltage / current calculation unit 24 uses these inputs to calculate the vector voltage components v of the d axis and the q axis. dc ^、v qc ^ and vector current component i dc ^、i qc ^, and then calculate the speed detection value ω by differentiating the position θ rc .

[0103] Then, as a second step, the dq-axis voltage / current calculation unit 24 uses the calculated vector voltage component v dc ^、v qc ^ and vector current component i dc ^、i qc The output frequency command value ω1^ is calculated by using equation (15) or equation (16) as well as the circuit parameters of the induction motor 1. As the circuit parameters of the induction motor 1, for example, the result of offline automatic tuning or the design value installed in the general inverter is used.

[0104] [Mathematical formula 15]

[0105]

[0106] [Formula 16]

[0107]

[0108] The slip calculation unit 25 uses the output frequency command value ω1^ and the speed detection value ω rc , use formula (17) to calculate the first slip frequency value ω s In addition, the slip calculation unit 25 calculates the second slip frequency value ω using equation (18) which is equivalent to equation (11): s Here, when the power conversion device 20 of the first embodiment is used, the first slip frequency value ω s ^ and the second slip frequency value ω s ^^Agree.

[0109] [Mathematical formula 17]

[0110] ω s ^=ω1^-ω rc …(17)

[0111] [Mathematical formula 18]

[0112]

[0113] <Schematic structure of main parts of power conversion device>

[0114] Fig.10 Yes means Figure 1and Figure 4 1 is a block diagram of a configuration example in which a main part is extracted from the power conversion device 20 shown. Fig.10 In the example, the power converter 2 is based on the voltage command value v in the stationary coordinate system. u * 、v v * 、v w * , converting DC power into AC power and outputting it to the induction motor 1. The controller CT calculates the voltage command value v under the stationary coordinate through vector control. u * 、v v * 、v w * In brief, the controller CT uses the slip command operation unit 7a2 to convert the d-axis current i d , q-axis current i q The steady-state value of the slip frequency determined by the quadratic time constant T2 is determined by using the d-axis current i d and q-axis current i q The instantaneous value determined by the differential operation is added to calculate the slip frequency command value ω s ** .

[0115] Here, in equation (11), as the d-axis current i d and q-axis current i q , using the current command value i of the d-axis d * and the delayed current command value i of the q axis q * td , but not limited to this, for example, the current detection value i of the d-axis can also be used dc and the q-axis current detection value i qc That is, ideally, it is preferable to use the current detection value i of the d-axis dc and the q-axis current detection value i qc However, the detected value may be more unstable than the command value, so the command value is used here. q * The current detection value i closer to the q axis qc , and the delayed current command value i of the q axis is used q * td .

[0116] Then, the controller CT uses the adding unit 7a4 to convert the slip frequency command value ω s **The speed value of the induction motor 1, for example, the estimated speed value ω r ^Add to calculate the output frequency command value ω1 * The speed value is obtained by detection or calculation. Figure 1 In the case of speed sensorless vector control shown in the figure, the speed value is the speed estimated value ω obtained by calculation r ^, when performing vector control with a speed sensor, the speed value is the speed detection value obtained by detection.

[0117] On the other hand, the controller CT uses the coordinate conversion unit 11 to convert the d-axis current command value i d * and the q-axis current command value i q * Perform coordinate transformation to calculate the current command value i of the m-axis m * and the current command value i of the t-axis t * In addition, the controller CT uses the coordinate conversion unit 6 to convert the current detection value i of the d-axis dc and the q-axis current detection value i qc Perform coordinate transformation to calculate the current detection value i of the m-axis mc and the current detection value i of the t-axis tc .

[0118] Then, the controller CT uses the mt-axis vector control calculation unit 13 to control the current command value i of the m-axis. m * and the current command value i of the t-axis t * , m-axis current detection value i mc and the current detection value i of the t-axis tc , and the output frequency command value ω1 * , to calculate the voltage command value v of the m-axis mc ** and the voltage command value v of the t-axis tc ** That is, the mt-axis vector control calculation unit 13 calculates equation (7). Then, the controller CT calculates the voltage command value v for the m-axis. mc ** and the voltage command value v of the t-axis tc ** , coordinate transformation units 14 and 15 are used to perform coordinate transformation in sequence, thereby calculating the voltage command value v under the stationary coordinates. u * 、v v * 、v w * .

[0119] <Main effects of embodiment 1>

[0120] In the above embodiment 1, by setting the slip frequency command value ω s ** The calculation includes transient values, and even if the speed response is improved, stable speed control without vibration can be achieved. That is, high-precision and high-response speed control can be achieved. Furthermore, by using the mt coordinate to estimate the speed of the induction motor 1, it is possible to suppress torque shortage and torque pulsation caused by the change of the primary resistance value in the induction motor 1 according to the winding temperature, and high-precision and stable speed control can be achieved.

[0121] (Implementation Method 2)

[0122] <Details of frequency estimation and phase calculation unit>

[0123] Fig.11 This is a diagram showing a power conversion device according to Embodiment 2. Figure 1 A block diagram of a detailed configuration example of the frequency estimation and phase calculation unit 7 in FIG. Fig.11 The frequency estimation and phase calculation unit 7b shown in FIG. Figure 4 Compared with the illustrated configuration example, the processing content of the slip command calculation unit 7b2 is different. Figure 4 In the case of slip frequency command value ω s ** In the calculation, it is necessary to use the current command value of the dq axis to calculate the phase angle θ φ The inverse tangent of "tan -1 ",but Fig.11 There is no need to calculate the inverse tangent "tan -1 ”.

[0124] Specifically, the slip command calculation unit 7b2 uses the delayed current command value i of the q-axis from the low-pass filter (LPF) 7a1. q * td , d-axis current command value i d * and the quadratic time constant T2, the slip frequency command value ω is calculated according to formula (19): s ** Formula (19) is the inverse tangent "tan -1 The differential operation of " is expanded to be equivalent to formula (11).

[0125] [Mathematical formula 19]

[0126]

[0127] <Main effects of Embodiment 2>

[0128] By using the method of the above-mentioned embodiment 2, the same effects as those described in the embodiment 1 can be obtained. Furthermore, in the controller CT, the slip frequency command value ω can be calculated without providing a calculation table or the like. s ** That is, to calculate the inverse tangent "tan -1 ", usually it is necessary to refer to a predetermined calculation table, etc., but in the method of implementation mode 2, there is no need to refer to such a calculation table.

[0129] (Implementation method 3)

[0130] <Details of frequency estimation and phase calculation unit>

[0131] Fig.12 This is a diagram showing a power conversion device according to Embodiment 3. Figure 1 A block diagram of a detailed configuration example of the frequency estimation and phase calculation unit 7 in FIG. Fig.12 The frequency estimation and phase calculation unit 7c shown in FIG. Figure 4 Compared with the illustrated configuration example, the processing content of the slip command calculation unit 7c2 is different. Fig.12 In, with Fig.11 The same is true for the case where the slip frequency command value ω s ** In the calculation of the inverse tangent, the inverse tangent "tan -1 " operation.

[0132] The slip command calculation unit 7c2 uses the delayed current command value i of the q-axis from the low-pass filter (LPF) 7a1. q * td , d-axis current command value i d * , secondary time constant T2, t-axis current command value i t * , calculate the slip frequency command value ω according to formula (20) s ** Formula (20) is the result of replacing “i d *2 +i q * td 2 " is replaced by "i t *2 The obtained formula is equivalent to formula (19).

[0133] [Mathematical formula 20]

[0134]

[0135] <Main effects of implementation method 3>

[0136] By using the method of the above embodiment 3, the same effects as those described in the embodiments 1 and 2 can be obtained. d *2 +i q * td 2 " is replaced by "i t *2 ", compared with the case of implementation mode 2, the calculation can be simplified.

[0137] (Implementation 4)

[0138] <Details of frequency estimation and phase calculation unit>

[0139] Fig.13 This is a diagram showing a power conversion device according to a fourth embodiment. Figure 1 A block diagram of a detailed configuration example of the frequency estimation and phase calculation unit 7 in FIG. Fig.13 The frequency estimation and phase calculation unit 7d shown in FIG. Figure 4 Compared with the illustrated configuration example, the processing content of the slip command calculation unit 7d2 is different. Fig.13 In, with Fig.11 The same is true for the case where the slip frequency command value ω s ** In the calculation of , there is no need to calculate the inverse tangent "tan -1 ”.

[0140] The slip command calculation unit 7d2 uses the delayed current command value i of the q-axis from the low-pass filter (LPF) 7a1. q * td , d-axis current command value i d * , quadratic time constant T2, calculate the slip frequency command value ω according to formula (21) s ** Formula (21) is the result of replacing the “tan -1 (i q * td / i d * )" is approximately equal to "i q * td / i d * ” Got it.

[0141] [Mathematical formula 21]

[0142]

[0143] <Main effects of embodiment 4>

[0144] By using the above embodiment 4, the same effects as those described in embodiments 1 and 2 can be obtained. In addition, the calculation can be further simplified compared with the case of embodiment 3. However, from the viewpoint of calculation accuracy, it is preferable to use the method of embodiments 1, 2, and 3.

[0145] (Implementation method 5)

[0146] <Overview of Motor Control System>

[0147] Fig.14 This is a block diagram showing a configuration example of a motor control system according to the fifth embodiment. Figure 1 Speed ​​sensorless control is used in Fig.14 Speed ​​sensor control is used. Therefore, Fig.14 In the embodiment, an encoder 1a is provided in the induction motor 1. The encoder 1a detects the phase of the induction motor 1 and outputs an encoder signal θ indicating the detection result. enc .in addition, Fig.14 In place of Figure 1 The frequency estimation and phase calculation unit 7 in the embodiment is provided with a frequency detection and phase calculation unit 7e. The frequency detection and phase calculation unit 7e is based on the encoder signal θ enc The speed value of the induction motor 1 is detected.

[0148] <Details of frequency estimation and phase calculation unit>

[0149] Fig.15 Yes means Fig.14 A block diagram of a detailed configuration example of the frequency detection and phase calculation unit 7e. Fig.15 In the frequency detection and phase calculation unit 7e shown in FIG. Figure 4 Compared with the configuration example shown in FIG. 1 , a frequency detection unit 7e3 is provided in place of the frequency estimation operation unit 7a3, and a holding circuit 7e6 is further provided. The holding circuit 7e6 holds the encoder signal θ in the current detection cycle. enc [n], and output the encoder signal θ in the previous detection cycle enc [n-1].

[0150] The frequency detection unit 7e3 uses the encoder signal θ in the current detection cycle enc [n] and the encoder signal θ in the previous detection cycle enc [n-1], the speed of the induction motor 1 is detected as the speed detection value ω according to formula (22) rc In formula (22), t smpIt is the value of the speed detection cycle. Speed ​​detection value ω rc Used to replace Figure 1 , Figure 4 The estimated speed value ω shown in the figure r ^.

[0151] [Mathematical formula 22]

[0152] ω rc =(θ enc [n]-θ enc [n-1]) / t smp …(twenty two)

[0153] <Main effects of embodiment 5>

[0154] By using the method of the above-mentioned embodiment 5, the same effects as the various effects described in embodiment 1 can be obtained. That is, not limited to speed sensorless control, the same effects can be obtained when using speed sensor control. In addition, the method of embodiment 5 can of course also be combined with the methods of embodiments 2, 3, and 4.

[0155] (Implementation method 6)

[0156] <Overview of Motor Control System>

[0157] Fig.16 : is a block diagram showing a configuration example of a motor control system according to Embodiment 6. For example, Figure 1 In the configuration example of, for example, the controller CT uses a method in which the circuit parameters of the induction motor 1 are fixedly set in advance in a microcontroller or the like. Fig.16 In the example, an IoT controller 16 implemented by a cloud computer is provided as a superior control device.

[0158] The IoT controller 16 is connected to the power conversion device including the controller CT via a communication line to control the power conversion device. Specifically, the IoT controller 16 inputs the control state quantity calculated in the controller CT via the communication line, calculates the circuit parameters through machine learning, and resets the calculated circuit parameters to the controller CT via the communication line.

[0159] exist Fig.16 In the figure, it is briefly shown Fig.10 The IoT controller 16 inputs, for example, voltage command values ​​v of the m-axis and the t-axis. mc ** 、v tc ** and the current detection value i mc 、i tc , speed estimation value ω r ^, and the slip frequency command value ωs ** , as the state quantity of control calculated in the controller CT. The IoT controller 16 performs machine learning using these inputs, and sequentially corrects the circuit parameters (R1) of the induction motor 1 in a manner that satisfies the state equation of the induction motor 1 equivalent to equation (7), for example. * 、R2' * , Lσ * 、M * 、L2 * ). Then, the IoT controller 16 performs feedback control on the controller CT by resetting the corrected circuit parameters to the mt-axis vector control operation unit 13 and the like.

[0160] <Main effects of embodiment 6>

[0161] By using the method of the sixth embodiment, the same effects as those described in the first embodiment can be obtained. Furthermore, the circuit parameters of the induction motor 1 can be determined more accurately by machine learning, so that the speed control accuracy of the induction motor 1 can be further improved.

[0162] (Implementation 7)

[0163] <Overview of Motor Control System>

[0164] Fig.17 This is a block diagram showing a configuration example of a motor control system according to a seventh embodiment. Fig.17 In the embodiment, the induction motor 1 drives industrial equipment such as a fan, a pump, a crane, etc. The power conversion device 20 is composed of a body having a controller CT, a power converter 2, and a digital operator 20b as a user interface. Fig.10 The controller CT shown has been added Figure 1 The configuration of the speed control calculation unit 10 is shown. The controller CT is implemented by, for example, a microcontroller or a programmable logic controller.

[0165] Specifically, the controller CT is mainly realized by a processor built into a microcontroller or a programmable logic controller executing a program stored in a built-in memory. However, the controller CT may also be realized by using an FPGA (Field Programmable Gate Array) or an ASIC (Application Specific Integrated Circuit).

[0166] The power converter 2 is implemented, for example, by a three-phase inverter circuit including switching elements such as MOSFET or IGBT. The switching element may be a Si (silicon) semiconductor element, or a wide-bandgap semiconductor element such as SiC (silicon carbide) or GaN (gallium nitride). Figure 1 The current detector 4 shown is realized by, for example, a resistor element or a current transformer for current detection and a combination of an analog-to-digital converter for converting the detected value into a digital value.

[0167] In addition, Fig.17 2, a user terminal such as a personal computer 26, a tablet computer 27, and a smartphone 28 is shown. The user can use such a user terminal or a digital operator 20b of the power conversion device 20 to set various parameters for the controller CT. For example, the power conversion device 20 and such a user terminal are directly connected via a local area network or the like, or via a digital operator such as a digital operator 20b. Fig.16 The IoT controller 16 is shown connected indirectly.

[0168] The user can set the control parameters and circuit parameters of the induction motor 1 in the controller CT via the user terminal and the local area network. In this example, the user instructs the speed control operation unit 10 in the controller CT via the user terminal to set the response angular frequency ω of the speed control system. ASR , that is, the proportional gain Kp_ASR, the integral gain Ki_ASR and the speed command value ω for the induction motor 1 of the speed control shown in equation (6) r * wait.

[0169] In addition, Fig.17 In the embodiment 1, the structure based on the embodiment 1 is shown, but it can also be a structure based on any one of the embodiments 2 to 6. In the embodiments 1 to 6, as shown in equations (11), (19) to (21), when the slip frequency command value ω s ** In the calculation of , the current command value i of the d-axis is used d * and the delayed current command value i of the q axis q * td However, the current detection values ​​i of the d-axis and q-axis can also be used dc 、i qc .

[0170] In the first to sixth embodiments, as shown in equation (8), the current command values ​​i of the m-axis and the t-axis are used. m * 、i t * and the current detection value i mc 、i tcCalculate the voltage correction value Δv m * , Δv t * Then, as shown in equation (9), by mc * 、v tc * and voltage correction value Δv m * , Δv t * Add to calculate the voltage command value v of the m-axis and t-axis mc ** 、v tc ** .

[0171] Alternatively, the current command values ​​i of the m-axis and t-axis can be used. m * 、i t * and the current detection value i mc 、i tc , calculate the voltage command value v of the m-axis and t-axis according to formula (23) mc *** 、v tc *** In formula (23), Kp_m1 is the proportional gain of the current control of the m-axis, Ki_m1 is the integral gain of the current control of the m-axis, Kp_t1 is the proportional gain of the current control of the t-axis, and Ki_t1 is the integral gain of the current control of the t-axis.

[0172] [Mathematical formula 23]

[0173]

[0174] Alternatively, the current command values ​​i of the m-axis and t-axis may be used. m * 、i t * , and the current detection value i mc 、i tc , calculate the intermediate current command value i of the m-axis and t-axis according to formula (24) m ** 、i t ** In formula (24), Kp_m2 is the proportional gain of the current control of the m-axis, Ki_m2 is the integral gain of the current control of the m-axis, Kp_t2 is the proportional gain of the current control of the t-axis, and Ki_t2 is the integral gain of the current control of the t-axis.

[0175] [Mathematical formula 24]

[0176]

[0177] When using equation (24), the speed estimated value ω is further used r ^, output frequency command value ω1 * , the flux command value φ of the m-axis and t-axis m * ,φ t * , and the circuit parameters of the induction motor 1 (R * , Lσ * 、M * 、L2 * ), calculate the voltage command value v of the m-axis and t-axis according to formula (25) mc **** 、v tc **** In formula (25), R * Reference formula (7) is "R1 * +R2' * ”,T ACR It is the time constant equivalent to the delay of current control.

[0178] [Mathematical formula 25]

[0179]

[0180] Alternatively, the current command values ​​i of the m-axis and t-axis may be used. m * 、i t * And the current detection value i mc 、i tc , calculate the voltage correction value Δv of the proportional operation component of the m-axis according to formula (26): mp * And the voltage correction value Δv of the integral operation component mi * , and the voltage correction value Δv of the proportional operation component of the t-axis tp * And the voltage correction value Δv of the integral operation component ti * In formula (26), Kp_m3 is the proportional gain of the current control of the m-axis, Ki_m3 is the integral gain of the current control of the m-axis, Kp_t3 is the proportional gain of the current control of the t-axis, and Ki_t3 is the integral gain of the current control of the t-axis.

[0181] [Mathematical formula 26]

[0182]

[0183] When using equation (26), the speed estimated value ω is further used r ^, output frequency command value ω1 * , the flux command value φ of the m-axis and t-axis m * ,φ t * , and the circuit parameters of the induction motor 1, the voltage command values ​​v of the m-axis and t-axis are calculated according to formula (27): mc ***** 、v tc ***** In formula (27), the circuit parameters are the same as those in formula (25).

[0184] [Mathematical formula 27]

[0185]

[0186] <Main effects of embodiment 7>

[0187] The same effects as those described in the first embodiment and the like can also be obtained by using the method of the seventh embodiment. In particular, the response angular frequency ω of the speed control system can be changed by the user according to the application of the induction motor 1. ASR Even in the case of , stable speed control without vibration can be achieved.

[0188] The invention obtained by the inventor is specifically described above based on the embodiments, but the present invention is not limited to the above embodiments, and various changes can be made within the scope of the main purpose. For example, the above embodiments are described in detail to explain the present invention in an easy-to-understand manner, and are not limited to all the structures that must be described. In addition, a part of the structure of a certain embodiment can be replaced with the structure of other embodiments, and the structure of other embodiments can be added to the structure of a certain embodiment. In addition, for a part of the structure of each embodiment, other structures can be added, deleted, or replaced.

[0189] Description of Reference Numerals

[0190] 1…induction motor, 2…power converter, 3…DC power supply, 4…current detector, 5…coordinate transformation unit, 6…coordinate transformation unit, 7, 7a to 7d…frequency estimation and phase calculation unit, 7a1…low-pass filter, 7a2 to 7d2…slip command calculation unit, 7e…frequency detection and phase calculation unit, 8…reference phase calculation unit, 9…field current / flux setting unit, 10…speed control calculation unit, 11…coordinate transformation unit, 12…coordinate transformation unit, 13…mt-axis vector control calculation unit, 14…coordinate transformation unit, 15…coordinate transformation unit, 16…IoT controller, 20…power conversion device, 23…encoder, CT…controller, i d* ...d-axis current command value, i q * ...q-axis current command value, i q * td ...delayed current command value of the q-axis, i dc ...current detection value of d-axis, i qc ...q-axis current detection value, v dc ** ...voltage command value of the d-axis, v qc ** ...q-axis voltage command value, i m * ...current command value of the m-axis, i t * ...t-axis current command value, i mc ...current detection value of the m-axis, i tc ...current detection value of t axis, v mc ** ...voltage command value of the m-axis, v tc ** ...t-axis voltage command value, ω r * ...speed command value, ω r ^…Speed ​​estimation value, ω rc ...speed detection value, ω s ** ...slip frequency command value, ω1 * ...output frequency command value, ω ASR …the response angular frequency of the speed control system, v u * 、v v * 、v w * ...three-phase AC voltage command value, i uc 、i vc 、i wc …Three-phase current detection values.

Claims

1. A power conversion device, characterized in that: include: A power converter that converts direct current power into alternating current power based on a voltage command value in a stationary coordinate system and outputs the converted power to the induction motor; and A controller, which calculates the voltage command value under the stationary coordinates through vector control, the controller, The invention further comprises the steps of: calculating a slip frequency command value by adding a steady-state value of a slip frequency determined by a d-axis current, a q-axis current, and a quadratic time constant to a transient value determined by a differential operation using the d-axis current and the q-axis current, using a rotating coordinate system, i.e., a dq coordinate system, in which an axis representing a magnetic flux direction of the induction motor is a d-axis system and an axis representing a direction orthogonal to the d-axis system is a q-axis system. The output frequency command value is calculated by adding the slip frequency command value to the detected or calculated speed value of the induction motor, Using mt coordinates in which the axis representing the direction of the primary current is the t-axis and the axis representing the direction orthogonal to the t-axis is the m-axis, coordinate transformation is performed on the current command value of the d-axis and the current command value of the q-axis to calculate the current command value of the m-axis and the current command value of the t-axis, and coordinate transformation is performed on the current detection value of the d-axis and the current detection value of the q-axis to calculate the current detection value of the m-axis and the current detection value of the t-axis, The m-axis voltage command value and the t-axis voltage command value are calculated based on the m-axis current command value and the t-axis current command value, the m-axis current detection value and the t-axis current detection value, and the output frequency command value, and the m-axis voltage command value and the t-axis voltage command value are transformed into coordinates to calculate the voltage command value under the stationary coordinates.

2. The power conversion device according to claim 1, characterized in that: The controller uses a low-pass filter to delay the current command value of the q-axis to calculate the delayed current command value of the q-axis, and sets the current command value of the d-axis to be "i d * ", the delayed current command value of the q-axis is use To calculate the transient value of the slip frequency command value.

3. The power conversion device according to claim 1, wherein: The controller uses a low-pass filter to delay the current command value of the q-axis to calculate the delayed current command value of the q-axis, and sets the current command value of the d-axis to be "i d * ", the delayed current command value of the q-axis is use To calculate the transient value of the slip frequency command value.

4. The power conversion device according to claim 1, wherein: The controller uses a low-pass filter to delay the current command value of the q-axis to calculate the delayed current command value of the q-axis, and sets the current command value of the d-axis to be "i d * ", the delayed current command value of the q-axis is The current command value of the t-axis is "i t * ",use To calculate the transient value of the slip frequency command value.

5. The power conversion device according to claim 1, characterized in that: The controller uses a low-pass filter to delay the current command value of the q-axis to calculate the delayed current command value of the q-axis, and sets the current command value of the d-axis to be "i d * ", the delayed current command value of the q-axis is use To calculate the transient value of the slip frequency command value.

6. The power conversion device according to claim 1, characterized in that: The controller calculates a speed value of the induction motor using the m-axis current detection value and the t-axis current detection value.

7. The power conversion device according to claim 1, characterized in that: The induction motor is provided with an encoder for detecting the phase of the induction motor. The controller detects a speed value of the induction motor based on an output signal of the encoder.

8. The power conversion device according to any one of claims 1 to 7, characterized in that: The controller can variably set the response frequency of the speed control system according to a user's instruction.

9. A motor control system, characterized in that: include: A power conversion device, comprising a power converter for converting DC power into AC power based on a voltage command value under stationary coordinates and outputting the AC power to an induction motor, and a controller for calculating the voltage command value under stationary coordinates by vector control; and a superior control device for controlling the power conversion device, the controller, The invention further comprises the steps of: calculating a slip frequency command value by adding a steady-state value of a slip frequency determined by a d-axis current, a q-axis current, and a quadratic time constant to a transient value determined by a differential operation using the d-axis current and the q-axis current, using a rotating coordinate system, i.e., a dq coordinate system, in which an axis representing a magnetic flux direction of the induction motor is a d-axis system and an axis representing a direction orthogonal to the d-axis system is a q-axis system. The output frequency command value is calculated by adding the slip frequency command value to the detected or calculated speed value of the induction motor, Using mt coordinates in which the axis representing the direction of the primary current is the t-axis and the axis representing the direction orthogonal to the t-axis is the m-axis, coordinate transformation is performed on the current command value of the d-axis and the current command value of the q-axis to calculate the current command value of the m-axis and the current command value of the t-axis, and coordinate transformation is performed on the current detection value of the d-axis and the current detection value of the q-axis to calculate the current detection value of the m-axis and the current detection value of the t-axis, calculating the m-axis voltage command value and the t-axis voltage command value based on the m-axis current command value and the t-axis current command value, the m-axis current detection value and the t-axis current detection value, and performing coordinate transformation on the m-axis voltage command value and the t-axis voltage command value to calculate the voltage command value in the stationary coordinates, The upper control device at least inputs the m-axis voltage command value and the t-axis voltage command value, as well as the m-axis current detection value and the t-axis current detection value calculated by the controller, corrects the circuit parameters of the induction motor used in the calculation within the controller by machine learning using the input values, and resets the corrected circuit parameters to the controller.

10. The motor control system according to claim 9, characterized in that: The controller uses a low-pass filter to delay the current command value of the q-axis to calculate the delayed current command value of the q-axis, and sets the current command value of the d-axis to be "i d * ", the delayed current command value of the q-axis is use To calculate the transient value of the slip frequency command value.

11. The motor control system according to claim 9 or 10, characterized in that: The controller can variably set the response frequency of the speed control system according to a user's instruction.

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