Dual-band waveguide conversion layer for non-periodic large-interval feed array

Abstract

This invention discloses a dual-band waveguide conversion layer adapted to aperiodic, large-pitch feed arrays. From the RF front-end to the feed array elements, it sequentially includes an RF interface module, a dual-band optimized chamfered waveguide module, a uniform-length trace module, and a layered arrangement module. These modules are connected to a calibration and verification module. The waveguide conversion layer is adapted for dual-band signal transmission in the 17.5GHz-20GHz transmission band and the 27.5GHz-30GHz reception band. It connects the feed array elements radially with a central distribution node as the core, and is adapted to a helical arrangement for aperiodic, large-pitch feed arrays. This invention adapts to aperiodic helical layouts, avoiding spatial conflicts between waveguides and improving assembly flexibility and engineering feasibility.

Description

Technical Field

[0001] This invention belongs to the field of microwave radio frequency technology, specifically relating to a dual-band waveguide conversion layer adapted to aperiodic large-pitch feed array. Background Technology

[0002] In a dual-band aperiodic large-pitch feed array system, the waveguide conversion network is the key hub connecting the RF transceiver module and the feed array elements. It needs to meet three core technical requirements at the same time: dual-band impedance matching, multi-channel phase consistency, and aperiodic layout adaptation. Its performance will directly affect the signal transmission efficiency and multi-beam synthesis accuracy of the entire system.

[0003] In existing technologies, dual-band waveguide conversion networks are mainly divided into two categories: discrete dual-band networks and integrated dual-band networks. Discrete dual-band networks consist of two independent single-band waveguide conversion networks operating in parallel, switching the operating frequency band via a frequency band selection switch. However, they suffer from complex structures and large sizes, making them difficult to adapt to the compact layout requirements of non-periodic arrays. Integrated dual-band networks use probe-slot composite coupling and other wideband coupling structures to achieve dual-band coverage, but in multi-channel application scenarios, it is difficult to guarantee phase consistency between channels. Impedance discontinuities at right-angle bends in waveguides of integrated dual-band networks can easily excite higher-order modes, causing a deterioration in the reflection coefficient, especially at high frequencies where signal leakage is likely to occur.

[0004] Among the existing related technical solutions, the large-scale high-stability waveguide network disclosed in CN117766969A achieves multi-port signal transmission through matching symmetrical structures and paths. However, its topology is fixed, making it difficult to adapt to feed arrays with aperiodic spiral arrangement, and it cannot simultaneously meet the impedance matching requirements of two large frequency bands, 17.5GHz-20GHz and 27.5GHz-30GHz. The low-loss ridge gap waveguide-gap waveguide miniaturized converter and feed network disclosed in CN117594968A mainly solves the design problem of feed networks with too small array spacing. Its structural optimization direction is miniaturization, but it does not comprehensively design for the phase consistency and dual-band cooperative performance of aperiodic large-gap feed arrays.

[0005] In summary, existing waveguide conversion networks have the following key drawbacks when applied to dual-band aperiodic large-spacing feed arrays: 1. Frequency band impedance mismatch: The coupling structure of traditional integrated networks is difficult to achieve good matching in both the 17.5GHz-20GHz and 27.5GHz-30GHz dual frequency bands at the same time. It often results in excellent performance in one frequency band but excessive reflection coefficient in the other frequency band. The reflection coefficient S11 is often higher than -15dB. 2. Poor phase consistency between channels: The array elements of the non-periodic array are scattered, and the waveguide paths of the existing network are of varying lengths, resulting in a phase difference between channels exceeding ±5°, which seriously affects the accuracy of multi-beam synthesis. 3. Severe reflection at bends: The discontinuity at right-angle bends in the waveguide will excite higher-order modes, leading to impedance abrupt changes and return loss greater than -15dB in both frequency bands; 4. Poor layout adaptability: The fixed topology of traditional networks is difficult to match the spiral arrangement of non-periodic feed arrays, waveguide routing is prone to spatial conflicts, and system assembly is difficult; 5. Insufficient frequency band performance synergy: The existing design does not optimize the structural parameters for the wavelength difference between the two frequency bands, resulting in the inability to simultaneously meet the design requirements for phase consistency in the low-frequency band and impedance matching in the high-frequency band.

[0006] Therefore, it is necessary to design a dual-band waveguide conversion layer that can solve the above problems simultaneously, adapt to non-periodic large-spacing feed arrays, and realize low-loss, high-isolation, and high-phase-consistency signal transmission within the dual-band. Summary of the Invention

[0007] To overcome the shortcomings of the existing technology, the present invention aims to provide a dual-band waveguide conversion layer adapted to aperiodic large-pitch feed arrays. This dual-band waveguide conversion layer achieves dual-band adaptation of 17.5GHz-20GHz and 27.5GHz-30GHz, and achieves a reflection coefficient S11≤-22dB and a phase difference between channels≤±3° across the entire frequency band. It solves the impedance matching problem at waveguide bends, ensuring that the reflection coefficient at bends within the dual-band is ≤-20dB. It adapts to aperiodic spiral layouts, avoids spatial conflicts of waveguides, and improves assembly flexibility and engineering feasibility.

[0008] To achieve the above objectives, the technical solution adopted by the present invention is as follows: A dual-band waveguide conversion layer adapted to a non-periodic large-pitch feed array includes, from the RF front end to the feed array element, an RF interface module, a dual-band optimized chamfered waveguide module, a uniform length trace module, and a layered arrangement module. The above modules are respectively connected to the calibration and verification module. The waveguide conversion layer is adapted to dual-band signal transmission of 17.5GHz-20GHz transmission frequency band and 27.5GHz-30GHz reception frequency band. It connects the feed array elements radially with the central distribution node as the core, and is adapted to the non-periodic large-spacing feed array in a spiral arrangement.

[0009] A non-periodic, large-spacing, spiral-arranged feed array is an array-type feed unit combination structure used in radio frequency microwave systems such as satellite communication and phased array radar. Its core is based on a spiral (helical) arrangement trajectory, breaking the traditional array's equal spacing and regular periodic arrangement. The spatial distance between array elements is designed with large spacing and no fixed repetition period. It is a core feed component suitable for large-aperture antennas and multi-beam combining scenarios. In this invention, the array is specifically a spiral-arranged array composed of 96 feed array elements.

[0010] The RF interface module serves as the signal input and output terminal of the dual-band waveguide conversion network, enabling waveguide interface adaptation between the network and the RF front-end and feed array elements. It adopts a standard rectangular waveguide interface and is matched with the dual-band transmission master mode TE. 10 The module provides a stable transmission access and output channel for dual-band radio frequency signals.

[0011] The dual-band optimized chamfered waveguide module is a dual-band waveguide conversion layer functional module adapted to aperiodic large-pitch helical feed arrays. It is a key structure for achieving low reflection and continuous impedance transmission in both bands at waveguide bends. The module is seamlessly integrated with the RF interface module and the uniform-length trace module. The main body consists of a waveguide substrate, dual-type chamfered structures, transition connection sections, and impedance matching surfaces. By designing differentiated chamfered parameters for the 17.5GHz-20GHz transmit frequency band (Tx band) and the 27.5GHz-30GHz receive frequency band (Rx band), and combining reactance compensation and field shaping principles, it suppresses high-order mode excitation at bends, achieving smooth transition of field distribution and impedance continuity at bends, ultimately making the reflection coefficient at bends ≤-20dB in both bands.

[0012] The dual-band optimized chamfered waveguide module designs two different chamfer parameters for 17.5GHz-20GHz (Tx band) and 27.5GHz-30GHz (Rx band), and uses the principles of reactance compensation and field shaping to suppress the excitation of higher-order modes at the bend, thereby achieving a smooth transition of field distribution and impedance continuity at the bend, making the reflection coefficient at the bend ≤-20dB in the dual-band mode. The dual-band optimized chamfered waveguide module is used to solve the dual-band impedance matching problem at the waveguide bend. Two types of chamfered structures are set at the waveguide bend: chamfer 1 at the z-direction and the xoy plane, and chamfer 2 at the xoy plane. Both chamfer 1 and chamfer 2 adopt the method of chamfering at equal distances on both sides.

[0013] The uniform length routing module ensures phase consistency between channels connecting each element of the sparse array. By unifying the total routing length of all channel waveguides in the xoy plane to 120mm and controlling the length deviation to ≤±0.1mm, the module eliminates the phase difference caused by the different physical lengths of each channel, under the premise of consistent waveguide material and cross-sectional dimensions, and achieves a phase difference between channels ≤±3° across the entire frequency band. The total length of all waveguide traces in the xoy plane is controlled to a certain length, which is determined by the number of sparse arrays, including the total length of straight segments and turning segments, and the physical length deviation of each channel waveguide is ≤ ±0.1mm.

[0014] For non-periodic spiral layouts, each channel waveguide achieves a uniform length of 120mm through a combination of "straight segments + optimized chamfered turning segments of dual-band optimized chamfered waveguide modules" and the number of turns is ≤2. Under the premise of consistent waveguide material and cross-sectional dimensions, the uniform physical length ensures that the phase change of each channel is consistent.

[0015] The layered arrangement module is adapted to the spiral arrangement of aperiodic large-spacing feed arrays. It adopts a five-layer layered structure to adapt to the aperiodic spiral layout; the interlayer waveguides maintain a safe distance of ≥5 mm.

[0016] The layer division principle of the layered layout module is: For waveguides whose upper and lower surfaces overlap in the top view, the upper surface waveguide is assigned to the upper layer and the lower surface waveguide to the lower layer; the upper surface waveguide closer to the center origin is assigned to the upper layer and the lower surface waveguide to the lower layer to avoid too many waveguides in the central area; waveguides that are close in location should be assigned to the same layer as much as possible to reduce the loss caused by the transition between layers; the upper layer waveguides should be assigned and connected first, and the lower layer waveguides should be assigned after the utilization rate of the upper layer is saturated.

[0017] The same-layer arrangement principle of the layered arrangement module is as follows: the waveguides in the same layer have the same direction, all facing the x or y direction; the waveguides in the same layer maintain the same spacing and angle to ensure consistent mutual coupling between channels; the waveguide traces in the same layer avoid the spatial areas of the upper and lower surfaces and maintain a safe distance from the upper and lower waveguides.

[0018] The calibration and verification module is used to verify the phase consistency and impedance matching performance under dual frequency bands, forming a closed-loop optimization, including a phase calibration unit and an impedance verification unit. The phase calibration unit measures the channel phase value at the characteristic frequency point of the dual-band, and the waveguide length is finely adjusted to compensate for the channel that does not meet the phase requirements. The impedance verification unit measures the reflection coefficient and standing wave ratio in the dual-band, and the chamfer parameters are finely adjusted for the parts with substandard impedance matching, so as to ensure that S11≤-22dB and the phase difference between channels≤±3 in the entire frequency band of the network.

[0019] If the phase difference between channels is greater than ±3°, the waveguide length deviation is finely adjusted with a fine adjustment amount ≤ ±0.05mm for phase compensation. The impedance verification unit measures the reflection coefficient and standing wave ratio of each waveguide in the dual-band. If the reflection coefficient S11 is greater than -18dB, the chamfer parameters of the dual-band optimized chamfer waveguide module are finely adjusted with a fine adjustment amount ≤ ±0.05mm until the reflection coefficient S11 is ≤ -22dB and the standing wave ratio is ≤ 1.3 in the entire frequency band.

[0020] The waveguide conversion layer has an insertion loss of ≤0.8dB across the entire frequency band of 17.5GHz-20GHz and 27.5GHz-30GHz, and the assembly tolerance is improved to ±1mm, thus enhancing the engineering feasibility of the system.

[0021] A dual-band waveguide conversion layer adapted to aperiodic large-gap feed array is applicable to satellite communication, radio astronomy, phased array radar and ultra-wideband imaging systems, enabling dual-band low-loss and high-phase-consistency signal transmission between the radio frequency front-end and the aperiodic large-gap feed array.

[0022] The beneficial effects of this invention are: 1. Excellent dual-band adaptability: Through differentiated chamfer parameter design and impedance matching optimization, reflection coefficient S11≤-22dB, VSWR≤1.3, and insertion loss≤0.8dB are achieved in the full frequency band of 17.5GHz-20GHz and 27.5GHz-30GHz. Compared with traditional dual-band networks, the insertion loss is reduced by more than 0.7dB and the reflection coefficient is optimized by more than 5dB, which completely solves the problem of impedance matching imbalance in traditional dual-band networks. 2. Outstanding channel phase consistency: By standardizing the 120 waveguide trace length and strictly controlling the physical length deviation to ≤±0.1mm, the phase difference between channels in the entire frequency band is ≤±3°, which is 40% higher than the existing network phase consistency and meets the multi-beam synthesis accuracy requirements of 96-channel aperiodic feed array. 3. Good matching at the turning point: The dual-band optimized chamfer structure is used to replace the traditional right-angle turn. Through reactance compensation and field shaping to suppress high-order mode excitation, the reflection coefficient at the turning point in the dual-band is ≤-22dB, which solves the problems of impedance change and severe reflection at right-angle turns. 4. Strong adaptability of aperiodic layout: It adopts a five-layer layered arrangement structure, follows the exclusive layer division and same-layer arrangement principle, perfectly matches the spiral aperiodic layout of the feed array, avoids waveguide space conflict, and improves the assembly tolerance to ±1mm, which greatly reduces the assembly difficulty and improves the feasibility of the project. 5. Good frequency band performance synergy: The chamfering parameters and routing structure are optimized to address the wavelength differences between the two frequency bands, achieving both phase consistency in the low-frequency band and impedance matching in the high-frequency band at the same time, thus solving the technical shortcomings of insufficient frequency band performance synergy in existing designs. 6. Closed-loop optimization ensures performance: A calibration and verification module is set up. Through phase calibration and impedance verification, detection and fine-tuning form a closed-loop optimization to ensure that the actual performance of the waveguide conversion layer is consistent with the design specifications, thereby improving the stability and reliability of the product. Attached Figure Description

[0023] Figure 1 This is a top view of the dual-band waveguide conversion layer adapted to the non-periodic large-spacing feed array of the present invention.

[0024] Figure 2 This is a bottom view of the dual-band waveguide conversion layer adapted to the non-periodic large-spacing feed array of the present invention.

[0025] Figure 3 This is a front view of the dual-band waveguide conversion layer adapted to the non-periodic large-spacing feed array of the present invention.

[0026] Figure 4 The left view shows the dual-band waveguide conversion layer adapted to the non-periodic large-spacing feed array of this invention.

[0027] Figure 5 This is a global diagram of the dual-band waveguide conversion layer adapted to the non-periodic large-spacing feed array of the present invention.

[0028] Figure 6 shows four corner-cutting structures of the dual-band optimized corner-cutting waveguide module of the present invention, wherein... Figure 6(a) , 6(b) 6(c) and 6(d) are chamfered structures in different orientations.

[0029] Figure 7 This is a statistical result diagram of the phase difference between waveguides in the waveguide conversion layer of the present invention.

[0030] Figure 8 This is a statistical result of the inter-waveguide phase difference in the waveguide conversion layer of this invention.

[0031] Figure 9 This is a simulation diagram (I) of the return loss of the waveguide conversion layer of the present invention in the 27.5GHz-30GHz receiving frequency band.

[0032] Figure 10 This is a simulation diagram (II) of the return loss of the waveguide conversion layer of the present invention in the 27.5GHz-30GHz receiving frequency band.

[0033] Figure 11 This is a simulation diagram (III) of the return loss of the waveguide conversion layer of the present invention in the 27.5GHz-30GHz receiving frequency band.

[0034] Figure 12 This is a simulation diagram (I) of the return loss of the waveguide conversion layer of the present invention in the 17.5GHz-20GHz transmission frequency band.

[0035] Figure 13 The simulation diagram (II) shows the return loss of the waveguide conversion layer of the present invention in the 17.5GHz-20GHz transmission frequency band.

[0036] Figure 14 The simulation diagram (IV) shows the return loss of the waveguide conversion layer of the present invention in the 27.5GHz-30GHz receiving frequency band.

[0037] Figure 15 The simulation diagram (V) shows the return loss of the waveguide conversion layer of the present invention in the 27.5GHz-30GHz receiving frequency band.

[0038] Figure 16 This is a simulation diagram (III) of the return loss of the waveguide conversion layer of the present invention in the 17.5GHz-20GHz transmission frequency band.

[0039] Figure 17 This is a simulation diagram (VI) of the return loss of the waveguide conversion layer of the present invention in the 27.5GHz-30GHz receiving frequency band.

[0040] Figure 18 This is a simulation diagram (VII) of the return loss of the waveguide conversion layer of the present invention in the 27.5GHz-30GHz receiving frequency band.

[0041] Figure 19 The simulation diagram (IV) shows the return loss of the waveguide conversion layer of the present invention in the 17.5GHz-20GHz transmission frequency band.

[0042] Figure 20 The simulation diagram (V) shows the return loss of the waveguide conversion layer of the present invention in the 17.5GHz-20GHz transmission frequency band.

[0043] Figure 21 The simulation diagram (VI) shows the return loss of the waveguide conversion layer of this invention in the 17.5GHz-20GHz transmission frequency band.

[0044] Figure 22 This is a simulation diagram (VII) of the return loss of the waveguide conversion layer of the present invention in the 17.5GHz-20GHz transmission frequency band.

[0045] Figure 23 This is a simulation diagram (VIII) of the return loss of the waveguide conversion layer of the present invention in the 17.5GHz-20GHz transmission frequency band.

[0046] Figure 24 The simulation diagram (IX) shows the return loss of the waveguide conversion layer of the present invention in the 17.5GHz-20GHz transmission frequency band.

[0047] Figure 25The simulation diagram (10) shows the return loss of the waveguide conversion layer of the present invention in the 17.5GHz-20GHz transmission frequency band.

[0048] Figure 26 The simulation diagram (XI) shows the return loss of the waveguide conversion layer of this invention in the 17.5GHz-20GHz transmission frequency band.

[0049] Figure 27 The simulation diagram (XII) shows the return loss of the waveguide conversion layer of the present invention in the 17.5GHz-20GHz transmission frequency band.

[0050] Figure 28 This is a comprehensive simulation diagram of the dual-band return loss of the waveguide conversion layer in this invention. Detailed Implementation

[0051] The present invention will now be described in further detail with reference to the accompanying drawings.

[0052] like Figure 1 As shown in Figure 6, a dual-band waveguide conversion layer adapted to a non-periodic large-pitch feed array includes, from the RF front end to the feed array element, an RF interface module, a dual-band optimized chamfered waveguide module, a uniform length trace module, a layered arrangement module, and a calibration and verification module that works in conjunction with the above modules. The waveguide conversion layer is adapted to dual-band signal transmission of 17.5GHz-20GHz transmission frequency band and 27.5GHz-30GHz reception frequency band. It connects the feed array elements radially with the central distribution node as the core, and is adapted to the spiral arrangement layout of non-periodic large-spacing feed arrays.

[0053] The RF interface module serves as the signal input and output terminal of the dual-band waveguide conversion network, enabling waveguide interface adaptation between the network and the RF front-end and feed array elements. It adopts a standard rectangular waveguide interface and is matched with the dual-band transmission master mode TE. 10 The module provides a stable transmission access and output channel for dual-band radio frequency signals; it is the fundamental hub for signals entering and leaving the conversion network.

[0054] Furthermore, the radio frequency interface module serves as the signal input and output terminal of the waveguide conversion layer, adapting to the waveguide interface of the radio frequency front-end and the feed array element. It adopts a standard rectangular waveguide interface with a length of 12.7 mm and a width of 6.35 mm.

[0055] The dual-band optimized chamfered waveguide module solves the impedance matching problem at waveguide bends in both bands. By designing two different chamfering parameters for 17.5GHz-20GHz (Tx band) and 27.5GHz-30GHz (Rx band), and utilizing the principles of reactance compensation and field shaping, it suppresses the excitation of higher-order modes at bends, achieving a smooth transition of field distribution and impedance continuity at bends, making the reflection coefficient at bends ≤-20dB in both bands; thus completely solving the problems of impedance abrupt change and severe reflection in traditional right-angle bends.

[0056] Furthermore, the dual-band optimized chamfered waveguide module is used to solve the dual-band impedance matching problem at the waveguide bend. Two types of chamfered structures are set at the waveguide bend: chamfer 1 at the z-direction and the xoy plane, and chamfer 2 at the xoy plane. Both chamfer 1 and chamfer 2 adopt the method of chamfering at equal distances on both sides.

[0057] Different chamfer parameters are adopted for the dual-band operation. Chamfer 1 has a chamfer of 6.57mm in the 17.5GHz-20GHz transmitting band and 4.26mm in the 27.5GHz-30GHz receiving band; Chamfer 2 has a chamfer of 3.21mm in the 17.5GHz-20GHz transmitting band and 2.34mm in the 27.5GHz-30GHz receiving band. The specific values ​​of these parameters can be optimized using HFSS simulation software and can fluctuate by 0.1mm.

[0058] Furthermore, the dual-band optimized chamfered waveguide module achieves dual-band impedance matching through chamfered reactance compensation and field shaping. By optimizing the chamfer distance and angle, the equivalent reactance at the bend cancels the parasitic reactance, while ensuring a smooth transition of the field distribution, suppressing high-order mode excitation, and ensuring impedance continuity within the dual-band, thereby achieving a reflection coefficient ≤-20dB at the bend within the dual-band.

[0059] The unified length routing module eliminates phase differences caused by different physical lengths of each channel by unifying the total routing length of all channel waveguides in the xoy plane to 120mm and controlling the length deviation to ≤±0.1mm, provided that the waveguide material and cross-sectional dimensions are consistent. At the same time, by optimizing the routing path and reducing the number of bends, additional phase deviations are avoided, ultimately achieving a phase difference between channels of ≤±3° across the entire frequency band, which meets the phase accuracy requirements of multi-beam synthesis.

[0060] Furthermore, the uniform length routing module is the core module that ensures phase consistency between the channels connecting the elements of the sparse array. The total length of all waveguides in the xoy plane is controlled to a certain length. This length is determined by the number of sparse arrays and includes the total length of straight segments and turning segments. The physical length deviation of each channel waveguide is ≤ ±0.1mm. For non-periodic spiral layouts, each channel waveguide achieves a uniform length of 120mm through a combination of "straight segments + optimized chamfered turning segments of dual-band optimized chamfered waveguide modules" and the number of turns is ≤2, avoiding excessive turns that introduce additional phase deviations; under the premise of consistent waveguide material and cross-sectional dimensions, the uniform physical length ensures that the phase change of each channel is consistent, achieving a phase difference between channels ≤±3° across the entire frequency band.

[0061] The layered layout module is adapted to the helical arrangement of non-periodic, large-spacing feed arrays, solving the problems of waveguide routing space conflicts and assembly difficulties caused by the fixed topology of traditional networks. By adopting a five-layer layered structure, following specific layer division and same-layer arrangement principles, the spatial layout of waveguides is rationally planned, safe distances are maintained between layers to avoid electromagnetic coupling and processing interference, and the standard routing and spacing of waveguides in the same layer ensure mutual coupling consistency. At the same time, the assembly tolerance is improved, significantly enhancing the network layout adaptability and engineering feasibility.

[0062] Furthermore, the layered arrangement module adopts a five-layer structure to adapt to non-periodic spiral layouts and avoid waveguide space conflicts; the waveguides between layers maintain a safe distance of ≥5 mm to avoid electromagnetic coupling and processing interference.

[0063] Furthermore, the layer division principle of the layered arrangement module is as follows: for waveguides whose upper and lower surfaces overlap in the top view, the upper surface waveguide is divided into the upper layer and the lower surface waveguide is divided into the lower layer; the upper surface waveguide closer to the center origin is divided into the upper layer and the lower surface waveguide is divided into the lower layer to avoid too many waveguides in the central area; waveguides that are close in position are divided into the same layer as much as possible to reduce the loss caused by the transition between layers; the upper layer waveguides are divided and connected first, and the lower layer waveguides are divided after the utilization rate of the upper layer is saturated.

[0064] Furthermore, the same-layer arrangement principle of the layered arrangement module is as follows: the waveguides in the same layer maintain the same direction, both facing the x or y direction, to reduce the risk of waveguide collision; the waveguides in the same layer maintain the same spacing and angle to ensure consistent mutual coupling between channels; the waveguide traces in the same layer avoid the spatial areas of the upper and lower surfaces and maintain a safe distance from the upper and lower waveguides.

[0065] The calibration and verification module performs performance testing, calibration, and closed-loop optimization on the dual-band waveguide conversion network to ensure that all network performance indicators meet the standards. The phase calibration unit measures the channel phase values ​​at characteristic frequencies of the dual-band network and compensates for channels that do not meet phase requirements by fine-tuning the waveguide length. The impedance verification unit measures the reflection coefficient and VSWR within the dual-band network and fine-tunes the chamfer parameters for areas with substandard impedance matching. Ultimately, this ensures that S11 ≤ -22dB and the phase difference between channels ≤ ±3° across the entire network frequency band, achieving low-loss, high-consistency signal transmission in the dual-band environment.

[0066] Furthermore, the calibration and verification module is used to verify the phase consistency and impedance matching performance under dual-band conditions, forming a closed-loop optimization. It includes a phase calibration unit and an impedance verification unit. The phase calibration unit measures the phase values ​​of each channel at six characteristic frequency points of 18GHz, 19GHz, 20GHz, 27.7GHz, 28.7GHz, and 29.7GHz using a vector network analyzer (these six frequency points are the values ​​of the two edge points and the middle point within the corresponding frequency band, and the values ​​are evenly distributed and representative). If the phase difference between channels is greater than ±3°, the waveguide length deviation is finely adjusted with a fine adjustment amount ≤ ±0.05mm for phase compensation. The impedance verification unit measures the reflection coefficient and standing wave ratio of each waveguide in the dual-band. If the reflection coefficient S11 is greater than -18dB, the chamfer parameters of the dual-band optimized chamfer waveguide module are finely adjusted with a fine adjustment amount ≤ ±0.05mm until the reflection coefficient S11 is ≤ -22dB and the standing wave ratio is ≤ 1.3 in the entire frequency band.

[0067] Furthermore, the waveguide conversion layer has an insertion loss of ≤0.8dB across the entire frequency band of 17.5GHz-20GHz and 27.5GHz-30GHz, and the assembly tolerance is improved to ±1mm, thereby enhancing the engineering feasibility of the system.

[0068] A dual-band waveguide conversion layer adapted to aperiodic large-gap feed array is applicable to satellite communication, radio astronomy, phased array radar and ultra-wideband imaging systems, enabling dual-band low-loss and high-phase-consistency signal transmission between the radio frequency front-end and the aperiodic large-gap feed array.

[0069] Example This embodiment provides a dual-band waveguide conversion layer adapted to aperiodic large-pitch feed arrays, which is used for signal connection between aperiodic large-pitch feed arrays and radio frequency front-ends in satellite communication systems, and is adapted to dual-band signal transmission in the 17.5GHz-20GHz transmission band and the 27.5GHz-30GHz reception band.

[0070] The waveguide conversion layer, from the RF front-end to the feed array elements, includes an RF interface module, a dual-band optimized chamfered waveguide module, a uniform-length trace module, a layered layout module, and a calibration and verification module. The entire layer is radiating outwards from the central distribution node, connecting 96 feed array elements. RF interface module: adopts a standard rectangular waveguide interface with a wide side of 12.7mm and a narrow side of 6.35mm, enabling docking with the RF front end and feed array elements; Unified length routing module: The total routing length of all waveguides in the xoy plane is 120mm, and the physical length deviation is controlled within ±0.1mm. The waveguides of each channel achieve uniform length through a combination of "straight segments + optimized chamfered turning segments", with 1-2 turns in each channel, ensuring that the phase difference between channels is ≤±3° across the entire frequency band. Dual-band optimized chamfered waveguide module: Chamfer 1 is set at the z-direction turning angle with the xoy plane, and chamfer 2 is set at the turning angle within the xoy plane, both of which are equidistant chamfers on both sides; the parameters of chamfer 1 are 6.57mm in the transmitting frequency band and 4.26mm in the receiving frequency band, and the parameters of chamfer 2 are 3.21mm in the transmitting frequency band and 2.34mm in the receiving frequency band. After optimization by HFSS simulation, the reflection coefficient at the turning point in the dual-band is ≤-20dB; Layered arrangement module: It adopts a five-layer layered structure, with each layer having a height of 9.02mm and an overall height of 400mm. The waveguide spacing between layers is ≥5mm. The waveguide layers are divided according to the principle of "higher layers on the upper surface and lower layers on the lower surface". The waveguides on the upper surface of the central area are all divided into higher layers. The waveguides in the same layer are all routed along the x-direction, and the spacing and corners are consistent, effectively avoiding waveguide space conflicts under aperiodic spiral layout. Calibration and verification module: The phase values ​​of six characteristic frequency points (18GHz, 19GHz, 20GHz, 27.7GHz, 28.7GHz, and 29.7GHz) are measured using a vector network analyzer. For channels with excessive phase differences, the waveguide length is fine-tuned (fine-tuning amount ≤ ±0.05mm). The reflection coefficient and standing wave ratio (SWR) are measured in both frequency bands. For waveguides with a reflection coefficient S11 > -18dB, the chamfer parameters are fine-tuned (fine-tuning amount ≤ ±0.05mm). Ultimately, S11 ≤ -22dB and SWR ≤ 1.3 are achieved across the entire frequency band.

[0071] The waveguide conversion layer in this embodiment has been tested and found to have an insertion loss of ≤0.8dB across the entire frequency band, an assembly tolerance of ±1mm, a phase difference between all 96 channels within ±3°, and a reflection coefficient of ≤-20dB at the bend. It is perfectly adapted to the spiral layout of the non-periodic large-pitch feed array, greatly reducing the assembly difficulty and enabling stable application in the engineering implementation of satellite communication systems.

[0072] The working principle of the dual-band waveguide conversion layer of this invention is as follows: After the RF signal is input from the RF interface module, it is transmitted with low reflection at the bend through the dual-band optimized chamfered waveguide module. The phase consistency of each channel is ensured by the uniform length trace module. The signal is then transmitted to the corresponding feed element through the radiation path of the layered arrangement module. The calibration and verification module monitors the phase, reflection coefficient and insertion loss of each channel in real time to ensure that the performance meets the standards in the dual-band. Specifically, the signal impedance matching is achieved in the 17.5GHz-20GHz transmission band through the chamfered transmission band parameters, and in the 27.5GHz-30GHz reception band through the chamfered reception band parameters. The uniform length trace ensures that the phase difference between the channels in the two bands meets the requirement of ≤±3°, and finally achieves high-performance signal transmission in the dual-band.

[0073] like Figure 7As shown, the phase difference values ​​of the waveguide channels corresponding to the 96 feed array elements are statistically analyzed at the characteristic frequency points of the dual frequency bands. This quantitatively reflects the phase consistency between channels and verifies the design specification of phase difference ≤ ±3°.

[0074] like Figure 8 As stated, Figure 7 In addition, the phase difference values ​​of 96 waveguide channels at different frequency points in the dual-band were further statistically analyzed to comprehensively verify the phase consistency performance between channels across the entire frequency band.

[0075] like Figure 9 As shown, the reflection coefficient (S11) of multiple waveguide channels in this frequency band varies with frequency, verifying the design specification of high-frequency return loss ≤-22dB.

[0076] like Figure 10 As shown, the return loss characteristics of different waveguide channels in the high-frequency band are further illustrated, demonstrating the overall performance of impedance matching in the high-frequency band.

[0077] like Figure 11 As shown, this further verifies the compliance of return loss in the high-frequency multi-channel band, reflecting the impedance matching effect after the chamfer optimization.

[0078] like Figure 12 As shown, the reflection coefficient of the waveguide channel in the low-frequency band varies with frequency, verifying the design specification of low-frequency return loss ≤ -22dB.

[0079] like Figure 13 As shown, the return loss characteristics of different waveguide channels in the low-frequency band are further illustrated, demonstrating the overall performance of impedance matching in the low-frequency band.

[0080] like Figure 14 As shown, we continue to verify the compliance of return loss in multiple channels of the high-frequency band, reflecting the overall high-frequency transmission performance of the network.

[0081] like Figure 15 As shown, further simulation data of return loss for different waveguide channels in the high-frequency band are provided to improve the performance verification in the high-frequency band.

[0082] like Figure 16 As shown, we continue to verify the compliance of the return loss of multiple channels in the low-frequency band, reflecting the effect of chamfer optimization on the impedance matching of the low-frequency band.

[0083] like Figure 17 As shown, simulation data of return loss at the edge frequencies of the high-frequency band are supplemented to verify the impedance matching performance across the entire high-frequency band.

[0084] like Figure 18 As shown, the return loss characteristics of different waveguide traces in the high-frequency band are further verified, demonstrating the impact of traces of uniform length on high-frequency transmission.

[0085] like Figure 19 As shown, the return loss details at characteristic frequency points in the low-frequency band are displayed, reflecting the signal transmission loss characteristics of the waveguide in the transmission frequency band.

[0086] like Figure 20 As shown, the reflection coefficient changes of multiple waveguide channels in the low-frequency band are displayed in the form of multiple curves, which fully verifies the impedance matching effect of multiple channels in the low-frequency band.

[0087] like Figure 21 As shown, simulation data of return loss for different waveguide channels in the low-frequency band are provided to demonstrate the impact of layered arrangement on low-frequency transmission.

[0088] like Figure 22 As shown, the return loss characteristics of multilayer waveguides in the low-frequency band are demonstrated, and the low-frequency transmission performance under layered arrangement is verified.

[0089] like Figure 23 As shown, further supplementary data on return loss of multiple layers and channels in the low-frequency band are provided to improve the performance verification of the low-frequency band.

[0090] like Figure 24 As shown, the return loss characteristics of the low-frequency end waveguide channel are demonstrated, verifying the end transmission performance of the radial trace.

[0091] like Figure 25 As shown, the distribution of low-frequency reflection coefficients is displayed in a multi-channel comparison format, which intuitively reflects the compliance status of S11≤-22dB across the entire low-frequency band.

[0092] like Figure 26 As shown, simulation data of return loss at high frequencies in the low-frequency band are supplemented to verify the impedance matching performance at the upper frequency limit of the transmission frequency band.

[0093] like Figure 27 As shown, the return loss characteristics of the low-frequency band low-layer waveguide channel are demonstrated, and the transmission performance of the low-layer waveguide under layered arrangement is verified.

[0094] like Figure 28 As shown, the simulation data of return loss in the 17.5GHz-20GHz transmit band and the 27.5GHz-30GHz receive band are integrated to fully demonstrate the overall impedance matching and signal transmission performance of the dual-band waveguide conversion layer.

[0095] The above description is merely a preferred embodiment of the present invention and is not intended to limit the scope of the present invention. Any modifications or equivalent substitutions made to the present invention without departing from the spirit and scope thereof should be covered within the protection scope of the claims of the present invention.

Claims

1. A dual-band waveguide conversion layer adapted to aperiodic large-spacing feed arrays, characterized in that, From the RF front-end to the feed array element, it includes an RF interface module, a dual-band optimized chamfered waveguide module, a uniform length trace module, and a layered layout module. The above modules are connected to the calibration and verification module respectively. The waveguide conversion layer is adapted to dual-band signal transmission of 17.5GHz-20GHz transmission frequency band and 27.5GHz-30GHz reception frequency band. It connects feed array elements radially with the central distribution node as the core, and is adapted to non-periodic, large-spacing feed arrays arranged in a spiral pattern.

2. The dual-band waveguide conversion layer adapted to aperiodic large-spacing feed array according to claim 1, characterized in that, The RF interface module serves as the signal input and output terminal of the dual-band waveguide conversion network, enabling waveguide interface adaptation between the network and the RF front-end and feed array elements. It adopts a standard rectangular waveguide interface and is matched with the dual-band transmission master mode TE. 10 The module provides a stable transmission access and output channel for dual-band radio frequency signals; The dual-band optimized chamfered waveguide module designs two different types of chamfer parameters for the 17.5GHz-20GHz Tx band and the 27.5GHz-30GHz Rx band. By utilizing the principles of reactance compensation and field shaping, it suppresses the excitation of higher-order modes at the bend, achieves a smooth transition of the field distribution and impedance continuity at the bend, and makes the reflection coefficient at the bend ≤-20dB under dual-band conditions.

3. A dual-band waveguide conversion layer adapted to aperiodic large-spacing feed array according to claim 2, characterized in that, Two types of chamfer structures are set at the bend of the waveguide: chamfer 1 at the bend between the z-direction and the xoy plane, and chamfer 2 at the bend within the xoy plane. Both chamfer 1 and chamfer 2 adopt the method of chamfering at equal distances on both sides.

4. A dual-band waveguide conversion layer adapted to aperiodic large-spacing feed array according to claim 3, characterized in that, The uniform length routing module ensures phase consistency between channels connecting the elements of the sparse array. By unifying the total routing length of all channel waveguides in the xoy plane to 120mm and controlling the length deviation to ≤±0.1mm, the module eliminates the phase difference caused by the different physical lengths of each channel, under the premise of consistent waveguide material and cross-sectional dimensions, and achieves a phase difference between channels ≤±3° across the entire frequency band.

5. A dual-band waveguide conversion layer adapted to aperiodic large-spacing feed array according to claim 4, characterized in that, For non-periodic spiral layouts, each channel waveguide achieves a uniform length of 120mm through a combination of "straight segment + optimized chamfered turning segment of dual-band optimized chamfered waveguide module" and the number of turns is ≤2; under the premise of consistent waveguide material and cross-sectional size, the uniform physical length ensures that the phase change of each channel is consistent.

6. A dual-band waveguide conversion layer adapted to aperiodic large-spacing feed array according to claim 5, characterized in that, The layered arrangement module is adapted to the spiral arrangement of aperiodic large-spacing feed arrays. It adopts a five-layer layered structure to adapt to the aperiodic spiral layout; the interlayer waveguides maintain a safe distance of ≥5 mm.

7. A dual-band waveguide conversion layer adapted to aperiodic large-spacing feed array according to claim 6, characterized in that, The layer division principle of the layered layout module is: For waveguides whose upper and lower surfaces overlap in the top view, the upper surface waveguide is assigned to the upper layer and the lower surface waveguide to the lower layer; the upper surface waveguide closer to the center origin is assigned to the upper layer and the lower surface waveguide to the lower layer to avoid too many waveguides in the central area; waveguides that are close in location should be assigned to the same layer as much as possible to reduce the loss caused by the transition between layers; the upper layer waveguides should be assigned and connected first, and the lower layer waveguides should be assigned after the utilization rate of the upper layer is saturated.

8. A dual-band waveguide conversion layer adapted to aperiodic large-spacing feed array according to claim 6, characterized in that, The same-layer arrangement principle of the layered arrangement module is as follows: the waveguides in the same layer have the same direction, all facing the x or y direction; the waveguides in the same layer maintain the same spacing and angle to ensure consistent mutual coupling between channels; the waveguide traces in the same layer avoid the spatial areas of the upper and lower surfaces and maintain a safe distance from the upper and lower waveguides.

9. A dual-band waveguide conversion layer adapted to aperiodic large-spacing feed array according to claim 6, characterized in that, The calibration and verification module is used to verify the phase consistency and impedance matching performance under dual frequency bands, forming a closed-loop optimization, including a phase calibration unit and an impedance verification unit. The phase calibration unit measures the channel phase value at the characteristic frequency point of the dual-band, and the waveguide length is finely adjusted to compensate for the channel that does not meet the phase requirements; the impedance verification unit measures the reflection coefficient and standing wave ratio in the dual-band, and the chamfer parameters are finely adjusted for the parts where the impedance matching is not up to standard, so as to ensure that S11≤-22dB and the phase difference between channels≤±3 in the entire frequency band of the network. If the phase difference between channels is greater than ±3°, the waveguide length deviation is finely adjusted with a fine adjustment amount ≤ ±0.05mm for phase compensation. The impedance verification unit measures the reflection coefficient and standing wave ratio of each waveguide in the dual-band. If the reflection coefficient S11 is greater than -18dB, the chamfer parameters of the dual-band optimized chamfer waveguide module are finely adjusted with a fine adjustment amount ≤ ±0.05mm until the reflection coefficient S11 is ≤ -22dB and the standing wave ratio is ≤ 1.3 in the entire frequency band.

10. The application of a dual-band waveguide conversion layer adapted to aperiodic large-spacing feed arrays as described in any one of claims 1-9, characterized in that, The dual-band waveguide conversion layer adapted to aperiodic large-pitch feed arrays is suitable for satellite communication, radio astronomy, phased array radar and ultra-wideband imaging systems, enabling dual-band low-loss and high-phase-consistency signal transmission between the RF front-end and the aperiodic large-pitch feed array.

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