Surface-mounted permanent magnet synchronous motor full-speed domain position sensorless control method

Abstract

A sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed domain is disclosed, belonging to the field of motor control technology. This invention addresses the technical problems of poor stability and unstable switching in traditional sensorless control strategies for surface-mounted permanent magnet synchronous motors across the entire speed domain. The invention includes: S1, constructing a dual-closed-loop control system for the surface-mounted permanent magnet synchronous motor using SVPWM modulation; S2, acquiring the motor's position information using a high-frequency pulse signal injection method when the motor is running at low speed; S3, employing a linear weighted method for switching operating conditions when the motor switches from low speed to high speed to ensure a smooth transition to high-speed operation; S4, acquiring the motor's position information using a sliding mode observer method combined with a phase-locked loop when the motor is running at high speed.

Description

Technical Field

[0001] This invention relates to the field of motor control technology, and more specifically, to a sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range. Background Technology

[0002] As the core device for energy conversion and power transmission, the motor system is widely used in industrial manufacturing, transportation, new energy power generation, and smart homes. Its performance directly affects the operating efficiency and reliability of the equipment. In the vector control system of permanent magnet synchronous motors, mechanical position sensors (such as photoelectric encoders and Hall sensors) are typically used to obtain real-time speed and rotor position information. However, the use of mechanical position sensors inevitably leads to increased system cost, larger motor size, and inconvenient installation. In addition, environmental factors such as temperature, humidity, and vibration can also seriously affect the service life and measurement accuracy of mechanical sensors.

[0003] In recent years, sensorless control algorithms have become a research hotspot to improve the reliability of vector control systems for permanent magnet synchronous motors and reduce production costs and equipment weight. In existing technologies, position estimation is typically performed using physical quantities such as stator voltage and stator current. However, due to the small back electromotive force at low speeds, there are differences between low-speed and high-speed sensorless control methods; therefore, speed-domain control strategies are often employed.

[0004] like Figure 1 As shown, in an existing sensorless control method for permanent magnet synchronous motors, a current-frequency ratio starting control method is used at low speeds, switching to a sensorless control method based on a sliding mode observer at high speeds. This strategy accelerates the motor to 30% of its rated speed through current-frequency ratio control, and then switches to high-speed sensorless control mode via a current switching function. However, this method cannot stabilize the speed at low speeds, mainly because the current-frequency ratio starting method cannot achieve stable operation at low speeds. This results in limited control of the surface-mounted permanent magnet synchronous motor at low speeds, affecting the system's operational stability and operating range.

[0005] Therefore, how to ensure stable operation of surface-mounted permanent magnet synchronous motors at high speeds while achieving stable operation and smooth switching at low speeds has become a pressing technical problem to be solved in current full-speed-domain sensorless control strategies. Summary of the Invention

[0006] To address the technical problems of poor stability and unstable switching in low-speed conditions in traditional sensorless control strategies for surface-mounted permanent magnet synchronous motors across the entire speed range, this invention provides a sensorless control method for surface-mounted permanent magnet synchronous motors across the entire speed range.

[0007] The present invention discloses a sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range, the method comprising the following steps:

[0008] S1. Construct a dual closed-loop control system for the surface-mounted permanent magnet synchronous motor, using the SVPWM modulation method;

[0009] S2. When the motor is running at low speed, the position information of the motor is obtained by high-frequency pulse signal injection method;

[0010] S3. When the motor operating condition changes from low speed to high speed, a linear weighted method is used to switch the operating condition to ensure that the motor can smoothly transition to the high-speed operating condition.

[0011] S4. When the motor is running at high speed, the position information of the motor is obtained by using the sliding mode observer method combined with the phase-locked loop.

[0012] Preferably, the construction of the dual closed-loop control system for the surface-mounted permanent magnet synchronous motor in step S1 specifically involves: constructing a simulation model of the surface-mounted permanent magnet synchronous motor, adopting a current-speed dual closed-loop control structure, and using a control method with Id=0; adjusting the PI regulation parameters, and by providing the motor speed and torque, ensuring that the overshoot and response time of the motor control meet engineering standards.

[0013] Preferably, the motor operation is divided into three speed ranges: speeds below... Zero-low speed zone to Speed ​​switching zone, speed higher than The medium-to-high speed zone, among which To switch the starting speed, To complete the speed switch;

[0014] At speed lower At that time, only the high-frequency pulse signal injection method described in step S2 is used to obtain rotor position information;

[0015] At a speed higher At that time, only the sliding mode observer method described in step S4 combined with the phase-locked loop is used to obtain the rotor position information;

[0016] exist to In the speed switching zone, both S2 and S4 control algorithms are used. The rotor position estimate is obtained by weighting the high-frequency pulse signal injection method and the sliding mode observer method.

[0017] Preferably, in step S2, the position information of the motor is obtained using a high-frequency pulse signal injection method, specifically: in the estimated synchronous rotating coordinate system of the motor... In the middle, towards A high-frequency pulsed voltage signal is injected into the shaft. The given injected high-frequency pulsed voltage signal is:

[0018]

[0019] In the formula, To estimate the direct-axis high-frequency voltage, To estimate the quadrature-axis high-frequency voltage, The magnitude of the injected voltage;

[0020] The frequency of the injected voltage;

[0021] The rotor position error signal is obtained after passing through a low-pass filter. :

[0022]

[0023] In the formula, For average inductance, These are direct-axis and quadrature-axis inductors, respectively. It is a half-differential inductor; This represents the rotor position estimation error.

[0024] when When the error approaches 0, the system gain is:

[0025]

[0026] The rotor position error signal As the input to the PI controller, the output of the PI controller is used as the speed estimate, which is then integrated to obtain the rotor position estimate.

[0027] Preferably, the linear weighted method is used for operating condition switching in step S3, specifically as follows:

[0028] In the speed switching zone, the high-frequency pulse signal injection method described in step S2 and the sliding mode observer method described in step S4 are combined with a phase-locked loop. The rotor position estimate is obtained by weighting the two methods. The specific algorithm formula is as follows:

[0029]

[0030] In the formula, and The electrical angle is the electrical angular velocity estimated by the pulsed high-frequency voltage signal injection method. and The electric angular velocity and electric angle estimated for the nonlinear observer. and The electric angular velocity and electric angle are calculated using the linear weighting method. and The weighting coefficients for the high-frequency injection method and the nonlinear observer satisfy the following conditions: .

[0031] Preferably, the weighting coefficients The calculation formula is:

[0032] .

[0033] Preferably, step S4 uses a sliding mode observer method combined with a phase-locked loop to obtain the motor's position information, specifically including the following steps:

[0034] Step 1: In the stationary coordinate system Below, a sliding mode observer model of a surface-mounted permanent magnet synchronous motor is constructed to obtain an estimate of the back electromotive force;

[0035] Step 2: Input the obtained back EMF estimate into the phase-locked loop to obtain the rotor position estimate. and estimated electric angular velocity .

[0036] Preferably, the estimated value of the back electromotive force is:

[0037]

[0038] In the formula, They are respectively axis, Back electromotive force estimate of the shaft The gain coefficient of the sliding mode observer. They are respectively axis, shaft current estimate Compared with the true value The difference between them , ;

[0039] The sliding mode switching function is chosen as sigmoid(x):

[0040]

[0041] In the formula, is the steepness coefficient of the sigmoid function.

[0042] Preferably, the process of obtaining the rotor position estimate in step two is as follows:

[0043] First, the input error signal of the phase-locked loop is obtained according to the following formula. :

[0044]

[0045] In the formula, It is a permanent magnet flux chain. This is the actual rotor electrical angle. This represents the linearization gain coefficient of the phase-locked loop;

[0046] Then, the error signal The estimated electric angular velocity is obtained after processing by a PI controller. The rotor position estimate is obtained after integration. .

[0047] Preferably, steps S1 to S4 are performed in the following sequence:

[0048] The motor is first accelerated to the weighted composite control stage by the high-frequency pulse signal injection method described in step S2, then the transition stage control is achieved by the linear weighting method described in step S3, and finally the sliding mode observer method combined with phase-locked loop control stage described in step S4 is entered to achieve sensorless control in the full speed domain.

[0049] The beneficial effects of this invention are:

[0050] 1. This invention reduces the cost of position sensors in permanent magnet synchronous motor control by utilizing algorithms. By designing a sensorless algorithm for permanent magnet synchronous motors, this invention eliminates the need for mechanical position sensors (such as photoelectric encoders and Hall effect sensors) to acquire position information, thus saving the cost of mechanical position sensors.

[0051] 2. The reliability of the permanent magnet synchronous motor speed control system is improved by utilizing algorithms. This invention designs a sensorless algorithm for permanent magnet synchronous motors, avoiding sensor malfunctions caused by environmental factors such as temperature, humidity, and vibration in traditional control methods, thus improving the lifespan of the motor control system.

[0052] 3. Significantly improves the dynamic response and steady-state performance of surface-mounted permanent magnet synchronous motors. The high-frequency pulse signal injection method and sliding mode observer combined with phase-locked loop observation method involved in this design can effectively improve speed overshoot and response time performance. At the same time, simulation results show that using the method of this invention, torque fluctuation and quadrature-axis current fluctuation are within a reasonable range.

[0053] 4. This design effectively solves the instability problem caused by the switching of permanent magnet synchronous motor operating conditions from low speed to high speed in the original method. It employs a speed hysteresis comparator method, enabling stepless switching between the two control methods during motor operation transitions. Simulation results show that the motor can smoothly transition from low-speed to high-speed operating conditions.

[0054] 5. This design effectively solves the problem of low-speed control instability related to the current-frequency ratio starting of permanent magnet synchronous motors in the original method. It employs a high-frequency pulse signal injection method to obtain motor speed information, enabling the motor to operate stably at low speeds. Simultaneously, it resolves the issues of requiring the design of current switching functions and excessively long start-up times inherent in traditional current-frequency ratio control.

[0055] 6. The implementation method is simple and easy to implement in engineering. This invention only adjusts the vector action order according to the parity of the sector based on the existing modulation strategy, without increasing the additional hardware cost. The algorithm has low complexity and is easy to implement in existing digital controllers, and has good engineering promotion value.

[0056] In summary, this invention, while maintaining the high-speed control performance of the original permanent magnet synchronous motor, effectively solves the problem of speed maintenance in the traditional low-speed control method of frequency ratio control, significantly improving the output waveform quality and operational reliability of the inverter system, while effectively controlling speed-torque pulsation and speed fluctuation. It is particularly suitable for applications in aerospace, marine propulsion, and other fields with high requirements for cost savings and weight reduction. Attached Figure Description

[0057] Figure 1 This is a block diagram of a traditional frequency-controlled closed-loop start and sensorless dual closed-loop control.

[0058] Figure 2 This is a schematic diagram of the motor used in the specific implementation method described in this invention;

[0059] Figure 3 This is a block diagram of a sensorless low-speed control system using high-frequency pulse signal injection.

[0060] Figure 4 This is a block diagram of the sensorless full-speed-domain control of the built-in permanent magnet synchronous motor described in this invention;

[0061] Figure 5 This is a simulation waveform diagram of the high-speed speed control of the sensorless motor control described in this invention.

[0062] Figure 6 This is a simulation waveform diagram of the speed error of the sensorless high-speed control motor described in this invention.

[0063] Figure 7 This is a simulation waveform diagram of the high-speed control angle of the sensorless motor control described in this invention.

[0064] Figure 8 This is a simulation waveform diagram of the angle error of the sensorless high-speed control of the motor described in this invention.

[0065] Figure 9 This is a simulation waveform diagram of the high-speed control torque of the sensorless motor control described in this invention.

[0066] Figure 10 This is a simulation waveform diagram of the quadrature-axis current of the sensorless high-speed control motor described in this invention.

[0067] Figure 11The above is a simulation waveform diagram of the three-phase current of the sensorless high-speed control motor described in this invention.

[0068] Figure 12 This is a simulation waveform diagram of the full-speed-domain control speed of the motor without sensor control according to the present invention;

[0069] Figure 13 This is a simulation waveform diagram of the speed error of the sensorless full-speed domain control of the motor described in this invention;

[0070] Figure 14 This is a simulation waveform diagram of the full-speed-domain control angle of the sensorless motor control described in this invention;

[0071] Figure 15 This is a simulation waveform diagram of the angle error of the sensorless full-speed domain control of the motor described in this invention;

[0072] Figure 16 This is a simulation waveform diagram of the full-speed-domain control torque of the sensorless motor control described in this invention;

[0073] Figure 17 This is a schematic diagram of the switching process of the sensorless motor control linear weighted method described in this invention;

[0074] Figure 18 This is a flowchart of the sensorless full-speed-domain control of the motor described in this invention. Detailed Implementation

[0075] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0076] It should be noted that, unless otherwise specified, the embodiments and features described in the present invention can be combined with each other.

[0077] The present invention will be further described below with reference to the accompanying drawings and specific embodiments, but this is not intended to limit the scope of the invention.

[0078] Specific Implementation Method 1: The following is combined with... Figures 2 to 18 This embodiment describes a sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range. The method includes the following steps:

[0079] S1. Construct a dual closed-loop control system for the surface-mounted permanent magnet synchronous motor, using the SVPWM modulation method;

[0080] S2. When the motor is running at low speed, the position information of the motor is obtained by high-frequency pulse signal injection method;

[0081] S3. When the motor operating condition changes from low speed to high speed, a linear weighted method is used to switch the operating condition to ensure that the motor can smoothly transition to the high-speed operating condition.

[0082] S4. When the motor is running at high speed, the position information of the motor is obtained by using the sliding mode observer method combined with the phase-locked loop.

[0083] Steps S1 to S4 are executed in the following sequence:

[0084] The motor is first accelerated to the weighted composite control stage by the high-frequency pulse signal injection method described in step S2, then the transition stage control is achieved by the linear weighting method described in step S3, and finally the sliding mode observer method combined with phase-locked loop control stage described in step S4 is entered to achieve sensorless control in the full speed domain.

[0085] The surface-mounted permanent magnet synchronous motor structure involved in this embodiment is as follows: Figure 2 As shown. This surface-mount permanent magnet synchronous motor is a motor for electric aircraft, and its characteristics conform to the non-polar surface-mount permanent magnet synchronous motor involved in this invention. The important parameters of this motor are shown in the table below:

[0086]

[0087] Step S1: Build a dual closed-loop control system for the surface-mounted permanent magnet synchronous motor, using the SVPWM modulation method.

[0088] A dual-closed-loop control system for a surface-mounted permanent magnet synchronous motor was constructed. The control section consists of an outer speed loop and an inner current loop, employing SVPWM modulation with a sampling frequency of 10000Hz. The PI controller parameters for the current loop were obtained through empirical formulas, while the PI controller parameters for the speed loop were adjusted using a pre-built model to find suitable PI controller parameters corresponding to appropriate speed overshoot and response time. The motor uses a control method where Id=0. The control signal is obtained by inverse Park transform and SVPWM modulation to generate the control signal for the switching transistors. The three-phase response currents of the motor are obtained as Id and Iq through Clark and Park transforms, forming a closed loop.

[0089] Step S2: When the motor is running at low speed, the position information of the motor is obtained by using a high-frequency pulse signal injection method.

[0090] Under low-speed conditions, since the motor is analyzed after the injection of a high-frequency signal, it is necessary to first construct a high-frequency inductive reactance model of the motor. The voltage equation of the motor under high-frequency excitation is as follows:

[0091]

[0092] In the formula, Average inductance; It is a half-differential inductor;

[0093] To estimate the direct-axis high-frequency voltage and current;

[0094] To estimate the quadrature-axis high-frequency voltage and current.

[0095] Towards The shaft-injected pulsating high-frequency voltage signal is:

[0096]

[0097] In the formula, The magnitude of the injected voltage;

[0098] The frequency of the injected voltage.

[0099] exist The voltage signal injected into the shaft will The shaft generates a high-frequency response current:

[0100]

[0101] As can be seen from the above formula, the estimated amplitude of the quadrature-axis high-frequency current includes the rotor position estimation error. ,when When it is zero, that is Using a cutoff frequency of A high-pass filter is used to select the high-frequency components for estimating the quadrature-axis current. Then use a high-frequency signal of the same frequency. Multiplying it and modulating it, then passing it through a low-pass filter, yields the rotor position error signal:

[0102]

[0103] when When it approaches 0, Approaching 2 The above formula can be simplified to:

[0104]

[0105] The saturated salient polarity of a surface-mounted permanent magnet synchronous motor can be determined by the ratio. To characterize this, for ease of analysis, the error system gain is:

[0106]

[0107] The rotor position error signal As the input to the PI controller, the output of the PI controller is used as the speed estimate, which is then integrated to obtain the rotor position estimate.

[0108] Figure 3 This is a block diagram of a sensorless low-speed control system using high-frequency pulse signal injection. Given the direct-axis current and the quadrature-axis current, Given a rotational speed.

[0109] Step S3: When the motor operating condition switches from low speed to high speed, a linear weighted method is used to switch the operating condition to ensure that the motor can smoothly transition to the high-speed operating condition.

[0110] To achieve sensorless vector control of a surface-mounted permanent magnet synchronous motor across the entire speed range, different rotor position estimation strategies can be employed in different speed ranges, and the switching between the two control strategies can be achieved through a switching algorithm. This invention uses a linear weighting method to switch between the sliding mode observer and the high-frequency pulse voltage signal injection method.

[0111] The motor operating range is divided into three stages: below the speed... The zero-low speed zone arrive The speed switching zone between them, the speed is higher than The medium to high speed range. Speeds below... The pulsed high-frequency voltage signal injection method is used, and the rotational speed is higher than that. The position is observed using a sliding mode observer phase-locked loop. During algorithm switching, the position is determined jointly by two algorithms, and a weighted function is used to achieve composite control of the two algorithms, such as... Figure 16 As shown.

[0112] Speed ​​below During the start-up and low-speed phases, the control algorithm employs a pulsed high-frequency voltage signal injection method, with weighting coefficients... A value of 1 indicates that a high-speed control method is not used, and the weighting coefficient is... The value is 0; the rotational speed is higher than 0. During the medium-to-high speed phase, the control algorithm employs the sliding mode observer method, with weighting coefficients... A value of 1 indicates that the pulsed high-frequency voltage signal injection method is not used, and the weighting coefficient is 1. The value is 0; arrive Two control algorithms are used simultaneously in the speed switching zone. The rotor position estimate is jointly determined by the pulsed high-frequency voltage signal injection method and the nonlinear observer. The participation degree is represented by a weighted coefficient. The specific algorithm formula is as follows:

[0113]

[0114] In the formula, and The electrical angle is the electrical angular velocity estimated by the pulsed high-frequency voltage signal injection method. and The electric angular velocity and electric angle estimated for the nonlinear observer. and The electric angular velocity and electric angle are calculated using the linear weighting method. and The weighting coefficients for the high-frequency injection method and the nonlinear observer satisfy the following conditions: , and The calculation is as follows:

[0115]

[0116] Figure 17 This is a schematic diagram of the switching process of the sensorless control linear weighted method for motors described in this invention.

[0117] Step S4: When the motor is running at high speed, the position information of the motor is obtained by using the sliding mode observer method combined with the phase-locked loop.

[0118] When the motor is running at high speed, the sliding mode observer method combined with a phase-locked loop is used to obtain the motor's position information. The sliding mode observer model for the surface-mounted permanent magnet synchronous motor is constructed as follows:

[0119]

[0120] In the formula Stator current The estimated value of the component, the induced electromotive force The estimated value of the components, in surface-mounted motors, is the stator inductance. R is the stator resistance. for The voltage on the shaft. Based on the concept of equivalent control, the estimated value of the induced electromotive force can be obtained as follows:

[0121]

[0122] In the formula, They are respectively axis, Back electromotive force estimate of the shaft The gain coefficient of the sliding mode observer. They are respectively axis, shaft current estimate Compared with the true value The difference between them , Therefore, the sliding surface function S(x) is defined as follows:

[0123]

[0124] To ensure that the current in the sliding mode observer can move stably towards S(x), the following condition must be met:

[0125]

[0126] In the formula, for transpose, for The transpose of .

[0127] The sliding mode variable structure control function is chosen to be the sigmoid(x) function:

[0128]

[0129] After the sliding mode observer obtains the estimated induced electromotive force, the required position signal is then obtained through a PLL phase-locked loop:

[0130]

[0131] The error signal The estimated electric angular velocity is obtained after processing by a PI controller. The rotor position estimate is obtained after integration. .

[0132] Figure 5 and Figure 6 The graphs show the observed and actual motor speeds and their error waveforms under high-speed conditions. It can be seen that the motor speed error is less than 20 r / min under high-speed conditions. Figure 7 and Figure 8 The graphs show the observed and actual angles of the motor under high-speed conditions, along with their error waveforms. It can be observed that the motor angle error is less than 3° under high-speed conditions. In conclusion, the motor performs well in sensorless control under high-speed conditions.

[0133] Figure 9 and Figure 10 The electromagnetic torque and quadrature-axis current waveforms of the motor are shown. When a torque fluctuation of 10 Nm is applied at 0.6 s, it can be found that the pulsation is very small, indicating that the motor can work under load and resist certain load torque disturbances. Figure 11 The waveforms of the three-phase currents of the motor show that the waveform envelopes of the three-phase currents are normal.

[0134] Combining steps S2, S3, and S4, a sensorless control method for the full speed range of a surface-mounted permanent magnet synchronous motor is obtained. Figure 18 This is a flowchart of a sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range. Figure 4This is a block diagram of the sensorless full-speed-domain control of the built-in permanent magnet synchronous motor described in this invention, wherein... This is a function representing the given speed of the motor.

[0135] Implementation results demonstration:

[0136] Figure 12 The waveform diagram for the full-speed domain control of the sensorless motor described in this invention is as follows: A low-speed, high-frequency pulse signal injection strategy is used for the motor from 0-1s; a weighted function is used for composite control from 1-2s; and after 2s, a high-speed sliding mode observer control is employed. The observed speed of the motor can closely match its actual speed. Figure 13 Assuming the motor speed error at the same moment, it can be observed that the motor speed error is close to 0 in steady state.

[0137] Figure 14 This is a simulation waveform diagram of the full-speed-domain control angle of the sensorless motor control described in this invention. It can be observed that the observed angle of the motor can follow the actual angle of the motor very well. Figure 15 The waveform diagram shown is a simulation of the angle error of the sensorless full-speed-domain control of the motor described in this invention. The angle error of the motor is less than 3° during low-speed and high-speed control. Figure 16 The simulation waveform of the full-speed-domain control torque of the sensorless motor control described in this invention shows that the torque can be stabilized near the target torque with minimal fluctuations.

[0138] While the invention has been described herein with reference to specific embodiments, it should be understood that these embodiments are merely examples of the principles and applications of the invention. Therefore, it should be understood that many modifications can be made to the exemplary embodiments, and other arrangements can be designed without departing from the spirit and scope of the invention as defined by the appended claims. It should be understood that different dependent claims and features described herein can be combined in ways different from those described in the original claims. It is also understood that features described in conjunction with individual embodiments can be used in other described embodiments.

Claims

1. A sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range, characterized in that, The method includes the following steps: S1. Construct a dual closed-loop control system for the surface-mounted permanent magnet synchronous motor, using the SVPWM modulation method; S2. When the motor is running at low speed, the position information of the motor is obtained by high-frequency pulse signal injection method; S3. When the motor operating condition changes from low speed to high speed, a linear weighted method is used to switch the operating condition to ensure that the motor can smoothly transition to the high-speed operating condition. S4. When the motor is running at high speed, the position information of the motor is obtained by using the sliding mode observer method combined with the phase-locked loop.

2. The sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range according to claim 1, characterized in that, In step S1, the dual closed-loop control system of the surface-mounted permanent magnet synchronous motor is constructed as follows: a simulation model of the surface-mounted permanent magnet synchronous motor is constructed, a current-speed dual closed-loop control structure is adopted, and the control method with Id=0 is used; the PI regulation parameters are adjusted, and the motor control overshoot and response time meet the engineering standards by giving the motor speed and torque.

3. The sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range according to claim 2, characterized in that, The motor's operation is divided into three speed ranges: below [speed range] Zero-low speed zone to Speed ​​switching zone, speed higher than The medium-to-high speed zone, among which To switch the starting speed, To complete the speed switch; At speed lower At that time, only the high-frequency pulse signal injection method described in step S2 is used to obtain rotor position information; At a speed higher At that time, only the sliding mode observer method described in step S4 combined with the phase-locked loop is used to obtain the rotor position information; exist to In the speed switching zone, both S2 and S4 control algorithms are used. The rotor position estimate is obtained by weighting the high-frequency pulse signal injection method and the sliding mode observer method.

4. The sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range according to claim 3, characterized in that, In step S2, the position information of the motor is obtained using a high-frequency pulse signal injection method. Specifically, this involves: in the estimated synchronous rotating coordinate system of the motor... In the middle, towards A high-frequency pulsed voltage signal is injected into the shaft. The given injected high-frequency pulsed voltage signal is: In the formula, To estimate the direct-axis high-frequency voltage, To estimate the quadrature-axis high-frequency voltage, The magnitude of the injected voltage; The frequency of the injected voltage; The rotor position error signal is obtained after passing through a low-pass filter. : In the formula, For average inductance, These are direct-axis and quadrature-axis inductors, respectively. It is a half-differential inductor; This represents the rotor position estimation error. when When the error approaches 0, the system gain is: The rotor position error signal As the input to the PI controller, the output of the PI controller is used as the speed estimate, which is then integrated to obtain the rotor position estimate.

5. The sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range according to claim 3, characterized in that, In step S3, a linear weighted method is used for switching operating conditions, specifically as follows: In the speed switching zone, the high-frequency pulse signal injection method described in step S2 and the sliding mode observer method described in step S4 are combined with a phase-locked loop. The rotor position estimate is obtained by weighting the two methods. The specific algorithm formula is as follows: In the formula, and The electrical angle is the electrical angular velocity estimated by the pulsed high-frequency voltage signal injection method. and The electric angular velocity and electric angle estimated for the nonlinear observer. and The electric angular velocity and electric angle are calculated using the linear weighting method. and The weighting coefficients for the high-frequency injection method and the nonlinear observer satisfy the following conditions: .

6. The sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range according to claim 5, characterized in that, The weighting coefficients The calculation formula is: 。 7. The sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range according to claim 3, characterized in that, Step S4 uses a sliding mode observer method combined with a phase-locked loop to obtain the motor's position information, specifically including the following steps: Step 1: In the stationary coordinate system Below, a sliding mode observer model of a surface-mounted permanent magnet synchronous motor is constructed to obtain an estimate of the back electromotive force; Step 2: Input the obtained back EMF estimate into the phase-locked loop to obtain the rotor position estimate. and estimated electric angular velocity .

8. The sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range according to claim 7, characterized in that, The estimated value of the back electromotive force is: In the formula, They are respectively axis, Back electromotive force estimate of the shaft The gain coefficient of the sliding mode observer. They are respectively axis, shaft current estimate Compared with the true value The difference between them , ; The sliding mode switching function is chosen as sigmoid(x): In the formula, is the steepness coefficient of the sigmoid function.

9. A sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range according to claim 7, characterized in that, The process of obtaining the rotor position estimate in step two is as follows: First, the input error signal of the phase-locked loop is obtained according to the following formula. : In the formula, It is a permanent magnet flux chain. This is the actual rotor electrical angle. This represents the linearization gain coefficient of the phase-locked loop; Then, the error signal The estimated electric angular velocity is obtained after processing by a PI controller. The rotor position estimate is obtained after integration. .

10. A sensorless control method for a surface-mounted permanent magnet synchronous motor across the entire speed range according to claim 1, characterized in that, Steps S1 to S4 are executed in the following sequence: The motor is first accelerated to the weighted composite control stage by the high-frequency pulse signal injection method described in step S2, then the transition stage control is achieved by the linear weighting method described in step S3, and finally the sliding mode observer method combined with phase-locked loop control stage described in step S4 is entered to achieve sensorless control in the full speed domain.

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