Power conversion circuit and power converter to which it is applied

Abstract

The application discloses a power conversion circuit and a power converter. The power conversion circuit comprises a first end, a second end, a first switch power conversion unit comprising a first switch and a third switch connected in series, a second switch power conversion unit comprising a second switch and a fourth switch connected in series, the third switch being connected in series with the first switch, and the fourth switch being connected in series with the second switch, a flying capacitor, and a magnetic assembly comprising two first windings and a second winding electromagnetically coupled, the two first windings being connected to each other at the same end and connected to the second end, one of the first windings being connected in series between the flying capacitor and the second end, the other first winding being connected in series between the second switch and the second end, the second winding being connected in series with the flying capacitor and the first windings, and the first end of the second winding and the first end of one of the first windings being connected to each other at the same end, wherein the turns ratio of the second winding and the two first windings is N:1:1.

Description

Technical Field

[0001] This case relates to a power conversion circuit, and more particularly to a power conversion circuit with a flexible voltage gain ratio and the power converter to which it is applied. Background Technology

[0002] In existing non-isolated high-current power conversion circuits, resonant scalable duty cycle topologies can be used. These resonant scalable duty cycle power conversion circuits can be symmetrical or asymmetrical. However, currently, regardless of whether the resonant scalable duty cycle power conversion circuit is symmetrical or asymmetrical, its input and output voltage gain ratio is fixed. This fixed voltage gain ratio limits the application range of resonant scalable duty cycle power conversion circuits. Utility Model Content

[0003] The purpose of this invention is to provide a power conversion circuit and a power converter suitable for it, which allows for flexible design of the voltage gain ratio of the power conversion circuit, thus expanding the application range of the power conversion circuit.

[0004] To achieve the above objectives, this invention provides a power conversion circuit, comprising: a first terminal including a first positive terminal and a first negative terminal; a second terminal including a second positive terminal and a second negative terminal, wherein the second negative terminal is electrically connected to the first negative terminal; a first switching power conversion unit including a first switch and a third switch connected in series; a second switching power conversion unit including a second switch and a fourth switch connected in series, wherein the first terminal of the first switch is electrically connected to the first terminal of the second switch, the second terminal of the first switch is electrically connected to the first positive terminal, the third switch is connected in series with the first switch, the fourth switch is connected in series with the second switch, the first terminals of the third switch and the fourth switch are electrically connected to the first negative terminal, and the second terminal of the fourth switch is connected to the second terminal of the second switch; the first, second, third, and fourth switches operate periodically according to a switching cycle, the switching cycle including a duty cycle; a flying capacitor; and a magnetic component including two first windings. The first and second windings are electromagnetically coupled to each other. The second ends of the two first windings are opposite ends and electrically connected to each other and to the second positive terminal. The first end of one of the first windings is connected in series to the second end of the third switch. The first end of the other first winding is electrically connected to the second end of the fourth switch and the second end of the second switch. The second winding and the flying capacitor are connected in series between the first end of the first switch and the first end of one of the first windings. The first end of the second winding and the first end of one of the first windings are the same end. The turns ratio of the second winding and the two first windings is N:1:1, where N is a positive number. The switching cycle includes two working intervals: a first working interval and a second working interval. In the first working interval, the current flowing through the second winding is equal to the current flowing through one of the first windings. In the second working interval, the current flowing through the second winding is equal to the current flowing through the other first winding.

[0005] In some embodiments, the power conversion circuit is a bidirectional power conversion circuit, with the first terminal of the power conversion circuit being an input terminal or an output terminal, and the second terminal of the power conversion circuit being an output terminal or an input terminal.

[0006] In some embodiments, the first end of the second winding is electrically connected to the first end of one of the first windings, and the second end of the second winding is electrically connected to one end of the flying capacitor; or one end of the flying capacitor is electrically connected to the first end of one of the first windings, and the other end of the flying capacitor is electrically connected to the first end of the second winding.

[0007] In some embodiments, the switching state of the first switch is the same as the switching state of the fourth switch, the switching state of the second switch is the same as the switching state of the third switch, and the switching state of the first switch is 180 degrees out of phase with the switching state of the second switch, and the conduction time of the first switch and the second switch is less than or equal to half of the switching period and greater than or equal to four times the switching period.

[0008] In some embodiments, the power conversion circuit has an equivalent resonant inductance that resonates with the flying capacitor. The resulting resonance has a resonant frequency and a resonant period. The switching period of the power conversion circuit is less than or equal to the resonant period, and the switching period is greater than or equal to half of the resonant period.

[0009] In some embodiments, the equivalent resonant inductance is the leakage inductance of the magnetic component and the parasitic inductance of the circuit; or the equivalent resonant inductance is at least one external inductor connected in series between the first terminal of the first switch and the second positive terminal.

[0010] In some embodiments, the two first windings and the second winding are wound on the same magnetic post of the same magnetic core assembly, and the coupling coefficient between each pair of the two first windings and the second winding is greater than 9.

[0011] In some embodiments, during a switching cycle, the flying capacitor synchronously stores energy and synchronously transfers energy to the output of the power conversion circuit as the corresponding switch is turned on and off, and the flying capacitor has a DC voltage, the ratio of which to the terminal voltage of the first terminal of the power conversion circuit is between 4 and 6.

[0012] In some embodiments, the voltage gain ratio of the terminal voltages of the first and second terminals of the power conversion circuit is (4-2N):1.

[0013] To achieve the above objectives, this invention further provides a power conversion circuit, comprising: a first terminal, including a first positive terminal and a first negative terminal; a second terminal, including a second positive terminal and a second negative terminal, wherein the second negative terminal is electrically connected to the first negative terminal; a first flying capacitor; a second flying capacitor; a first switching power conversion unit, including a first switch, a second switch, and a third switch, wherein the first terminal of the first switch is electrically connected to the first positive terminal, the second terminal of the second switch is electrically connected to the first terminal of the third switch, and the second terminal of the third switch is electrically connected to the second negative terminal; a second switching power conversion unit, including a fourth switch, a fifth switch, and a sixth switch, wherein the first terminal of the fourth switch is electrically connected to the first positive terminal, the second terminal of the fifth switch is electrically connected to the first terminal of the sixth switch, and the second terminal of the sixth switch is electrically connected to the second negative terminal, and the second terminal of the fourth switch is electrically connected to the first terminal of the second switch, and the first terminal of the fifth switch is electrically connected to the second terminal of the first switch; wherein the first, second, third, fourth, fifth, and sixth switches are periodically switched according to a switching cycle. The operation includes a switching cycle with a duty cycle; and a magnetic component comprising two first windings and two second windings, which are electromagnetically coupled to each other. The second ends of the two first windings are opposite terminals and electrically connected to each other and to a second positive terminal. The first end of one of the first windings is electrically connected to the second end of a fifth switch and the first end of a sixth switch, and the first end of the other first winding is electrically connected to the second end of a second switch and the first end of a third switch. One of the second windings is connected in series with a second flying capacitor between the second ends of a fourth switch and the second ends of a fifth switch. The other second winding is connected in series with a first flying capacitor between the second ends of a first switch and the second ends of a second switch. The first end of one of the second windings is the same terminal as the first end of the first winding and is electrically connected. The first end of the other second winding is the same terminal as the first end of the other first winding and is electrically connected. The turns ratio of the two second windings and the two first windings is N:N:1:1, where N is a positive number. The switching cycle includes two working intervals: a first working interval and a second working interval. In the first working interval, the sum of the currents flowing through the two second windings is equal to the current flowing through one of the first windings. In the second working interval, the sum of the currents flowing through the two second windings is equal to the current flowing through the other first winding.

[0014] In some embodiments, the power conversion circuit is a bidirectional power conversion circuit, with the first terminal of the power conversion circuit being an input terminal or an output terminal, and the second terminal of the power conversion circuit being an output terminal or an input terminal.

[0015] In some embodiments, the second end of one of the second windings is electrically connected to a second flying capacitor, and the second end of the other second winding is electrically connected to a first flying capacitor.

[0016] In some embodiments, one end of the second flying capacitor is electrically connected to the first end of one of the first windings, the other end of the second flying capacitor is electrically connected to the first end of one of the second windings, and one end of the first flying capacitor is electrically connected to the first end of another first winding, and the other end of the first flying capacitor is electrically connected to the first end of another second winding.

[0017] In some embodiments, the switching states of the first switch, the second switch, and the sixth switch are the same, the switching states of the fourth switch, the fifth switch, and the third switch are the same, and the switching states of the first switch and the fourth switch are out of phase by 180 degrees, and the conduction time of the first switch and the fourth switch is less than or equal to half of the switching period and greater than or equal to 0.4 times the switching period.

[0018] In some embodiments, the power conversion circuit has an equivalent resonant inductance, which resonates with the first flying capacitor and the second flying capacitor. The resulting resonance has a resonant frequency and a resonant period. The switching period of the power conversion circuit is less than or equal to the resonant period, and the switching period of the power conversion circuit is greater than or equal to half of the resonant period.

[0019] In some embodiments, the equivalent resonant inductance is the leakage inductance of the magnetic component and the parasitic inductance of the circuit, or the equivalent resonant inductance is at least one external inductor connected in series between the first terminal of the first switch and the second positive terminal.

[0020] In some embodiments, the two first windings and the two second windings are wound on the same magnetic post of the same magnetic core assembly, and the coupling coefficient between each pair of the two first windings and the two second windings is greater than 0.9.

[0021] In some embodiments, during a switching cycle, the first flying capacitor and the second flying capacitor synchronously store energy and synchronously transfer energy to the output terminal of the power conversion circuit as the corresponding switches are turned on and off. Both the first flying capacitor and the second flying capacitor have DC voltages, and the ratio of each DC voltage to the terminal voltage of the first terminal of the power conversion circuit is between 0.4 and 0.6.

[0022] In some embodiments, the voltage gain ratio of the first and second terminals of the power conversion circuit is (4-2N):1.

[0023] To achieve the above objectives, this invention further provides a power converter, comprising: N power conversion circuits, wherein each power conversion circuit includes: a first terminal, comprising a first positive terminal and a first negative terminal; a second terminal, comprising a second positive terminal and a second negative terminal, wherein the second negative terminal is electrically connected to the first negative terminal; a first switching power conversion unit, comprising a first switch and a third switch connected in series; a second switching power conversion unit, comprising a second switch and a fourth switch connected in series, wherein the first terminal of the first switch is electrically connected to the first terminal of the second switch, the second terminal of the first switch is electrically connected to the first positive terminal, the third switch is connected in series with the first switch, and the fourth switch is connected in series with the second switch, wherein the third switch's first terminal is connected in series with the first switch, and the fourth switch is connected in series with the second switch, wherein the third switch's second terminal is connected in series with the first positive terminal; the third terminal of the third switch is connected in series with the first positive terminal; the fourth terminal of the third switch is connected in series with the first positive terminal; the fifth ... One end of the first switch and the first end of the fourth switch are electrically connected to the first negative terminal. The second end of the fourth switch is electrically connected to the second end of the second switch. The first, second, third, and fourth switches operate periodically according to a switching cycle, which includes a duty cycle; a flying capacitor; and a magnetic component, including two first windings and a second winding, which are electromagnetically coupled to each other. The second ends of the two first windings are opposite ends and electrically connected to each other and to the second positive terminal. The first end of one of the first windings is connected in series to the second end of the third switch, and the first end of the other first winding is electrically connected to the second end of the fourth switch and the second end of the second switch. The second winding is connected to the flying capacitor. A capacitor is connected in series between the first terminal of the first switch and the first terminal of one of the first windings. The first terminal of the second winding and the first terminal of the first winding are of the same name. The turns ratio of the second winding and the two first windings is N:1:1, where N is a positive number. The switching cycle includes two working intervals: a first working interval and a second working interval. In the first working interval, the current flowing through the second winding is equal to the current flowing through one of the first windings. In the second working interval, the current flowing through the second winding is equal to the current flowing through the other first winding. N first terminals of the power conversion circuit are connected in parallel, and N second terminals of the power conversion circuit are connected in parallel. When N is greater than 1 and is odd, the N power conversion circuits are controlled by N sets of control signals. Each set of control signals controls each power conversion circuit. The corresponding control signals in the N sets are sequentially out of phase by any value between [360 / N-20, 360 / N+20] degrees. When N is greater than 1 and is even, the N power conversion circuits are controlled by N / 2 sets of control signals. The nth set of control signals controls the nth and N / 2+nth power conversion circuits. The corresponding control signals in the N / 2 sets are sequentially out of phase by any value between [360 / N-20, 360 / N+20] degrees.

[0024] To achieve the above objectives, this application also provides a power converter, comprising: N power conversion circuits, each power conversion circuit comprising: a first terminal, including a first positive terminal and a first negative terminal; a second terminal, including a second positive terminal and a second negative terminal, wherein the second negative terminal is electrically connected to the first negative terminal; a first flying capacitor; a second flying capacitor; a first switching power conversion unit, comprising a first switch, a second switch, and a third switch, wherein the first terminal of the first switch is electrically connected to the first positive terminal, the second terminal of the second switch is electrically connected to the first terminal of the third switch, and the second terminal of the third switch is electrically connected to the second negative terminal; a second switching power conversion unit, comprising a fourth switch, a fifth switch, and a sixth switch, wherein the first terminal of the fourth switch is electrically connected to the first positive terminal, the second terminal of the fifth switch is electrically connected to the first terminal of the sixth switch, the second terminal of the sixth switch is electrically connected to the second negative terminal, and the second terminal of the fourth switch is electrically connected to the first terminal of the second switch, and the first terminal of the fifth switch is electrically connected to the second terminal of the first switch. The first, second, third, fourth, fifth, and sixth switches operate periodically according to a switching cycle, which includes a duty cycle. A magnetic component is also included, comprising two first windings and two second windings, which are electromagnetically coupled to each other. The second ends of the two first windings are opposite ends and electrically connected to each other and to a second positive electrode. The first end of one of the first windings is electrically connected to the second end of the fifth switch and the first end of the sixth switch, while the first end of the other first winding is electrically connected to the second end of the second switch and the first end of the third switch. One of the second windings is connected in series with a second flying capacitor between the second ends of the fourth and fifth switches, and the other second winding is connected in series with a first flying capacitor between the second ends of the first and second switches. The turns ratio of the two second windings and the two first windings is N:N:1:1, where N is a positive number. The switching cycle includes two working intervals: a first working interval and a second working interval. In the first working interval, the sum of the currents flowing through the two second windings is equal to the current flowing through one of the first windings. In the second working interval, the sum of the currents flowing through the two second windings is equal to the current flowing through the other first winding. The first terminals of N power conversion circuits are connected in parallel, and the second terminals of N power conversion circuits are connected in parallel, where N is greater than 1. The N power conversion circuits are controlled by N sets of control signals. Each set of control signals is used to control each power conversion circuit. The corresponding control signals in the N sets of control signals are sequentially out of phase by an angle of any value between [360 / 2N-20, 360 / 2N+20] degrees. Attached Figure Description

[0025] Figure 1a This is a schematic diagram of the power conversion circuit in the first preferred embodiment of this case;

[0026] Figure 1b for Figure 1aThe waveform diagrams of the switching, voltage, and current operation of the power conversion circuit are shown.

[0027] Figure 1c for Figure 1a The power conversion circuit shown is in Figure 1b The equivalent circuit diagram shown is for the time interval t0-t1.

[0028] Figure 1d for Figure 1a The power conversion circuit shown is in Figure 1b The equivalent circuit diagram shown is for the time interval t2-t3.

[0029] Figure 1e This is a schematic diagram of the power conversion circuit in the second preferred embodiment of this case;

[0030] Figure 2a This is a schematic diagram of the power conversion circuit in the third preferred embodiment of this case;

[0031] Figure 2b for Figure 2a The waveform diagrams of the switching, voltage, and current operation within the power conversion circuit are shown.

[0032] Figure 2c for Figure 2a The power conversion circuit shown is in Figure 2b The equivalent circuit diagram shown is for the time interval t0-t1.

[0033] Figure 2d for Figure 2a The power conversion circuit shown is in Figure 2b The equivalent circuit diagram for the t2-t3 interval is shown below;

[0034] Figure 2e This is a schematic diagram of the power conversion circuit in the fourth preferred embodiment of this case;

[0035] Figure 3 This is a schematic diagram of the circuit structure of the power converter in the first preferred embodiment of this case;

[0036] Figure 4 This is a schematic diagram of the circuit structure of the power converter in the second preferred embodiment of this case;

[0037] The reference numerals in the attached figures are explained as follows:

[0038] 1, 2: Power conversion circuit

[0039] V1+: First positive electrode

[0040] V1-: First negative electrode

[0041] V2+: Second positive electrode

[0042] V2-: Second negative electrode

[0043] C1: First capacitor

[0044] C2: Second capacitor

[0045] Cb11: Flying Capacitor

[0046] T-1: Magnetic Components

[0047] S11, S21: First switch

[0048] S12, S24: Second switches

[0049] Sr11, Sr22: Third switch

[0050] Sr12, S23: Fourth Switch

[0051] T11, T12, T21, T22: First winding

[0052] T13, T23, T24: Second winding

[0053] Ts: Switching cycle

[0054] T11', T12', T21', T22': Equivalent first winding

[0055] T13', T23', T24': Equivalent second winding

[0056] Vc11: Voltage across flying capacitor Cb11

[0057] Lr11, Lr12, Lr13, Lr21, Lr22, Lr23, Lr24: Equivalent leakage inductance

[0058] Lm1, Lm2: Magnetizing inductors

[0059] iLm1, iLm2: Excitation current

[0060] Vlm1, Vlm2: Excitation voltage

[0061] io: Output current

[0062] iLr11, iLr12, iLr13, iLr21, iLr22, iLr23, iLr24: Resonant Current

[0063] Cb21: First Flying Capacitor

[0064] Cb22: Second Flying Capacitor

[0065] S22: Fifth Switch

[0066] Sr21: Sixth Switch

[0067] Vc22: Voltage of the second flying capacitor Cb22

[0068] Vc21: Voltage of the first flying capacitor Cb21

[0069] 100, 110: Power converter Detailed Implementation

[0070] Some typical embodiments that embody the features and advantages of this invention will be described in detail in the following description. It should be understood that this invention can have various variations in different implementations, all of which do not depart from the scope of this invention, and the descriptions and illustrations therein are for illustrative purposes only and are not intended to limit this invention.

[0071] Figure 1a This is a schematic diagram of the power conversion circuit in the first preferred embodiment of this case. Figure 1b for Figure 1a The diagram shows the switching, voltage, and current waveforms of the power conversion circuit. Figure 1c for Figure 1a The power conversion circuit shown is in Figure 1b The equivalent circuit diagram shown is for the time interval t0-t1. Figure 1d for Figure 1a The power conversion circuit shown is in Figure 1b The equivalent circuit diagram is shown for the time interval t2-t3. The power conversion circuits described below can all perform bidirectional power conversion. The first terminal of the power conversion circuit is the input terminal, and the second terminal is the output terminal; or, the first terminal is the output terminal, and the second terminal is the input terminal. Furthermore, the power conversion circuit is a resonant circuit topology with an expandable duty cycle. Figure 1aThe power conversion circuit 1 of the first preferred embodiment shown has an asymmetrical circuit topology and includes a first terminal (containing a first positive terminal V1+ and a first negative terminal V1-), a second terminal (containing a second positive terminal V2+ and a second negative terminal V2-), a first switching power conversion unit, a second switching power conversion unit, a first capacitor C1, a second capacitor C2, a flying capacitor Cb11, and a magnetic component T-1. The first negative terminal V1- and the second negative terminal V2- are grounded together. The first switching power conversion unit includes a first switch S11 and a third switch Sr11 connected in series. The second switching power conversion unit includes a second switch S12 and a fourth switch Sr12 connected in series. The first switch S11, the second switch S12, the third switch Sr11, and the fourth switch Sr12 operate periodically according to a switching cycle, and the switching cycle includes a duty cycle. Furthermore, the first terminal of the first switch S11 is electrically connected to the first terminal of the second switch S12, the second terminal of the first switch S11 is electrically connected to the first positive terminal V1+, the second terminal of the fourth switch Sr12 is electrically connected to the second terminal of the second switch S12, and the first terminals of the third switch Sr11 and the fourth switch Sr12 are electrically connected and also electrically connected to the first negative terminal V2-. In addition, the switching state of the first switch S11 is the same as the switching state of the fourth switch Sr12, the switching state of the second switch S12 is the same as the switching state of the third switch Sr11, and the switching states of the first switch S11 and the second switch S12 are 180 degrees out of phase. The conduction time of the first switch S11 and the second switch S12 is less than or equal to half of the switching cycle and greater than or equal to 0.4 times the switching cycle. The first capacitor C1 is electrically connected between the first positive terminal V1+ and the first negative terminal V1-, and the second capacitor C2 is electrically connected between the second positive terminal V2+ and the second negative terminal V2-.

[0072] The magnetic component T-1 includes two first windings T11 and T12 and a second winding T13. The two first windings T11 and T12 and the second winding T13 are electromagnetically coupled through the same magnetic core assembly and wound on the same magnetic post. The second ends of the two first windings T11 and T12, which are opposite-named terminals, are electrically connected to the second positive terminal V2+. The first end of the first winding T11 is electrically connected to the second end of the third switch Sr11, and the first end of the other first winding T12 is electrically connected between the second ends of the fourth switch Sr12 and the second end of the second switch S12. The second winding T13 is connected in series with a flying capacitor Cb11 and then electrically connected between the first end of the first switch S11, the second end of the third switch Sr11, and the first winding T11. Furthermore, the turns ratio of the second winding T13 and the two first windings T11 and T12 is N:1:1, where N is a positive number, preferably a positive integer. In this case, there are no restrictions on the positional order of the second winding T13 and the flying capacitor Cb11, as long as they are in series.

[0073] In one embodiment, the first end of the second winding T13 is electrically connected to the first end of the first winding T11, and the first end of the second winding T13 and the first end of the first winding T11 are terminals with the same name. The second end of the second winding T13 is electrically connected to the flying capacitor Cb11. In another embodiment, one end of the flying capacitor Cb11 is electrically connected to the first end of the first winding T11, and the other end of the flying capacitor Cb11 is electrically connected to the first end of the second winding T13, and the first end of the second winding T13 and the first end of the first winding T11 are terminals with the same name.

[0074] The voltage gain ratio of power conversion circuit 1 will be explained below, and for now, we will tentatively designate the first terminal of power conversion circuit 1 as the input voltage terminal and the second terminal as the output voltage terminal. Please refer to [further details needed]. Figures 1b-1d When the power conversion circuit 1 operates in steady state, the time interval t0-t4 is one switching cycle Ts. During the time interval t0-t1, the first switch S11 and the fourth switch Sr12 are turned on, which can be marked as the first operating interval. At this time, the input voltage V1 charges the flying capacitor Cb11 through the first switch S11, and simultaneously transfers energy to the output terminal through the second winding T13 and the first winding T11. The first winding T12 freewheels through the fourth switch Sr12. At this time, the current flowing through the second winding T13 is equal to the current flowing through the first winding T11. Its equivalent circuit is referenced [reference needed]. Figure 1c Where T11', T12', and T13' are the ideal windings corresponding to each winding, Lr11, Lr12, and Lr13 are the equivalent leakage inductances corresponding to each winding, and Lm1 is the equivalent magnetizing inductance of the magnetic component T-1. In one embodiment, the equivalent resonant inductance formed by the equivalent leakage inductances Lr11, Lr12, and Lr13 resonates with the flying capacitor Cb11, generating resonant currents iLr11 and iLr12. The magnetizing current in the magnetic component is equivalent to iLm1. Within this range, the corresponding node voltage relationships of the power conversion circuit 1 can be referred to... Figure 1c As shown, the voltage across the ideal first winding T12' is equal to the voltage at the second terminal of power conversion circuit 1, let's assume it's V2. Since the turns ratio between the second winding T13 and the first windings T11 and T12 is N:1:1, the voltage across the ideal first winding T11' is also V2, and the voltage across the ideal second winding T13' is N·V2. Therefore, the voltage V1 at the first terminal of power conversion circuit 1 can be deduced as:

[0075] V1=Vc11+(2-N)·V2…(1);

[0076] Where Vc11 is the terminal voltage of the flying capacitor Cb11. At time t1, when the resonant currents iLr11 and iLr12 are equal to the excitation currents iLm1 and -iLm1 respectively, the first switch S11 and the fourth switch Sr12 are turned off to achieve zero-current turn-off (ZCS) of the switches, thereby reducing the turn-off loss of the switches and improving the energy transfer efficiency of the power conversion circuit 1.

[0077] During the interval from time t1 to time t2, all switches are turned off, and the excitation current iLm1 flowing through the magnetic component T-1 continues to draw charge from the parasitic capacitances across the second switch S12 and the third switch Sr11, causing the voltages across the second switch S12 and the third switch Sr11 to drop. In one embodiment, when the voltages across the second switch S12 and the third switch Sr11 drop to less than 50% of their respective initial voltages (i.e., the voltages across their respective switches at time t1), the second switch S12 and the third switch Sr11 can be turned on. This reduces the switching losses, thereby improving the energy conversion efficiency and power density of the power conversion circuit. In another embodiment, the inductance of the magnetic component T-1 can be controlled so that the inductance of the equivalent magnetizing inductance Lm1 of the magnetic component T-1 is small enough, thereby making the magnetizing current iLm1 flowing through the equivalent magnetizing inductance Lm1 large enough to completely remove the charge on the parasitic capacitances at the two ends of the second switch S12 and the third switch Sr11, causing the voltage across the two ends of the second switch S12 and the third switch Sr11 to drop to zero. This allows the body diodes of the second switch S12 and the third switch Sr11 to be turned on, thereby achieving zero-voltage switching (ZVS). This can further reduce the switching losses and improve the energy conversion efficiency and power density of the power conversion circuit 1.

[0078] During the time interval t2-t3, the second switch S12 and the third switch Sr11 are turned on, which can be marked as the second operating interval. At this time, the energy stored in the flying capacitor Cb11 is transferred to the output terminal via the second switch S12, the first winding T12, the third switch Sr11, and the second winding T13. The first winding T11 freewheels through the third switch Sr11. The current flowing through the second winding T13 is equal to the current flowing through the first winding T12. Its equivalent circuit is referenced [reference needed]. Figure 1d As shown, the equivalent resonant inductance formed by leakage inductances Lr11, Lr12, and Lr13 resonates with the flying capacitor Cb11, generating resonant currents iLr11 and iLr12. The excitation current in this magnetic component is equivalent to iLm1. Within this range, the corresponding node voltage relationships of the power conversion circuit 1 can be referenced. Figure 1dThe ideal voltage across the first winding T11' is equal to the voltage V2 at the second terminal of the power conversion circuit 1. Since the turns ratio between the second winding T13 and the first windings T11 and T12 is N:1:1, the ideal voltage across the first winding T12' is also V2. The ideal voltage across the second winding T13' is N·V2. Therefore, the voltage Vc11 of the flying capacitor Cb11 can be derived as:

[0079] Vc11=(2-N)·V2…(2);

[0080] Since the energy stored in the flying capacitor Cb11 in the t0-t1 interval is transferred to the output terminal in the t2-t3 interval, substituting equation (2) into equation (1) yields the voltage V1 = (4-2N)·V2 at the first terminal of the power conversion circuit 1. At time t3, when the resonant currents iLr11 and iLr12 are equal to the magnetizing currents iLm1 and -iLm1 respectively, the second switch S12 and the third switch Sr11 are turned off to achieve zero-current turn-off (ZCS), thereby reducing the turn-off loss of the switches and improving the energy transfer efficiency of the power conversion circuit 1.

[0081] During the interval from time t3 to time t4, all switches are turned off, and the excitation current iLm1 flowing through the first windings T11 and T12 continues to draw charge from the parasitic capacitances across the first switch S11 and the fourth switch Sr12, causing the voltage across the first switch S11 and the fourth switch Sr12 to drop. In one embodiment, when the voltage across the first switch S11 and the fourth switch Sr12 drops to less than 50% of their respective initial voltages (i.e., the voltages across their respective switches at time t1), the first switch S11 and the fourth switch Sr12 can be turned on. This reduces the switching losses, thereby improving the energy conversion efficiency and power density of the power conversion circuit. In another embodiment, the inductance of the magnetic component T-1 can be controlled so that the inductance of the equivalent magnetizing inductance Lm1 of the magnetic component T-1 is small enough, thereby making the magnetizing current iLm1 flowing through the magnetizing inductance Lm1 large enough to completely remove the charge on the parasitic capacitances at the ends of the first switch S11 and the fourth switch Sr12, causing the voltage across the first switch S11 and the fourth switch Sr12 to drop to zero. This allows the body diodes of the first switch S11 and the fourth switch Sr12 to be turned on, thereby achieving zero-voltage switching (ZVS). This further reduces the switching losses and improves the energy conversion efficiency and power density of the power conversion circuit 1.

[0082] During the intervals from time t0 to time t1 and from time t2 to time t3, the first windings T11 and T12 simultaneously carry resonant currents iLr11 and iLr12, and the frequencies of the resonant currents iLr11 and iLr12 are equal to the switching frequency. In this embodiment, the resonant period and the switching period are nearly equal. In other embodiments, taking advantage of the large capacitance of the flying capacitor Cb11 and the small inductance of the equivalent resonant inductor, the corresponding switches can be turned off when the resonant currents iLr11 and iLr12 are greater than the excitation currents iLm1 and -iLm1, respectively, during the t0-t1 interval; and when the resonant currents iLr11 and iLr12 are greater than the excitation currents -iLm1 and iLm1, respectively, during the t2-t3 interval. Although the turn-off current is greater than zero, the loss caused by the non-zero current turn-off can be ignored because the inductance of the equivalent resonant inductor is small. This makes the switching period of the power conversion circuit 1 less than or equal to the resonant period of the resonant current. However, in order to balance loss and energy transfer efficiency, the switching period Ts is better than or equal to 0.5 times the resonant period.

[0083] In this embodiment, the voltage gain ratio of the input voltage V1 to the output voltage V2 of the power conversion circuit 1 is (4-2N):1, which can be adjusted by changing N. In other words, by adding a second winding T13 to the magnetic component T-1, the voltage gain ratio of the power conversion circuit 1 can be flexibly designed by changing the number of turns of the second winding T13 and the first windings T11 and T12, based on the magnetic coupling relationship between the second winding T13 and the first windings T11 and T12, and by setting the turns ratio to N:1:1. Furthermore, since the voltage gain ratio of the input voltage V1 to the output voltage V2 of the power conversion circuit 1 in this embodiment is (4-2N):1, it can be applied in application areas requiring lower voltage gain, such as automotive vehicles.

[0084] Additionally, in this embodiment, as Figure 1c , 1d As shown, the electromagnetic coupling of the first winding T11, T12 and the second winding T13 generates corresponding equivalent leakage inductances Lr11, Lr12 and Lr13, respectively. For the sake of simplicity and ease of analysis of the relationship between each resonant current, the excitation current iLm1 and excitation voltage Vlm1 of the equivalent magnetizing inductance Lm1 are temporarily ignored. During the time intervals t0-t1 and t2-t3, the flying capacitor Cb11 resonates with the equivalent leakage inductances Lr11, Lr12 and Lr13. In this embodiment, the equivalent resonant capacitance of the power conversion circuit 1 is the flying capacitor Cb11, and the equivalent resonant inductance is the sum of the equivalent leakage inductances Lr11, Lr12 and Lr13. If the excitation current iLm1 is ignored, the output current io of the power conversion circuit 1 can be equivalently represented as:

[0085] io=iLr11+iLr12…(3);

[0086] Where iLr11 is the resonant current flowing through the equivalent leakage inductance Lr11, and iLr12 is the resonant current flowing through the equivalent leakage inductance Lr12. During the time interval t0-t1, the resonant current iLr13 flowing through the equivalent leakage inductance Lr13 is equal to the resonant current iLr11 flowing through the equivalent leakage inductance Lr11.

[0087] Right now:

[0088] iLr13=iLr11…(4);

[0089] Furthermore, based on the magnetic potential balance, we can obtain:

[0090] -N·iLr13+iLr11=iLr12…(5);

[0091] From the above equations (3), (4) and (5), it can be deduced that the resonant current iLr12 is (1-N)·io / (2-N) and the resonant current iLr11 is io / (2-N).

[0092] Furthermore, within the time interval t2-t3, the resonant current iLr13 flowing through the equivalent leakage inductance Lr13 is equal to the resonant current iLr12 flowing through the equivalent leakage inductance Lr12, that is:

[0093] iLr13=iLr12…(6);

[0094] Furthermore, based on the magnetic potential balance, we can obtain:

[0095] -N·iLr13+iLr12=iLr11…(7);

[0096] From the above equations (3), (6) and (7), it can be derived that the resonant current iLr11=(1-N)·io / (2-N) and the resonant current iLr12=io / (2-N).

[0097] As can be seen from the above, the power conversion circuit 1 of this embodiment not only achieves the technical effect of flexibly designing the voltage gain ratio through the second winding T13 of the magnetic component T-1, but also the sum of the resonant currents iLr11 and iLr12 in the t0-t1 interval is equal to the sum of the resonant currents iLr11 and iLr12 in the t2-t3 interval. It can be seen that the addition of the second winding T13 does not affect the resonance effect of the power conversion circuit 1. At the same time, the terminal voltage Vc11 of the flying capacitor is a DC voltage superimposed with an AC resonant voltage. The typical value of the DC voltage is Vin / 2, so the typical ratio of the DC voltage to the input voltage is 0.5. Considering factors such as the distribution of device parameters, the ratio of the DC voltage to the input voltage (the terminal voltage of the first terminal of the power conversion circuit 1) is between 0.4 and 0.6. The amplitude of the AC resonant voltage of the terminal voltage Vc11 is determined by the inductance of the equivalent resonant inductor, the capacitance of the equivalent resonant capacitor, the switching frequency of the power conversion circuit, and the load size.

[0098] In the foregoing embodiments, the equivalent resonant inductance includes the leakage inductance generated by the coupling between the two first windings T11 and T12 and the second winding T13 of the magnetic component T-1, as well as the parasitic inductance in the circuit. Considering the resonant period, the switching period, and the capacitance of the flying capacitor Cb11, the coupling coefficient between each pair of the two first windings T11 and T12 and the second winding T13 is preferably greater than 0.9, but is not limited thereto. Furthermore, in some embodiments, the equivalent resonant inductance includes an external inductor, specifically as follows: Figure 1e Specifically, an external inductor can be connected in series between the electrical connection point of the opposite ends of the two first windings T11 and T12 and the second positive terminal V2+ (point A); or two external inductors with the same inductance can be connected in series with the two first windings T11 and T12 respectively (points B1 or B2); or an external inductor can be connected in series with the flying capacitor Cb11 and the second winding T13 in the series branch (points C1, C2, or C3). Furthermore, external inductors can be simultaneously placed in two or all three of the above positions to obtain a suitable resonant period. Figure 1e It can be seen that the external inductor is actually connected in series between the first terminal of the first switch S11 and the second positive terminal V2+.

[0099] In some embodiments, diodes can be used to replace the third switch Sr11 and the fourth switch Sr12, respectively. The diodes serve as freewheeling diodes. The switching period of this power conversion circuit is less than or equal to the resonant period. Its equivalent circuit and current waveform can be derived from the aforementioned embodiments, and therefore will not be repeated here. The switches in the above power conversion circuit can be controllable switches such as MOS, SiC, and GaN.

[0100] Of course, since the power conversion circuit 1 in this embodiment is bidirectional, the first end of the power conversion circuit 1 can be the output voltage end, and the second end of the power conversion circuit 1 can be the input voltage end. However, its working principle is similar to that of the aforementioned power conversion circuit 1 where the first end is the input voltage end and the second end is the output voltage end. The only difference is that when the first end of the power conversion circuit 1 is the output voltage end and the second end is the input voltage end, the voltage gain ratio between the input and output of the power conversion circuit 1 is 1:(4-2N). Therefore, it will not be described in detail here.

[0101] Figure 2a This is a schematic diagram of the power conversion circuit in the third preferred embodiment of this case. Figure 2b for Figure 2a The diagram shows the operating waveforms of the switches, voltage, and current within the power conversion circuit. Figure 2c for Figure 2a The power conversion circuit shown is in Figure 2b The equivalent circuit diagram shown is for the time interval t0-t1. Figure 2d for Figure 2a The power conversion circuit shown is in Figure 2b The equivalent circuit diagram for the t2-t3 interval is shown. The power conversion circuit 2 in this embodiment has a symmetrical topology and includes a first terminal (containing a first positive terminal V1+ and a first negative terminal V1-), a second terminal (containing a second positive terminal V2+ and a second negative terminal V2-), a first capacitor C1, a second capacitor C2, a first flying capacitor Cb21, a second flying capacitor Cb22, a first switching power conversion unit, a second switching power conversion unit, and a resonant circuit T-2. The first negative terminal V1- and the second negative terminal V2- are grounded together.

[0102] The first switching power conversion unit includes a first switch S21, a second switch S24, and a third switch Sr22. The structure of the second switching power conversion unit is similar to that of the first switching power conversion unit, namely, it includes a fourth switch S23, a fifth switch S22, and a sixth switch Sr21. The first terminal of the first switch S21 is electrically connected to the first positive terminal V1+, the second terminal of the first switch S21 is electrically connected to the first terminal of the fifth switch S22, the second terminal of the fifth switch S22 is electrically connected to the first terminal of the sixth switch Sr21, and the second terminal of the sixth switch Sr21 is electrically connected to the second negative terminal V2-. The first terminal of the fourth switch S23 is electrically connected to the first positive terminal V1+ and is connected in parallel with the first switch S21. The second terminal of the fourth switch S23 is electrically connected to the first terminal of the second switch S24, the second terminal of the second switch S24 is electrically connected to the first terminal of the third switch Sr22, and the second terminal of the third switch Sr22 is electrically connected to the second negative terminal V2-. The first flying capacitor Cb21 is electrically connected between the second terminal of the first switch S21, the second terminal of the second switch S24, and the first terminal of the third switch Sr22. The second flying capacitor Cb22 is electrically connected between the second terminal of the fourth switch S23, the second terminal of the fifth switch S22, and the first terminal of the sixth switch Sr21. Furthermore, the first switch S21, the second switch S24, the third switch Sr22, the fourth switch S23, the fifth switch S22, and the sixth switch Sr21 operate periodically according to a switching cycle, and this switching cycle includes a duty cycle.

[0103] Furthermore, the switching states of the first switch S21, the second switch S24, and the sixth switch Sr21 are the same; the switching states of the fourth switch S23 and the fifth switch S22 are the same as the third switch Sr22; and the switching states of the first switch S21 and the fourth switch S23 are 180 degrees out of phase. The conduction time of the first switch S21 and the fourth switch S23 is less than or equal to half of the switching cycle and greater than or equal to 0.4 times the switching cycle. The first capacitor C1 is electrically connected between the first positive terminal V1+ and the first negative terminal V1-, and the second capacitor C2 is electrically connected between the second positive terminal V2+ and the second negative terminal V2-.

[0104] Magnetic component T-2 includes two first windings T21 and T22 and two second windings T23 and T24. The two first windings T21 and T22 and the two second windings T23 and T24 are electromagnetically coupled through the same magnetic core assembly and wound on the same magnetic post. The second ends of the two first windings T21 and T22, which are opposite-named terminals, are electrically connected to the second positive terminal V2+. The first end of the first winding T21 is electrically connected to the second end of the fifth switch S22 and the first end of the sixth switch Sr21. The first end of the first winding T22 is also electrically connected to the second end of the second switch S24 and the first end of the third switch Sr22. The second winding T23 is connected in series with the second flying capacitor Cb22 and then electrically connected between the second ends of the fourth switch S23 and the fifth switch S22. The second winding T24 is connected in series with the first flying capacitor Cb21 and then electrically connected between the second ends of the first switch S21 and the second switch S24. Even more remarkably, the turns ratio of the two second windings T23 and T24 and the two first windings T21 and T22 is N:N:1:1, where N is a positive number.

[0105] In this embodiment, the order of the second winding T23 and the second flying capacitor Cb22 is not limited. In one embodiment, the first end of the second winding T23 is connected in series with the first end of the first winding T21, and the first end of the second winding T23 and the first end of the first winding T21 are terminals with the same name. The second end of the second winding T23 is connected to the second flying capacitor Cb22. In another embodiment, one end of the second flying capacitor Cb22 is connected to the first end of the first winding T21, and the other end of the second flying capacitor Cb22 is connected to the first end of the second winding T23, and the first end of the second winding T23 and the first end of the first winding T21 are terminals with the same name.

[0106] In this case, the positional order of the second winding T24 and the first flying capacitor Cb21 is not limited. In one embodiment, the first end of the second winding T24 is connected in series with the first end of the first winding T22, and the first end of the second winding T24 and the first end of the first winding T22 are terminals with the same name. The second end of the second winding T24 is connected to the first flying capacitor Cb21. In another embodiment, one end of the first flying capacitor Cb21 is connected to the first end of the first winding T22, and the other end of the first flying capacitor Cb21 is connected to the first end of the second winding T24, and the first end of the second winding T24 and the first end of the first winding T22 are terminals with the same name.

[0107] The voltage gain ratio of power conversion circuit 2 will be explained below, and for now, we will tentatively designate the first terminal of power conversion circuit 2 as the input voltage terminal and the second terminal as the output voltage terminal. Please refer to [further details needed]. Figures 2b-2dWhen the power conversion circuit 2 operates in steady state, the time interval t0-t4 is one switching cycle Ts. During the time interval t0-t1, the first switch S21, the second switch S24, and the sixth switch Sr21 are turned on, which can be marked as the first operating interval. At this time, the input voltage V1 charges the first flying capacitor Cb21 through the first switch S21, and simultaneously transfers energy to the output terminal through the second winding T24 and the first winding T22. The energy stored in the second flying capacitor Cb22 is transferred to the output terminal through the second switch S24, the first winding T22, the sixth switch Sr21, and the second winding T23. The first winding T21 freewheels through the sixth switch Sr21. At this time, the sum of the current flowing through the second winding T23 and the current flowing through the second winding T24 is equal to the current flowing through the first winding T22. Its equivalent circuit is as follows: Figure 2c As shown, T21', T22', T23', and T24' are the ideal windings corresponding to each winding, Lr21, Lr22, Lr23, and Lr24 are the leakage inductances corresponding to each winding, and Lm2 is the excitation inductance of the magnetic component T-2. In one embodiment, the equivalent resonant inductance formed by the leakage inductances Lr21, Lr22, Lr23, and Lr24 resonates with the first flying capacitor Cb21 and the second flying capacitor Cb22, generating resonant currents iLr21 and iLr22. The excitation current in the magnetic component is equivalent to iLm2. Within this range, the corresponding node voltage relationships of the power conversion circuit 1 can be referred to as follows: Figure 2c As shown, the voltage across the ideal first winding T21' is equal to the voltage across the second terminal of the power conversion circuit 2, let's assume it's V2. Since the turns ratio of the two second windings T23 and T24 and the two first windings T21 and T22 is N:N:1:1, the voltage across the ideal first winding T22' is also V2. The voltages across the ideal second winding T23' and the ideal second winding T24' are N·V2 respectively. Therefore, we can deduce that the voltage V1 at the first terminal of the power conversion circuit 2 and the voltage Vc22 at the second flying capacitor Cb22 are respectively:

[0108] V1=Vc21+(2-N)·V2…(8);

[0109] Vc22=(2-N)·V2…(9);

[0110] Where Vc21 is the voltage of the first flying capacitor Cb21, and Vc22 is the voltage of the second flying capacitor Cb22. At time t1, when the resonant currents iLr21 and iLr22 are equal to the magnetizing currents iLm2 and -iLm2 respectively, the first switch S21, the second switch S24, and the sixth switch Sr21 are turned off to achieve zero-current turn-off (ZCS), thereby reducing the turn-off losses of the switches and improving the energy transfer efficiency of the power conversion circuit 2.

[0111] During the interval from time t1 to time t2, all switches are turned off, and the excitation current iLm2 flowing through the magnetic component T-2 continues to draw charge from the parasitic capacitances across the fourth switch S23, the fifth switch S22, and the third switch Sr22. This causes the voltages across the fourth switch S23, the fifth switch S22, and the third switch Sr22 to drop. In one embodiment, when the voltages across the fourth switch S23, the fifth switch S22, and the third switch Sr22 drop to less than 50% of their respective initial voltages (i.e., the voltages across their respective switches at time t1), the fourth switch S23, the fifth switch S22, and the third switch Sr22 can be turned on. This reduces the switching losses, thereby improving the energy conversion efficiency and power density of the power conversion circuit. In another embodiment, the magnetizing inductance of the magnetic component T-2 can be controlled to make the inductance of the equivalent magnetizing inductance Lm2 of the magnetic component T-2 sufficiently small, thereby making the magnetizing current iLm2 flowing through the magnetizing inductance Lm2 sufficiently large. This can completely remove the charge from the parasitic capacitances at the ends of the fourth switch S23, the fifth switch S22, and the third switch Sr22, causing the voltage across the fourth switch S23, the fifth switch S22, and the third switch Sr22 to drop to zero. This allows the body diodes of the fourth switch S23, the fifth switch S22, and the third switch Sr22 to be turned on, thereby achieving zero-voltage switching (ZVS). This can further reduce the switching losses and improve the energy conversion efficiency and power density of the power conversion circuit 2.

[0112] During the time interval t2-t3, the fourth switch S23, the fifth switch S22, and the third switch Sr22 are turned on, which can be marked as the second operating interval. At this time, the input voltage V1 charges the second flying capacitor Cb22 through the fourth switch S23, and simultaneously transfers energy to the output terminal through the second winding T23 and the first winding T21. The energy stored in the first flying capacitor Cb21 is transferred to the output terminal through the fifth switch S22, the first winding T21, the third switch Sr22, and the second winding T24. The first winding T22 freewheels through the third switch Sr22. At this time, the sum of the current flowing through the second winding T23 and the current flowing through the second winding T24 is equal to the current flowing through the first winding T21. Its equivalent circuit is as follows: Figure 2d As shown, in one embodiment, the equivalent resonant inductance formed by the equivalent leakage inductances Lr21, Lr22, Lr23, and Lr24 resonates with the first flying capacitor Cb21 and the second flying capacitor Cb22, generating resonant currents iLr21 and iLr22. The excitation current in this magnetic component is equivalent to iLm2. Within this range, the corresponding node voltage relationships of the power conversion circuit 2 can be referenced. Figure 2dThe voltage across the ideal first winding T22' is equal to the voltage V2 at the second terminal of the power conversion circuit 2. Since the turns ratio of the two second windings T23 and T24 and the two first windings T21 and T22 is N:N:1:1, the voltage across the ideal first winding T21' is also V2. The voltages across the ideal second winding T23' and the ideal second winding T24' are N·V2 respectively. Therefore, the voltage V1 at the first terminal of the power conversion circuit 2 and the voltage Vc21 at the first flying capacitor Cb21 can be deduced as follows:

[0113] V1=Vc22+(2-N)·V2…(10);

[0114] Vc21=(2-N)·V2…(11);

[0115] Since the energy stored in the first flying capacitor Cb21 in the t0-t1 interval is transferred to the output terminal in the t2-t3 interval, and the energy stored in the second flying capacitor Cb22 in the t2-t3 interval is transferred to the output terminal in the t0-t1 interval, the voltage V1 = (4-2N)·V2 at the first terminal of the power conversion circuit 2 can be derived according to equations (8)-(11).

[0116] At time t3, when the resonant currents iLr21 and iLr22 are equal to the excitation currents iLm2 and -iLm2 respectively, the fourth switch S23, the fifth switch S22 and the third switch Sr22 are turned off to achieve zero current turn-off (ZCS) of the switches, thereby reducing the turn-off loss of the switches and improving the energy transfer efficiency of the power conversion circuit 2.

[0117] During the interval from time t3 to time t4, all switches are turned off, and the excitation current iLm2 flowing through the magnetic component T-2 continues to draw charge from the parasitic capacitances across the first switch S21, the second switch S24, and the sixth switch Sr21, causing the voltages across these switches to drop. In one embodiment, when the voltages across these switches drop to less than 50% of their initial voltages (i.e., the voltages across their respective switches at time t1), the first switch S21, the second switch S24, and the sixth switch Sr21 can be turned on. This reduces switching losses, thereby improving the energy conversion efficiency and power density of the power conversion circuit. In another embodiment, the inductance of the magnetic component T-2 can be controlled so that the inductance of the equivalent magnetizing inductance Lm2 of the magnetic component T-2 is sufficiently small, thereby making the magnetizing current iLm2 flowing through the magnetizing inductance Lm2 sufficiently large. This can completely remove the charge from the parasitic capacitances at the ends of the first switch S21, the second switch S24, and the sixth switch Sr21, causing the voltage across the first switch S21, the second switch S24, and the sixth switch Sr21 to drop to zero. This allows the body diodes of the first switch S21, the second switch S24, and the sixth switch Sr21 to be turned on, thereby achieving zero-voltage switching (ZVS). This can further reduce the switching losses and improve the energy conversion efficiency and power density of the power conversion circuit 2.

[0118] During the intervals from time t0 to time t1 and from time t2 to time t3, the first windings T21 and T22 simultaneously carry resonant currents iLr21 and iLr22, and the resonant frequencies of the resonant currents iLr21 and iLr22 are equal to the switching frequency. In this embodiment, the resonant period and the switching period are nearly equal. In other embodiments, taking advantage of the large capacitance values ​​of the first flying capacitor Cb21 and the second flying capacitor Cb22 and the small inductance of the equivalent resonant inductor, the corresponding switches can be turned off when the resonant currents iLr22 and iLr21 are greater than the excitation currents iLm2 and -iLm2, respectively, during the t0-t1 interval; and when the resonant currents iLr22 and iLr21 are greater than the excitation currents -iLm2 and iLm2, respectively, during the t2-t3 interval. Although the turn-off current is greater than zero, the loss caused by the non-zero current turn-off can be ignored because the inductance of the equivalent resonant inductor is small. This makes the switching period of the power conversion circuit 2 less than or equal to the resonant period of the resonant current. However, in order to balance loss and energy transfer efficiency, the switching period Ts is better than or equal to 0.5 times the resonant period.

[0119] Therefore, in this embodiment, the voltage gain ratio of the input to the output of the power conversion circuit 2 is (4-2N):1, rather than fixed. In other words, by adding a second winding T23 and T24 to the magnetic component T-2, the turns ratio can be set to N:N:1:1 based on the magnetic coupling relationship between the second winding T23 and T24 and the first winding T21 and T22. Furthermore, the second winding T23 and T24 are positioned at specific locations in the power conversion circuit 2. Thus, the voltage gain ratio of the power conversion circuit 2 can be flexibly designed by changing the number of turns of the second winding T23 and T24, thereby expanding the application range of the power conversion circuit 2.

[0120] Additionally, in this embodiment, as Figure 2c , Figure 2d As shown, the electromagnetic coupling of the first winding T21 and T22 and the second winding T23 and T24 produces the corresponding equivalent leakage inductances Lr21, Lr22, Lr23 and Lr24, respectively. For the sake of convenience and simplicity in analyzing the relationship between each resonant current, the excitation current iLm2 and the excitation voltage Vlm2 of the equivalent magnetizing inductance Lm2 are ignored here. During the time intervals t0-t1 and t2-t3, the first flying capacitor Cb21 and the second flying capacitor Cb22 resonate with the equivalent leakage inductances Lr21, Lr22, Lr23, and Lr24. The equivalent resonant capacitance of the power conversion circuit 2 is the sum of the capacitances of the first flying capacitor Cb21 and the second flying capacitor Cb22. The equivalent resonant inductance of the power conversion circuit 2 is the equivalent leakage inductances Lr23 and Lr24 connected in parallel and then added to Lr21 and Lr22 (i.e., the equivalent resonant inductance of the power conversion circuit 2 is Lr23||Lr24+Lr21+Lr22). If the magnetizing current iLm2 is ignored, the output current io of the power conversion circuit 2 can be equivalently expressed as:

[0121] io=iLr21+iLr22…(12);

[0122] Where iLr21 is the resonant current flowing through the equivalent leakage inductance Lr21, and iLr22 is the resonant current flowing through the equivalent leakage inductance Lr22. During the time interval t0-t1, the resonant current iLr23 flowing through the equivalent leakage inductance Lr23 is equal to the resonant current iLr24 flowing through the equivalent leakage inductance Lr24, that is:

[0123] iLr23=iLr24…(13);

[0124] Furthermore, the sum of the resonant current iLr23 flowing through the equivalent leakage inductance Lr23 and the resonant current flowing through the equivalent leakage inductance Lr24 is equal to the resonant current iLr22 of the equivalent leakage inductance Lr22, that is...

[0125] iLr23+iLr24=iLr22…(14);

[0126] Furthermore, based on the magnetic potential balance, we can obtain:

[0127] -N·iLr23-N·iLr24+iLr22=iLr21…(15);

[0128] From the above equations (13), (14) and (15), it can be derived that the resonant current iLr22=io / (2-N) and the resonant current iLr21=(1-N)·io / (2-N).

[0129] Furthermore, within the time interval t2-t3, the resonant current iLr23 flowing through the equivalent leakage inductance Lr23 is equal to the resonant current iLr24 flowing through the equivalent leakage inductance Lr24 (as shown in equation (13)), and the sum of the resonant current iLr23 flowing through the equivalent leakage inductance Lr23 and the resonant current flowing through the equivalent leakage inductance Lr24 is equal to the resonant current iLr21 flowing through the equivalent leakage inductance Lr21 (as shown in equation (14)). Based on the magnetomotive force balance, we can obtain:

[0130] -N·iLr23-N·iLr24+iLr21=iLr22…(16);

[0131] From the above equations (13), (14) and (16), it can be derived that the resonant current iLr21=io / (2-N) and the resonant current iLr22=(1-N)·io / (2-N).

[0132] As can be seen from the above, the power conversion circuit 2 of this embodiment not only achieves the technical effect of flexibly designing the voltage gain ratio through the second windings T23 and T24 of the magnetic component T-2, but also ensures that the sum of the resonant currents iLr21 and iLr22 in the t0-t1 interval is equal to the sum of the resonant currents iLr21 and iLr22 in the t2-t3 interval. This shows that the addition of the second windings T23 and T24 does not affect the resonance effect of the power conversion circuit 2. Meanwhile, the terminal voltage Vc21 of the first flying capacitor Cb21 and the terminal voltage Vc22 of the second flying capacitor Cb22 are both DC voltages superimposed with an AC resonant voltage. The typical value of each DC voltage is Vin / 2, so the typical ratio of each DC voltage to the input voltage (e.g., the terminal voltage of the first terminal of the power conversion circuit 2) is 0.5. Considering factors such as the distribution of device parameters, the ratio of each DC voltage to the input voltage is between 0.4 and 0.6. The amplitudes of the AC resonant voltages Vc21 and Vc22 are determined by the inductance of the equivalent resonant inductor, the capacitance of the equivalent resonant capacitor, the switching frequency of the power conversion circuit, and the load size.

[0133] In the foregoing embodiments, the equivalent resonant inductance includes the leakage inductance generated by the coupling between the two first windings T21 and T22 and the two second windings T23 and T24 of the magnetic component T-2, as well as the parasitic inductance in the circuit. Considering the resonant period, the switching period, and the capacitance values ​​of the flying capacitors Cb21 and Cb22, the coupling coefficient between each pair of the two first windings T21 and T22 and the two second windings T23 and T24 is preferably greater than 0.9, but is not limited thereto. Furthermore, in some embodiments, the equivalent resonant inductance includes an external inductor, specifically as follows: Figure 2e Specifically, an external inductor can be connected in series between the electrical connection point of the opposite ends of the two first windings T21 and T22 and the second positive terminal V2+ (point D); or two external inductors with the same inductance can be connected in series with the two first windings T21 and T22 respectively (two points E1 or two points E2); or two external inductors with the same inductance can be connected in series with each flying capacitor and a second winding respectively (two points F1, two points F2, or two points E2). Furthermore, external inductors can be simultaneously placed in two or all three of the above positions to obtain a suitable resonant period. Figure 2e It can be seen that the external inductor is actually connected in series between the first terminal of the first switch S21 and the second positive terminal V2+.

[0134] In some embodiments, diodes can be used to replace the third switch Sr22 and the sixth switch Sr21, respectively. The diodes serve as freewheeling diodes. The switching period of this power conversion circuit is less than or equal to the resonant period. Its equivalent circuit and current waveform can be derived from the aforementioned embodiments, and therefore will not be repeated here. The switches in the above power conversion circuit can be controllable switches such as MOS, SiC, and GaN.

[0135] Of course, since the power conversion circuit 2 in this embodiment is bidirectional, the first end of the power conversion circuit 2 can be the output voltage end, and the second end of the power conversion circuit 2 can be the input voltage end. However, its working principle is similar to that of the above-mentioned first end being the input voltage end and the second end being the output voltage end. The only difference is that when the first end of the power conversion circuit 2 is the output voltage end and the second end is the input voltage end, the voltage gain ratio between the input and output of the power conversion circuit 2 is 1:(4-2N). Therefore, it will not be described again here.

[0136] Of course, to accommodate high-current applications, the aforementioned N power conversion circuits can be interleaved to increase the load capacity of the power conversion system, where N is greater than 1. The first terminals of the N power conversion circuits are connected in parallel, the second terminals of the N power conversion circuits are connected in parallel, and the N power conversion circuits have the same circuit structure and essentially the same circuit parameters.

[0137] Reference Figure 1a In the circuit topology shown, when N is odd, the N power conversion circuits are controlled by N sets of control signals. Each set of control signals controls one power conversion circuit, and the corresponding control signals in the N sets are sequentially out of phase by any value between [360 / N-20, 360 / N+20] degrees. When N is even, the N power conversion circuits are controlled by N / 2 sets of control signals. Each set of control signals controls two power conversion circuits, for example, the nth set of control signals controls the nth and N / 2+nth power conversion circuits. The corresponding control signals in the N / 2 sets are sequentially out of phase by any value between [360 / N-20, 360 / N+20] degrees, where n is an integer greater than 1 and less than N. (Refer to...) Figure 2a The circuit topology shown has N power conversion circuits controlled by N sets of control signals. Each set of control signals is used to control each power conversion circuit. The corresponding control signals in the N sets of control signals are out of phase by an arbitrary value between [360 / 2N-20, 360 / 2N+20] degrees.

[0138] The following example illustrates the use of two power conversion circuits connected in parallel with alternating connections. Figure 3 This is a schematic diagram of the circuit structure of the power converter according to the first preferred embodiment of this invention. The power converter 100 of this embodiment includes two... Figure 1a The power conversion circuit 1 shown has its first terminals connected in parallel and its second terminals also connected in parallel.

[0139] At Figure 3 In the illustrated embodiment, each of the two power conversion circuits of the power converter 100 contains a first capacitor C1 and a second capacitor C2. However, in other embodiments, the first terminals of the two power conversion circuits 1 may share a single first capacitor C1, and the second terminals of the two power conversion circuits 1 may share a single second capacitor C2.

[0140] Figure 4 This is a schematic diagram of the circuit structure of the power converter according to the second preferred embodiment of this invention. The power converter 110 of this embodiment includes two... Figure 2a The power conversion circuit 2 shown has its first terminals connected in parallel and its second terminals also connected in parallel.

[0141] At Figure 4In the illustrated embodiment, the two power conversion circuits of the power converter 110 each contain a first capacitor C1 and a second capacitor C2. However, in other embodiments, the first terminals of the two power conversion circuits 1 may share a single first capacitor C1, and the second terminals of the two power conversion circuits 1 may share a single second capacitor C2.

[0142] In summary, this invention provides a power conversion circuit and a power converter applicable thereto. The magnetic component in the power conversion circuit includes a second winding that is electromagnetically coupled to the first winding, in addition to the first winding. Therefore, the voltage gain ratio of the power conversion circuit can be flexibly designed by changing the number of turns of the second winding, based on the turns ratio of N:1 between the second winding and the first winding, thus expanding the application range of the power conversion circuit.

[0143] It should be noted that the above are merely preferred embodiments for illustrating this case, and this case is not limited to the described embodiments. The scope of this case is determined by the appended claims. Furthermore, this case can be modified in various ways by those skilled in the art, but all such modifications will not depart from the protection sought by the appended claims.

Claims

1. A power conversion circuit, characterized by, Include: The first end includes a first positive electrode and a first negative electrode; A second terminal includes a second positive electrode and a second negative electrode, wherein the second negative electrode is electrically connected to the first negative electrode; A first switching power conversion unit includes a first switch and a third switch connected in series; A second switching power conversion unit includes a second switch and a fourth switch connected in series. A first terminal of the first switch is electrically connected to a first terminal of the second switch, and a second terminal of the first switch is electrically connected to a first positive terminal. A third switch is connected in series with the first switch, and a fourth switch is connected in series with the second switch. A first terminal of the third switch and a first terminal of the fourth switch are electrically connected to a first negative terminal, and a second terminal of the fourth switch is electrically connected to a second terminal of the second switch. The first switch, the second switch, the third switch, and the fourth switch operate periodically according to a switching cycle, which includes a duty cycle. A flying capacitor; and A magnetic component includes two first windings and a second winding. The two first windings and the second winding are electromagnetically coupled to each other. The second ends of the two first windings are opposite ends and are electrically connected to each other and electrically connected to a second positive electrode. One first end of one of the first windings is connected in series to a second end of a third switch. One first end of the other first winding is electrically connected to the second end of a fourth switch and a second end of a second switch. The second winding and the flying capacitor are connected in series between the first end of the first switch and the first end of one of the first windings. The first end of the second winding and the first end of one of the first windings are the same end. The turns ratio of the second winding and the two first windings is N:1:1, where N is a positive number. The switching cycle includes two working intervals: a first working interval and a second working interval. In the first working interval, the current flowing through the second winding is equal to the current flowing through one of the first windings. In the second working interval, the current flowing through the second winding is equal to the current flowing through the other first winding.

2. The power conversion circuit as described in claim 1, characterized in that, The power conversion circuit is a bidirectional power conversion circuit. The first terminal of the power conversion circuit is an input terminal or an output terminal, and the second terminal of the power conversion circuit is an output terminal or an input terminal.

3. The power conversion circuit as described in claim 1, characterized in that, The first end of the second winding is electrically connected to the first end of one of the first windings, and the second end of the second winding is electrically connected to one end of the flying capacitor; or one end of the flying capacitor is electrically connected to the first end of one of the first windings, and the other end of the flying capacitor is electrically connected to the first end of the second winding.

4. The power conversion circuit as described in claim 1, characterized in that, The switching state of the first switch is the same as that of the fourth switch, the switching state of the second switch is the same as that of the third switch, the switching state of the first switch is 180 degrees out of phase with the switching state of the second switch, and the conduction time of the first switch and the second switch is less than or equal to half of the switching cycle and greater than or equal to 0.4 times the switching cycle.

5. The power conversion circuit as described in claim 1, characterized in that, The power conversion circuit has an equivalent resonant inductance that resonates with the flying capacitor. The resulting resonance has a resonant frequency and a resonant period. The switching period of the power conversion circuit is less than or equal to the resonant period, and the switching period is greater than or equal to half of the resonant period.

6. The power conversion circuit as described in claim 5, characterized in that, The equivalent resonant inductance is the leakage inductance of the magnetic component and the parasitic inductance of the circuit; or the equivalent resonant inductance is at least one external inductor connected in series between the first terminal of the first switch and the second positive terminal.

7. The power conversion circuit as described in claim 1, characterized in that, The two first windings and the second winding are wound on the same magnetic post of the same magnetic core assembly, and the coupling coefficient between each pair of the two first windings and the second winding is greater than 0.

9.

8. The power conversion circuit as described in claim 1, characterized in that, During the switching cycle, the flying capacitor synchronously stores energy and transfers the energy to the output terminal of the power conversion circuit as the corresponding switch is turned on and off. The flying capacitor has a DC voltage, and the ratio of the DC voltage to the terminal voltage of the first terminal of the power conversion circuit is between 0.4 and 0.

6.

9. The power conversion circuit as described in claim 1, characterized in that, The voltage gain ratio of the terminal voltages of the first and second terminals of the power conversion circuit is (4-2N):

1.

10. A power conversion circuit, characterized in that, Include: A first end includes a first positive electrode and a first negative electrode; A second terminal includes a second positive electrode and a second negative electrode, wherein the second negative electrode is electrically connected to the first negative electrode; The first flying capacitor; A second flying capacitor; A first switching power conversion unit includes a first switch, a second switch and a third switch. A first terminal of the first switch is electrically connected to the first positive terminal, a second terminal of the second switch is electrically connected to a first terminal of the third switch, and a second terminal of the third switch is electrically connected to the second negative terminal. A second switching power conversion unit includes a fourth switch, a fifth switch and a sixth switch. A first terminal of the fourth switch is electrically connected to the first positive terminal, a second terminal of the fifth switch is electrically connected to the first terminal of the sixth switch, a second terminal of the sixth switch is electrically connected to the second negative terminal, and the second terminal of the fourth switch is electrically connected to the first terminal of the second switch, and the first terminal of the fifth switch is electrically connected to the second terminal of the first switch. The first switch, the second switch, the third switch, the fourth switch, the fifth switch, and the sixth switch operate periodically according to a switching cycle, the switching cycle including a duty cycle; and A magnetic component includes two first windings and two second windings, which are electromagnetically coupled to each other. The second ends of the two first windings are opposite to each other and electrically connected to each other and to a second positive terminal. One first winding's first end is electrically connected to the second terminal of a fifth switch and the first terminal of a sixth switch. The other first winding's first end is electrically connected to the second terminal of a second switch and the first terminal of a third switch. One of the second windings is connected in series with a second flying capacitor. Between the second terminal of the fourth switch and the second terminal of the fifth switch, another second winding and the first flying capacitor are connected in series between the second terminal of the first switch and the second terminal of the second switch. One first terminal of one of the second windings and one first terminal of one of the first windings are the same terminal and are connected. One first terminal of the other second winding and one first terminal of the other first winding are the same terminal and are connected. The turns ratio of the two second windings and the two first windings is N:N:1:1, where N is a positive number. The switching cycle includes two working intervals: a first working interval and a second working interval. In the first working interval, the sum of the currents flowing through the two second windings is equal to the current flowing through one of the first windings. In the second working interval, the sum of the currents flowing through the two second windings is equal to the current flowing through the other first winding.

11. The power conversion circuit as described in claim 10, characterized in that, The power conversion circuit is a bidirectional power conversion circuit. The first terminal of the power conversion circuit is an input terminal or an output terminal, and the second terminal of the power conversion circuit is an output terminal or an input terminal.

12. The power conversion circuit as described in claim 10, characterized in that, One of the second windings has a second terminal electrically connected to the second flying capacitor, and the other of the second windings has a second terminal electrically connected to the first flying capacitor.

13. The power conversion circuit as described in claim 10, characterized in that, One end of the second flying capacitor is electrically connected to the first end of one of the first windings, the other end of the second flying capacitor is electrically connected to the first end of one of the second windings, and one end of the first flying capacitor is electrically connected to the first end of the other first winding, and the other end of the first flying capacitor is electrically connected to the first end of the other second winding.

14. The power conversion circuit as described in claim 10, characterized in that, The switching states of the first switch, the second switch, and the sixth switch are the same. The switching states of the fourth switch, the fifth switch, and the third switch are the same. The switching states of the first switch and the fourth switch are out of phase by 180 degrees. The conduction time of the first switch and the fourth switch is less than or equal to half of the switching cycle and greater than or equal to 0.4 times the switching cycle.

15. The power conversion circuit as described in claim 10, characterized in that, The power conversion circuit has an equivalent resonant inductance that resonates with the first flying capacitor and the second flying capacitor. The resulting resonance has a resonant frequency and a resonant period. The switching period of the power conversion circuit is less than or equal to the resonant period, and the switching period of the power conversion circuit is greater than or equal to half of the resonant period.

16. The power conversion circuit as described in claim 15, characterized in that, The equivalent resonant inductance is the leakage inductance of the magnetic component and the parasitic inductance of the circuit, or the equivalent resonant inductance is at least one external inductor connected in series between the first terminal of the first switch and the second positive terminal.

17. The power conversion circuit as described in claim 10, characterized in that, The two first windings and the two second windings are wound on the same magnetic post of the same magnetic core assembly, and the coupling coefficient between each pair of the two first windings and the two second windings is greater than 0.

9.

18. The power conversion circuit as described in claim 10, characterized in that, During the switching cycle, the first flying capacitor and the second flying capacitor synchronously store energy and transfer the energy to the output terminal of the power conversion circuit as the corresponding switch is turned on and off. Both the first flying capacitor and the second flying capacitor have a DC voltage, and the ratio of each DC voltage to the terminal voltage of the first terminal of the power conversion circuit is between 0.4 and 0.

6.

19. The power conversion circuit as described in claim 10, characterized in that, The voltage gain ratio of the first and second terminals of the power conversion circuit is (4-2N):

1.

20. A power converter, characterized in that, Include: N power conversion circuits as described in any one of claims 1-9, wherein the first terminals of the N power conversion circuits are connected in parallel, and the second terminals of the N power conversion circuits are connected in parallel, wherein when N is greater than 1 and is odd, the N power conversion circuits are controlled by N sets of control signals, each set of control signals is used to control each power conversion circuit, and the corresponding control signals in the N sets of control signals are sequentially out of phase by any value between [360 / N-20, 360 / N+20] degrees; wherein when N is greater than 1 and is even, the N power conversion circuits are controlled by N / 2 sets of control signals, the nth set of control signals is used to control the nth and N / 2+nth power conversion circuits, and the corresponding control signals in the N / 2 sets of control signals are sequentially out of phase by any value between [360 / N-20, 360 / N+20] degrees.

21. A power converter, characterized in that, Include: N power conversion circuits as described in any one of claims 10-19, wherein the first terminals of the N power conversion circuits are connected in parallel, and the second terminals of the N power conversion circuits are connected in parallel, wherein N is greater than 1, and the N power conversion circuits are controlled by N sets of control signals, each set of control signals being used to control each power conversion circuit, wherein the corresponding control signals in the N sets of control signals are sequentially out of phase by an arbitrary value between [360 / 2N-20, 360 / 2N+20] degrees.

Images