Wireless Transmitter

Abstract

To provide a radio transmission device and a radio reception device capable of reducing error rate degradation.SOLUTION: In a radio transmission device, an OFDM signal forming unit forms an OFDM signal by arranging pilot signals with a pilot arrangement pattern. In the pilot arrangement pattern, pilot signals for phase noise estimation are arranged onto all resource elements in a plurality of first pilot arrangement subcarriers having a subcarrier interval of NPilot_SPA, within a plurality of pilot arrangement subcarriers having a subcarrier interval of NPilot_Freq. Information symbols and pilot symbols are arranged onto different resource elements. In a plurality of second pilot arrangement subcarriers except the plurality of first pilot arrangement subcarriers, within the plurality of pilot arrangement subcarriers having the subcarrier interval of NPilot_Freq, the pilot signals for phase noise estimation are arranged with a resource element period of NPilot_Time pieces on time domain.SELECTED DRAWING: Figure 5

Description

[Technical Field]

[0001] This disclosure relates to a wireless transmitter and a wireless receiver. [Background technology]

[0002] With the rapid spread of 4G Long Term Evolution (LTE) and LTE-Advanced, the provision of full-fledged mobile broadband services has become feasible. To cope with the rapidly increasing traffic on cellular networks, the fifth-generation (5G) New Radio (NR) system requires even greater ultra-high speed and capacity, as well as increased frequency utilization efficiency, compared to LTE. In addition to heterogeneous networks that overlay small cells to efficiently accommodate the uneven traffic within macrocells, highly efficient wireless access technology is required. In addition to an ultra-high-speed, high-capacity wireless access network that enables gigabit-class services to user equipment (UE), even greater ultra-high speed and capacity are required for the backhaul between base stations and the Serving-Gateway (S-GW) of the Evolved Packet Core (EPC) network. Backhaul links consist of E1 / T1 dedicated lines, fiber optic networks, microwave wireless backhaul, etc. Wireless backhaul has the advantage of lower network costs compared to wired backhaul. The same applies to a configuration that includes a base station composed of remote radio equipment (RRE) and a central base station that processes the baseband physical layer and higher layers, specifically in the case of a fronthaul connecting the RRE and the central base station.

[0003] In wireless backhaul, the millimeter-wave frequency spectrum is applied. In the 5G NR system, the application of the millimeter-wave frequency spectrum, which can secure a wide bandwidth for access links, is being considered. In Japan, the 28 GHz frequency spectrum has been allocated. In the millimeter-wave frequency spectrum, phase noise caused by frequency fluctuations in the local oscillators of base stations and user terminals can degrade the error rate. Local oscillators (frequency synthesizers) with very high frequency stability are expensive. Therefore, the general approach is to estimate and compensate for the phase noise generated in the transmitter's local oscillator through signal processing during transmission or reception. Similarly, the general approach is to estimate and compensate for the phase noise generated in the receiver's local oscillator through signal processing during reception.

[0004] The phase noise generated in the local oscillator is modeled using a discrete-time Wiener process. The phase noise at sample timing k is θ k =θ k-1 +Δ k It is represented by Δ k The mean is 0, and the variance is 0.

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[0005] A pilot signal is necessary to estimate phase noise. The modulation phase and amplitude of this pilot signal are known by the receiver. Phase noise varies with each sample timing in the time domain. Therefore, it is desirable to continuously multiplex the pilot signal for all symbols in the time domain. On the other hand, continuously multiplexing the pilot signal for all symbols increases the insertion loss (i.e., overhead) of the pilot signal, leading to a deterioration in frequency utilization efficiency.

[0006] In addition to phase noise, waveform distortion due to multipath fading is a major factor in the degradation of error rates in the millimeter-wave frequency spectrum. Wireless access links are primarily characterized by non-line-of-sight (NLOS) propagation paths. In NLOS propagation paths, each path with a different delay time is modeled by a propagation path that undergoes independent Rayleigh fading. The amplitude and phase of a signal subjected to Rayleigh fading fluctuate randomly. It is known that the amplitude can be approximated by a Rayleigh distribution, and the phase can be approximated by a uniform distribution. When multiple paths with different delay times are combined, frequency-selective fading occurs, which has time-varying frequency selectivity. On the other hand, wireless backhaul links are generally line-of-sight (LOS) propagation paths where the direct wave exists. In wireless backhaul links with high antenna heights and long site-to-site distances, in addition to the direct wave, delayed waves generated by reflections from the ground, etc., are received. Multipath fading (i.e., frequency-selective fading) occurs due to these delayed waves. When the distance between sites is long, the delay time of delayed waves due to reflection from the ground is short compared to the symbol length. As a result, a large notch occurs in the frequency domain. Furthermore, in the backhaul link between macrocell sites and small cell sites in a heterogeneous network (HetNet), the antenna height at the small cell site is low. Therefore, in addition to the direct wave, delayed waves with large delay times are received due to reflection and scattering from buildings and other objects around the small cell site, and multipath fading occurs due to these delayed waves. As described above, both wireless access links and wireless backhaul links experience frequency-selective multipath fading. For this reason, an equalizer is required in the receiver to compensate for waveform distortion caused by inter-symbol interference (ISI). Equalizers are classified into time-domain equalizers (TDEs) and frequency-domain equalizers (FDEs).

[0007] In wireless backhaul, Time Deposition Experiments (TDEs) have generally been commonly used. TDEs can be implemented using transversal filters or Finite Impulse Response (FIR) filters, as shown in Figure 1. Transversal filters with a number of taps greater than or equal to the maximum delay time of the delayed wave are used for discrete-time sampling. The weight coefficients (equalization weights) of the transversal filter are updated using an adaptive algorithm for the time-varying delayed wave. The Minimum Mean Square Error (MMSE) criterion is used to control the weight coefficients. TDEs require a number of taps with a time range sufficiently long compared to the maximum delay time of the delayed wave (multipath). As shown in Figure 1, TDEs require convolution, which involves complex multiplication by the number of taps for each sample value. Therefore, as the maximum delay time of the delayed wave increases, the number of taps increases, and the computational load of the convolution becomes enormous.

[0008] To reduce the computational complexity of time-domain equalizers, FDEs have been proposed (e.g., Non-Patent Document 1). Figure 2 shows the configuration of an FDE. A time-domain received signal is converted into a frequency-domain signal by a Discrete Fourier Transform (DFT) or a Fast Fourier Transform (FFT). The number of samples in the time-domain FFT corresponds to the number of subcarriers in the frequency-domain signal. In this specification, the frequency components obtained when a single-carrier signal is converted into a frequency-domain signal by FFT are sometimes referred to as "subcarriers". Then, each subcarrier component in the frequency domain is multiplied by an equalization weight (weighting coefficient). Then, the complex channel response at subcarrier k is expressed as h k When expressed in this way, the equalization weights for the mean squared error minimum (MMSE) standard are given by the following equation (1):

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[0009] The equalized signal is converted into a time-domain signal by an Inverse Discrete Fourier Transform (IDFT) or an Inverse Fast Fourier Transform (IFFT). Although FDE requires FFT (DFT) and IFFT (IDFT), the equalization process at each sub-carrier position is realized by multiplication processing. Therefore, the overall computational complexity can be reduced compared to TDE. Thus, in the single-carrier FDMA (Frequency Division Multiple Access) of the uplink of LTE, a radio interface assuming the application of FDE is adopted.

[0010] Both TDE and FDE require a pilot signal for channel estimation. In the 3GPP (registered trademark) NR radio interface using OFDMA, a reference signal for demodulation (including equalization) and a reference signal for reception quality measurement are defined. The reference signal for demodulation (including equalization) is called a Demodulation Reference Signal (DM-RS), and the reference signal for reception quality measurement is called a Channel State Information Reference Signal (CSI-RS).

[0011] Furthermore, NR radio interfaces specify a Phase Tracking Reference Signal (PTRS) for estimating phase noise and frequency offset in OFDMA. Figure 3 shows the PTRS multiplexing method in an NR radio interface. This PTRS multiplexing method is disclosed in Non-Patent Literature 2. In the frequency domain, PTRS are multiplexed at a minimum interval of 12 subcarriers. Phase noise varies over time on an OFDM sample-by-sample basis. Therefore, in the time domain, PTRS are densely multiplexed. Four thresholds are specified to determine the time-domain multiplexing density of PTRS according to the modulation scheme and coding scheme (MCS). MCS Index I MCS If ptrs-MCS1 is lower than ptrs-MCS1, PTRS will not be duplicated. Also, ptrs-MCS1 ≤ I MCS If ptrs-MCS2 ≤ I, then PTRS is multiplexed for every 4 OFDM symbols. Also, ptrs-MCS2 ≤ I MCS If ptrs-MCS3 ≤ I, then PTRS is multiplexed for every 2 OFDM symbols. Also, ptrs-MCS3 ≤ I MCS In the case of ≤ptrs-MCS4, it is specified that PTRS is multiplexed at all OFDM symbol positions consecutively in the time domain.

[0012] DFT-spread OFDM, which performs frequency allocation (mapping) of single-carrier FDMA signals using frequency domain processing, is employed in the uplink of LTE. In NR radio interfaces, DFT-spread OFDM is used when transmitting one stream on the uplink. Because DFT-spread OFDM is a single-carrier system, it has the advantage of being able to lower the peak-to-average power ratio (PAPR) compared to OFDMA. Therefore, in DFT-spread OFDM, the transmit backoff of the power amplifier (PA) can be set lower, resulting in a higher average power. Consequently, when using the same PA with the same peak power, DFT-spread OFDM has a longer radio wave range and a wider coverage area compared to OFDMA. In DFT-spread OFDM, to achieve a low PAPR, pilot symbols are usually time-division multiplexed (TDM) between information symbols. Assuming an insertion loss of about 10% for pilot symbols, one pilot symbol would be multiplexed for every 10 symbols. When the symbol length is short, it can adequately track propagation path variations caused by multipath fading. Therefore, TDM is used in DMRS for LTE and NR radio interfaces. However, TDM-type pilot symbols with an insertion loss of about 10% cannot adequately track phase noise. To enable tracking of phase noise, pilot symbols need to be multiplexed more densely in the time domain. However, TDM-type pilot symbol multiplexing significantly increases insertion loss, drastically degrading frequency utilization efficiency.

[0013] Therefore, in single-carrier DFT-spread OFDM, a frequency division multiplexing (FDM) type pilot symbol multiplexing that is continuously multiplexed in the time direction has been proposed (Non-Patent Literature 3). Figure 4A shows the frequency domain superimposed pilot technique (FDSPT), and Figure 4B shows the frequency expanding technique (FET). In FDSPT, pilot symbols are superimposed on information symbols at a constant tone interval in the frequency domain. Here, the tone corresponds to the subcarrier in OFDMA. The pilot symbols are multiplexed onto consecutive symbols in the time domain. Since the pilot symbols are multiplexed onto the same resource element (RE) as the information symbols, the bandwidth efficiency of the information symbols does not decrease. On the other hand, interference from the pilot symbols leads to a slight degradation in the error rate of the information symbols and an increase in PAPR. The degradation in the error rate of the information symbols can be kept low by applying error correction coding. Therefore, interference from information symbols to pilot symbols actually degrades the accuracy of phase noise estimation. To address this, methods have been proposed to reduce the transmission power of information symbols at tone positions where pilot symbols are superimposed. In FETs, pilot symbols and information symbols are orthogonally multiplexed, so they do not interfere with each other; however, the insertion loss of pilot symbols degrades the bandwidth efficiency of information symbols. Both FDSPT and FETs result in an increase in PAPR compared to TDM-type pilot symbol multiplexing. [Prior art documents] [Non-patent literature]

[0014] [Non-Patent Document 1] D. Falconer, SL Ariyavisitakul, A. Benyamin-Seeyar, and B. Eidson, “Frequency domain equalization for single-carrier broadband wireless systems,” IEEE Commun. Mag., vol 40, no. 4, pp. 58-66, April 2002. [Non-Patent Document 2] 3GPP TS 38.211, NR; Physical channels and modulation (Release 16), V16.7.0, Sept. 2021. [Non-Patent Document 3] C.-T. Lam, DD Falconer, F. Danilo-Lemoine, and R. Dinis, “Channel Estimation for SC-FDE Systems Using Frequency Domain Multiplexed Pilots,” Proc. IEEE VTC2006-Fall, Sept. 2006. [Non-Patent Document 4] RA Casas, SL Biracree, and AE Youtz, “Time domain phase noise correction for OFDM signals,” IEEE Transactions on Broadcasting, vol. 48, no. 3, pp. 230 - 236, Sept. 2002. [Non-Patent Document 5] U. Sorger, I. De Broeck, and M. Schnell, “Interleaved FDMA - a new spread-spectrum multiple-access scheme,” Proc. 1998 IEEE International Conference on Communications. Conference, ICC '98, June 1998. [Non-Patent Document 6] DC Chu, “Polyphase codes with good periodic correlation properties,” IEEE Trans. Inform. Theory, vol. IT-18, pp. 531 - 532, July 1972. [Non-Patent Document 7] S. Suyama, J. Onodera, H. Suzuki, and K. Fukawa, “Decision-directed phase noise compensation for millimeter-wave single carrier transmission systems with frequency-domain equalization,” Proc. 2009 European Wireless Technology Conference. [Non-Patent Document 8] N. Kamiya and E. Sasaki, “Pilot-Symbol Assisted and Code-Aided Phase Error Estimation for High-Order QAM Transmission,” IEEE Trans. on Commun., vol. 61, no. 10, pp. 4369-4380, Oct. 2013. [Non-Patent Document 9] S. Wu and Y. Bar-Ness, “A phase noise suppression algorithm for OFDM-based WLANs,” IEEE Commun. Lett., vol. 6, no. 12, pp. 535-537, Dec. 2002. [Non-Patent Document 10] D. Petrovic, W. Rave, and G. Fettweis, “Effects of phase noise in OFDM systems with and withoutput PLL: characterization and compensation,” IEEE Trans. on Commun., vol. 55, no. 8, pp. 1607-1616, Aug. 2007. [Overview of the project] [Problems that the invention aims to solve]

[0015] The LTE radio interface specifies a maximum system bandwidth of 20 MHz (transmit or receive bandwidth at the base station). LTE-Advanced (3GPP Release 11 specification), an advanced version of LTE, specifies a system bandwidth of up to 100 MHz through carrier aggregation (CA), which combines multiple component carriers (CCs) using the LTE system bandwidth as a component carrier.

[0016] The NR radio interface scalably extends the subcarrier spacing of LTE, defining a system bandwidth of up to 400 MHz. While the allocated bandwidth per user terminal is narrower compared to the system bandwidth, it will increase as the system bandwidth widens. The 5G system is expected to use the millimeter-wave frequency spectrum, which can secure a wide bandwidth, and the allocation of 28 GHz frequencies to the four carriers in Japan has already been completed. Furthermore, the application of even higher frequencies is anticipated, with the Beyond 5G (6G) system anticipating the application of frequency spectra up to 1 THz. The wireless backhaul link already uses the millimeter-wave frequency spectrum, and the application of higher frequency spectra, which can secure even wider bandwidths, is anticipated.

[0017] The main factors contributing to the degradation of the error rate in the millimeter-wave frequency spectrum are waveform distortion caused by multipath interference from delayed waves, and phase noise caused by frequency fluctuations and jitter (time-domain fluctuations) occurring in local oscillators at base stations and user terminals.

[0018] A pilot signal is used to estimate the phase noise generated in a local oscillator. Since the phase noise fluctuates on a sample-by-sample basis for OFDM signals and on a single-carrier symbol basis for DFT-spread OFDM signals, the pilot signal needs to be densely multiplexed in the time domain. Numerous phase noise estimation and compensation methods for time-domain or frequency-domain processing have been published (e.g., Non-Patent Documents 4-10).

[0019] Each symbol is subject to waveform distortion caused by multipath interference from preceding symbols, i.e., inter-symbol interference. Time-domain equalizer (TDE) or frequency-domain equalizer (FDE) are used to equalize the waveform distortion caused by inter-symbol interference. For updating the equalization weight coefficients of TDE, algorithms such as the LMS algorithm or RLS algorithm are used. The judgment value of the information symbol is used as the reference signal to generate the error signal of the LMS or RLS algorithm. However, since information symbols are prone to decoding errors, it is necessary to periodically multiplex pilot signals between information symbols in order to operate the LMS or RLS algorithm with high accuracy. In addition, MMSE equalization weights are used in FDE, and these MMSE equalization weights are generated from the channel response of each subcarrier (tone) position estimated by the pilot signal.

[0020] When using a single-carrier signal, or an OFDM with a large subcarrier spacing, a broadband pilot signal for phase noise estimation will result in the pilot signal being subjected to phase noise and frequency-selective multipath fading. Using a pilot signal subject to phase noise fluctuations makes it impossible to generate equalization weights with high accuracy. Conversely, using a pilot signal subject to multipath fading makes it impossible to estimate phase noise with high accuracy. Therefore, it is necessary to repeatedly perform phase noise estimation / compensation and equalization processes alternately multiple times. As a result, the computational load of these repeated phase noise estimation / compensation and equalization processes increases significantly in order to keep phase noise and residual equalization at sufficiently low levels.

[0021] The object of this disclosure is to provide a wireless transmitter and a wireless receiver that can separate phase noise estimation from multipath fading and reduce the degradation of the error rate by using a narrowband pilot signal for phase noise estimation. [Means for solving the problem]

[0022] A wireless transmitter according to a first aspect of this disclosure includes a signal forming unit that forms an OFDM (Orthogonal Frequency Division Multiplexing) signal including a pilot signal, or a DFT-spread OFDM signal, A transmitting radio unit that forms a radio signal from the OFDM signal or DFT-spread OFDM signal formed above, It is equipped with, The signal forming unit, Of the total subcarriers of the OFDM signal or DFT-spread OFDM signal, N Pilot_Freq The pilot signals are placed on multiple pilot-placed subcarriers at subcarrier intervals, Among the aforementioned multiple pilot-placed subcarriers, N Pilot_SPAIn a subcarrier with multiple first pilot placement subcarriers at subcarrier intervals, each resource element places the pilot signal on all resource elements defined by the combination of subcarrier and OFDM symbol. Among the plurality of pilot-configured subcarriers, the plurality of second pilot-configured subcarriers excluding the plurality of first pilot-configured subcarriers, N in the time domain Pilot_Time The pilot signals are arranged in a period of one resource element. The pilot signals are arranged according to the pilot placement pattern.

[0023] A wireless transmitter according to a second aspect of this disclosure is N TxAnt A single transmitting antenna, A signal forming unit that forms an OFDM (Orthogonal Frequency Division Multiplexing) signal including a pilot signal, or a DFT-spread OFDM signal, A transmitting radio unit that forms a radio signal from the OFDM signal or DFT-spread OFDM signal formed above, It is equipped with, The signal forming unit, Of the total subcarriers of the OFDM signal or DFT-spread OFDM signal, each set is N TxAnt It contains a set of consecutive subcarriers, and the interval between each set of two adjacent subcarriers is N. Pilot_Freq The pilot signals are placed on a set of multiple pilot-placed subcarriers, which are subcarriers. Among the aforementioned multiple pilot placement subcarrier sets, N Pilot_SPA In a set of multiple first pilot-placed subcarriers with subcarrier spacing, each resource element is defined by a combination of a subcarrier and an OFDM symbol or a single-carrier symbol block, and the pilot signal is placed on all resource elements. In the plurality of second pilot-configured subcarrier sets, excluding the plurality of first pilot-configured subcarrier sets, N in the time domain Pilot_Time The pilot signals are arranged in a period of one resource element. The pilot signals are arranged according to the pilot placement pattern. The aforementioned N TxAnt The multiple pilot signals transmitted from each of the transmitting antennas are spread out by a spread sequence specific to each of the different transmitting antennas.

[0024] A wireless transmitter according to a third aspect of this disclosure includes a signal forming unit that forms an OFDM (Orthogonal Frequency Division Multiplexing) signal including a pilot signal, or a DFT-spread OFDM signal, A transmitting radio unit that forms a radio signal from the OFDM signal or DFT-spread OFDM signal formed above, It is equipped with, The signal forming unit, Of the total subcarriers of the OFDM signal or DFT-spread OFDM signal, N Pilot_Freq The pilot signals are placed on multiple pilot-placed subcarriers at subcarrier intervals, In each pilot-configured subcarrier, N in the time domain Pilot_Time The pilot signals are arranged in a cycle of the resource element, In each of two adjacent pilot placement subcarriers, the pilot signals are arranged such that the resource elements on which the pilot signals are placed do not overlap in the time domain. The pilot signals are arranged according to the pilot placement pattern.

[0025] A wireless transmitter according to the fourth aspect of this disclosure is N TxAnt A single transmitting antenna, A signal forming unit that forms an OFDM (Orthogonal Frequency Division Multiplexing) signal including a pilot signal, or a DFT-spread OFDM signal, A transmitting radio unit that forms a radio signal from the OFDM signal or DFT-spread OFDM signal formed above, It is equipped with, The signal forming unit, Of the total subcarriers of the OFDM signal or DFT-spread OFDM signal, each set is N TxAnt It contains a set of consecutive subcarriers, and the interval between each set of two adjacent subcarriers is N. Pilot_Freq The pilot signals are placed on a set of multiple pilot-placed subcarriers, which are subcarriers. In each pilot-configured subcarrier set, N in the time domain Pilot_Time The pilot signals are arranged in a cycle of the resource element, In each of two adjacent pilot placement subcarrier sets, the pilot signals are arranged such that the resource elements on which the pilot signals are placed do not overlap in the time domain. The pilot signals are arranged according to the pilot placement pattern. The aforementioned N TxAnt The multiple pilot signals transmitted from each of the transmitting antennas are spread out by a spread sequence specific to each of the different transmitting antennas.

[0026] A wireless receiver according to a fifth aspect of this disclosure comprises a signal processing unit that performs signal processing on an OFDM (Orthogonal Frequency Division Multiplexing) signal or a DFT-spread OFDM signal, The signal processing unit, A phase noise compensation unit that estimates the phase noise of the received OFDM signal or DFT-spread OFDM signal based on a pilot signal contained in the received OFDM signal or DFT-spread OFDM signal and compensates for the phase noise of the received OFDM signal or DFT-spread OFDM signal, A conversion unit that receives a first time-domain signal and converts the received first time-domain signal into a frequency-domain signal, An equalization unit that performs equalization processing on the frequency domain signal obtained by the conversion unit, A demapping unit that demmaps the information symbols contained in the frequency domain signal after the equalization process and reconstructs the information bits, An extraction unit that extracts a pilot signal included in the frequency domain signal obtained by the conversion unit, A channel estimation unit that estimates the channel response using the extracted pilot signal, A symbol estimation unit that maps the regenerated information bits to symbols to obtain regenerated symbols, A first multiplication unit multiplies the obtained regenerated symbol by the estimated channel response, An inverse transformer unit that converts the regenerated symbols multiplied by the channel response into a second time-domain signal, A phase noise compensation signal generation unit that estimates the phase noise using the first time-domain signal and the second time-domain signal obtained by the inverse transform unit and outputs the inverse characteristics of the estimated phase noise, A second multiplier is provided at the input stage of the conversion unit, which compensates for the phase noise of the received OFDM signal or DFT-spread OFDM signal after the phase noise has been compensated by the phase noise compensation unit by multiplying the inverse characteristic of the output phase noise by the received OFDM signal or DFT-spread OFDM signal, and outputs the received OFDM signal or DFT-spread OFDM signal after phase noise compensation to the conversion unit as the first time-domain signal. Includes.

[0027] A wireless receiver according to a sixth aspect of this disclosure comprises a signal processing unit that performs signal processing on an OFDM (Orthogonal Frequency Division Multiplexing) signal or a DFT-spread OFDM signal, The signal processing unit, A phase noise compensation unit that estimates the phase noise of the received OFDM signal or DFT-spread OFDM signal based on a pilot signal contained in the received OFDM signal or DFT-spread OFDM signal and compensates for the phase noise of the OFDM signal or DFT-spread OFDM signal, A conversion unit that receives a first time-domain signal and converts the received first time-domain signal into a frequency-domain signal, An equalization unit that performs equalization processing on the frequency domain signal obtained by the conversion unit, A demapping unit that demmaps the information symbols contained in the frequency domain signal after the equalization process and calculates the reliability information of each bit in each symbol, A decoding unit that performs error correction decoding using the confidence information of each bit and outputs the obtained decoded bits, An extraction unit that extracts a pilot signal included in the frequency domain signal obtained by the conversion unit, A channel estimation unit that estimates the channel response using the extracted pilot signal, A symbol estimation unit maps the decoded bits output from the decoding unit to symbols to obtain a reconstructed symbol, A first multiplication unit multiplies the obtained regenerated symbol by the estimated channel response, An inverse transformer unit that converts the regenerated symbols multiplied by the channel response into a second time-domain signal, A phase noise compensation signal generation unit that estimates the phase noise using the first time-domain signal and the second time-domain signal obtained by the inverse transform unit and outputs the inverse characteristics of the estimated phase noise, A second multiplier is provided at the input stage of the conversion unit, which compensates for the phase noise of the received OFDM signal or DFT-spread OFDM signal after the phase noise has been compensated by the phase noise compensation unit by multiplying the inverse characteristic of the output phase noise by the received OFDM signal or DFT-spread OFDM signal, and outputs the received OFDM signal or DFT-spread OFDM signal after phase noise compensation to the conversion unit as the first time-domain signal. Includes.

[0028] A wireless receiver according to a seventh aspect of this disclosure comprises a signal processing unit that performs signal processing on an OFDM (Orthogonal Frequency Division Multiplexing) signal or a DFT-spread OFDM signal, The signal processing unit, A phase noise compensation unit that estimates the phase noise of the received OFDM signal or DFT-spread OFDM signal based on a pilot signal contained in the received OFDM signal or DFT-spread OFDM signal and compensates for the phase noise of the received OFDM signal or DFT-spread OFDM signal, A first conversion unit that receives a first time-domain signal and converts the received first time-domain signal into a first frequency-domain signal, An equalization unit that performs an equalization process on the first frequency domain signal obtained by the first conversion unit, A first inverse transformer converts the frequency domain signal after the equalization process into an information symbol, which is a second time domain signal. A demapping unit that demmaps the aforementioned information symbols and reconstructs the information bits, An extraction unit that extracts a pilot signal included in the first frequency domain signal obtained by the first conversion unit, A channel estimation unit that estimates the channel response using the extracted pilot signal, A symbol estimation unit that maps the regenerated information bits to symbols to obtain regenerated symbols, A second conversion unit that converts the aforementioned regenerated symbol into a symbol which is a second frequency domain signal, A mapping unit that maps the symbol, which is the second frequency domain signal, to a subcarrier, A first multiplication unit multiplies the symbols mapped to the subcarriers by the estimated channel response, A second inverse transform unit that converts the symbol multiplied by the channel response into a third time-domain signal, A phase noise compensation signal generation unit that estimates the phase noise using the first time-domain signal and the third time-domain signal obtained by the second inverse transform unit, and outputs the inverse characteristics of the estimated phase noise, A second multiplier is provided at the input stage of the first conversion unit, which compensates for the phase noise of the received OFDM signal or DFT-spread OFDM signal after the phase noise has been compensated by the phase noise compensation unit by multiplying the inverse characteristic of the output phase noise by the received OFDM signal or DFT-spread OFDM signal, and outputs the received OFDM signal or DFT-spread OFDM signal after phase noise compensation to the first conversion unit as the first time-domain signal. Includes.

[0029] A wireless receiver according to the eighth aspect of this disclosure comprises a signal processing unit that performs signal processing on an OFDM (Orthogonal Frequency Division Multiplexing) signal or a DFT-spread OFDM signal, The signal processing unit, A phase noise compensation unit that estimates the phase noise of the received OFDM signal or DFT-spread OFDM signal based on a pilot signal contained in the received OFDM signal or DFT-spread OFDM signal and compensates for the phase noise of the received OFDM signal or DFT-spread OFDM signal, A first conversion unit that receives a first time-domain signal and converts the received first time-domain signal into a first frequency-domain signal, An equalization unit that performs an equalization process on the first frequency domain signal obtained by the first conversion unit, A first inverse transformer converts the frequency domain signal after the equalization process into an information symbol, which is a second time domain signal. A demapping unit that demmaps the aforementioned information symbols and calculates the reliability information of each bit in each symbol, A decoding unit that performs error correction decoding using the confidence information of each bit and outputs the obtained decoded bits, An extraction unit that extracts a pilot signal included in the first frequency domain signal obtained by the first conversion unit, A channel estimation unit that estimates the channel response using the extracted pilot signal, A symbol estimation unit maps the decoded bits output from the decoding unit to symbols to obtain a reconstructed symbol, A second conversion unit that converts the aforementioned regenerated symbol into a symbol which is a second frequency domain signal, A mapping unit that maps the symbol, which is the second frequency domain signal, to a subcarrier, A first multiplication unit multiplies the symbols mapped to the subcarriers by the estimated channel response, A second inverse transform unit that converts the symbol multiplied by the channel response into a third time-domain signal, A phase noise compensation signal generation unit that estimates the phase noise using the first time-domain signal and the third time-domain signal obtained by the second inverse transform unit, and outputs the inverse characteristics of the estimated phase noise, A second multiplier is provided at the input stage of the first conversion unit, which compensates for the phase noise of the received OFDM signal or DFT-spread OFDM signal after the phase noise has been compensated by the phase noise compensation unit by multiplying the inverse characteristic of the output phase noise by the received OFDM signal or DFT-spread OFDM signal, and outputs the received OFDM signal or DFT-spread OFDM signal after phase noise compensation to the first conversion unit as the first time-domain signal. Includes.

[0030] A wireless receiver according to the ninth aspect of this disclosure comprises a signal processing unit that performs signal processing on an OFDM (Orthogonal Frequency Division Multiplexing) signal or a DFT-spread OFDM signal, The signal processing unit, A phase noise compensation unit that estimates the phase noise of the received OFDM signal or DFT-spread OFDM signal based on a pilot signal contained in the received OFDM signal or DFT-spread OFDM signal and compensates for the phase noise of the received OFDM signal or DFT-spread OFDM signal, A conversion unit that receives the received OFDM signal or DFT-spread OFDM signal after the phase noise has been compensated by the phase noise compensation unit, and converts the received OFDM signal or DFT-spread OFDM signal into a frequency domain signal, An equalization unit that performs equalization processing on the frequency domain signal obtained by the conversion unit, A demapping unit that demmaps the information symbols contained in the frequency domain signal after the equalization process and calculates the reliability information of each bit in each symbol, A decoding unit that performs error correction decoding using the confidence information of each bit and outputs the obtained decoded bits, A detection unit for detecting the phase difference between the frequency domain signal after the equalization process and the reference signal, A loop filter that outputs the phase difference obtained by reducing the noise component of the detected phase difference, A phase noise compensation processing unit is provided at the input stage of the demapping unit and uses the output phase difference to compensate for the phase noise of the frequency domain signal after the equalization process, A symbol estimation unit that maps the decoded bits to symbols to obtain a reconstructed symbol and outputs the reconstructed symbol as the reference signal, Includes.

[0031] A wireless receiver according to a tenth aspect of this disclosure comprises a signal processing unit that performs signal processing on an OFDM (Orthogonal Frequency Division Multiplexing) signal or a DFT-spread OFDM signal, A phase noise compensation unit that estimates the phase noise of the received OFDM signal or DFT-spread OFDM signal based on a pilot signal contained in the received OFDM signal or DFT-spread OFDM signal and compensates for the phase noise of the received OFDM signal or DFT-spread OFDM signal, A conversion unit that receives the received OFDM signal or DFT-spread OFDM signal after the phase noise has been compensated by the phase noise compensation unit, and converts the received OFDM signal or DFT-spread OFDM signal into a frequency domain signal, An equalization unit that performs equalization processing on the frequency domain signal obtained by the conversion unit, An extraction unit that extracts a pilot signal included in the frequency domain signal obtained by the conversion unit, A channel estimation unit that estimates the channel response using the extracted pilot signal, A demapping unit that demmaps the information symbols contained in the frequency domain signal obtained by the conversion unit and reconstructs the information bits, A symbol estimation unit that maps the regenerated information bits to symbols to obtain regenerated symbols, A calculation unit calculates the discrete Fourier transform coefficients of the phase noise based on the information symbols included in the frequency domain signal obtained by the conversion unit, the estimated channel response, and the reproduced symbols, using the mean square error minimum criterion. Distributed at the input stage of the equalization unit, an inter-carrier interference compensation unit performs convolution processing using the frequency domain signal obtained in the conversion unit and the calculated discrete Fourier transform coefficients to compensate for the phase noise of the received OFDM signal or DFT-spread OFDM signal after the phase noise has been compensated by the phase noise compensation unit, Includes. [Effects of the Invention]

[0032] This disclosure provides a wireless transmitter and a wireless receiver that can reduce the degradation of the error rate. [Brief explanation of the drawing]

[0033] [Figure 1] This is a diagram showing the configuration of a transversal filter. [Figure 2] This diagram shows the configuration of a frequency domain equalizer. [Figure 3] This diagram shows the multiplexing method for phase tracking reference signals in an NR radio interface. [Figure 4A] This is a diagram showing the frequency domain superimposed pilot multiplexing method. [Figure 4B] This is a diagram showing the frequency-extended pilot multiplexing method. [Figure 5] This is a block diagram showing an example of a wireless transmission device in the first embodiment. [Figure 6] This is a block diagram showing an example of an OFDM signal generation unit. [Figure 7] Another example of a block diagram of an OFDM signal generation unit. [Figure 8] This is a diagram illustrating example 1 of the pilot placement pattern. [Figure 9] This figure illustrates example 2 of the pilot placement pattern. [Figure 10] This diagram illustrates example 3 of the pilot placement pattern. [Figure 11] This is a diagram illustrating example 4 of the pilot placement pattern. [Figure 12] This figure illustrates example 5 of the pilot placement pattern. [Figure 13] This figure illustrates an example of CDM pilot signal multiplexing in single-carrier DFT-spread OFDM. [Figure 14] This is a diagram illustrating example 6 of the pilot placement pattern. [Figure 15] This figure illustrates example 7 of the pilot placement pattern. [Figure 16] This figure illustrates example 7 of the pilot placement pattern. [Figure 17] This figure illustrates example 8 of the pilot placement pattern. [Figure 18] This figure shows an example of the basic configuration of a wireless receiver in the second embodiment. [Figure 19] This block diagram shows an example of a phase noise estimation and compensation section using pilot symbols. [Figure 20] Block diagram showing an example of a wireless receiver in the second embodiment. [Figure 21] This is a block diagram showing an example of a wireless receiver in the third embodiment. [Figure 22] Block diagram showing an example of a wireless receiver in the fourth embodiment. [Figure 23] This is a block diagram showing an example of a pilot signal extraction unit. [Figure 24] This diagram illustrates an example of the processing operation of the pilot signal extraction unit. [Figure 25] Block diagram showing an example of a wireless receiver in the fifth embodiment. [Figure 26] This figure illustrates an example of the averaging process in the phase noise compensation signal generation unit. [Figure 27] This is a block diagram showing an example of a wireless receiver in the eighth embodiment. [Figure 28] This is a block diagram showing an example of a wireless receiver using OFDM in the ninth embodiment. [Figure 29] Block diagram showing an example of a wireless receiver in the tenth embodiment. [Figure 30] This is a block diagram showing an example of a wireless receiver in the 11th embodiment. [Figure 31] This block diagram shows an example of a wireless receiver in the 12th embodiment. [Figure 32] This figure shows an example of the hardware configuration of a wireless transmitter. [Figure 33]This figure shows an example of the hardware configuration of a wireless receiver. [Modes for carrying out the invention]

[0034] Embodiments will be described below with reference to the drawings. In the drawings of this disclosure, connections between blocks are indicated by arrows, but these are for convenience in explanation and the connections between blocks are not necessarily in the order of the arrows. In addition, the following descriptions and drawings have been omitted and simplified as appropriate for clarity of explanation. Also, in the following drawings, the same elements are denoted by the same reference numerals, and redundant explanations have been omitted as necessary. Furthermore, in this disclosure, unless otherwise specified, "at least one of A or B (A / B)" may mean any one of A or B, or both A and B. Similarly, when "at least one" is used for three or more elements, it may mean any one of these elements, or any multiple elements (including all elements).

[0035] <First Embodiment> <Example of a wireless transmitter configuration> Figure 5 is a block diagram showing an example of a wireless transmitter in the first embodiment. In Figure 5, the wireless transmitter 10 includes an information symbol output unit 11, a pilot signal output unit 12, an OFDM signal formation unit 13, and a transmitting wireless unit 14. The wireless transmitter 10 may have the same number of sets of the information symbol output unit 11, pilot signal output unit 12, OFDM signal formation unit 13, and transmitting wireless unit 14 as the number of antennas in the wireless transmitter 10. In each set, a wireless signal of the OFDM signal transmitted from the corresponding antenna may be formed.

[0036] The information symbol output unit 11 forms an information symbol and outputs it to the OFDM signal formation unit 13. For example, the information symbol output unit 11 channel-codes the information bit sequence using, for example, a low-density parity-check (LDPC) code. The information symbol output unit 11 then forms an information symbol by bit-interleaving the channel-coded bits to obtain encoded bits and bit-mapping them to the signal space arrangement (constellation) of the modulation scheme assigned according to the reception quality.

[0037] The pilot signal output unit 12 forms a pilot signal and outputs it to the OFDM signal formation unit 13.

[0038] The OFDM signal formation unit 13 forms an OFDM signal that includes information symbols and pilot signals. For example, the OFDM signal formation unit 13 forms an OFDM signal by arranging pilot signals according to a "pilot arrangement pattern". A "pilot arrangement pattern" is an arrangement pattern of resource elements in which pilot signals are arranged in an OFDM signal. Resource elements are defined (specified) by a combination of subcarriers and OFDM symbols. In this disclosure, multicarrier OFDM and single-carrier DFT-spread OFDM may be collectively referred to simply as "OFDM". Also, in the explanation of single-carrier DFT-spread OFDM, for convenience, the subcarriers of OFDM may be referred to as "tones (frequency components)". Also, in the explanation of single-carrier DFT-spread OFDM, for convenience, OFDM symbols may be referred to as "single-carrier symbol blocks".

[0039] (In the case of multi-carrier OFDM) Figure 6 is a block diagram showing an example of an OFDM signal formation unit. In Figure 6, the OFDM signal formation unit 13 includes series-parallel conversion units 13A1 and 13A2, a subcarrier mapping unit 13B, an IFFT unit 13C, and a CP insertion unit 13D.

[0040] The series-to-parallel conversion unit 13A1 converts multiple information symbols in series into parallel information symbols using series-to-parallel conversion (S / P conversion).

[0041] The series-parallel conversion unit 13A2 performs S / P conversion on multiple pilot symbols in series into pilot symbols in parallel.

[0042] The subcarrier mapping unit 13B maps the parallel information symbols received from the serial-parallel conversion unit 13A1 and the parallel pilot symbols received from the serial-parallel conversion unit 13A2 to the subcarriers according to the resource allocation. FFT A frequency-domain OFDM signal (i.e., an OFDM symbol) of a certain size is formed. As described above, the subcarrier mapping unit 13B maps the pilot symbol to the subcarrier (resource element) according to the "pilot placement pattern".

[0043] The IFFT section 13C processes the frequency domain OFDM signal formed in the subcarrier mapping section 13B. FFT The signal is converted to a time-domain OFDM signal using the inverse fast Fourier transform (IFFT) of the size.

[0044] The CP insertion unit 13D adds the signal at the end of the OFDM symbol as a Cyclic Prefix (CP) to the beginning of the OFDM symbol. This forms the OFDM signal.

[0045] (Case of single-carrier DFT-spread OFDM) Figure 7 is a block diagram showing another example of an OFDM signal formation unit. In Figure 7, the OFDM signal formation unit 13 includes DFT units 13E1 and 13E2, a subcarrier mapping unit 13F, an IDFT unit 13G, and a CP insertion unit 13H.

[0046] The DFT section 13E1 processes multiple information symbols in series, N DFT Each symbol is blocked. The DFT section 13E1 processes this single-carrier symbol block into NDFT The signal is converted to a frequency domain signal using a DFT (Digital Scale Format).

[0047] The DFT section 13E2 has multiple pilot symbols in series N DFT Each symbol is blocked. The DFT section 13E1 processes this single-carrier symbol block into N DFT The signal is converted to a frequency domain signal using a DFT (Digital Scale Format).

[0048] The subcarrier mapping unit 13F maps the frequency domain signal (information symbol) received from the DFT unit 13E1 and the frequency domain signal (pilot symbol) received from the DFT unit 13E2 to tones according to resource allocation. FFT A frequency-domain OFDM signal (i.e., a single-carrier symbol block) of a certain size is formed. As described above, the subcarrier mapping unit 13B maps pilot symbols to tones (resource elements) according to the "pilot placement pattern".

[0049] The IDFT section 13G receives the frequency domain OFDM signal formed in the subcarrier mapping section 13F. FFT The signal is converted to a time-domain OFDM signal using the inverse fast Fourier transform (IFFT) of the size.

[0050] The CP insertion unit 13H adds the signal at the end of the OFDM symbol as a Cyclic Prefix (CP) to the beginning of the OFDM symbol. This forms the OFDM signal.

[0051] Returning to the explanation of Figure 5, the transmitting radio unit 14 applies transmitting radio processing (analog-to-digital conversion, upconversion, etc.) to the OFDM signal formed in the OFDM signal formation unit 13 to form a radio signal. This radio signal is transmitted from an antenna (not shown) connected to the transmitting radio unit 14.

[0052] <Example of pilot placement pattern 1> Figure 8 is a diagram illustrating example 1 of the pilot placement pattern. The OFDM signal forming unit 13 may place the pilot signal according to the pilot placement pattern shown in Figure 8. In pilot placement pattern 1 shown in Figure 8, N is among the total subcarriers (tones) of the OFDM signal. Pilot_Freq Multiple pilot-placed subcarriers at the subcarrier (tone) intervals are equipped with pilot signals for phase noise estimation. Pilot_Freq In Figure 8,

number

[0053] Furthermore, in the pilot arrangement pattern 1 shown in Figure 8, N Pilot_Freq Among the multiple pilot-placed subcarriers (tones) with subcarrier spacing, N Pilot_SPA In the multiple first pilot-placed subcarriers (tones) at subcarrier (tone) intervals, a pilot signal for phase noise estimation is placed in all resource elements (in Figure 8, OFDM symbols (single carrier symbol blocks) #1 to #13) (except for the leading OFDM symbol (leading single carrier symbol block)). That is, N Pilot_SPA In multiple first pilot-placed subcarriers (tones) with subcarrier (tone) intervals, pilot signals for phase noise estimation are placed on multiple time-continuous resource elements.

[0054] Furthermore, in the pilot arrangement pattern 1 shown in Figure 8, N Pilot_Freq Among the multiple pilot-placed subcarriers (tones) with subcarrier (tone) intervals, the multiple second-placed subcarriers (tones) excluding the multiple first-placed subcarriers (tones) have N in the time domain. Pilot_Time A pilot signal for phase noise estimation is placed at the period of each resource element. That is, in the second pilot-placed subcarrier (tone), the pilot signals are placed discretely. Pilot_Timen is an integer value greater than or equal to 2, and is mainly changed depending on the modulation scheme (i.e., the signal space arrangement that maps the channel coding bits). For example, when the modulation multi-level number is small (i.e., when the number of signal points in the signal space arrangement is small), the phase margin between adjacent signal points is large, so N Pilot_Time By setting a large value, the insertion loss of the pilot signal is reduced. On the other hand, when the modulation level is large (i.e., when there are many signal points), the phase margin between adjacent signal points is small, so N Pilot_Time By reducing the value of N, it becomes possible to estimate the phase noise using a pilot signal at short time intervals. Pilot_SPA In modes where pilot signals are not multiplexed onto subcarriers (tones) other than the multiple first pilot-placed subcarriers (tones) in the subcarrier (tone) interval (i.e., N Pilot_Time There may also be a mode where (=∞).

[0055] In DFT-spread OFDM, FDM multiplexing of a pilot signal with a single-carrier information symbol signal results in a multi-carrier signal, leading to an increase in PAPR. However, in pilot signal multiplexing with pilot placement pattern 1, the tone interval of the pilot signal FDM-multiplexed in each single-carrier symbol block is constant. Therefore, although PAPR increases compared to a perfect single-carrier signal, the increase in PAPR can be kept low because it results in interleaved FDMA with different signal bandwidths.

[0056] As shown in Figure 8, the pilot signal multiplexed on the first OFDM symbol (or the first single-carrier symbol block) is a demodulation pilot signal for estimating the channel response of synchronous detection. In 5G and Beyond 5G systems, the subcarrier spacing is scalably widened from 15 kHz in LTE to achieve low transmission delay, and the slot length is shortened. Therefore, the amplitude and phase fluctuations caused by multipath fading in the propagation path within the slot section are very small. For this reason, the channel response of each subcarrier (tone) position estimated at the beginning of the slot can be used for all OFDM symbols (single-carrier symbol blocks) within the slot. In the embodiment shown in Figure 8, the demodulation pilot signal is consecutively multiplexed on all subcarrier (tone) positions of the OFDM symbol (or single-carrier symbol block).

[0057] According to the pilot placement pattern 1 described above, N Pilot_Freq Among the multiple pilot-placed subcarriers (tones) with a subcarrier (tone) interval, N Pilot_SPA In the multiple first pilot-placed subcarriers (tones) at subcarrier (tone) intervals, a pilot signal for phase noise estimation is placed in all resource elements (except the leading OFDM symbol (single-carrier symbol block)). That is, N Pilot_SPA In multiple first pilot-placed subcarriers (tones) with subcarrier (tone) intervals, pilot signals for phase noise estimation are placed on multiple time-continuous resource elements.

[0058] Furthermore, according to pilot placement pattern 1, information symbols and pilot symbols are mapped to different resource elements. As a result, unlike the FDSPT multiplexing method, pilot symbols are orthogonal to information symbols. Therefore, it is possible to estimate phase noise with high accuracy using pilot symbols that are not affected by interference from information symbols.

[0059] Furthermore, according to pilot placement pattern 1, N Pilot_FreqAmong the multiple pilot-placed subcarriers (tones) with subcarrier (tone) intervals, the multiple second-placed subcarriers (tones) excluding the multiple first-placed subcarriers (tones) have N in the time domain. Pilot_Time A pilot signal for phase noise estimation is placed at the period of each resource element. This reduces the insertion loss of the pilot signal compared to FET multiplexing.

[0060] Furthermore, according to pilot placement pattern 1, in the single-carrier symbol block of DFT-spread OFDM, the pilot signal is multiplexed between information symbols at a constant frequency interval in the frequency domain. Therefore, although the peak-to-average power ratio (PAPR) increases compared to a complete single-carrier, similar to interleaved FDMA and FET, the increase in PAPR can be kept lower compared to OFDM.

[0061] <Example of pilot placement pattern 2> Figure 9 illustrates example 2 of the pilot placement pattern. The pilot placement pattern 2 shown in Figure 9 differs from pilot placement pattern 1 in Figure 8 in the placement of the demodulation pilot signal. Specifically, as shown in Figure 9, the demodulation pilot signal is multiplexed by thinning out the subcarrier (tone) positions in the frequency domain. When the demodulation pilot signal is multiplexed in this way, control information can be multiplexed, for example, in resource elements (REs) that do not have the demodulation pilot signal multiplexed. In single-carrier DFT-spread OFDM, when the pilot signal and control information symbols are interleaved FDMA multiplexed in a comb-like manner in the frequency domain, there is no increase in PAPR.

[0062] <Example 3 of pilot placement patterns> Figure 10 is a diagram illustrating example 3 of pilot placement patterns. Pilot placement pattern 3 shown in Figure 10 is an example of a pilot placement pattern in the case of multi-antenna transmission. That is, the wireless transmitter 10 includes a first antenna and a second antenna N TXAnt This assumes that there are (2 or more integers) antennas. In Figure 10, in particular, N TXAnt The case where =2 is shown.

[0063] According to pilot configuration pattern 3 shown in Figure 10, in the OFDM signal transmitted by the first antenna, the pilot signal for phase noise estimation is placed using the same pattern as pilot configuration pattern 1. Furthermore, according to pilot configuration pattern 3, in the OFDM signal transmitted by the second antenna, the pilot signal for phase noise estimation is placed using the same pattern as pilot configuration pattern 1 on multiple pilot configuration subcarriers (tones) obtained by shifting the multiple pilot configuration subcarriers (tones) of pilot configuration pattern 1 by one subcarrier (tone) in the frequency direction. For the third antenna and beyond, the pilot signal for phase noise estimation should be placed using the same pattern as pilot configuration pattern 1 on multiple pilot configuration subcarriers (tones) obtained by sequentially shifting the multiple pilot configuration subcarriers (tones) of the previous antenna by one subcarrier (tone) in the frequency direction. In other words, according to pilot configuration pattern 3 shown in Figure 10, the pilot signals of different transmitting antennas are orthogonally multiplexed using FDM. TxAnt The pilot signals transmitted from (2 or more integers) transmitting antennas are orthogonal. Therefore, if a pilot signal transmitted from one transmitting antenna is multiplexed to a resource element, the OFDM signals transmitted from other transmitting antennas to that resource element will be muted.

[0064] Furthermore, regarding the demodulation pilot signal multiplexed on the leading OFDM symbol (or the leading single-carrier symbol block), the demodulation pilot signal is N in the frequency domain. TXAntTwo approaches are possible: FDM multiplexing for each subcarrier, and CDM multiplexing of the demodulation pilot signal between antennas. When FDM multiplexing of the demodulation pilot signal, the channel response at subcarrier locations where the demodulation pilot signal is not multiplexed can be estimated by interpolating the estimated channel response at subcarrier locations where the pilot signal is multiplexed.

[0065] <Example 4 of pilot placement patterns> Figure 11 is a diagram illustrating example 4 of the pilot placement pattern. Unlike pilot placement pattern 3 in Figure 10, pilot placement pattern 4 in Figure 11 has N Pilot_Freq Regarding resource elements on multiple second pilot-placed subcarriers, excluding multiple first pilot-placed subcarriers, among multiple pilot-placed subcarriers with subcarrier spacing, the resource elements of the OFDM signal transmitted by the first antenna and the resource elements of the OFDM signal transmitted by the second antenna do not overlap in time. Specifically, N Pilot_Freq Regarding resource elements on multiple second-pilot-configured subcarriers, excluding multiple first-pilot-configured subcarriers, among the multiple pilot-configured subcarriers with subcarrier spacing, the resource elements of the OFDM signal transmitted by the first antenna are shifted by one resource element in the time domain from the resource elements of the OFDM signal transmitted by the second antenna. The same rule can be used to configure pilot signals even when there are three or more transmitting antennas. By configuring pilot signals according to this rule, N TxAnt When considering the entire OFDM signal transmitted from (2 or more integers) transmitting antennas, the pilot signals are arranged (multiplexed) continuously in the time domain. In the above explanation, the term "subcarrier" may be replaced with "tone."

[0066] <Example of pilot placement pattern 5> Figure 12 is a diagram illustrating example 5 of pilot placement patterns. The pilot placement pattern 5 shown in Figure 12 is an example of a pilot placement pattern in the case of multi-antenna transmission. That is, the wireless transmitter 10 includes a first antenna and a second antenna N TXAnt This assumes that there are (2 or more integers) antennas. In Figure 12, in particular, N TXAnt The case where =2 is shown. In pilot arrangement pattern 5 shown in Figure 12, each set is N of all subcarriers in the OFDM signal. TxAnt It contains a set of consecutive subcarriers, and the interval between each set of two adjacent subcarriers is N. Pilot_Freq Pilot signals for phase noise estimation are placed on multiple pilot-placed subcarrier sets, which are subcarriers.

[0067] Furthermore, in the pilot placement pattern 5 shown in Figure 12, among the above multiple pilot placement subcarrier sets, N Pilot_SPA In a subcarrier set with multiple first pilot placements at subcarrier intervals, pilot signals for phase noise estimation are placed on all resource elements (OFDM symbols (single carrier symbol blocks) #1 to #13 in Figure 12).

[0068] Furthermore, in the pilot configuration pattern 5 shown in Figure 12, among the multiple pilot configuration subcarrier sets described above, the multiple second pilot configuration subcarrier sets, excluding the multiple first pilot configuration subcarrier sets described above, have N in the time domain. Pilot_Time A pilot signal for phase noise estimation is placed at each resource element period. That is, in the second pilot-placed subcarrier set, the pilot signals are placed discretely. Pilot_Time This is an integer value greater than or equal to 2, and is primarily modified depending on the modulation scheme (i.e., the signal space arrangement that maps the channel coding bits).

[0069] The pilot signals transmitted from each transmitting antenna are spread out, and then N in the frequency domain. TxAntThe signals are orthogonally multiplexed by CDM across a series of consecutive subcarriers. That is, the pilot signal output unit 12 may spread the pilot signals transmitted from each transmitting antenna and output the spread pilot signals to the OFDM signal formation unit 13. Two methods can be used as the method for CDM orthogonal multiplexing.

[0070] In the first method, the pilot signals of each transmitting antenna are transmitted over a sequence length of N. TxAnt Spread across different Walsh-Hadamard codes, in the frequency domain N TxAnt This is a method for mapping individual subcarriers to a pilot deployment subcarrier set.

[0071] The second method is to use a cyclic shift of a constant-amplitude Zadoff-Chu sequence shown in equation (2) below for multiplexing.

number

[0072] Furthermore, regarding the demodulation pilot signal multiplexed on the leading OFDM symbol (or the leading single-carrier symbol block), the demodulation pilot signal is N in the frequency domain. TXAnt When FDM multiplexing is performed for each subcarrier, and when the demodulation pilot signal is N TxAntOne possible scenario is CDM multiplexing between antennas using a series of consecutive subcarriers. In the above explanation, "subcarrier" can be interpreted as "tone."

[0073] Figure 13 illustrates an example of CDM pilot signal multiplexing in single-carrier DFT-spread OFDM. Figure 13 shows a method corresponding to the second method described above. Here, different cyclic shifts are applied to Zadoff-Chu sequences with the same root index to generate orthogonal pilot signals for phase noise estimation. Furthermore, assuming a Zadoff-Chu sequence, the sequence length is N. ZC This is represented as follows. The pilot signal output unit 12 may have a spread sequence generation unit and a cyclic shift generation unit as shown in Figure 13.

[0074] The spreading code generation unit generates spreading codes such as the Zadoff-Chu sequence. The cyclic shift unit receives the spreading codes from the spreading code generation unit and uses these spreading codes to generate a signal for a number of transmitting antennas N. TxAnt Generate the same number of cyclic shift sequences as the number of cyclic shifts, which is equivalent to the number of cyclic shifts. CS Therefore, the cyclic shift sequence length (i.e., the cyclic shift amount) of the cyclic shift index is N ΔCS =N ZC / N CS This is the result. N TxAnt As the value of increases, the number of cyclic shifts increases, and the amount of shift N between different cyclic shift sequences increases. ΔCS Therefore, the sequence length becomes shorter. Sequence length N ΔCS The time must be longer than the maximum delay time of the multipath. The multipath delay time is the amount of the cyclic shift N. ΔCS If the shift length becomes longer than this, intersymbol interference will occur between codes that use different cyclic shifts.

[0075] <Example 6 of pilot placement patterns> Figure 14 is a diagram illustrating example 6 of the pilot placement pattern. In pilot placement pattern 6 shown in Figure 14, N is among the total subcarriers of the OFDM signal. Pilot_FreqPilot signals are arranged on a plurality of pilot placement sub-carriers with a sub-carrier interval.

[0076] Also, in the pilot placement pattern 6 shown in FIG. 14, in each pilot placement sub-carrier, N Pilot_Time pilot signals are arranged at resource element periods in the time domain.

[0077] Also, in the pilot placement pattern 6 shown in FIG. 14, for each two adjacent pilot placement sub-carriers, pilot signals are arranged so that the resource elements where the pilot signals are arranged do not overlap in the time domain. In particular, in the pilot placement pattern 6 shown in FIG. 14, for the first pilot placement sub-carrier and the second pilot placement sub-carrier arranged side by side in the frequency direction among each two adjacent pilot placement sub-carriers, the position in the time domain of the first resource element where a pilot is arranged in the second pilot placement sub-carrier is adjacent in the time domain to the position in the time domain of the first resource element where a pilot is arranged in the first pilot placement sub-carrier, and pilot signals are arranged. The sub-carriers described above may be read as tones.

[0078] <Example 7 of Pilot Placement Pattern> FIGS. 15 and 16 are diagrams for explaining Example 7 of the pilot placement pattern. The pilot placement pattern 7 shown in FIGS. 15 and 16 is an example of a pilot placement pattern in the case of multi-antenna transmission. That is, it is assumed that the wireless transmission device 10 has N TXAnt (an integer of 2 or more) antennas. In FIGS. 15 and 16, in particular, the case of N TXAnt = 2 is shown.

[0079] According to the pilot arrangement pattern 7 shown in FIGS. 15 and 16, in the OFDM signal transmitted by the first antenna, pilot signals for phase noise estimation are arranged according to the same pattern as the above-described pilot arrangement pattern 6. Also, according to the pilot arrangement pattern 7, in the OFDM signal transmitted by the second antenna, pilot signals for phase noise estimation are arranged according to the same pattern as the pilot arrangement pattern 6 in a plurality of pilot arrangement sub-carriers obtained by shifting, in the frequency direction, the plurality of pilot arrangement sub-carriers of the pilot arrangement pattern 6 by only one sub-carrier. Regarding the third antenna and subsequent antennas, pilot signals for phase noise estimation may be arranged according to the same pattern as the pilot arrangement pattern 6 in a plurality of pilot arrangement sub-carriers obtained by sequentially shifting, in the frequency direction, the plurality of pilot arrangement sub-carriers of the immediately preceding antenna by only one sub-carrier. That is, according to the pilot arrangement pattern 7 shown in FIGS. 15 and 16, the pilot signals of different transmitting antennas are orthogonally multiplexed by FDM. N TxAnt (An integer of 2 or more) pilot signals transmitted from transmitting antennas are orthogonal. For this reason, as shown in FIG. TxAnt , when a pilot signal transmitted from one transmitting antenna is multiplexed in a certain resource element, muting is performed in that resource element of the OFDM signal transmitted from other transmitting antennas. The sub-carriers in the above description may be read as tones.

[0080] <Example 8 of Pilot Arrangement Pattern> FIG. 17 is a diagram for explaining Example 8 of a pilot arrangement pattern. The pilot arrangement pattern 6 shown in FIG. 17 is an example of a pilot arrangement pattern in the case of multi-antenna transmission. That is, it is premised that the wireless transmission device 10 has N TXAnt (An integer of 2 or more) antennas. In FIG. 17, particularly the case of N TXAnt =2 is shown. In the pilot arrangement pattern 8 shown in FIG. 17, among all the sub-carriers of the OFDM signal, each set includes N TxAnt consecutive sub-carriers and the interval between each two adjacent sets is N Pilot_FreqPilot signals for phase noise estimation are placed on multiple pilot-placed subcarrier sets, which are subcarriers.

[0081] Furthermore, according to the pilot configuration pattern 8 shown in Figure 17, in each pilot configuration subcarrier set, N in the time domain Pilot_Time Pilot signals are placed at the interval of each resource element.

[0082] Furthermore, according to pilot placement pattern 8 shown in Figure 17, in each of two adjacent pilot placement subcarrier sets, the pilot signals are arranged such that the resource elements on which the pilot signals are placed do not overlap in the time domain. In particular, in pilot placement pattern 8 shown in Figure 17, in each pilot placement subcarrier set, the pilot signals are arranged such that the resource elements on which the pilot signals are placed overlap in time. And, with respect to the first and second pilot placement subcarrier sets, which are aligned in the frequency direction among each of two adjacent pilot placement subcarrier sets, the pilot signals are arranged such that the time domain position of the first resource element on which the pilot is placed in the second pilot placement subcarrier set is adjacent in the time domain to the time domain position of the first resource element on which the pilot is placed in the first pilot placement subcarrier set.

[0083] The pilot signals transmitted from each transmitting antenna are spread out, and then N in the frequency domain. TxAnt The signals are orthogonally multiplexed by CDM across a series of consecutive subcarriers. That is, the pilot signal output unit 12 may spread the pilot signals transmitted from each transmitting antenna and output the spread pilot signals to the OFDM signal formation unit 13. As a method of CDM orthogonal multiplexing, the two methods described above in the explanation of pilot arrangement pattern example 5 can be used.

[0084] <Second Embodiment> The second embodiment relates to an example of a wireless receiving device.

[0085] <Example of basic configuration for a wireless receiver> Figure 18 shows an example of the basic configuration of a wireless receiver in the second embodiment. In Figure 18, the wireless receiver 20 has a receiving wireless unit 21 and a signal processing unit 22. The signal processing unit 22 has a phase noise estimation and compensation unit 23 using pilot symbols, a fast Fourier transform (discrete Fourier transform) unit 24, an equalization processing unit 25, a phase noise estimation and compensation unit 26 using decision feedback symbols, and an error correction decoding unit 27.

[0086] The receiving wireless unit 21 processes the received wireless signal received via the antenna (not shown) of the wireless receiving device 20 by performing wireless reception processing (down-conversion, analog-to-digital conversion, etc.) and outputs the resulting received signal (time-domain OFDM signal) to the signal processing unit 22.

[0087] The phase noise estimation and compensation unit 23 receives a received signal (time-domain OFDM signal) from the receiving radio unit 21, estimates the phase noise of the received signal using a pilot signal, and compensates for it.

[0088] The Fast Fourier Transform (Discrete Fourier Transform) unit 24 converts a time-domain signal with phase noise compensation into a frequency-domain signal.

[0089] The equalization processing unit 25 estimates the channel response of each subcarrier signal of the frequency domain signal and performs equalization processing to compensate for fluctuations in the phase and amplitude of the propagation path. In the case of OFDM, the equalization processing unit 25 can be implemented with an equalizer with one tap per subcarrier. That is, the equalization processing unit 25 can be implemented with a synchronous detection and demodulation unit. Furthermore, the equalization processing unit 25 demapping the equalized symbols to reconstruct the hard decision bits before error correction decoding. After phase noise compensation for iterative processing, the equalization processing unit 25 generates confidence information for each bit from the equalized symbols.

[0090] The phase noise estimation and compensation unit 26 remaps the hard decision bits to generate symbols, multiplies them by the estimated value of the channel response, and then converts them into a time-domain signal by IFFT. Further, the phase noise estimation and compensation unit 26 estimates and compensates for the residual phase noise from the received signal and the above time-domain signal. Therefore, the phase noise estimation and compensation unit 26 that uses the decision feedback symbols includes an IFFT that converts the frequency-domain signal into a time-domain signal.

[0091] The error correction decoding unit 27 receives the reliability information of each bit from the equalized symbols after the phase noise compensation of the repetition process, and performs error correction decoding.

[0092] FIG. 19 is a block diagram showing an example of a phase noise estimation and compensation unit using pilot symbols. In FIG. 19, the phase noise estimation and compensation unit 23 includes an orthogonal basis function generation unit (output unit) 23A, a weight coefficient generation unit (weight generation unit) 23B, a phase noise calculation unit 23C, and a multiplication unit 23D.

[0093] Let the number of samples in the OFDM symbol section in the time domain be N FFT be represented by. And the phase noise at each sample position is represented by the following formula. In this formula, T represents transpose.

Equation

[0094] The case of using the discrete cosine transform (DCT: Discrete Cosine Transform) for the basis function will be described. D basis functions of length N FFT are

Equation

Equation

[0095] V(D) =[v0,v1,···,v D-1 For ], the weight coefficient γ(=[γ0,γ1,···γ) satisfies the following equation (4). D-1 ] T ) defines.

number

[0096] The weight coefficient γ can be determined by the following equation (5) using the least mean squared error (LS) criterion.

number

[0097] For example, the weight coefficient γ is estimated using pilot symbols that are continuously multiplexed in the time domain. For example, N as explained in Example 1 of the pilot placement patterns above. Pilot_SPA Pilot signals placed on multiple first pilot-placed subcarriers at subcarrier intervals may be used to estimate the weight coefficient γ. Furthermore, in example 6 of the pilot placement patterns described above, although the subcarriers on which pilot signals for phase noise estimation are placed differ depending on the OFDM symbol, pilot signals for phase noise estimation are placed on all OFDM symbols. Therefore, when viewed as a whole across multiple subcarriers, the pilot signals are continuously multiplexed in the time domain. For this reason, in example 6 of the pilot placement patterns described above, these pilot signals can be used as the weight coefficient γ.

[0098] The matrix W is defined by the following equation (6).

number

[0099] The number of rows in matrix W corresponds to the number of subcarriers, which is equivalent to the number of FFT samples, and the number of columns corresponds to the number of basis functions. Furthermore, Λ represents a diagonal matrix whose diagonal elements are the channel coefficients corresponding to each subcarrier position.

number

[0100] According to equation (5), the estimated value of the weight coefficient γ is,

number

number

[0101] In equation (7), W P This is a transmit pilot symbol s multiplexed from W using FDM. P This represents a matrix in which rows corresponding to the subcarrier positions of the multiplexed subcarrier are extracted. Also, the superscript symbol (·) is used. H This represents the complex conjugate transpose. Also, (·) -1 This represents the inverse matrix.

[0102] The weight coefficient generation unit 23B uses the pilot signal extracted by the pilot signal extraction unit 41 to calculate an estimated value of the weight coefficient γ, for example, according to equation (7).

[0103] The phase noise calculation unit 23C uses the estimated weight coefficient γ calculated by the weight coefficient generation unit 23B and the basis function generated by the orthogonal basis function generation unit 23A to calculate the estimated phase noise e according to equation (4). -jΦ Calculate the estimated value of phase noise e. -jΦ This is calculated on a sample-by-sample basis and output to the multiplication unit 23D.

[0104] The multiplication unit 23D calculates the estimated phase noise e -jΦ Phase noise is compensated for by multiplying the received signal by a factor of 1 / 2 on a sample-by-sample basis.

[0105] Furthermore, the eigenvectors of the covariance matrix of the phase noise can also be used as basis functions. The complex representation of the phase noise is given by ψ(t)=e jΦ(t) Defined as follows: The covariance r of ψ(t) ψ (τ) can be calculated using the following formula.

number

[0106] N is the number of OFDM symbol samples (or symbols within a single carrier symbol block). FFT The discrete ψ(t) in this case is expressed by the following equation.

number

[0107] Here, T S If we let be the sampling interval, the following equation holds:

number

[0108] The covariance matrix of the phase noise is defined by the following equation (8).

number

[0109] The covariance matrix is ​​estimated using a pilot signal whose modulation phase and modulation amplitude are known. The covariance matrix in equation (8) is then decomposed into eigenvalues ​​(singular value decomposition) as shown in equation (9).

number

[0110] In equation (9),

number

number

[0111] As described above, since the weight coefficient γ in equation (7) is calculated on a sample-by-sample basis for OFDM symbols and the phase noise is estimated from equation (4), it can be directly applied to single-carrier signals by performing the same process on a symbol-by-symbol basis for single-carrier symbol blocks.

[0112] <Example of a wireless receiver configuration> Figure 20 is a block diagram showing an example of a wireless receiver in the second embodiment. In Figure 20, the wireless receiver 20 includes an FFT unit 31, an equalizer (synchronous detector) 32, a demapping unit 33, and an error correction decoding unit 34. The wireless receiver 20 also includes a pilot signal extraction unit 41, a channel response estimation unit 42, a symbol estimate generation unit 43, a pilot symbol generation unit 44, a subcarrier mapping unit 45, a multiplication unit 46, an IFFT unit 47, a phase noise compensation signal generation unit 48, and a multiplication unit 49. The FFT unit 31 corresponds to the Fast Fourier Transform (Discrete Fourier Transform) unit 24 described above. The equalizer (synchronous detector) 32 and the demapping unit 33 correspond to the equalization processing unit 25 described above. The error correction decoding unit 34 corresponds to the error correction decoding unit 27 described above. Furthermore, the pilot signal extraction unit 41, the channel response estimation unit 42, the symbol estimate value generation unit 43, the pilot symbol generation unit 44, the subcarrier mapping unit 45, the multiplication unit 46, the IFFT unit 47, the phase noise compensation signal generation unit 48, and the multiplication unit 49 correspond to the above-mentioned phase noise estimation and compensation unit 26.

[0113] The FFT section 31 receives the received signal (time-domain OFDM signal (hereinafter sometimes referred to as the "first time-domain signal")) after the phase noise has been compensated by the phase noise estimation and compensation section 23 using pilot symbols, and converts the received first time-domain signal into a frequency-domain signal.

[0114] Here, consider a case where in the wireless transmission device 10, an information symbol corresponding to the number of stages of the FFT unit 31 is blocked to generate an OFDM signal. The information symbol s k is subjected to IFFT processing to generate an OFDM signal d n (0 ≤ k, n ≤ N FFT -1). d n is expressed by the following equation (10). [Number]

[0115] At this time, the received signal r n at the sample timing n within the FFT block is expressed by the following equation (11). [Number]

[0116] In equation (11), h l is the channel impulse response in path l (0 ≤ l ≤ L - 1). Also, W n represents an AWGN (Additive White Gaussian Noise) component. Also, Φ n represents residual phase noise. In equation (11), the term of the information symbol that has undergone multipath fading among r n is represented by y n . That is, y n is expressed by the following equation. [Number]

[0117] The FFT unit 31 converts the received signal r n into a frequency domain signal R k by FFT. R k is expressed by the following equation (12). [Number] ​​In equation (12), H k ,p k ,W k These are, respectively, h n ,e jΦn ,w n This is the FFT transform. As shown in equations (11) and (12), the phase noise that gives random phase rotation in the time domain is represented in the frequency domain by the convolution of inter-subcarrier interference (ICI).

[0119] The pilot signal extraction unit 41 extracts the pilot signal included in the frequency domain signal obtained by the FFT unit 31.

[0120] In the case of OFDM, the equalizer 32 consists of a single-tap equalizer and performs synchronous detection and demodulation. The channel response at each subcarrier position is estimated using a demodulation pilot signal multiplexed at the OFDM symbol position at the beginning of the slot. Alternatively, in addition to the demodulation pilot signal multiplexed at the OFDM symbol position at the beginning of the first slot, a pilot signal multiplexed between information symbols may be used.

[0121] The demapping unit 33 demmaps each symbol in the symbol sequence after synchronous detection and reconstructs the transmitted bits. When used for phase noise estimation and compensation, the demapping unit 33 generates hard decision bits. For the error correction decoding unit 35, the demapping unit 33 generates a log-likelihood ratio (LLR), which is the logarithm of the ratio of the probabilities that each bit is "1" or "0 (-1)". When a receiver uses antenna diversity reception, the output signals from multiple antennas, the receiving radio unit, and the equalizer (synchronous detector) are added together independently as in-phase and quadrature components.

[0122] The error correction decoding unit 34 receives the log-likelihood ratio (LLR), which is the logarithm of the ratio of the probabilities that each bit generated by the demapping unit 33 is "1" or "0 (-1)," and performs error correction decoding processing, outputting the resulting decoded bits.

[0123] The symbol estimate generation unit 43 generates an estimate of the information symbol by mapping the hard decision bit of the transmitted bit.

number

[0124] The subcarrier mapping unit 45 maps the estimated values ​​of information symbols obtained by the symbol estimation value generation unit 43 and the pilot symbols generated by the pilot symbol generation unit 26 to the subcarriers. The pilot symbol sequence is known to the base station equipment, and is either known to the user terminal in advance or notified in advance by the base station via the control channel.

[0125] The channel response estimation unit 42 estimates the channel response in subcarrier k.

number

[0126] The multiplication unit 46 multiplies the estimated value of the information symbol generated by the symbol estimate generation unit 43 and mapped to subcarrier k with the estimated value of the channel response in subcarrier k, as shown in the following equation. The multiplication result obtained by the multiplication unit 46 can be called the "received signal replica" in the frequency domain.

number

[0127] The IFFT section 47 performs an IFFT transformation on the frequency domain received signal replica to obtain a time domain received signal replica (which may be referred to below as the "second time domain signal").

number

[0128] The phase noise compensation signal generation unit 48 estimates phase noise using the received signal and the received signal replica in the time domain (the above-mentioned "second time domain signal"), and outputs a phase noise compensation signal having an inverse characteristic of the estimated phase noise.

[0129] For example, the signal after decision feedback at loop count i,

Equation

Equation

[0130] As shown in Equation (13),

Equation

Equation

Equation

Equation

Equation

Equation

[0131] The phase noise at each sample timing of Equation (13),

Equation

[0132] The phase noise at each sample timing after averaging is

number

[0133]

number

number

[0134] The multiplier 49 calculates the compensation value for phase noise at each of the above sample timings, which is the received signal including residual phase noise.

number

[0135] <Third Embodiment> The third embodiment relates to another example of a wireless receiver. Specifically, in the second embodiment, a symbol estimate is generated using hard decision bits obtained by demapping the signal before error correction decoding (before channel coding decoding), whereas in the third embodiment, a symbol count estimate is generated from bits after error correction decoding. The basic configuration of the wireless receiver in the third embodiment is the same as the basic configuration of the wireless receiver 20 in the second embodiment.

[0136] Figure 21 is a block diagram showing an example of a wireless receiver in the third embodiment. In Figure 21, the wireless receiver 50 includes an FFT unit 31, an equalizer (synchronous detector) 32, a demapping unit 33, and an error correction decoding unit 34. The wireless receiver 50 also includes a pilot signal extraction unit 41, a channel response estimation unit 42, a symbol estimate generation unit 51, a pilot symbol generation unit 44, a subcarrier mapping unit 52, a multiplication unit 46, an IFFT unit 47, a phase noise compensation signal generation unit 48, and a multiplication unit 49. The FFT unit 31 corresponds to the Fast Fourier Transform (Discrete Fourier Transform) unit 24 described above. The equalizer (synchronous detector) 32 and the demapping unit 33 correspond to the equalization processing unit 25 described above. The error correction decoding unit 34 corresponds to the error correction decoding unit 27 described above. Furthermore, the pilot signal extraction unit 41, the channel response estimation unit 42, the symbol estimate value generation unit 51, the subcarrier mapping unit 52, the pilot symbol generation unit 44, the multiplication unit 46, the IFFT unit 47, the phase noise compensation signal generation unit 48, and the multiplication unit 49 correspond to the above-mentioned phase noise estimation and compensation unit 26.

[0137] The demapping unit 33 in the wireless receiver 50 demmaps the information symbols contained in the time-domain signal after equalization processing and calculates the log-likelihood ratio (LLR), which is the logarithm of the ratio of the probabilities that each bit in each symbol is "1" or "0 (-1)".

[0138] The error correction decoding unit 34 in the wireless receiver 50 receives the confidence level information LLR for each bit in each symbol obtained by the demapping unit 33 and performs error correction decoding processing. The error correction decoding unit 34 then outputs the obtained decoded bits.

[0139] For example, the error correction decoding unit 34 deinterleaves the log-likelihood ratio of each bit, inputs the deinterleaved log-likelihood ratio, performs error correction decoding processing, and reconstructs the transmitted bits by making a hard decision on the log-likelihood ratio of each bit with improved reliability. Alternatively, it calculates the probability that each bit is "1" or "0 (-1)" from the LLR of each bit in the output of the error correction decoder. Assuming that each bit of each symbol (signal point) in the signal space arrangement is independent, the probability of the symbol is calculated from the probability that each bit is "1" or "0 (-1)". It is also possible to use a soft decision symbol generated in this way.

[0140] The symbol estimation generation unit 51 generates an estimated value of an information symbol by mapping the hard determination bit of the transmitted bit reconstructed by the error correction decoding unit 34.

[0141] The subcarrier mapping unit 52 maps the estimated values ​​of information symbols obtained by the symbol estimation value generation unit 51 and the pilot symbols generated by the pilot symbol generation unit 44 to the subcarriers.

[0142] The multiplication unit 46 multiplies the estimated value of the information symbol generated by the symbol estimate generation unit 51 and mapped to the subcarrier k by the estimated value of the channel response in the subcarrier k.

[0143] <Fourth Embodiment> The fourth embodiment relates to another example of a wireless receiver. In particular, the fourth embodiment relates to a configuration that corresponds to single-carrier DFT-spread OFDM. That is, the wireless receiver of the fourth embodiment has a configuration that corresponds to single-carrier DFT-spread OFDM, compared to the configuration of the wireless receiver 20 of the second embodiment that corresponds to multi-carrier OFDM. Note that the basic configuration of the wireless receiver of the fourth embodiment is the same as the basic configuration of the wireless receiver 20 of the second embodiment.

[0144] Figure 22 is a block diagram showing an example of a wireless receiver in the fourth embodiment. In Figure 22, the wireless receiver 60 includes an FFT unit 31, an equalizer 32, an IDFT unit 61, a demapping unit 62, and an error correction decoding unit 34. The wireless receiver 60 also includes a pilot signal extraction unit 41, a channel response estimation unit 42, a symbol estimate generation unit 43, a DFT unit 63, a pilot signal generation unit 64, a subcarrier mapping unit 45, a multiplication unit 46, an IFFT unit 47, a phase noise compensation signal generation unit 48, and a multiplication unit 49. The FFT unit 31 corresponds to the Fast Fourier Transform (Discrete Fourier Transform) unit 24 described above. The equalizer 32, IDFT unit 61, and demapping unit 62 correspond to the equalization processing unit 25 described above. The error correction decoding unit 34 corresponds to the error correction decoding unit 27 described above. The pilot signal extraction unit 41, the channel response estimation unit 42, the symbol estimate value generation unit 43, the DFT unit 63, the pilot signal generation unit 64, the subcarrier mapping unit 45, the multiplication unit 46, the IFFT unit 47, the phase noise compensation signal generation unit 48, and the multiplication unit 49 correspond to the above-mentioned phase noise estimation and compensation unit 26.

[0145] The equalizer 32 in the wireless receiver 60 performs waveform equalization on the frequency domain signal obtained by the FFT unit 31. Typically, equalization weight coefficients based on the Minimum Mean Square Error (MMSE) algorithm are used.

[0146] The IDFT section 61 converts the equalized frequency domain signal into a time domain signal.

[0147] The demapping unit 62 demmaps each symbol in the information symbol sequence and reconstructs the transmitted bits. When used for phase noise estimation and compensation, it generates hard decision bits. For the error correction decoding unit 34, it generates the log-likelihood ratio (LLR), which is the logarithm of the ratio of the probabilities of each bit being "1" or "0 (-1)". When the receiver uses antenna diversity reception, the output signals from multiple antennas, the receiving radio unit, and the equalizer (synchronous detector) are added independently using in-phase and quadrature components.

[0148] The symbol estimate generation unit 43 generates an estimate of the information symbol by mapping the hard decision bit of the transmitted bit.

[0149] The DFT unit 63 converts multiple information symbol estimates generated by the symbol estimate generation unit 43 into frequency domain signals.

[0150] The subcarrier mapping unit 45 maps the frequency domain signal obtained by the DFT unit 63 to tones. The subcarrier mapping unit 45 also multiplexes the pilot signal generated by the pilot signal generation unit 64 onto the tone positions between the tones of the information symbol.

[0151] The pilot signal extraction unit 41 extracts the pilot signal included in the frequency domain signal obtained by the FFT unit 31.

[0152] Figure 23 is a block diagram showing an example of the pilot signal extraction unit. Figure 24 is a diagram illustrating an example of the processing operation of the pilot signal extraction unit. Here, we will explain the extraction of the pilot signal sequence when the pilot signal is transmitted using an orthogonal sequence specific to the transmitting antenna due to the cyclic shift of the Zadoff-Chu sequence shown in Figure 13.

[0153] As shown in Figure 23, the pilot signal extraction unit 41 includes a subcarrier demapping unit 41A and a pilot extraction processing unit 41B.

[0154] The subcarrier demapping unit 41A extracts pilot signals that are multiplexed between information symbols in the frequency domain signal obtained by the FFT unit 31.

[0155] The pilot extraction processing unit 41B multiplies the pilot signal extracted by the subcarrier demapping unit 41A by the complex conjugate of the cyclic shift sequence of the frequency domain pilot signal, and obtains the N CS By adding these signals in phase (i.e., by despreading), a pilot signal specific to the transmitting antenna is generated (extracted).

[0156] Here, the time-domain shift, performed using the Discrete Fourier Transform (Fast Fourier Transform), corresponds to a phase rotation in the frequency domain. The number of cyclic shifts in the time domain is N. CS In contrast, in the frequency domain, 2π / N per tone. CS Only a phase shift occurs. Therefore, discretely mapped N CS Since the phase rotation amount between these tones is 2π, N CS The cross-correlation of signs between individual tones is zero.

[0157] The channel response estimation unit 42 estimates the channel response at tone k.

[0158] The multiplication unit 46 multiplies the estimated value of the information symbol generated by the symbol estimate generation unit 43 and mapped to tone k by the estimated value of the channel response in tone k.

[0159] <Fifth Embodiment> The fifth embodiment relates to another example of a wireless receiver. Specifically, in the fourth embodiment, a symbol estimate is generated using hard decision bits obtained by demapping the signal before error correction decoding (before channel coding decoding), whereas in the fifth embodiment, a symbol count estimate is generated from bits after error correction decoding. The basic configuration of the wireless receiver in the fifth embodiment is the same as the basic configuration of the wireless receiver 20 in the second embodiment.

[0160] Figure 25 is a block diagram showing an example of a wireless receiver in the fifth embodiment. In Figure 25, the wireless receiver 70 includes an FFT unit 31, an equalizer 32, an IDFT unit 61, a demapping unit 62, and an error correction decoding unit 34. The wireless receiver 70 also includes a pilot signal extraction unit 41, a channel response estimation unit 42, a symbol estimate generation unit 51, a DFT unit 63, a pilot signal generation unit 64, a subcarrier mapping unit 45, a multiplication unit 46, an IFFT unit 47, a phase noise compensation signal generation unit 48, and a multiplication unit 49. The FFT unit 31 corresponds to the Fast Fourier Transform (Discrete Fourier Transform) unit 24 described above. The equalizer 32, IDFT unit 61, and demapping unit 62 correspond to the equalization processing unit 25 described above. The error correction decoding unit 34 corresponds to the error correction decoding unit 27 described above. The pilot signal extraction unit 41, the channel response estimation unit 42, the symbol estimate value generation unit 51, the DFT unit 63, the pilot signal generation unit 64, the subcarrier mapping unit 45, the multiplication unit 46, the IFFT unit 47, the phase noise compensation signal generation unit 48, and the multiplication unit 49 correspond to the above-mentioned phase noise estimation and compensation unit 26.

[0161] The demapping unit 62 in the wireless receiver 70 demapping the information symbols contained in the time-domain signal after equalization processing generates a log-likelihood ratio (LLR), which is the logarithm of the ratio of the probabilities that each bit in each symbol is "1" or "0 (-1)".

[0162] The error correction decoding unit 34 in the wireless receiver 70 receives the log-likelihood ratio (LLR), which is the logarithm of the ratio of the probabilities that each bit in each symbol obtained by the demapping unit 62 is "1" or "0 (-1)," and performs error correction decoding. It then outputs the LLR with improved reliability after error correction decoding.

[0163] The symbol estimation unit 51 in the wireless receiver 70 generates an estimated value of an information symbol by mapping bits that have been hard-determined from the log-likelihood ratio (LLR) output from the error correction decoding unit 34.

[0164] The DFT unit 63 in the wireless receiver 70 converts multiple information symbol estimates generated by the symbol estimate generation unit 51 into frequency domain signals.

[0165] The subcarrier mapping unit 45 in the wireless receiver 70 maps the frequency domain signal obtained by the DFT unit 63 and the pilot signal generated by the pilot signal generation unit 64 to their respective tone positions.

[0166] The multiplication unit 46 multiplies the estimated value of the information symbol generated by the symbol estimate generation unit 51 and mapped to tone k by the estimated value of the channel response in tone k.

[0167] <Sixth Embodiment> The sixth embodiment relates to an example of averaging processing in the phase noise compensation signal generation unit 48 described in the second to fifth embodiments.

[0168] Figure 26 illustrates an example of averaging processing in the phase noise compensation signal generation unit. Since this averaging processing is performed on a sample-by-sample basis, it can be applied to both OFDM and DFT-spread OFDM.

[0169]

number

number

number

[0170] This is an estimate of the phase noise at the center position of a series of averaging windows.

number

number

[0171] For example, the phase noise compensation signal generation unit (second phase noise compensation processing unit) 48 generates an N of OFDM symbols corresponding to the FFT block. FFT N samples Blk N of each block divided into individual blocks FFT / N Blk The phase noise estimates for each sample are independently averaged using the common-mode component and the orthogonal component. Then, the phase noise compensation signal generation unit (second phase noise compensation processing unit) 48 processes the obtained N Blk The phase noise at each sample point is estimated by interpolating the average of the phase noise estimates for each block using linear interpolation, quadratic interpolation, or interpolation using a higher-order function.

[0172] <Seventh Embodiment> The seventh embodiment relates to another example of the averaging process in the phase noise compensation signal generation unit 48 described in the second to fifth embodiments. The following describes the case of OFDM, but the same procedure can be used to implement a single-carrier DFT-spread OFDM.

[0173] The relative phase shift of the phase noise at sampling timing n compared to the phase noise one sample prior is defined as follows:

number

[0174] The error signal e between the received signal and the symbol estimate considering channel response and phase noise. n This is expressed by the following equation (16).

number

[0175] The phase noise compensation processing unit (second phase noise compensation processing unit) 49 uses the LMS algorithm to perform the relative phase shift ψ as shown in equation (17) below. n Update.

number

[0176] <Eighth Embodiment> The eighth embodiment relates to another example of a wireless receiver. In particular, the wireless receiver of the eighth embodiment relates to phase noise estimation and phase noise compensation using a phase-locked loop (PLL) when an OFDM is used. The basic configuration of the wireless receiver of the eighth embodiment is the same as the basic configuration of the wireless receiver 20 of the second embodiment.

[0177] Figure 27 is a block diagram showing an example of a wireless receiver in the eighth embodiment. In Figure 27, the wireless receiver 80 includes an FFT unit 31, an equalizer (synchronous detector) 32, a demapping unit 33, and an error correction decoding unit 34. The wireless receiver 80 also includes a bit mapping unit 81, a phase detector (PD) 82, a loop filter 83, a phase noise compensation processing unit (second phase noise compensation processing unit) 84, and a switch 85. The FFT unit 31 corresponds to the Fast Fourier Transform (Discrete Fourier Transform) unit 24 described above. The equalizer (synchronous detector) 32 and the demapping unit 33 correspond to the equalization processing unit 25 described above. The error correction decoding unit 34 corresponds to the error correction decoding unit 27 described above. Furthermore, the bit mapping unit 81, the phase detector (PD) 82, the loop filter 83, and the phase noise compensation processing unit (second phase noise compensation processing unit) 84 correspond to the phase noise estimation and compensation unit 26 described above.

[0178] In the wireless receiver 80, when the demapping unit 33 is connected to the equalizer (synchronous detector) 32 by switch 85, it demmaps the information symbols contained in the time-domain signal after equalization processing and calculates the reliability information (i.e., the log-likelihood ratio (LLR)) for each bit in each symbol. Also, when the demapping unit 33 is connected to the phase noise compensation processing unit 84 by switch 85, it demmaps the information symbols contained in the time-domain signal after compensation processing and calculates the reliability information for each bit in each symbol.

[0179] The error correction decoding unit 34 in the wireless receiver 80 receives confidence information (LLR) for each bit in each symbol obtained by the demapping unit 33 and performs error correction decoding processing. The error correction decoding unit 34 then outputs the log-likelihood ratio (LLR) of each bit with improved confidence.

[0180] The bit mapping unit 81 generates an estimated value of an information symbol by remapping the bits that have been hard-determined as the log-likelihood ratio (LLR) of the output of the error correction decoding unit 34.

[0181] The phase detector (PD) 82 detects the phase difference between a signal obtained by the phase noise compensation processing unit 84, which compensates for phase fluctuations caused by phase noise for the information symbol of interest, and a reference signal (the estimated value of the information symbol generated by the bit mapping unit 81 is used as the reference signal).

[0182] Since the phase difference signal output of the phase detector (PD) 82 contains a noise (AWGN) component, the loop filter 83 generates an estimate of the residual phase noise by averaging the phase difference signal output of the phase detector (PD) 82 to suppress phase fluctuations caused by the noise.

[0183] The phase noise compensation processing unit 84 compensates for residual phase noise by multiplying the output signal of the equalizer (synchronous detector) 32 by the complex conjugate of the estimated residual phase noise of the output of the loop filter 83. By repeatedly performing this PLL process with the phase noise compensation processing unit 84 and the demapping unit 33 connected by switch 85, the residual phase noise can be suppressed to a low level.

[0184] Furthermore, by inserting an IDFT after the equalizer 32, a wireless receiver applicable to DFT-spread OFDM can be realized.

[0185] <Ninth Embodiment> The ninth embodiment relates to another example of a wireless receiving device. The basic configuration of the wireless receiving device of the ninth embodiment is the same as the basic configuration of the wireless receiving device 20 of the second embodiment.

[0186] Figure 28 is a block diagram showing an example of a wireless receiver using OFDM in the ninth embodiment. In Figure 28, the wireless receiver 90 includes an FFT unit 31, an equalizer (synchronous detector) 32, a demapping unit 33, and an error correction decoding unit 34. The wireless receiver 90 also includes a pilot signal extraction unit 41, a channel response estimation unit 42, a common phase error compensation unit 91, a demapping unit 92, a symbol estimation unit 93, an inter-carrier interference (ICI) estimation unit 94, and an inter-carrier interference (ICI) compensation unit 95. The FFT unit 31 corresponds to the Fast Fourier Transform (Discrete Fourier Transform) unit 24 described above. The equalizer (synchronous detector) 32 and the demapping unit 33 correspond to the equalization processing unit 25 described above. The error correction decoding unit 34 corresponds to the error correction decoding unit 27 described above. The pilot signal extraction unit 41, the channel response estimation unit 42, the common phase error compensation unit 91, the demapping unit 92, the symbol estimation unit 93, the inter-carrier interference estimation unit 94, and the inter-carrier interference compensation unit 95 correspond to the above-mentioned phase noise estimation and compensation unit 26.

[0187] As described above, phase noise imparts random phase rotation to the time-domain transmitted or received signal, resulting in a common phase error (phase rotation) (CPE) across all subcarriers in the frequency domain, as well as different inter-carrier interference between subcarriers. This inter-carrier interference is simply called inter-carrier interference (ICI). Therefore, for each subcarrier signal after the OFDM FFT processing, the CPE of the equalized (synchronously detected) symbol is compensated for. After demapping the CPE-compensated symbols, symbol estimates are generated, and inter-carrier interference (ICI) is estimated from the received symbol at each subcarrier position, the symbol estimates, the estimated channel response, and the weighting coefficients of the Minimum Mean Square Error (MMSE) standard. The ICI is then removed (compensated) from each symbol.

[0188] The time-domain block of the received signal that has undergone multipath fading is represented by the following equation (18).

number

[0189] In equation (18), x n This represents a pilot signal or information symbol. Also, h n This represents the channel impulse response. Also, w n This represents the background noise component. The received signal r, when the background noise component is not considered, is expressed as follows:

number

number

number

[0190] Received signal r n Frequency domain signal R k (k=0,1,···,N FFT -1) is represented by the following formula (19), as disclosed in Non-Patent Document 9.

number

[0191] In equation (19), X k ,H k ,η k These represent the information symbol, channel response, and background noise component of subcarrier k, respectively. k As shown in equation (20), the time-domain phase noise

number

number

[0192] In equation (20), the zero-frequency component J0 can be expressed as shown in equation (21).

number

[0193] In equation (21), Φ0 represents the average phase shift between blocks. Also, Δφ n Δφ represents the phase shift from Φ0 at the sample point. n Since is a very small value, the approximation in equation (21) holds. The zero-frequency component J0 is a phase rotation common to all subcarrier positions and is therefore called the Common Phase Error (CPE), and can be easily estimated. Also, the second term on the right-hand side of equation (19) is the inter-subcarrier interference (ICI), which differs depending on the subcarrier position. As shown in equation (19), the phase variation caused by phase noise in the time domain becomes the CPE and the inter-carrier interference (ICI), which differs at each subcarrier position, in the frequency domain. Inter-carrier interference J k Since estimating for all k would increase the computational cost, we estimate and compensate for up to a certain k-th order inter-carrier interference.

[0194] Equation (19) shows ε k This is the sum of the uncompensated residual ICI term and the external noise, with a mean of 0 and a variance of

number

number

[0195] From equation (22),

number

number

[0196] The common phase error compensation unit 91 of the wireless receiver 90 shown in Figure 28 estimates the CPE (Common Phase Error) according to equation (23), for example, using a pilot signal.

[0197] Using the CPE estimated by equation (23), the equalization weights for MMSE are given by the following equation (24).

number

[0198] In equation (24),

number

number

[0199] Using the CPE estimated by equation (23), the Maximal Ratio Combining (MRC) weight is given by the following equation (25).

number

[0200] The channel response estimation unit 42 estimates the channel response in subcarrier k and outputs it to the common phase error compensation unit 91 and the ICI estimation unit 94.

[0201] The common phase error compensation unit 91 compensates for CPE by multiplying the frequency domain received signal (frequency domain OFDM signal) by an equalization weighting coefficient calculated according to equation (25), for example.

[0202] The demapping unit 92 demmaps each symbol in the symbol sequence with CPE compensated and reconstructs the transmit bits.

[0203] The symbol estimation unit 93 generates an estimated value of the information symbol by remapping the hard decision bit of the transmitted bit.

[0204] As mentioned above, phase noise generates inter-subcarrier interference (ICI) in the frequency domain, which interferes with multiple subcarriers. Next, inter-carrier interference (ICI) is estimated using each symbol after FFT processing of the received signal, the estimated value of each symbol, and the estimated channel response at each symbol position. In actual systems, compensating for ICI of about second order can provide an estimation accuracy of phase noise that is close to the time-varying fluctuations of phase noise. Subcarrier index l

number

number

[0205] The frequency-domain received signal R for a subset L of the subcarrier is

number

[0206]

number

number

[0207] In equation (26),

number

number

[0208] Equation (26) is written in matrix notation,

number

number

[0209] Here, the component ε, which is the sum of uncompensated higher-order residual inter-carrier interference and background noise,

number

[0210] The matrix M can be obtained by the following equation (28).

number

[0211] In equation (28),

number

number

[0212] The carrier-to-carrier interference estimation unit 94 calculates an estimated value of the carrier-to-carrier interference using, for example, equations (27) and (28).

[0213] The inter-carrier interference compensation unit 95 generates symbols with compensated carrier interference by subtracting the inter-carrier interference estimated by the inter-carrier interference estimation unit 94 from each symbol of the output signal of the FFT unit 31.

[0214] <Tenth Embodiment> The tenth embodiment relates to another example of a wireless receiver. Specifically, in the ninth embodiment, a symbol estimate is generated using hard decision bits obtained by demapping the signal before error correction decoding (before channel coding decoding), whereas in the tenth embodiment, a symbol count estimate is generated from the decoded bits after error correction decoding. The basic configuration of the wireless receiver in the tenth embodiment is the same as the basic configuration of the wireless receiver 20 in the second embodiment.

[0215] Figure 29 is a block diagram showing an example of a wireless receiver in the tenth embodiment. In Figure 29, the wireless receiver 100 includes an FFT unit 31, an equalizer (synchronous detector) 32, a demapping unit 33, and an error correction decoding unit 34. The wireless receiver 100 also includes a pilot signal extraction unit 41, a channel response estimation unit 42, a common phase error compensation unit 91, a demapping unit 92, an error correction decoding unit 101, a symbol estimation unit 93, an inter-carrier interference generation unit 94, and an inter-carrier interference compensation unit 95. The FFT unit 31 corresponds to the Fast Fourier Transform (Discrete Fourier Transform) unit 24 described above. The equalizer (synchronous detector) 32 and the demapping unit 33 correspond to the equalization processing unit 25 described above. The error correction decoding unit 34 corresponds to the error correction decoding unit 27 described above. The pilot signal extraction unit 41, the channel response estimation unit 42, the common phase error compensation unit 91, the demapping unit 92, the error correction decoding unit 101, the symbol estimation unit 93, the inter-carrier interference generation unit 94, and the inter-carrier interference compensation unit 95 correspond to the above-mentioned phase noise estimation and compensation unit 26.

[0216] The demapping unit 92 in the wireless receiver 100 demapping the symbols after common phase error (CPE) compensation and calculating the log-likelihood ratio (LLR), which is the logarithm of the ratio of the probabilities that each bit in each symbol is "1" or "0 (-1)".

[0217] The error correction decoding unit 101 in the wireless receiver 100 deinterleaves the log-likelihood ratio of each bit in each symbol obtained by the demapping unit 92, inputs it, and performs error correction decoding. The error correction decoding unit 101 then outputs the decoded bits with improved reliability.

[0218] The symbol estimation unit 93 in the wireless receiver 100 generates an estimated value of an information symbol by mapping the hard decision bit of the transmitted bit reconstructed by the error correction decoding unit 101.

[0219] <Embodiment 11> The 11th embodiment relates to another example of a wireless receiver. In particular, the 11th embodiment relates to a configuration that corresponds to single-carrier DFT-spread OFDM. That is, the wireless receiver of the 11th embodiment has a configuration that corresponds to single-carrier DFT-spread OFDM, compared to the configuration of the wireless receiver 90 of the 9th embodiment that corresponds to multi-carrier OFDM. The basic configuration of the wireless receiver of the 11th embodiment is the same as the basic configuration of the wireless receiver 20 of the second embodiment.

[0220] Figure 30 is a block diagram showing an example of a wireless receiver in the 11th embodiment. In Figure 30, the wireless receiver 110 includes an FFT unit 31, an equalizer 32, an IDFT unit 61, a demapping unit 62, and an error correction decoding unit 34. The wireless receiver 110 also includes a pilot signal extraction unit 41, a channel response estimation unit 42, a frequency domain equalizer (FDE) 111, a common phase error compensation unit 91, an IDFT unit 112, a demapping unit 92, a symbol estimation unit 93, a DFT unit 113, an inter-carrier interference generation unit 94, and an inter-carrier interference compensation unit 95. The FFT unit 31 corresponds to the Fast Fourier Transform (Discrete Fourier Transform) unit 24 described above. The frequency domain (FDE) equalizer 32, the IDFT unit 61, and the demapping unit 62 correspond to the equalization processing unit 25 described above. The error correction decoding unit 34 corresponds to the error correction decoding unit 27 described above. The pilot signal extraction unit 41, the channel response estimation unit 42, the frequency domain (FDE) equalizer 111, the common phase error compensation unit 91, the IDFT unit 112, the demapping unit 92, the symbol estimation unit 93, the DFT unit 113, the inter-carrier interference generation unit 94, and the inter-carrier interference compensation unit 95 correspond to the above-mentioned phase noise estimation and compensation unit 26.

[0221] In the wireless receiver 110, the frequency domain equalizer (FDE) 111 and the common phase error compensation unit 91 equalize the signal by multiplying it by an equalization weighting coefficient calculated according to, for example, equation (24), to the single-carrier received signal in the frequency domain. Equalization processing to compensate for amplitude and phase fluctuations in the propagation path and CPE compensation are performed simultaneously.

[0222] In the wireless receiver 110, the IDFT unit 112 converts the frequency domain signal after equalization and CPE compensation into a time domain signal (a single-carrier symbol sequence).

[0223] The demapping unit 92 demmaps each symbol of the single carrier and reconstructs the transmitted bits.

[0224] The symbol estimation unit 93 generates an estimated value of the information symbol by remapping the hard decision bits.

[0225] The DFT unit 113 converts multiple information symbol estimates generated by the symbol estimation unit 93 into frequency domain signals.

[0226] <Twelfth Embodiment> The twelfth embodiment relates to another example of a wireless receiver. Specifically, in the eleventh embodiment, a symbol estimate was generated using hard decision bits obtained by demapping symbols before error correction decoding (before channel coding decoding), whereas in the twelfth embodiment, a symbol count estimate is generated from bits after error correction decoding. The basic configuration of the wireless receiver in the twelfth embodiment is the same as the basic configuration of the wireless receiver 20 in the second embodiment.

[0227] Figure 31 is a block diagram showing an example of a wireless receiver in the twelfth embodiment. In Figure 31, the wireless receiver 120 includes an FFT unit 31, a frequency-domain equalizer 32, an IDFT unit 61, a demapping unit 62, and an error correction decoding unit 34. The wireless receiver 120 also includes a pilot signal extraction unit 41, a channel response estimation unit 42, a frequency-domain equalizer 111, a common phase error compensation unit 91, an IDFT unit 112, a demapping unit 92, an error correction decoding unit 101, a symbol estimation unit 93, a DFT unit 113, an inter-carrier interference generation unit 94, and an inter-carrier interference compensation unit 95. The FFT unit 31 corresponds to the Fast Fourier Transform (Discrete Fourier Transform) unit 24 described above. The frequency-domain equalizer 32, the IDFT unit 61, and the demapping unit 62 correspond to the equalization processing unit 25 described above. The error correction decoding unit 34 corresponds to the error correction decoding unit 27 described above. The pilot signal extraction unit 41, the channel response estimation unit 42, the frequency domain equalizer 111, the common phase error compensation unit 91, the IDFT unit 112, the demapping unit 92, the error correction decoding unit 101, the symbol estimation unit 93, the DFT unit 113, the inter-carrier interference generation unit 94, and the inter-carrier interference compensation unit 95 correspond to the above-mentioned phase noise estimation and compensation unit 26.

[0228] In the wireless receiver 120, the IDFT unit 112 converts the frequency domain signal after equalization and common phase error (CPE) compensation into a time domain single carrier signal.

[0229] The mapping unit 92 demaps the time-domain information symbols after equalization and common phase error (CPE) compensation, and calculates the log-likelihood ratio (LLR), which is the logarithm of the ratio of the probabilities that each bit in each symbol is "1" or "0 (-1)".

[0230] The error correction decoding unit 101 takes the log-likelihood ratio of each bit in each symbol obtained by the demapping unit 92 as input after deinterleaving and performs error correction decoding. The error correction decoding unit 101 then outputs the decoded bits with improved reliability.

[0231] The symbol estimation unit 93 generates an estimated value of the information symbol by mapping the hard decision bit of the transmitted bit reconstructed by the error correction decoding unit 101.

[0232] The DFT unit 113 converts the single-carrier information symbol estimates generated by the symbol estimation unit 93 into frequency domain signals.

[0233] <Other Embodiments> Figure 32 shows an example of the hardware configuration of a wireless transmitter. In Figure 32, the wireless transmitter 200 includes a communication circuit 201, a processor 202, and a memory 203. The processor 202 may be, for example, a microprocessor, an MPU (Micro Processing Unit), or a CPU (Central Processing Unit). The processor 202 may include multiple processors. The memory 203 is composed of a combination of volatile memory and non-volatile memory. The memory 203 may include storage located away from the processor 202. In this case, the processor 202 may access the memory 203 via an I (input) / O (output) interface, which is not shown.

[0234] The wireless transmitter 10 of the first embodiment may have the hardware configuration shown in Figure 32. The information symbol output unit 11, the pilot signal output unit 12, and the OFDM signal formation unit 13 of the wireless transmitter 10 of the first embodiment may be realized by a processor 202 reading and executing a program stored in memory 203. The transmitting wireless unit 14 may be realized by a communication circuit 201. The program can be stored using various types of non-transitory computer-readable media and supplied to the wireless transmitter 10. Examples of non-transitory computer-readable media include magnetic recording media (e.g., flexible disks, magnetic tapes, hard disk drives) and magneto-optical recording media (e.g., magneto-optical disks). Furthermore, examples of non-transitory computer-readable media include CD-ROMs (Read Only Memory), CD-Rs, and CD-R / Ws. Furthermore, examples of non-transitory computer-readable media include semiconductor memory. Semiconductor memory includes, for example, mask ROM, PROM (Programmable ROM), EPROM (Erasable PROM), flash ROM, and RAM (Random Access Memory). The program may also be supplied to the wireless transmitter 10 by various types of transient computer-readable medium. Examples of transient computer-readable medium include electrical signals, optical signals, and electromagnetic waves. The transient computer-readable medium can supply the program to the wireless transmitter 10 via wired communication channels such as electric wires and optical fibers, or via wireless communication channels.

[0235] Figure 33 shows an example of the hardware configuration of a wireless receiver. In Figure 33, the wireless receiver 300 includes a communication circuit 301, a processor 302, and a memory 303. The processor 302 may be, for example, a microprocessor, an MPU (Micro Processing Unit), or a CPU (Central Processing Unit). The processor 302 may include multiple processors. The memory 303 is composed of a combination of volatile memory and non-volatile memory. The memory 303 may include storage located away from the processor 302. In this case, the processor 302 may access the memory 303 via an I (input) / O (output) interface, which is not shown.

[0236] The wireless receivers 20, 50, 60, 70, 80, 90, 100, 110, and 120 of the second to twelfth embodiments may each have the hardware configuration shown in Figure 33. The signal processing unit 22 of the wireless receivers 20, 50, 60, 70, 80, 90, 100, 110, and 120 of the second to twelfth embodiments may be implemented by a processor 302 reading and executing a program stored in memory 303. The receiving wireless unit 21 may be implemented by a communication circuit 301. The program can be stored using various types of non-transitory computer-readable medium and supplied to the wireless receivers 20, 50, 60, 70, 80, 90, 100, 110, and 120. Examples of non-transitory computer-readable mediums include magnetic recording media (e.g., flexible disks, magnetic tapes, hard disk drives) and magneto-optical recording media (e.g., magneto-optical disks). Furthermore, examples of non-transitory computer-readable media include CD-ROMs (Read Only Memory), CD-Rs, and CD-R / Ws. Additionally, examples of non-transitory computer-readable media include semiconductor memory. Semiconductor memory includes, for example, mask ROMs, PROMs (Programmable ROMs), EPROMs (Erasable PROMs), flash ROMs, and RAMs (Random Access Memory). Programs may also be supplied to wireless receivers 20, 50, 60, 70, 80, 90, 100, 110, and 120 by various types of transient computer-readable media. Examples of transient computer-readable media include electrical signals, optical signals, and electromagnetic waves. Transitory computer-readable media can supply programs to wireless receivers 20, 50, 60, 70, 80, 90, 100, 110, and 120 via wired communication channels such as electric wires and optical fibers, or via wireless communication channels.

[0237] Although the present invention has been described above with reference to embodiments, the present invention is not limited thereto. Various modifications to the structure and details of the present invention can be made that are understandable to those skilled in the art within the scope of the invention.

[0238] Some or all of the above embodiments may also be described as follows, but are not limited to the following: (Note 1) A signal forming unit that forms an OFDM (Orthogonal Frequency Division Multiplexing) signal including a pilot signal, A transmitting radio unit that forms a wireless signal from the OFDM signal formed above, It is equipped with, The signal forming unit, Of the total subcarriers of the aforementioned OFDM signal, N Pilot_Freq The pilot signals are placed on multiple pilot-placed subcarriers at subcarrier intervals, Among the aforementioned multiple pilot-placed subcarriers, N Pilot_SPA In a subcarrier with multiple first pilot placement subcarriers at subcarrier intervals, each resource element places the pilot signal on all resource elements defined by the combination of subcarrier and OFDM symbol. Among the plurality of pilot-configured subcarriers, the plurality of second pilot-configured subcarriers excluding the plurality of first pilot-configured subcarriers, N in the time domain Pilot_Time The pilot signals are arranged in a period of one resource element. The pilot signals are arranged according to the pilot placement pattern. Wireless transmitter. (Note 2) The aforementioned OFDM signal is a DFT (Discrete Fourier Transformation) spread OFDM signal. The signal forming unit, A conversion unit that converts pilot symbol blocks, which are blocks of time-domain pilot symbols, into frequency-domain signals using a discrete Fourier transform, A mapping unit that maps the frequency domain signal to the pilot-arranged subcarrier, The inverse transform unit converts the mapped frequency domain signal into a time domain signal by an inverse discrete Fourier transform, Equipped with, The wireless transmitting device described in Appendix 1. (Note 3) The aforementioned wireless transmitting device comprises a plurality of antennas, including a first antenna and a second antenna. The signal forming unit, In the OFDM signal transmitted by the first antenna, the pilot signal is positioned according to the pilot placement pattern. In the OFDM signal transmitted by the second antenna, the pilot signal is placed on multiple pilot-arranged subcarriers obtained by shifting the multiple pilot-arranged subcarriers of the pilot-arranged pattern by one subcarrier in the frequency direction, using the same pattern as the pilot-arranged pattern. The wireless transmitting device described in Appendix 1. (Note 4) N TxAnt This transmits an antenna, A signal forming unit that forms an OFDM (Orthogonal Frequency Division Multiplexing) signal including a pilot signal, A transmitting radio unit that forms a wireless signal from the OFDM signal formed above, It is equipped with, The signal forming unit, Of the total subcarriers of the OFDM signal, each set is N TxAnt It contains a set of consecutive subcarriers, and the interval between each set of two adjacent subcarriers is N. Pilot_Freq The pilot signals are placed on a set of multiple pilot-placed subcarriers, which are subcarriers. Among the aforementioned multiple pilot placement subcarrier sets, N Pilot_SPA In a set of multiple first pilot-placed subcarriers with subcarrier spacing, each resource element is defined by a combination of a subcarrier and an OFDM symbol, and the pilot signal is placed on all resource elements. In the plurality of second pilot-configured subcarrier sets, excluding the plurality of first pilot-configured subcarrier sets, N in the time domain Pilot_Time The pilot signals are arranged in a period of one resource element. The pilot signals are arranged according to the pilot placement pattern. The aforementioned N TxAnt The multiple pilot signals transmitted from each of the transmitting antennas are spread out by a spread sequence specific to each of the different transmitting antennas. Wireless transmitter. (Note 5) A signal forming unit that forms an OFDM (Orthogonal Frequency Division Multiplexing) signal including a pilot signal, A transmitting radio unit that forms a wireless signal from the OFDM signal formed above, It is equipped with, The signal forming unit, Of the total subcarriers of the aforementioned OFDM signal, N Pilot_Freq The pilot signals are placed on multiple pilot-placed subcarriers at subcarrier intervals, In each pilot-configured subcarrier, N in the time domain Pilot_Time The pilot signals are arranged in a cycle of the resource element, In each of two adjacent pilot placement subcarriers, the pilot signals are arranged such that the resource elements on which the pilot signals are placed do not overlap in the time domain. The pilot signals are arranged according to the pilot placement pattern. Wireless transmitter. (Note 6) The signal forming unit arranges the pilot signal with respect to the first and second pilot placement subcarriers, which are aligned in the frequency direction among the two adjacent pilot placement subcarriers, such that the time domain position of the first resource element on which the pilot signal is placed in the second pilot placement subcarrier is adjacent in time domain to the time domain position of the first resource element on which the pilot signal is placed in the first pilot placement subcarrier. The wireless transmitting device described in Appendix 5. (Note 7) The aforementioned wireless transmitting device comprises a plurality of antennas, including a first antenna and a second antenna. The signal forming unit, In the OFDM signal transmitted by the first antenna, the pilot signal is positioned according to the pilot placement pattern. In the OFDM signal transmitted by the second antenna, the pilot signal is placed on multiple pilot subcarriers obtained by shifting the multiple pilot subcarriers of the pilot placement pattern by one subcarrier in the frequency direction, using the same pattern as the pilot placement pattern. The wireless transmitting device described in Appendix 5. (Note 8) N TxAnt This transmits an antenna, A signal forming unit that forms an OFDM (Orthogonal Frequency Division Multiplexing) signal including a pilot signal, A transmitting radio unit that forms a wireless signal from the OFDM signal formed above, It is equipped with, The signal forming unit, Of the total subcarriers of the OFDM signal, each set is N TxAnt It contains a set of consecutive subcarriers, and the interval between each set of two adjacent subcarriers is N. Pilot_Freq The pilot signals are placed on a set of multiple pilot-placed subcarriers, which are subcarriers. In each pilot-configured subcarrier set, N in the time domain Pilot_TimeThe pilot signals are arranged in a cycle of the resource element, In each of two adjacent pilot placement subcarrier sets, the pilot signals are arranged such that the resource elements on which the pilot signals are placed do not overlap in the time domain. The pilot signals are arranged according to the pilot placement pattern. The aforementioned N TxAnt The multiple pilot signals transmitted from each of the transmitting antennas are spread out by a spread sequence specific to each of the different transmitting antennas. Wireless transmitter. (Note 9) The signal forming unit, In each pilot placement subcarrier set, the pilot signals are arranged such that the resource elements for which the pilot signals are placed overlap in time in each pilot placement subcarrier. With respect to the first and second pilot placement subcarrier sets, which are aligned in the frequency direction among the two adjacent pilot placement subcarrier sets, the pilot signal is positioned such that the time domain position of the first resource element on which the pilot signal is placed in the second pilot placement subcarrier set is adjacent in time domain to the time domain position of the first resource element on which the pilot signal is placed in the first pilot placement subcarrier set. The wireless transmitting device described in Appendix 8. (Note 10) It comprises a signal processing unit that performs signal processing on OFDM (Orthogonal Frequency Division Multiplexing) signals, The signal processing unit, A phase noise compensation unit that estimates the phase noise of the received OFDM signal based on the pilot signal included in the received OFDM signal and compensates for the phase noise of the received OFDM signal, A conversion unit that receives a first time-domain signal and converts the received first time-domain signal into a frequency-domain signal, An equalization unit that performs equalization processing on the frequency domain signal obtained by the conversion unit, A demapping unit that demmaps the information symbols contained in the frequency domain signal after the equalization process and reconstructs the information bits, An extraction unit that extracts a pilot signal included in the frequency domain signal obtained by the conversion unit, A channel estimation unit that estimates the channel response using the extracted pilot signal, A symbol estimation unit that maps the regenerated information bits to symbols to obtain regenerated symbols, A first multiplication unit multiplies the obtained regenerated symbol by the estimated channel response, An inverse transformer unit that converts the regenerated symbols multiplied by the channel response into a second time-domain signal, A phase noise compensation signal generation unit that estimates the phase noise using the first time-domain signal and the second time-domain signal obtained by the inverse transform unit, and outputs the inverse characteristics of the estimated phase noise, A second multiplier is provided at the input stage of the conversion unit, which compensates for the phase noise of the received OFDM signal by multiplying the inverse characteristic of the output phase noise by the received OFDM signal after the phase noise has been compensated by the phase noise compensation unit, and outputs the received OFDM signal after phase noise compensation to the conversion unit as the first time-domain signal. including, Wireless receiving device. (Note 11) It comprises a signal processing unit that performs signal processing on OFDM (Orthogonal Frequency Division Multiplexing) signals, The signal processing unit, A phase noise compensation unit that estimates the phase noise of the received OFDM signal based on the pilot signal included in the received OFDM signal and compensates for the phase noise of the OFDM signal, A conversion unit that receives a first time-domain signal and converts the received first time-domain signal into a frequency-domain signal, An equalization unit that performs equalization processing on the frequency domain signal obtained by the conversion unit, A demapping unit that demmaps the information symbols contained in the frequency domain signal after the equalization process and calculates the reliability information of each bit in each symbol, A decoding unit that performs error correction decoding using the confidence information of each bit and outputs the obtained decoded bits, An extraction unit that extracts a pilot signal included in the frequency domain signal obtained by the conversion unit, A channel estimation unit that estimates the channel response using the extracted pilot signal, A symbol estimation unit maps the decoded bits output from the decoding unit to symbols to obtain a reconstructed symbol, A first multiplication unit multiplies the obtained regenerated symbol by the estimated channel response, An inverse transformer unit that converts the regenerated symbols multiplied by the channel response into a second time-domain signal, A phase noise compensation signal generation unit that estimates the phase noise using the first time-domain signal and the second time-domain signal obtained by the inverse transform unit, and outputs the inverse characteristics of the estimated phase noise, A second multiplier is provided at the input stage of the conversion unit, which compensates for the phase noise of the received OFDM signal by multiplying the inverse characteristic of the output phase noise by the received OFDM signal after the phase noise has been compensated by the phase noise compensation unit, and outputs the received OFDM signal after phase noise compensation to the conversion unit as the first time-domain signal. including, Wireless receiving device. (Note 12) It comprises a signal processing unit that performs signal processing on OFDM (Orthogonal Frequency Division Multiplexing) signals, The signal processing unit, A phase noise compensation unit that estimates the phase noise of the received OFDM signal based on the pilot signal included in the received OFDM signal and compensates for the phase noise of the received OFDM signal, A first conversion unit that receives a first time-domain signal and converts the received first time-domain signal into a first frequency-domain signal, An equalization unit that performs an equalization process on the first frequency domain signal obtained by the first conversion unit, A first inverse transformer converts the frequency domain signal after the equalization process into an information symbol, which is a second time domain signal. A demapping unit that demmaps the aforementioned information symbols and reconstructs the information bits, An extraction unit that extracts a pilot signal included in the first frequency domain signal obtained by the first conversion unit, A channel estimation unit that estimates the channel response using the extracted pilot signal, A symbol estimation unit that maps the regenerated information bits to symbols to obtain regenerated symbols, A second conversion unit that converts the aforementioned regenerated symbol into a symbol which is a second frequency domain signal, A mapping unit that maps the symbol, which is the second frequency domain signal, to a subcarrier, A first multiplication unit multiplies the symbols mapped to the subcarriers by the estimated channel response, A second inverse transform unit that converts the symbol multiplied by the channel response into a third time-domain signal, A phase noise compensation signal generation unit that estimates the phase noise using the first time-domain signal and the third time-domain signal obtained by the second inverse transform unit, and outputs the inverse characteristics of the estimated phase noise, A second multiplier is provided at the input stage of the first conversion unit, which compensates for the phase noise of the received OFDM signal by multiplying the inverse characteristic of the output phase noise by the received OFDM signal after the phase noise has been compensated by the phase noise compensation unit, and outputs the received OFDM signal after phase noise compensation to the first conversion unit as the first time-domain signal. including, Wireless receiving device. (Note 13) It comprises a signal processing unit that performs signal processing on OFDM (Orthogonal Frequency Division Multiplexing) signals, The signal processing unit, A phase noise compensation unit that estimates the phase noise of the received OFDM signal based on the pilot signal included in the received OFDM signal and compensates for the phase noise of the received OFDM signal, A first conversion unit that receives a first time-domain signal and converts the received first time-domain signal into a first frequency-domain signal, An equalization unit that performs an equalization process on the first frequency domain signal obtained by the first conversion unit, A first inverse transformer converts the frequency domain signal after the equalization process into an information symbol, which is a second time domain signal. A demapping unit that demmaps the aforementioned information symbols and calculates the reliability information of each bit in each symbol, A decoding unit that performs error correction decoding using the confidence information of each bit and outputs the obtained decoded bits, An extraction unit that extracts a pilot signal included in the first frequency domain signal obtained by the first conversion unit, A channel estimation unit that estimates the channel response using the extracted pilot signal, A symbol estimation unit maps the decoded bits output from the decoding unit to symbols to obtain a reconstructed symbol, A second conversion unit that converts the aforementioned regenerated symbol into a symbol which is a second frequency domain signal, A mapping unit that maps the symbol, which is the second frequency domain signal, to a subcarrier, A first multiplication unit multiplies the symbols mapped to the subcarriers by the estimated channel response, A second inverse transform unit that converts the symbol multiplied by the channel response into a third time-domain signal, A phase noise compensation signal generation unit that estimates the phase noise using the first time-domain signal and the third time-domain signal obtained by the second inverse transform unit, and outputs the inverse characteristics of the estimated phase noise, A second multiplier is provided at the input stage of the first conversion unit, which compensates for the phase noise of the received OFDM signal by multiplying the inverse characteristic of the output phase noise by the received OFDM signal after the phase noise has been compensated by the phase noise compensation unit, and outputs the received OFDM signal after phase noise compensation to the first conversion unit as the first time-domain signal. including, Wireless receiving device. (Note 14) The phase noise compensation unit is An output unit that outputs multiple basis functions, A weight generation unit that uses the pilot signal included in the received OFDM signal and the plurality of basis functions to calculate the mean squared error of the linear sum of the phase noise included in the received OFDM signal and the result of multiplying each basis function by a weight coefficient, and generates weight coefficients for each basis function to minimize the calculated mean squared error, A phase noise calculation unit calculates the phase noise by obtaining a linear sum of basis functions using the generated weight coefficients, including, The wireless receiving device described in Appendix 10. (Note 15) The phase noise compensation signal generation unit generates N OFDM symbols corresponding to the FFT block. FFT N samples Blk N of each block divided into individual blocks FFT / N Blk The estimated phase noise values ​​of each sample are independently averaged with respect to the in-phase component and the orthogonal component, and the obtained N Blk The phase noise at each sample point is estimated by interpolating the average of the phase noise estimates for each block using linear interpolation, quadratic interpolation, or interpolation using a higher-order function. The wireless receiving device described in Appendix 10. (Note 16) The phase noise compensation signal generation unit is Using the first time-domain signal and the second time-domain signal obtained by the inverse transform unit, the N OFDM symbol corresponding to the FFT block is calculated. FFT A means for estimating the phase noise of individual sample positions, A means for averaging the estimated phase noise of each sample across OFDM symbols using an adaptive algorithm for the least mean squared error criterion, including, The wireless receiving device described in Appendix 10. (Note 17) It comprises a signal processing unit that performs signal processing on OFDM (Orthogonal Frequency Division Multiplexing) signals, The signal processing unit, A phase noise compensation unit that estimates the phase noise of the received OFDM signal based on the pilot signal included in the received OFDM signal and compensates for the phase noise of the received OFDM signal, A conversion unit that receives the received OFDM signal after the phase noise has been compensated by the phase noise compensation unit, and converts the received OFDM signal into a frequency domain signal, An equalization unit that performs equalization processing on the frequency domain signal obtained by the conversion unit, A demapping unit that demmaps the information symbols contained in the frequency domain signal after the equalization process and calculates the reliability information of each bit in each symbol, A decoding unit that performs error correction decoding using the confidence information of each bit and outputs the obtained decoded bits, A detection unit for detecting the phase difference between the frequency domain signal after the equalization process and the reference signal, A loop filter that outputs the phase difference obtained by reducing the noise component of the detected phase difference, A phase noise compensation processing unit is provided at the input stage of the demapping unit and uses the output phase difference to compensate for the phase noise of the frequency domain signal after the equalization process, A symbol estimation unit that maps the decoded bits to symbols to obtain a reconstructed symbol and outputs the reconstructed symbol as the reference signal, including, Wireless receiving device. (Note 18) It comprises a signal processing unit that performs signal processing on OFDM (Orthogonal Frequency Division Multiplexing) signals, A phase noise compensation unit that estimates the phase noise of the received OFDM signal based on the pilot signal included in the received OFDM signal and compensates for the phase noise of the received OFDM signal, A conversion unit that receives the received OFDM signal after the phase noise has been compensated by the phase noise compensation unit, and converts the received OFDM signal into a frequency domain signal, An equalization unit that performs equalization processing on the frequency domain signal obtained by the conversion unit, An extraction unit that extracts a pilot signal included in the frequency domain signal obtained by the conversion unit, A channel estimation unit that estimates the channel response using the extracted pilot signal, A demapping unit that demmaps the information symbols contained in the frequency domain signal obtained by the conversion unit and reconstructs the information bits, A symbol estimation unit that maps the regenerated information bits to symbols to obtain regenerated symbols, A calculation unit calculates the discrete Fourier transform coefficients of the phase noise based on the information symbols included in the frequency domain signal obtained by the conversion unit, the estimated channel response, and the reproduced symbols, using the mean square error minimum criterion. Distributed at the input stage of the equalization unit, an inter-carrier interference compensation unit performs convolution processing using the frequency domain signal obtained in the conversion unit and the calculated discrete Fourier transform coefficients to compensate for the phase noise of the received OFDM signal after the phase noise has been compensated by the phase noise compensation unit, including, Wireless receiving device. [Explanation of Symbols]

[0239] 10 Wireless Transmitter 11. Information Symbol Output Unit 12 Pilot signal output section 13 OFDM signal forming section 14. Transmitting Radio Unit 20, 50, 60, 70, 80, 90, 100, 110, 120 Wireless receiver 21 Receiving Radio Unit 22 Signal Processing Unit

Claims

1. A signal forming unit that forms an OFDM (Orthogonal Frequency Division Multiplexing) signal including a pilot signal, A transmitting radio unit that forms a wireless signal from the OFDM signal formed above, It is equipped with, The signal forming unit, Of the total subcarriers of the OFDM signal, N Pilot_Freq The pilot signals are placed on multiple pilot-placed subcarriers at subcarrier intervals, Among the aforementioned multiple pilot-placed subcarriers, N Pilot_ＳＰＡ In a subcarrier with multiple first pilot placement subcarriers at subcarrier intervals, each resource element places the pilot signal on all resource elements defined by the combination of subcarrier and OFDM symbol. Among the plurality of pilot-configured subcarriers, the plurality of second pilot-configured subcarriers excluding the plurality of first pilot-configured subcarriers, N in the time domain Pilot_Ｔｉｍｅ The pilot signals are arranged such that the resource elements on which the pilot signals are arranged overlap in the time domain, with each resource element period and with each of two adjacent second pilot-arranged subcarriers. The pilot signals are arranged according to the pilot placement pattern. Wireless transmitter.

2. The OFDM signal is a DFT (Discrete Fourier Transformation) spread OFDM signal. The signal forming unit, A conversion unit that converts pilot symbol blocks, which are blocks of time-domain pilot symbols, into frequency-domain signals using a discrete Fourier transform, A mapping unit that maps the frequency domain signal to the pilot-arranged subcarrier, The inverse transform unit converts the mapped frequency domain signal into a time domain signal by an inverse discrete Fourier transform, Equipped with, The wireless transmitting device according to claim 1.

3. The wireless transmitting device comprises a plurality of antennas, including a first antenna and a second antenna. The signal forming unit, In the OFDM signal transmitted by the first antenna, the pilot signal is positioned according to the pilot placement pattern. In the OFDM signal transmitted by the second antenna, the pilot signal is placed on multiple pilot-arranged subcarriers obtained by shifting the multiple pilot-arranged subcarriers of the pilot-arranged pattern by one subcarrier in the frequency direction, using the same pattern as the pilot-arranged pattern. The wireless transmitting device according to claim 1.

4. N ＴｘＡｎｔ This transmits an antenna, A signal forming unit that forms an OFDM (Orthogonal Frequency Division Multiplexing) signal including a pilot signal, A transmitting radio unit that forms a wireless signal from the OFDM signal formed above, It is equipped with, The signal forming unit, Of the total subcarriers of the OFDM signal, each set is N ＴｘＡｎｔ It contains a set of consecutive subcarriers, and the interval between each set of two adjacent subcarriers is N. Pilot_Freq The pilot signals are placed on a set of multiple pilot-placed subcarriers, which are subcarriers. Among the aforementioned multiple pilot placement subcarrier sets, N Pilot_ＳＰＡ In a set of multiple first pilot-placed subcarriers with subcarrier spacing, each resource element is defined by a combination of a subcarrier and an OFDM symbol, and the pilot signal is placed on all resource elements. In the plurality of second pilot-arranged subcarrier sets, excluding the plurality of first pilot-arranged subcarrier sets, N in the time domain Pilot_Ｔｉｍｅ The pilot signals are arranged in a period of one resource element. The pilot signals are arranged according to the pilot placement pattern. The foregoing N ＴｘＡｎｔ The plurality of pilot signals transmitted respectively from the N transmission antennas are spread by spreading sequences unique to the different transmission antennas from each other. Wireless transmitter.

5. A signal forming unit that forms an OFDM (Orthogonal Frequency Division Multiplexing) signal including a pilot signal, A transmitting radio unit that forms a wireless signal from the OFDM signal formed above, It is equipped with, The signal forming unit, Of the total subcarriers of the OFDM signal, N Pilot_Freq The pilot signals are placed on multiple pilot-placed subcarriers at subcarrier intervals, In each pilot-configured subcarrier, N in the time domain Pilot_Ｔｉｍｅ The pilot signals are arranged in a cycle of the resource element, In each of the two adjacent pilot placement subcarriers, the pilot signals are arranged such that the resource elements on which the pilot signals are placed do not overlap in the time domain. The pilot signals are arranged according to the pilot placement pattern. Wireless transmitter.

6. The signal forming unit arranges the pilot signals with respect to the first and second pilot placement subcarriers, which are aligned in the frequency direction among the two adjacent pilot placement subcarriers, such that the time domain position of the first resource element on which the pilot signal is placed in the second pilot placement subcarrier is adjacent in time domain to the time domain position of the first resource element on which the pilot signal is placed in the first pilot placement subcarrier. The wireless transmitting device according to claim 5.

7. The wireless transmitting device comprises a plurality of antennas, including a first antenna and a second antenna. The signal forming unit, In the OFDM signal transmitted by the first antenna, the pilot signal is positioned according to the pilot placement pattern. In the OFDM signal transmitted by the second antenna, the pilot signal is placed on multiple pilot subcarriers obtained by shifting the multiple pilot subcarriers of the pilot placement pattern by one subcarrier in the frequency direction, using the same pattern as the pilot placement pattern. The wireless transmitting device according to claim 5.

8. N ＴｘＡｎｔ This transmits an antenna, A signal forming unit that forms an OFDM (Orthogonal Frequency Division Multiplexing) signal including a pilot signal, A transmitting radio unit that forms a wireless signal from the OFDM signal formed above, It is equipped with, The signal forming unit, Of the total subcarriers of the OFDM signal, each set is N ＴｘＡｎｔ It contains a set of consecutive subcarriers, and the interval between each set of two adjacent subcarriers is N. Pilot_Freq The pilot signals are placed on a set of multiple pilot-placed subcarriers, which are subcarriers. In each pilot-configured subcarrier set, N in the time domain Pilot_Ｔｉｍｅ The pilot signals are arranged in a cycle of the resource element, In each of two adjacent pilot placement subcarrier sets, the pilot signals are arranged such that the resource elements on which the pilot signals are placed do not overlap in the time domain. The pilot signals are arranged according to the pilot placement pattern. The aforementioned N ＴｘＡｎｔ The multiple pilot signals transmitted from each of the transmitting antennas are spread out by a spread sequence specific to each of the different transmitting antennas. Wireless transmitter.

9. The signal forming unit, In each pilot placement subcarrier set, the pilot signals are arranged such that the resource elements for which the pilot signals are placed overlap in time in each pilot placement subcarrier. With respect to the first and second pilot placement subcarrier sets, which are aligned in the frequency direction among the two adjacent pilot placement subcarrier sets, the pilot signal is positioned such that the time domain position of the first resource element on which the pilot signal is placed in the second pilot placement subcarrier set is adjacent in time domain to the time domain position of the first resource element on which the pilot signal is placed in the first pilot placement subcarrier set. The wireless transmitting device according to claim 8.

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