A broadband load modulation balanced amplifier with harmonic suppression function

Abstract

The present invention discloses a broadband load modulation balanced amplifier with harmonic suppression function. The power amplifier is characterized by the use of a dual transmission line bandpass matching circuit. In the target design frequency band, this structure can achieve broadband matching of the optimal complex impedance of the power amplifier, and insert multiple transmission zeros in the second and third harmonic frequency bands to achieve the effect of suppressing harmonics. The circuit consists of ten sections of transmission lines, namely, pads, two groups of dual transmission lines on the main transmission path, two half-wavelength stub transmission lines connected in parallel at the same node, a stepped stub transmission line located between the two groups of dual transmission lines, and a quarter-wavelength stub transmission line grounded. The power amplifier proposed by the present invention is a broadband load modulation balanced amplifier with harmonic suppression function, which is applicable to the broadband and multi-band application scenarios of communication systems and can meet their requirements for communication bandwidth and linearity.

Description

Technical Field

[0001] The present invention relates to the technical field of radio frequency power amplifiers, and particularly to a broadband load modulation balanced amplifier with harmonic suppression function. Background Art

[0002] In recent years, various high-speed broadband applications, such as virtual reality, high-definition video, etc., have continuously promoted the development of wireless communication, and put forward higher requirements for all aspects of wireless communication technology and communication equipment. In order to improve the data transmission rate and spectrum utilization rate, orthogonal frequency division multiplexing (OFDM) technology, MIMO technology, and high-order modulation technologies such as M-QAM are widely used, resulting in a significant increase in the peak-to-average power ratio (PAPR) of signals. In order to ensure that high PAPR signals are amplified efficiently, it is necessary to improve the efficiency of the power amplifier back-off region. As one of the methods to improve the back-off performance of the power amplifier, the broadband load modulation balanced amplifier is particularly suitable for the efficient amplification of high PAPR signals. In addition, due to the inherent non-linear characteristics of the power amplifier, high PAPR signals will generate serious non-linear distortion after being amplified by the power amplifier, resulting in an increase in the bit error rate in the band and adjacent channel interference outside the band. Therefore, improving the linearity of the power amplifier is a major problem to be solved.

[0003] In the design of power amplifier circuits, suppressing harmonics is the most direct way to improve the output linearity. Usually, a filtering network is used as part of the matching network to improve the linearity of the power amplifier. However, in the design of broadband power amplifiers, due to the wide operating bandwidth of the power amplifier, it is still difficult to suppress the harmonic frequency bands of the power amplifier. As one of the alternative matching network schemes, the band-pass matching network has good out-of-band suppression effect. On this basis, the author of the literature "Implementation of Flat Gain Broadband Power Amplifier With Impedance Rotation Compensation" introduced a band-pass filter structure into the design of a band-pass broadband matching network based on filter technology, and derived detailed calculation formulas. In order not to introduce lumped elements, the author used a half-wavelength stub microstrip line to replace the original grounded quarter-wavelength stub transmission line. This structure can achieve good fundamental wave impedance matching and has obvious advantages in terms of bandwidth. However, there are great deficiencies in the harmonic impedance regulation part. Currently, the optimization schemes for the harmonic impedance of the band-pass matching network are as follows:

[0004] 1) Convert the half-wavelength stub transmission line of the band-pass matching network into a stepped stub transmission line;

[0005] 2) Cascade a low-pass network and a band-pass network to jointly achieve fundamental wave impedance matching and harmonic impedance matching optimization.

[0006] The effects of the two methods are not satisfactory. Therefore, the band-pass matching network can still be further improved in terms of harmonic impedance optimization. Summary of the Invention

[0007] In order to meet the requirements of modern wireless mobile communication systems for the linearity of power amplifiers, the present invention provides a new solution. By suppressing harmonics of the power amplifier based on a dual-transmission-line band-pass matching circuit, the linearity of a broadband load-modulated balanced amplifier is improved.

[0008] The technical solution adopted by the present invention to solve the technical problem is as follows:

[0009] A broadband load-modulated balanced amplifier with harmonic suppression function, comprising an input port Port1, an output port Port2, a power divider, a phase-shift line, a control signal amplifier, a balanced amplifier, and a post-matching network; the input port Port1 is connected to the input end of the power divider, the output ends of the power divider are respectively connected to the control signal amplifier and the phase-shift line, the phase-shift line is connected to the input port of the coupler (Z0 = 50) of the balanced amplifier, the control signal amplifier is connected to the isolation port of the output coupler (Z0 = 30) of the balanced amplifier, the output port of the output coupler (Z0 = 30) of the balanced amplifier is connected to the post-matching network, and the post-matching network is connected to the output port Port2; the control signal amplifier includes a control signal amplifier input matching network, a control signal amplifier power transistor, and a control signal amplifier output matching network connected in sequence; the balanced amplifier includes a coupler (Z0 = 50) connected in sequence, the isolation end of the coupler (Z0 = 50) is connected to a 50Ω resistor, the through and coupled output ends of the coupler (Z0 = 50) are connected to the same power amplifier BA and a coupler (Z0 = 30); the power amplifier BA includes a power amplifier BA input matching network, a power amplifier BA power transistor, and a dual-transmission-line band-pass matching network based on the power amplifier BA connected in sequence;

[0010] The working bandwidths of the power divider, the post-matching network, the coupler (Z0 = 50), and the coupler (Z0 = 30) can all cover the working frequency band of the broadband load-modulated balanced amplifier; the power divider is a non-equal power divider; the coupler (Z0 = 50) and the coupler (Z0 = 30) are broadband branch-line couplers, the terminating load of the coupler (Z0 = 50) is 50 ohms, and the terminating load of the coupler (Z0 = 30) is 30 ohms; the post-matching network is an impedance conversion circuit for realizing the conversion from 30 ohms to 50 ohms;

[0011] The control signal amplifier input matching network and the power amplifier BA input matching network are hybrid circuits including a stability circuit, a DC bias circuit, a DC blocking circuit, and a matching circuit based on distributed parameters, which are respectively used to achieve the termination of the characteristic impedance and the target source impedance (Z of the control signal amplifier power transistorS,OPT1 ) and the impedance transformation of the power amplifier BA power tube target source impedance (Z S,OPT2 ) on the working frequency band. The stabilization circuit is used to increase the circuit stability within the target frequency band. The DC bias circuit is used to provide the gate bias voltage. The DC blocking circuit refers to the DC blocking capacitor on the main transmission path, allowing the RF signal to pass through normally and blocking the gate DC current;

[0012] The output matching network of the control signal amplifier is a hybrid circuit formed by a DC bias circuit, a DC blocking circuit, and a distributed-parameter-based matching circuit, which are respectively used to achieve the impedance transformation between the terminating characteristic impedance of the coupler (Z0 = 30) and the target load impedance (Z L,OPT1 ) on a specific frequency band. The DC bias circuit is used to provide the drain bias voltage;

[0013] The dual-transmission-line bandpass matching circuit of the power amplifier BA is used to achieve the impedance transformation between the terminating characteristic impedance of the coupler (Z0 = 30) and the target load impedance (Z L,OPT2 ) on a specific frequency band;

[0014] Furthermore, the dual-transmission-line bandpass matching circuit is composed of ten sections of transmission lines, namely, a pad, two half-wavelength stub transmission lines TL1 and TL2 with the same characteristic impedance connected in parallel at the same node, a group of dual transmission lines TL3 and TL4 located on the transmission path and composed of two transmission lines with the same characteristic impedance and electrical lengths of 60° and 120° respectively connected in parallel, a stepped stub transmission line composed of two quarter-wavelength transmission lines TL5 and TL6 with different characteristic impedances connected in series and connected in parallel after TL3 and TL4, a group of dual transmission lines TL7 and TL8 located on the transmission path and composed of two transmission lines with the same characteristic impedance and electrical lengths of 60° and 120° respectively connected in parallel, and a quarter-wavelength stub transmission line TL9 connected to the ground in parallel; one end of the pad is connected to the output end of the power amplifier BA power tube, and the other end is connected to one end of the dual transmission line composed of TL3 and TL4 connected in parallel and two symmetrically distributed half-wavelength stub transmission lines TL1 and TL2. The other end of the dual transmission line composed of TL3 and TL4 connected in parallel is connected to one end of the dual transmission line composed of TL7 and TL8 connected in parallel and a stepped stub transmission line composed of two quarter-wavelength transmission lines TL5 and TL6 with different characteristic impedances connected in series. The other end of the dual transmission line composed of TL7 and TL8 connected in parallel is connected to a quarter-wavelength stub transmission line TL9 connected to the ground and the input port of the coupler (Z0 = 30); through this structure, the impedance transformation of the working frequency band can be achieved, and at the same time, transmission zeros are inserted in the second and third harmonic frequency bands to achieve the effect of suppressing the second and third harmonics.

[0015] Furthermore, the functions of microstrip lines TL3 and TL4, TL7 and TL8 in the dual - transmission - line band - pass matching circuit are to insert two transmission zeros at the second harmonic 2f0 and the third harmonic 3f0 of the center frequency f0 of the working frequency band.

[0016] Furthermore, the functions of microstrip lines TL5 and TL6 in the dual - transmission - line band - pass matching circuit are to insert two symmetric transmission zeros on both sides of the second harmonic 2f0.

[0017] Furthermore, transmission lines TL1, TL2, and TL9 in the dual - transmission - line band - pass matching circuit can insert transmission zeros near the pass - band frequency band, retaining the out - of - band rejection effect of the dual - transmission - line band - pass matching network. TL9 is a quarter - wavelength stub transmission line grounded, which can also serve as a DC bias circuit to provide the drain bias voltage.

[0018] The beneficial effects of the present invention are as follows:

[0019] The innovation of the present invention lies in using a band - pass matching circuit based on a dual - transmission line. In the target design frequency band, this structure can achieve impedance transformation of the working frequency band, and at the same time insert transmission zeros in the second - and third - harmonic frequency bands to achieve the effect of suppressing the second and third harmonics.

[0020] The broadband load - modulated balanced amplifier designed by the present invention can achieve a back - off range greater than or equal to 7.5 dB in the frequency range of 2.3 - 3.6 GHz. When backing off, the drain efficiency is 47.5% - 55.8% (when the output power is 38 dBm), the drain efficiency at saturation is 61.2% - 73%, the gain at saturation is greater than 10.5 dB, and the output power at saturation is 45.5 dBm - 46.2 dBm. Description of the Drawings

[0021] Figure 1 It is a schematic diagram of a broadband load - modulated balanced amplifier with harmonic suppression function according to the present invention.

[0022] Figure 2 It is a schematic diagram of the band - pass matching circuit (ideal microstrip line).

[0023] Figure 3 It is a schematic diagram of the S - parameter simulation results of the band - pass matching circuit (ideal microstrip line) in the frequency range of 1 - 11 GHz.

[0024] Figure 4 It is a schematic diagram of the dual - transmission - line band - pass matching circuit (ideal microstrip line).

[0025] Figure 5 It is a schematic diagram of the S - parameter simulation results of the dual - transmission - line band - pass matching circuit (ideal microstrip line) in the frequency range of 1 - 11 GHz.

[0026] Figure 6 It is a schematic diagram of the actual microstrip line circuit of the dual - transmission - line band - pass matching network applied to the broadband load - modulation balanced power amplifier embodiment with a working frequency band of 2.3 GHz - 3.6 GHz.

[0027] Figure 7 It is a schematic diagram of the actual microstrip line circuit of the input matching network of the BA power amplifier applied to the broadband load - modulation balanced power amplifier embodiment with a working frequency band of 2.3 GHz - 3.6 GHz.

[0028] Figure 8 It is a schematic diagram of the actual microstrip line circuit of the input matching network of the control - signal amplifier applied to the broadband load - modulation balanced power amplifier embodiment with a working frequency band of 2.3 GHz - 3.6 GHz.

[0029] Figure 9 It is a schematic diagram of the actual microstrip line circuit of the output matching network of the control - signal amplifier applied to the broadband load - modulation balanced power amplifier embodiment with a working frequency band of 2.3 GHz - 3.6 GHz.

[0030] Figure 10 It is a schematic diagram of the actual microstrip line circuit of the post - matching network applied to the broadband load - modulation balanced power amplifier embodiment with a working frequency band of 2.3 GHz - 3.6 GHz.

[0031] Figure 11 It is a schematic diagram of the S - parameter simulation results of the broadband load - modulation balanced power amplifier embodiment of the present invention in the frequency range of 1 - 11 GHz.

[0032] Figure 12 It is a schematic diagram of the simulation results of the drain - efficiency vs. output - power curves at each frequency point in the frequency range of 2.3 GHz - 3.6 GHz for the broadband load - modulation balanced power amplifier embodiment of the present invention.

[0033] Figure 13 It is a schematic diagram of the simulation results of the gain vs. output - power curves at each frequency point in the frequency range of 2.3 GHz - 3.6 GHz for the broadband load - modulation balanced power amplifier embodiment of the present invention.

[0034] Figure 14 It is a schematic diagram of the dual - transmission - line band - pass matching circuit. The input impedance of the matching circuit shown on the left side of the figure represents the target load impedance (ZL , OPT), and the terminated load (Z0) of the matching circuit shown on the right side. Detailed implementation manners

[0035] See Figure 1, the structural composition of the main circuit of the embodiments of the present invention is introduced, including an input port Port1, an output port Port2, a power divider (01), a phase shift line (05), a control signal amplifier, a balanced amplifier, and a post-matching network (11); the input port Port1 is connected to the input end of the power divider (01), the output ends of the power divider (01) are respectively connected to the control signal amplifier and the phase shift line (05), the phase shift line (05) is connected to the input port of the input coupler (06) of the balanced amplifier, the control signal amplifier is connected to the isolation port of the output coupler (10) of the balanced amplifier, the output port of the output coupler (10) of the balanced amplifier is connected to the post-matching network (11), and the post-matching network (11) is connected to the output port Port2; the control signal amplifier includes a control signal amplifier input matching network (02), a control signal amplifier power transistor (03), and a control signal amplifier output matching network (04) connected in sequence; the balanced amplifier includes an input coupler (06) connected in sequence, the isolation end of the input coupler (06) is connected to a 50Ω resistor, and the through and coupled output ends of the input coupler (06) are connected to the same power amplifier BA and output coupler (10); the power amplifier BA includes a power amplifier BA input matching network (07), a power amplifier BA power transistor (08), and a power amplifier BA dual-transmission-line bandpass matching network (09) connected in sequence;

[0036] It should be noted that the working bandwidths of the power divider (01), the post-matching network (11), the input coupler (06), and the output coupler (10) can all cover the working frequency band of the broadband load modulation balanced amplifier; the power divider (01) is a non-equal power divider; the input coupler (06) and the output coupler (10) are broadband branch-line couplers, the terminating load of the input coupler (06) is 50 ohms, and the terminating load of the output coupler (10) is 30 ohms; the post-matching network (11) is an impedance conversion circuit that realizes the impedance conversion from 30 ohms to 50 ohms;

[0037] It should be noted that the control signal amplifier input matching network (02) and the power amplifier BA input matching network (07) are hybrid circuits formed by including a stability circuit, a DC bias circuit, a DC blocking circuit, and a distributed-parameter-based matching circuit, which are respectively used to realize the impedance transformation between the terminating characteristic impedance and the target source impedance (Z S,OPT1 ) of the control signal amplifier power transistor and the target source impedance (Z S,OPT2 ) of the power amplifier BA power transistor on the working frequency band. The stability circuit is to increase the circuit stability within the target frequency band, the DC bias circuit is used to provide the gate bias voltage, and the DC blocking circuit refers to the DC blocking capacitor on the main transmission path to allow the RF signal to pass normally and block the gate DC current;

[0038] It should be noted that the output matching network (04) of the control signal amplifier is a hybrid circuit including a DC bias circuit, a DC blocking circuit, and a matching circuit based on distributed parameters, which are respectively used to implement the impedance transformation between the termination characteristic impedance of the output coupler (10) and the target load impedance (Z L,OPT1 ) in a specific frequency band. The DC bias circuit is used to provide the drain bias voltage;

[0039] It should be noted that the dual - transmission - line band - pass matching network (09) of the power amplifier BA is used to implement the impedance transformation between the termination characteristic impedance of the output coupler (10) and the target load impedance (Z L,OPT2 ) in a specific frequency band;

[0040] Generally speaking, a hybrid circuit formed by a lumped - parameter - based bias circuit, an RC stabilization circuit, a DC blocking circuit, a distributed - parameter - based matching circuit, a power divider, and a coupler realizes the broadband load - modulation balanced amplifier with harmonic suppression function described in this patent.

[0041] In this embodiment, the selected dielectric board is Rogers 5880, with a dielectric constant of 2.2 and a board thickness of 20 mil. The simulation software is Advanced Design System. The power transistors all use CG2H40010F produced by CREE. The target frequency band is selected as 2.3 - 3.6 GHz, and the center frequency f0 is 2.95 GHz. The band - pass matching circuit of the power amplifier BA based on dual - transmission lines is the core part of the power amplifier of the present invention and will be described in detail. Other parts, including the power divider, coupler, bias circuit, stabilization circuit, input matching circuit, post - matching network, and control signal amplifier output matching circuit, are designed according to the well - known methods in the art. To avoid confusing the concept of the present invention, the description of well - known structures and technologies is omitted in this article.

[0042] Step 1: In the target frequency band, obtain the target source impedance Z S,OPT and the target load impedance Z T,OPT required for the BA power transistor in the balanced amplifier through source - pulling and load - pulling techniques.

[0043] Step 2: Considering obtaining the target load impedance Z T,OPTThe variation ranges of the real and imaginary parts. According to the published article "Implementation of Flat Gain Broadband Power Amplifier With Impedance Rotation Compensation", we select R = 9Ω, L = 0.71nH, C = 0.43pf to design the band-pass matching network. The band-pass matching structure and the characteristic impedance values of each microstrip line obtained through formula calculation and optimization are shown in Figure 2 . The band-pass matching network consists of two quarter-wavelength transmission lines on the main transmission path and three half-wavelength stub transmission lines. The characteristic impedance of the first half-wavelength stub transmission line is Z1 = 7.2Ω, the characteristic impedance of the second quarter-wavelength transmission line on the main transmission path is Z2 = 16.5Ω, the characteristic impedance of the third half-wavelength stub transmission line is Z3 = 77.4Ω, the characteristic impedance of the fourth quarter-wavelength transmission line on the main transmission path is Z4 = 29.1Ω, and the characteristic impedance of the fifth half-wavelength stub transmission line is Z5 = 62.8Ω. The S-parameter simulation results of the band-pass matching circuit (ideal microstrip line) in the frequency range of 1 - 11GHz are shown in Figure 3 .

[0044] Step 3: Use the formula (Z is the characteristic impedance value of the quarter-wavelength transmission line on the main transmission path of the band-pass matching network, and Z” is the characteristic impedance value of the dual transmission line after replacement), convert the quarter-wavelength transmission line on the main transmission path of the band-pass matching into a dual transmission line, and insert transmission zeros at 2f0 and 3f0; use the formula (fZ is the frequency point where the transmission zero is inserted to the left of 2f0, Z is the characteristic impedance value of the half-wavelength stub transmission line between two groups of dual transmission lines, Z ia is the characteristic impedance value of the first quarter-wavelength stub transmission line of the stepped stub transmission line after replacement, Z ib is the characteristic impedance value of the second quarter-wavelength stub transmission line of the stepped stub transmission line after replacement), convert the half-wavelength stub transmission line between two groups of dual transmission lines into a stepped stub transmission line, and insert symmetric transmission zeros on both sides of the second harmonic 2f0 frequency point. At the same time, for the convenience of circuit power-on, use the formula (where w is the relative bandwidth, Z is the characteristic impedance value of the half-wavelength stub transmission line where the band-pass matching network is connected to the output end, and Z” is the characteristic impedance value of the quarter-wavelength stub transmission line grounded after replacement), convert the half-wavelength stub transmission line where the band-pass matching network is connected to the output end into a quarter-wavelength stub transmission line grounded. Since the structure of the band-pass matching network has changed, the fundamental wave impedance trajectory within the target frequency band will also change. Therefore, we need to optimize the band-pass matching network after replacement to ensure that the fundamental wave impedance trajectory is as close as possible to that of the band-pass network, thereby ensuring the performance of the power amplifier. For the schematic diagram of the circuit structure of the optimized band-pass matching circuit based on dual transmission lines (ideal microstrip lines), see Figure 4 . The characteristic impedance of microstrip lines TL1 and TL2 is Z1’ = 14.4 Ω, and the electrical length is 180°. The characteristic impedances of microstrip lines TL5 and TL6 are Z 3a ’ = 91 Ω and Z 3b ’ = 13.9 Ω respectively, and the electrical length of both is 90°. The characteristic impedance of microstrip lines TL7 and TL8 is Z4’ = 67.2 Ω, and the electrical lengths are 60° and 120° respectively. The characteristic impedance of microstrip line TL9 is Z5’ = 98 Ω, and the electrical length is 90°. For the S-parameter simulation results of the band-pass matching circuit based on dual transmission lines (ideal microstrip lines) in the frequency range of 1 - 11 GHz, see Figure 5 .

[0045] Step 4: To obtain the band-pass matching circuit based on dual transmission lines, we need to add pads to the circuit and match the band-pass matching circuit based on dual transmission lines to the target load impedance ZT , OPT. Due to the introduction of capacitors and considering the simulation results of the power amplifier, a large amount of simulation work is required for the circuit and the circuit size needs to be iteratively optimized. Therefore, there will be a certain difference between the actual physical size obtained and the initial circuit value calculated based on theory. For the schematic diagram of the actual microstrip line circuit of the band-pass matching network based on dual transmission lines applied to the broadband load modulation balanced power amplifier with a working frequency band of 2.3 GHz - 3.6 GHz, see Figure 6。The line width of microstrip line TLM1 is 1.4 mm and the line length is 2.6 mm. The line width of microstrip line TLM2 is 7.9 mm and the line length is 36.6 mm. The line width of microstrip line TLM3 is 7.9 mm and the line length is 34.8 mm. The line width of microstrip line TLM4 is 2.5 mm and the line length is 12.4 mm. The line width of microstrip line TLM5 is 2.5 mm and the line length is 24 mm. The line width of microstrip line TLM6 is 0.4 mm and the line length is 18.8 mm. The line width of microstrip line TLM7 is 3.5 mm and the line length is 17.8 mm. The line width of microstrip line TLM8 is 2.5 mm and the line length is 12.4 mm. The line width of microstrip line TLM9 is 2.5 mm and the line length is 24 mm. The line width of microstrip line TLM10 is 0.4 mm and the line length is 17.2 mm. The line width of microstrip line TLM11 is 3.1 mm and the line length is 0.2 mm. CM1 = 9 pf and CM2 = 5.6 pf. As the output matching structure of a broadband power amplifier, the band-pass matching circuit based on dual transmission lines can achieve matching from the coupler terminated with a load of 30 Ω to the optimal impedance region at the output end of the balanced amplifier transistor.

[0046] Step Five: Add other circuit structures to verify the effectiveness of the design. The schematic diagram of the actual microstrip line circuit of the input matching network of the BA power amplifier applied to the broadband load modulation balanced power amplifier embodiment with a working frequency band of 2.3 GHz - 3.6 GHz is shown in Figure 7 。The line width of microstrip line TLN1 is 1.5 mm and the line length is 2 mm. The line width of microstrip line TLN2 is 1.6 mm and the line length is 5.5 mm. The line width of microstrip line TLN3 is 2 mm and the line length is 20.8 mm. The line width of microstrip line TLN4 is 3.5 mm and the line length is 36 mm. The line width of microstrip line TLN5 is 4.8 mm and the line length is 19.3 mm. The line width of microstrip line TLN6 is 19.5 mm and the line length is 35.4 mm. The line width of microstrip line TLN7 is 5 mm and the line length is 1.5 mm. The line width of microstrip line TLN8 is 0.5 mm and the line length is 8.8 mm. CN1 = 5.6 pf, CN2 = 9 pf, CN3 = 2.4 pf, RM1 = 1 Ω, RM2 = 100 Ω. The schematic diagram of the actual microstrip line circuit of the input matching network of the control signal amplifier applied to the broadband load modulation balanced power amplifier embodiment with a working frequency band of 2.3 GHz - 3.6 GHz is shown in Figure 8。The line width of microstrip line TLZ1 is 1.5 mm and the line length is 1 mm. The line width of microstrip line TLZ2 is 2.3 mm and the line length is 18.3 mm. The line width of microstrip line TLZ3 is 4.2 mm and the line length is 14.1 mm. The line width of microstrip line TLZ4 is 3.2 mm and the line length is 1.8 mm. The line width of microstrip line TLZ5 is 15.9 mm and the line length is 3.1 mm. The line width of microstrip line TLZ6 is 3.6 mm and the line length is 5.5 mm. The line width of microstrip line TLZ7 is 18 mm and the line length is 11.4 mm. The line width of microstrip line TLZ8 is 6 mm and the line length is 2 mm. The line width of microstrip line TLZ9 is 26.6 mm and the line length is 7.8 mm. The line width of microstrip line TLZ10 is 0.5 mm and the line length is 16 mm. CZ1 = 5.6 pf, CZ2 = 3.3 pf, CZ3 = 9 pf, RM1 = 47 Ω, RM2 = 30 Ω. For the output matching network of the control signal amplifier applied to the actual microstrip line circuit diagram of the broadband load modulation balanced power amplifier embodiment with a working frequency band of 2.3 GHz - 3.6 GHz, see Figure 9 。The line width of microstrip line TLX1 is 1.4 mm and the line length is 1.8 mm. The line width of microstrip line TLX2 is 12.1 mm and the line length is 35.4 mm. The line width of microstrip line TLX3 is 5.7 mm and the line length is 15.5 mm. The line width of microstrip line TLX4 is 4.2 mm and the line length is 11.8 mm. The line width of microstrip line TLX5 is 3.2 mm and the line length is 21 mm. The line width of microstrip line TLX6 is 2.6 mm and the line length is 1.9 mm. The line width of microstrip line TLX7 is 3.1 mm and the line length is 0.2 mm. The line width of microstrip line TLX8 is 0.4 mm and the line length is 17.2 mm. CX1 = 9 pf, CX2 = 5.6 pf. For the post-matching network applied to the actual microstrip line circuit diagram of the broadband load modulation balanced power amplifier embodiment with a working frequency band of 2.3 GHz - 3.6 GHz, see Figure 10 。The line width of microstrip line TLP1 is 2.3 mm and the line length is 8 mm. The line width of microstrip line TLP2 is 2.8 mm and the line length is 8 mm. The line width of microstrip line TLP3 is 1.6 mm and the line length is 8 mm. The line width of microstrip line TLP4 is 2 mm and the line length is 8 mm. The line width of microstrip line TLP5 is 1.5 mm and the line length is 5 mm. Among them, the phase shift line is a microstrip line with a width of 1.5 mm and a length of 152.2 mm.

[0047] See Figure 3 and Figure 5 。Comparing the simulation results of the S parameters, it can be seen that 1) in the target frequency band of 2.3 GHz - 3.6 GHz, the S of the band-pass matching network 11 <-30 dB, and the S of the band-pass matching circuit based on the dual transmission line 11 <-15 dB; 2) Through Figure 3It can be found that in the second harmonic and third harmonic frequency bands, the transmission coefficient S of the band-pass matching network 21 does not decay rapidly, which means that the harmonics are not suppressed and spurious harmonics still exist. Based on the dual transmission line band-pass matching circuit, in the second and third harmonic frequency bands, two sets of dual transmission lines introduce transmission zeros fTZ1 and fTZ2, and the stepped stub transmission line introduces transmission zeros fTZ3 and fTZ4. Due to the insertion of transmission zeros, the transmission coefficient S in the second and third harmonic frequency bands 21 decays rapidly, which means that the harmonic impedance is closer to the edge of the Smith chart, achieving the purpose of suppressing harmonics. Figure 11 The figure shows the simulation results of the S-parameters of the broadband load modulation balanced amplifier in the frequency range of 1-11 GHz. It can be seen that in the second harmonic and third harmonic frequency bands, except for individual frequency points, most of the S in the second harmonic and third harmonic frequency bands 21 is rapidly compressed, achieving the effect of suppressing harmonics. It should be noted that due to the optimization of the circuit structure, the problem of frequency offset of the inserted transmission zeros inevitably occurs, but the effect of suppressing harmonics remains. Figure 12 and Figure 13 are respectively the schematic diagrams of the simulation results of the drain efficiency and gain varying with the output power at each frequency point in the frequency range of 2.3 GHz - 3.6 GHz for the embodiment of the broadband load modulation balanced power amplifier of the present invention. From the simulation results, it can be seen that in the operating frequency band range of 2.3 GHz - 3.6 GHz, when the output power is 38 dBm, the drain efficiency is 47.5% - 55.8%, the back-off range is 7.5 - 8 dB, the drain efficiency at saturation is 61.2% - 73%, the gain at saturation is greater than 10.5 dB, and the output power at saturation is 45.5 dBm - 46.2 dBm.

[0048] The embodiment of the present invention verifies the design method of the broadband load modulation balanced amplifier with the function of harmonic suppression proposed by the present invention, achieving the invention purpose. In the embodiment of the present invention, S 21 decays rapidly in the second and third harmonic frequency bands, achieving the purpose of suppressing harmonics. And the embodiment of the present invention realizes a working bandwidth of 1.3 GHz in the frequency range of 2.3 - 3.6 GHz, having an obvious advantage in bandwidth, with a back-off range of 7.5 - 8 dB, meeting the requirements of modern communication systems for the linearity of power amplifiers.

[0049] It should be noted that the above specific embodiments of the present invention are only for illustrative purposes and do not constitute a limitation to the present invention. Therefore, any modifications, equivalent replacements, improvements, etc. made without departing from the spirit and scope of the present invention shall be included within the protection scope of the present invention.

Claims

1. A broadband load modulation balanced amplifier with harmonic suppression function, characterized in that , including an input port Port1, an output port Port2, a power divider (01), a phase offset line (05), a control signal amplifier, a balanced amplifier, and a post-matching network (11); the input port Port1 is connected to the input end of the power divider (01), the output ends of the power divider (01) are respectively connected to the control signal amplifier and the phase offset line (05), the phase offset line (05) is connected to the input port of the balanced amplifier coupler (6), the control signal amplifier is connected to the isolation port of the balanced amplifier output coupler (10), the output port of the balanced amplifier output coupler (10) is connected to the post-matching network (11), and the post-matching network (11) is connected to the output port Port2; the control signal amplifier includes a control signal amplifier input matching network (02), a control signal amplifier power transistor (03), and a control signal amplifier output matching network (04) connected in sequence; the balanced amplifier includes an input coupler (06) connected in sequence, the isolation end of the input coupler (06) is connected to a 50Ω resistor, and the through and coupled output ends of the input coupler (06) are connected to the same power amplifier BA and the output coupler (10); the power amplifier BA includes a power amplifier BA input matching network (07), a power amplifier BA power transistor (08), and a power amplifier BA dual-transmission-line bandpass matching network (09) connected in sequence; The working bandwidths of the power divider (01), the post-matching network (11), the input coupler (06), and the output coupler (10) can all cover the working frequency band of the broadband load modulation balanced amplifier; the power divider (01) is a non-equal power divider; the input coupler (06) and the output coupler (10) are broadband branch-line couplers, the terminating load of the input coupler (06) is 50 ohms, and the terminating load of the output coupler (10) is 30 ohms; the post-matching network (11) is an impedance conversion circuit that realizes the impedance conversion from 30 ohms to 50 ohms; The input matching network (02) of the control signal amplifier and the input matching network (07) of the power amplifier BA are hybrid circuits formed by including a stability circuit, a DC bias circuit, a DC blocking circuit, and a distributed-parameter-based matching circuit, which are respectively used to achieve impedance transformation between the terminated characteristic impedance and the target source impedance (Z S,OPT1 ) of the power transistor of the control signal amplifier and the target source impedance (Z S,OPT2 ) of the power transistor of the power amplifier BA within the operating frequency band. The stability circuit is for increasing circuit stability within the target frequency band. The DC bias circuit is used to provide the gate bias voltage. The DC blocking circuit refers to the DC blocking capacitor on the main transmission path, allowing the RF signal to pass normally and blocking the gate DC current; The output matching network (04) of the control signal amplifier is a hybrid circuit formed by a DC bias circuit, a DC blocking circuit, and a distributed-parameter-based matching circuit, which are respectively used to achieve impedance transformation between the termination characteristic impedance of the output coupler (10) and the target load impedance (Z L,OPT1 ) in a specific frequency band. The DC bias circuit is used to provide a drain bias voltage; The dual - transmission - line band - pass matching network (09) of the power amplifier BA is used to achieve impedance transformation between the termination characteristic impedance of the output coupler (10) and the target load impedance (Z L,OPT2 ) in a specific frequency band; The harmonic suppression function means that for the designed broadband load modulation balanced amplifier, within the second and third harmonic frequency bands, S 21 rapidly decays to achieve the effect of harmonic suppression.

2. The broadband load modulation balanced amplifier with harmonic suppression function according to claim 1, wherein, The described dual - transmission - line band - pass matching network (09) is composed of ten sections of transmission lines, namely, a pad, two half - wavelength stub transmission lines TL1 and TL2 with the same characteristic impedance in parallel at the same node, a pair of dual - transmission lines TL3 and TL4 composed of two transmission lines with the same characteristic impedance and electrical lengths of 60° and 120° respectively in parallel on the transmission path, a stepped - stub transmission line composed of two quarter - wavelength transmission lines TL5 and TL6 with different characteristic impedances in series and in parallel after TL3 and TL4, a pair of dual - transmission lines TL7 and TL8 composed of two transmission lines with the same characteristic impedance and electrical lengths of 60° and 120° respectively in parallel on the transmission path, and a quarter - wavelength stub transmission line TL9 in parallel to the ground; one end of the pad is connected to the output end of the power amplifier BA power transistor (08), and the other end is connected to one end of the dual - transmission line composed of TL3 and TL4 in parallel and two symmetrically distributed half - wavelength stub transmission lines TL1 and TL2. The other end of the dual - transmission line composed of TL3 and TL4 in parallel is connected to one end of the dual - transmission line composed of TL7 and TL8 in parallel and a stepped - stub transmission line composed of two quarter - wavelength transmission lines TL5 and TL6 with the same characteristic impedance in series. The other end of the dual - transmission line composed of TL7 and TL8 in parallel is connected to a quarter - wavelength stub transmission line TL9 grounded and the input port of the output coupler (10); through this structure, impedance transformation in the working frequency band can be achieved, and at the same time, transmission zeros are inserted in the second - and third - harmonic frequency bands to achieve the effect of suppressing the second and third harmonics.

3. The broadband load modulation balanced amplifier with harmonic suppression function according to claim 1, wherein In the described dual - transmission - line band - pass matching network (09), the functions of the microstrip lines TL3 and TL4, TL7 and TL8 are to insert two transmission zeros at the second - harmonic 2f0 and third - harmonic 3f0 of the center frequency point f0 of the working frequency band.

4. A broadband load modulation balanced amplifier with harmonic suppression function according to claim 1, characterized in that, In the described dual - transmission - line band - pass matching network (09), the functions of the microstrip lines TL5 and TL6 are to insert two symmetric transmission zeros on both sides of the second - harmonic 2f0.

5. A broadband load modulation balanced amplifier with harmonic suppression function according to claim 1, characterized in that, In the described dual - transmission - line band - pass matching network (09), the transmission lines TL1, TL2, and TL9 can insert transmission zeros near the pass - band frequency band, retaining the out - of - band suppression effect of the dual - transmission - line band - pass matching network. TL9 is a quarter - wavelength stub transmission line in parallel to the ground and can also be used as a DC bias circuit to provide the drain bias voltage.

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