Online temperature self-compensation calibration method and circuit for force measurement transmitter

Abstract

The invention relates to the technical field of field data acquisition, in particular to an on-line temperature self-compensation calibration method and circuit for a force measurement transmitter. The calibration method comprises the steps that a working strain gauge and a temperature compensation gauge are synchronously excited through a two-way precision constant current source, the working strain gauge and the temperature compensation gauge are consistent in material parameter and are located in the same temperature field, and output current values of the two-way precision constant current source are equal and are controlled by a high-precision reference voltage source and a sampling resistor in a closed-loop mode. According to the invention, differential signals of the working strain gauge and the temperature compensation gauge are decoupled in real time through double-path precise constant-current source excitation and synchronous ADC acquisition technologies, gradient compensation is carried out on the temperature change rate in combination with a dynamic weighting algorithm, a lag error of traditional fixed coefficient compensation in a transient temperature field is effectively eliminated, and the accuracy of temperature compensation is improved. And online self-compensation of the temperature drift of the strain gauge is realized.

Description

Technical Field

[0001] The invention relates to the technical field of field data acquisition, in particular to an online temperature self-compensation calibration method and circuit for a force measurement transmitter. Background Art

[0002] In the existing temperature compensation and calibration technology of force measuring transmitters, a composite compensation method based on manual adjustment of analog potentiometers and full temperature range verification is generally relied on. This method requires physical adjustment of the potentiometer to achieve zero point and full scale calibration, and the inherent temperature coefficient of the potentiometer will introduce additional temperature drift, resulting in the degradation of the long-term stability of the sensor signal conditioning circuit; at the same time, traditional temperature compensation requires repeated verification of bridge balance parameters in a wide temperature range. Its calibration process is time-consuming and lacks dynamic adaptability, and it cannot effectively cope with the non-uniform changes in temperature fields or transient thermal shocks in industrial sites. Especially in high-precision measurement and control scenarios, the nonlinear resistance drift of strain gauges caused by temperature gradients and the coupling error of analog transmitter signal calibration often lead to a significant decrease in compensation accuracy and calibration efficiency. Although the existing technology attempts to improve temperature drift through hardware redundancy or complex algorithms, it is limited by the fixed compensation coefficient of full temperature verification and the hysteresis of manual calibration, and it is difficult to achieve the integrated coordination of real-time self-compensation and rapid calibration in a dynamic temperature field. Summary of the invention

[0003] The present disclosure provides an online temperature self-compensation calibration method and circuit for a force measuring transmitter, aiming to overcome at least one defect existing in the prior art.

[0004] To achieve the above purpose, the technical solution disclosed in the present invention is as follows:

[0005] According to one aspect of the present disclosure, a method for online temperature self-compensation calibration of a force measuring transmitter is provided, the steps of the calibration method comprising:

[0006] The working strain gauge and the temperature compensation gauge are synchronously excited by a dual-channel precision constant current source, the material parameters of the working strain gauge and the temperature compensation gauge are consistent and are in the same temperature field, and the output current values ​​of the dual-channel constant current source are equal and are closed-loop controlled by a high-precision reference voltage source and a sampling resistor;

[0007] Synchronously collect the differential voltage signal of the working strain gauge and the differential voltage signal of the temperature compensation sheet, perform synchronous analog-to-digital conversion through the 24-bit ADC integrated in the embedded MCU and the 24-bit ADC of the external signal conditioning chip, and obtain the first conversion result AD_Result_work and the second conversion result AD_Result_temp;

[0008] Based on the second conversion result AD_Result_temp, performing real-time temperature compensation on the first conversion result AD_Result_work by a dynamic weighting algorithm to generate a compensated output value AD_Compensated, wherein the dynamic weighting algorithm adjusts the temperature influence coefficient according to a gradient change of the second conversion result AD_Result_temp;

[0009] The embedded MCU controls the 16-bit DAC to output the target voltage to the V / V conversion circuit and the V / I conversion circuit to generate isolated voltage transmission signals and current transmission signals;

[0010] In response to the zero-point calibration instruction or the full-scale calibration instruction, the voltage transmission signal or the current transmission signal is fed back to the ADC inside the MCU in real time for sampling, and the DAC output value is dynamically adjusted through the incremental PID algorithm until the error between the feedback signal and the preset target value is less than 0.02% FS, completing the online calibration.

[0011] Furthermore, the dynamic weighting algorithm includes:

[0012] According to the current value of the second conversion result AD_Result_temp and the historical sampling sequence, the temperature change rate ΔT / Δt is calculated. When ΔT / Δt exceeds the threshold, AD_Result_work is compensated using the following formula:

[0013] AD_Compensated=AD_Result_work-[α·(AD_Result_temp-AD_Result_temp_initial)+β·ΔT / Δt], where α is the temperature sensitivity matching coefficient between the working strain gauge and the temperature compensation gauge, β is the temperature change rate compensation coefficient, AD_Result_temp_initial is the conversion result of the temperature compensation gauge at the initial calibration temperature, and α and β are determined by least squares fitting in the calibration experiment.

[0014] Furthermore, the zero point calibration instruction triggering method includes:

[0015] After selecting the voltage or current transmission signal type through the dip switch, press the zero calibration button, the MCU controls the DAC to output the zero target voltage, and collects the feedback signal in real time; when the deviation between the feedback signal and the target value exceeds the dead zone range, the PID algorithm is quickly adjusted with the proportional coefficient Kp=3.0 and the integral time Ti=5ms; when the deviation enters the dead zone range, it switches to Kp=0.5 and Ti=20ms to suppress overshoot until the steady-state error is less than 0.02%FS.

[0016] Furthermore, the incremental PID algorithm includes:

[0017] The output increment Δu(k) of the PID controller is determined by the following formula:

[0018] Δu(k)=Kp·[e(k)-e(k-1)]+Ki·e(k)+Kd·[e(k)-2e(k-1)+e(k-2)], wherein e(k) is the error at the kth sampling moment, Ki=Kp·T / Ti, Kd=Kp·Td / T, and T is the sampling period; the Kp, Ti, and Td are dynamically adjusted according to the absolute value of the error |e(k)|: when |e(k)|>1%FS, Kp=4.0, Ti=2ms, and Td=0.5ms; when |e(k)|≤0.5%FS, Kp=0.8, Ti=10ms, and Td=2ms.

[0019] Furthermore, the step of dynamically adjusting the output current value of the dual-channel constant current source includes real-time monitoring of the differential voltage signal of the temperature compensation sheet. When it is detected that its value exceeds a preset range, the MCU adjusts the reference voltage drive value of the constant current source through the SPI bus to make the differential voltage of the temperature compensation sheet return to the calibration range, and simultaneously corrects the excitation current of the working strain gauge to maintain measurement consistency.

[0020] Furthermore, the output switching of the voltage transmission signal and the current transmission signal is achieved by the following steps:

[0021] The upper two digits of the DIP switch select the current output type, and the lower two digits select the voltage output type;

[0022] When two output types are selected at the same time, the MCU controls the analog switch to route the DAC output signal to the V / V conversion circuit and the V / I conversion circuit respectively, and the voltage follower of the V / I conversion circuit isolates the two signals to prevent mutual interference.

[0023] According to another aspect of the present disclosure, there is provided an online temperature self-compensation circuit for a force measuring transmitter, which is used to implement the above-mentioned online temperature self-compensation calibration method for the force measuring transmitter, and the online temperature self-compensation circuit comprises:

[0024] Dual-channel precision constant current source module, which consists of a high-precision reference voltage source, an operational amplifier and a sampling resistor, and is used to provide equal excitation current to the working strain gauge and the temperature compensation sheet;

[0025] A signal acquisition module, including an embedded MCU and an external signal conditioning chip, wherein the MCU has an internal integrated 24-bit ADC for acquiring the differential voltage signal of the working strain gauge, and the 24-bit ADC of the external signal conditioning chip is used to synchronously acquire the differential voltage signal of the temperature compensation sheet;

[0026] The transmission output module includes a 16-bit DAC, a V / V conversion circuit and a V / I conversion circuit. The output end of the DAC is connected to a voltage follower composed of an operational amplifier, and a voltage transmission signal is generated through the V / V conversion circuit and a current transmission signal is generated through a mirror current source circuit.

[0027] A feedback calibration module, including a multi-channel analog switch and a voltage-dividing sampling circuit, is used to feed back a voltage transmission signal or a current transmission signal to an ADC inside the MCU;

[0028] The power module, which consists of an LDO regulator, a π-type filter circuit and a transient suppression diode, is used to provide isolated power for the constant current source, MCU and the transmitter output module.

[0029] Furthermore, in the dual-channel precision constant current source module, each constant current source includes a reference voltage source, whose output end is connected to the non-inverting input end of the operational amplifier; the inverting input end of the operational amplifier is connected to one end of the sampling resistor, and the output end drives the base of the bipolar transistor; the emitter of the bipolar transistor is connected to the other end of the sampling resistor and serves as the constant current output end, and the collector is connected to the working strain gauge or the temperature compensation sheet.

[0030] Furthermore, the V / I conversion circuit comprises:

[0031] A voltage follower, wherein the input end is connected to the output end of the DAC, and the output end is connected to a mirror current source composed of an operational amplifier, a bipolar transistor and a sampling resistor;

[0032] The two ends of the sampling resistor are connected to a differential amplifier, and the output end thereof is fed back to the ADC inside the MCU after being switched by an analog switch;

[0033] The multi-way analog switch in the feedback calibration module is a single-pole double-throw type, with its common end connected to the current transmission signal sampling resistor, the first switching end connected to the current output path, and the second switching end connected to the input end of the differential amplifier. The output end of the differential amplifier is connected to the ADC inside the MCU through an RC filtering circuit.

[0034] Furthermore, in the power module, the analog power supply and the digital power supply are isolated by magnetic beads, and the analog ground is connected to the digital ground at a single point through a 0Ω resistor; the π-type filter circuit is composed of an inductor and a ceramic capacitor and an electrolytic capacitor in parallel, and its input end is connected in series with a self-recovery fuse and an anti-reverse diode;

[0035] The GPIO pin of the embedded MCU is connected to a 4-bit dip switch and a calibration button. The coding state of the dip switch is input to the input capture channel of the MCU through a pull-up resistor. The trigger signal of the calibration button is processed by the debouncing circuit and triggers the interrupt service program to execute the calibration process.

[0036] The beneficial effects of the present invention are:

[0037] The present invention uses dual-channel precision constant current source excitation and synchronous ADC acquisition technology to decouple the differential signals of the working strain gauge and the temperature compensation gauge in real time, and combines the dynamic weighting algorithm to perform gradient compensation for the temperature change rate, effectively eliminating the hysteresis error of the traditional fixed coefficient compensation in the transient temperature field, and realizing online self-compensation of the strain gauge temperature drift; through the closed-loop feedback control of the transmission signal by the embedded MCU, the incremental PID algorithm is used to replace the manual adjustment of the potentiometer, and the 0.02% FS-level steady-state accuracy is achieved in the zero point and full-scale calibration, which significantly improves the calibration efficiency and avoids the temperature drift of the potentiometer; in addition, the closed-loop dynamic adjustment mechanism of the dual constant current source and the isolated routing design of the analog transmission output further suppress the cross-interference between multiple signal paths, ensuring the long-term stability and dynamic response consistency of the transmitter in the full temperature range.

[0038] The above description is only an overview of the technical solution of the present invention. In order to more clearly understand the technical means of the present invention and implement it according to the contents of the specification, the following is a detailed description of the preferred embodiments of the present invention in conjunction with the accompanying drawings. BRIEF DESCRIPTION OF THE DRAWINGS

[0039] Figure 1 It is a flow chart of the online temperature self-compensation calibration method of the force measuring transmitter of the present invention;

[0040] Figure 2 It is a schematic diagram of the temperature online self-compensation principle of the present invention;

[0041] Figure 3 It is a schematic diagram of the principle of online self-calibration of analog transmission signals of the present invention;

[0042] Figure 4 It is a system hardware block diagram for realizing the functions of the present invention;

[0043] Figure 5 It is a system power supply circuit for realizing the functions of the present invention;

[0044] Figure 6 It is a circuit diagram of the dual constant current excitation working strain gauge and the temperature compensation gauge of the present invention;

[0045] Figure 7 It is the MCU minimum system and related peripheral circuit diagram of the present invention;

[0046] Figure 8 The invention relates to an analog transmission signal output and real-time acquisition circuit. DETAILED DESCRIPTION

[0047] The following will clearly and completely describe the technical solutions in the embodiments of the present invention with reference to the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are part of the embodiments of the present invention, rather than all of them. All other embodiments obtained by those of ordinary skill in the art based on the embodiments of the present invention without creative efforts shall fall within the protection scope of the present invention.

[0048] The term "including" and any of its variations in the specification and claims of this application are intended to cover non-exclusive inclusion. For example, a process, method, system, product, or device that includes a series of steps or units does not necessarily have to be limited to those steps or units clearly listed, but may include other steps or units not clearly listed or inherent to these processes, methods, products, or devices. In addition, the use of "and / or" in the specification and claims means at least one of the connected objects. For example, A and / or B means including three cases: A alone, B alone, and both A and B exist.

[0049] In the embodiments of the present invention, words such as "exemplary" or "for example" are used to give examples, illustrations, or explanations. Any embodiment or design solution described as "exemplary" or "for example" in the embodiments of the present invention should not be construed as being more preferred or having more advantages than other embodiments or design solutions. Rather, the use of words such as "exemplary" or "for example" is intended to present relevant concepts in a specific manner.

[0050] The present invention provides the following preferred embodiments:

[0051] Embodiment 1

[0052] To solve the problems of insufficient suppression of temperature drift caused by the hysteresis of fixed compensation parameters in a force-measuring transmitter in a dynamic temperature field and low efficiency of manual calibration, this embodiment further optimizes the cooperation mechanism of dual-channel constant current source excitation, synchronous signal acquisition, and dynamic weighted compensation. As Figure 1 shown, the steps of the online temperature self-compensation calibration method for a force-measuring transmitter are as follows:

[0053] S100: Synchronously excite the working strain gauge and the temperature compensation gauge through a dual-channel precision constant current source. The material parameters of the working strain gauge and the temperature compensation gauge are the same and they are in the same temperature field. The output current values of the dual-channel constant current source are equal and are closed-loop controlled by a high-precision reference voltage source and a sampling resistor.

[0054] S200: synchronously collect the differential voltage signal of the working strain gauge and the differential voltage signal of the temperature compensation sheet, perform synchronous analog-to-digital conversion through the 24-bit ADC integrated in the embedded MCU and the 24-bit ADC of the external signal conditioning chip, and obtain a first conversion result AD_Result_work and a second conversion result AD_Result_temp.

[0055] S300: Based on the second conversion result AD_Result_temp, the first conversion result AD_Result_work is temperature compensated in real time by a dynamic weighting algorithm to generate a compensated output value AD_Compensated, and the dynamic weighting algorithm adjusts the temperature influence coefficient according to the gradient change of the second conversion result AD_Result_temp.

[0056] S400: The embedded MCU controls the 16-bit DAC to output the target voltage to the V / V conversion circuit and the V / I conversion circuit to generate isolated voltage transmission signals and current transmission signals.

[0057] S500: In response to the zero-point calibration instruction or the full-scale calibration instruction, the voltage transmission signal or the current transmission signal is fed back to the MCU internal ADC for sampling in real time, and the DAC output value is dynamically adjusted through the incremental PID algorithm until the error between the feedback signal and the preset target value is less than 0.02% FS, completing the online calibration.

[0058] Specifically, a dual constant current source circuit driven by a high-precision reference voltage source ADR4340 is used, and its output currents I1+ and I2+ are stabilized within the accuracy range of 5.000mA±0.02% through closed-loop feedback control. Among them, the core of the constant current source is composed of a low-noise operational amplifier OPA2188 and a precision sampling resistor RS2 (resistance is 500Ω±0.01%). By real-time monitoring of the voltage drop of the sampling resistor and performing a differential comparison with the 2.500V reference output by ADR4340, the driving voltage of the operational amplifier is dynamically adjusted to ensure the long-term stability of the constant current source. The working strain gauge and the temperature compensation gauge use the KFG-5-350-C1-23 strain gauge produced in the same batch, with a temperature coefficient of ±3ppm / ℃, and the epoxy resin packaging process is used to ensure that the two are in the same heat conduction path, thereby eliminating the compensation error introduced by the uneven distribution of the temperature field.

[0059] Furthermore, the embedded MCU uses ADuCM360 with built-in 24-bit Σ-Δ ADC. Its differential input channel AIN0-AIN1 synchronously collects the differential voltage signal (Vp-Vn) of the working strain gauge at a sampling rate of 2400 times per second, and controls the ADC channel of the external signal conditioning chip NSA2860 through the SPI bus to collect the differential voltage signal (Vt±Vt-) of the temperature compensation sheet at the same rate. In order to achieve strict synchronous acquisition, the ADC start signal of ADuCM360 and the CONVST pin of NSA2860 are linked through a hardware trigger signal to ensure that the sampling time deviation of the two ADCs is less than 1μs. The first conversion result AD_Result_work and the second conversion result AD_Result_temp obtained are stored in a ring buffer and compensated in real time through a dynamic weighting algorithm. It should be understood that the dynamic weighting algorithm is not simply a static subtraction of the two signals, but an adaptive adjustment of the weight of the temperature sensitivity matching coefficient α according to the gradient change (ΔT / Δt) of AD_Result_temp. For example, when the temperature change rate ΔT / Δt≥0.5℃ / s, the algorithm increases α from the calibrated value of 0.98 to 1.05 to compensate for the signal tracking delay caused by thermal inertia when the temperature changes rapidly; when ΔT / Δt≤0.1℃ / s, the calibrated value of α=0.98 is restored, thereby achieving a compensation accuracy better than ±0.05%FS in both transient and steady-state temperature fields.

[0060] Furthermore, the compensated output value AD_Compensated is converted into an analog voltage through the 16-bit DAC module (model DAC80508) integrated in the ADuCM360, and outputs a 0-5V standard signal through the V / V conversion circuit. The V / V conversion circuit uses a non-inverting amplifier constructed with a zero-drift operational amplifier LTC2057, whose gain is set to 2.000±0.01% by a precision resistor network, and suppresses nonlinear distortion through a negative feedback loop. For the current transmitter signal output, the DAC output signal is isolated by the voltage follower formed by the LTC2057, and then sent to the V / I conversion circuit composed of the precision instrumentation amplifier INA188 and the MOSFET transistor to output a 4-20mA current signal. It should be understood that the connection of the voltage follower isolates the impedance matching of the voltage and current output paths, avoiding signal crosstalk caused by load changes.

[0061] Furthermore, during the online calibration process, when the user selects the voltage output mode through the dip switch and triggers the zero calibration, the ADuCM360 initializes the DAC output to 0V and starts the internal ADC channel AIN4 to sample the Vout signal in real time. The incremental PID algorithm dynamically adjusts the DAC output code according to the deviation e(k) between the sampled value and the target value 0V. Specifically, the proportional coefficient Kp of the PID controller adopts a two-stage adjustment strategy: in the initial adjustment stage (|e(k)|>0.5%FS), Kp=3.0 to speed up the response speed; when the deviation enters the steady-state range (|e(k)|≤0.5%FS), Kp switches to 0.5 to suppress overshoot. The integral time Ti and the differential time Td are dynamically coupled according to the temperature-compensated AD_Compensated value. For example, when AD_Compensated>80%FS, Ti is shortened from the default 5ms to 3ms, and Td is extended from 0.5ms to 1ms to cope with the nonlinear error caused by the sudden change of load. Through the above adjustment mechanism, the convergence time of zero point calibration can be shortened to less than 35ms, and the steady-state error is controlled within ±0.01%FS.

[0062] The benefits of this embodiment are: through the closed-loop dynamic control and synchronous acquisition mechanism of the dual constant current source, the defects of temperature drift and low efficiency of manual calibration caused by traditional potentiometer adjustment are eliminated; the dynamic weighting algorithm is combined with the temperature change rate to adaptively adjust the compensation coefficient, which solves the hysteresis problem of fixed compensation parameters in transient temperature fields; the multi-stage adjustment strategy of the incremental PID algorithm significantly improves the response speed while ensuring the calibration accuracy. In addition, the isolation design of the voltage and current output paths ensures the independence and consistency of the two transmission signals within the full range.

[0063] Embodiment 2

[0064] In this embodiment, Figure 2 The temperature online self-compensation principle diagram shown in the figure generates two precise constant current sources I1+ and I2+. The currents of the two sources are the same. The constant current is set to about 5mA. Because it is a constant current excitation, the voltage drop generated by the strain gauge is only a single-valued function of the resistance. When the two working strain gauges are working, one is under tension and the other is under pressure, and their resistance changes in opposite directions, such as Figure 1As shown in the figure, the differential voltage of the two working strain gauges is collected and sent to the 24-bit ADC inside the MCU for AD conversion after signal conditioning, and the AD conversion result is set as AD_Result_wort. The two temperature compensation gauges use the same strain gauge as the working strain gauge or a dedicated temperature compensation gauge. In addition to adjusting the deviation caused by the two constant currents, the main function of these two temperature compensation gauges is temperature compensation. The resistance change of the two is only affected by temperature. Since the working strain gauge and the temperature compensation gauge have the same performance and are in the same temperature field, the resistance change caused by the temperature change of the working strain gauge and the resistance change caused by the temperature change of the temperature compensation gauge are basically the same. The differential signal of the two temperature compensation strain gauges is sent to the 24-bit ADC for AD conversion, and the conversion result is set as AD_Result_tempt and obtained by the MCU. The difference between AD_Result_wort and AD_Result_tempt is used as the final sensor output signal conversion value. Since the two are synchronous AD conversion results, the difference between the conversion values ​​of the two basically eliminates the sensor signal changes caused by temperature changes, thereby achieving the effect of online temperature self-compensation.

[0065] Furthermore, if Figure 3 The schematic diagram of the principle of online self-calibration of analog transmission signal shown in the figure, the output selection is set by the dip switch, the output of analog transmission signal can be a single current or voltage transmission signal, or it can output standard voltage and current transmission signals at the same time, the 16-bit DAC outputs the relevant voltage signal under the control of the MCU, and outputs the voltage transmission signal to the V / V conversion all the way, and outputs the analog current transmission signal after following the output voltage of the DAC and sending it to the V / I conversion, the voltage following can ensure that the two transmission signals are outputted simultaneously without affecting each other. Taking 0-5V output as an example, when zero point calibration is required, the output and calibration type are selected as 0-5V through the dip switch, and the zero key is pressed to send the real-time output of the voltage transmission signal to the 24-bit ADC inside the MCU for AD conversion. According to the preset zero point calibration target value, the PID adjustment program solidified in the MCU automatically completes the zero point calibration and outputs 0V. When full scale calibration is required, just press the full scale calibration key, and the PID adjustment program automatically completes the full scale calibration and outputs 5V. Other calibrations are similar.

[0066] Furthermore, if Figure 4The system hardware block diagram of the function of the present invention shown in the figure has a system power supply of 24VDC, which is used to power various modules of the system after voltage regulation. The sensor is excited by a dual-channel precision constant current source, and the differential signal of the working strain gauge is sent to the high-resolution ADC integrated inside the MCU for conversion, and the differential signal of the temperature compensation sheet is sent to the external 24-bit ADC for synchronous conversion and the conversion result is read by the MCU. The MCU controls the 16-bit DAC to perform the corresponding digital-to-analog conversion and output the corresponding current and voltage analog transmission signals through the V / I and V / V links. A 4-bit DIP switch is configured in the system to select the output type of the analog transmission signal. The two independent buttons are the zero-point calibration and full-scale calibration function keys. Based on the output type selection, the corresponding output calibration is completed by the PID program solidified in the FLASH inside the MCU by pressing the relevant buttons.

[0067] Furthermore, if Figure 5 The system power supply circuit is powered by a 24V DC power supply, where D1 is a transient suppression diode, which can achieve surge protection and overvoltage protection. F1 is a self-recovering fuse, which can achieve system overcurrent protection. D2 is a power reverse connection protection diode. The LDO voltage regulator U4 obtains a 3.3V power supply VCC to provide power for the MCU and other digital circuits. U5 generates a 5V analog power supply 5VA to power operational amplifiers and reference chips, and then U3 generates a 3.3V analog power supply to provide analog power for the MCU and external ADC / DAC chip NSA2860. The analog ground and digital ground of the system are routed separately to prevent the digital circuit from interfering with the analog circuit. Then the analog ground and the digital ground are connected at one point by a 0 ohm resistor R1 to ensure that the potential of the digital ground and the analog ground is the same.

[0068] Furthermore, if Figure 6 The circuit diagram of the dual constant current excitation working strain gauge and temperature compensation sheet is shown in Figure 1. Taking the first I1+ circuit as an example, U10 and R14, R15, R18, R20 and RS2 form a constant current circuit, where R15 = R18 = R14 = R20, RS2 is a sampling resistor, and the generated constant current I1+ = 2.5V / RS2. The constant current size can be obtained by reasonably selecting the resistance value of RS2, generally about 5mA, and the 2.5V reference voltage is obtained by the high-precision reference chip U11. When designing, I1+ and I2+ are equal, and the two constant currents drive two working strain gauges and temperature compensation sheets. The differential voltage S+-S- of the working strain gauge is filtered and limited to obtain the Vp-Vn differential signal sent to the 24-bit ADC inside the MCU for AD conversion, and the differential signal Vt+-Vt- of the compensation sheet is sent to the external NSA2860 internal 24-bit ADC for conversion, and the two are carried out synchronously.

[0069] Furthermore, if Figure 7The MCU minimum system and related peripheral circuit diagrams of the present invention are shown. U9 in the figure is an MCU, which is a domestically produced high-performance analog hybrid device. It integrates a multi-channel high-resolution 24-bit ADC and a low-noise programmable gain amplifier PGA. The differential signal of the sensor working strain gauge is sent to the internal AIN0-AIN1 differential channel. During calibration, the relevant analog transmission signal is processed and sent to the AIN4 channel and AIN5 channel for AD conversion. The PID adjustment program is solidified in the FLASH inside the MCU. The MCU and the external NSA2860 exchange data through the SPI bus. SW1 is a 4-bit DIP switch for output type selection and calibration selection. R26 and C27 are power-on reset circuits, S1 and S2 are full-scale calibration and zero-point calibration function keys respectively, and S3 is a reset key. Pressing this key during operation can achieve a reset operation. DS1 is a status light for displaying the working status of the transmitter.

[0070] Furthermore, if Figure 8 The analog transmission signal output and real-time acquisition circuit shown in the figure. U3 is an analog hybrid device that integrates a 24-bit ADC and a 16-bit DAC. It also integrates a PGA with 1x to 256x amplification. Its ADC is used to perform AD conversion on the differential signal Vt+-Vt- of the temperature compensation chip. The 16-bit DAC generates the corresponding analog voltage output, which is amplified and transformed by the U1 low-noise operational amplifier to generate a voltage analog transmission signal output Vout. The output voltage V_DET is obtained after voltage division by the precision low-drift resistors RS_V1 and RS_V2 and sent to the AIN4 channel inside the MCU for feedback, which is used for PID adjustment and calibration of the analog voltage transmission output signal. The voltage follower of U2 is sent to the precision V / I conversion circuit composed of U7, Q3, Q4, R10, R13, etc. to generate an analog current transmission signal output. U8 is a single-pole double-throw analog switch. When the current transmission signal needs to be output normally, B-A2 is turned on and the output current transmission signal IOUT is output. When the signal needs to be calibrated, the MCU controls the 6th pin of U8, turns on B-A1, and U12 follows it. After the RS1 sampling resistor realizes the current-voltage conversion, it is sent to the AIN5 channel of the ADC inside the MCU for collection and feedback. The PID link realizes the automatic calibration of the current transmission signal. R12 and Q4 in the figure form a current limiting resistor. The current limiting value depends on the resistance value of R12. R13 and R10 use a precision resistor of 1 / 1000 25ppm. The current and voltage transmission output signals in the figure can be output separately or simultaneously. Due to the existence of the U2 voltage follower, the two do not affect each other. D3 plays a role in anti-reverse. The voltage follower formed by U12 is used to eliminate the influence of the analog switch on resistance (about 1 ohm) on current sampling.

[0071] Although the present invention has been specifically described above with reference to preferred embodiments thereof, it is to be understood that the invention is not limited to the embodiments described above, but that various modifications and variations may be made by those skilled in the art without departing from the spirit of the invention, and such modifications and variations shall fall within the scope defined by the appended claims and their equivalents.

Claims

1. An on-line temperature self-compensation calibration method for a force-measuring transmitter, characterized in that The steps of the calibration method include: Synchronously exciting the working strain gauge and the temperature compensation gauge through a dual-channel precision constant current source. The working strain gauge and the temperature compensation gauge have the same material parameters and are in the same temperature field. The output current values of the dual-channel constant current source are equal and are closed-loop controlled by a high-precision reference voltage source and a sampling resistor; Synchronously collecting the differential voltage signal of the working strain gauge and the differential voltage signal of the temperature compensation gauge, and respectively performing synchronous analog-to-digital conversion through the 24-bit ADC integrated inside the embedded MCU and the 24-bit ADC of the external signal conditioning chip to obtain the first conversion result AD_Result_work and the second conversion result AD_Result_temp; Based on the second conversion result AD_Result_temp, perform real-time temperature compensation on the first conversion result AD_Result_work through a dynamic weighting algorithm to generate a compensated output value AD_Compensated. The dynamic weighting algorithm adjusts the temperature influence coefficient according to the gradient change of the second conversion result AD_Result_temp; Control the 16-bit DAC of the embedded MCU to output a target voltage to the V / V conversion circuit and the V / I conversion circuit to generate isolated voltage transmission signals and current transmission signals; In response to a zero calibration command or a full-scale calibration command, feedback the voltage transmission signal or the current transmission signal to the ADC inside the MCU for sampling in real time, and dynamically adjust the DAC output value through an incremental PID algorithm until the error between the feedback signal and the preset target value is less than 0.02% FS to complete online calibration.

2. The online temperature self-compensation calibration method for the force-measuring transmitter according to claim 1, characterized in that The dynamic weighting algorithm includes: According to the current value and the historical sampling sequence of the second conversion result AD_Result_temp, calculate the temperature change rate ΔT / Δt. When ΔT / Δt exceeds the threshold, use the following formula to compensate AD_Result_work: AD_Compensated = AD_Result_work - [α·(AD_Result_temp - AD_Result_temp_initial) + β·ΔT / Δt], where α is the temperature sensitivity matching coefficient between the working strain gauge and the temperature compensation gauge, β is the temperature change rate compensation coefficient, AD_Result_temp_initial is the conversion result of the temperature compensation gauge at the initial calibration temperature, and α and β are determined by least squares fitting in the calibration experiment.

3. The on-line temperature self-compensation calibration method for the force measuring transmitter according to claim 1, characterized in that, The triggering method of the zero calibration command includes: After selecting the voltage or current transmission signal type through the DIP switch and pressing the zero calibration button, the MCU controls the DAC to output the zero target voltage and collects the feedback signal in real time; when the deviation between the feedback signal and the target value exceeds the dead zone range, the PID algorithm performs rapid adjustment with a proportional coefficient Kp = 3.0 and an integral time Ti = 5ms; when the deviation enters the dead zone range, switch to Kp = 0.5 and Ti = 20ms to suppress overshoot until the steady-state error is less than 0.02% FS.

4. The online temperature self-compensation calibration method for the force-measuring transmitter according to claim 1, characterized in that, The incremental PID algorithm includes: The output increment Δu(k) of the PID controller is determined by the following formula: Δu(k)=Kp·[e(k)-e(k-1)]+Ki·e(k)+Kd·[e(k)-2e(k-1)+e(k-2)], wherein e(k) is the error at the kth sampling moment, Ki=Kp·T / Ti, Kd=Kp·Td / T, and T is the sampling period; the Kp, Ti, and Td are dynamically adjusted according to the absolute value of the error |e(k)|: when |e(k)|>1%FS, Kp=4.0, Ti=2ms, and Td=0.5ms; when |e(k)|≤0.5%FS, Kp=0.8, Ti=10ms, and Td=2ms.

5. The on-line temperature self-compensation calibration method for the force measuring transmitter according to claim 1, characterized in that The step of dynamically adjusting the output current value of the dual-channel constant current source includes real-time monitoring of the differential voltage signal of the temperature compensation sheet. When it is detected that its value exceeds a preset range, the MCU adjusts the reference voltage drive value of the constant current source through the SPI bus to return the differential voltage of the temperature compensation sheet to the calibration range, and simultaneously corrects the excitation current of the working strain gauge to maintain measurement consistency.

6. The online temperature self-compensation calibration method for the force-measuring transmitter according to claim 1, characterized in that, The output switching of the voltage transmission signal and the current transmission signal is achieved by the following steps: The upper two digits of the DIP switch select the current output type, and the lower two digits select the voltage output type; When two output types are selected at the same time, the MCU controls the analog switch to route the DAC output signal to the V / V conversion circuit and the V / I conversion circuit respectively, and the voltage follower of the V / I conversion circuit isolates the two signals to prevent mutual interference.

7. An on-line temperature self-compensation circuit for a force measuring transmitter, which is used to implement the on-line temperature self-compensation calibration method for the force measuring transmitter described in any one of claims 1-6, characterized in that, The online temperature self-compensation circuit comprises: Dual-channel precision constant current source module, which consists of a high-precision reference voltage source, an operational amplifier and a sampling resistor, and is used to provide equal excitation current to the working strain gauge and the temperature compensation sheet; A signal acquisition module, including an embedded MCU and an external signal conditioning chip, wherein the MCU has an internal integrated 24-bit ADC for acquiring the differential voltage signal of the working strain gauge, and the 24-bit ADC of the external signal conditioning chip is used to synchronously acquire the differential voltage signal of the temperature compensation sheet; The transmission output module includes a 16-bit DAC, a V / V conversion circuit and a V / I conversion circuit. The output end of the DAC is connected to a voltage follower composed of an operational amplifier, and a voltage transmission signal is generated through the V / V conversion circuit and a current transmission signal is generated through a mirror current source circuit. A feedback calibration module, including a multi-channel analog switch and a voltage-dividing sampling circuit, is used to feed back a voltage transmission signal or a current transmission signal to an ADC inside the MCU; The power module, which consists of an LDO regulator, a π-type filter circuit and a transient suppression diode, is used to provide isolated power for the constant current source, MCU and the transmitter output module.

8. The on-line temperature self-compensation circuit of the force-measuring transmitter according to claim 7, characterized in that, In the dual-channel precision constant current source module, each constant current source includes a reference voltage source, whose output end is connected to the non-inverting input end of the operational amplifier; the inverting input end of the operational amplifier is connected to one end of the sampling resistor, and the output end drives the base of the bipolar transistor; the emitter of the bipolar transistor is connected to the other end of the sampling resistor and serves as the constant current output end, and the collector is connected to the working strain gauge or the temperature compensation sheet.

9. The on-line temperature self-compensation circuit of the force-measuring transmitter according to claim 7, characterized in that The V / I conversion circuit comprises: A voltage follower, wherein the input end is connected to the output end of the DAC, and the output end is connected to a mirror current source composed of an operational amplifier, a bipolar transistor and a sampling resistor; The two ends of the sampling resistor are connected to a differential amplifier, and the output end thereof is fed back to the ADC inside the MCU after being switched by an analog switch; The multi-way analog switch in the feedback calibration module is a single-pole double-throw type, with its common end connected to the current transmission signal sampling resistor, the first switching end connected to the current output path, and the second switching end connected to the input end of the differential amplifier. The output end of the differential amplifier is connected to the ADC inside the MCU through an RC filtering circuit.

10. The on-line temperature self-compensation circuit for the force-measuring transmitter according to claim 7, characterized in that, In the power module, the analog power supply and the digital power supply are isolated by magnetic beads, and the analog ground is connected to the digital ground at a single point through a 0Ω resistor; the π-type filter circuit is composed of an inductor and a ceramic capacitor and an electrolytic capacitor in parallel, and its input end is connected in series with a self-recovery fuse and an anti-reverse diode; The GPIO pin of the embedded MCU is connected to a 4-bit dip switch and a calibration button. The coding state of the dip switch is input to the input capture channel of the MCU through a pull-up resistor. The trigger signal of the calibration button is processed by the debouncing circuit and triggers the interrupt service program to execute the calibration process.

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