High-frequency loss characteristic characterization method and system of non-closed magnetic core

Abstract

The invention discloses a high-frequency loss characteristic characterization method of a non-closed magnetic core. The method comprises the following steps: constructing a test unit; the test unit comprises an inductor arranged around the non-closed magnetic core; under the condition of high-frequency large signals enabling the non-closed magnetic core to enter a nonlinear working state, a test unit is used for collecting and preprocessing voltage signals and current signals at the two ends of the non-closed magnetic core; performing Fourier transform on the preprocessed voltage signal and current signal to generate voltage frequency spectrum information and current frequency spectrum information; calibrating the parasitic parameters of the test unit and the time delay of the system; calculating inductor coil winding loss parameters; acquiring impedance parameters, inductance parameters and resistance parameters of the inductor at different frequency points according to the ratio of the voltage spectrum information to the current spectrum information; and performing fitting operation and analysis on the impedance parameter, the inductance parameter and the resistance parameter to generate a magnetic core loss characteristic curve under the high-frequency large-signal condition for enabling the non-closed magnetic core to enter a nonlinear working state.

Description

TECHNICAL FIELD

[0001] The present application relates to the technical field of magnetic material characterization, and particularly relates to a high-frequency loss characterization method and system for a non-closed magnetic core. BACKGROUND

[0002] With the rapid evolution of the electronic information industry towards high frequency, miniaturization and integration, the performance requirements of high-frequency magnetic devices in the fields of 5G communication, new energy vehicles, aerospace, etc. are becoming increasingly stringent. Soft magnetic materials have unique advantages such as high magnetic permeability, low coercivity, high-frequency low loss, and easy compatibility with microelectronic processes, and have become key magnetic materials for preparing core devices such as high-frequency inductors, micro transformers, and magnetic isolators. The high-frequency magnetic core loss characteristics of soft magnetic materials directly determine the energy conversion efficiency, temperature rise control and long-term working reliability of the devices.

[0003] Magnetic core loss is the energy loss of soft magnetic materials under the action of alternating magnetic field due to magnetic hysteresis, eddy current and residual loss. Especially under high-frequency working conditions (such as MHz to GHz frequency band), eddy current loss and abnormal skin effect lead to complex frequency dependence of loss characteristics. Precise characterization of high-frequency magnetic core loss of soft magnetic thin films has become a key link in material research and development and device design. However, there are still many problems to be solved in the existing magnetic core loss characterization technology:

[0004] On the one hand, traditional loss measurement methods are mostly for bulk or thick film materials, such as using B-H loop instrument to measure magnetic hysteresis loss. However, the thickness of soft magnetic thin film is usually only nanometers to microns, and the magnetic signal is weak. The existing equipment is difficult to accurately capture the details of the magnetic hysteresis loop under high-frequency dynamic magnetic field, resulting in large measurement error. At the same time, although the conventional impedance analysis method can measure the impedance of the device to inverse the loss, it is affected by the parasitic parameters (such as probe inductance, cable distributed capacitance) and test delay of the test unit. Data distortion easily occurs at high frequency band, and the magnetic core loss and other parasitic losses cannot be accurately separated.

[0005] On the other hand, the existing characterization system lacks a special test architecture for soft magnetic materials. The signal generation module is difficult to output stable high-frequency large signals to simulate the actual working conditions. The signal-to-noise ratio improvement capability of the data acquisition module for weak magnetic and electric signals is insufficient, and the system error is not effectively calibrated and compensated in the data processing process, resulting in that the repeatability and accuracy of the measurement results are difficult to meet the requirements of engineering applications. In addition, the current loss characterization is mostly focused on a single frequency point or a narrow frequency range, and cannot realize continuous characterization of loss characteristics in a wide frequency band, which is difficult to support the wide frequency design requirements of high-frequency magnetic devices.

[0006] Therefore, developing a method and system capable of accurately and efficiently characterizing the core loss under high-frequency conditions to solve the problems of low measurement accuracy, large system error, and narrow applicable frequency band in the prior art is of great significance for promoting the research and development iteration of new soft magnetic materials and the performance optimization of high-frequency magnetic devices, and is also a technical bottleneck that needs to be broken through in the field of electronic materials and devices. SUMMARY

[0007] The purpose of the present application is to provide a method and system for characterizing the high-frequency loss characteristics of a non-closed magnetic core, which generates a high-frequency large signal excitation that can make the non-closed magnetic core enter a nonlinear working state, accurately collects voltage and current signals, extracts fundamental and harmonic components, and combines digital signal processing technology and software simulation technology to realize accurate measurement of the loss of a non-closed magnetic core under a maximum frequency of 100 Mhz and a maximum excitation current of 200 mA, thereby solving the problem that the prior art cannot accurately measure the loss of a soft magnetic thin film under a large signal condition, and also providing a feasible solution for calculating the complex permeability and loss under different magnetic field strengths under an applied bias magnetic field.

[0008] To solve the above technical problems, the present application provides the following technical solutions:

[0009] In a first aspect, a method for characterizing the high-frequency loss characteristics of a non-closed magnetic core includes the following steps:

[0010] S1, constructing a test unit; the test unit includes inductors arranged around the non-closed magnetic core;

[0011] S2, under a high-frequency large signal condition that makes the non-closed magnetic core enter a nonlinear working state, collecting and preprocessing the voltage signal and current signal across the non-closed magnetic core using the test unit; performing Fourier transform on the preprocessed voltage signal and current signal to generate voltage spectrum information and current spectrum information;

[0012] S3, calibrating the parasitic parameters of the test unit and the system's own time delay; calculating the inductor coil winding loss parameters;

[0013] S4, according to the ratio of the voltage spectrum information and the current spectrum information, obtaining the impedance parameters, inductance parameters, and resistance parameters of the inductor at different frequency points;

[0014] S5, performing fitting operation and analysis on the impedance parameters, inductance parameters, and resistance parameters to generate a core loss characteristic curve under the high-frequency large signal condition that makes the non-closed magnetic core enter a nonlinear working state.

[0015] Preferably, in S2, the preprocessing includes:

[0016] The Daubechies wavelet basis is selected as the wavelet basis for high-frequency signal denoising; the number of wavelet decomposition levels is determined according to the signal frequency range; the wavelet coefficients are thresholded using a soft thresholding method to remove high-frequency noise components; the processed wavelet coefficients are reconstructed using wavelet to obtain the denoised voltage signal and the current signal.

[0017] A digital filter is used, and the window function method is employed to filter the denoised voltage signal and the current signal.

[0018] Preferably, S3 includes: conducting a preliminary experiment using an air-core inductor to obtain voltage and current data of the air-core inductor at different frequencies and the system's own time delay; generating initial resistance and inductance values ​​of the air-core inductor at different frequencies using the voltage and current data of the air-core inductor; changing the phase difference by altering the impedance angle to obtain the inductance and resistance values ​​of the air-core inductor that are closest to those measured under small-signal conditions, thereby generating the final phase difference and impedance angle.

[0019] Preferably, S3 includes defining the expressions for the input voltage V and output current I in the equivalent circuit as follows:

[0020]

[0021] Where V0 is the peak voltage and I0 is the peak current. Let V and I be the initial phases of voltage and current, respectively, and ω be the angular frequency. The complex domain expressions for V and I are: for:

[0022]

[0023] Where j is the imaginary unit;

[0024] Therefore, the expression for the impedance X of this circuit is:

[0025]

[0026] Right now:

[0027]

[0028] in, The phase difference between the input and output is the phase angle of the inductor itself, where the real part is... To test the equivalent resistance R1 of the inductor loss, the imaginary part... For inductively coupled XL, probe mismatch and phase shift of the test unit itself occurred during the testing process. This will affect the calculation of the inductor's equivalent resistance and inductance, thus causing deviations in the loss calculation. Considering the phase shift, the complex domain expression of the actual output current I' becomes... for:

[0029]

[0030] in, This is due to the probe mismatch and the phase shift caused by the test unit.

[0031] According to the above formula, the actual impedance X' is:

[0032]

[0033] Among them, the real part To account for the phase-shifted resistance R' of the test circuit, the imaginary part... To account for the inductive reactance XL' of the test circuit after phase shift, the error formula can be derived from the loss calculation formula as follows:

[0034]

[0035] Where R and R' represent the resistance of the test circuit before and after considering the phase shift, respectively; P and P' represent the loss of the test circuit before and after considering the phase shift, respectively; and ΔP represents the error factor of the effect of phase shift on loss calculation.

[0036] The inductance and permeability tests are correlated using the following formula:

[0037]

[0038] Where N is the number of turns in the inductor coil, w M t M These represent the width and thickness of the magnetic core, respectively. M N is the effective length of the magnetic core. d It is the demagnetizing factor. For a finite-sized ferromagnetic material, when magnetized in an external magnetic field H, the surface magnetic poles cause a magnetic field to be generated inside the magnet that is opposite in direction to the magnetization intensity M. This magnetic field is called the demagnetizing field, denoted by H. d Indicates. Demagnetizing field H d The size of H depends on the shape of the magnet and the strength of the magnetic poles, and is directly proportional to the magnetization M, i.e., H d =-N d M, where N d It is a demagnetization factor that is only related to the shape of the sample.

[0039] Preferably, S4 also includes: measuring the circuit impedance X to address parasitic effects within the voltage probe and inductor. / The expression is:

[0040]

[0041] Where R1 is the inductor resistance and ωL1 is the inductor reactance. The capacitive reactance of the parasitic circuit is introduced, where C is the parasitic parameter and L1 is the inductance value. In the inductance testing circuit, Therefore, the formula can be further derived as follows:

[0042]

[0043] The real part is represented as:

[0044]

[0045] That is, the resistance R' of the test circuit after considering parasitic parameters, which can be obtained from the formula derived above:

[0046]

[0047] Preferably, S5 includes: using the least squares method to perform curve fitting and calculation on the data to obtain the impedance characteristic curve of the inductor under high frequency with a large signal, which causes the non-closed magnetic core to enter a nonlinear working state, as well as the inductor resistance data at different frequencies; generating the inductor's inductance, quality factor, and total loss density by analyzing the shape and data characteristics of the impedance characteristic curve; obtaining the inductor's winding resistance and winding loss through simulation software; generating the core loss by subtracting the total loss and winding loss; obtaining the magnetic induction intensity value of the magnetic core under the same current excitation through simulation software, and plotting the core loss characteristic curve of the magnetic core under high frequency with a large signal.

[0048] Secondly, a high-frequency loss characterization system for a non-closed magnetic core includes a testing unit; the testing unit includes:

[0049] The system includes a signal generation module, a data acquisition module, a data processing module, a simulation design module, and an inductor located around the non-closed magnetic core.

[0050] The signal generation module is used to generate a large high-frequency signal that causes the non-closed magnetic core to enter a nonlinear operating state.

[0051] After the data acquisition module acquires the voltage and current signals of the inductor, the data processing module preprocesses the voltage and current signals and generates voltage spectrum information and current spectrum information through Fourier transform.

[0052] The simulation design module obtains the winding coil resistance, winding loss, and magnetic induction intensity value of the inductor under the same current excitation through simulation.

[0053] Preferably, the signal generation module includes a signal generator and a power amplifier; the signal generator is used to generate a large signal under high-frequency conditions that causes the non-closed magnetic core to enter a nonlinear operating state; the power amplifier is used to amplify the signal and apply it to the inductor under test; the data acquisition module includes a voltage probe, a current probe, a coaxial cable, and an oscilloscope; the voltage probe and the current probe are used to acquire the voltage signal across the inductor and the current signal flowing through the inductor, respectively, and the coaxial cable is used to transmit the signal to the oscilloscope; the oscilloscope uses a high-precision ADC chip to capture the subtle features of the high-frequency signal.

[0054] Preferably, the signal generator uses direct digital frequency synthesis technology to generate high-frequency signals with a frequency range of 1μHz to 150MHz and an amplitude range of 1mVPP to 20VPP; the power amplifier is a broadband linear amplifier that achieves power output across the entire frequency range, with a maximum output power of 100W; the current probe is a non-contact type, relying on a Hall effect sensor, with a bandwidth of up to 100MHz, a maximum sensitivity of 10mA / div, and an AC accuracy of ±3%; the voltage probe has a bandwidth of up to 200MHz, a high input impedance of 10MΩ, and a low parasitic capacitance of 15pF.

[0055] Compared with the prior art, the beneficial effects achieved by the present invention are:

[0056] (1) Through specially designed test units and signal processing technology, it is possible to accurately measure the high-frequency loss characteristics of magnetic cores under high frequency (up to 100MHz) and large signal (up to 200mA excitation current) conditions, covering most of the operating frequencies of integrated magnetic cores in practical applications;

[0057] (2) By using wavelet transform, Fourier transform, least squares fitting and other methods, the impact of external interference on measurement accuracy is greatly reduced;

[0058] (3) Through system calibration, the effects of time delay and parasitic parameters in the test unit are effectively eliminated, the sensitivity of loss to high frequency is reduced, and the measurement accuracy is further improved;

[0059] (4) The test frequency range is from 10MHz to 100MHz, which can cover most of the operating frequencies of the magnetic core in practical applications. By analyzing the loss characteristics at different frequencies, a comprehensive understanding of the frequency response characteristics of the magnetic core can be obtained, providing an important reference for circuit design;

[0060] (5) By comparing the loss characteristics under different signal intensities, the nonlinear characteristics of soft magnetic films under large signal conditions are studied, providing an important basis for the design and optimization of inductors;

[0061] (6) The inductor adopts a PCB hole design, which can realize the replacement and testing of different magnetic core materials, and facilitates the performance evaluation of different magnetic core inductors under high frequency and large signal.

[0062] (7) It has the characteristic of being able to apply an external bias magnetic field, which can be used to evaluate the μ-f curve characteristics and BH curve characteristics under different magnetic field strengths. Attached Figure Description

[0063] The accompanying drawings are provided to further illustrate the invention and form part of the specification. They are used in conjunction with embodiments of the invention to explain the invention and do not constitute a limitation thereof. In the drawings:

[0064] Figure 1 This is a flowchart of the method steps in Embodiment 1 of the present invention;

[0065] Figure 2 This is a finite element simulation model diagram of an inductor with a magnetic core to be tested inside, as shown in Embodiment 1 of the present invention.

[0066] Figure 3 This is a schematic diagram of the test unit in Embodiment 1 of the present invention;

[0067] Figure 4 This is a system framework diagram of Embodiment 2 of the present invention;

[0068] Figure 5 This is a system configuration diagram of Embodiment 2 of the present invention;

[0069] Figure 6 This is a schematic diagram of a complex permeability testing device suitable for frequencies below 1 GHz according to Embodiment 2 of the present invention;

[0070] Figure 7 This is a schematic diagram of the complex permeability test results applicable to frequencies below 1 GHz in Embodiment 2 of the present invention;

[0071] Figure 8 This is a schematic diagram of loss testing for an externally applied bias magnetic field applicable to 10-100MHz, according to Embodiment 2 of the present invention.

[0072] Figure 9 This is a graph showing the loss test results provided in Embodiment 2 of the present invention. Detailed Implementation

[0073] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0074] Example 1:

[0075] like Figure 1 As shown, a method for characterizing the high-frequency loss characteristics of a non-closed magnetic core includes the following steps:

[0076] S1. Construct a test unit; the test unit includes inductors arranged around the non-closed magnetic core;

[0077] S2. Use the test unit to collect and preprocess the voltage and current signals at both ends of the non-closed magnetic core; perform Fourier transform on the preprocessed voltage and current signals to generate voltage spectrum information and current spectrum information.

[0078] During signal acquisition, the acquired voltage and current signals often contain a lot of noise due to environmental noise, equipment noise, and interference during signal transmission. Therefore, signal preprocessing is a key step to ensure the accuracy of subsequent analysis. Preprocessing steps, including denoising and filtering, are performed on the acquired voltage and current signals.

[0079] S21. Among them, the removal of high-frequency noise by employing wavelet transform specifically includes:

[0080] A suitable wavelet basis for high-frequency signal processing is selected; since the Daubechies wavelet basis (db4) has good localization properties, it is chosen for denoising high-frequency signals. The number of wavelet decomposition levels is determined based on the signal frequency range; in this embodiment, the number of decomposition levels is set to 5. A soft thresholding method is used to threshold the wavelet coefficients to remove high-frequency noise components; the processed wavelet coefficients are then reconstructed using wavelet decomposition to obtain the denoised signal.

[0081] S22. In order to further remove spurious frequency components in the signal, a digital filter is used for filtering: that is, a bandpass filter is selected with a passband range of 10MHz to 100MHz and a stopband attenuation greater than 60dB. By using an FIR filter and designing it using the window function method, the Hamming window is selected as the window function to filter the denoised signal, remove out-of-band noise, and achieve filtering.

[0082] S23. Subsequently, a Fourier transform is performed on the preprocessed signal to obtain its spectral characteristics. Further, to obtain the signal's spectral information, a Fourier transform is performed on the preprocessed signal:

[0083] Windowing: Blackman-Harris windowing is used to reduce spectral leakage.

[0084] FFT calculation: Perform a Fast Fourier Transform (FFT) on the windowed signal to obtain the signal's spectrum.

[0085] Spectrum interpolation: The interpolation FFT algorithm is used to improve the spectrum resolution, and the interpolation factor is selected as 4 times.

[0086] S3. Calibrate the parasitic parameters and self-delay within the test unit, calculate the inductor coil winding loss parameters, including simulating the inductor using Ansys Maxwell software to obtain the resistance parameters, and then using P... wire =I rms 2 ×R yields the final winding loss;

[0087] In this embodiment, before calculating the impedance of the magnetic core inductor, a preliminary experiment was conducted using an air core inductor to obtain the time delay of the test unit within the range of 4.2-6 ns (10M-100Mhz). Figure 1 The resistance and inductance introduced by the copper winding were obtained using finite element simulation software. The simulation model is as follows: Figure 2 As shown.

[0088] For example, by numerically calculating the voltage and current waveforms, the inductor impedance, inductance, and resistance can be accurately obtained: based on Figure 3 The equivalent circuit is shown, where C1 is the DC blocking capacitor, R1 is the equivalent resistance for testing inductor losses, L1 is the inductance of the test inductor, R2 is the winding resistance, L2 is the winding inductance, and C2 includes the parasitic capacitance of the voltage probe and the parasitic capacitance of the test inductor winding. The expressions for the input V and output I are as follows:

[0089]

[0090] Where V0 is the peak voltage, I0 is the peak current, φ1 and φ2 are the initial phases of the voltage and current, respectively, ω is the angular frequency, and V and I are expressed in the complex domain. for:

[0091]

[0092] Therefore, the expression for the impedance X of this circuit is:

[0093]

[0094] Right now:

[0095]

[0096] Where Δφ is the phase difference between the input and output (the phase angle of the inductor itself), it can be seen that the real part... To test the equivalent resistance R1 of the inductor loss, the imaginary part... The inductive reactance is XL. However, during the test, probe mismatch and the phase shift φ3 of the test unit itself will affect the calculation of the inductor's equivalent resistance and inductance, thus causing deviations in the loss calculation. Considering the phase shift, the complex domain expression of the actual output current I' is... for:

[0097]

[0098] in, This is due to the probe mismatch and the phase shift caused by the test unit.

[0099] According to the above formula, the actual impedance X' is:

[0100]

[0101] Among them, the real part To account for the phase-shifted resistance R' of the test circuit, the imaginary part... To account for the inductive reactance XL' of the test circuit after phase shift, the error formula can be derived from the loss calculation formula as follows:

[0102]

[0103] Where R and R' represent the resistance of the test circuit before and after phase shift, respectively, P and P' represent the loss of the test circuit before and after phase shift, respectively, and ΔP represents the error factor of the effect of phase shift on loss calculation.

[0104] It is worth mentioning that the inductance can also be correlated with the permeability test, and the specific formula is as follows:

[0105]

[0106] Where N is the number of turns in the inductor coil, w M t M These represent the width and thickness of the magnetic core, respectively. M N is the effective length of the magnetic core. d It is the demagnetizing factor. For a finite-sized ferromagnetic material, when magnetized in an external magnetic field H, the surface magnetic poles cause a magnetic field to be generated inside the magnet that is opposite in direction to the magnetization intensity M. This magnetic field is called the demagnetizing field, denoted by H. d Indicates. Demagnetizing field H d The size of H depends on the shape of the magnet and the strength of the magnetic poles, and is directly proportional to the magnetization M, i.e., H d =-N d M, where N d It is a demagnetization factor that is only related to the shape of the sample.

[0107] S4. By calculating the ratio of the voltage spectrum to the current spectrum, obtain the impedance parameters, inductance parameters, and resistance parameters of the inductor at different frequency points.

[0108] Furthermore, considering the parasitic effects inside the voltage probe and inductor, the measured circuit impedance X / The expression is:

[0109]

[0110] Where R1 is the inductor resistance and ωL1 is the inductor reactance. The capacitive reactance of the parasitic circuit is introduced, where C is the parasitic parameter and L1 is the inductance value. In the inductance testing circuit, Therefore, the formula can be further derived as follows:

[0111]

[0112] The real part is represented as:

[0113]

[0114] That is, the resistance R' of the test circuit after considering parasitic parameters, which can be obtained from the formula derived above:

[0115]

[0116] S5. The obtained parameters are fitted and analyzed to finally generate the core loss characteristic curve of the inductor under high frequency and large signal conditions.

[0117] In this embodiment, the compensated impedance data is finally fitted and analyzed: the least squares method is used to fit the data to a curve and calculate the impedance characteristic curve of the inductor at high frequency and the inductor resistance data at different frequencies. By analyzing the shape of the curve and the data characteristics, key parameters such as the inductor's inductance, quality factor, and total loss density are obtained. At the same time, the inductor winding resistance is obtained through simulation software and calculated using formula P. wire =I rms 2 The winding loss is obtained by multiplying the total loss by R. The final core loss can be obtained by subtracting the total loss from the winding loss. The B value of the core under the same current excitation can then be obtained by simulation software, and the BP loss characteristic curve of the core under high frequency and large signal can be plotted.

[0118] Example 2:

[0119] like Figure 1 As shown, a high-frequency loss characterization system for a non-closed magnetic core includes a test unit; the test unit includes: a signal generation module, a data acquisition module, a data processing module, and a simulation design module;

[0120] The signal generation module is used to generate large signals at high frequencies, enabling non-closed magnetic cores to enter a nonlinear operating state.

[0121] The data acquisition module is used to acquire the voltage signal across the inductor and the current signal flowing through the inductor;

[0122] Data processing module: used to preprocess the acquired voltage and current signals and perform Fourier transform and other methods to obtain the signal's spectrum information;

[0123] The simulation design module is used to simulate the winding coil resistance of the inductor, thereby obtaining the winding loss, and to simulate the B value under the same circuit excitation.

[0124] Furthermore, Figure 3 In the schematic diagram of the test unit shown, C1 is the DC blocking capacitor, R1 is the equivalent resistance of the test inductor loss, L1 is the inductance of the test inductor, R2 is the winding resistance, L2 is the winding inductance, and C2 includes the parasitic capacitance of the voltage probe and the parasitic capacitance of the test inductor winding.

[0125] In this embodiment, combined with Figure 2 As shown, the simulated external coil is an inductor winding coil, with a non-closed area in the middle that can be used to place the magnetic core. It should be emphasized that the solenoid cross-section can be rectangular to accommodate rectangular cross-section magnetic cores, or circular or other shapes to accommodate cylindrical or other shaped cross-section magnetic cores.

[0126] Specifically, in combination Figure 5 As shown, the hardware design of the test unit is key to realizing the large-signal core loss test of non-closed magnetic cores at high frequencies. The signal generation module consists of a signal generator and a power amplifier. The signal generator generates a large signal in a high-frequency environment, enabling the non-closed magnetic core to enter a nonlinear operating state. The power amplifier is responsible for amplifying the signal to meet the test requirements and applying it to the inductor under test. The data acquisition module includes a voltage probe, a current probe, a coaxial cable, and an oscilloscope. The voltage probe and current probe are used to acquire the voltage signal across the inductor and the current signal flowing through the inductor, respectively. The coaxial cable transmits the signal to the oscilloscope, which uses a high-precision ADC chip to capture the subtle features of the high-frequency signal.

[0127] Specifically, in combination Figure 6 , Figure 7 As shown, the hardware design of this test unit can also be applied to situations where an external bias magnetic field is applied, to measure the relationship between the permeability and frequency of the magnetic core under different bias magnetic field strengths. Figure 6As shown, a Helmholtz coil providing a strength of 0.1-1T is placed at both radial ends of the inductor. The soft magnetic core sample under test is placed inside the solenoid coil, forming a solenoid inductor. A pair of Helmholtz coils are placed on both sides of the solenoid inductor. The Helmholtz coils generate a DC bias magnetic field, parallel to the plane of the magnetic core and perpendicular to the axial direction of the solenoid inductor. The inductance value L is related to the magnetic permeability μ of the magnetic core. r The relationship is shown in the following formula:

[0128]

[0129] Where N is the number of turns in the inductor coil, w M t M These represent the width and thickness of the magnetic core, respectively. M N is the effective length of the magnetic core. d It is the demagnetizing factor. For a finite-sized ferromagnetic material, when magnetized in an external magnetic field H, the surface magnetic poles cause a magnetic field to be generated inside the magnet that is opposite in direction to the magnetization intensity M. This magnetic field is called the demagnetizing field, denoted by H. d Indicates. Demagnetizing field H d The size of H depends on the shape of the magnet and the strength of the magnetic poles, and is directly proportional to the magnetization M, i.e., H d =-N d M, where N d This is the demagnetizing factor, which depends only on the shape of the sample. μ can be calculated by measuring the inductance value at different inductance values ​​(f) under different H intensities. r The curve characteristics of f. Figure 7 In the equation, H1, H2, and H3 increase sequentially, representing different bias magnetic field strengths. As the magnetic field strength increases, the material's resonance point is delayed, and the permeability decreases accordingly.

[0130] Specifically, such as Figure 8 As shown, under an applied bias magnetic field, the hardware design of this test unit can also simulate the actual working state of the magnetic core, obtaining the true core loss at frequencies from 10MHz to 100MHz. The soft magnetic core sample under test is placed inside a solenoid coil, forming a solenoid inductor. A pair of Helmholtz coils are placed at both ends of the solenoid inductor along its axial direction. The Helmholtz coils generate a DC bias magnetic field in the same direction as the axial direction of the solenoid inductor. The solenoid inductor core itself has an AC magnetic field generated by the alternating current in the coil. Therefore, the magnetic field on the core is the superposition of the two magnetic fields mentioned above, which is close to the working state of the inductor core in an actual circuit. Thus, the measured loss is also closer to the inductor core loss in actual applications.

[0131] In this implementation, a series of experimental tests were conducted to verify the effectiveness and accuracy of the method:

[0132] Test objects: Inductors with different magnetic cores selected, including ferrite, thin film and air core inductors (preliminary experiment);

[0133] Test frequency range: 10MHz to 100MHz;

[0134] Excitation signal range: 50mA to 200mA;

[0135] Test environment: Conducted in a shielded room at room temperature;

[0136] Combination Figure 9 As shown in the figure, the test results demonstrate that the method of this invention can accurately measure the core loss characteristics of non-closed magnetic cores under high-frequency, high-signal conditions. The measurement results agree well with the theoretical calculations. Specific results are as follows:

[0137] Ferrite inductors: impedance measurement error is less than 8% in the range of 10MHz to 100MHz;

[0138] Air-core inductors: impedance measurement error is less than 2% in the range of 10MHz to 100MHz;

[0139] Thin-film inductors: impedance measurement error is less than 6% in the range of 10MHz to 100MHz.

[0140] It should be noted that, in this invention, relational terms such as "first" and "second" are used merely to distinguish one entity or operation from another, and do not necessarily require or imply any such actual relationship or order between these entities or operations. Furthermore, the terms "comprising," "including," or any other variations thereof are intended to cover non-exclusive inclusion, such that a process, method, article, or apparatus that comprises a list of elements includes not only those elements but also other elements not expressly listed, or elements inherent to such a process, method, article, or apparatus.

[0141] Finally, it should be noted that the above are merely preferred embodiments of the present invention and are not intended to limit the present invention. Although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art can still modify the technical solutions described in the foregoing embodiments or make equivalent substitutions for some of the technical features. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the protection scope of the present invention.

Claims

1. A method for characterizing the high-frequency loss characteristics of a non-closed magnetic core, characterized in that, The method comprises the following steps: S1, constructing a test unit; the test unit comprises an inductor arranged around a non-closed magnetic core; S2, under the condition of a high-frequency large signal causing the non-closed magnetic core to enter a nonlinear working state, collecting and preprocessing voltage signals and current signals at both ends of the non-closed magnetic core by using the test unit; performing Fourier transform on the preprocessed voltage signals and current signals to generate voltage spectrum information and current spectrum information; S3, calibrating parasitic parameters of the test unit and system self time delay; and calculating inductor coil winding loss parameters; S4, obtaining impedance parameters, inductance parameters and resistance parameters of the inductor at different frequency points according to the ratio of the voltage spectrum information and the current spectrum information; S5, performing fitting operation and analysis on the impedance parameters, the inductance parameters and the resistance parameters to generate a magnetic core loss characteristic curve under the condition of the high-frequency large signal causing the non-closed magnetic core to enter the nonlinear working state.

2. The method of claim 1, wherein the non- closed magnetic core is a toroidal core. In S2, the preprocessing comprises: selecting a Daubechies wavelet base as a wavelet base for high-frequency signal denoising processing; determining a wavelet decomposition layer number according to a signal frequency range; performing threshold value processing on wavelet coefficients by using a soft threshold value method to remove high-frequency noise components; and performing wavelet reconstruction on the processed wavelet coefficients to obtain the denoised voltage signals and current signals; using a digital filter to filter the denoised voltage signals and current signals by using a window function method.

3. The method of claim 1, wherein the method is characterized in that, S3 comprises: obtaining voltage and current data of the air core inductor and the system self time delay at different frequencies by using pre-experiment of the air core inductor; generating resistance initial values and inductance initial values of the air core inductor at different frequencies based on the voltage and current data of the air core inductor; and changing the phase difference by changing the impedance angle, thereby obtaining inductance values and resistance values of the air core inductor closest to the measured values under a small signal condition to generate a final phase difference and impedance angle.

4. The method of claim 3, wherein the high-frequency loss characteristic of the non- closed magnetic core is characterized by, In S3, expressions of input voltage V and output current I in an equivalent circuit are defined as: where V0is the voltage peak value, I0is the current peak value, are the initial phase of voltage and current, respectively, and ω is the angular frequency, and the complex domain expressions for V and I are are: wherein j is an imaginary unit; Therefore, an expression of circuit impedance X is: That is: wherein, is the phase difference of the input and output, i.e. the phase angle of the inductance itself, wherein the real part is the equivalent resistance R1 testing the inductor losses, the imaginary part is the inductive reactance XL, the probe mismatch during the test and the phase shift of the test unit itself will influence the calculation of the equivalent resistance and inductance of the inductor, and thus the losses calculation will be biased, considering the phase shift, the actual output current I' in the complex domain is expressed as ​ wherein is the phase shift generated by the probe mismatch and the test unit. According to the above formula, the actual impedance X' is: where the real part is the resistance R' of the test circuit considering the phase shift is the inductive reactance XL' of the test circuit considering the phase shift; according to the loss calculation formula, the error formula can be derived as follows: wherein R and R' respectively represent resistances of the test circuit before and after considering the phase shift, P and P' respectively represent losses of the test circuit before and after considering the phase shift, and ΔP represents an error factor of the influence of the phase shift on the loss calculation; The test of the inductance and the permeability is related to each other, and a specific formula is: where N is the number of turns of the inductor coil, w M t M are the width and thickness of the magnetic core, respectively, l M is the effective length of the magnetic core, N d is the demagnetization factor; for a finite-sized ferromagnetic material, when magnetized in an external magnetic field H, a magnetic field is generated inside the magnet, opposite to the direction of the magnetization M, due to the surface poles, this magnetic field is called demagnetizing field, denoted by H d ; the magnitude of the demagnetizing field H d is related to the shape of the magnet and the strength of the poles, and is proportional to the magnetization M, i.e. H d = -N d M, where N d is the demagnetization factor, which is only related to the shape of the sample.

5. The method of claim 4, wherein the method further comprises: S4 also includes: for voltage probe and inductor internal parasitic, measured circuit impedance X / The expression is: where R1 is the inductor resistance, ωL1 is the inductor inductance, C is the parasitic capacitance, L1 is the inductor inductance, and Thus the formula continues to derive: wherein the real part is represented as: That is, the resistance R' of the test circuit considering the parasitic parameters, and according to the above formula, it is obtained that:

6. The method of claim 1, wherein the method is a method of characterizing high frequency loss properties of a non-closed magnetic core, the method comprising: S5 includes: using least squares method to curve fitting and calculation, get the inductor in the high frequency of making the non-closed magnetic core into the nonlinear working state Large signal impedance characteristic curve and inductance resistance data under different frequencies; By analyzing the shape and data characteristics of the impedance characteristic curve, the inductance value, quality factor and total loss density of the inductor are generated; At the same time, the winding resistance and winding loss of the inductor are obtained by simulation software; The core loss is generated by subtracting the total loss and the winding loss; The magnetic induction intensity value of the magnetic core under the same current excitation is obtained by simulation software, and the core loss characteristic curve of the magnetic core under high frequency is drawn.

7. A high-frequency loss characteristic characterization system for a non-closed magnetic core, characterized in that, It comprises a test unit; the test unit comprises: a signal generation module, a data acquisition module, a data processing module, a simulation design module and an inductor arranged around the non-closed magnetic core; the signal generation module is used for generating a large signal under high frequency to make the non-closed magnetic core enter a nonlinear working state; the data acquisition module acquires voltage signals and current signals of the inductor, the data processing module pre-processes the voltage signals and the current signals, and generates voltage spectrum information and current spectrum information through Fourier transform; the simulation design module obtains the winding coil resistance, winding loss and magnetic induction intensity value under the same current excitation of the inductor through simulation means.

8. The system for characterizing high-frequency loss properties of a non- closed magnetic core of claim 7, wherein, The signal generation module comprises a signal generator and a power amplifier; the signal generator is used for generating a large signal under high frequency conditions to make the non-closed magnetic core enter a nonlinear working state; the power amplifier is used for amplifying the signal and applying it to the inductor to be tested; the data acquisition module comprises a voltage probe, a current probe, a coaxial cable and an oscilloscope; the voltage probe and the current probe are respectively used for acquiring voltage signals and current signals flowing through the inductor, and the coaxial cable is used for signal transmission to the oscilloscope; the oscilloscope captures the fine features of high frequency signals through high precision ADC chips.

9. The system for characterizing the high-frequency loss properties of a non- closed magnetic core of claim 8, wherein, The signal generator uses direct digital frequency synthesis technology to generate high frequency signals with a frequency interval of 1 μHz to 150 MHz and an amplitude range of 1 mVPP to 20 VPP; the power amplifier is a wideband linear amplifier, which realizes power output in the full frequency range, and the maximum output power reaches 100 W; the current probe is a non-contact type, relying on a Hall effect sensor, with a bandwidth of up to 100 MHz, a maximum sensitivity of 10 mA / div, and an AC accuracy of ±3%; the voltage probe has a bandwidth of up to 200 MHz, a high input impedance of 10 MΩ and a low parasitic capacitance of 15 pF.

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