Complex permittivity measurement device, complex permittivity measurement method, and resonator

Abstract

To accurately measure the frequency dependence of the microwave band and the like of the relative permittivity εrt and dielectric tangent tanδt.SOLUTION: A complex permittivity measurement device comprises: a conductive first housing that has a first recess located on a first surface; a conductive second housing that has a second recess located on a second surface; and an electromagnetic field generation unit that forms an electromagnetic field in the first recess and second recess. Both the first recess and second recess have a rectangular parallelepiped shape. When the side along the depth direction of the first recess and second recess is defined as a first side, the shorter side of the two sides orthogonal to the first side is defined as a second side, the longer side is defined as a third side, the dimension of the first side is defined as h / 2, the dimension of the second side is defined as w, and the dimension of the third side is defined as L, the relationship of w<h<L is satisfied. A measurement sample is sandwiched between the first surface and second surface so that the first recess and second recess form a rectangular parallelepiped shape, one or more resonance modes having an electric field parallel to the second side are formed in the first recess and the second recess by the electromagnetic field generation unit, therefore, the resonance frequency and no-load Q of the one or more resonance modes are measured.SELECTED DRAWING: Figure 1

Description

Technical Field

[0001] The present disclosure mainly relates to the complex permittivity (specifically, relative permittivity and dielectric loss tangent) of a dielectric used in a high-frequency region such as a microwave band or a millimeter-wave band, and a measuring apparatus, a measuring method, and a resonator used for measuring the frequency dependence of this complex permittivity.

Background Art

[0002] With the recent progress of mobile communication and sensing technologies, a wide frequency band from the microwave band to the millimeter-wave band, such as 5G communication or in-vehicle millimeter-wave radar, has come to be used. Therefore, accurately measuring the frequency dependence of the relative permittivity and dielectric loss tangent of a dielectric substrate in a wide frequency region from the microwave band to the millimeter-wave band is important for substrate material development and planar circuit design.

[0003] Furthermore, an organic substrate having a glass cloth or molecular orientation is known to have anisotropic relative permittivity and dielectric loss tangent in a direction parallel to the surface of the substrate (hereinafter also referred to as the in-plane direction) and a direction perpendicular to the surface of the substrate (hereinafter also referred to as the normal direction). Therefore, it is necessary to separately measure the relative permittivity and dielectric loss tangent in the in-plane direction and the normal direction of the substrate.

[0004] The relative permittivity ε rn and dielectric loss tangent tanδ n in the normal direction of a dielectric substrate in the microwave band or millimeter-wave band can be measured, for example, by the balanced disk resonator method (see, for example, Non-Patent Documents 1 to 3).

[0005] In this method, since only the excitation and detection of the TM 0m0 resonance mode are selectively performed, the relative permittivity ε rn and dielectric loss tangent tanδ n in a wide frequency region using a plurality of resonance modes, and the measurement of their frequency dependence are possible. However, in this method, the relative permittivity ε rt and dielectric loss tangent tanδt cannot be measured.

[0006] The relative permittivity ε in the in-plane direction of the substrate in the microwave band or millimeter-wave band rt and the dielectric loss tangent tanδ t can be measured by the split-cylinder cavity resonator method (see, for example, Non-Patent Documents 4 and 5) or the cutoff-cylinder waveguide resonator method (see, for example, Non-Patent Document 6).

[0007] In these resonator methods, generally, the relative permittivity ε 011 at a single frequency using the TE rt mode and the dielectric loss tangent tanδ t are measured. This is because the higher-order TE 0mp mode is easily interfered with by unwanted modes in the vicinity.

[0008] That is, in the measurement methods such as Non-Patent Documents 4 to 6 above, the frequency dependence of the relative permittivity ε rt and the dielectric loss tangent tanδ t cannot be measured. Also, in the measurement methods such as Non-Patent Documents 4 to 6 above, at frequencies around 100 GHz and above, the measurement of the relative permittivity ε rt and the dielectric loss tangent tanδ t is difficult.

[0009] On the other hand, according to a Fabry-Perot resonator (also called an open resonator), it has been reported that the relative permittivity ε in the in-plane direction of the dielectric substrate rt and the dielectric loss tangent tanδ t can be measured even at frequencies of 100 GHz or higher (see, for example, Non-Patent Document 7).

[0010] However, in the Fabry-Perot resonator method, since only a part of the electric field distribution of the TEM 00q resonance mode excited between two concave mirrors is distributed in the sample, the relative permittivity ε rt and the dielectric loss tangent tanδ tThere is a problem that the measurement accuracy is insufficient. In addition, since the Fabry-Perot resonator is an open resonator, there is also a problem that it is easily affected by disturbances in the measurement environment such as temperature and humidity.

Prior Art Documents

Non-Patent Documents

[0011]

Non-Patent Document 1

Non-Patent Document 2

Non-Patent Document 3

Non-Patent Document 4

Non-Patent Document 5

Non-Patent Document 6

Non-Patent Document 7

Summary of the Invention

Problems to be Solved by the Invention

[0012] As described above, in the conventional prior art, there has been a problem that it is difficult to accurately measure the frequency dependence of the relative permittivity ε rt and the dielectric loss tangent tanδ t of a dielectric substrate from the microwave band to the millimeter wave band.

[0013] The present disclosure has been devised in view of the problems in such conventional technologies, and an object thereof is to provide a method for highly accurately measuring the frequency dependence of the relative permittivity ε rt and the dielectric loss tangent tan δ t from the microwave band to the millimeter wave band.

Means for Solving the Problems

[0014] The complex permittivity measuring apparatus of the present disclosure includes a first housing, a second housing, and an electromagnetic field forming unit. The conductive first housing has a first surface and a first recess located on the first surface. The conductive second housing has a second surface and a second recess located on the second surface. The electromagnetic field forming unit is provided in at least one of the first recess and the second recess and forms an electromagnetic field in the first recess and the second recess. Both the first recess and the second recess are in the shape of a rectangular parallelepiped. When the side along the depth direction of the first recess and the second recess is defined as the first side, the shorter side of the two sides orthogonal to the depth direction of the first recess and the second recess is defined as the second side, and the longer side is defined as the third side, and the dimension of the first side is h / 2, the dimension of the second side is w, and the dimension of the third side is L, they have a relationship of w < h < L. The plate-shaped measurement sample is sandwiched between the first surface of the first housing and the second surface of the second housing so that the first recess and the second recess are in the shape of a rectangular parallelepiped, and one or more resonance modes having an electric field parallel to the second side are formed in the first recess and the second recess by the electromagnetic field forming unit, thereby measuring the resonance frequency and the unloaded Q of the one or more resonance modes.

Advantages of the Invention

[0015] In the present disclosure, since a relatively strong electric field is generated in the dielectric substrate by inserting the dielectric substrate, which is an object to be measured, parallel to the electric field on the surface where the electric field strength of the TE 10p resonance mode is maximized, the relative permittivity ε rt and the dielectric loss tangent tan δ t can be measured with high accuracy.

[0016] Also, in the present disclosure, TE10p Since the excitation and detection of resonance modes can be selectively performed in a wide frequency range, by using a plurality of resonance modes, the relative permittivity ε of the substrate rt and the dielectric loss tangent tanδ t can be measured. Furthermore, since a high unloaded Q value of about several thousand can be obtained in the present disclosure, the dielectric loss tangent tanδ in a wide frequency range from 10 -5 to 10 -2 is suitable for measuring a dielectric substrate having t .

Brief Description of the Drawings

[0017]

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Embodiments for Carrying Out the Invention

[0018] Hereinafter, embodiments for carrying out a complex dielectric constant measuring device, a complex dielectric constant measuring method, and a resonator according to the present disclosure (hereinafter referred to as "embodiments") will be described in detail with reference to the drawings. Note that the present disclosure is not limited by these embodiments.

[0019] In addition, each embodiment can be appropriately combined within a range that does not conflict with the processing content. Also, in the following embodiments, the same parts are denoted by the same reference numerals, and redundant explanations are omitted.

[0020] Also, note that the drawings are schematic, and it is necessary to be aware that the dimensional relationships between elements, the ratios of the elements, etc. may be different from reality. Furthermore, even between the drawings, there may be parts where the dimensional relationships and ratios between each other are different.

[0021] In addition, in the embodiments described below, expressions such as "identical", "perpendicular", or "parallel" may be used, but these expressions do not necessarily require strict "identity", "perpendicularity", or "parallelism". That is, each of the above expressions shall allow for deviations such as manufacturing accuracy and installation accuracy.

[0022] <Measuring device> First, the configuration of the complex permittivity measuring device 1 according to the embodiment will be described with reference to FIGS. 1 to 8. FIG. 1 is a perspective view showing an example of the configuration of the complex permittivity measuring device 1 and the resonator 2 according to the embodiment. FIG. 2 is a plan view showing an example of the configuration of the complex permittivity measuring device 1 and the resonator 2 according to the embodiment. FIG. 3 is a cross-sectional view taken along the line A-A shown in FIG. 2. FIG. 4 is a cross-sectional view taken along the line B-B shown in FIG. 2.

[0023] As shown in FIGS. 1 to 4, the complex permittivity measuring device 1 according to the embodiment (hereinafter, also simply referred to as the measuring device 1) includes a resonator 2 and an electromagnetic field forming unit 30. Further, the resonator 2 according to the embodiment includes a first housing 10 and a second housing 20.

[0024] In each of the drawings referred to below, for the sake of easy understanding of the description, an orthogonal coordinate system is defined in which the X-axis direction, the Y-axis direction, and the Z-axis direction are orthogonal to each other, the Z-axis direction is the longitudinal direction of the first housing 10 and the second housing 20, and the direction in which the first housing 10 and the second housing 20 face each other is the X-axis direction.

[0025] The first housing 10 is made of a conductive material. The first housing 10 may be made of, for example, a highly conductive metal such as silver or copper, or an alloy containing these highly conductive metals.

[0026] The first housing 10 is, for example, in the shape of a rectangular parallelepiped and has a first surface 11 (see FIG. 3) on the negative X-axis side. In the present disclosure, an example in which the first housing 10 is in the shape of a rectangular parallelepiped is shown, but the present disclosure is not limited to such an example, and any shape may be used as long as the first surface 11, which is a plane, is located on the negative X-axis side.

[0027] Further, the first housing 10 has a first recess 12 and a first spacer 13 on the first surface 11. FIG. 5 is a diagram for explaining the configuration of the first recess 12 of the first housing 10 according to the embodiment. In FIG. 5, the description of the first spacer 13, the through hole 14, and the through hole 15 is omitted.

[0028] As shown in FIG. 5, the first recess 12 has a first side E 11 and a second side E 12 and a third side E 13 and is in the shape of a rectangular parallelepiped. The first side E 11 is a side along the depth direction of the first recess 12 and is a side along the X-axis direction.

[0029] The second side E 12 is the shorter side of the two sides orthogonal to the depth direction of the first recess 12 and is a side along the Y-axis direction. The third side E 13 is the longer side of the two sides orthogonal to the depth direction of the first recess 12 and is a side along the Z-axis direction.

[0030] Returning to the description of FIGS. 1 to 4. The first spacer 13 has dielectric or conductive properties. The first spacer 13 may be made of, for example, a resin material or a metal material. The first spacer 13 is positioned on the first surface 11 so as to surround the first recess 12 and has a thickness t s The first spacer 13 is, for example, in a frame shape in plan view. In the present disclosure, "plan view" means the viewpoint when the measuring device 1 is viewed in the X-axis direction.

[0031] As shown in FIGS. 3 and 4, the first spacer 13 comes into contact with the measurement sample 100 when the measurement sample 100 is sandwiched between the first housing 10 and the second housing 20. The measurement sample 100 is, for example, composed of a plate-shaped dielectric and has a thickness t. Note that the measurement sample 100 may be a non-woven fabric, a woven fabric, a film, or the like.

[0032] As shown in FIG. 3, the first housing 10 has a through hole 14 and a through hole 15. The through hole 14 and the through hole 15 penetrate, for example, between the bottom surface of the first recess 12 and the surface of the first housing 10 on the side opposite to the first surface 11.

[0033] The second housing 20 is made of a conductive material. The second housing 20 may be composed of, for example, a highly conductive metal such as silver or copper, or an alloy containing such a highly conductive metal.

[0034] The second housing 20 has, for example, a rectangular parallelepiped shape and has a second surface 21 on the positive X-axis direction side. In the present disclosure, an example in which the second housing 20 has a rectangular parallelepiped shape is shown, but the present disclosure is not limited to such an example, and any shape may be used as long as the second surface 21, which is a plane, is located on the positive X-axis direction side.

[0035] Further, the second housing 20 has a second recess 22 and a second spacer 23 on the second surface 21. FIG. 6 is a diagram for explaining the configuration of the second recess 22 of the second housing 20 according to the embodiment. Note that the description of the second spacer 23 is omitted in FIG. 6.

[0036] As shown in FIG. 6, the second recess 22 has a first side E 21 and a second side E 22 and a third side E 23 and has a rectangular parallelepiped shape. The first side E 21 is a side along the depth direction of the second recess 22 and is a side along the X-axis direction.

[0037] The second side E 22 is the shorter side of the two sides orthogonal to the depth direction of the second recess 22 and is a side along the Y-axis direction. The third side E 23is the long side of two sides orthogonal to the depth direction of the second recess 22 and is the side along the Z-axis direction.

[0038] In the embodiment, the first recess 12 and the second recess 22 have the same dimensions. That is, in the embodiment, the first side E of the first recess 12 11 and the first side E of the second recess 22 21 have the same dimension h / 2, the second side E of the first recess 12 12 and the second side E of the second recess 22 22 have the same dimension w, and the third side E of the first recess 12 13 and the third side E of the second recess 22 23 have the same dimension L.

[0039] Here, "the same dimensions" or "the same dimension" means that the difference in dimensions is 10% or less. If the difference in dimensions between the first recess 12 and the second recess 22 exceeds 10%, in the measurement process of the complex dielectric constant described later, the symmetry of the electromagnetic field is impaired, and the electromagnetic field radiates in the Y-axis direction and the Z-axis direction through the part holding the measurement sample 100, making it difficult to measure the complex dielectric constant.

[0040] On the other hand, if the difference in dimensions between the first recess 12 and the second recess 22 is 10% or less, it is possible to measure the complex dielectric constant. If the difference in dimensions is 1% or less, it is possible to make the error in the measurement of the complex dielectric constant due to the difference in dimensions generally 1% or less.

[0041] Also, at the corner portions of the first recess 12 and the second recess 22 having a rectangular parallelepiped shape, an R that is difficult to avoid in processing occurs, but such an R is desirably 0.3 mm or less, and 1 mm or less is the allowable range.

[0042] Returning to the description of FIGS. 1 to 4. The second spacer 23 has dielectric or conductive properties. The second spacer 23 may be made of, for example, a resin material or a metal material. The second spacer 23 is positioned on the second surface 21 so as to surround the second recess 22 and has a thickness t s and, for example, has a frame shape in plan view.

[0043] Further, as shown in FIGS. 3 and 4, when the measurement sample 100 is sandwiched between the first housing 10 and the second housing 20, the second spacer 23 comes into contact with the measurement sample 100.

[0044] The measuring device 1 has a moving mechanism (not shown) for moving at least one of the first housing 10 and the second housing 20 described so far. Such a moving mechanism moves the first housing 10 in the X-axis direction, Y-axis direction, and Z-axis direction with respect to the second housing 20, and rotates the first housing 10 about an axis along the X-axis direction.

[0045] In the complex dielectric constant measurement method according to the embodiment (hereinafter, also simply referred to as the measurement method), as shown in FIG. 1, the measurement sample 100 is disposed between the first housing 10 and the second housing 20 spaced apart from each other in the X-axis direction.

[0046] Next, in the measurement method according to the embodiment, as shown in FIGS. 3 and 4, the first housing 10 and the second housing 20 are brought closer to each other, and the measurement sample 100 is sandwiched between the first housing 10 and the second housing 20 so that the measurement sample 100 faces the first surface 11 and the second surface 21.

[0047] At this time, the first housing 10 and the second housing 20 are aligned so that the first recess 12 and the second recess 22 have a substantially integral rectangular parallelepiped shape (a measurement unit including the measurement sample 100 and formed with the first recess 12 and the second recess 22 facing each other), that is, so that the first recess 12 and the second recess 22 overlap without deviation in plan view.

[0048] Continuing the description of the configuration of the measuring device 1. The electromagnetic field forming unit 30 forms an electromagnetic field in the first recess 12 and the second recess 22. For example, as shown in FIGS. 3 and 4, the electromagnetic field forming unit 30 forms a magnetic field H (TE 101 resonance mode) and an electric field E (TE 101 ) in the substantially integral first recess 12 and second recess 22. 101 )

[0049] Here, TE mnpWhen described as a resonance mode, the subscript mnp represents a mode index, where m is the number of changes in the electric field strength in the X-axis direction (i.e., the number of extrema of the electric field strength), n is the number of changes in the electric field strength in the Y-axis direction, and p is the number of changes in the electric field strength in the Z-axis direction.

[0050] The magnetic field H(TE 101 ) is formed, for example, in a single loop shape that encircles the entire outer periphery in the XZ cross-section of the first recess 12 and the second recess 22, as shown in FIG. 3. Also, the electric field E(TE 101 ) is formed along the central axis of the loop of the magnetic field H(TE 101 ).

[0051] Accordingly, in the measuring device 1 according to the embodiment, as shown in FIG. 4 and the like, the electric field E(TE 101 ) when measuring the complex dielectric constant of the measurement sample 100 can be formed along the in-plane direction of the measurement sample 100 inside the measurement sample 100.

[0052] Note that the resonance mode formed in the measuring device 1 according to the embodiment is not limited to the TE 101 resonance mode. FIGS. 7 and 8 are diagrams showing an example of another resonance mode formed in the complex dielectric constant measuring device 1 according to the embodiment.

[0053] As shown in FIG. 7, in the measuring device 1 according to the embodiment, by appropriately controlling the network analyzer of the electromagnetic field forming unit 30, the magnetic field H(TE 102 ) and the electric field E(TE 102 ) of the TE 102 resonance mode can be formed.

[0054] Also, as shown in FIG. 8, in the measuring device 1 according to the embodiment, by appropriately controlling the network analyzer of the electromagnetic field forming unit 30, the magnetic field H(TE 103 ) and the electric field E(TE 103 ) of the TE 103 resonance mode can be formed.

[0055] Thus, in the measuring device 1 according to the embodiment, by appropriately controlling the network analyzer of the electromagnetic field forming unit 30, TE 10p magnetic field H (TE 10p ) and electric field E (TE 10p ) (p is a positive integer) of the resonance mode can be formed.

[0056] Returning to the description of FIGS. 1 to 4. The electromagnetic field forming unit 30 has, for example, a coaxial cable 31 and a coaxial cable 32. As shown in FIG. 3, the coaxial cables 31 and 32 are respectively inserted into the through holes 14 and 15 of the first housing 10.

[0057] The coaxial cable 31 has a transmission antenna (not shown) at the tip on the side of the first recess 12. Further, the coaxial cable 32 has a reception antenna (not shown) at the tip on the side of the first recess 12. These transmission antenna and reception antenna are configured to be movable along the X-axis direction.

[0058] In the electromagnetic field forming unit 30, the network analyzer (not shown) is electrically connected to the transmission antenna and the reception antenna by the coaxial cables 31 and 32. Then, by operating this network analyzer, an electromagnetic field of a desired TE 10p resonance mode is formed in the first recess 12 and the second recess 22.

[0059] The transmission antenna and the reception antenna of the electromagnetic field forming unit 30 may be loop antennas having a loop surface that opens in the Z-axis direction. Thereby, since the magnetic field H (TE 10p ) of the TE 10p resonance mode rotates parallel to the XZ cross section, the excitation and detection of the TE 10p resonance mode are efficiently performed.

[0060] In the examples of FIGS. 1 to 4, an example in which the coaxial cables 31 and 32, the transmitting antenna, and the receiving antenna of the electromagnetic field forming unit 30 are all provided in the first housing 10 has been shown. However, the present disclosure is not limited to such an example. For example, the coaxial cable 31 and the transmitting antenna may be provided in the second housing 20, or the coaxial cable 32 and the receiving antenna may be provided in the second housing 20.

[0061] Here, in the embodiment, when the dimension of the first side E 11 , E 21 of the first recess 12 and the second recess 22 is h / 2, the dimension of the second side E 12 , E 22 is w, and the dimension of the third side E 13 , E 23 is L, it is preferable to have a relationship of w < h < L.

[0062] Thereby, the frequency dependencies of the relative permittivity ε rt in the in-plane direction and the dielectric loss tangent tanδ t of the measurement sample 100, which is a dielectric, can be measured with high accuracy from the microwave band to the millimeter wave band.

[0063] This is because, since the dimensions of the first recess 12 and the second recess 22 have a relationship of w < h < L, TM 110 , TM 11p , TE 11p resonance modes resonate on the high-frequency side of the resonance frequency f 10p of the TE 0 resonance mode. Therefore, the measurement of the complex permittivity by a plurality of TE 10p resonance modes can be performed without being affected by these resonance modes.

[0064] Further, in the embodiment, the dimensions of the first recess 12 and the second recess 22 may have a relationship of 0.1 ≦ w / h ≦ 0.7.

[0065] Since the dimensions of the first recess 12 and the second recess 22 have a relationship of w / h ≦ 0.7, TM 110 , TM 11p , TE 11pThe resonance frequency f of the resonance mode 0 can be shifted to the high-frequency side. As a result, the frequency range in which only a plurality of TE 10p resonance modes to be measured resonate can be widened.

[0066] Also, by having the relationship of 0.1 ≦ w / h for the dimensions of the first recess 12 and the second recess 22, the conductor loss during complex permittivity measurement due to the conductive first housing 10 and second housing 20 can be reduced.

[0067] Therefore, according to the embodiment, by having the relationship of 0.1 ≦ w / h ≦ 0.7 for the dimensions of the first recess 12 and the second recess 22, the relative permittivity ε rt and the dielectric loss tangent tanδ t of the measurement sample 100 in the in-plane direction can be measured with higher accuracy over the frequency range from the microwave band to the millimeter wave band.

[0068] Also, in the embodiment, the dimensions of the first recess 12 and the second recess 22 may have the relationship of L / h ≧ 2, and it is more preferable to have the relationship of L / h ≧ 3. Thereby, in the first recess 12 and the second recess 22, excitation and detection of more TE 10p resonance modes can be efficiently performed.

[0069] Therefore, according to the embodiment, the relative permittivity ε rt and the dielectric loss tangent tanδ t of the measurement sample 100 in the in-plane direction can be measured with higher accuracy over the frequency range from the microwave band to the millimeter wave band.

[0070] Also, in the embodiment, when measuring the complex permittivity of the measurement sample 100, instead of directly sandwiching the measurement sample 100 between the first housing 10 and the second housing 20, it may be sandwiched via the first spacer 13 and the second spacer 23.

[0071] When the measurement sample 100 is directly sandwiched between the first housing 10 and the second housing 20, due to the electric field along the normal direction (X-axis direction) that locally occurs at the contact surface between the first surface 11 or the second surface 21 and the measurement sample 100, a slight and hard-to-detect gap at the contact surface may have an adverse effect on the measurement of the relative permittivity ε rt and the dielectric loss tangent tanδ t There is a risk of affecting the measurement.

[0072] Also, this electric field in the normal direction may mix the influence of the relative permittivity ε rn and the dielectric loss tangent tanδ n of the measurement sample 100 into the measurement of the relative permittivity ε rt and the dielectric loss tangent tanδ t in the in-plane direction. There is a risk of mixing.

[0073] Therefore, in the embodiment, by disposing the first spacer 13 or the second spacer 23 between the measurement sample 100 and the first surface 11 or the second surface 21, the relative permittivity ε rt and the dielectric loss tangent tanδ t in the in-plane direction can be accurately measured.

[0074] Also, in the embodiment, the first spacer 13 and the second spacer 23 may be located away from the first recess 12 and the second recess 22, respectively. In the present disclosure, positioning the first spacer 13 and the second spacer 23 away from the first recess 12 and the second recess 22, respectively, means that it does not affect the electromagnetic field in the TE 10p resonance mode, that is, separating them to such an extent that the resonance waveform does not change.

[0075] Thereby, it is possible to reduce the influence of the first spacer 13 and the second spacer 23 on the electromagnetic field in the TE 10p resonance mode. Therefore, according to the embodiment, the relative permittivity ε rt and the dielectric loss tangent tanδ t in the in-plane direction can be accurately measured.

[0076] For example, in the embodiment, the first spacer 13 and the second spacer 23 may be positioned about 1 mm to 3 mm away from the first recess 12 and the second recess 22, respectively.

[0077] Also, in the embodiment, the first spacer 13 and the second spacer 23 may be made of a dielectric. Thereby, the influence of the materials of the first spacer 13 and the second spacer 23 on the measurement of the relative permittivity ε rt and the dielectric loss tangent tanδ t can be reduced from being mixed in. That is, the influence of the first spacer 13 and the second spacer 23 on the electromagnetic field of the TE 10p resonance mode can be reduced.

[0078] Therefore, according to the embodiment, the relative permittivity ε rt in the in-plane direction and the dielectric loss tangent tanδ t can be accurately measured.

[0079] Note that in the examples of FIGS. 1 to 4, an example in which the first spacer 13 and the second spacer 23 are each in an integral frame shape is shown, but the present disclosure is not limited to such an example. For example, the first spacer 13 or the second spacer 23 may be composed of a plurality of members, or the planar shape may be a rectangular parallelepiped shape, a circular shape, or the like.

[0080] Also, in the examples of FIGS. 1 to 4, by arranging the first spacer 13 and the second spacer 23, the measurement sample 100 is installed at the center between the first housing 10 and the second housing 20, but the present disclosure is not limited to such an example.

[0081] For example, in the present disclosure, the measurement sample 100 is held by a sample holder connected to a vertical stage mounted on the first housing 10, and the sample holder is driven up and down by this vertical stage, and the TE 10p resonance frequency f 0 of the resonance mode has an extreme value, and the measurement sample 100 may be installed at the center between the first housing 10 and the second housing 20. That is, the measurement sample 100 and the sample holder are not in contact with either the first housing 10 or the second housing 20.

[0082] Also, in the embodiment, the transmission antenna and the reception antenna may be arranged at a distance of L / 4 or more from each other, and it is more preferable to arrange the transmission antenna and the reception antenna at a distance of L / 2 or more from each other. Thereby, since direct coupling of electromagnetic waves between the transmission antenna and the reception antenna can be reduced, TE 10p In the measurement of the resonance mode, an increase in the background level (noise level) can be reduced.

[0083] Therefore, according to the embodiment, the relative permittivity ε in the in-plane direction rt and the dielectric loss tangent tanδ t can be accurately measured.

[0084] Also, in the embodiment, the transmission antenna and the reception antenna may be located on a plane passing through the centers of the plurality of second sides E 12 , E 22 .

[0085] Thereby, since the magnetic field formed by the transmission antenna and the reception antenna efficiently couples with the magnetic field of the TE 10p resonance mode, the electromagnetic field of the TE 10p resonance mode is favorably formed. Therefore, according to the embodiment, the relative permittivity ε in the in-plane direction rt and the dielectric loss tangent tanδ t can be accurately measured.

[0086] <Measurement Process> Next, regarding the details of the complex permittivity measurement process using the measurement device 1 according to the embodiment, in addition to FIGS. 1 to 4 described so far, FIGS. 9 to 12 will also be referred to for explanation. FIG. 9 is a flowchart showing the procedure of the complex permittivity measurement process according to the embodiment. FIG. 10 is a flowchart showing the procedure of the concave portion alignment process according to the embodiment.

[0087] As shown in FIG. 9, in the complex permittivity measurement process according to the embodiment (hereinafter, also simply referred to as the measurement process), first, a concave portion alignment process for aligning the relative positions between the first concave portion 12 and the second concave portion 22 is performed (step S101).

[0088] Specifically, in this concave portion alignment process, the measurement sample 100 is not sandwiched, and the first housing 10 and the second housing 20 are separated slightly in the X-axis direction (for example, by about the thickness t of the measurement sample 100).

[0089] Next, in the concave portion alignment process, as shown in FIG. 10, a first alignment process is performed (step S111). Specifically, in this first alignment process, the electromagnetic field forming unit 30 is operated to measure the resonance frequency f 10p of the TE 0 resonance mode, and the relative position in the Z-axis direction between the first concave portion 12 and the second concave portion 22 is aligned so that this resonance frequency f 0 becomes a minimum value.

[0090] Also, in the concave portion alignment process, a second alignment process is performed (step S112). Specifically, in this second alignment process, the electromagnetic field forming unit 30 is operated to measure the resonance frequency f 10p of the TE 0 resonance mode, and the relative position in the Y-axis direction between the first concave portion 12 and the second concave portion 22 is aligned so that this resonance frequency f 0 becomes a maximum value.

[0091] Also, in the concave portion alignment process, a third alignment process is performed (step S113). Specifically, in this third alignment process, the electromagnetic field forming unit 30 is operated to measure the resonance frequency f 10p of the TE 0 resonance mode, and the relative position in the rotational direction between the first concave portion 12 and the second concave portion 22 is aligned so that this resonance frequency f 0 becomes a maximum value.

[0092] Note that the first to third alignment processes in the recess alignment process may be performed in any order. Also, in the recess alignment process, the positions of the transmission antenna and the reception antenna in the X direction are such that the resonance frequency f 0 of the resonator 2 at 21 the peak intensity of the transmission power S is adjusted to a predetermined value (for example, about -50 dB to -30 dB).

[0093] Returning to the description of FIG. 9. Following the recess alignment process described so far, in the measurement process according to the embodiment, a dimension measurement process for measuring the dimensions of each part in the measurement device 1 is performed (step S102).

[0094] Specifically, in this dimension measurement process, without sandwiching the measurement sample 100 and without arranging the first spacer 13 and the second spacer 23, the first surface 11 of the first housing 10 and the second surface 21 of the second housing 20 are brought into contact with each other.

[0095] Next, the electromagnetic field forming unit 30 is operated to measure the resonance frequencies f 101 of the TE 102 resonance mode, the TE 110 resonance mode, and the TM 0 resonance mode, respectively. Then, by performing finite element method analysis using the obtained plurality of resonance frequencies f 0 as input parameters, the dimensions h, w, and L of the hollow portion (the first recess 12 and the second recess 22) are obtained.

[0096] At this time, the resonance frequency f 110 of the TM 0 resonance mode is measured with the loop surfaces of the transmission antenna and the reception antenna parallel to the Z axis.

[0097] Also, in the dimension measurement process, without sandwiching the measurement sample 100 and with the first spacer 13 and the second spacer 23 arranged, the first spacer 13 and the second spacer 23 are brought into contact with each other.

[0098] Next, the electromagnetic field forming unit 30 is operated to perform TE 101Resonance frequency f of the resonance mode 0 is measured. Then, by performing a finite element method analysis with the obtained resonance frequency f 0 as an input parameter, the thickness t s of the first spacer 13 and the second spacer 23 is obtained.

[0099] Following the dimension measurement process described so far, in the measurement process according to the embodiment, a sandwiching process is performed to sandwich the measurement sample 100 between the first surface 11 of the first housing 10 and the second surface 21 of the second housing 20 so that the first recess 12 and the second recess 22 have a rectangular parallelepiped shape (step S103). Note that the recess alignment process and the dimension measurement process do not need to be performed prior to each complex dielectric constant measurement process. For example, they may be performed once prior to a series of multiple measurement processes.

[0100] Next, an antenna alignment process is performed to align the positions of the transmitting antenna and the receiving antenna of the electromagnetic field forming unit 30 (step S104). Specifically, in this antenna alignment process, the electromagnetic field forming unit 30 is operated to measure the resonance frequency f 10p of the TE 0 resonance mode.

[0101] Then, the positions of the transmitting antenna and the receiving antenna in the X-axis direction are aligned so that the peak intensity of the transmitted power S 0 of the resonator 2 at this resonance frequency f 21 becomes a predetermined value (for example, about -50 dB to -30 dB).

[0102] Next, a measurement process is performed to control the electromagnetic field forming unit 30 to measure the resonance frequency f 10p of one or more TE 0 resonance modes and the unloaded Q (step S105). In this measurement process, for example, from the measured values of the resonance frequency f 0 , the power half-width Δf, and the insertion loss IL 0 at the resonance frequency f 0 , the unloaded Q (Q u ) is obtained using the following equation (1). Note that the insertion loss IL 0 is at the resonance frequency f0 The transmission power S of the resonator 2 in 21 is synonymous with.

[0103]

Number

[0104] Next, based on the measured resonance frequency f 0 and the unloaded Q, a calculation process is performed to calculate the complex permittivity in the in-plane direction of the measurement sample 100, specifically, the relative permittivity ε rt in the in-plane direction and the dielectric loss tangent tanδ t (step S106), and a series of measurement processes of the complex permittivity is terminated.

[0105] Here, an example of such a calculation process will be described with reference to FIGS. 11 and 12. FIG. 11 is a diagram showing an approximate analysis model for explaining the measurement process of the complex permittivity according to the embodiment. In the approximate model shown in FIG. 11, the measurement sample 100 is inserted into the conductor constituting the resonator 2.

[0106] FIG. 12 is a diagram showing an accurate analysis model for explaining the measurement process of the complex permittivity according to the embodiment. In the analysis model shown in FIG. 12, since the conductor constituting the resonator 2 is integrated (short-circuited) at a position where the electromagnetic field at the edge of the measurement sample 100 is sufficiently attenuated, the electromagnetic field of the resonator 2 can be analyzed more accurately.

[0107] First, using the characteristic equation of the TE 10p resonance mode represented by the following equation (2) that holds in the approximate model of FIG. 11, an approximate value of the relative permittivity ε 0 in the in-plane direction is calculated from the measured value of the resonance frequency f rt .

[0108]

Number

[0109] However, in this case, the following equations (3) and (4) hold.

[0110]

Number

[0111]

Number

[0112] Here, k x1 , k x2 are the wavenumbers in the X-axis direction of the dielectric region (the region of the measurement sample 100) and the air region (the region of the hollow part other than the measurement sample 100), respectively, and k z (=pπ / L) is the wavenumber in the Z-axis direction, and c is the speed of light.

[0113] Next, using an analysis model that includes the effect of the edge part of the measurement sample 100 shown in FIG. 12, an approximate relative permittivity ε rt around the relative permittivity ε rt and the relationship between the resonance frequency f 0 are obtained. Then, from this relational expression and the measured value of the resonance frequency f 0 , an accurate relative permittivity ε rt is obtained.

[0114] After obtaining the relative permittivity ε rt in the in-plane direction, the dielectric loss tangent tanδ t in the in-plane direction is calculated by the following equation (5) from the measured value of the unloaded Q 10p of the TE u resonance mode.

[0115]

Number

[0116] Here, P e is the electric field energy concentration ratio of the measurement sample 100, G is the shape factor of the resonator 2, and σ 0is the conductivity of 5.8×10 of the international standard soft copper specified in IEC60028 at 20 °C 7 [S / m], σ r is the effective relative conductivity of the recess surface, and μ is the magnetic permeability of the conductor. The electric field energy concentration ratio P e and the shape factor G are calculated by using the accurate analysis model shown in Fig. 12 and by means of the finite element method, etc., for example, by the method disclosed in Non-Patent Document 8 below.

[0117] Non-Patent Document 8: N. Hirayama, A. Nakayama, Y. Hiromichi, T. Shimizu, and Y. Kogami, “Measurement Technique for Interface and Surface Conductivities at Millimeter-Wave Frequencies Using Dielectric Rod Resonator Excited by Nonradiative Dielectric Waveguide,” IEEE Trans. Microw. Theory Techn., vol. 70, no. 5, pp. 2750-2761, May. 2022.

[0118] Next, an example of the measurement process of the complex permittivity performed by the inventor of the present application will be described below. First, the measurement device 1 having the configuration shown in FIGS. 1 to 6 was prepared. As the network analyzer of the electromagnetic field forming unit 30, N5291A manufactured by Keysight was used.

[0119] The resonator 2 was excited and detected by a transmitting antenna and a receiving antenna each having a loop surface of φ0.5 mm formed at the tips of the coaxial cables 31 and 32 of φ0.9 mm. The insertion depths of the transmitting antenna and the receiving antenna into the through holes 14 and 15 were such that the insertion loss IL 0 was adjusted to be about -50 dB to -30 dB.

[0120] In this example, in order to evaluate the complex relative permittivity in the range of 20 GHz to 60 GHz, the cross-sectional dimensions of the resonator 2 were designed to be h = 5.690 mm and w = 2.845 mm (i.e., w / h = 0.5). The length of the resonator 2 was designed to be L = 20.000 mm (L / h = 3.5) so that the interval between the TE 10p resonance modes is about 5 GHz.

[0121] Here, the basic design method of the dimensions w, h, and L of the hollow portion (the first recess 12 and the second recess 22) in the resonator 2 will be described in detail. In the resonator 2 of the present disclosure, when the measurement sample 100 as the object to be measured and the first spacer 13 and the second spacer 23 are removed and the two housings are brought into contact without displacement, a rectangular parallelepiped cavity resonator 2 having the dimensions w, h, and L of the hollow portion is formed.

[0122] In this case, the resonance frequency f mnp of the TE mnp resonance mode and the TM 0 resonance mode is calculated by the following formula (6), where c is the speed of light.

[0123]

Equation

[0124] The resonance frequency f 101 of the TE 0 resonance mode, which is the lower limit of the measurement of the frequency dependence of the complex permittivity, is determined by h and L because the mode index n is 0 as shown in the above formula (6). However, under the condition that h < L, especially when L is several times larger than h, it is approximately c / 2h. Therefore, the approximate value of h is designed from the desired lower limit of the measurement of the frequency dependence of the complex permittivity.

[0125] For example, when h = 5.690 mm, c / 2h = 26 GHz is the approximate value of the lower limit of the measurement of the frequency dependence of the complex permittivity. Actually, by inserting a dielectric substrate between the split surfaces of the resonator 2, the relative permittivity ε rt and the thickness t of the measurement sample 100 cause the resonance frequency f 101 of the TE0 Since it decreases, the design value of h is determined in consideration of this fact.

[0126] Next, the design of w / h will be described. FIG. 13 shows the TE 10p resonance frequency f of the resonance mode of the resonator 2 according to the embodiment 0 and the TM 110 and the TM 11p and the TE 11p resonance frequency f of the resonance mode 0 plotted as functions of w / h.

[0127] Note that the TM 110 and the TM 11p and the TE 11p resonance modes are resonance modes that are obstructive to complex permittivity measurement. Also, the TM 11p resonance mode and the TE 11p resonance mode are degenerate and have the same resonance frequency f 0 .

[0128] Also, in the state where the dividing surface of the hollow portion is closed, a TE 110 and a TE 11p resonance mode having a resonance frequency f 11p lower than the TM 0 resonance mode appears as an interfering mode, but since these are resonance modes in which current flows across the dividing surface, they disappear the moment the dividing surface is opened. Therefore, these resonance modes are not shown in FIG. 13. 20p resonance mode or a TE 01p resonance mode

[0129] Also, FIG. 13 shows the resonance frequency f 0 as a function of w / h when h = 5.690 mm, L = 20.000 mm, and L / h = 3.5.

[0130] From this FIG. 13, in the frequency range where there are no resonances of various interfering modes and only the TE 10p resonance mode resonates (hereinafter, the TE 10p(Also referred to as the selective resonance region).) The upper limit can be seen to increase by reducing w / h. Therefore, in principle, TE 10p It is possible to design w / h such that the upper limit of the selective resonance region becomes the desired frequency.

[0131] On the other hand, if w / h is made too small, it will cause an increase in the conductor loss of the resonator 2, which is disadvantageous for measuring the dielectric tangent tanδ of the low-loss measurement sample 100. t In the embodiment, considering these factors, it is desirable that w / h is in the range of 0.1 to 0.7.

[0132] And in this example, the resonator 2 was fabricated with w / h = 0.5. In this case, TE 10p The upper limit of the selective resonance region is 59 GHz.

[0133] Next, the design of L will be described. FIG. 14 shows the frequency interval of the TE 10p resonance mode when the dimension L is changed in the resonator 2 according to the embodiment. Note that the frequency interval of the TE 10p resonance mode refers to the difference between the resonance frequency f 10(p+1) of the TE 0 resonance mode and the resonance frequency f 10p of the TE 0 resonance mode.

[0134] Also, FIG. 14 shows the frequency interval of the TE 10p resonance mode as a function of the mode index p when h = 5.690 mm.

[0135] From this FIG. 14, it can be seen that by increasing L, the frequency interval of the TE 10p resonance mode can be reduced. In this example, the resonator 2 was fabricated with L = 20.000 mm such that the frequency interval of the TE 10p resonance mode is about 6 GHz.

[0136] On the other hand, when L is made excessively long, the adjacent TE 10pThere is a risk that interference occurs between resonance modes, making accurate measurement difficult. Especially for a measurement sample with a large dielectric loss tangent tanδ t , specifically, for a measurement sample with a dielectric loss tangent tanδ t of 1×10 -2 or more, in the measurement sample 100, the unloaded Q of the TE 10p resonance mode becomes as small as about 100, and the peak width of the resonance peak spreads, making it easier for adjacent TE 10p resonance modes to interfere with each other.

[0137] Taking the TE 104 resonance mode of this example as an example, since the resonance frequency f 0 is about 40 GHz, when the unloaded Q is about 100, the width of the resonance peak spreads to about 400 MHz. In this case, if the frequency interval between adjacent TE 105 resonance modes is 2 GHz or more, interference can be avoided, so it can be seen from FIG. 14 that the upper limit of L is about 30 mm.

[0138] Note that if it is limited to the measurement sample 100 with a dielectric loss tangent tanδ t of 1×10 -2 or less, the upper limit of L may be even larger.

[0139] As described above, the method for measuring the frequency dependence of the complex dielectric constant by the resonator 2 of the present disclosure can measure in a frequency region (TE 10p selective resonance region) that is not affected by interference from unnecessary adjacent modes, the upper and lower limits of the measurement frequency interval (the frequency interval between adjacent TE 10p resonance modes), and the number of TE 10p resonance modes appearing in the TE 10p selective resonance region can be adjusted somewhat freely by appropriately designing w, h, and L, which is an excellent feature.

[0140] FIG. 15 is a diagram showing the resonance frequency f 0 of each resonance mode as a function of w / h when h = 5.69 mm, L = 17.07 mm, and L / h = 3. As shown in FIG. 15, in the resonator 2 that satisfies h and L under the above conditions, at w / h = 0.5, five TE10p The resonance mode is TE 10p It can be seen that it appears in the selected resonance region.

[0141] Also, in the resonator 2 that satisfies h and L under the above conditions, at w / h = 0.7, TE 10p The selected resonance region is in the range of 28 GHz to 45 GHz, and four TE 10p The resonance mode is TE 10p It can be seen that it appears in the selected resonance region.

[0142] Fig. 16 is a diagram showing the resonance frequency f 0 of each resonance mode as a function of w / h when h = 5.69 mm, L = 11.38 mm, and L / h = 2. As shown in Fig. 16, in the resonator 2 that satisfies h and L under the above conditions, at w / h = 0.5, three TE 10p The resonance mode is TE 10p It can be seen that it appears in the selected resonance region.

[0143] Also, in the resonator 2 that satisfies h and L under the above conditions, at w / h = 0.7, TE 10p The selected resonance region is in the range of 30 GHz to 45 GHz, and two TE 10p The resonance mode is TE 10p It can be seen that it appears in the selected resonance region.

[0144] Fig. 17 is a diagram showing the resonance frequency f 0 of each resonance mode as a function of w / h when h = 5.69 mm, L = 5.69 mm, and L / h = 1. As shown in Fig. 17, in the resonator 2 that satisfies h and L under the above conditions, at w / h = 0.5, one TE 10p The resonance mode is TE 10p It can be seen that it appears in the selected resonance region.

[0145] Also, in the resonator 2 that satisfies h and L under the above conditions, at w / h = 0.7, TE 10p The selected resonance region is in the range of 38 GHz to 45 GHz, and one TE 10p The resonance mode is TE 10p It can be seen that it appears in the selected resonance region.

[0146] FIG. 18 is a diagram showing the resonance frequency f of each resonance mode when h = 2.00 mm, L = 6.00 mm, and L / h = 3. 0 It is a diagram showing f as a function of w / h. As shown in FIG. 18, in the resonator 2 satisfying h and L under the above conditions, at w / h = 0.5, five TE 10p resonance modes appear in the TE 10p selective resonance region.

[0147] Also, in the resonator 2 satisfying h and L under the above conditions, at w / h = 0.7, the TE 10p selective resonance region is in the range of 80 GHz to 130 GHz, and four TE 10p resonance modes appear in the TE 10p selective resonance region.

[0148] FIG. 19 is a diagram showing the resonance frequency f of each resonance mode when h = 2.00 mm, L = 4.00 mm, and L / h = 2. 0 It is a diagram showing f as a function of w / h. As shown in FIG. 19, in the resonator 2 satisfying h and L under the above conditions, at w / h = 0.5, three TE 10p resonance modes appear in the TE 10p selective resonance region.

[0149] Also, in the resonator 2 satisfying h and L under the above conditions, at w / h = 0.7, the TE 10p selective resonance region is in the range of 80 GHz to 130 GHz, and two TE 10p resonance modes appear in the TE 10p selective resonance region.

[0150] FIG. 20 is a diagram showing the resonance frequency f of each resonance mode when h = 2.00 mm, L = 2.00 mm, and L / h = 1. 0 It is a diagram showing f as a function of w / h. As shown in FIG. 20, in the resonator 2 satisfying h and L under the above conditions, at w / h = 0.5, one TE 10p resonance mode appears in the TE 10p selective resonance region.

[0151] Also, in the resonator 2 that satisfies h and L of the above conditions, at w / h = 0.7, TE 10p The selected resonance region is in the range of 110 GHz to 130 GHz, and one TE 10p resonance mode is TE 10p It can be seen that it appears in the selected resonance region.

[0152] FIG. 21 is a diagram showing the resonance frequency f 0 of each resonance mode as a function of w / h when h = 0.70 mm, L = 2.10 mm, and L / h = 3. As shown in FIG. 21, in the resonator 2 that satisfies h and L of the above conditions, at w / h = 0.5, five TE 10p resonance modes are TE 10p It can be seen that it appears in the selected resonance region.

[0153] Also, in the resonator 2 that satisfies h and L of the above conditions, at w / h = 0.7, TE 10p The selected resonance region is in the range of 220 GHz to 370 GHz, and four TE 10p resonance modes are TE 10p It can be seen that it appears in the selected resonance region.

[0154] FIG. 22 is a diagram showing the resonance frequency f 0 of each resonance mode as a function of w / h when h = 0.70 mm, L = 1.40 mm, and L / h = 2. As shown in FIG. 22, in the resonator 2 that satisfies h and L of the above conditions, at w / h = 0.5, three TE 10p resonance modes are TE 10p It can be seen that it appears in the selected resonance region.

[0155] Also, in the resonator 2 that satisfies h and L of the above conditions, at w / h = 0.7, TE 10p The selected resonance region is in the range of 240 GHz to 370 GHz, and two TE 10p resonance modes are TE 10p It can be seen that it appears in the selected resonance region.

[0156] FIG. 23 shows the resonance frequency f of each resonance mode when h = 0.70 mm, L = 0.70 mm, and L / h = 1. 0 This is a diagram showing it as a function of w / h. As shown in FIG. 23, in the resonator 2 satisfying h and L under the above conditions, at w / h = 0.5, one TE 10p resonance mode appears in the TE 10p selective resonance region.

[0157] Also, in the resonator 2 satisfying h and L under the above conditions, at w / h = 0.7, the TE 10p selective resonance region is in the range of 300 GHz to 370 GHz, and one TE 10p resonance mode appears in the TE 10p selective resonance region.

[0158] As shown so far, the inventor of the present application has found that when L / h ≥ 2 under the condition of 0.1 ≤ w / h ≤ 0.7, two or more TE 10p resonance modes appear in the TE 10p selective resonance region.

[0159] The description of an example of the complex dielectric constant measurement process performed by the inventor of the present application will be continued. First, as shown in FIG. 9, the inventor performed the concave portion alignment process (step S101). Since the details of this process have been described above, the description will be omitted.

[0160] Next, as the dimension measurement process (step S102 in FIG. 9), the inventor measured the dimensions w, h, L of the resonator 2, the effective conductivity σ of the resonator 2 r , and the thickness t with the first spacer 13 and the second spacer 23. s At this time, the inventor removed the measurement sample 100, the first spacer 13, and the second spacer 23 from the resonator 2, brought the first housing 10 and the second housing 20 into contact without displacement, and in the state of forming the empty resonator 2, the TE

[0161] resonance mode, the TE 101 resonance mode, and the resonance frequency f of the TM 102 resonance mode and the TM 110 resonance mode0 Each was measured five times, and the dimensions h, w, and L were obtained by finite element method analysis.

[0162] However, TM 110 When measuring the resonance mode, the loop surfaces of the transmitting antenna and the receiving antenna were made parallel to the Z-axis direction to facilitate excitation of the TM 110 resonance mode.

[0163] In addition, the inventor arranged the first spacer 13 and the second spacer 23 in the resonator 2, brought the first housing 10 and the second housing 20 into contact without misalignment, and in the state where the empty resonator 2 was formed, TE 101 The resonance frequency f of the resonance mode 0 was measured five times, and the thickness t of the first spacer 13 and the second spacer 23 s was obtained by finite element method analysis. These measurement results are shown in FIG. 24.

[0164] Furthermore, the inventor measured the unloaded Qu of the TE 10p resonance modes (p = 1 to 8) of the empty resonator 2 in which the first spacer 13 and the second spacer 23 were arranged five times each, and the effective relative conductivity σ of the inner walls of the first recess 12 and the second recess 22 r was obtained by finite element method analysis. FIG. 25 shows these results and the fitting line of the effective relative conductivity σ r with respect to the frequency.

[0165] The inventor used this fitting line of the effective relative conductivity σ r to obtain the dielectric loss tangent tan δ of the measurement sample 100. The effective relative conductivity σ t when obtaining the dielectric loss tangent tan δ t The uncertainty of was obtained from the uncertainty of the fitting. r

[0166] Next, as shown in FIG. 9, the inventor performed the clamping process (step S103) and the antenna alignment process (step S104) in order. Since the details of these processes have been described above, the description is omitted.

[0167] Next, the inventor performed measurement processing (step S105 in FIG. 9). FIG. 26 is a diagram showing an example of the result of the measurement processing according to the embodiment. Specifically, a single crystal sapphire substrate perpendicular to the c-axis with a thickness of 0.318 mm was inserted into the resonator 2 through the first spacer 13 and the second spacer 23 with a thickness of t s to show the measurement result of the frequency response of the transmitted power S 21 in the resonator 2.

[0168] As shown in FIG. 26, in the technology of the present disclosure, in the frequency range of 20 GHz to 50 GHz, it was found that the TE 10p resonance modes with p = 1 to 10 resonate without interference modes.

[0169] Next, the inventor performed calculation processing (step S106 in FIG. 9). Specifically, from the resonance frequencies f 0 and the unloaded Q u measured from the measurement results of each resonance peak of the frequency response as shown in FIG. 26, the relative permittivity ε rt and the dielectric loss tangent tan δ t of the measurement sample 100 were calculated by the method described above.

[0170] In this example, for a single crystal sapphire substrate perpendicular to the c-axis with a thickness of 0.318 mm, a polytetrafluoroethylene (PTFE) substrate with a thickness of 0.202 mm, and a liquid crystal polymer (LCP) substrate with a thickness of 0.100 mm, the resonance frequencies f 0 and the unloaded Q u were measured using the measuring device 1, and the relative permittivity ε rt and the dielectric loss tangent tan δ t of each sample were calculated from this measurement result. The calculation results are shown in FIG. 27.

[0171] Note that in FIG. 27, as a reference example, the resonance frequencies f 0 and the unloaded Q u measured by the conventional split cylindrical cavity resonator are also shown, along with the calculation results of the relative permittivity ε rt and the dielectric loss tangent tan δ t of each sample.

[0172] As shown in Fig. 27, the relative permittivity ε of each sample rt shows a value that is almost constant with respect to the frequency. Also, the dielectric loss tangent tanδ of the sapphire substrate and the LCP substrate t shows a tendency to increase with the frequency. Further, the dielectric loss tangent tanδ of the PTFE substrate t shows a value that is almost constant with respect to the frequency.

[0173] Also, as shown in Fig. 27, even for a low-loss material such as sapphire or PTFE, the relative permittivity ε rt and the dielectric loss tangent tanδ t calculated by the present disclosed technique are in agreement with the relative permittivity ε rt and the dielectric loss tangent tanδ t calculated by the conventional split cylindrical resonator.

[0174] This result shows the high effectiveness of the technique of the present disclosure in the measurement of the frequency dependence of the relative permittivity ε rt and the dielectric loss tangent tanδ t .

[0175] Note that the uncertainties Δη rt and Δtanδ t of the measured values of the relative permittivity ε rt and Δtanδ t (error bars in Fig. 27) were calculated by the following equations (7) and (8).

[0176]

Equation

[0177]

Equation

[0178] However, Δη rt,f , Δη rt,t , Δη rt,h , Δη rt,w , Δη rt,L and Δηrt,ts is the permittivity ε 0 due to the uncertainties in the resonance frequency f s , thickness t, dimensions h, w, L, and thickness t rt . Also, Δtanδ t,Q and Δtanδ t,σ are the uncertainties in the dielectric tangent tanδ u due to the uncertainties in the no-load Q r and the effective conductivity σ t , respectively.

[0179] <Another Embodiment> Next, another embodiment will be described with reference to FIGS. 28 and 29. FIG. 28 is a cross-sectional view showing an example of the configuration of a complex permittivity measuring apparatus 1 and a resonator 2 according to another embodiment, and corresponds to FIG. 3 of the embodiment.

[0180] As shown in FIG. 28, in another embodiment, the configuration of the electromagnetic field forming unit 30 is different from that of the above-described embodiment. Specifically, in another embodiment, the electromagnetic field forming unit 30 may include a first waveguide 33 and a second waveguide 34.

[0181] The first waveguide 33 transmits electromagnetic waves to the first recess 12 or the second recess 22. The second waveguide 34 receives electromagnetic waves from the first recess 12 or the second recess 22.

[0182] And, in another embodiment, by using these first waveguide 33 and second waveguide 34 to perform excitation and detection of the resonator 2, excitation and detection of the resonator 2 can be performed well even in a frequency band of 140 GHz or higher.

[0183] In another embodiment, for example, the space between the first waveguide 33 and the first recess 12 may be connected via a metal plate 35 and a slit 36, and the space between the second waveguide 34 and the first recess 12 may be connected via a metal plate 37 and a slit 38. The metal plates 35 and 37 are each configured to be movable along the Z-axis direction by a moving mechanism (not shown).

[0184] In the measuring device 1 according to another embodiment, by moving the metal plates 35 and 37 in the Z-axis direction, the widths of the slits 36 and 38 are adjusted, and the coupling amount of the TE 10p resonance mode can be adjusted. In the present disclosure, either the first waveguide 33 or the second waveguide 34 for excitation and detection may be provided in the second housing 20 together with the metal plate and the slit.

[0185] FIG. 29 is a diagram showing an example of the result of measurement processing according to another embodiment. Note that FIG. 29 shows the transmission power S in the resonator 2 according to another embodiment 21 which is the result of simulation by the finite element method. The simulation conditions are as follows. ·h = 0.7 mm, w = 0.432 mm, L = 10 mm, h / w = 0.62, L / h = 14.3 ·Width of the waveguide in the Z-axis direction = 0.864 mm, width of the waveguide in the Y-axis direction = 0.432 mm ·Relative permittivity ε of the measurement sample rt = 2.05, dielectric loss tangent tanδ t = 0.0002, t = 0.1 mm ·Effective relative conductivity σ of the resonator r = 100% ·Thickness t of the first spacer and the second spacer s = 0.05 mm

[0186] As shown in FIG. 29, in the measuring device 1 according to another embodiment, excitation and detection of the TE 10p resonance mode can be performed in the frequency range of 220 GHz to 330 GHz. That is, in the measuring device 1 according to another embodiment, in the frequency range of 140 GHz or higher, the relative permittivity ε rt and the dielectric loss tangent tanδ t of the measurement sample 100 can be measured.

[0187] As described above in detail, the present disclosure is not limited to the above-described embodiments, and various changes and improvements can be made without departing from the gist of the present disclosure.

[0188] The embodiments disclosed this time should be considered illustrative in all respects and not restrictive. Indeed, the above-described embodiments can be embodied in various forms. Also, the above embodiments may be omitted, replaced, or changed in various forms without departing from the scope and spirit of the appended claims.

[0189] Note that the present technology can adopt the following configurations. (1) A conductive first housing having a first surface and a first recess located on the first surface, A conductive second housing having a second surface and a second recess located on the second surface, An electromagnetic field forming unit provided in at least one of the first recess and the second recess to form an electromagnetic field in the first recess and the second recess, Comprising, Both the first recess and the second recess are in the shape of a rectangular parallelepiped. Taking the side along the depth direction of the first recess and the second recess as the first side, and taking the shorter side of the two sides orthogonal to the depth direction of the first recess and the second recess as the second side and the longer side as the third side. When the dimension of the first side is h / 2, the dimension of the second side is w, and the dimension of the third side is L, they have the relationship of w < h < L, The first surface of the first housing and the second surface of the second housing sandwich a plate-shaped measurement sample so that the first recess and the second recess are in the shape of a rectangular parallelepiped, By forming one or more resonance modes having an electric field parallel to the second side in the first recess and the second recess by the electromagnetic field forming unit, measuring the resonance frequency and the unloaded Q of the one or more resonance modes A complex permittivity measuring device. (2) w and h satisfy 0.1 ≦ w / h ≦ 0.7 The complex permittivity measuring device according to (1) above. (3) L and h satisfy L / h ≧ 2 The complex permittivity measuring device according to (1) or (2) above. (4) By forming a plurality of resonance modes having an electric field parallel to the second side in the first concave portion and the second concave portion by the electromagnetic field forming unit, the resonance frequencies and the unloaded Q of the plurality of resonance modes are measured. The complex permittivity measuring device according to any one of (1) to (3) above. (5) The first housing is located so as to surround the first concave portion, and has a first spacer made of a dielectric or a conductor that contacts the measurement sample when sandwiching the measurement sample. The second housing is located so as to surround the second concave portion, and has a second spacer made of a dielectric or a conductor that contacts the measurement sample when sandwiching the measurement sample. The complex permittivity measuring device according to any one of (1) to (4) above. (6) The first spacer and the second spacer are located away from the first concave portion and the second concave portion, respectively. The complex permittivity measuring device according to (5) above. (7) The electromagnetic field forming unit has a transmission antenna and a reception antenna. The transmission antenna and the reception antenna are located on a plane passing through the centers of the plurality of second sides. The complex permittivity measuring device according to any one of (1) to (6) above. (8) The electromagnetic field forming unit has a transmission antenna and a reception antenna. The transmission antenna and the reception antenna are configured to be movable in a direction perpendicular to the in-plane direction of the measurement sample. The complex permittivity measuring device according to any one of (1) to (7) above. (9) The electromagnetic field forming unit has a first waveguide that transmits electromagnetic waves to the first concave portion or the second concave portion, and a second waveguide that receives electromagnetic waves from the first concave portion or the second concave portion. The complex permittivity measuring device according to any one of (1) to (6) above. (10) A conductive first housing having a first surface and a first recess located on the first surface, A conductive second housing having a second surface and a second recess located on the second surface, An electromagnetic field forming unit provided in at least one of the first recess and the second recess, and forming an electromagnetic field in the first recess and the second recess, Comprising, Both the first recess and the second recess are rectangular parallelepiped-shaped. A side along the depth direction of the first recess and the second recess is defined as the first side, and among the two sides orthogonal to the depth direction of the first recess and the second recess, the shorter side is defined as the second side and the longer side is defined as the third side. When the dimension of the first side is h / 2, the dimension of the second side is w, and the dimension of the third side is L, using a measuring device having a relationship of w < h < L, A sandwiching step of sandwiching a plate-shaped measurement sample between the first surface of the first housing and the second surface of the second housing so that the first recess and the second recess are rectangular parallelepiped-shaped, A forming step of forming one or more resonance modes having an electric field parallel to the second side in the first recess and the second recess by the electromagnetic field forming unit, A measuring step of measuring the resonance frequency and the unloaded Q of the one or more resonance modes, A calculating step of calculating the complex permittivity in the in-plane direction of the measurement sample based on the resonance frequency and the unloaded Q measured in the measuring step, A method for measuring complex permittivity including. (11) Before the sandwiching step, further including a recess alignment step of aligning the relative positions between the first recess and the second recess, The method for measuring complex permittivity according to (10) above. (12) The recess alignment step includes a first alignment step of aligning the positions in the direction parallel to the third side between the first recess and the second recess so that the resonance frequency of the one or more resonance modes having an electric field parallel to the second side becomes a minimum value, The method for measuring complex permittivity according to (11) above. (13) The recess alignment step includes a second alignment step of aligning the positions in the direction parallel to the second side between the first recess and the second recess such that the resonance frequencies of the one or more resonance modes having an electric field parallel to the second side reach maximum values. The method for measuring complex permittivity according to (11) or (12) above. (14) The recess alignment step includes a third alignment step of aligning the positions in the rotational direction between the first recess and the second recess such that the resonance frequencies of the one or more resonance modes having an electric field parallel to the second side reach maximum values. The method for measuring complex permittivity according to any one of (11) to (13) above. (15) The electromagnetic field forming unit includes a transmitting antenna and a receiving antenna configured to be movable in a direction perpendicular to the in-plane direction of the measurement sample. Further including an antenna alignment step that is performed between the clamping step and the measurement step and aligns the positions of the transmitting antenna and the receiving antenna. The method for measuring complex permittivity according to any one of (10) to (14) above. (16) A conductive first housing having a first surface and a first recess located on the first surface. A conductive second housing having a second surface and a second recess located on the second surface. Comprising: Both the first recess and the second recess are in the shape of a rectangular parallelepiped. A side along the depth direction of the first recess and the second recess is defined as the first side, and among the two sides orthogonal to the depth direction of the first recess and the second recess, the shorter side is defined as the second side and the longer side is defined as the third side. When the dimension of the first side is h / 2, the dimension of the second side is w, and the dimension of the third side is L, they have the relationship of w < h < L. The first surface of the first housing and the second surface of the second housing sandwich a plate-shaped measurement sample so that the first recess and the second recess are in the shape of a rectangular parallelepiped, and the electromagnetic field forming unit is configured to be able to form one or more resonance modes having an electric field parallel to the second side in the first recess and the second recess. Resonator. (17) The w and the h satisfy 0.1 ≦ w / h ≦ 0.7 The resonator according to (16). (18) The L and the h satisfy L / h ≧ 2 The resonator according to (16) or (17). (19) The first housing is positioned to surround the first recess, and has a first spacer made of a dielectric or a conductor that contacts the measurement sample when sandwiching the measurement sample. The second housing is positioned to surround the second recess, and has a second spacer made of a dielectric or a conductor that contacts the measurement sample when sandwiching the measurement sample. The resonator according to any one of (16) to (18). (20) The first spacer and the second spacer are respectively positioned away from the first recess and the second recess. The resonator according to (19).

Explanation of reference numerals

[0190] 1 Measuring device 2 Resonator 10 First housing 11 First surface 12 First recess 13 First spacer 20 Second housing 21 Second surface 22 Second recess 23 Second spacer 30 Electromagnetic field forming unit 33 First waveguide 34 Second waveguide 100 Measurement sample E 11 、E 21 First side E 12 、E 22 Second side E 13 、E 23 Third side

Claims

1. A conductive first housing having a first surface and a first recess located on the first surface, A conductive second housing having a second surface and a second recess located on the second surface, An electromagnetic field forming unit provided in at least one of the first recess and the second recess, for forming an electromagnetic field in the first recess and the second recess, Comprising, Both the first recess and the second recess are in the shape of a rectangular parallelepiped. A side along the depth direction of the first recess and the second recess is defined as the first side, and among the two sides orthogonal to the depth direction of the first recess and the second recess, the shorter side is defined as the second side and the longer side is defined as the third side. When the dimension of the first side is h / 2, the dimension of the second side is w, and the dimension of the third side is L, they have the relationship of w < h < L, The first surface of the first housing and the second surface of the second housing sandwich a plate-like measurement sample so that the first recess and the second recess are in the shape of a rectangular parallelepiped, By forming one or more resonance modes having an electric field parallel to the second side in the first recess and the second recess by the electromagnetic field forming unit, measuring the resonance frequency and the unloaded Q of the one or more resonance modes A complex permittivity measuring device.

2. Wherein w and h satisfy 0.1 ≦ w / h ≦ 0.7 The complex permittivity measuring device according to Claim 1.

3. Wherein L and h satisfy L / h ≧ 2 The complex permittivity measuring device according to Claim 2.

4. By forming a plurality of resonance modes having an electric field parallel to the second side in the first recess and the second recess by the electromagnetic field forming unit, measuring the resonance frequency and the unloaded Q of the plurality of resonance modes The complex permittivity measuring device according to any one of Claims 1 to 3.

5. The first housing has a dielectric or conductive first spacer that is located so as to surround the first recess and contacts the measurement sample when sandwiching the measurement sample, The second housing has a dielectric or conductive second spacer that is located so as to surround the second recess and contacts the measurement sample when sandwiching the measurement sample The complex permittivity measuring device according to any one of Claims 1 to 3.

6. The first spacer and the second spacer are located away from the first recess and the second recess respectively The complex permittivity measuring device according to Claim 5.

7. The electromagnetic field forming unit has a transmitting antenna and a receiving antenna, The transmitting antenna and the receiving antenna are located on a plane passing through the centers of a plurality of the second sides. The complex permittivity measuring device according to any one of claims 1 to 3.

8. The electromagnetic field forming unit includes a transmitting antenna and a receiving antenna. The transmitting antenna and the receiving antenna are configured to be movable in a direction perpendicular to the in-plane direction of the measurement sample. The complex permittivity measuring device according to any one of claims 1 to 3.

9. The electromagnetic field forming unit includes a first waveguide that transmits electromagnetic waves to the first recess or the second recess, and a second waveguide that receives electromagnetic waves from the first recess or the second recess. The complex permittivity measuring device according to any one of claims 1 to 3.

10. A conductive first housing having a first surface and a first recess located on the first surface; A conductive second housing having a second surface and a second recess located on the second surface; An electromagnetic field forming unit provided in at least one of the first recess and the second recess, and configured to form an electromagnetic field in the first recess and the second recess; Comprising Both the first recess and the second recess are in the shape of a rectangular parallelepiped. A side along the depth direction of the first recess and the second recess is defined as the first side, and among the two sides orthogonal to the depth direction of the first recess and the second recess, the shorter side is defined as the second side and the longer side is defined as the third side. When the dimension of the first side is h / 2, the dimension of the second side is w, and the dimension of the third side is L, a measuring device having a relationship of w < h < L is used. A sandwiching step of sandwiching a plate-shaped measurement sample between the first surface of the first housing and the second surface of the second housing so that the first recess and the second recess are in the shape of a rectangular parallelepiped; A forming step of forming one or more resonance modes having an electric field parallel to the second side in the first recess and the second recess by the electromagnetic field forming unit; A measuring step of measuring the resonance frequency and the unloaded Q of the one or more resonance modes; A calculating step of calculating the complex permittivity in the in-plane direction of the measurement sample based on the resonance frequency and the unloaded Q measured in the measuring step; A method for measuring complex permittivity including the above steps.

11. Further including a recess alignment step of aligning the relative positions between the first recess and the second recess before the sandwiching step. The method for measuring complex permittivity according to claim 10.

12. The concave portion alignment step includes a first alignment step of aligning the positions in a direction parallel to the third side between the first concave portion and the second concave portion such that the resonance frequency of the one or more resonance modes having an electric field parallel to the second side becomes a minimum value. The method for measuring complex permittivity according to claim 11.

13. The concave portion alignment step includes a second alignment step of aligning the positions in a direction parallel to the second side between the first concave portion and the second concave portion such that the resonance frequency of the one or more resonance modes having an electric field parallel to the second side becomes a maximum value. The method for measuring complex permittivity according to claim 11 or 12.

14. The concave portion alignment step includes a third alignment step of aligning the positions in the rotational direction between the first concave portion and the second concave portion such that the resonance frequency of the one or more resonance modes having an electric field parallel to the second side becomes a maximum value. The method for measuring complex permittivity according to claim 11 or 12.

15. The electromagnetic field forming unit has a transmitting antenna and a receiving antenna configured to be movable in a direction perpendicular to the in-plane direction of the measurement sample. The method further includes an antenna alignment step that is performed between the sandwiching step and the measurement step and aligns the positions of the transmitting antenna and the receiving antenna. The method for measuring complex permittivity according to any one of claims 10 to 12.

16. A conductive first housing having a first surface and a first concave portion located on the first surface. A conductive second housing having a second surface and a second concave portion located on the second surface. Comprising Both the first concave portion and the second concave portion are in the shape of a rectangular parallelepiped. The side along the depth direction of the first concave portion and the second concave portion is defined as the first side, and among the two sides orthogonal to the depth direction of the first concave portion and the second concave portion, the shorter side is defined as the second side and the longer side is defined as the third side. When the dimension of the first side is h / 2, the dimension of the second side is w, and the dimension of the third side is L, they have a relationship of w < h < L. The first surface of the first housing and the second surface of the second housing sandwich a plate-shaped measurement sample so that the first concave portion and the second concave portion are in the shape of a rectangular parallelepiped, and the first concave portion and the second concave portion are configured to be able to form one or more resonance modes having an electric field parallel to the second side by an electromagnetic field forming unit. Resonator

17. For the w and the h, 0.1 ≦ w / h ≦ 0.

7. The resonator according to claim 16.

18. The L and the h satisfy L / h ≥ 2 The resonator according to claim 17 **Claim 19** The first housing is positioned so as to surround the first recess, and has a first spacer made of dielectric or conductive material that contacts the measurement sample when sandwiching the measurement sample The second housing is positioned so as to surround the second recess, and has a second spacer made of dielectric or conductive material that contacts the measurement sample when sandwiching the measurement sample The resonator according to any one of claims 16 to 18 **Claim 20** The first spacer and the second spacer are respectively positioned away from the first recess and the second recess The resonator according to claim 19

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