Motor control devices, hybrid systems, electromechanical integrated units, electric vehicle systems

Abstract

To sufficiently reduce a power loss, which is generated during motor drive, respectively in cases where a motor performs co-rotation drive and does not perform it.SOLUTION: A motor control device 1 comprises: a carrier wave generation section 17 which generates a carrier wave signal Tr; a carrier frequency adjustment section 16 for adjusting a carrier frequency fc which is a frequency of a carrier wave; and a PWM control section 18 which uses the carrier wave signal Tr to perform pulse width modulation on three-phase voltage commands Vu*, Vv* and Vw* and generates a PWM pulse signal for controlling the operation of an inverter. The carrier frequency adjustment section 16 adjusts the carrier frequency fc in such a manner that the carrier frequency fc while a motor 2 performs co-rotation drive is higher than the carrier frequency fc while the motor does not perform the co-rotation drive.SELECTED DRAWING: Figure 2

Description

Technical Field

[0001] The present invention relates to a motor control device, a hybrid system, an electromechanical unit, and an electric vehicle system.

Background Art

[0002] Conventionally, a motor control device that controls the operation of an inverter that converts DC power into AC power using a plurality of switching elements and drives an AC motor using the AC power output from the inverter to control the motor is known. Such motor control devices are widely used for controlling motors in electric vehicles such as railway vehicles and electric automobiles. [[ID=I4]]

[0003] Among the motors mounted on electric vehicles, permanent magnet synchronous motors in which permanent magnets are attached to rotors are widely adopted. In a region where the load on the motor is small, so-called drag drive of the motor occurs, in which the rotor of the motor is rotationally driven by the rotation of the motor drive shaft accompanying the running of the electric vehicle. When the motor is under drag drive, an alternating magnetic field is generated in the stator as the rotor of the motor is rotationally driven, resulting in a problem of no-load iron loss (drag loss). [[ID=I8]]

[0004] Regarding reduction of iron loss in a motor, for example, the technique of Patent Document 1 is known. Patent Document 1 describes a control device for an AC motor that reduces iron loss in a motor by calculating in advance, by electromagnetic field analysis, a current waveform for reducing the iron loss in the motor and performing energization control of the motor according to the calculated current waveform.

Prior Art Documents

Patent Documents

[0005]

Patent Document 1

Summary of the Invention

Problems to be Solved by the Invention

[0006] Power losses generated during motor operation mainly consist of inverter switching losses and motor iron losses. These losses vary depending on the inverter switching frequency and the motor load condition. However, the control device described in Patent Document 1 does not take this into consideration. Therefore, it is not possible to sufficiently reduce power losses generated during motor operation, both when the motor is driven in a co-rotating manner and when it is not. [Means for solving the problem]

[0007] A motor control device according to one aspect of the present invention is connected to an inverter that converts DC power to AC power and outputs it to a motor, and controls the operation of the inverter in accordance with a torque command, thereby controlling the drive of the motor using the inverter, and comprises a carrier wave generation unit that generates a carrier wave, a carrier frequency adjustment unit that adjusts the carrier frequency which is the frequency of the carrier wave, and a PWM control unit that pulse-width modulates a voltage command using the carrier wave and generates a PWM pulse signal for controlling the operation of the inverter, wherein the carrier frequency adjustment unit adjusts the carrier frequency such that the carrier frequency when the motor is driven in conjunction with the motor is higher than the carrier frequency when the motor is not driven in conjunction with the motor. 。 Book The hybrid system according to the invention comprises a motor control device, an inverter connected to the motor control device, a motor driven by the inverter, and an engine system connected to the motor. The electromechanical unit according to the present invention comprises a motor control device, an inverter connected to the motor control device, a motor driven by the inverter, and a gear that transmits the rotational driving force of the motor, wherein the motor, the inverter, and the gear are integrated into a single structure. The electric vehicle system according to the present invention comprises a motor control device, an inverter connected to the motor control device, and a motor driven by the inverter, and travels using the rotational driving force of the motor. [Effects of the Invention]

[0008] According to the present invention, power loss during motor operation can be sufficiently reduced in both cases, whether the motor is driven in a co-rotating manner or not. [Brief explanation of the drawing]

[0009] [Figure 1] An overall configuration diagram of a motor drive system equipped with a motor control device according to one embodiment of the present invention. [Figure 2] A block diagram showing the functional configuration of a motor control device according to the first embodiment of the present invention. [Figure 3] A diagram illustrating the relationship between motor losses, inverter losses, and the combined system losses. [Figure 4] A figure showing an example of the simulation results of a current waveform. [Figure 5] This figure shows the ratio of motor losses to inverter losses in the system losses. [Figure 6] A diagram illustrating an example of the relationship between motor rotation speed and motor torque during vehicle operation. [Figure 7] A diagram illustrating an example of system loss when the carrier frequency is changed. [Figure 8] A flowchart illustrating the processing of the carrier frequency adjustment unit in the first embodiment of the present invention. [Figure 9] A diagram showing an example of carrier frequency adjustment in the first embodiment of the present invention. [Figure 10] This figure shows an example of the calculation results of system losses in motor control when applying the present invention compared to conventional motor control. [Figure 11] This diagram illustrates the relationship between the carrier signal in conventional motor control and the current control and current command output performed within the microcontroller. [Figure 12] A diagram showing the relationship between the carrier signal and the current control and current command output implemented in the microcomputer in the motor control device of the present embodiment. [Figure 13] A block diagram showing the functional configuration of a motor control device according to a second embodiment of the present invention. [Figure 14] A block diagram of a command correction unit according to a second embodiment of the present invention. [Figure 15] A diagram showing an example of iron loss for each time order when a d-axis current is applied to the motor. [Figure 16] A flowchart showing the processing of the command correction unit, switching unit, and carrier frequency adjustment unit in the second embodiment of the present invention. [Figure 17] A configuration diagram of a hybrid system in a third embodiment of the present invention. [Figure 18] An external perspective view of an electromechanical integrated unit in a fourth embodiment of the present invention. [Figure 19] A configuration diagram of a hybrid vehicle system according to a fifth embodiment of the present invention.

Embodiments for Carrying Out the Invention

[0010] (First Embodiment) Hereinafter, a first embodiment of the present invention will be described with reference to the drawings.

[0011] FIG. 1 is an overall configuration diagram of a motor drive system including a motor control device according to an embodiment of the present invention. In FIG. 1, the motor drive system 100 includes a motor control device ①, a permanent magnet synchronous motor (hereinafter simply referred to as "motor") ②, an inverter ③, a rotational position detector ④, and a high-voltage battery ⑤.

[0012] The motor control device 1 controls the operation of the inverter 3 based on a torque command T* corresponding to the target torque requested from the vehicle to the motor 2, thereby generating a PWM pulse signal to control the drive of the motor 2. The generated PWM pulse signal is then output to the inverter 3. Further details of the motor control device 1 will be explained later.

[0013] The inverter 3 comprises an inverter circuit 31, a gate drive circuit 32, and a smoothing capacitor 33. The gate drive circuit 32 generates gate drive signals to control each switching element in the inverter circuit 31 based on a PWM pulse signal input from the motor control device 1, and outputs these signals to the inverter circuit 31. The inverter circuit 31 has switching elements corresponding to the upper and lower arms of the U, V, and W phases, respectively. By controlling these switching elements according to the gate drive signals input from the gate drive circuit 32, the DC power supplied from the high-voltage battery 5 is converted to AC power and output to the motor 2. The smoothing capacitor 33 smooths the DC power supplied from the high-voltage battery 5 to the inverter circuit 31.

[0014] Motor 2 is a synchronous motor that is rotationally driven by AC power supplied from inverter 3, and has a stator and a rotor. When AC power input from inverter 3 is applied to the armature coils Lu, Lv, and Lw provided on the stator, three-phase AC currents Iu, Iv, and Iw conduct in motor 2, and armature magnetic flux is generated in each armature coil. Attraction and repulsion forces are generated between the armature magnetic flux of each armature coil and the magnetic flux of the permanent magnets arranged on the rotor, which generates torque in the rotor and drives the rotor to rotate.

[0015] A rotational position sensor 8 is attached to the motor 2 to detect the rotational position θ of the rotor. The rotational position detector 4 calculates the rotational position θ from the input signal of the rotational position sensor 8. The result of the rotational position θ calculation by the rotational position detector 4 is input to the motor control device 1 and used in AC power phase control, which is performed by the motor control device 1 generating a PWM pulse signal in accordance with the phase of the induced voltage of the motor 2.

[0016] Here, a resolver consisting of an iron core and windings is more preferable for the rotational position sensor 8, but a magnetoresistive element such as a GMR sensor or a sensor using a Hall element is also acceptable. Furthermore, the rotational position detector 4 may estimate the rotational position θ using the three-phase AC currents Iu, Iv, Iw flowing through the motor 2, or the three-phase AC voltages Vu, Vv, Vw applied to the motor 2 from the inverter 3, without using the input signal from the rotational position sensor 8.

[0017] A current detection unit 7 is positioned between the inverter 3 and the motor 2. The current detection unit 7 detects the three-phase AC currents Iu, Iv, and Iw (U-phase AC current Iu, V-phase AC current Iv, and W-phase AC current Iw) that energize the motor 2. The current detection unit 7 is configured using, for example, a Hall current sensor. The detection results of the three-phase AC currents Iu, Iv, and Iw by the current detection unit 7 are input to the motor control device 1 and used to generate the PWM pulse signal performed by the motor control device 1. Although Figure 2 shows an example in which the current detection unit 7 is composed of three current detectors, the current detectors can be limited to two, and the remaining one-phase AC current can be calculated from the fact that the sum of the three-phase AC currents Iu, Iv, and Iw is zero. Alternatively, the pulsed DC current flowing from the high-voltage battery 5 to the inverter 3 may be detected by a shunt resistor or the like inserted between the smoothing capacitor 33 and the inverter 3, and the three-phase AC currents Iu, Iv, and Iw may be determined based on this DC current and the three-phase AC voltages Vu, Vv, and Vw applied from the inverter 3 to the motor 2.

[0018] Next, the motor control device 1 will be described in detail. Figure 2 is a block diagram showing the functional configuration of the motor control device 1 according to the first embodiment of the present invention. In Figure 2, the motor control device 1 has the following functional blocks: a current command generation unit 11, a speed calculation unit 12, a current conversion unit 13, a current control unit 14, a three-phase voltage conversion unit 15, a carrier frequency adjustment unit 16, a carrier wave generation unit 17, and a PWM control unit 18. The motor control device 1 is configured, for example, by a microcomputer, and these functional blocks can be realized by executing a predetermined program in the microcomputer. Alternatively, some or all of these functional blocks may be realized using hardware circuits such as logic ICs or FPGAs.

[0019] The current command generation unit 11 calculates the d-axis current command Id* and the q-axis current command Iq* based on the input torque command T* and the voltage Hvdc of the high-voltage battery 5. Here, for example, a pre-set current command map or mathematical formula is used to determine the d-axis current command Id* and the q-axis current command Iq* corresponding to the torque command T*.

[0020] The speed calculation unit 12 calculates the motor rotation speed ωr, which represents the rotational speed (rotational speed) of the motor 2, from the time change of the rotational position θ. Note that the motor rotation speed ωr may be expressed as either angular velocity (rad / s) or rotational speed (rpm). These values ​​may also be converted to and used interchangeably.

[0021] The current conversion unit 13 performs a dq conversion on the three-phase AC currents Iu, Iv, and Iw detected by the current detection unit 7, based on the rotation position θ determined by the rotation position detector 4, and calculates the d-axis current value Id and the q-axis current value Iq.

[0022] The current control unit 14 calculates the d-axis voltage command Vd* and q-axis voltage command Vq* corresponding to the torque command T*, based on the deviation between the d-axis current command Id* and q-axis current command Iq* output from the current command generation unit 11 and the d-axis current value Id and q-axis current value Iq output from the current conversion unit 13, so that these values ​​match. Here, for example, using a control method such as PI control, the d-axis voltage command Vd* corresponding to the deviation between the d-axis current command Id* and the d-axis current value Id, and the q-axis voltage command Vq* corresponding to the deviation between the q-axis current command Iq* and the q-axis current value Iq are determined at predetermined calculation cycles Tv.

[0023] The three-phase voltage conversion unit 15 performs a three-phase conversion based on the rotation position θ determined by the rotation position detector 4, using the d-axis voltage command Vd* and q-axis voltage command Vq* calculated by the current control unit 14, and calculates three-phase voltage commands Vu*, Vv*, and Vw* (U-phase voltage command value Vu*, V-phase voltage command value Vv*, and W-phase voltage command value Vw*). This generates three-phase voltage commands Vu*, Vv*, and Vw* corresponding to the torque command T*.

[0024] The carrier frequency adjustment unit 16 adjusts the carrier frequency fc, which is the frequency of the carrier wave used to generate the PWM pulse signal, based on the rotational speed ωr determined by the speed calculation unit 12. At this time, the carrier frequency adjustment unit 16 determines whether or not the motor 2 is being driven in a co-rotational manner based on the torque command T* or the current command generation unit 11. If it is determined that the motor 2 is being driven in a co-rotational manner, the carrier frequency fc is adjusted so that it is higher than the carrier frequency fc when the motor 2 is not being driven in a co-rotational manner. This reduces the power loss that occurs when the motor 2 is being driven in a co-rotational manner, in both cases. Details of the carrier frequency adjustment unit 16 will be described later.

[0025] The carrier wave generation unit 17 generates a carrier wave signal (triangular wave signal) Tr based on the carrier frequency fc calculated by the carrier frequency adjustment unit 16.

[0026] The PWM control unit 18 uses the carrier signal Tr output from the carrier wave generation unit 17 to pulse-width modulate the three-phase voltage commands Vu*, Vv*, and Vw* output from the three-phase voltage conversion unit 15, respectively, and generates PWM pulse signals to control the operation of the inverter 3. Specifically, based on the comparison result between the three-phase voltage commands Vu*, Vv*, and Vw* output from the three-phase voltage conversion unit 15 and the carrier signal Tr output from the carrier wave generation unit 17, it generates pulsed voltages for each of the U, V, and W phases. Then, based on the generated pulsed voltages, it generates PWM pulse signals for the switching elements of each phase of the inverter 3. At this time, the PWM pulse signals Gup, Gvp, and Gwp of the upper arm of each phase are logically inverted to generate the PWM pulse signals Gun, Gvn, and Gwn of the lower arm. The PWM pulse signals generated by the PWM control unit 18 are output from the motor control device 1 to the gate drive circuit 32 of the inverter 3, and are converted into gate drive signals by the gate drive circuit 32. This controls the on / off state of each switching element in the inverter circuit 31, thereby adjusting the output voltage of the inverter 3.

[0027] Next, the operation of the carrier frequency adjustment unit 16 in the motor control device 1 will be described. As described above, the carrier frequency adjustment unit 16 determines whether or not the motor 2 is being driven in a traversal manner based on the torque command T* or the d-axis current command Id* and q-axis current command Iq* generated by the current command generation unit 11. If it is determined that the motor 2 is being driven in a traversal manner, the carrier frequency fc is adjusted so that the carrier frequency fc is higher than when the motor 2 is not being driven in a traversal manner. By sequentially controlling the frequency of the carrier signal Tr generated by the carrier wave generation unit 17 according to this carrier frequency fc, the PWM control unit 18 generates a PWM pulse signal in such a way that it reduces the power loss that occurs when the motor 2 is being driven in a traversal manner, in both cases.

[0028] The losses of the motor 2 and inverter 3 that constitute the motor drive system 100 are described below. There are two main types of motor losses in the motor 2: copper loss and iron loss. Copper loss is the loss that occurs when current flows through the copper wire coil connected to the stator, and it increases in proportion to the square of the current amplitude. This copper loss is not affected by the step size of the PWM pulse signal output from the motor control device 1 to the inverter 3. On the other hand, iron loss is the loss that occurs due to fluctuations in the magnetic flux flowing through the stator and rotor. It is widely known that the finer the step size of the PWM pulse signal, the more the fluctuations in the magnetic flux generated from the copper wire coil of the stator are suppressed, and therefore the iron loss decreases.

[0029] Furthermore, the inverter losses generated in inverter 3 can be broadly divided into two types: conduction losses and switching losses. Conduction losses are losses that occur when each switching element conducts, and they increase in proportion to the current flowing through inverter 3. On the other hand, switching losses are losses that occur due to the on / off operation of each switching element. It is widely known that the finer the step size of the PWM pulse signal, the more times the switching elements are turned on and off, and therefore the greater the switching losses.

[0030] Figure 3 is a diagram illustrating the relationship between motor losses, inverter losses, and the combined system losses in a motor drive system 100. In Figure 3, the vertical axis represents the magnitude of each loss, and the horizontal axis represents the switching frequency that determines the step size of the PWM pulse signal, i.e., the carrier frequency fc. From Figure 3, it can be seen that as the switching frequency increases, motor losses decrease while inverter losses increase, indicating a trade-off relationship between these losses. Therefore, in conventional motor control methods, it was common to adjust the carrier frequency fc to aim for the minimum point (the point of highest system efficiency) where system losses are minimized.

[0031] However, the inventors of this invention have found that the above-mentioned trade-off relationship does not necessarily hold true depending on the motor's torque and rotational speed. This point will be explained in detail below.

[0032] Figure 4 shows an example of the simulation results of the current waveform when an 8-pole permanent magnet synchronous motor is driven at 8,000 r / min using a PWM pulse signal with a carrier frequency fc of 8 kHz. Figure 6(a) shows an example of the U-phase current waveform obtained by simulation, and Figure 6(b) shows the results of analyzing the frequency components of the current waveform in Figure 6(a) using FFT (Fast Fourier Transformation). From these figures, it can be seen that even if the amplitude of the fundamental wave corresponding to the current command is not very large, around a few A, current distortion (current distortion caused by time harmonics) in the frequency components near the carrier frequency fc occurs with an amplitude similar to that of the fundamental wave, and this causes large fluctuations in the current waveform.

[0033] Figure 5 shows the ratio of motor loss to inverter loss in the system loss under the motor drive conditions shown in Figure 4. Figure 5 shows examples of the motor loss and inverter loss generated by the current waveform exemplified in Figure 4(a), and the system loss which is the sum of these, calculated by electromagnetic field analysis. From Figure 5, it can be seen that in the region where the current flowing through the motor is not very large, around a few amperes, motor loss accounts for the vast majority of the system loss at 99.88%, while inverter loss is extremely small at 0.12%. Furthermore, a detailed analysis of the breakdown of motor loss revealed that various motor losses originating from harmonics (harmonic iron loss, magnet loss, AC copper loss) account for 18% of the total motor loss. It was also found that these harmonic-derived motor losses (harmonic iron loss, magnet loss, AC copper loss) are caused by current distortion due to the time harmonics mentioned above.

[0034] Next, an example of motor operation during vehicle operation will be described below. Figure 6 shows an example of the relationship between motor rotation speed and motor torque during vehicle operation. In Figure 6, the relationship between motor rotation speed and motor torque when a vehicle equipped with the motor drive system 100 is driven in WLTC (Worldwide harmonized Light vehicle Test Cycles) mode is shown on an NT characteristic curve with motor rotation speed (r / min) on the horizontal axis and motor torque (Nm) on the vertical axis. From Figure 6, it can be seen that the motor torque operating points in WLTC driving mode are mostly located within a certain range centered on a torque value of 0, that is, in the region where the motor load is below a certain level. In particular, in the region near a torque value of 0, motor 2 is driven in a co-rotational manner, and it can be seen that many torque operating points exist within this co-rotational driving region.

[0035] Generally, as motor torque increases, the modulation ratio also increases. Therefore, as shown in Figure 6, it can be said that the operating point of motor torque in WLTC driving mode is largely located within a certain range of modulation ratio. Modulation ratio is a parameter that represents the ratio of DC voltage to AC voltage, and is also called voltage utilization rate. The modulation ratio is calculated using the following equation (1) based on the d-axis voltage Vd, the q-axis voltage Vq, and the voltage Hvdc of the high-voltage battery 5. H=√(Vd 2 +Vq 2 ) / Hvdc ···(1)

[0036] As explained above, in WLTC driving mode, the majority of the region is below a certain value in absolute terms (modulation frequency of 1.25 or less), and this region includes many areas where motor 2 is driven by rotation. When the rotational speed of motor 2 is above a certain value, it is necessary to supply a field weakening current to motor 2 so that the voltage Hvdc applied from the high-voltage battery 5 to the inverter 3 does not saturate due to the induced voltage of the rotor magnet. However, as can be seen from Figure 6, the motor torque is generally small in WLTC driving mode, and therefore the field weakening current is not supplied for much of the time while the vehicle is running.

[0037] Therefore, in this embodiment, when motor 2 is driven in a co-operated manner, time harmonics are improved by increasing the carrier frequency fc, to the extent that inverter losses do not increase. As described above, since the modulation rate does not often exceed 1.25 during vehicle operation, improving the carrier frequency fc when motor 2 is driven in a co-operated manner yields a greater effect in reducing system losses.

[0038] In this embodiment, the motor control device 1 selects the characteristics of each component of the motor 2 so that the induced voltage induced in each armature coil of the stator when the motor 2 is at its maximum rotational speed does not exceed the withstand voltage of the switching elements of the inverter 3. In other words, the motor control device 1 of this embodiment controls the driving of the motor 2 so that the induced voltage generated by the rotation of the motor 2 is less than the withstand voltage of the switching elements of the inverter 3.

[0039] Figure 7 shows an example of system loss (sum of motor loss and inverter loss) when the carrier frequency fc is changed. Figure 7 illustrates the relationship between switching frequency and system loss when the carrier frequency fc is changed under the same motor driving conditions as in Figure 4. In Figure 7, graph 41 shows an example of switching frequency and system loss when an IGBT (Insulated Gate Bipolar Transistor) is used as the switching element, and graph 42 shows an example of switching frequency and system loss when a SiC (Silicon Carbide) semiconductor is used as the switching element.

[0040] Furthermore, when motor 2 is driven in a co-rotational manner, inverter losses are minimal as explained in Figure 5, and therefore there is no minimum point where the system is most efficient, as shown in Figure 3. For this reason, as shown in graphs 41 and 42 in Figure 7, system losses decrease monotonically with increasing switching frequency. Thus, the inventors of the present invention have discovered that when the motor load is relatively small, system losses decrease monotonically with respect to the carrier frequency fc. In other words, when motor 2 is driven in a co-rotational manner, motor harmonic losses can be minimized by increasing the carrier frequency fc as much as possible, within the limits of the processing load constraints of the microcontroller realizing the motor control device 1 and the capacity constraints of the gate power supply (not shown) that supplies power to the gate drive circuit 32 of the inverter 3.

[0041] The branching point between the monotonically decreasing curve shown in Graphs 41 and 42 and the curve with the minimum point where the system achieves maximum efficiency is determined by the motor torque and current. Therefore, it is necessary to determine the torque and current conditions for switching the control of the carrier frequency fc by performing simulations using electromagnetic field analysis and actual machine verification in advance. By using these determined torque and current conditions as thresholds and switching whether or not to increase the carrier frequency fc before and after the threshold, it is possible to sufficiently reduce the power loss that occurs during motor operation, both when motor 2 is driven in a rotational manner and when it is not.

[0042] Figure 8 is a flowchart showing the processing of the carrier frequency adjustment unit 16 in the first embodiment of the present invention. The processing shown in the flowchart of Figure 8 is performed in the carrier frequency adjustment unit 16, for example, at predetermined processing cycles.

[0043] In step S101, the torque command T*, or the values ​​of the d-axis current command Id* and q-axis current command Iq* generated by the current command generation unit 11 are obtained. Both of these may be obtained, or only one of them may be obtained.

[0044] In step S102, the absolute value of the torque command T* or current command (d-axis current command Id* and q-axis current command Iq*) obtained in step S101 is compared with a predetermined threshold to determine whether the absolute value of the torque command T* or current command is less than or equal to the threshold. In this case, if a torque command T* was obtained in step S101, the absolute value of that torque command T* is compared with the threshold for the torque command, and if a current command was obtained, the absolute value of that current command is compared with the threshold for the current command. The threshold used in the determination in step S102 is determined based on the results of a simulation or experiment conducted in advance using electromagnetic field analysis, as described above, and is stored in the motor control device 1.

[0045] In step S102, if the absolute value of the torque command T* or current command is less than or equal to the threshold, it is determined that motor 2 is being driven in a cascading manner, and the process proceeds to step S110. On the other hand, if the absolute value of the torque command T* or current command is greater than the threshold, it is determined that motor 2 is not being driven in a cascading manner, and the process shown in the flowchart of Figure 8 is terminated. In this case, the carrier frequency adjustment unit 16 adjusts the carrier frequency fc based on the rotational speed ωr, similar to normal synchronous PWM control.

[0046] In step S110, the carrier frequency fc is increased within a predetermined constraint range compared to the carrier frequency fc in normal synchronous PWM control. This adjusts the carrier frequency fc so that when the motor 2 is driven in a co-rotational manner, it is higher than when the motor 2 is not driven in a co-rotational manner. As a result, if the absolute value of the torque command T* or current command is below a predetermined threshold, the PWM control unit 18 can generate a PWM pulse signal to control the operation of the inverter 3 so as to suppress harmonic pulsations of the gap magnetic flux density between the stator and rotor of the motor 2. The constraint range of the carrier frequency fc is determined based on factors such as the processing load of the microcontroller implementing the motor control device 1 and the capacity of the gate power supply that supplies power to the gate drive circuit 32 of the inverter 3, as described above, and is stored in the motor control device 1.

[0047] After adjusting the carrier frequency fc in step S110, the process shown in the flowchart in Figure 8 is terminated.

[0048] Figure 9 shows an example of carrier frequency adjustment in the first embodiment of the present invention. Figure 9(a) shows an example of the time change of a torque command T* or a current command, with time on the horizontal axis and the absolute value of the torque command T* or current command on the vertical axis. Figure 9(b) shows an example of the time change of the adjusted carrier frequency fc compared to Figure 9(a), with time on the horizontal axis and the carrier frequency fc on the vertical axis.

[0049] As shown in Figure 9(a), motor 2 is driven normally until time t1, and the absolute value of the torque command T* or current command at this time is relatively large. On the other hand, after time t1, motor 2 is driven in a co-rotational manner, and the absolute value of the torque command T* or current command at this time is smaller than during normal driving and falls below a predetermined threshold. As a result, as shown in Figure 9(b), during co-rotational driving after time t1, the carrier frequency fc changes so that the carrier frequency fc is higher within a predetermined constraint range compared to the normal driving before time t1.

[0050] Furthermore, if the carrier frequency fc is increased all at once at time t1, the control amount of motor 2 will also change abruptly, which can cause a sudden change in the driving state of motor 2 and lead to vibration and noise. To avoid this, when changing the carrier frequency fc in response to a change from normal driving to rotational driving, an upper limit may be set on the range of change of the carrier frequency fc so that the rate of change of the carrier frequency fc per unit time is below a predetermined value.

[0051] The motor control device 1 of this embodiment, by performing the operations described above, can suppress the increase in inverter losses due to the increase in the switching frequency of the inverter 3, while also suppressing motor losses (harmonic iron loss, magnet loss, AC copper loss) in the motor 2, both when the motor 2 is driven in a rotational manner and when it is not. As a result, it becomes possible to reduce system losses.

[0052] Figure 10 shows an example of the calculation results of system losses in both conventional motor control without the application of the present invention and motor control with the application of the present invention. In the example in Figure 10, the calculation results of system losses are shown when the vehicle's driving pattern is in WLTC mode.

[0053] Figure 10 shows that when the present invention is applied to motor control, system losses can be reduced by 2.7% compared to conventional motor control.

[0054] Next, the method for reducing the microcontroller processing load in this embodiment will be described below.

[0055] In the motor control device 1 of this embodiment, in order to minimize system losses during rotational drive, as described above, it is necessary to increase the carrier frequency fc as much as possible and improve the switching frequency within the constraints of the microcontroller's processing load and the gate power supply capacity. To achieve this, it is desirable to reduce the microcontroller's processing load as much as possible. Below, with reference to Figures 11 and 12, an example of a method for reducing the microcontroller's processing load in the motor control device 1 of this embodiment will be described.

[0056] Figure 11 shows the relationship between the carrier signal Tr in conventional motor control and the current control and current command output performed within the microcontroller, which is the motor control device 1. In conventional motor control, for example, current control by the microcontroller is started at the peak (the point where it changes from rising to falling) and trough (the point where it changes from falling to rising) of the carrier signal Tr, and the calculated duty cycle voltage commands (d-axis voltage command Vd* and q-axis voltage command Vq*) are output for the period of the peak or trough of the carrier signal Tr corresponding to the next current control period. This makes it possible to generate a PWM pulse signal with fine steps and few time harmonics.

[0057] However, in the conventional motor control method shown in Figure 11, for example, if the carrier frequency fc is 20 kHz, the interval for starting current control is 25 μs. Therefore, the processing load for current control in the microcontroller is relatively large, which reduces the time available for other processing.

[0058] Figure 12 shows the relationship between the carrier signal Tr in the motor control device 1 of this embodiment and the current control and current command output performed within the microcontroller that is the motor control device 1. In the motor control device 1 of this embodiment, for example as shown in Figure 12, the microcontroller starts current control at a rate of once every three peaks and troughs of the carrier signal Tr. Then, the calculated duty cycle voltage commands (d-axis voltage command Vd* and q-axis voltage command Vq*) are repeatedly output for the period of the carrier signal Tr corresponding to the next current control period, that is, for the period of three consecutive peaks and troughs. This separates the period of current control from the period of the carrier signal Tr, reducing the processing load of current control in the microcontroller, while enabling the generation of a PWM pulse signal with fine steps and few time harmonics.

[0059] In Figure 12, an example is shown where the microcontroller controls the current at a rate of once every three peaks and troughs of the carrier signal Tr, but other rates may be used. At a minimum, the above effect can be achieved if the calculation period of the voltage command performed by the current control unit 14 is longer than half the period of the carrier signal Tr, i.e., the interval between peaks and troughs. In other words, the carrier frequency adjustment unit 16 can reduce the processing load of current control and further improve the switching frequency by adjusting the carrier frequency fc when the motor 2 is driven in conjunction with the carrier so that the calculation period of the voltage command performed by the current control unit 14 is longer than half the period of the carrier signal Tr. If the microcontroller has sufficient processing power, it is not necessary to adopt the motor control method shown in Figure 12, and the conventional motor control method shown in Figure 11 may be used instead.

[0060] According to the first embodiment of the present invention described above, the following effects are achieved.

[0061] (1) The motor control device 1 is connected to an inverter 3 that converts DC power to AC power and outputs it to the motor 2, and controls the operation of the inverter 3 according to the torque command T*, thereby controlling the drive of the motor 2 using the inverter 3. The motor control device 1 includes a carrier wave generation unit 17 that generates a carrier wave signal Tr, a carrier frequency adjustment unit 16 that adjusts the carrier frequency fc, which is the frequency of the carrier wave, and a PWM control unit 18 that generates a PWM pulse signal for controlling the operation of the inverter 3 by pulse width modulating the three-phase voltage commands Vu*, Vv*, and Vw* using the carrier wave signal Tr. The carrier frequency adjustment unit 16 adjusts the carrier frequency fc so that the carrier frequency fc when the motor 2 is driven in a co-rotation manner is higher than the carrier frequency fc when the motor 2 is not driven in a co-rotation manner (step S110). In this way, power loss that occurs when the motor is driven in a co-rotation manner can be sufficiently reduced in both cases.

[0062] (2) The carrier frequency adjustment unit 16 compares the absolute value of the torque command T* with a predetermined threshold (step S102), and if the absolute value of the torque command T* is less than or equal to the threshold (step S102: Yes), it determines that the motor 2 is being driven in a co-rotational manner. In this way, it is easy to determine whether or not the motor 2 is being driven in a co-rotational manner.

[0063] (3) The above threshold is determined based on the results of electromagnetic field analysis simulations or experiments conducted in advance. In this way, an appropriate threshold can be set.

[0064] (4) The carrier frequency fc when the motor 2 is driven in a co-rotational manner is determined based on at least one of the processing load of the motor control device 1 and the capacity of the gate power supply that supplies power to the gate drive circuit 32 of the inverter 3. In this way, the carrier frequency fc when the motor 2 is driven in a co-rotational manner can be increased within a reasonable range.

[0065] (5) The motor control device 1 controls the drive of the motor 2 so that the induced voltage generated by the rotation of the motor 2 is less than the withstand voltage of the switching elements of the inverter 3. In this way, even when the motor 2 is driven at high speed, it is possible to prevent the switching elements of the inverter 3 from being destroyed by the induced voltage.

[0066] (6) The motor control device 1 includes a current control unit 14 that calculates the d-axis voltage command Vd* and the q-axis voltage command Vq* at predetermined calculation cycles. The carrier frequency adjustment unit 16 can adjust the carrier frequency fc when the motor 2 is driven in a co-rotation manner so that the calculation cycle of the voltage command by the current control unit 14 is longer than half the period of the carrier wave signal Tr. In this way, when implementing the motor control device 1 using a microcontroller, it is possible to generate a PWM pulse signal with fine steps and few time harmonics while reducing the processing load of current control in the microcontroller.

[0067] (7) The carrier frequency adjustment unit 16 may adjust the carrier frequency fc so that the rate of change of the carrier frequency fc is less than or equal to a predetermined value. In this way, vibration and noise can be prevented when the drive state of the motor 2 is switched from the normal drive state to the rotational drive state.

[0068] (8) The motor control device 1 is connected to an inverter 3 that converts DC power to AC power and outputs it to the motor 2, and controls the operation of the inverter 3 in accordance with the torque command T*, thereby controlling the drive of the motor 2 using the inverter 3. When the absolute value of the torque command T* is below a predetermined threshold, the motor control device 1 generates a PWM pulse signal to control the operation of the inverter 3 so as to suppress harmonic pulsations of the gap magnetic flux density between the stator and rotor of the motor 2. In this way, when the motor 2 is driven in a co-rotational manner, power loss that occurs when the motor is driven can be reduced.

[0069] (Second embodiment) Next, a second embodiment of the present invention will be described with reference to the drawings. In the first embodiment described above, the inverter loss is small when the motor 2 is driven in a controlled rotational manner, and a motor control method was described in which time harmonics are reduced by increasing the carrier frequency fc, thereby reducing motor losses (magnet loss, AC copper loss, iron loss) originating from harmonics and reducing system losses. In contrast, the second embodiment will describe a motor control method that further reduces iron loss during field weakening control.

[0070] Figure 13 is a block diagram showing the functional configuration of a motor control device 1A according to a second embodiment of the present invention. In Figure 13, the motor control device 1A has the same configuration as the motor control device 1 described in the first embodiment, except that it further comprises a command correction unit 11A and a switching unit 11B.

[0071] The command correction unit 11A calculates corrected d-axis current command Ihd* and corrected q-axis current command Ihq* to correct the d-axis current command Id* and q-axis current command Iq* generated by the current command generation unit 11, respectively. At this time, the command correction unit 11A calculates current commands to superimpose pulsations corresponding to a predetermined time order onto the d-axis current command Id* and q-axis current command Iq*, respectively, and outputs the calculation results as corrected d-axis current command Ihd* and corrected q-axis current command Ihq*. Details of the calculation method of corrected d-axis current command Ihd* and corrected q-axis current command Ihq* by the command correction unit 11A will be described later.

[0072] The switching unit 11B switches the connection state between the current command generation unit 11 and the command correction unit 11A. When the current command generation unit 11 and the command correction unit 11A are connected by the switching unit 11B, the corrected d-axis current command Ihd* and corrected q-axis current command Ihq* output from the command correction unit 11A are superimposed on the d-axis current command Id* and q-axis current command Iq* output from the current command generation unit 11, respectively, and the d-axis current command Id* and q-axis current command Iq* are corrected. The corrected d-axis current command Id* and q-axis current command Iq* are then input to the current control unit 14 and used in the calculation of the d-axis voltage command Vd* and q-axis voltage command Vq*.

[0073] In this embodiment, the motor control device 1A switches the switching unit 11B to connect the current command generation unit 11 and the command correction unit 11A when the motor 2 is being driven in a co-rotational manner and field weakening control of the motor 2 is being performed. This ensures that the d-axis current command Id* and the q-axis current command Iq* are corrected.

[0074] Next, the operation of the command correction unit 11A in the motor control device 1A will be described. As described above, the command correction unit 11A calculates corrected d-axis current command Ihd* and corrected q-axis current command Ihq* to superimpose pulsations corresponding to a predetermined time order onto the d-axis current command Id* and q-axis current command Iq*, respectively. At this time, the command correction unit 11A calculates the corrected d-axis current command Ihd* and corrected q-axis current command Ihq* by adjusting the amplitude and phase of the pulsations superimposed on the current command based on the motor rotation speed ωr and torque command T* in order to cancel out vibrations and noise generated in the motor 2.

[0075] Figure 14 is a block diagram of the command correction unit 11A according to a second embodiment of the present invention. The command correction unit 11A includes a superimposed dq axis current amplitude calculation unit 111, a superimposed dq axis current phase calculation unit 112, and a corrected dq axis current command generation unit 113.

[0076] The superimposed dq-axis current amplitude calculation unit 111 calculates the amplitude of the pulsation superimposed on the d-axis current command Id* and the q-axis current command Iq*, respectively, based on the torque command T*, the voltage Hvdc of the high-voltage battery 5, and the motor rotation speed ωr. In this embodiment, for example, the superimposed dq-axis current amplitude calculation unit 111 calculates the amplitude of the pulsation superimposed on the d-axis current command Id* and the q-axis current command Iq* for each time order from 6 to 24 times the electrical angular frequency, i.e., time order 6 (rotation 24th order), time order 12 (rotation 48th order), time order 18 (rotation 72nd order), and time order 24 (rotation 96th order), respectively, for a motor 2 with 8 poles and 48 slots. In Figure 14, the amplitude of the pulsation for the d-axis current command Id* and the amplitude of the pulsation for the q-axis current command Iq* are shown together for each order. In other words, the superimposed dq-axis current amplitudes Idq6, Idq12, Idq18, and Idq24 shown in Figure 14 represent the amplitudes of pulsations at the 6th, 12th, 18th, and 24th time orders, respectively, for the d-axis current command Id* and the q-axis current command Iq*.

[0077] The superimposed dq-axis current phase calculation unit 112 calculates the phase of the pulsation superimposed on the d-axis current command Id* and the q-axis current command Iq*, respectively, based on the torque command T*, the voltage Hvdc of the high-voltage battery 5, the motor rotation speed ωr, and the rotation position θ. In this embodiment, for example, the superimposed dq-axis current phase calculation unit 112 calculates the phase of the pulsation superimposed on the d-axis current command Id* and the q-axis current command Iq* for each time order from 6 to 24 times the electrical angular frequency, i.e., time order 6 (rotation 24th order), time order 12 (rotation 48th order), time order 18 (rotation 72nd order), and time order 24 (rotation 96th order), respectively, for a motor 2 with 8 poles and 48 slots. In Figure 14, the phase of the pulsation for the d-axis current command Id* and the phase of the pulsation for the q-axis current command Iq* are shown together for each order. In other words, the superimposed dq-axis current phases θdq6, θdq12, θdq18, and θdq24 shown in Figure 14 represent the phases of pulsations at the 6th, 12th, 18th, and 24th time orders, respectively, for the d-axis current command Id* and the q-axis current command Iq*.

[0078] The corrected dq-axis current command generation unit 113 generates superimposed d-axis current commands Ihd* and superimposed q-axis current commands Ihq* based on the amplitude of each order of pulsation calculated by the superimposed dq-axis current amplitude calculation unit 111, i.e., superimposed dq-axis current amplitudes Idq6, Idq12, Idq18, Idq24, and the phase of each order of pulsation calculated by the superimposed dq-axis current phase calculation unit 112, i.e., superimposed dq-axis current phases θdq6, θdq12, θdq18, θdq24, corresponding to the pulsation.

[0079] The superimposed d-axis current command Ihd* and superimposed q-axis current command Ihq* generated by the corrected dq-axis current command generation unit 113 are input to the output side of the current command generation unit 11 via the switching unit 11B, and these values ​​are subtracted from the d-axis current command Id* and q-axis current command Iq* generated by the current command generation unit 11. As a result, the superimposed d-axis current command Ihd* and superimposed q-axis current command Ihq* are superimposed on the d-axis current command Id* and q-axis current command Iq* as pulsations corresponding to the rotation of the motor 2. The obtained calculation results are then input to the current control unit 14 as the corrected d-axis current command Id* and q-axis current command Iq*.

[0080] Furthermore, the calculations of the superimposed dq-axis current amplitudes Idq6, Idq12, Idq18, and Idq24 in the superimposed dq-axis current amplitude calculation unit 111, and the calculations of the superimposed dq-axis current phases θdq6, θdq12, θdq18, and θdq24 in the superimposed dq-axis current phase calculation unit 112, can be performed, for example, based on pre-stored map information. Each map information can be created in advance by determining the amplitude and phase shift of pulsations that can effectively reduce iron loss generated in the motor 2 during field weakening control for various combinations of torque command T*, high-voltage battery 5 voltage Hvdc, and motor rotation speed ωr, for each order through simulation or actual measurement.

[0081] Next, we will explain how to reduce iron loss during field weakening control in this embodiment. In the first embodiment, as illustrated in Figure 6, we described a method for reducing system losses during rotational drive, focusing on motor 2, which often does not exceed a modulation ratio of 1.25 while the vehicle is running. However, in recent years, there has been an increase in motors with structures that improve induced voltage and reduce motor loss per unit current. When such a motor is used as motor 2 in the motor drive system 100 in Figure 1, the motor control method described in the first embodiment alone may not be sufficient to reduce system losses. The reason for this will be explained below with reference to Figure 15.

[0082] Figure 15 shows an example of iron loss for each time order when a d-axis current Id is applied to motor 2. Focusing on the relationship between the time order and the d-axis current Id, it can be seen that when the d-axis current Id is 0A, a large amount of iron loss occurs in the first time order. Furthermore, it can be seen that as the d-axis current Id is gradually increased from 0A, the iron loss of the fifth time order component increases, while the iron loss of the first time order decreases due to the field weakening effect caused by the application of the d-axis current Id.

[0083] As described above, in motor 2, the iron loss of the fifth time component changes significantly due to field weakening. Therefore, the iron loss during field weakening can be suppressed by a pulsating current command of this time component (the sixth time component when converted to the dq axis). In other words, by pre-calculating the amplitude and phase of the sixth-order component of the dq-axis pulsating current through prior electromagnetic field analysis, and performing current control to follow that current command, it is possible to reduce the iron loss increased by field weakening.

[0084] In this embodiment, the command correction unit 11A and the switching unit 11B described in Figures 13 and 14 realize the current control described above. Specifically, when the motor 2 is under field weakening control, the switching unit 11B connects the current command generation unit 11 and the command correction unit 11A, and the superimposed d-axis current command Ihd* and superimposed q-axis current command Ihq* generated by the command correction unit 11A are used to superimpose pulsations corresponding to the rotation of the motor 2 onto the d-axis current command Id* and q-axis current command Iq*, respectively. Then, by inputting the corrected d-axis current command Id* and q-axis current command Iq* to the current control unit 14 and performing current control, a PWM pulse signal capable of reducing the iron loss increased by field weakening is generated in the PWM control unit 18.

[0085] Figure 16 is a flowchart showing the processing of the command correction unit 11A, the switching unit 11B, and the carrier frequency adjustment unit 16 in a second embodiment of the present invention. The processing shown in the flowchart of Figure 16 is performed in the command correction unit 11A, the switching unit 11B, and the carrier frequency adjustment unit 16, for example, at predetermined processing cycles.

[0086] In steps S101 and S102, the same processes as those described in the flowchart of Figure 8 in the first embodiment are performed, respectively. In the process of step S102, if the absolute value of the torque command T* or current command is less than or equal to the threshold, it is determined that the motor 2 is being driven in a cascading manner, and the process proceeds to step S103. On the other hand, if the absolute value of the torque command T* or current command is greater than the threshold, it is determined that the motor 2 is not being driven in a cascading manner, and the process shown in the flowchart of Figure 16 is terminated. In this case, the carrier frequency adjustment unit 16 adjusts the carrier frequency fc based on the rotational speed ωr, similar to normal synchronous PWM control.

[0087] In step S103, it is determined whether or not field weakening control is being performed on motor 2. If the PWM control unit 18 is performing field weakening control on motor 2 by generating a PWM pulse signal to weaken the magnetic flux of motor 2, the process proceeds to step S120; otherwise, the process proceeds to step S110.

[0088] If the process proceeds from step S103 to step S110, the carrier frequency fc is increased within a predetermined constraint range, similar to the flowchart in Figure 8. In this case as well, similar to the first embodiment, the constraint range of the carrier frequency fc is determined based on factors such as the processing load of the microcontroller implementing the motor control device 1A and the capacity of the gate power supply that supplies power to the gate drive circuit 32 of the inverter 3, and is stored in the motor control device 1A.

[0089] After adjusting the carrier frequency fc in step S110, the process shown in the flowchart in Figure 16 is terminated.

[0090] On the other hand, if the process proceeds from step S103 to step S120, in step S120, the switching unit 11B is switched to the connection side, and the command correction unit 11A is connected to the output side of the current command generation unit 11.

[0091] In step S121, the command correction unit 11A corrects the current command. At this time, the command correction unit 11A generates superimposed d-axis current command Ihd* and superimposed q-axis current command Ihq* as described above, and uses these to correct the d-axis current command Id* and q-axis current command Iq* respectively, thereby superimposing the pulsation corresponding to the rotation of the motor 2 onto the d-axis current command Id* and q-axis current command Iq*.

[0092] After correcting the current command in step S121, the process shown in the flowchart in Figure 16 is terminated.

[0093] In Figure 13, an example is shown in which the command correction unit 11A corrects the d-axis current command Id* and q-axis current command Iq* generated by the current command generation unit 11 using the superimposed d-axis current command Ihd* and superimposed q-axis current command Ihq* generated by the command correction unit 11A. However, instead of correcting the d-axis current command Id* and q-axis current command Iq*, the current control unit 14 may be used to correct the d-axis voltage command Vd* and q-axis voltage command Vq* generated by the current control unit 14. In this case, instead of generating the superimposed d-axis current command Ihd* and superimposed q-axis current command Ihq*, the command correction unit 11A may generate superimposed d-axis voltage command Vhd* and superimposed q-axis voltage command Vhq* as voltage commands to superimpose pulsations corresponding to a predetermined time order onto the d-axis voltage command Vd* and q-axis voltage command Vq*, respectively. Furthermore, the generation of superimposed d-axis voltage commands Vhd* and superimposed q-axis voltage commands Vhq* can be performed, for example, based on pre-stored map information, similar to the generation of superimposed d-axis current commands Ihd* and superimposed q-axis current commands Ihq*.

[0094] According to the second embodiment of the present invention described above, in addition to the effects and advantages described in the first embodiment, the following further effects and advantages are achieved.

[0095] (9) The motor control device 1A includes a current command generation unit 11 that generates a d-axis current command Id* and a q-axis current command Iq* based on a torque command T*, a current control unit 14 that calculates a d-axis voltage command Vd* and a q-axis voltage command Vq* based on the d-axis current command Id* and the q-axis current command Iq*, and a command correction unit 11A that corrects the d-axis current command Id* and the q-axis current command Iq*, or the d-axis voltage command Vd* and the q-axis voltage command Vq*, so that a specific harmonic component is superimposed on the current flowing through the motor 2. The PWM control unit 18 is capable of performing field weakening control by generating a PWM pulse signal to weaken the magnetic flux of the motor 2. The command correction unit 11A corrects the d-axis current command Id* and q-axis current command Iq*, or the d-axis voltage command Vd* and q-axis voltage command Vq* when the motor 2 is driven in a cascading manner (step S102: Yes) and the PWM control unit 18 is performing field weakening control (step S103: Yes) (step S121). The carrier frequency adjustment unit 16 adjusts the carrier frequency fc when the motor 2 is driven in a cascading manner to be higher than the carrier frequency fc when the motor 2 is not driven in a cascading manner (step S110) when the PWM control unit 18 is not performing field weakening control (step S103: No). In this way, power loss generated during motor driving can be sufficiently reduced in both cases, whether the motor 2 is driven in a cascading manner or not, and iron loss during field weakening control can also be reduced.

[0096] (10) The specific orders mentioned above are, for example, 6th, 12th, 18th, and 24th orders, which are multiples of 6 in terms of electrical angle. In this way, it is possible to effectively reduce the order component of the iron loss for each time order when the d-axis current Id is applied to the motor 2, which is greatly affected by field weakening.

[0097] In the first and second embodiments described above, examples were given in which the motor control devices 1 and 1A control the drive of the motor 2 based on a torque command T* input from an external source. However, the drive of the motor 2 may also be controlled based on something other than a torque command T*, such as an accelerator command corresponding to the operation of the accelerator pedal by the vehicle driver, or a torque command output from an automatic driving control device that performs automatic driving control of the vehicle.

[0098] Furthermore, in the first and second embodiments described above, if the absolute value of the torque command T* is below a predetermined threshold and the d-axis current command Id* and q-axis current command Iq* output from the current command generation unit 11 can be considered to be approximately zero, the output of the PWM pulse signal from the motor control device 1,1A to the inverter 3 may be stopped. In this way, the current flowing in the motor 2 during rotational drive is rectified by the diode, thereby further reducing system losses.

[0099] Alternatively, in the first and second embodiments described above, a circuit breaker may be provided between the inverter 3 and the motor 2. If the absolute value of the torque command T* is below a predetermined threshold and the d-axis current command Id* and q-axis current command Iq* output from the current command generation unit 11 can be considered to be approximately zero, the circuit breaker may be turned off to disconnect the connection between the inverter 3 and the motor 2. In this way, current cannot flow in the motor 2 during rotational drive, thereby minimizing system losses.

[0100] (Third embodiment) Next, a third embodiment of the present invention will be described with reference to the drawings.

[0101] Figure 17 is a diagram showing the configuration of a hybrid system 72 in a third embodiment of the present invention.

[0102] As shown in Figure 17, the hybrid system 72 is composed of the motor drive system 100 (motor control device 1 or 1A, motor 2, inverter 3, rotational position detector 4, high-voltage battery 5, current detection unit 7) described in the first and second embodiments, and a similar motor drive system 101 (motor control device 1 or 1A, motor 2a, inverter 3a, rotational position detector 4a, high-voltage battery 5, current detection unit 7a). The motor drive systems 100 and 101 share the motor control devices 1 and 1A and the high-voltage battery 5.

[0103] A rotational position sensor 8a is attached to the motor 2a to detect the rotational position θa of the rotor. The rotational position detector 4a calculates the rotational position θa from the input signal of the rotational position sensor 8a and outputs it to the motor control device 1,1A. A current detection unit 7a is located between the inverter 3a and the motor 2a. The torque generated in the rotor of the motor 2a is transmitted to the outside of the motor drive system 101 from the rotating shaft fixed to the rotor.

[0104] Inverter 3a includes an inverter circuit 31a, a gate drive circuit 32a, and a smoothing capacitor 33a. The gate drive circuit 32a is connected to a motor control device 1,1A common to the gate drive circuit 32 of inverter 3, and generates a gate drive signal to control each switching element of the inverter circuit 31a based on a PWM pulse signal input from the motor control device 1,1A, and outputs it to the inverter circuit 31a. The inverter circuit 31a and the smoothing capacitor 33a are connected to a high-voltage battery 5 common to the inverter circuit 31 and the smoothing capacitor 33.

[0105] The motor control devices 1 and 1A receive a torque command T* for motor 2 and a torque command Ta* for motor 2a. Based on these torque commands, the motor control devices 1 and 1A generate PWM pulse signals to control the drive of motors 2 and 2a in the manner described in the first or second embodiment, and output them to inverters 3 and 3a, respectively. Specifically, the carrier frequency adjustment unit 16 of the motor control devices 1 and 1A adjusts the carrier frequency fc so that when motors 2 and 2a are driven in a coordinated manner, the carrier frequency fc is higher than when they are not driven in a coordinated manner. This reduces system losses. The carrier frequency adjustment unit 16 may set the carrier frequency fc to different values ​​for motors 2 and 2a.

[0106] The motor 2 is connected to the engine system 721 and the engine control unit 722. The engine system 721 is driven by the engine control unit 722, causing the motor 2 to rotate. The motor 2, rotated by the engine system 721, acts as a generator and generates AC power. The AC power generated by the motor 2 is converted to DC power by the inverter 3 and charged to the high-voltage battery 5. This allows the hybrid system 72 to function as a series hybrid system. The engine system 721 and the engine control unit 722 may also be connected to the motor 2a.

[0107] According to this embodiment, the hybrid system 72 shown in Figure 17 is realized using the motor control device 1 or motor control device 1A described in the first and second embodiments, respectively. This provides the same effect as in the first and second embodiments, by reducing system losses for both the motor drive system 100 and the motor drive system 101.

[0108] (Fourth embodiment) Next, a fourth embodiment of the present invention will be described with reference to the drawings.

[0109] Figure 18 is an external perspective view of the electromechanical unit 71 in the fourth embodiment of the present invention. The electromechanical unit 71 is composed of the motor drive system 100 (motor control device 1 or 1A, motor 2, and inverter 3) described in the first and second embodiments. Motor 2 and inverter 3 are connected at a coupling part 713 via a busbar 712. The output of motor 2 is transmitted via a gear 711 to a differential gear (not shown) and then to the axle. Although motor control devices 1 and 1A are not shown in Figure 18, they can be placed at any position.

[0110] A key feature of this electromechanical unit 71 is its integrated structure, which combines the motor 2, inverter 3, and gear 711. The electromechanical unit 71 requires a reduction in the combined system loss of the motor 2 and inverter 3. Therefore, by using the motor control device 1 or motor control device 1A described in the first and second embodiments, system losses can be reduced, thereby realizing a highly efficient electromechanical unit.

[0111] (Fifth embodiment) Next, an embodiment in which the motor drive system 100 described in the first embodiment is applied to a vehicle will be described using Figure 19.

[0112] Figure 19 is a diagram showing the configuration of a hybrid vehicle system according to a fifth embodiment of the present invention. As shown in Figure 19, the hybrid vehicle system of this embodiment has a powertrain in which motor 2 is used as a motor / generator.

[0113] In the hybrid vehicle system shown in Figure 19, a front axle 801 is rotatably supported at the front of the vehicle body 800, and front wheels 802 and 803 are provided at both ends of the front axle 801. A rear axle 804 is rotatably supported at the rear of the vehicle body 800, and rear wheels 805 and 806 are provided at both ends of the rear axle 804.

[0114] A differential gear 811, which is a power distribution mechanism, is provided in the center of the front axle 801, and distributes the rotational driving force transmitted from the engine 810 via the transmission 812 to the left and right front axles 801.

[0115] A pulley on the crankshaft of engine 810 and a pulley on the rotating shaft of motor 2 are mechanically connected via a belt.

[0116] This allows the rotational driving force of motor 2 to be transmitted to engine 810, and the rotational driving force of engine 810 to be transmitted to motor 2. Motor 2 generates rotational driving force corresponding to the three-phase AC power, when three-phase AC power output from inverter 3 is supplied to the stator coil of the stator in accordance with the control of motor control device 1 or 1A, causing the rotor to rotate.

[0117] In other words, motor 2 operates as an electric motor using three-phase AC power output from inverter 3 under the control of motor control device 1,1A, while simultaneously operating as a generator that generates three-phase AC power when the rotor rotates in response to the rotational driving force of engine 810, thereby inducing an electromotive force in the stator coils of the stator.

[0118] The inverter 3 is a power conversion device that converts DC power supplied from a high-voltage battery 5, which is a high-voltage (42V or 300V) power source, into three-phase AC power. It controls the three-phase AC current flowing through the stator coil of the motor 2 according to the operating command value and the magnetic pole position of the rotor.

[0119] The three-phase AC power generated by motor 2 is converted to DC power by inverter 3 to charge the high-voltage battery 5. The high-voltage battery 5 is electrically connected to the low-voltage battery 823 via DC-DC converter 824. The low-voltage battery 823 constitutes the low-voltage (14V) power supply system of the automobile and is used to power the starter 825 for initial starting (cold start) of the engine 810, as well as the radio, lights, etc.

[0120] When the vehicle is stopped, such as at a traffic light (idle stop mode), the engine 810 is stopped. When restarting the engine 810 (hot start) upon re-entry, the inverter 3 drives the motor 2 to restart the engine 810. However, in idle stop mode, if the high-voltage battery 5 is not sufficiently charged or if the engine 810 is not sufficiently warmed up, the engine 810 will not stop and will continue to run. Also, in idle stop mode, it is necessary to secure a power source for auxiliary equipment that uses the engine 810 as a power source, such as the air conditioner compressor. In this case, the motor 2 is driven to power the auxiliary equipment.

[0121] Even in acceleration mode or high-load driving mode, the motor 2 is driven to assist the engine 810. Conversely, when the vehicle is in charging mode where the high-voltage battery 5 needs to be charged, the engine 810 generates electricity for the motor 2 to charge the high-voltage battery 5. In other words, it performs regenerative braking and deceleration modes.

[0122] In the hybrid vehicle system shown in Figure 19, which is realized using the motor drive system 100 described in the first and second embodiments, the motor control devices 1 and 1A adjust the carrier frequency fc so that when the motor 2 is driven in conjunction with other motors, the carrier frequency fc is higher than when it is not driven in conjunction with other motors. This makes it possible to reduce system losses.

[0123] In each of the embodiments described above, the functions of each component within the motor control devices 1 and 1A (Figures 2 and 13, etc.) may be realized by a CPU and a program, rather than by hardware. Realizing the components within the motor control devices 1 and 1A using a CPU and a program has the advantage of reducing the cost because the number of hardware components is reduced. Furthermore, this program can be provided in advance by being stored in the storage medium of the inverter control device. Alternatively, the program can be provided by being stored in an independent storage medium, or it can be recorded and stored in the storage medium of the inverter control device via a network connection. It may also be supplied as a computer program product that can be read by a computer in various forms, such as data signals (carrier waves).

[0124] It should be noted that the present invention is not limited to the embodiments described above, and various modifications are possible without departing from the spirit of the invention. [Explanation of symbols]

[0125] 1,1A...Motor control device, 2...Motor, 3...Inverter, 4...Rotation position detector, 5...High-voltage battery, 7...Current detection unit, 8...Rotation position sensor, 11...Current command generation unit, 11A...Command correction unit, 11B...Switching unit, 12...Speed ​​calculation unit, 13...Current conversion unit, 14...Current control unit, 15...Three-phase voltage conversion unit, 16...Carrier frequency adjustment unit, 17...Carrier wave generation unit, 18...PWM control unit, 31...Inverter circuit, 32...Gate drive circuit, 33...Smoothing key 71...Mechatronics unit, 72...Hybrid system, 100, 101...Motor drive system, 711...Gear, 712...Bus bar, 713...Coupling, 800...Body, 801...Front axle, 802...Front wheel, 803...Front wheel, 804...Rear axle, 805...Rear wheel, 806...Rear wheel, 810...Engine, 811...Differential gear, 812...Transmission, 823...Low-voltage battery, 824...DC-DC converter, 825...Starter

Claims

1. A motor control device that is connected to an inverter that converts DC power to AC power and outputs it to a motor, and controls the operation of the inverter in accordance with a torque command, thereby controlling the drive of the motor using the inverter, A carrier wave generation unit that generates a carrier wave, A carrier frequency adjustment unit adjusts the carrier frequency, which is the frequency of the carrier wave. The system includes a PWM control unit that uses the carrier wave to pulse-width modulate a voltage command and generates a PWM pulse signal for controlling the operation of the inverter, The carrier frequency adjustment unit is a motor control device that adjusts the carrier frequency such that the carrier frequency when the motor is driven in a co-rotational manner is higher than the carrier frequency when the motor is not driven in a co-rotational manner.

2. In the motor control device according to claim 1, The carrier frequency adjustment unit compares the absolute value of the torque command with a predetermined threshold, and determines that the motor is being driven in a co-rotation manner if the absolute value of the torque command is less than or equal to the threshold.

3. In the motor control device according to claim 2, The threshold value is determined based on the results of a prior electromagnetic field analysis simulation or experiment in the motor control device.

4. In the motor control device according to claim 2, A motor control device that stops outputting the PWM pulse signal to the inverter when the absolute value of the torque command is less than or equal to the threshold value.

5. In the motor control device according to claim 2, A motor control device that disconnects the connection between the inverter and the motor when the absolute value of the torque command is less than or equal to the threshold.

6. In the motor control device according to claim 1, A motor control device in which the carrier frequency when the motor is being driven in a co-rotational manner is determined based on at least one of the processing load of the motor control device and the capacity of the gate power supply that supplies power to the gate drive circuit of the inverter.

7. In the motor control device according to claim 1, A motor control device that controls the drive of the motor such that the induced voltage generated by the rotation of the motor is less than the withstand voltage of the switching element of the inverter.

8. In the motor control device according to claim 1, The system includes a current control unit that calculates the voltage command at predetermined calculation cycles, The carrier frequency adjustment unit is a motor control device that adjusts the carrier frequency when the motor is driven in a cascading manner such that the calculation period is longer than half the period of the carrier wave.

9. In the motor control device according to claim 1, The carrier frequency adjustment unit is a motor control device that adjusts the carrier frequency so that the rate of change of the carrier frequency is less than or equal to a predetermined value.

10. In the motor control device according to claim 1, A current command generation unit that generates a current command based on the torque command, A current control unit that calculates the voltage command based on the current command, The system includes a command correction unit that corrects the current command or the voltage command so that a specific harmonic component is superimposed on the current flowing through the motor. The PWM control unit is capable of performing field weakening control, which generates the PWM pulse signal to weaken the magnetic flux of the motor. The command correction unit corrects the current command or the voltage command when the motor is driven in a co-rotational manner and the PWM control unit is performing the field weakening control. The carrier frequency adjustment unit is a motor control device that adjusts the carrier frequency such that, when the PWM control unit is not performing the field weakening control, the carrier frequency when the motor is driven in a co-rotational manner is higher than the carrier frequency when the motor is not driven in a co-rotational manner.

11. In the motor control device according to claim 10, The aforementioned specific order is a motor control device whose order is a multiple of 6 in terms of electrical angle.

12. A motor control device according to any one of claims 1 to 11, The inverter connected to the motor control device, The motor driven by the inverter, A hybrid system comprising an engine system connected to the aforementioned motor.

13. A motor control device according to any one of claims 1 to 11, The inverter connected to the motor control device, The motor driven by the inverter, The motor comprises a gear that transmits the rotational driving force of the motor, A mechatronic unit comprising the motor, the inverter, and the gear in an integrated structure.

14. A motor control device according to any one of claims 1 to 11, The inverter connected to the motor control device, The motor is driven by the inverter, An electric vehicle system that moves using the rotational driving force of the aforementioned motor.

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