Control device for inverter, control method for inverter, and electric vehicle equipped with said control device

Abstract

In order to constantly remain stable during a transient response at the time of regenerative braking by an electric vehicle using a DC power source and increase regenerative electric power, this control device for an inverter which converts, into AC power, DC power supplied from the DC power source and smoothed via a smoothing capacitor to drive a motor comprises: a generation unit for generating a first motor current operation amount by extracting an AC component of a smoothing capacitor voltage detected from the smoothing capacitor; a generation unit for generating a second motor current command value according to the smoothing capacitor voltage; a generation unit for generating a second motor current operation amount by smoothing the difference between a set first motor current command value and the second motor current command value; and an update unit for updating the first motor current command value by adding the sum of the first motor current operation amount and the second motor current operation amount to the first motor current command value. The control device for the inverter controls an output voltage of the inverter on the basis of a motor current detection value of the motor and the updated first motor current command value.

Description

Inverter control device, inverter control method, and electric vehicle equipped with said control device

[0001] The present invention relates to an inverter control device and an inverter control method, and is particularly suitable for controlling an inverter mounted on an electric vehicle in a DC overhead line.

[0002] There are two types of overhead line systems for supplying power to electric vehicles: AC overhead lines and DC overhead lines. Of these, the DC overhead line system is widely used, mainly in urban areas with high vehicle traffic density, due to its advantage of simplifying the design of vehicle-side equipment. DC substations convert the three-phase AC power supplied by the power company into DC power, mainly using diode rectifiers. As a result, the regenerative energy generated by the braking of electric vehicles cannot be returned to the power company, and in most cases is consumed by powered vehicles traveling nearby.

[0003] However, if there are no powered trains nearby, or if the regenerative power generated by braking is greater than the power consumed by the powered train, the regenerative power cannot be fully consumed, resulting in an increase in overhead line voltage. The stopping of inverter operation due to high overhead line voltage is called regeneration lapse, and once regeneration lapse occurs, all of the regenerative energy is wasted. To prevent this regeneration lapse, a light-load regeneration control technology is used that adjusts the regenerative power according to the power consumed by the powered train.

[0004] There are two main methods for this light-load regenerative control. The first method is a method that configures a feedback control system that throttles the motor current command according to the amount of excess when the overhead line voltage or smoothing capacitor voltage (hereinafter, both are collectively referred to as DC voltage) exceeds a reference voltage. This feedback control system is generally configured using proportional-integral control (PI control). For example, the technology described in Patent Document 1 (JP-A-2005-102626) allows the DC voltage to be increased to the reference voltage level, thereby increasing regenerative power. Furthermore, a feedback control system configured using PI control has two adjustment parameters, a P-term gain and an I-term gain, which facilitates adjustment. For example, the P-term gain is adjusted to address transient response, such as a sudden change in overhead line voltage, while the I-term gain is adjusted to address the need to quickly converge the deviation between the DC voltage and the reference voltage or to address severe overshoot.

[0005] On the other hand, there is a problem that DC voltage detection errors between inverters in the same train set can cause imbalances in the amount of motor current throttling. In particular, even slight variations in the detected values ​​can result in extreme motor current differences such as a 100:0 ratio (current flows from one inverter but not the other). Within the same train set, the technology described in Patent Document 1 can be used to correct imbalances by exchanging information between inverters. However, this is impossible for rolling stock that does not have a means of transmitting information within the train set, and it is also impossible to correct imbalances between train sets that are physically separated and do not have a means of communication.

[0006] The second method sets a derating characteristic that defines an upper limit or throttling coefficient of the motor current according to the DC voltage, and when the DC voltage exceeds a first reference voltage (throttling start voltage), the motor current command begins to be throttled according to the derating characteristic, and when it exceeds a second reference voltage (throttling end voltage), it is set to zero.As an example, the technology described in Patent Document 2 has the advantage that even if there is no information transmission means within the train set, it is possible to achieve balance between other inverters, and if the derating characteristic is the same, it is possible to achieve balance between other physically distant train sets.

[0007] JP 2001-157303 A JP 2013-70611 A

[0008] The technology described in Patent Document 2 has two problems. First, adjustment is extremely difficult. This is because the change in motor current in response to a change in DC voltage is determined by the derating characteristics, and there are virtually no adjustment parameters for transient response. Second, because the DC voltage is balanced between a first reference voltage (voltage at which voltage throttling begins) and a second reference voltage (voltage at which voltage throttling ends), regenerative power is reduced compared to the technology described in Patent Document 1, which can increase the DC voltage to the reference voltage level.

[0009] Therefore, an object of the present invention is to provide a technology that can always maintain a stable state even with respect to a transient response during regenerative braking and increase regenerative power.

[0010] In order to solve the above problems, one representative inverter control device according to the present invention is a control device for an inverter that converts DC power supplied from a DC power source and smoothed through a smoothing capacitor into AC power to drive an electric motor, and includes: a first control input generator that extracts an AC component of a smoothing capacitor voltage detected from the smoothing capacitor to generate a first motor current control input; a command value generator that generates a second motor current command value in accordance with the smoothing capacitor voltage; a second control input generator that smooths a difference between a set first motor current command value and a set second motor current command value to generate the second motor current control input; and a command value update unit that adds the sum of the first motor current control input and the second motor current control input to the first motor current command value to update the first motor current command value, and controls the output voltage of the inverter based on a detected motor current value of the electric motor and the updated first motor current command value.

[0011] According to the present invention, while the second motor current command value corresponding to the DC voltage is followed in a steady state, the first motor current control amount and the second motor current control amount are used for transient stabilization, and in addition, when the second motor current command value is generated from the constant stabilization state, the interval between the first reference value and the second reference value is narrowed, thereby making it possible to increase regenerative power. Problems, configurations, and effects other than those described above will become apparent from the description of the following embodiments.

[0012] 1 is a block diagram showing the overall configuration of an inverter and a control device for an inverter according to the present invention. FIG. 2 is a block diagram showing the schematic configuration of the light load regeneration control shown in FIG. 1. FIG. 3 is a block diagram showing the detailed configuration of a light load regeneration control unit according to a first embodiment of the present invention, i.e., the details of the schematic configuration shown in FIG. 2. FIG. 4 is a diagram showing the characteristics of a generation unit for determining the maximum q-axis current shown in FIG. 3. FIG. 5 is a block diagram showing the detailed configuration of a light load regeneration control unit according to a second embodiment of the present invention, i.e., the details of the schematic configuration shown in FIG. 5. FIG. 6 is a diagram showing the characteristics of a generation unit for determining the maximum power supply current shown in FIG. 7. FIG. 8 is a diagram showing operation waveforms when the modulation rate is 100% in the first embodiment. FIG. 9 is a diagram showing operation waveforms when the modulation rate is 50% in the first embodiment. FIG. 10 is a diagram showing operation waveforms when the modulation rate is 100% in the second embodiment. FIG. 11 is a diagram showing operation waveforms when the modulation rate is 50% in the second embodiment.

[0013] Hereinafter, as modes for carrying out the present invention, Examples 1 and 2 will be described with reference to the drawings. Note that the present invention is not limited to these Examples. In addition, in the description of the drawings, the same parts are denoted by the same reference numerals.

[0014] 1 is a block diagram showing the overall configuration of an inverter and an inverter control device according to the present invention. DC power supplied by a DC voltage source (DC power supply), not shown, is stabilized by a smoothing capacitor 1. Examples of the DC voltage source (DC power supply) include overhead lines that supply power to electric vehicles and storage batteries (not shown). Semiconductor switching elements 2 and 3, 4 and 5, and 6 and 7 are connected in series to form a three-phase bridge circuit as an inverter, which is connected in parallel to the smoothing capacitor 1. The AC output terminals of this three-phase bridge circuit (inverter) are connected to an AC motor 8.

[0015] The sensors provided are a voltage sensor 9 that detects the terminal voltage Ecf of the smoothing capacitor 1 and a current sensor 10 that detects the phase currents iu, iv, and iw of the AC motor 8. The conduction states of the semiconductor switching elements 2 to 7 that form a three-phase bridge circuit as an inverter are controlled by gate signals Gpu, Gnu, Gpv, Gnv, Gpw, and Gnw from a control device 11.

[0016] The control device 11 is given a d-axis current command value Id* and a q-axis current command value Iq* as command values, and is composed of a coordinate conversion unit 12, a current control unit 13, a PWM control unit 14, a light load regeneration control unit 15, and an adder 16, and controls the inverter. When the AC motor 8 is mounted on an electric vehicle to drive the electric vehicle, the control device 11 for this inverter is also mounted on the electric vehicle together with the inverter.

[0017] The coordinate conversion unit 12 generates a d-axis current detection value Id and a q-axis current detection value Iq from the phase currents iu, iv, and iw detected by the current sensor 10. The light-load regenerative control unit 15 generates a q-axis current manipulation variable ΔIq* from the DC voltage Ecf detected by the voltage sensor 9 and the q-axis current command value Iq*. The adder 16 adds the q-axis current manipulation variable ΔIq* generated by the light-load regenerative control unit 15 to the q-axis current command value Iq*, and updates the q-axis current command value Iq* to a new q-axis current command value Iq**.

[0018] The current control unit 13 generates a d-axis voltage Vd and a q-axis voltage Vq from the d-axis current command value Id*, the new q-axis current command value Iq**, the d-axis current detection value Id, and the q-axis current detection value Iq. The PWM control unit 14 generates gate signals Gpu, Gnu, Gpv, Gnv, Gpw, and Gnw from the d-axis voltage Vd and the q-axis voltage Vq.

[0019] Fig. 2 is a block diagram showing a schematic configuration of the light-load regenerative control unit 15 shown in Fig. 1. The light-load regenerative control unit 15 is composed of a high-pass filter 21 that extracts the AC component of the DC voltage Ecf, a generation unit 22 that generates a maximum q-axis current Iqmax according to the DC voltage Ecf, a subtractor 23 that calculates the difference between the maximum q-axis current Iqmax and the q-axis current command value Iq*, a low-pass filter 24 that smoothes the output of the subtractor 23, and an adder 25 that calculates the q-axis current manipulated variable ΔIq* from the sum of the outputs of the high-pass filter 21 and the low-pass filter 24.

[0020] Fig. 3 is a block diagram showing the detailed configuration of the light load regenerative control unit 15 according to the first embodiment of the present invention, i.e., the details of the schematic configuration shown in Fig. 2. The high-pass filter 21 that extracts the AC component of the DC voltage Ecf is made up of a first-order high-pass filter 31 with a low cutoff angular frequency ωp, a multiplier 32 that calculates the product of the output of the first-order high-pass filter 31 and a gain g, and a multiplier 33 that multiplies the output of the multiplier 32 by Kp.

[0021] Low-pass filter 24, which smoothes the output of subtractor 23, is composed of subtractor 34, multiplier 35, multiplier 36, and integrator 37. Subtractor 34 calculates the difference between the output of subtractor 23 and the output of integrator 37, multiplier 35 calculates the product of the output of subtractor 34 and gain g, multiplier 36 multiplies the output of multiplier 35 by Ki, and integrator 37 integrates the output of multiplier 36. During this integration, the integral value is limited by a lower limit of 0 and an upper limit of the range of the absolute value |Iq*| of the q-axis current command value.

[0022] 4 is a diagram showing the characteristics of the generator 22 that calculates the maximum q-axis current Iqmax shown in FIG. 3. Here, the q-axis current Iq is throttled down as the DC voltage Ecf increases, which is also called the derating characteristic. When the DC voltage Ecf is equal to or less than the throttle-down start voltage Er0, Iqmax is constant at Iqr = Iqr. When the DC voltage Ecf is equal to or greater than the throttle-down end voltage Er1, Iqmax = 0. Between Er0 and Er1, the rate of change (slope) is |Iqr| / (Er1-Er0), with "Iqr" and "0" connected.

[0023] Next, a description will be given of the settings of various constants in Example 1. Although light load regenerative control is originally a control for adjusting power, it is a control system that commands a q-axis current command, i.e., torque, and therefore the control response varies depending on the speed or DC voltage or output voltage.

[0024] To prevent this, the gain g is set to the following [Equation 1], where DC voltage Ecf, number of motors driven collectively by one inverter Nm, and q-axis voltage Vq. [Equation 1] g = Ecf / (3 Nm Vq)

[0025] The constant Kp of the multiplier 33 is expressed by the following [Equation 2], where C [F] is the capacitance of the smoothing capacitor 1 and ωd [rad / s] is the desired control response. [Equation 2] Kp=ωd·C

[0026] On the other hand, the constant Ki of the multiplier 36 is expressed by the following [Equation 3] for the desired integral response ωi [rad / s]. According to this, the integral response ωi can be regarded as being set in accordance with (Er1-Er0) / |Iqr|, that is, the reciprocal of the rate of change (slope, |Iqr| / (Er1-Er0)) when calculating the maximum q-axis current value Iqmax shown in FIG. 4. [Equation 3] Ki=Kp・ωi・(Er1-Er0) / |Iqr| Here, the relationship between the two responses ωd and ωi must be ωd≧ωi.

[0027] Next, it will be explained how stabilization of control can be achieved by using the above-described configuration and constant settings in Example 1. High-frequency components of the DC voltage Ecf pass through the high-pass filter 21 but cannot pass through the low-pass filter 24. As a result, stabilization control that suppresses oscillations in the DC voltage Ecf is achieved.

[0028] On the other hand, the low frequency components of the DC voltage Ecf cannot pass through the high pass filter 21 but pass through the low pass filter 24. Here, since the gain of the low pass filter 24 is 1, the steady-state value of the q-axis current manipulated variable ΔIq* is expressed by the following [Equation 4]: ΔIq*=Iqmax−Iq*

[0029] In this case, the new q-axis current command value Iq** in the control device 11 of Fig. 1 is expressed by the following [Equation 5], and steadily follows the derating characteristics shown in Fig. 4. [Equation 5] Iq** = Iq* + ΔIq* = Iqmax As described above, it is possible to steadily follow the desired derating characteristics while also achieving stabilization control that suppresses oscillations in the DC voltage Ecf.

[0030] Furthermore, when trying to increase regenerative power, it is generally sufficient to reduce the difference between the throttling start voltage Er0 and the end voltage Er1. However, if this difference, Er1 - Er0, is reduced, the amount of fluctuation in Iqmax (δIqmax) in response to a slight fluctuation in DC voltage (δEcf) becomes large, resulting in a so-called oversensitive characteristic and making it difficult to stabilize control. This is clear from the relational expression [Equation 6] below. [Equation 6] δIqmax = |Iqr| / (Er1 - Er0) × δEcf

[0031] However, in the first embodiment, the constant Ki is proportional to (Er1-Er0) according to the above formula 3, so the two are cancelled out, and fluctuations in the output of the low-pass filter 24 itself are suppressed. This allows for stable control. As a result, there is also the advantage that regenerative power can be increased.

[0032] Next, the operating waveforms in Example 1 will be described with reference to Fig. 7 and Fig. 8. Fig. 7 is a diagram showing the operating waveforms in Example 1 when the inverter modulation factor Ymr is 100% (i.e., the output voltage is 100%), and Fig. 8 is a diagram showing the operating waveforms in Example 1 when the inverter modulation factor Ymr is 50% (i.e., the output voltage is 50%).

[0033] In both Figures 7 and 8, the load that absorbs regenerative power is simulated by a resistor RL, which is changed in a stepwise manner. The two inverters G1 and G2 are inverters within the same train set, and are operated in parallel using a common power supply voltage, such as overhead line voltage. However, an error is intentionally introduced into the DC voltage detection value, so that the G1 side detects a lower value and the G2 side detects a higher value.

[0034] As a result, the q-axis current command value Iq** is more reduced on the G2 side, which detects a higher value. The q-axis current command value Iq** differs between the inverters G1 and G2, causing a so-called imbalance, but this is not a complete imbalance and settles at a value corresponding to the error in the DC voltage detection value. This characteristic itself is due to the derating characteristic.

[0035] Therefore, as shown in the waveforms at the bottom of Figures 7 and 8, attention will be focused on the transient response (sudden change in regenerative load) when the load resistance RL is changed in a stepwise manner. Although the DC voltage Ecf temporarily overshoots or undershoots, it converges stably in both Figures 7 and 8. However, strictly speaking, the convergence in Figure 8 is faster than that in Figure 7. Originally, the design was such that a constant response could be obtained regardless of the output voltage, but because the gain g is applied to the integral response ωi rather than the gain of the low-pass filter 24, the responses are not exactly the same.

[0036] Furthermore, when focusing on the DC voltage Ecf, although the load resistance values ​​are the same in Figures 7 and 8, the DC voltage Ecf in Figure 7 is higher than that in Figure 8. This also means that the regenerative power is large because it is proportional to the square of the DC voltage. The maximum regenerative power is normally determined by the magnitude of the load resistance RL, but in Example 1, it has the characteristic of fluctuating depending on the speed or output voltage (or modulation factor Ymr).

[0037] A second embodiment of the present invention aims to solve the problem of the regenerative power varying depending on the speed or the output voltage (or the modulation factor Ymr) in the first embodiment. Fig. 5 is a block diagram showing a detailed configuration of a light load regenerative control unit 15 according to the second embodiment of the present invention (i.e., the detailed configuration of Fig. 2).

[0038] In the second embodiment, the high-pass filter 21 that extracts the AC component of the DC voltage Ecf and the subtractor 23 that calculates the difference between the maximum q-axis current Iqmax and the specified q-axis current value Iq* are configured in the same manner as in the first embodiment shown in FIG. 3 .

[0039] The generating unit 22 that generates the maximum q-axis current Iqmax in accordance with the DC voltage Ecf is composed of a generating unit 42 that generates the maximum power supply current Ismax for a DC power supply such as an overhead line in accordance with the DC voltage Ecf, and a multiplier 45 that calculates the product of the maximum power supply current Ismax and a gain g.

[0040] Low-pass filter 24, which smooths the output of subtractor 23, is composed of subtractor 34, multiplier 46, and integrator 37. Subtractor 34 finds the difference between the output of subtractor 23 and the output of integrator 37, multiplier 46 multiplies the output of subtractor 34 by Ki', and integrator 37 integrates the output of multiplier 46. During this integration, the integral value is limited by a lower limit of 0 and an upper limit of the range of the absolute value |Iq*| of the q-axis current command value.

[0041] 6 is a diagram showing the characteristics of the generator 42 that calculates the maximum power supply current Ismax shown in FIG. 5. This is also called a derating characteristic because the power supply current Is is reduced as the DC voltage Ecf increases. When the DC voltage Ecf is equal to or less than the throttling start voltage Er0, Ismax is set to a constant value of Isr, and when the DC voltage Ecf is equal to or greater than the throttling end voltage Er1, Ismax is set to 0. Between Er0 and Er1, the slope is |Isr| / (Er1-Er0).

[0042] Next, a description will be given of the settings of various constants in the embodiment 2. The gain g and the constant Kp of the multiplier 33 are the same as in the embodiment 1, and are expressed by the above [Equation 1] and [Equation 2], respectively.

[0043] The constant Ki' of the multiplier 46 is expressed by the following [Equation 7] for the desired integral response ωi [rad / s]: [Equation 7] Ki'=Kp·ωi·(Er1−Er0) / |Isr| Here, the relationship between the two control responses ωd ([Equation 2]) and ωi must be ωd≧ωi, as in the first embodiment.

[0044] Next, in Example 2, it will be described how stabilization of control can be achieved by using the above-described configuration and constant settings. High-frequency components of the DC voltage Ecf pass through the high-pass filter 21 but cannot pass through the low-pass filter 24. As a result, the control operates as a stabilization control that suppresses oscillations of the DC voltage Ecf.

[0045] On the other hand, the low frequency components of the DC voltage Ecf cannot pass through the high pass filter 21 but pass through the low pass filter 24. Here, since the gain of the low pass filter 24 is 1, the steady-state value of the q-axis current manipulated variable ΔIq* is given by the following [Equation 8] [Equation 8] ΔIq*=g·Ismax−Iq*

[0046] Then, the new q-axis current command value Iq** in the control device 11 in Fig. 1 is given by the following [Equation 9], and steadily follows the derating characteristics shown in Fig. 6. [Equation 9] Iq** = Iq* + ΔIq = g·Ismax As described above, similar to the first embodiment, it is possible to steadily follow the desired derating characteristics while also achieving stabilization control that suppresses oscillations in the DC voltage Ecf.

[0047] Furthermore, when trying to increase regenerative power, it is generally sufficient to reduce the difference between the throttling start voltage Er0 and the end voltage Er1. However, if this difference, Er1 - Er0, is reduced, the amount of fluctuation in Ismax (δIsmax) relative to a slight fluctuation in DC voltage (δEcf) increases, resulting in so-called oversensitive characteristics and making it difficult to stabilize control. This is clear from the relational expression [Equation 10] below. [Equation 10] δIsmax = |Isr| / (Er1 - Er0) × δEcf

[0048] However, in the second embodiment, the constant Ki' is proportional to (Er1-Er0) according to [Equation 7], so the two are cancelled out, and fluctuations in the output of the low-pass filter 24 itself are suppressed. This makes it possible to stabilize control. As a result, the second embodiment also has the advantage of being able to increase regenerative power.

[0049] Next, the operating waveforms in Example 2 will be described with reference to Fig. 9 and Fig. 10. Fig. 9 is a diagram showing the operating waveforms in Example 2 when the modulation factor Ymr = 100% (i.e., the output voltage is 100%), and Fig. 10 is a diagram showing the operating waveforms in Example 2 when the modulation factor Ymr = 50% (i.e., the output voltage is 50%).

[0050] In both Figures 9 and 10, the load absorbing regenerative power is simulated by a resistor RL and changed in a stepwise manner. The two inverters G1 and G2 are in the same train and operate in parallel using a common power supply voltage, such as overhead line voltage. However, an error is intentionally introduced into the detected DC voltage, so that the G1 side detects a lower voltage and the G2 side detects a higher voltage. As a result, the G2 side, which detects a higher voltage, narrows the q-axis current command value Iq** more. The q-axis current command values ​​Iq** of the inverters G1 and G2 differ, resulting in a so-called imbalance. However, this is not a complete imbalance, but rather settles to a value corresponding to the error in the detected DC voltage. This characteristic itself is due to the derating characteristic. Here, under conditions where the modulation factor Ymr (i.e., output voltage) is low, the difference in imbalance is smaller in Example 1 than in Example 2.

[0051] Therefore, attention will be focused on the transient response (sudden change in regenerative load) when the load resistance RL is changed stepwise, as shown in the waveforms at the bottom of each of Figures 9 and 10. The DC voltage Ecf temporarily overshoots or undershoots, but stabilizes and converges in both Figures 9 and 10. However, while there is a difference between the response waveforms of the transient responses shown in Figures 7 and 8 in Example 1, the transient response waveforms shown in Figures 9 and 10 in Example 2 are almost the same.

[0052] Furthermore, when focusing on the DC voltage Ecf, the DC voltage Ecf in Figures 9 and 10 is approximately equal except during the rise of the q-axis current command Iq**, which means that the regenerative power is also approximately equal. Compared to the first embodiment (Figures 7 and 8), the second embodiment can increase the regenerative power even under conditions (Figure 10) where the modulation factor Ymr (i.e., the output voltage) is low.

[0053] Although the first and second embodiments have been described above as modes for carrying out the present invention, the present invention is not limited to these embodiments, and various modifications are possible within the scope of the present invention.

[0054] 1...Smoothing capacitor, 2 to 7...Semiconductor switching element, 8...AC motor, 9...Voltage sensor, 10...Current sensor, 11...Control device, 12...Coordinate conversion unit, 13...Current control unit, 14...PWM control unit, 15...Light load regeneration control unit, 21...High-pass filter, 22...Generation unit that generates a maximum q-axis current Iqmax according to a DC voltage Ecf, 23, 34...Subtractor, 24...Low-pass filter, 25...Adder, 31...First-order high-pass filter, 32, 33, 35, 36, 45, 46...Multiplier, 37...Integrator, 42...Generation unit that generates a maximum power supply current Ismax according to a DC voltage Ecf

Claims

1. A control device for an inverter that drives a motor by converting DC power supplied from a DC power source and smoothed through a smoothing capacitor into AC power, comprising: a first control input generation unit that extracts an AC component of a smoothing capacitor voltage detected from the smoothing capacitor to generate a first motor current control input; a command value generation unit that generates a second motor current command value in accordance with the smoothing capacitor voltage; a second control input generation unit that smooths a difference between a set first motor current command value and the second motor current command value to generate a second motor current control input; and a command value update unit that adds the sum of the first motor current control input and the second motor current control input to the first motor current command value to update the first motor current command value, and the inverter control device controls an output voltage of the inverter based on a motor current detection value of the motor and the updated first motor current command value.

2. An inverter control device according to claim 1, wherein the first control variable generation unit generates the first motor current control variable by adjusting the extracted AC component in accordance with the smoothing capacitor voltage and the inverter output voltage.

3. An inverter control device according to claim 1 or 2, wherein the command value generating unit sets the second motor current command value to a constant value when the smoothing capacitor voltage is equal to or less than a first reference value, to zero when the smoothing capacitor voltage is equal to or greater than a second reference value, and to a value that narrows from the constant value to zero at a constant rate of change when the smoothing capacitor voltage is between the first reference value and the second reference value.

4. An inverter control device according to claim 3, wherein the second operation amount generating unit performs the smoothing using an integrator, and adjusts the integral response of the integrator in accordance with the smoothing capacitor voltage and the inverter output voltage.

5. An inverter control device according to claim 4, wherein the second operation amount generating unit sets the integral response according to the reciprocal of the constant rate of change used in the command value generating unit.

6. An inverter control device according to claim 1 or 2, wherein the command value generating unit sets the power supply current command value of the DC power supply to a constant value when the smoothing capacitor voltage is equal to or less than a first reference value, to zero when the smoothing capacitor voltage is equal to or greater than a second reference value, and to a value that narrows from the constant value to zero at a constant rate of change between the first reference value and the second reference value, and further adjusts the power supply current command value in accordance with the smoothing capacitor voltage and the output voltage of the inverter to generate the second motor current command value.

7. An inverter control device according to claim 6, wherein the second operation amount generating unit performs the smoothing using an integrator, and sets the integral response of the integrator according to the reciprocal of the constant rate of change used in the command value generating unit.

8. An electric vehicle equipped with an inverter control device according to any one of claims 1 to 7.

9. A control method for an inverter that drives a motor by converting DC power supplied from a DC power source and smoothed through a smoothing capacitor into AC power, the control method comprising: extracting an AC component of a smoothing capacitor voltage detected from the smoothing capacitor to generate a first motor current control amount; generating a second motor current command value in accordance with the smoothing capacitor voltage; smoothing a difference between a set first motor current command value and the second motor current command value to generate a second motor current control amount; adding the sum of the first motor current control amount and the second motor current control amount to the first motor current command value to update the first motor current command value; and controlling an output voltage of the inverter based on a motor current detection value of the motor and the updated first motor current command value.

10. A method for controlling an inverter according to claim 9, characterized in that, when generating the first motor current control variable, the extracted AC component is adjusted in accordance with the smoothing capacitor voltage and the inverter output voltage to generate the first motor current control variable.

11. A method for controlling an inverter as claimed in claim 9 or 10, characterized in that when generating the second motor current command value, the second motor current command value is generated as a constant value when the smoothing capacitor voltage is equal to or less than a first reference value, zero when the smoothing capacitor voltage is equal to or greater than a second reference value, and a value that narrows from the constant value to zero at a constant rate of change when the smoothing capacitor voltage is between the first reference value and the second reference value.

12. A method for controlling an inverter as claimed in claim 9 or 10, characterized in that, when generating the second motor current command value, the power supply current command value of the DC power supply is set to a constant value when the smoothing capacitor voltage is equal to or less than a first reference value, to zero when the smoothing capacitor voltage is equal to or greater than a second reference value, and to a value that narrows from the constant value to zero at a constant rate of change between the first reference value and the second reference value, and further, the power supply current command value is adjusted in accordance with the smoothing capacitor voltage and the output voltage of the inverter to generate the second motor current command value.

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