Dual-polarization low-profile phased array antenna based on miniaturized reflective phase shifter

Abstract

The invention discloses a dual-polarization low-profile phased array antenna based on a miniaturized reflection type phase shifter, which belongs to the field of wireless communication, and uses the miniaturized reflection type phase shifter which is independently developed, small in size, low in cost, low in power consumption and low in insertion loss to replace a traditional digital phase shifter. The digital phase shifter is designed based on a cross-direction coupling line coupler, and compared with a branch line coupler, the cross-direction coupling line coupler has more compact size and wider bandwidth; compared with a Lange coupler, the cross-direction coupling line coupler is more compact in size and lower in manufacturing difficulty; the phase shifter can realize 360-degree continuous phase shift within the bandwidth, and has the advantages of small size, low insertion loss, wide bandwidth and the like; therefore, the phase shift between the array units is regulated and controlled through the direct current bias voltage, the phased array antenna provided by the invention can realize + / -45-degree beam scanning on a two-dimensional plane, and can realize the functions of beam directional emission, beam continuous scanning and the like in a wireless communication system.

Description

Technical Field

[0001] The present invention belongs to the field of wireless communications, and more particularly, relates to a dual-polarization low-profile phased array antenna based on a miniaturized reflective phase shifter. Background Art

[0002] With the development of fifth-generation (5G) mobile communications and broadband satellite networks, phased array antennas have become recognized as a key technology. They are widely used in military radar, measurement, and civilian satellite communications. Traditional phased array antennas offer advantages such as fast beamforming, continuous scanning, low sidelobe levels, and a wide scanning angle. However, they typically use a large number of high-precision digital phase shifters to achieve beamforming. Digital phase shifters are expensive and consume a lot of power. Their extensive use in phased array antennas results in high manufacturing and maintenance costs. Disadvantages such as high complexity, high production costs, difficulty in miniaturization, and difficult maintenance make traditional phased array antennas difficult to promote and apply in commercial applications. Summary of the Invention

[0003] In response to the above-mentioned defects or improvement needs of the existing technology, the present invention provides a dual-polarized low-profile phased array antenna based on a miniaturized reflective phase shifter, thereby solving the technical problems of traditional phased array antennas such as high complexity, high production cost, difficulty in miniaturization and difficult maintenance.

[0004] To achieve the above objectives, according to a first aspect of the present invention, a dual-polarization low-profile phased array antenna based on a miniaturized reflective phase shifter is provided, comprising: N phase shifter groups, a power division network, N dual-polarization array elements, and first and second input ports;

[0005] Each phase shifter group includes two phase shifters; the output ends of the two phase shifters of the nth phase shifter group are respectively connected to the two input ports of the nth dual-polarization array unit in a one-to-one correspondence; N≥2, n=1,2,…,N;

[0006] The power division network includes a first power division network and a second power division network; the first and second power division networks are respectively used to equally divide the radio frequency signal input through the first and second input ports into N signals of equal amplitude and phase; the nth signal equally divided by the first power division network is inputted one-to-one to the input port of one of the phase shifters connected to the nth dual-polarization array unit; the nth signal equally divided by the second power division network is inputted one-to-one to the input port of another phase shifter connected to each dual-polarization array unit;

[0007] The phase shifter includes: a cross-coupled line coupler, first and second π-type reflective loads, and a first inductor. The cross-coupled line coupler includes two parallel coupled lines and a plurality of first capacitors periodically connected in parallel between the two parallel coupled lines. The first and second π-type reflective loads each include two varactor diodes and a microstrip line connected therebetween. The first and second π-type reflective loads are respectively connected to the coupled port and the through port of the cross-coupled line coupler in a one-to-one correspondence. The coupled port of the cross-coupled line coupler is on the same side as the input port, and the through port is on the same side as the output port. One end of the first inductor is connected to the microstrip line of the first or second π-type reflective load, and the other end is used to input a DC bias voltage to adjust the phase difference between the input port and the output port.

[0008] In general, the above technical solutions conceived by the present invention can achieve the following beneficial effects compared with the prior art:

[0009] 1. The phased array antenna proposed in the present invention uses a independently developed small-size, low-cost, low-power, and low-insertion-loss miniaturized reflective phase shifter to replace the traditional digital phase shifter. The digital phase shifter is designed based on a cross-coupled line coupler. Compared with the branch line coupler, the cross-coupled line coupler has a more compact size and a wider bandwidth; compared with the Lange coupler, the cross-coupled line coupler has a more compact size and lower manufacturing difficulty; it has been verified by actual measurements that the insertion loss of the phase shifter provided by the present invention within a bandwidth of 2.3GHz to 2.8GHz is 1.5±0.6dB. By changing the DC bias voltage applied to the varactor diode, the phase shifter can achieve a continuous phase shift of 360° within the bandwidth, and has the advantages of small size, low insertion loss, and wide bandwidth; therefore, by regulating the phase shift between array units through the DC bias voltage, the phased array antenna proposed in the present invention can achieve ±45° beam scanning on a two-dimensional plane, and can realize functions such as beam directional transmission and continuous beam scanning in wireless communication systems.

[0010] 2. As a further preferred embodiment, the phased array antenna provided by the present invention uses a low-profile, high-gain, and easily machined coupled-fed microstrip antenna as a dual-polarization array element, exhibiting a good front-to-back ratio and high array gain. The dual-polarization array element utilizes mutually perpendicular H-shaped slots to excite dual polarizations, exhibiting good port isolation and polarization isolation. An air dielectric layer is used to increase the array's operating bandwidth. Actual measurements show that the 3dB gain bandwidth of both polarizations of the dual-polarization array element exceeds 26.9% (the 3dB gain bandwidth is the frequency range corresponding to a 3dB drop from the maximum gain value in the antenna's gain curve).

[0011] 3. As a further preferred solution, the power splitter network, phase shifter, and dual-polarization array unit coupling feed port in the phased array antenna provided by the present invention are centrally designed on a single-layer dielectric PCB, which has the advantages of simple design and manufacturing, low cost, high gain, and small space occupation. According to actual measurements, in the 2.3GHz-2.8GHz frequency band, the phased array antenna provided by the embodiment of the present invention has a height of 6mm and a cross-section of only 0.06λ (λ is the air wavelength at the array operating center frequency).

[0012] In summary, the phased array antenna proposed in the present invention has the advantages of low profile, easy processing, low power consumption, high gain, high bandwidth, and dual polarization, and can solve the cost and power consumption problems of wireless communication equipment deployment. BRIEF DESCRIPTION OF THE DRAWINGS

[0013] Figure 1 A circuit schematic diagram of a reflective phase shifter provided in an embodiment of the present invention;

[0014] Figure 2 A schematic diagram of the stacking of a reflective phase shifter provided in an embodiment of the present invention;

[0015] Figure 3 A layout diagram of a reflective phase shifter provided in an embodiment of the present invention;

[0016] Figure 4 An isometric view of a reflective phase shifter provided by an embodiment of the present invention;

[0017] Figure 5 A diagram marking key dimensions of a reflective phase shifter provided in an embodiment of the present invention;

[0018] Figure 6 A physical diagram of a reflective phase shifter provided in an embodiment of the present invention;

[0019] Figure 7 A schematic diagram of the stacked structure of a dual-polarization array unit provided in an embodiment of the present invention;

[0020] Figure 8 A diagram illustrating key dimensions of a dual-polarization array unit provided in an embodiment of the present invention;

[0021] Figure 9 The simulated S-parameters and gain curves of the dual-polarization array unit provided in the embodiment of the present invention;

[0022] Figure 10 The simulated radiation pattern of the dual-polarized array unit provided in an embodiment of the present invention at 2.6 GHz frequency when excited by PortV;

[0023] Figure 11The simulated radiation pattern of the dual-polarized array unit provided in an embodiment of the present invention at 2.6 GHz frequency when excited by PortH;

[0024] Figure 12 A 3D schematic diagram of a dual-polarization low-profile phased array antenna provided by an embodiment of the present invention;

[0025] Figure 13 A bottom view of a dual-polarization low-profile phased array antenna provided in an embodiment of the present invention;

[0026] Figure 14 The simulated and measured S-parameter curves of the phased array antenna provided in the embodiment of the present invention;

[0027] Figure 15 The simulated and measured vertically polarized normalized radiation patterns of the phased array antenna provided in an embodiment of the present invention at 2.6 GHz;

[0028] Figure 16 The simulated and measured horizontally polarized normalized radiation patterns of the phased array antenna provided in an embodiment of the present invention at 2.6 GHz;

[0029] Figure 17 The measured vertical polarization scanning radiation pattern of the phased array antenna provided in an embodiment of the present invention at 2.6 GHz;

[0030] Figure 18 The measured horizontal polarization scanning radiation pattern of the phased array antenna provided in an embodiment of the present invention at 2.6 GHz;

[0031] Figure 19 Simulated and measured gain curves for two polarizations of the phased array antenna provided in an embodiment of the present invention;

[0032] Figure 20 A schematic diagram of a microwave darkroom test scenario provided in an embodiment of the present invention. DETAILED DESCRIPTION

[0033] In order to make the objectives, technical solutions and advantages of the present invention more clearly understood, the present invention is further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely for the purpose of explaining the present invention and are not intended to limit the present invention. In addition, the technical features involved in the various embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.

[0034] To address the cost and power consumption issues of traditional phased arrays, one approach is to research and design low-cost, low-power, and low-insertion-loss phase shifters. Reflective analog phase shifters can meet these requirements. Reflective phase shifters typically consist of a 3dB coupler and two reflective loads. The reflective loads, connected between the coupler's through port and coupled port, typically consist of controlled varactor diodes. By adjusting the varactor diode's bias voltage, analog phase shifting of the RF signal can be achieved. Reflective phase shifters offer advantages such as simple manufacturing processes and low cost, and are expected to address the cost and power consumption issues faced by phased array antennas.

[0035] However, the majority of current reflective phase shifters use branch-line couplers. These couplers, consisting of four microstrip transmission lines connected at 90° angles, are typically large and have a small bandwidth, making them unsuitable for applications requiring a compact design and wide bandwidth. Replacing the branch-line couplers used in traditional reflective phase shifters with cross-coupled-line couplers can significantly reduce size and increase bandwidth.

[0036] Based on this, an embodiment of the present invention provides a dual-polarization low-profile phased array antenna based on a miniaturized reflective phase shifter, comprising: N phase shifter groups, a power division network, N dual-polarization array units, and first and second input ports;

[0037] Each phase shifter group includes two phase shifters; the output ends of the two phase shifters of the nth phase shifter group are respectively connected to the two input ports of the nth dual-polarization array unit in a one-to-one correspondence; N≥2, n=1,2,…,N;

[0038] The power division network includes a first power division network and a second power division network; the first power division network is used to equally divide the radio frequency signal input through the first input port into N signals of equal amplitude and phase, and the second power division network is used to equally divide the radio frequency signal input through the second input port into N signals of equal amplitude and phase; the nth signal equally divided by the first power division network is inputted into one input port of a phase shifter connected to the nth dual-polarization array unit in a one-to-one correspondence; the nth signal equally divided by the second power division network is inputted into one input port of another phase shifter connected to each dual-polarization array unit in a one-to-one correspondence;

[0039] like Figure 1 As shown, the phase shifter includes: a transverse coupled line coupler, a first and a second π-type reflective load and a first inductor L1;

[0040] The cross-coupling line coupler includes two parallel coupling lines and a plurality of first capacitors C1 periodically connected in parallel between the two parallel coupling lines;

[0041] The first and second π-type reflective loads each include two varactor diodes and a microstrip transmission line connected therebetween; the first and second π-type reflective loads are connected to the coupled port and the through port of the cross-coupled line coupler in a one-to-one correspondence; the coupled port and the input port of the cross-coupled line coupler are on the same side, and the through port and the output port are on the same side; the input port and the output port are on one coupled line, and the through port and the output port are on another coupled line;

[0042] One end of the first inductor L1 is connected to the microstrip transmission line of the first or second π-type reflective load, and the other end is used to input a DC bias voltage to adjust the phase difference between the input port and the output port.

[0043] Specifically, the trans-coupling line coupler provided by the embodiment of the present invention can be obtained by periodically loading a plurality of parallel capacitors C1 between two parallel coupling lines.

[0044] A first π-type reflective load and a second π-type reflective load are connected to the coupled port and through port of the coupler, respectively, to form a reflective phase shifter. The first and second π-type reflective loads are each composed of two varactor diodes and a microstrip transmission line connected between the varactor diodes.

[0045] Considering an ideal transmission line and an ideal varactor diode (the varactor diode equivalent series resistance is 0Ω), the input impedance of the load port is Z IN The reflection coefficients of the and ports are:

[0046]

[0047] Where C is the equivalent series capacitance of the four varactors, Z1 and θ1 are the characteristic impedance and electrical length of the microstrip transmission line within the π-type reflective load, respectively (the characteristic impedance and electrical length of the microstrip transmission line of the first and second π-type reflective loads are Z1 and θ1, respectively). Z0 is the characteristic impedance of the coupler port.

[0048] The reflective phase shifter can be equivalent to a two-port network, and its scattering matrix is as follows:

[0049]

[0050] Therefore, the S of the reflective phase shifter 21 The parameter is: jΓ, and its insertion loss is:

[0051] |S 21 |=-20log|Γ|

[0052] Phase shift degree for:

[0053]

[0054] By adjusting the DC bias voltage of the phase shifter, the equivalent capacitance C of the four varactor diodes can be changed simultaneously, thereby changing the input impedance Z of the load port. IN , to achieve the phase difference between the control ports (i.e. ) purpose. The relationship between the DC bias voltage and the equivalent capacitance C of the four varactor diodes varies depending on the type of varactor and can be found in the varactor data sheet. Using a π-type reflective load increases the range of the reflection coefficient, thereby increasing the maximum phase shift range of the phase shifter.

[0055] Preferably, the cross-coupling line coupler further comprises at least one second capacitor C2 connected in series between the midpoints of the two parallel coupling lines, for adjusting the coupling coefficient of the coupling lines.

[0056] Preferably, each varactor diode of the first and second π-type reflective loads is connected in series with a second inductor L2. The series inductor and the capacitance of the varactor diode form a resonance, thereby increasing the maximum phase shift range of the phase shifter.

[0057] In order to minimize the size of the phase shifter and take into account the performance of the coupler and the specific processing and manufacturing, the minimum length of the cross-coupling line coupler is preferably set to 0.25λ g ,λ g is the medium wavelength at the working center frequency of the phase shifter.

[0058] When the operating frequency band is the S band, in order to minimize the size of the phase shifter, preferably, the input port and the output port are both located on one side of the two parallel coupling lines and are both perpendicular to the two parallel coupling lines;

[0059] The transmission lines of the first and second π-type reflective loads are placed on the other side of the two parallel coupling lines after being bent at least once;

[0060] The bent transmission line is located in a space formed by the input transmission line and the output transmission line in the vertical direction; the width of the space is the sum of the width of the input transmission line, the output transmission line and the length of the parallel coupling line.

[0061] Those skilled in the art will appreciate that the characteristic impedance of the input and output transmission lines is typically set to 50Ω, so the widths of the input and output transmission lines are fixed. The bent transmission lines are located within the space formed by the input and output transmission lines in the perpendicular direction. In other words, the length of the phase shifter is equal to the sum of the length of the coupler and the widths of the input and output transmission lines. The length of the coupler is the length of the parallel coupled lines. Therefore, the length of the phase shifter is primarily determined by the length of the coupler across the coupled lines.

[0062] In order to facilitate integration with the power division network of the array antenna, minimize the size of the phase shifter and obtain good performance, preferably, the circuit formed by connecting the cross-coupled line coupler and the first and second π-type reflective loads is an axisymmetric structure.

[0063] When the operating frequency band is the C band, in order to minimize the size of the phase shifter, considering that the C band frequency is relatively high, and to avoid coupling affecting performance due to the RF connectors at the input port and the output port being too close, preferably, the input port and the output port are both arranged parallel to the two parallel coupling lines;

[0064] The microstrip lines of the first and second π-type reflective loads are both perpendicular to the two parallel coupling lines and are both located on the same side of the two parallel coupling lines.

[0065] Accordingly, the length of the phase shifter is equal to the sum of the length of the coupler and the width of the microstrip line of the first and second π-type reflective loads. Therefore, the length of the phase shifter mainly depends on the length of the cross-coupling line coupler.

[0066] In order to further reduce the size of the phase shifter in the direction perpendicular to the parallel coupling lines, preferably, without affecting signal transmission, the microstrip lines of the first and second π-type reflective loads can be bent at least once to reduce the size of the phase shifter in the direction perpendicular to the two parallel coupling lines.

[0067] The first and second power dividing networks both use existing power dividing networks, such as Wilkinson power dividing, T-type power dividing, etc., which can be selected according to actual application requirements.

[0068] With spectrum resources becoming increasingly scarce, dual-polarized antennas are becoming increasingly advantageous in wireless communications. They can increase channel capacity without increasing antenna size and mitigate multipath fading through polarization diversity. Dual-polarized phased array antennas improve transmission efficiency between transmit and receive antennas and are widely used in applications such as mobile satellite communications, weather radar, and synthetic aperture radar. Therefore, the present invention employs dual-polarized antennas as phased array elements.

[0069] It is understandable that any existing dual-polarization array unit can be used. Taking into account factors such as design and manufacturing difficulty, cost, space occupation and gain, preferably, the dual-polarization array unit includes a radiation layer, an air dielectric layer, and a feed layer from top to bottom;

[0070] The air dielectric layer is located between the metal radiation layer and the feed layer;

[0071] The feed layer includes, from top to bottom, a metal ground layer, a first dielectric substrate, and first and second microstrip lines. The metal ground layer is provided with first and second H-shaped coupling slots, and the middle slots of the first and second H-shaped coupling slots are perpendicular to each other.

[0072] The first and second microstrip lines are respectively in one-to-one correspondence and perpendicular to the centers of the middle slots of the first and second H-shaped coupling slots; the signal input ports of the first and second microstrip lines are the two input ports of the dual-polarization unit.

[0073] Preferably, the structural parameters of the first and second H-shaped coupling slots are the same, so that the two polarizations maintain consistent performance parameters such as bandwidth, gain, and radiation pattern, thereby stimulating polarized radiation with the same performance.

[0074] The radiation layer may adopt any conventional structure. For example, the radiation layer includes a second dielectric substrate and a metal patch, and the metal patch is located above the second dielectric substrate.

[0075] The first dielectric substrate can be made of high-frequency, low-loss materials such as PTFE, RO4350B, or RO5880 to improve the gain and radiation efficiency of the array antenna. The second dielectric substrate can be made of common materials such as FR4 to reduce costs.

[0076] The thickness of the air dielectric layer can be set according to an empirical value, which is usually obtained through simulation optimization based on performance parameters such as the bandwidth and front-to-back ratio of the antenna unit.

[0077] In order to make the core design of the dual-polarization array unit more compact, preferably, the ends of the first and second microstrip lines are both fan-shaped.

[0078] Preferably, the N phase shifter groups, the power division network, the N dual-polarization array units and the first and second input ports are integrated on a single-layer dielectric PCB, which has the advantages of compact structure, low manufacturing difficulty, and low profile.

[0079] The following is a specific example of the phased array antenna provided by the present invention. The following example is a dual-polarized low-profile phased array antenna based on a miniaturized reflective phase shifter designed with preferred parameters and working in the 2.3GHz-2.8GHz frequency band. By simply adjusting the parameters (for example, scaling or amplifying all dimensions of the antenna proportionally and using electromagnetic simulation software for appropriate optimization to obtain better performance), the solution proposed in the present invention can realize a dual-polarized low-profile phased array antenna working in the Sub-6G high frequency band, 5G, 6G, WLAN and other frequency bands.

[0080] like Figure 1-3 As shown, four parallel capacitors C1 are periodically loaded between two parallel coupling lines of the cross-coupling line coupler, and two second capacitors C2 are also connected in series between the midpoints of the two parallel coupling lines.

[0081] The phase shifter operates in the 2.3GHz-2.8GHz frequency band and is integrated on a single-layer dielectric PCB. By adjusting the DC bias voltage, the phase difference between the two ports can be controlled. Figure 2 To reduce insertion loss, the dielectric substrate used can be common PTFE, which has a dielectric constant of 2.94, a loss tangent of 0.0016, and a thickness of 0.76 mm.

[0082] Figure 1 Where Z0' is the characteristic impedance of the transmission line between the parallel capacitors; 2*θ' is the electrical length of the transmission line between the parallel capacitors, and Z2 and θ2 are the characteristic impedance and electrical length of the transmission line loaded between the parallel line couplers, respectively.

[0083] like Figure 3 As shown in the dashed area, four capacitors C1 are connected in parallel between two parallel coupled lines to form a transverse coupled line coupler. Furthermore, two capacitors C2 and a transmission line are loaded between the parallel coupled lines to adjust the coupling coefficient of the coupled lines. A π-type reflective load is connected to the Coupled and Through ports of the coupler to form a reflective phase shifter. Each π-type reflective load consists of two varactor diodes and a microstrip transmission line connected between the varactors. In this example, the transmission line is bent three times to reduce the space occupied by the phase shifter. A second inductor L2 is connected in series with each varactor diode to increase the maximum phase shift range.

[0084] A DC bias voltage is applied to the reflective phase shifter via a first inductor L1, where a DC bias voltage is input to one end of L1 and the other end is connected to a π-type reflective load transmission line. By adjusting the DC bias voltage, the capacitance of all varactor diodes is dynamically adjusted, thereby changing the reflection coefficients of the Coupled port and the Through port to achieve the purpose of regulating the phase difference between Port1 and Port2 (i.e., the input port and the output port). Due to the layout characteristics of the cross-coupling line coupler, only one DC bias voltage input is required to dynamically regulate the capacitance of the four varactor diodes. In addition, the input bias voltage can be isolated from the RF signal through the periodically loaded capacitors in the cross-coupling line coupler, eliminating the need to add DC blocking capacitors at the two ports of the phase shifter, thereby simplifying the circuit design of the bias voltage input.

[0085] The key dimensions of the phase shifter are as follows: Figure 5 As shown, the overall design layout is axially symmetrical. Figure 5 In the equation ( 4 ), L3 is the coupler length, L4 is the length of the transmission line between the parallel capacitors; L3 = 4*L4, L5 is the length of the transmission line loaded between the parallel line couplers, L6 + L7 + 2L8 is the length of the loaded transmission line, W1 is the width of the parallel coupled lines, W2 is the distance between the parallel coupled lines, W3 is the width of the transmission line loaded between the parallel line couplers, W4 is the width of the loaded transmission line, Slot is the slot opened in the reference ground, Ws and Ls are the width and length of the slot, and W50 is the width of the input and output transmission lines (for performance considerations, W4 is usually set to less than W50). The length of the phase shifter is equal to the length of the coupler plus the width of the microstrip transmission lines of the first and second π-type reflective loads, that is, L3 + 2*W50.

[0086] The above dimensions are calculated based on transmission line theory and adjusted and optimized using full-wave electromagnetic simulation software. First, the basic dimensions W1, W2, L3, L4 of the cross-coupled line coupler and the approximate range of the shunt capacitance are determined based on the operating band and transmission line theory. Then, full-wave electromagnetic simulation software is used to simulate and optimize these parameters. Finally, full-wave electromagnetic simulation software is used to adjust and optimize the characteristic impedance Z1, electrical length θ1, and microstrip transmission line within the π-type reflective load (corresponding to W4, L6, L7, and L8) to minimize the insertion loss and size of the phase shifter while ensuring that the maximum phase shift is greater than 360°.

[0087] The actual picture of the phase shifter is as follows Figure 6 According to actual measurements, the core design of the reflective phase shifter occupies a size of 19.3mm*8.5mm. The air wavelength λ at 2.6GHz is 115.4mm. Therefore, the size expressed in λ is: (19.3 / 115.4)*(8.5 / 115.4)λ2 , that is 0.17*0.07λ 2 , it can be seen that the reflective phase shifter occupies a total of 0.17×0.07λ 2 PCB size.

[0088] The schematic diagram of the stacked structure of the dual-polarization array unit is as follows: Figure 7 As shown, from top to bottom, the stacked layers are the radiation layer, air dielectric layer, and feed layer. The radiation layer's dielectric is made of FR4, with a thickness of 1.49 mm, a relative dielectric constant of 4.6, and a loss tangent of 0.017. The feed layer's dielectric is made of PTFE, with a thickness of 0.76 mm, a relative dielectric constant of 2.94, and a loss tangent of 0.0016. The air dielectric between the radiation layer and the feed layer is 5 mm thick. A square patch is located above the second dielectric substrate, forming the radiation layer; the microstrip feed line is located below the first dielectric substrate; and the coupling slot is designed to be H-shaped to reduce the array's back radiation. The H-shaped coupling slots are placed perpendicular to each other to excite both vertical and horizontal polarizations. To make the core design of the array unit more compact, the end of the microstrip feed line is designed to be fan-shaped. By adjusting the size of the coupling slot and the fan shape, the array unit port can achieve good impedance matching.

[0089] After measurement, the cross-section of the array unit is 7.2mm. The wavelength of air at the 2.6GHz frequency point is 115.4mm. Therefore, the cross-section of the array unit is: 7.2mm / 115.4mm*λ, which is 0.062λ.

[0090] The key dimensions of the dual-polarization array element are as follows: Figure 8 As shown, where L1 is the side length of the array unit, L2 is the side length of the radiating metal patch, and W s1 is the height of the middle slot of the H-type coupling slot, Ws2 is the width of the two end slots of the H-type coupling slot, L S1 is the width of the H-type coupling slot, L S2 is the height of the H-type coupling slot, W 50 The width of the feed line is usually set to make its impedance 50Ω.

[0091] The RF signal is fed into the two ports PortH and PortV, and coupled to the radiation layer through the mutually perpendicular H-shaped coupling slots, thereby stimulating vertical and horizontal polarized radiation respectively. Figure 9 The following are the simulated S-parameters and gain curves for the dual-polarization array element. The simulation results show that the -10dB coincident impedance bandwidth of the array element's two ports is 7.7% (2.5GHz-2.7GHz), and the isolation between Port H and Port V is greater than 30dB within the operating frequency band. Furthermore, the simulated gain of the array element is 7.9dBi for vertical polarization and 7.8dBi for vertical polarization.

[0092] The normalized simulated radiation patterns of the array elements when PortV and PortH are excited are as follows: Figure 10 、 11 The simulated cross-polarization levels of the two polarizations are both lower than -30dB, indicating that the array elements have good polarization isolation.

[0093] Based on the above dual-polarization antenna unit and miniaturized reflective phase shifter, the 3D schematic diagram and bottom view of the dual-polarization low-profile phased array antenna provided by the embodiment of the present invention are respectively as follows: Figure 12 、 Figure 13 As shown. Each array element is equipped with two miniaturized reflective phase shifters to provide continuous phase shift from 0° to 360° for the two polarizations. Considering the isolation between the output ports and the compactness of the overall structure, the 1-to-16 power splitter network used at each port is composed of a 1-to-4 Wilkinson power splitter and four T-type power splitters. Figure 13 As shown, two one-to-sixteen power splitting networks are placed alternately, and PortH and PortV excite the horizontal polarization and vertical polarization of the array respectively.

[0094] The simulated and measured S parameters of the two ports of the phased array antenna are as follows: Figure 14 The simulation results show that the coincidence impedance bandwidth of the two polarization ports is 16.9% (2.40GHz-2.84GHz), and the measured results show that the coincidence impedance bandwidth of the two polarization ports is 21.1% (2.32GHz-2.87GHz). Within the working bandwidth, the isolation between the two ports exceeds 32dB. The simulated and measured normalized radiation patterns of the array antenna at 2.6GHz for two polarizations are shown in Figure 2. Figure 15 and Figure 16 The simulated and measured cross-polarization levels are both lower than -28dB in the main lobe direction. The measured scanning radiation patterns of the array antenna at 2.6GHz for the two polarizations are shown in Figure 2. Figure 17 and Figure 18 The phased array antenna can achieve ±45° continuous beam scanning in the E and H planes. The simulated and measured gains of the two polarizations of the phased array antenna are shown in Figure 2. Figure 19 As shown in the figure, it can be seen that the 3dB gain bandwidth of the two polarizations of the array exceeds 26.9%. Figure 20 shown.

[0095] It will be easily understood by those skilled in the art that the above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included in the scope of protection of the present invention.

Claims

1. A dual-polarization low-profile phased array antenna based on a miniaturized reflective phase shifter, characterized in that: include: N phase shifter groups, a power division network, N dual-polarization array elements, and first and second input ports; Each phase shifter group includes two phase shifters; the output ends of the two phase shifters of the nth phase shifter group are respectively connected to the two input ports of the nth dual-polarization array unit in a one-to-one correspondence; N≥2, n=1,2,…,N; The power division network includes a first power division network and a second power division network; the first and second power division networks are respectively used to equally divide the radio frequency signal input through the first and second input ports into N signals of equal amplitude and phase; the nth signal equally divided by the first power division network is inputted one-to-one to the input port of one of the phase shifters connected to the nth dual-polarization array unit; the nth signal equally divided by the second power division network is inputted one-to-one to the input port of another phase shifter connected to each dual-polarization array unit; The phase shifter includes: a cross-coupled line coupler, first and second π-type reflective loads, and a first inductor. The cross-coupled line coupler includes two parallel coupled lines and a plurality of first capacitors periodically connected in parallel between the two parallel coupled lines. The first and second π-type reflective loads each include two varactor diodes and a microstrip line connected therebetween. The first and second π-type reflective loads are respectively connected to the coupled port and the through port of the cross-coupled line coupler in a one-to-one correspondence. The coupled port of the cross-coupled line coupler is on the same side as the input port, and the through port is on the same side as the output port. One end of the first inductor is connected to the microstrip line of the first or second π-type reflective load, and the other end is used to input a DC bias voltage to adjust the phase difference between the input port and the output port.

2. The phased array antenna according to claim 1, wherein: The input port and the output port are both arranged in parallel with the two parallel coupling lines; The microstrip lines of the first and second π-type reflective loads are both perpendicular to the two parallel coupling lines and are both located on the same side of the two parallel coupling lines.

3. The phased array antenna according to claim 2, wherein: The microstrip lines of the first and second π-type reflective loads are bent at least once to reduce the size of the phase shifter in a direction perpendicular to the two parallel coupling lines.

4. The phased array antenna according to claim 1, wherein: The input port and the output port are both located on one side of the two parallel coupling lines and are both perpendicular to the two parallel coupling lines; The transmission lines of the first and second π-type reflective loads are placed on the other side of the two parallel coupling lines after being bent at least once; The bent transmission line is located in a space formed by the input transmission line and the output transmission line in the vertical direction; the width of the space is the sum of the width of the input transmission line, the output transmission line and the length of the parallel coupling line.

5. The phased array antenna according to any one of claims 1 to 4, wherein: The length of the cross-coupled line coupler is 0.25λ g ,λ g is the medium wavelength at the working center frequency of the phase shifter.

6. The phased array antenna according to claim 1, wherein: The cross-coupling line coupler further includes at least one second capacitor connected in series between the midpoints of the two parallel coupling lines; and each varactor diode of the first and second π-type reflective loads is connected in series with a second inductor.

7. The phased array antenna according to claim 1, wherein: The circuit formed by connecting the cross-coupled line coupler and the first and second π-type reflective loads is an axisymmetric structure.

8. The phased array antenna according to any one of claims 1 to 7, wherein: The dual-polarization array unit includes, from top to bottom, a radiation layer, an air dielectric layer, and a feed layer; The air dielectric layer is located between the radiation layer and the feed layer; The feed layer includes, from top to bottom, a metal ground layer, a first dielectric substrate, and first and second microstrip lines. The metal ground layer is provided with first and second H-shaped coupling slots, and the first and second H-shaped coupling slots are perpendicular to each other. The first and second microstrip lines are respectively perpendicular to the centers of the first and second H-shaped coupling slots in a one-to-one correspondence.

9. The phased array antenna according to claim 8, wherein: The structural parameters of the first and second H-shaped coupling slots are the same; The ends of the first and second microstrip lines are both fan-shaped.

10. The phased array antenna according to claim 8, wherein: The N phase shifter groups, the power division network, the N dual-polarization array units and the first and second input ports are integrated on a single-layer dielectric PCB.

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