A self-adjusting zero fluxgate current sensor

Abstract

This invention discloses a self-zeroing fluxgate current sensor, belonging to the field of current sensor technology, aiming to solve the problem of zero-point bias error caused by the asymmetry of amplitude or phase of the two excitation signals in existing fluxgate current sensors. It includes an excitation signal generation circuit, two excitation cores and coils, an adder, an adjustable bandpass filter, an amplitude adjustment branch, a phase adjustment branch, a zero-adjustment controller, and a signal conditioner. The amplitude adjustment branch extracts the amplitude matching error of the two excitation induction signals, the phase adjustment branch extracts the phase deviation of the two excitation induction signals, the adjustment controller calculates the gain adjustment value and the phase adjustment value, and the signal conditioner uses the gain adjustment value and the phase adjustment value to adjust the gain and phase of one of the excitation drive signals in a closed loop, making the two excitation drive signals equal in amplitude and opposite in phase, directly eliminating the amplitude error and phase error from the excitation source, realizing mutual cancellation of the fundamental wave and odd harmonics, and fundamentally suppressing zero-point bias.

Description

Technical Field

[0001] This invention belongs to the field of current sensor technology, specifically, it relates to a self-zeroing fluxgate current sensor based on fluxgate technology. Background Technology

[0002] Fluxgate current sensors require two excitation cores, each with an excitation coil wound on it. Two opposing excitation signals are used to excite the in-phase coil, or in-phase excitation signals are used to excite the oppositely wound excitation coil. The purpose is to create magnetic fields in opposite directions in the two excitation coils, causing the cores to enter a periodic bidirectional oversaturation operating state. In this way, the even-order harmonic components of the induced signal can reflect the magnetic field generated by the measured current, while the fundamental and odd-order harmonics of the excitation can cancel each other out.

[0003] However, in reality, due to the differences in the performance of the two excitation circuits and the magnetic core, as well as the differences in the winding process, it is impossible to completely cancel out the fundamental and odd harmonics. This will result in the presence of fundamental and odd harmonic components in the induced signal, which manifests as zero-point offset error.

[0004] Traditional methods for reducing the impact of zero-point offset include the following:

[0005] (1) Adjust the excitation signal manually based on the final output bias of the sensor. However, this method can only adjust the signal at the very end, changing the zero-point bias of the entire system. It cannot change the root cause zero-point bias caused by the excitation circuit and magnetic core, cannot guarantee the symmetry of bidirectional excitation, cannot determine the amplitude and phase of the excitation, and therefore cannot guarantee the zero flux state, affecting the zero-point bias and accuracy of the sensor. Furthermore, it is easily affected by environmental interference, requires manual operation, is time-consuming, and increases production and maintenance costs.

[0006] (2) Use a microcontroller to acquire the dual-channel excitation output signal, or acquire the single-channel signal after the two channels are added together. Determine whether the excitation is symmetrical by measuring the amplitude, and then reduce the output bias by adjusting the excitation circuit. If the dual-channel output signal is acquired, theoretically the amplitude and phase difference of the two signals can be extracted. However, due to factors such as the operating speed of the microcontroller, there will be a large error. If the single-channel signal after the two channels are added together is acquired, it can only be adjusted according to the amplitude, and the phase difference of the excitation cannot be determined. Summary of the Invention

[0007] The purpose of this invention is to design a self-adjusting zero fluxgate current sensor that can monitor the amplitude and phase differences of dual excitation signals in real time, and adjust the amplitude and phase of the excitation drive signal accordingly, fundamentally solving the zero-point deviation problem caused by excitation asymmetry due to long-term operation. Furthermore, by adding a temperature sensor to record the adjustment values ​​at different temperatures, high-precision measurement over a wide temperature range is ensured.

[0008] The present invention is implemented using the following technical solutions: A self-adjusting zero fluxgate current sensor is proposed, comprising: The excitation signal generation circuit is used to generate two excitation drive signals; Two excitation cores are used, each with an excitation coil wound on it. The two excitation coils are excited by two excitation drive signals respectively. An adder is used to receive the excitation induction signals of two excitation coils and add them together for output. An adjustable bandpass filter receives the output of an adder and extracts the excitation frequency from the output of the adder in self-zeroing mode. The amplitude adjustment branch receives the excitation frequency signal and the output of the adjustable bandpass filter, and extracts the first DC component; the first DC component linearly reflects the amplitude matching error of the two excitation induction signals. The phase adjustment branch receives the quadrature excitation frequency signal and the output of the adjustable bandpass filter, and extracts the second DC component; the second DC component linearly reflects the phase deviation of the two excitation induction signals. The zero-adjustment controller receives a first DC component and a second DC component, generates a gain adjustment value based on the first DC component, and generates a phase adjustment value based on the second DC component. The signal conditioner receives gain control signal and phase control signal in self-zeroing mode, and adjusts the gain and phase of one excitation drive signal based on the gain adjustment value and phase adjustment value, so that the two excitation drive signals are equal in amplitude and opposite in phase.

[0009] Compared with the prior art, the advantages and positive effects of the present invention are as follows: The self-zeroing fluxgate current sensor proposed in this invention includes an excitation signal generation circuit, two excitation magnetic cores and coils, an adder, an adjustable bandpass filter, an amplitude adjustment branch, a phase adjustment branch, a zero-adjustment controller, and a signal conditioner. The amplitude adjustment branch extracts the amplitude matching error of the two excitation induction signals, the phase adjustment branch extracts the phase deviation of the two excitation induction signals, the adjustment controller calculates the gain adjustment value and the phase adjustment value, and the signal conditioner uses the gain adjustment value and the phase adjustment value to adjust the gain and phase of one of the excitation drive signals in a closed loop, so that the two excitation drive signals are equal in amplitude and opposite in phase, directly eliminating the amplitude error and phase error from the excitation source, realizing the mutual cancellation of the fundamental wave and odd harmonics, and fundamentally suppressing zero-point bias.

[0010] In some embodiments of the present invention, the amplitude adjustment branch includes a first multiplier and a first low-pass filter; the adjustment controller includes a first PI controller; The first multiplier is connected to the excitation frequency signal and the output of the adjustable bandpass filter. The output of the first multiplier is connected to the input of the first low-pass filter. The output of the first low-pass filter is connected to the first PI controller.

[0011] In some embodiments of the present invention, the phase adjustment branch includes a second multiplier and a second low-pass filter; the adjustment controller includes a second PI controller; The second multiplier is connected to the excitation frequency quadrature signal and the output of the adjustable bandpass filter. The output of the second multiplier is connected to the input of the second low-pass filter. The output of the second low-pass filter is connected to the second PI controller.

[0012] In the above embodiments, the amplitude adjustment branch demodulates the excitation frequency signal in phase, extracts the DC component I reflecting the amplitude error of the two excitation drive signals, and the first PI controller uses this DC component I as feedback and a target value of 0 to output the gain parameter of the direct correction signal conditioner to ensure that the amplitude error eventually converges to zero. The phase adjustment branch demodulates the excitation frequency signal in quadrature, extracts the DC component Q reflecting the phase error of the two excitation drive signals, and the second PI controller uses this DC component Q as feedback and a target value of 0 to output the phase parameter of the direct correction signal conditioner to ensure that the phase error eventually converges to zero. The two loops do not interfere with each other.

[0013] In some embodiments of the present invention, the current sensor further includes a gating switch network; the gating switch network includes: The first selection switch is connected between the excitation signal generation circuit and the amplitude adjustment branch; The second selector switch is connected between the amplitude adjustment branch and the zero adjustment controller; In normal operating mode, the first gating switch is configured to connect a frequency twice the excitation frequency to the amplitude adjustment branch; the second gating switch is configured to connect the amplitude adjustment branch to the feedback branch; and the cutoff frequency of the first low-pass filter is set to half the excitation frequency, and the center frequency of the adjustable bandpass filter is set to twice the excitation frequency. In self-zeroing mode, the first gating switch is configured to connect the excitation frequency signal to the amplitude adjustment branch; the second gating switch is configured to connect the amplitude adjustment branch to the zeroing controller; and the adjustable bandpass filter is set as a low-pass filter with a cutoff frequency exceeding the excitation frequency.

[0014] In this embodiment, a gating switch network switches between normal operating mode and self-zeroing mode, with different filter parameters and signal paths configured in different modes. The amplitude adjustment branch is reused in both normal operating mode and self-zeroing mode, reducing circuit complexity and cost. Based on this design, the self-zeroing mode can be triggered at any time, adapting to different operating conditions and calibration requirements after long-term operation.

[0015] In some embodiments of the present invention, the current sensor switches to verification mode after completing self-zeroing; In verification mode, the first gating switch is configured to connect a second harmonic signal of the excitation frequency to the amplitude adjustment branch; the second gating switch is configured to connect the amplitude adjustment branch to the feedback branch; the center frequency of the adjustable bandpass filter is set to a second harmonic of the excitation frequency; and the cutoff frequency of the first low-pass filter is set to half the excitation frequency. In verification mode, monitor the sensor output when there is no current to be measured. If the output is not zero, it verifies that there is a fault in the sensor hardware circuit.

[0016] In this embodiment, after the self-zeroing mode is completed, the verification mode is used to verify whether the self-zeroing is truly effective, thus avoiding misjudgment caused by hardware failure.

[0017] In some embodiments of the present invention, when the current sensor outputs a voltage, it further includes: A signal compensator, connected to the output of the sampling resistor, is used to adjust the gain of the output voltage signal according to different input currents to be measured, so as to adjust the input-output ratio to a set ratio; and to increase the system gain when operating at high frequency, so as to increase the system bandwidth of the sensor; wherein, the sampling resistor is used to convert the output current of the feedback coil into voltage.

[0018] In this embodiment, when the sensor is a voltage output type, a signal compensator is configured for the sensor. The gain of the signal compensator is adjusted according to the output voltage under different input currents to achieve an ideal input-output ratio. Furthermore, when the system gain decays with frequency, the signal compensator is used to increase the gain in the attenuation frequency band to improve the system bandwidth, thereby effectively increasing the system's operating bandwidth.

[0019] In some embodiments of the present invention, the current sensor is further configured with a temperature sensor and connected to a zero-adjustment controller; The zero-adjustment controller is configured to perform self-zeroing at different temperatures and store the corresponding gain adjustment value and phase adjustment value in the memory; in normal operating mode, it automatically calls the corresponding adjustment value according to the temperature read by the current temperature sensor and sends it to the signal conditioner.

[0020] In this embodiment, in order to ensure the performance of the sensor over a wide temperature range, the adjustment relationship between the gain adjustment value and the phase adjustment value at different temperatures is established. In normal working mode, if the ambient temperature changes, the corresponding adjustment data can be retrieved according to the current temperature to automatically adjust the signal and improve the accuracy of the sensor.

[0021] In some embodiments of the present invention, the current sensor further includes: An anomaly detection circuit, connected to the output of the sensor, is used to monitor whether the sensor can achieve zero magnetic flux. A feedback drive circuit is connected between the anomaly detection circuit and the feedback coil. When the anomaly detection circuit detects that the sensor cannot achieve zero magnetic flux, it injects large positive and negative currents into the feedback coil to force the sensor into a zero magnetic flux state. The feedback coil is wound on the main magnetic core.

[0022] In this embodiment, an anomaly detection circuit is configured to monitor whether the sensor has zero magnetic flux. When the input exceeds the limit or external interference causes the sensor to fail with zero magnetic flux, the feedback drive circuit injects large positive and negative currents into the feedback coil to force the sensor to enter the zero magnetic flux state, thereby preventing the sensor from working in an abnormal state for a long time and improving the robustness of the system.

[0023] Other features and advantages of the present invention will become clearer after reading the detailed description of the embodiments of the present invention in conjunction with the accompanying drawings. Attached Figure Description

[0024] The accompanying drawings, as part of this invention, are provided to further illustrate the invention. The illustrative embodiments and descriptions are used to explain the invention but do not constitute an undue limitation thereof. Clearly, the drawings described below are merely some embodiments; those skilled in the art can obtain other drawings based on these drawings without creative effort.

[0025] Figure 1 The circuit structure diagram of the self-zeroing fluxgate current sensor proposed in this invention is shown below; Figure 2 This is a schematic diagram of the circuit structure of the self-zeroing fluxgate current sensor of the present invention operating in the self-zeroing state; Figure 3 The circuit structure of the self-zeroing fluxgate current sensor of the present invention is used to switch to the verification mode after self-zeroing. Figure 4 This is the circuit structure of the self-adjusting zero fluxgate current sensor under normal operating mode according to the present invention; Figure 5 This is the excitation drive circuit under ideal conditions; Figure 6 This demonstrates the bidirectional excitation drive signal in an ideal excitation drive circuit. Figure 7 This is a demonstration of the output signal of the excitation coil in an ideal excitation drive circuit. Figure 8 This is a waveform display of the excitation induced current after being added by an adder under ideal conditions; Figure 9 for Figure 8 The frequency representation of the output waveform is shown below; Figure 10 Demonstration of the difference circuit for the excitation coil; Figure 11 for Figure 10 Display of excitation coil current under the specified conditions; Figure 12 for Figure 11 The spectral components of the output signal under the given state; Figure 13 Demonstration of a circuit with resistance differences; Figure 14 for Figure 13 The spectral components of the output signal under the given state; Figure 15 In response to Figure 10 The circuit after adjusting the signal conditioner; Figure 16 for Figure 15 The corresponding excitation coil current is displayed; Figure 17 for Figure 15 The spectral components of the output signal; Figure 18 For signal conditioning circuits with series capacitor C; Figure 19 for Figure 18 The corresponding excitation coil current is displayed; Figure 20 for Figure 18 The spectral components of the output signal.

[0026] It should be noted that these accompanying drawings and textual descriptions are not intended to limit the scope of the invention in any way, but rather to illustrate the concept of the invention to those skilled in the art by referring to specific embodiments. Detailed Implementation

[0027] To make the objectives, technical solutions, and advantages of the embodiments of the present invention clearer, the technical solutions in the embodiments will be clearly and completely described below with reference to the accompanying drawings. The following embodiments are used to illustrate the present invention, but are not intended to limit the scope of the present invention.

[0028] This invention aims to design a self-zeroing circuit for a fluxgate current sensor, enabling it to operate in both a normal working mode and a self-zeroing mode. When switching to self-zeroing mode, it monitors the amplitude and phase differences between the dual excitation drive signals and adjusts the amplitude and phase of one of the excitation drive signals accordingly. This fundamentally solves the zero-point deviation problem introduced by excitation asymmetry. Specifically, as shown... Figure 1 As shown, the self-zeroing fluxgate current sensor includes:

[0029] The excitation signal generation circuit, consisting of a crystal oscillator, a frequency divider, and an integrator, is used to generate two excitation drive signals. The frequency divider and integrator can be replaced by a digital-to-analog converter (ADC) or a direct digital synthesizer (DDS) to generate a sine wave or other complex waveform as the source of the excitation signal.

[0030] Two excitation cores, each with an excitation coil wound on it, and the two excitation coils are excited by two excitation drive signals respectively.

[0031] The main magnetic core has a feedback coil wound on it. The excitation core and the main magnetic core can be arranged side by side, or the excitation core can be placed inside the main magnetic core.

[0032] The adder is used to receive the excitation induction signals from two excitation coils, add the two excitation induction signals together, and then output the result.

[0033] An adjustable bandpass filter receives the output of an adder, extracts the excitation frequency from the adder's output in self-zeroing mode, and extracts a second harmonic signal of the excitation frequency from the adder's output in normal operating mode.

[0034] In self-zeroing mode, the amplitude adjustment branch receives the excitation frequency signal and the output of the adjustable bandpass filter, extracting the first DC component. This first DC component linearly reflects the amplitude matching error between the two excitation induction signals. In normal operating mode, it receives the second harmonic signal of the excitation frequency and the output of the adjustable bandpass filter, extracting the third DC component and outputting it to the hardware PID controller. This third DC component reflects the difference between the current actual value and the standard value. The hardware PID controller calculates the correction control quantity using proportional, integral, and derivative operations. The feedback drive circuit then sends the adjustment quantity calculated by the hardware PID controller back to the front-end circuit to correct the excitation induction signal, forming a closed-loop feedback.

[0035] The phase adjustment branch receives the quadrature excitation frequency signal and the output of the adjustable bandpass filter, and extracts the second DC component; this second DC component linearly reflects the phase deviation of the two excitation induction signals.

[0036] A zero-adjustment controller receives a first DC component and a second DC component, generates a gain adjustment value based on the first DC component, and generates a phase adjustment value based on the second DC component. In this embodiment of the invention, the zero-adjustment controller includes a first PI controller and a second PI controller. The first PI controller generates the gain adjustment value based on the first DC component, and the second PI controller generates the phase adjustment value based on the second DC component.

[0037] The signal conditioner receives gain control signal and phase control signal in self-zeroing mode, and adjusts the gain and phase of one excitation drive signal based on the gain adjustment value and phase adjustment value, so that the two excitation drive signals are equal in amplitude and opposite in phase.

[0038] like Figure 1 As shown, the current sensor provides current output or voltage output by switching the third selector switch; for voltage output, a sampling resistor is used to convert the excitation induced current output by the coil into voltage.

[0039] In this embodiment of the invention, the amplitude adjustment branch consists of a first multiplier and a first low-pass filter, which together form a phase-sensitive detector. The first multiplier is connected to the excitation frequency signal and the output of an adjustable bandpass filter, and the output of the first multiplier is connected to the input of the first low-pass filter; the output of the first low-pass filter is connected to the first PI controller of the zero-adjustment controller. Alternatively, a dedicated phase-sensitive detector chip can be used to implement this amplitude adjustment branch. The first low-pass filter is an adjustable low-pass filter, allowing it to be applied to both self-zeroing mode and normal operating mode.

[0040] The phase adjustment branch consists of a second multiplier and a second low-pass filter, which together form a phase-sensitive detector. The second multiplier is connected to the quadrature signal of the excitation frequency and the output of an adjustable bandpass filter. The output of the second multiplier is connected to the input of the second low-pass filter; the output of the second low-pass filter is connected to a second PI controller. This second low-pass filter is a fixed low-pass filter with a cutoff frequency set to an extremely low frequency.

[0041] In this invention, the current sensor operates in two states: normal operation and self-zeroing. The switching of the gating switch network and the configuration of components in each mode are controlled by the debugging system and the regulating controller. The gating switch network includes a first gating switch and a second gating switch. For example... Figure 1 As shown, the first selector switch is connected between the excitation signal generation circuit and the amplitude adjustment branch, and the second selector switch is connected between the amplitude adjustment branch and the zero adjustment controller.

[0042] The circuit combinations and operating logic for the two operating states are described below.

[0043] 1. Self-zeroing mode.

[0044] When hardware parameter debugging is required after the current sensor is assembled or during long-term use, the debugging system controls the regulating controller to switch it into self-zeroing state, combined with... Figure 2 As shown, the circuit state is as follows:

[0045] (1) The adjustable bandpass filter is adjusted to a low-pass filter with a cutoff frequency slightly higher than the excitation frequency f.

[0046] (2) The first selection switch is connected to the excitation frequency signal from the excitation signal generation circuit.

[0047] (3) The cutoff frequency of the first low-pass filter is adjusted to an extremely low frequency, which is consistent with the cutoff frequency of the second low-pass filter.

[0048] (4) The second selector switch is connected to the regulating controller.

[0049] (5) The phase shifter performs a 90-degree fixed phase shift on the excitation frequency signal.

[0050] In this state, the signal entering the first and second multipliers contains excitation imbalance information, and its frequency is consistent with the excitation frequency. Therefore, the amplitude and phase information of this signal can be obtained by using the phase-sensitive demodulation method (IQ demodulation method). That is, the signal is multiplied by two reference signals that are the same as and orthogonal to the excitation signal, and the DC component is obtained by using a very low-frequency low-pass filter. Then, the amplitude and phase are calculated in the regulating controller to obtain the amplitude and phase information of the excitation imbalance signal.

[0051] The regulating controller adjusts the signal conditioner based on amplitude and phase information, that is, it adjusts the amplitude and phase of one of the excitation signals to minimize the amplitude of the excitation imbalance signal, thereby reducing the zero-point deviation caused by imbalance factors such as excitation coils and circuits.

[0052] In this embodiment, two independent PI controls are used: the first PI controller controls the output amplitude, and the second PI controller controls the output phase, in order to reduce the amplitude and phase of the unbalanced signal.

[0053] The core of this process lies in the two DC components after IQ demodulation: the first DC component and the second DC component, which linearly correspond to the amplitude difference and phase difference of the two excitation channels, respectively. The first DC component (I component) reflects the amplitude matching error, and the second DC component (Q component) reflects the phase deviation relative to the two excitation signals. Therefore, the system can simultaneously operate two completely independent PI controllers: the amplitude loop uses I as feedback and a target value of 0, and its output directly corrects the gain parameter of the signal conditioner; the phase loop uses Q as feedback and a target value of 0, and its output directly corrects the phase parameter of the signal conditioner. If I > 0, it indicates that the amplitude of the modulated channel is too small, and the first PI controller will automatically increase the gain, and vice versa. If Q > 0, it indicates that the phase lags behind the ideal inversion, and the second PI controller will correspondingly increase the phase shift, and vice versa. The two loops complement each other and continuously adjust until I and Q converge to within the preset dead zone. At this time, the unbalanced error signal approaches zero, the two excitations recover to equal amplitude and inversion, and the zero-point deviation introduced by the drive asymmetry is effectively suppressed.

[0054] The mathematical explanation of the above process is as follows:

[0055] Assume reference excitation signal Directly drive coil L1 (forward wound), and simultaneously feed it into an amplitude-phase adjustable regulator; the regulator output... Drive coil L2 (reverse wound). The output signals of the two coils are added together to obtain the unbalanced error signal: ; In the formula, This represents the circuit gain. In ideal balance, , The two magnetic fields completely cancel each other out. .

[0056] remember For the phase deviation of the regulator output relative to the reference (target is 0), when When the angle is very small, using a small angle approximation, we have: ; With Same frequency and phase and orthogonal For reference, After multiplication demodulation and low-pass filtering at an extremely low cutoff frequency, two DC quantities are obtained: , ; in, The linear component reflects the amplitude matching error between the two coils. The linear component reflects the degree to which the phase deviates from the ideal in-phase state, and the two are completely decoupled. Therefore, the system operates two independent PI controllers in parallel:

[0057] Amplitude adjustment ring: As a feedback quantity, with a target of 0, the output directly corrects the regulator gain. ,like (express If the L2 amplitude is too small, it will automatically increase. ; Then decrease .

[0058] Phase adjustment loop: The feedback quantity is set to 0, and the output corrects the regulator phase. ,like (express (i.e., the L2 current phase lags behind the reference), then increasing Make it approach zero; Then decrease .

[0059] The two ring roads will continue to be adjusted until... and All converge to the preset minimum dead zone, at which point... , , The fundamental component approaches zero, the two excitation currents are strictly equal in amplitude and phase, and the magnetic field is completely canceled by the reverse winding of the coil. The zero-point deviation caused by the asymmetry of the drive and the difference of the coil is effectively suppressed.

[0060] In this embodiment of the invention, after zeroing, the system switches to verification mode to verify the adjustment result of the zero-point offset, such as... Figure 3 As shown, the circuit state is as follows:

[0061] (1) The center frequency of the adjustable bandpass filter is set to twice the excitation frequency, and the channel does not include the excitation frequency.

[0062] (2) The first selection switch is connected to the second harmonic signal of the excitation frequency.

[0063] (3) The cutoff frequency of the first low-pass filter is adjusted to half of the excitation frequency.

[0064] (4) The second selector switch is connected to the hardware PID.

[0065] At this point, the system is close to normal operating condition, but the debugging system is still connected to the sensor. The debugging system monitors the sensor output when there is no current to be measured. Under normal circumstances, the system output is 0 when there is no current. If it is not 0, the sensor hardware circuit needs to be troubleshooted.

[0066] Furthermore, if the third selection switch selects the voltage output channel, i.e., the sensor is configured as a voltage output type sensor, a signal compensator is connected to the output terminal of the sampling resistor. The gain of the signal compensator is adjusted according to the output voltage signal under different input currents, thus adjusting the input-output ratio to a set ratio to achieve the ideal input-output ratio. On another front, when the sensor operates at high frequencies, the gain of the signal compensator is adjusted to compensate for high-frequency output attenuation, ensuring that the sensor's high-frequency gain does not excessively decrease and increasing the sensor's operating bandwidth.

[0067] Furthermore, to ensure the sensor's performance over a wide temperature range, this embodiment of the invention includes a temperature sensor that performs self-zeroing at different temperatures and stores the gain and phase adjustment values ​​at the corresponding temperatures in a memory, thus establishing a relationship between temperature and adjustment. In this way, during normal operation, if the ambient temperature changes significantly, the signal conditioner can be automatically adjusted according to the current temperature to improve the sensor's accuracy.

[0068] The sensor designed in this invention can also enter a self-zeroing state as needed during production debugging, long-term operation, etc., to determine the initial state or to clear the accumulated error.

[0069] 2. Normal working mode.

[0070] Combination Figure 4 As shown, the circuit states in this mode include:

[0071] (1) The center frequency of the adjustable bandpass filter is set to twice the excitation frequency, and the passband does not include the excitation frequency.

[0072] (2) The first selection switch is connected to the second harmonic signal of the excitation frequency.

[0073] (3) The cutoff frequency of the first low-pass filter is adjusted to half of the excitation frequency.

[0074] (4) The second selector switch is connected to the hardware PID.

[0075] In normal operating mode, the circuit switches to the above state without dynamic adjustment. The current sensor operates in the conventional sensor state. In this embodiment of the invention, an abnormality detection circuit is added to the output terminal. When zero magnetic flux cannot be achieved due to reasons such as input exceeding limits, the sensor can be forced to enter the zero magnetic flux state by injecting large positive and negative currents into the feedback drive coil, so that the system can return to normal operating mode.

[0076] The excitation regulation circuit will be explained below to demonstrate that adjusting the amplitude and phase can effectively reduce the zero-point bias of the output signal.

[0077] (1) Ideally, such as Figure 5 As shown, U1 and U3 are followers, each driving two magnetic coils with the same inductance. The two drive signals are in opposite directions. The output signals are added using U2 to eliminate the fundamental signal and superimpose the effective signal to be measured. Ideally, the output of U2 should not contain the fundamental or odd harmonic signals of the excitation. The second and even harmonics are related to the current to be measured, and the envelope waveform of the second harmonic is consistent with the waveform of the current to be measured.

[0078] from Figure 6 and Figure 7 As can be seen, the bidirectional drive signal input to the coil ( Figure 6 ) and the signal output by the excitation coil ( Figure 7 The basic waveforms are those with the same amplitude but opposite phase. The waveforms after adding using an adder are as follows: Figure 8 As shown, its spectral components are as follows Figure 9 As shown, the waveform is not a simple fundamental wave; it contains odd harmonics. The magnitude of the fundamental wave is 72 nV. This value will be used as a reference for later analysis. Figure 1 The fundamental frequency is used as the basic data for phase-sensitive detection. The regulating controller adjusts the signal conditioner according to the magnitude of this value to ensure the consistency of the bidirectional excitation current.

[0079] (2) When the inductance of the two excitation coils differs due to the different permeability of the magnetic core, the circuit is as follows: Figure 10 As shown, L2 is adjusted to 105mH, and the excitation current of the two coils is as follows: Figure 11 The difference is evident; taking a peak difference of 59µA as an example, the fundamental frequency amplitude of the output signal exceeds that of... Figure 12 The value shown is 300µV, compared to Figure 9 The 72nV increased by more than 4000 times.

[0080] (3) If there are differences in the components in the excitation circuit, such as differences in the added resistors, for example... Figure 13 As shown, R8 is adjusted from 1KΩ to 1.01KΩ. In this circuit state, the spectrum of the output signal is as follows: Figure 14 As shown, even a tiny 1% change in resistance will affect the final output fundamental frequency amplitude. Figure 9 The voltage increased from 72nV to 64µV, and the peak difference in excitation current was 12µA.

[0081] Combining the two abnormal situations (2) and (3), it can be clearly seen that due to differences in the circuit or magnetic core, the excitation does not achieve absolute equal-amplitude and phase reversal, and the difference between the fundamental component of the output signal and the excitation current is positively correlated. Based on this, the accuracy of the excitation can be judged according to the amplitude of the fundamental wave.

[0082] To monitor the fundamental frequency, the circuit's self-zeroing mode can be used. The outputs of the first (adjustable) and second (fixed) low-pass filters are used to detect the in-phase and quadrature components of the fundamental frequency, respectively. The square root of the sum of the squares of the two components yields the amplitude of the fundamental frequency component. The regulating controller continuously adjusts the signal conditioner based on the amplitude of the fundamental frequency component, minimizing its intensity and achieving as symmetrical a bidirectional excitation as possible.

[0083] The following shows the adjustment effect of the signal conditioner.

[0084] exist Figure 10 Based on this, the controller detects an increase in the fundamental amplitude and adjusts the output gain or phase of the signal conditioner, assuming it is adjusted to 1.05KΩ via the adjustable resistor R8 (e.g., Figure 15 As shown), to increase the signal amplitude, the drive current of this path increases, and the excitation current at this time is as follows. Figure 16 As shown, the frequency of the output signal is as follows: Figure 17 As shown, with Figure 11 and 12 In comparison, the difference between the two excitation currents is now zero, and the fundamental amplitude has decreased from 300µV to 4.9µV, proving the effectiveness of the method. Theoretically, the signal conditioner can more precisely adjust the amplitude and phase of the output signal to ensure that the current of L2 is consistent with the ideal situation, that is, equal in magnitude and opposite in direction to the current of L1.

[0085] To adjust the phase of the excitation current, a capacitor can be connected in series in the signal conditioner to reduce the phase deviation caused by the increase in L2 inductance. An ideal inductor is purely inductive with a phase angle of 90°, and an ideal capacitor is -90°. Therefore, a 105mH inductor connected in series with a 5µF capacitor C is approximately equivalent to a 100mH inductor. The circuit is as follows: Figure 18 As shown, the excitation current and output impedance are as follows: Figure 19 and 20 As shown, the error in the capacitance value results in a current difference of 2µA, and the amplitude of the output fundamental frequency is 0.25µV.

[0086] Furthermore, the aforementioned circuit and analysis demonstrate that the difference in excitation current caused by the circuit and the magnetic core is reflected in the fundamental component of the output signal. By fine-tuning the signal conditioner circuit, the difference in excitation current can be effectively reduced, thus proving the effectiveness of the design scheme of this invention.

[0087] It should be noted that, in the specific implementation process, the control part mentioned above is implemented by the FPGA executing computer-executed instructions in software form stored in the memory. This will not be elaborated here. The programs corresponding to the actions performed by the control circuit can all be stored in software form in the FPGA-readable storage medium of the system, so that the FPGA can call and execute the corresponding operations of the above modules.

[0088] The computer-readable storage media mentioned above may include volatile memory, such as random access memory; may also include non-volatile memory, such as read-only memory, flash memory, hard disk or solid-state drive; and may also include combinations of the above types of memory.

[0089] The term "processor" as mentioned above can also refer to a collective of multiple processing elements. For example, a processor can be a central processing unit (CPU), or other general-purpose processors, digital signal processors, application-specific integrated circuits (ASICs), field-programmable gate arrays (FPGAs) or other programmable logic devices, discrete gate or transistor logic devices, discrete hardware components, etc. A general-purpose processor can be a microprocessor or any conventional processor, and it can also be a special-purpose processor.

[0090] It should be noted that the above description is not intended to limit the present invention, and the present invention is not limited to the examples given above. Any changes, modifications, additions or substitutions made by those skilled in the art within the scope of the present invention should also fall within the protection scope of the present invention.

Claims

1. A self-adjusting zero fluxgate current sensor, comprising: The excitation signal generation circuit is used to generate two excitation drive signals; Two excitation cores are used, each with an excitation coil wound on it. The two excitation coils are excited by two excitation drive signals respectively. Its characteristic is that it further includes: An adder is used to receive the excitation induction signals of two excitation coils and add them together for output. An adjustable bandpass filter receives the output of an adder and extracts the excitation frequency from the output of the adder in self-zeroing mode. The amplitude adjustment branch receives the excitation frequency signal and the output of the adjustable bandpass filter, and extracts the first DC component; the first DC component linearly reflects the amplitude matching error of the two excitation induction signals. The phase adjustment branch receives the quadrature excitation frequency signal and the output of the adjustable bandpass filter, and extracts the second DC component; the second DC component linearly reflects the phase deviation of the two excitation induction signals. The zero-adjustment controller receives a first DC component and a second DC component, generates a gain adjustment value based on the first DC component, and generates a phase adjustment value based on the second DC component. The signal conditioner receives gain control signal and phase control signal in self-zeroing mode, and adjusts the gain and phase of one excitation drive signal based on the gain adjustment value and phase adjustment value, so that the two excitation drive signals are equal in amplitude and opposite in phase.

2. The self-zeroing fluxgate current sensor according to claim 1, characterized in that, The amplitude adjustment branch includes a first multiplier and a first low-pass filter; the zero-adjustment controller includes a first PI controller; The first multiplier is connected to the excitation frequency signal and the output of the adjustable bandpass filter. The output of the first multiplier is connected to the input of the first low-pass filter. The output of the first low-pass filter is connected to the first PI controller.

3. The self-zeroing fluxgate current sensor according to claim 1, characterized in that, The phase adjustment branch includes a second multiplier and a second low-pass filter; the zero-adjustment controller includes a second PI controller. The second multiplier is connected to the quadrature signal of the excitation frequency and the output of the adjustable bandpass filter, and the output of the second multiplier is connected to the input of the second low-pass filter. The output of the second low-pass filter is connected to the second PI controller.

4. The self-zeroing fluxgate current sensor according to claim 2, characterized in that, The current sensor further includes a gating switch network; the gating switch network includes: The first selection switch is connected between the excitation signal generation circuit and the amplitude adjustment branch; The second selector switch is connected between the amplitude adjustment branch and the zero adjustment controller; In normal operating mode, the first gating switch is configured to connect a frequency twice the excitation frequency to the amplitude adjustment branch; the second gating switch is configured to connect the amplitude adjustment branch to the feedback branch; and the cutoff frequency of the first low-pass filter is set to half the excitation frequency, and the center frequency of the adjustable bandpass filter is set to twice the excitation frequency. In self-zeroing mode, the first gating switch is configured to connect the excitation frequency signal to the amplitude adjustment branch; the second gating switch is configured to connect the amplitude adjustment branch to the zeroing controller; and the adjustable bandpass filter is set as a low-pass filter with a cutoff frequency exceeding the excitation frequency.

5. The self-zeroing fluxgate current sensor according to claim 4, characterized in that, After completing self-zeroing, the current sensor switches to verification mode. In verification mode, the first gating switch is configured to connect a second harmonic signal of the excitation frequency to the amplitude adjustment branch; the second gating switch is configured to connect the amplitude adjustment branch to the feedback branch; the center frequency of the adjustable bandpass filter is set to a second harmonic of the excitation frequency; and the cutoff frequency of the first low-pass filter is set to half the excitation frequency. In verification mode, monitor the sensor output when there is no current to be measured. If the output is not zero, it verifies that there is a fault in the sensor hardware circuit.

6. The self-zeroing fluxgate current sensor according to claim 5, characterized in that, When the current sensor outputs a voltage, it also includes: A signal compensator, connected to the output of the sampling resistor, is used to adjust the gain of the output voltage signal according to different input currents to be measured, so as to adjust the input-output ratio to a set ratio; and to increase the system gain when operating at high frequency, so as to increase the system bandwidth of the sensor; wherein, the sampling resistor is used to convert the output current of the feedback coil into voltage.

7. The self-zeroing fluxgate current sensor according to claim 1, characterized in that, The current sensor is also equipped with a temperature sensor and is connected to the zero-adjustment controller; The zero-adjustment controller is configured to perform self-zeroing at different temperatures and store the gain adjustment value and phase adjustment value at the corresponding temperature in a memory. In normal operating mode, the corresponding adjustment value is automatically retrieved based on the temperature read by the current temperature sensor and sent to the signal conditioner.

8. The self-zeroing fluxgate current sensor according to claim 1, characterized in that, The current sensor also includes: An anomaly detection circuit, connected to the output of the sensor, is used to monitor whether the sensor can achieve zero magnetic flux. A feedback drive circuit is connected between the anomaly detection circuit and the feedback coil. When the anomaly detection circuit detects that the sensor cannot achieve zero magnetic flux, it injects large positive and negative currents into the feedback coil to force the sensor into a zero magnetic flux state. The feedback coil is wound on the main magnetic core.

Images