IQ imbalance calibration method, electronic device and storage medium

Abstract

The embodiment of the present application provides an IQ imbalance calibration method, an electronic device and a storage medium, wherein the above-mentioned random access method is applied to a wireless transceiver component, including: generating a first single-tone signal, converting the first single-tone signal into a first radio frequency single-tone signal, and transmitting the first radio frequency single-tone signal; obtaining the first baseband data transmitted by the I receiving path and the second baseband data transmitted by the Q receiving path; calculating the gain mismatch calibration coefficient and the phase mismatch calibration coefficient according to the first baseband data and the second baseband data; and compensating the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient. The IQ imbalance calibration method provided in the present application can realize self-calibration, automatically compensate according to the calibration coefficient in a non-business state, and improve calibration efficiency and user experience.

Description

[Technical field]

[0001] The embodiments of the present application relate to the field of wireless communication technology, and in particular to an IQ imbalance calibration method, an electronic device, and a storage medium. [Background technology]

[0002] In the receiver of a wireless communication system, IQ imbalance is the gain and phase mismatch between the signal in the in-phase receiving path and the signal in the quadrature-phase receiving path. IQ imbalance will produce image products, affecting the receiver's throughput, demodulation performance, etc.

[0003] At present, for the IQ imbalance of the receiver, a single tone signal is usually sent to the receiver with the help of an instrument before leaving the factory, and the gain mismatch value and phase mismatch value of the IQ two-way signal of the receiver are estimated through a calibration algorithm, and then the baseband signal of the receiver is compensated according to the mismatch value. However, this method takes up time and instrument costs and is inefficient. [Summary of the invention]

[0004] The embodiments of the present application provide an IQ imbalance calibration method, an electronic device, and a storage medium. For IQ imbalance, the electronic device or a wireless transceiver component can achieve self-calibration, reduce the impact of IQ imbalance on the communication performance of the electronic device, and improve the user experience.

[0005] In a first aspect, an embodiment of the present application provides an IQ imbalance calibration method, which is applied to a wireless transceiver component, and the method includes:

[0006] Generate a first single-tone signal, convert the first single-tone signal into a first radio frequency single-tone signal, and transmit the first radio frequency single-tone signal;

[0007] After transmitting the first radio frequency single tone signal, obtaining first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path;

[0008] Calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data;

[0009] The baseband signal of the I receiving path and / or the baseband signal of the Q receiving path are compensated according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient.

[0010] The embodiment of the present application can realize self-calibration of IQ imbalance without the need for an external transmitter or instrument through the above method, thereby saving calibration time and improving calibration efficiency. Moreover, after the self-calibration is completed, the compensation function can be automatically turned on to ensure the communication performance of the receiver.

[0011] In one possible implementation, the difference between the frequency of the first RF single-tone signal and the calibration frequency is less than half of the calibration bandwidth, the calibration frequency is the frequency used by the wireless transceiver component when down-converting the received signal, and the calibration bandwidth is the bandwidth of the filter used by the wireless transceiver component when filtering the received signal after down-conversion.

[0012] In one possible implementation, converting the first single-tone signal into a first radio frequency single-tone signal includes:

[0013] The first single-tone signal is up-converted into the first RF single-tone signal using a first local frequency, the difference between an image frequency corresponding to the first single-tone signal and a calibration frequency is greater than half of a calibration bandwidth, and the image frequency corresponding to the first single-tone signal is the first local frequency minus the frequency of the first single-tone signal.

[0014] In one possible implementation manner, a difference between a frequency of the first RF single-tone signal and a calibration frequency is greater than or equal to a first threshold.

[0015] In one possible implementation, the gain mismatch calibration coefficient includes a first gain mismatch calibration coefficient, the phase mismatch calibration coefficient includes a first phase mismatch calibration coefficient,

[0016] The calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data comprises:

[0017] The first gain mismatch calibration coefficient and the first phase mismatch calibration coefficient are calculated according to the first baseband data and the second baseband data.

[0018] In one possible implementation, the gain mismatch calibration coefficient includes a second gain mismatch calibration coefficient, and the phase mismatch calibration coefficient includes a second phase mismatch calibration coefficient.

[0019] After transmitting the first radio frequency single tone signal, the method further comprises:

[0020] Transmitting a second radio frequency single tone signal, wherein the difference between the frequency of the second radio frequency single tone signal and the calibration frequency is less than half of the calibration bandwidth, and the frequency of the second radio frequency single tone signal is different from the frequency of the first radio frequency single tone signal;

[0021] The obtaining of first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path comprises:

[0022] Acquire first baseband data and second baseband data corresponding to the first radio frequency single tone signal, and acquire first baseband data and second baseband data corresponding to the second radio frequency single tone signal;

[0023] The calculating, according to the first baseband data and the second baseband data, a gain mismatch calibration coefficient and a phase mismatch calibration coefficient comprises:

[0024] Calculate a first amplitude difference and a first phase difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path according to the first baseband data and the second baseband data corresponding to the first radio frequency single tone signal;

[0025] Calculating a second amplitude difference and a second phase difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path according to the first baseband data and the second baseband data corresponding to the second RF single tone signal;

[0026] A second gain mismatch calibration coefficient is calculated according to the first amplitude difference value and the second amplitude difference value, and a second phase mismatch calibration coefficient is calculated according to the first phase difference value and the second phase difference value.

[0027] In one possible implementation manner, a difference between a frequency of the second RF single-tone signal and the calibration frequency is greater than or equal to a first threshold.

[0028] In one possible implementation, transmitting the second radio frequency single tone signal includes:

[0029] Generate a second single-tone signal, up-convert the second single-tone signal into the second radio frequency single-tone signal using a third local frequency, and transmit the second radio frequency single-tone signal, wherein:

[0030] The difference between the image frequency corresponding to the second single-tone signal and the calibration frequency is greater than half of the calibration bandwidth, and the image frequency corresponding to the second single-tone signal is the third local frequency minus the frequency of the second single-tone signal.

[0031] In one possible implementation manner, before compensating the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient, the method includes:

[0032] Transmitting a first verification signal, and compensating a baseband signal of the I receiving path and / or a baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient;

[0033] Acquire the third baseband data transmitted by the I receiving path after compensation, and the fourth baseband data transmitted by the Q receiving path after compensation;

[0034] Calculating an image rejection ratio IRR according to the third baseband data and the fourth baseband data;

[0035] When the IRR is greater than a second threshold, the gain mismatch calibration coefficient and the phase mismatch calibration coefficient are saved.

[0036] In one possible implementation manner, calculating the image rejection ratio IRR according to the third baseband data and the fourth baseband data includes:

[0037] Calculating an amplitude difference and a phase difference between a baseband signal of the I receiving path and a baseband signal of the Q receiving path according to the third baseband data and the fourth baseband data;

[0038] The IRR is calculated according to the amplitude difference and the phase difference.

[0039] In one possible implementation, the method further includes:

[0040] Acquire current environmental parameters, where the current environmental parameters include one of temperature, standing wave ratio, adjacent channel leakage ratio, spurious radiation, receiving sensitivity, spurious response, return loss, and attenuation coefficient;

[0041] If the current environmental parameters meet the preset conditions, the step of generating the first single tone signal is entered.

[0042] In one possible implementation, the preset condition includes one or more of the following:

[0043] The current environmental parameter is greater than or equal to a third threshold;

[0044] The current environmental parameter is less than or equal to a fourth threshold;

[0045] The difference between the current environmental parameter and the historical environmental parameter is greater than or equal to a fifth threshold, and the historical environmental parameter is the environmental parameter obtained by the wireless transceiver component after generating the first single-tone signal and before calculating the gain mismatch calibration coefficient and the phase mismatch calibration coefficient.

[0046] In one possible implementation, transmitting the first radio frequency single tone signal includes:

[0047] Transmit the first radio frequency single tone signal respectively using each target power within a first power range;

[0048] After transmitting the first radio frequency single tone signal, obtaining first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path includes:

[0049] Respectively acquiring first baseband data and second baseband data corresponding to each of the target powers;

[0050] The calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data comprises:

[0051] According to the first baseband data and the second baseband data corresponding to each of the target powers, a gain mismatch calibration coefficient and a phase mismatch calibration coefficient corresponding to each of the target powers are calculated respectively.

[0052] In one possible implementation, the acquiring first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path includes:

[0053] using each target bandwidth value within the first bandwidth value range as a bandwidth value of the calibration bandwidth;

[0054] According to the calibration bandwidth corresponding to each target bandwidth value, respectively acquiring the first baseband data and the second baseband data corresponding to each target bandwidth value;

[0055] The calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data comprises:

[0056] A gain mismatch calibration coefficient and a phase mismatch calibration coefficient corresponding to each target bandwidth value are calculated respectively according to the first baseband data and the second baseband data corresponding to each target bandwidth value.

[0057] In one possible implementation, the acquiring first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path includes:

[0058] Using each target frequency value within the first calibration frequency value range as the frequency value of the calibration frequency;

[0059] According to the calibration frequency corresponding to each target frequency value, respectively acquiring the first baseband data and the second baseband data corresponding to each target frequency value;

[0060] The calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data comprises:

[0061] According to the first baseband data and the second baseband data corresponding to each of the target frequency values, a gain mismatch calibration coefficient and a phase mismatch calibration coefficient corresponding to each of the target frequency values ​​are calculated respectively.

[0062] In a second aspect, an embodiment of the present application provides a wireless transceiver component, including:

[0063] A signal generating module, used for generating a first single tone signal;

[0064] A first transmitting module, used for converting the first single-tone signal into a first radio frequency single-tone signal, and transmitting the first radio frequency single-tone signal;

[0065] A first acquisition module, configured to acquire first baseband data transmitted by an I receiving channel and second baseband data transmitted by a Q receiving channel after transmitting the first radio frequency single tone signal;

[0066] A first calculation module, used for calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data;

[0067] A compensation module is used to compensate the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient.

[0068] In one possible implementation, the difference between the frequency of the first RF single-tone signal and the calibration frequency is less than half of the calibration bandwidth, the calibration frequency is the frequency used by the wireless transceiver component when down-converting the received signal, and the calibration bandwidth is the bandwidth of the filter used by the wireless transceiver component when filtering the received signal after down-conversion.

[0069] In one possible implementation, the first transmitting module includes:

[0070] A first transmitting unit is used to up-convert the first single-tone signal into the first RF single-tone signal using a first local frequency, wherein a difference between an image frequency corresponding to the first single-tone signal and a calibration frequency is greater than half of a calibration bandwidth, and the image frequency corresponding to the first single-tone signal is the first local frequency minus a frequency of the first single-tone signal.

[0071] In one possible implementation manner, a difference between a frequency of the first RF single-tone signal and a calibration frequency is greater than or equal to a first threshold.

[0072] In one possible implementation, the gain mismatch calibration coefficient includes a first gain mismatch calibration coefficient, the phase mismatch calibration coefficient includes a first phase mismatch calibration coefficient,

[0073] The first computing module comprises:

[0074] The first calculation unit is used to calculate the first gain mismatch calibration coefficient and the first phase mismatch calibration coefficient according to the first baseband data and the second baseband data.

[0075] In one possible implementation, the gain mismatch calibration coefficient includes a second gain mismatch calibration coefficient, and the phase mismatch calibration coefficient includes a second phase mismatch calibration coefficient.

[0076] The electronic device further includes:

[0077] A second transmitting module, configured to transmit a second radio frequency single tone signal after transmitting the first radio frequency single tone signal, wherein the difference between the frequency of the second radio frequency single tone signal and the calibration frequency is less than half of the calibration bandwidth, and the frequency of the second radio frequency single tone signal is different from the frequency of the first radio frequency single tone signal;

[0078] The first acquisition module includes:

[0079] A first acquisition unit, configured to acquire first baseband data and second baseband data corresponding to the first radio frequency single tone signal, and to acquire first baseband data and second baseband data corresponding to the second radio frequency single tone signal;

[0080] The first calculation module further includes:

[0081] A second calculation unit, configured to calculate a first amplitude difference and a first phase difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path according to the first baseband data and the second baseband data corresponding to the first radio frequency single tone signal;

[0082] A third calculation unit, configured to calculate a second amplitude difference and a second phase difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path according to the first baseband data and the second baseband data corresponding to the second RF single tone signal;

[0083] The fourth calculation unit is used to calculate a second gain mismatch calibration coefficient according to the first amplitude difference and the second amplitude difference, and to calculate a second phase mismatch calibration coefficient according to the first phase difference and the second phase difference.

[0084] In one possible implementation manner, a difference between a frequency of the second RF single-tone signal and the calibration frequency is greater than or equal to a first threshold.

[0085] In one possible implementation, the second transmitting module includes:

[0086] The second transmitting unit is used to generate a second single-tone signal, up-convert the second single-tone signal into the second radio frequency single-tone signal using a third local frequency, and transmit the second radio frequency single-tone signal, wherein:

[0087] The difference between the image frequency corresponding to the second single-tone signal and the calibration frequency is greater than half of the calibration bandwidth, and the image frequency corresponding to the second single-tone signal is the third local frequency minus the frequency of the second single-tone signal.

[0088] In one possible implementation, the electronic device further includes:

[0089] a third transmitting module, configured to transmit a first verification signal before compensating the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient, and to compensate the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient;

[0090] A second acquisition module, used for acquiring the third baseband data transmitted by the I receiving path after compensation, and the fourth baseband data transmitted by the Q receiving path after compensation;

[0091] A second calculation module, used for calculating an image suppression ratio IRR according to the third baseband data and the fourth baseband data;

[0092] The storage module is used to save the gain mismatch calibration coefficient and the phase mismatch calibration coefficient when the IRR is greater than a second threshold.

[0093] In one possible implementation, the second calculation module includes:

[0094] a fifth calculation unit, configured to calculate an amplitude difference and a phase difference between a baseband signal of the I receiving path and a baseband signal of the Q receiving path according to the third baseband data and the fourth baseband data;

[0095] A sixth calculation unit is used to calculate the IRR according to the amplitude difference and the phase difference.

[0096] In one possible implementation, the electronic device further includes:

[0097] A third acquisition module is used to acquire current environmental parameters, where the current environmental parameters include one of temperature, standing wave ratio, adjacent channel leakage ratio, spurious radiation, receiving sensitivity, spurious response, return loss, and attenuation coefficient;

[0098] The trigger module is used to enter the step of generating the first single tone signal if the current environmental parameters meet the preset conditions.

[0099] In one possible implementation, the preset condition includes one or more of the following:

[0100] The current environmental parameter is greater than or equal to a third threshold;

[0101] The current environmental parameter is less than or equal to a fourth threshold;

[0102] The difference between the current environmental parameter and the historical environmental parameter is greater than or equal to a fifth threshold, and the historical environmental parameter is an environmental parameter acquired by the electronic device after generating the first single-tone signal and before calculating the gain mismatch calibration coefficient and the phase mismatch calibration coefficient.

[0103] In one possible implementation, the first transmitting module is further used to:

[0104] Transmit the first radio frequency single tone signal respectively using each target power within a first power range;

[0105] The first acquisition module is further used to respectively acquire the first baseband data and the second baseband data corresponding to each of the target powers;

[0106] The first calculation module is further used to calculate the gain mismatch calibration coefficient and the phase mismatch calibration coefficient corresponding to each target power according to the first baseband data and the second baseband data corresponding to each target power.

[0107] In one possible implementation, the first acquisition module is further configured to:

[0108] using each target bandwidth value within the first bandwidth value range as a bandwidth value of the calibration bandwidth;

[0109] According to the calibration bandwidth corresponding to each target bandwidth value, respectively acquiring the first baseband data and the second baseband data corresponding to each target bandwidth value;

[0110] The first calculation module is further used to calculate the gain mismatch calibration coefficient and the phase mismatch calibration coefficient corresponding to each target bandwidth value according to the first baseband data and the second baseband data corresponding to each target bandwidth value.

[0111] In one possible implementation, the first acquisition module is further configured to:

[0112] Using each target frequency value within the first calibration frequency value range as the frequency value of the calibration frequency;

[0113] According to the calibration frequency corresponding to each target frequency value, respectively acquiring the first baseband data and the second baseband data corresponding to each target frequency value;

[0114] The first calculation module is further used to calculate the gain mismatch calibration coefficient and the phase mismatch calibration coefficient corresponding to each target frequency value according to the first baseband data and the second baseband data corresponding to each target frequency value.

[0115] In a third aspect, an embodiment of the present application provides a chip system, comprising: a communication interface for inputting and / or outputting information; a processor for calling a computer program so that a device equipped with the chip system executes the method provided in the first aspect of the embodiment of the present application, or executes the method provided in the second aspect of the embodiment of the present application, or executes the method provided in the third aspect of the embodiment of the present application, or executes the method provided in the fourth aspect of the embodiment of the present application.

[0116] In a fourth aspect, an embodiment of the present application provides an electronic device, comprising: at least one processor; and at least one memory communicatively connected to the processor, wherein: the memory stores a computer program executable by the processor, and the processor runs the computer program to implement the method provided in the first aspect of the embodiment of the present application, or executes the method provided in the second aspect of the embodiment of the present application, or executes the method provided in the third aspect of the embodiment of the present application, or executes the method provided in the fourth aspect of the embodiment of the present application.

[0117] In a fifth aspect, an embodiment of the present application provides a computer-readable storage medium, which stores a computer program. When the computer program is run by a computer, it implements the method provided in the first aspect of the embodiment of the present application, or executes the method provided in the second aspect of the embodiment of the present application, or executes the method provided in the third aspect of the embodiment of the present application, or executes the method provided in the fourth aspect of the embodiment of the present application.

[0118] It should be understood that the sixth to eighth aspects of the embodiments of the present application are consistent with the technical solutions of the first aspect, or the second aspect, or the third aspect, or the fourth aspect of the embodiments of the present application, and the beneficial effects achieved by each aspect and the corresponding feasible implementation methods are similar and will not be repeated.

Brief Description of the Drawings

[0119] In order to more clearly illustrate the technical solutions of the embodiments of the present application, the drawings required for use in the embodiments will be briefly introduced below. Obviously, the drawings described below are only some embodiments of this specification. For ordinary technicians in this field, other drawings can be obtained based on these drawings without paying creative work.

[0120] Figure 1 is a structural diagram of a transmitter provided in an embodiment of the present application;

[0121] Figure 2 is a structural schematic diagram of another transmitter provided in an embodiment of the present application;

[0122] Figure 3 is a structural diagram of a receiver provided in an embodiment of the present application;

[0123] Figure 4 This is a schematic diagram of a mirror signal provided in an embodiment of the present application;

[0124] Figure 5 It is a schematic diagram of the structure of an electronic device provided in an embodiment of the present application;

[0125] Figure 6 This is a schematic diagram of an application scenario provided by an embodiment of the present application;

[0126] Figure 7 is a flow chart of an IQ imbalance calibration method provided in an embodiment of the present application;

[0127] Figure 8 is a schematic diagram of another electronic device structure provided by an embodiment of the present application;

[0128] Fig. 9 is a schematic diagram of another electronic device structure provided by an embodiment of the present application;

[0129] Fig.10 It is a schematic diagram of an image signal corresponding to a first single tone signal under the situation provided by an embodiment of the present application;

[0130] Fig.11 is a schematic diagram of an image signal corresponding to a first single tone signal under scenario 2 provided in an embodiment of the present application;

[0131] Fig.12 is a schematic diagram of an image signal corresponding to a first single tone signal under scenario three provided in an embodiment of the present application;

[0132] Fig.13 is a schematic diagram of an image signal corresponding to a first single tone signal under scenario 4 provided in an embodiment of the present application;

[0133] Fig.14 is a flow chart of another IQ imbalance calibration method provided in an embodiment of the present application;

[0134] Fig.15 is a flow chart of another IQ imbalance calibration method provided in an embodiment of the present application;

[0135] Fig.16 is a schematic diagram of another electronic device structure provided by an embodiment of the present application;

[0136] Fig.17 is a schematic diagram of another electronic device structure provided by an embodiment of the present application;

[0137] Fig.18 is a flow chart of another IQ imbalance calibration method provided in an embodiment of the present application;

[0138] Fig.19It is a schematic diagram of a process of sending a verification signal multiple times provided in an embodiment of the present application;

[0139] Fig. 20 is a flow chart of another IQ imbalance calibration method provided in an embodiment of the present application;

[0140] Fig.21 It is a schematic diagram of the structure of an electronic device provided by an embodiment of the present invention specification;

[0141] Fig. 22 It is a structural diagram of an electronic device provided by another embodiment of this specification. [Specific implementation method]

[0142] In the following, the terms "first" and "second" are used for descriptive purposes only and are not to be understood as indicating or implying relative importance or implicitly indicating the number of the indicated technical features. Thus, a feature defined as "first" or "second" may explicitly or implicitly include one or more of the features. In the description of this embodiment, unless otherwise specified, "plurality" means two or more.

[0143] It should be noted that, in the description of the embodiments of the present application, unless otherwise specified, " / " means or, for example, A / B can mean A or B; the "and / or" in the text is merely a description of the association relationship of associated objects, indicating that three relationships may exist, for example, A and / or B can mean: A exists alone, A and B exist at the same time, and B exists alone.

[0144] To facilitate understanding of the technical solutions provided by the embodiments of the present application, some technical terms involved in the embodiments of the present application are first explained.

[0145] Baseband Signal: The original signal sent by the transmitter without modulation (spectrum shifting). The baseband signal can carry useful information. The frequency of the baseband signal is near the zero frequency. The baseband signal can be divided into analog baseband signal and digital baseband signal.

[0146] Amplifier: used to amplify the signal amplitude. Exemplarily, in a receiver architecture, the amplifier may be a low noise amplifier (Low Noise Amplifier, LNA).

[0147] Mixer: used for up conversion (UC) or down conversion (DC) of a signal. In mathematical representation, it is the product of the signal and the sine signal / cosine signal generated by the local oscillator (LO). The sine signal / cosine signal generated by the LO can be called a local oscillator signal, or a local signal, or a local carrier, or a carrier.

[0148] For the convenience of description, in the embodiments of the present application, RX LO may be used to represent a receiver local oscillator, and TX LO may be used to represent a transmitter local oscillator.

[0149] It should be noted that, in some cases, the RX LO and the TX LO may be the same LO, and in other cases, the RX LO and the TX LO may be different LOs, which is not specifically limited in the embodiments of the present application.

[0150] Exemplarily, up-conversion can be to move the frequency of the baseband signal to near the high frequency. For example, in the transmitter, after the baseband signal passes through the mixer, the baseband signal is multiplied by the local signal generated by the TX LO, and the frequency of the baseband signal can be moved to near the local frequency of the local signal, and then converted into a radio frequency (RF) signal, and the frequency of the RF signal is the sum of the frequency of the baseband signal and the local frequency.

[0151] Exemplarily, down-conversion may be to move the frequency of the RF signal to near zero frequency. For example, after the RF signal passes through a mixer, that is, the RF signal is multiplied by the local signal generated by the RX LO, the frequency of the RF signal may be moved to near zero frequency, and then converted into a baseband signal, the frequency of the baseband signal being the difference between the frequency of the RF signal and the local frequency of the RX LO.

[0152] IQ signal: An IQ signal can be represented as two signals that are orthogonal to each other, where an I signal represents an in-phase signal and a Q signal represents a quadrature-phase signal.

[0153] In the transmitter, IQ modulation may be used to obtain an IQ signal. Exemplarily, the IQ signal may be obtained in the following manner:

[0154] See also Figure 1 In the transmitter, the baseband signal can be divided into two paths. The two signals I(t) and Q(t) are mixed separately and then summed by an adder to obtain the sum signal S(t) of I(t) and Q(t). The carriers used by the two corresponding mixers are orthogonal to each other. The mathematical formula is as follows:

[0155] S(t)=I(t)cos2πf txtQ(t)sin2πf tx t (1)

[0156] Among them, cos2πf tx t is the mathematical expression of the local signal generated by TX LO, sin2πf tx t is the mathematical expression of the local signal generated by TX LO after 90° phase shift, f tx is the local frequency of TX LO.

[0157] It should be noted that I(t) and Q(t) may be the same.

[0158] In some cases, Figure 1 The IQ modulation process shown can be simplified as Figure 2 The IQ modulation diagram is shown in Figure 1. Figure 2 As shown, the baseband signal can be expressed as I(t)+jQ(t), and the mathematical expression of the local signal generated by TX LO can be and then:

[0159] S(t)=(I(t)+jQ(t))(cos2πf tx t+j sin2πf tx t)

[0160] =I(t)cos2πf tx tQ(t)sin2πf tx t+j(I(t)sin2πf tx t+Q(t)cos2πf tx t) (2)

[0161] After taking the real part of formula (2), we can get formula (1).

[0162] Filter: Filters out signals outside a specific frequency range. Filters can include analog filters and digital filters. Analog filters are used to filter analog signals. Digital filters are used to process discrete signals.

[0163] Exemplarily, the filter may be an analog baseband (ABB) filter. In a transmitter, the ABB filter may filter the analog baseband signal, and then up-convert it to obtain a radio frequency signal. In a receiver, after the radio frequency signal is down-converted, it may be filtered by the ABB filter to obtain an analog baseband signal.

[0164] Digital-to-Analog Converter (DAC): can convert discrete digital signals into continuously changing analog signals.

[0165] Exemplarily, in a transmitter, a DAC may convert a digital baseband signal to obtain an analog baseband signal.

[0166] Analog-to-Digital Converter (ADC): can convert continuously changing analog signals into discrete digital signals.

[0167] Exemplarily, in a receiver, an ADC may convert the obtained analog baseband signal to obtain a digital baseband signal.

[0168] Receiver: used to convert RF signals into baseband signals. Taking a zero intermediate frequency receiver as an example, this article explains the basic structure of the receiver and the principle of converting RF signals into baseband signals.

[0169] See also Figure 3 The zero intermediate frequency receiver may include: an amplifier 301 , a first mixer 302 , a second mixer 303 , a first filter 304 , a second filter 305 , a first ADC 306 , and a second ADC 307 .

[0170] like Figure 3 As shown, after the amplifier 301, the receiving path is divided into an I receiving path and a Q receiving path. The RF signal of the I receiving path is down-converted by the first mixer 302, and after passing through the first filter 304, it is converted into an analog baseband signal I(t), and I(t) can be converted into a digital baseband signal I(n) by the first ADC 306. The RF signal of the Q receiving path is down-converted by the second mixer 303, and after passing through the second filter 305, it is converted into an analog baseband signal Q(t), wherein the carrier of the second mixer 303 is orthogonal to the local signal of the first mixer 302, and Q(t) can be converted into a digital baseband signal Q(n) by the second ADC 307.

[0171] IQ imbalance: In the architecture of a zero-IF receiver, due to the influence of mixers, filters, etc., the amplitude and phase of the signal in the I receiving path are not completely consistent with those in the Q receiving path. There is an amplitude difference and a phase difference, which in turn causes a gain mismatch and a phase mismatch between the signal in the I receiving path and the signal in the Q receiving path, generating a mirror signal.

[0172] In the spectrum, the frequency of the image signal and the frequency of the baseband signal are symmetrically distributed relative to the frequency of the DC signal. Figure 4 As shown in the figure, after the RF signal is down-converted to a baseband signal, its frequency is f0. If the IQ is unbalanced, the image signal will appear at -f0.

[0173] Image Reject Ratio (IRR): See Figure 4 , IRR can be the power difference or amplitude difference between the baseband signal and the image signal. IRR can be used to measure the IQ imbalance of the receiver. If IRR is greater than a threshold, it can be determined that the IQ imbalance of the receiver is not serious.

[0174] Single-tone signal: A single-tone signal is a signal with only one frequency, such as a sine wave signal or a cosine wave signal. It can be expressed by the mathematical formula: sin2πft, or e j2πft , where α is the amplitude of the single tone signal, is the phase of the single tone signal, and f is the frequency of the single tone signal.

[0175] The technical solution provided by this application is described below.

[0176] The IQ imbalance calibration method provided in the embodiment of the present application can be applied to a wireless transceiver component, which may include a baseband chip, a radio frequency chip, and a processing chip. In a possible implementation, the processing chip is integrated with the baseband chip, or the three are integrated, such as a system on chip. The IQ imbalance calibration method provided in the embodiment of the present application can also be applied to a chip system and an electronic device 500. The electronic device 500 can be a device or apparatus with a wireless transceiver function, such as a terminal, a base station, a data transmission device, a remote control device, a measurement and control device, etc.

[0177] The structure of the electronic device 500 is exemplarily described below. Figure 5 , the electronic device 500 may include: a transmitter 510 , a receiver 520 , and an antenna 530 .

[0178] The transmitter 510 may include a baseband module, a filter and a frequency converter. The baseband module is used to generate a baseband signal. After the baseband signal is processed by the filter and the mixer, it is converted into a radio frequency signal, which can be transmitted through the antenna 530.

[0179] The receiver 520 may include a mixer, a filter, an ADC, such as Figure 3 The receiver 520 can amplify, demodulate, filter, and process the received RF signal by using the mixer, filter, ADC, and the like to obtain a baseband signal.

[0180] It is to be understood that the structure of the electronic device 500 illustrated in the embodiment of the present application does not constitute a specific limitation on the electronic device 500. In some embodiments, the electronic device 500 may include: Figure 5 More or fewer components, or combining some components, or splitting some components, or different component arrangements. The components shown in the figure can be implemented in hardware, software, or a combination of software and hardware.

[0181] In order to facilitate understanding of the improvements made by the technical solution provided by the present application, the solutions in the existing related technologies are briefly described first. Based on the above description of the electronic device 500 and IQ imbalance, the electronic device 500 can test the receiver 520 before leaving the factory. If the IRR is lower than the threshold, the receiver 520 needs to be calibrated. Usually, at room temperature, a single-tone signal is generated by an instrument, and then the receiver 520 receives the single-tone signal, obtains the baseband signal in the I receiving path, and obtains the baseband signal in the Q receiving path. Through the baseband signals received by the IQ two paths and the calibration algorithm, the gain mismatch value and phase mismatch value of the IQ two paths in the receiver 520 are calculated respectively, and the compensation is applied to the baseband signal of the receiver 520, which can improve the IRR of the received signal.

[0182] However, calibrating the electronic device 500 by using the single-tone signal generated by the instrument requires time and instrument costs, which is inefficient.

[0183] Based on this, an embodiment of the present application proposes a solution. Compared with existing solutions, it does not rely on instruments and does not occupy factory calibration time, site and other resources. The electronic device 500 can self-calibrate for IQ imbalance, thereby improving calibration efficiency and user experience.

[0184] The following is a preliminary introduction to the solution provided by the embodiment of the present application, taking the electronic device 500 as a mobile phone 610, combined with two application scenario examples:

[0185] Example application scenario 1:

[0186] See also Figure 6 , the mobile phone 610 can communicate with the base station 620 through the radio access technology (Radio Access Technology, RAT). RAT can be Long Term Evolution (Long Term Evolution, LTE) technology, LTE Frequency Division Duplex (FDD) technology, LTE Time Division Duplex (TDD) technology, Universal Mobile Telecommunication System (UMTS), The 5th Generation Mobile Communication Technology (5G) or New Radio (NR) technology, or other evolved communication technologies.

[0187] The mobile phone 610 may enter the calibration process after being powered on, or the mobile phone 610 may enter the calibration process in a non-business state. For example, the non-business state may be an idle state in an idle mode after the mobile phone 610 is powered on, at which time the mobile phone 610 resides in a cell of the base station 620, reading system messages, monitoring paging information, etc. The calibration process of the mobile phone 610 includes:

[0188] The mobile phone 610 generates a single-tone signal, and then converts the single-tone signal into a radio frequency single-tone signal, which is transmitted through the antenna 530. In the process of transmitting the radio frequency single-tone signal by the transmitter 510, part of the radio frequency single-tone signal can leak into the receiving path of the mobile phone 610. After the receiver 520 receives the leaked radio frequency single-tone signal, it converts the radio frequency single tone into an IQ two-way baseband signal. The mobile phone 610 can obtain the baseband data of the I receiving path and the baseband data of the Q receiving path by calculating the calibration coefficient, and compensate the calibration coefficient on the baseband signal of the I receiving path and / or the Q receiving path to complete the calibration. In this way, the mobile phone 610 can be self-calibrated without the need for instruments and factory calibration time, thereby improving calibration efficiency.

[0189] Example application scenario 2:

[0190] After the mobile phone 610 calculates the calibration coefficient, the mobile phone 610 can re-enter the calibration process when it detects that the environmental parameters have changed. Environmental parameters may include temperature, performance parameters of the device, etc. It can be understood that when the temperature of the mobile phone 610 changes, or the device ages, the performance of the receiver 520 may change, such as the operating parameters of the amplifier, mixer, filter and other devices in the receiver 520 change, and then the IQ two-way baseband signal may be IQ unbalanced again after compensation according to the calibration coefficient. When the environmental parameters are detected to have changed, the mobile phone 610 re-enters the calibration process to better ensure the calibration effect. In addition, the mobile phone 610 can track and detect the device status of the mobile phone 610 in the power-on state or the working state to ensure the compensation effect of the calibration coefficient.

[0191] It should be noted that the electronic device 500 involved in the embodiment of the present application may be a mobile phone, a tablet computer, a computer with wireless transceiver function, a virtual reality (VR) terminal device, an augmented reality (AR) terminal device, a terminal in industrial control (Industrial Control), a vehicle-mounted terminal device, a terminal in unmanned driving, a terminal in assisted driving, a terminal in remote medical care, a terminal in smart grid (Smart Grid), a terminal in transportation safety (Transportation Safety), a terminal in smart city (Smart City), a terminal in smart home (Smart Home), etc. The embodiment of the present application does not limit the application scenario.

[0192] It can be understood that the electronic device 500 can also be called user equipment (UE), access terminal, vehicle-mounted terminal, industrial control terminal, UE unit, UE station, mobile station, mobile station, remote station, remote terminal, mobile device, UE terminal equipment, wireless communication equipment, machine terminal, UE agent or UE device, etc.

[0193] For example, in the embodiment of the present application, the electronic device 500 may also be a wearable device. Wearable devices may also be called wearable smart devices, which are a general term for wearable devices that are intelligently designed and developed using wearable technology for daily wear, such as glasses, gloves, watches, clothing, and shoes. A wearable device is a portable device that is worn directly on the body or integrated into the user's clothes or accessories.

[0194] In addition, in the embodiment of the present application, the electronic device 500 can also be a terminal device in the Internet of Things (IoT) system. IoT is an important part of the future development of information technology. Its main technical feature is to connect items to the network through communication technology, so as to realize the intelligent network of human-machine interconnection and object-to-object interconnection. The terminal device 500 of the present application can also be a vehicle-mounted module, a vehicle-mounted module, a vehicle-mounted component, a vehicle-mounted chip or a vehicle-mounted unit built into the vehicle as one or more components or units. The vehicle can implement the method of the present application through the built-in vehicle-mounted module, vehicle-mounted module, vehicle-mounted component, vehicle-mounted chip or vehicle-mounted unit. Therefore, the embodiment of the present application can be applied to the Internet of Vehicles, such as vehicle external connection (Vehicle To Everything, V2X), long-term evolution technology of workshop communication (Long Term Evolution-Vehicle, LTE-V), vehicle-to-vehicle (Vehicle-To-Vehicle, V2V), etc.

[0195] It can be understood that the electronic device 500 involved in the embodiment of the present application can also be other devices with wireless transceiver functions, such as a base station 620. The base station 620 can be a micro base station (Micro Site), a pico base station (Pico Site), a flying base station (FEMTO Site), etc. The base station 620 can also be called an access network device or a wireless access network device. The base station 620 can be, for example, a transmission reception point (Transmission Reception Point, TRP), or an evolved base station (Evolved NodeB, eNB) in LTE, or a home base station (Home Evolved NodeB), a base band unit (Base Band Unit, BBU), or a wireless controller in a cloud radio access network (Cloud Radio Access Network, CRAN) scenario. The base station 620 can be a relay station, an access point, a vehicle-mounted device, a wearable device, a network device in a 5G network, or a network device in a future evolved PLMN network, etc., for example, it can be a gNB in ​​NR.

[0196] The following takes the application of the calibration method to the electronic device 500 as an example to specifically describe the process of the calibration method provided in the embodiment of the present application.

[0197] Figure 7 is a flow chart of an IQ imbalance calibration method provided in an embodiment of the present application, such as Figure 7 As shown, the calibration method can be applied to the above electronic device 500, including:

[0198] Step 701: The electronic device 500 generates a first single-tone signal, converts the first single-tone signal into a first radio frequency single-tone signal, and transmits the first radio frequency single-tone signal.

[0199] Optionally, the first single-tone signal is a single-tone signal. The single-tone signal may be a sine signal including a frequency sin2πf0t, a cosine signal including a frequency cos2πf0t, or a superposition of the two. f0 is the frequency of the single tone signal. The single tone signal may also be a signal of other forms including one frequency, which is not specifically limited in the embodiment of the present application.

[0200] In a possible implementation, the first single tone signal may be an analog signal, and the electronic device 500 may filter the first single tone signal and transmit the filtered first single tone signal. For example, the electronic device 500 may up-convert the filtered first single tone signal into a first RF single tone signal through a mixer.

[0201] In another possible implementation, the first single-tone signal may be a digital signal, and the electronic device 500 performs digital / analog conversion on the first single-tone signal and may transmit the converted first single-tone signal.

[0202] The following takes the first single tone signal as a digital signal, and combines Figure 3 , exemplarily illustrating the structures of a transmitter 510 and a receiver 520 of an electronic device 500.

[0203] See also Figure 8 The transmitter 510 of the electronic device 500 includes: a single-tone signal generating module 801 and a transmit analog front-end (TX Analog Front-End, TX AFE) 802.

[0204] The single-tone signal generating module 801 is used to generate a single-tone signal, such as a first single-tone signal. It should be noted that the single-tone signal generated by the single-tone signal generating module 801 can be a low-frequency digital signal, and the low frequency here can refer to the frequency of the single-tone signal being near the zero frequency, for example, the difference between the frequency of the single-tone signal and the zero frequency is less than 10MHz, 20MHz, 100MHz, etc., and the embodiment of the present application does not specifically limit this.

[0205] The transmitting analog front end 802 includes a DAC 8021 , a third filter 8022 , and a third mixer 8023 .

[0206] DAC8021 is used to convert a digital signal into an analog signal. For example, DAC8021 can convert a first single tone signal into an analog signal

[0207] The third filter 8022 is used to filter the converted first single tone signal to remove useless components.

[0208] The third mixer 8023 is used to up-convert the filtered first single-tone signal into a first RF single-tone signal. The input of the third mixer 8023 includes the first single-tone signal and the local signal transmitted by TX LO, and the output of the third mixer 8023 is the first RF single-tone signal.

[0209] For convenience of expression, when transmitting the first RF single-tone signal, the first local signal is used to represent the local signal transmitted by TX LO, the frequency of the first local signal is the first local frequency, and the frequency of the first RF single-tone signal is the sum of the frequency of the first single-tone signal and the first local frequency.

[0210] It should be noted that the first RF single tone signal can pass through the RF path and finally be transmitted via the antenna 530. Figure 8 In the radio frequency path, a switch 803 or a duplexer or a switch 803 and a duplexer may be included. When the first radio frequency single tone signal passes through the switch 803 and the duplexer, part of the first radio frequency single tone signal may leak into the receiver 520, and then the receiver may receive the first radio frequency single tone signal.

[0211] See again Figure 8 , the receiver 520 of the electronic device 500 may include an amplifier 301 and a receive analog front end (RXAnalog Front-End, RX AFE) 804. The receive analog front end 804 may include a first mixer 302, a second mixer 303, a first filter 304, a second filter 305, a first ADC 306, and a second ADC 308.

[0212] The receiving path in the receiving analog front end 804 may include an I receiving path and a Q receiving path, wherein the first mixer 302, the first filter 304, and the first ADC 306 are arranged in the I receiving path, and the second mixer 304, the second filter 305, and the second ADC 307 are arranged in the Q receiving path.

[0213] It can be understood that the first RF single tone signal is converted into a digital baseband signal I(n) after passing through the first mixer 302, the first filter 304, and the first ADC 306. Correspondingly, the first RF single tone signal is converted into a digital baseband signal Q(n) after passing through the Q receiving path.

[0214] Need to explain, Figure 8 The structure shown does not constitute a specific limitation on the electronic device 500. In some embodiments, the electronic device 500 may include Figure 8More or fewer components, or combining some components, or splitting some components, or different component arrangements. The components shown in the figure can be implemented in hardware, software, or a combination of software and hardware.

[0215] also, Figure 8 The zero intermediate frequency receiver structure shown is only an exemplary description. The method provided in the present application can also be applied to a heterodyne receiver, a superheterodyne receiver, a near-zero intermediate frequency receiver, etc., and the embodiments of the present application are not limited to this.

[0216] Step 702: After sending the first radio frequency single tone signal, the electronic device 500 obtains the first baseband data transmitted by the I receiving path and the second baseband data transmitted by the Q receiving path.

[0217] Optionally, the first baseband data may be data of a digital baseband signal I(n) transmitted by the I receiving channel. For example, the first baseband data may be all data transmitted by the I receiving channel within a period of time. Alternatively, the first baseband data may be data within a data length transmitted by the I receiving channel. Alternatively, the first baseband data may be all data transmitted by the I receiving channel within a period of time or a data length, or N sample values ​​of all the data, where N is a positive integer. The embodiment of the present application does not limit the specific value of N. Similarly, the embodiment of the present application does not limit the specific value of the above-mentioned period of time or data length.

[0218] It can be understood that the second baseband data can be data of the digital baseband signal Q(n) transmitted by the Q receiving path. For example, the second baseband data can be all data transmitted by the Q receiving path within a period of time. Alternatively, the second baseband data can be all data within a data length transmitted by the Q receiving path. Alternatively, the second baseband data can be all data transmitted by the Q receiving path within a period of time or a data length, or N sample values ​​of all the data.

[0219] In step 703, the electronic device 500 calculates a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data.

[0220] Optionally, the electronic device 500 may obtain statistical data about I(n), statistical data about Q(n), and statistical data about the product of I(n) and Q(n) based on the first baseband data and the second baseband data. The electronic device 500 may calculate the gain mismatch value and the phase mismatch value between the baseband signal of the I receiving path and the baseband signal of the Q receiving path based on the above statistical data, and further calculate the gain mismatch calibration coefficient and the phase mismatch calibration coefficient of the baseband signal of the I receiving path and the baseband signal of the Q receiving path that need to be compensated.

[0221] To facilitate understanding of the technical solution for calculating the gain mismatch calibration coefficient and the phase mismatch calibration coefficient in step 703, the corresponding mathematical model is introduced below.

[0222] See also Figure 8 , since the gain mismatch value and phase mismatch value reflect the difference between the I receiving path signal and the Q receiving path signal, the gain mismatch value and the phase mismatch value can be placed in one of the paths for easy processing. Taking the example of placing the gain mismatch value and the phase mismatch value in the Q receiving path, the I receiving path is a normal signal, and the gain mismatch and phase mismatch are both reflected in the Q receiving path. The IQ imbalance relationship between the I receiving path and the Q receiving path is shown in the following formula:

[0223] I'(f)=I(f)

[0224]

[0225] Wherein, I(f) represents the first RF single tone signal received by the I receiving path, Q(f) represents the first RF single tone signal received by the Q receiving path, and f represents frequency. I'(f) represents the baseband signal after down-conversion by the first mixer 302 and filtering by the first filter 304, and Q'(f) represents the baseband signal after down-conversion by the second mixer 303 and filtering by the second filter 305. ε represents the amplitude difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path, Indicates the phase difference between the baseband signal of the I receive path and the baseband signal of the Q receive path.

[0226] It can be understood that the responses of the amplifier 301, mixer, filter and other devices in the receiver 520 to the I and Q signals are frequency-dependent, and the gain mismatch and phase mismatch caused by signals of different frequencies are different. Therefore, formula (3) is expressed as ε and Regarding the frequency f changes.

[0227] Optionally, the electronic device 500 can calculate ε and Then, the gain mismatch calibration coefficient can be obtained through ε, The phase mismatch calibration coefficient can be calculated.

[0228] In step 704, the electronic device 500 compensates for the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient.

[0229] It is understandable that due to ε and The electronic device 500 can convert ε and All compensation is performed on the baseband signal of the I receiving path or the baseband signal of the Q receiving path, or the electronic device 500 converts ε and It is divided into two parts, which compensate the baseband signal of the I receiving path and the baseband signal of the Q receiving path respectively.

[0230] Next, the electronic device 500 converts ε and Taking the baseband signal in the Q receiving path that is fully compensated as an example, the compensation method of the electronic device 500 is exemplified.

[0231] See also Fig. 9 , the electronic device 500 may sequentially set a first multiplier 901 and a first adder 902 after the second ADC 307 of the Q receiving path, and set a second multiplier 903 between the first adder 902 and the I receiving path. The gain mismatch calibration coefficient may be The phase mismatch calibration coefficient can be Then, the coefficient of the first multiplier 901 can be set to G, and the coefficient of the second multiplier 903 can be set to P, and then the baseband signal Q'(f) of the Q receiving path after compensation is:

[0232] Q'(f)=Q'(f)G+I'(f)P=Q(f) (4)

[0233] It should be noted that the above Fig. 9 The compensation method shown is only an exemplary description. Through the above exemplary description, it can be easily obtained that the first multiplier 901 and the second multiplier 903 are set on the I receiving path to realize the compensation of ε and The baseband signal technology scheme of the I receiving path is fully compensated. In addition, Fig. 9 The compensation method shown can also be easily obtained by converting ε and Compensation is implemented in the I receive path and the Q receive path separately.

[0234] The embodiment of the present application uses the above method to enable the electronic device 500 to self-calibrate the IQ imbalance after the first power-on without the need for an external transmitter or instrument, thereby saving calibration time and improving calibration efficiency. Moreover, after the electronic device 500 completes the self-calibration, the compensation function can be automatically turned on to ensure the performance of the receiver 520 of the electronic device 500.

[0235] It should be noted that IQ imbalance is related to frequency. At different frequencies, the amplitude difference and phase difference between the IQ two-way signals are different. It is necessary to balance the calibration results at different frequencies. Based on this, Figures 7 to 9 In the illustrated embodiment, the frequency of the first RF single tone signal needs to be determined by parameters used by components within the receiver 520 of the electronic device 500 .

[0236] In a possible implementation manner, a difference between a frequency of the first radio frequency single tone signal and a calibration frequency is less than half of a calibration bandwidth.

[0237] It should be noted that the calibration frequency is the frequency used by the electronic device 500 when down-converting the received signal. Figure 8 and Fig. 9 The calibration frequency may be the frequency of the local signal transmitted by the RX LO. The calibration frequency may be within the operating frequency band of the electronic device 500. The calibration bandwidth is the bandwidth of the filter used by the electronic device 500 to filter the received signal after down-conversion. For example, see Figure 8 and Fig. 9 , the calibration bandwidth may be the bandwidth of the first filter 304 or the second filter 305. The bandwidth of the first filter 304 or the second filter 305 may be 5 MHz, 10 MHz, 50 MHz, etc., which is not limited in the embodiment of the present application.

[0238] For example, the electronic device 500 uses 5G wireless access technology and the working frequency band is N41. The frequency range of N41 is 2496-2690MHz. The electronic device 500 can select one of the frequencies in 2496-2690MHz as the calibration frequency. If the calibration frequency is 2546MHz, the calibration bandwidth is 50MHz, and if the difference between the frequency of the first RF single-tone signal and the calibration frequency is less than half of the calibration bandwidth, the frequency of the first RF single-tone signal should be in the interval (2521MHz, 2571MHz).

[0239] In the calibration method provided in the embodiment of the present application, the frequency of the first RF single-tone signal transmitted by the transmitter 510 of the electronic device 500 is determined by the parameters used by the mixer and the filter in the receiver 520, so that the calibration parameters within a specific frequency range can be obtained.

[0240] It should be understood that since the performance of the components of transmitter 520 is related to the frequency, even at the same calibration frequency, different calibration bandwidths will result in different amplitude differences and phase differences between the IQ signals. In order to improve the calibration performance, the frequency of the first RF single-tone signal can be set within a specific range, for example, refer to the following embodiments.

[0241] In a possible implementation manner, a difference between a frequency of the first radio frequency single tone signal and a calibration frequency is greater than or equal to a first threshold.

[0242] Optionally, the first threshold may be greater than or equal to one fifth of the calibration bandwidth, for example, the first threshold may be one quarter of the calibration bandwidth, or the first threshold may be one third of the calibration bandwidth. In this way, the frequency of the down-converted baseband signal may be prevented from being too close to the zero frequency, so as to avoid reducing the calibration effect.

[0243] In order to prevent the non-ideal characteristics of the transmission path of the electronic device 500 from affecting the calibration, the first local frequency may be set, for example, refer to the following embodiment.

[0244] Figure 7 The step 701 of converting the first tone signal into a first radio frequency tone signal comprises:

[0245] The first single-tone signal is up-converted into a first RF single-tone signal using a first local frequency, the difference between the image frequency corresponding to the first single-tone signal and the calibration frequency is greater than half of the calibration bandwidth, and the image frequency corresponding to the first single-tone signal is the first local frequency minus the frequency of the first single-tone signal.

[0246] It can be understood that when the calibration bandwidth is large, when the frequency of the baseband signal after down-conversion is too close to the zero frequency, the gain mismatch calibration coefficient and the phase mismatch calibration coefficient of the baseband signal at this frequency cannot accurately represent the IQ imbalance of the frequency at the edge of the calibration bandwidth. The above calibration method provided in the embodiment of the present application can avoid the image frequency generated by the non-ideal characteristics of the transmission path, thereby improving the IQ imbalance calibration performance.

[0247] For the convenience of expression, f TX LO represents the first local frequency, expressed as f Tone represents the frequency of the first tone signal, expressed as f TX Tone represents the frequency of the first RF single tone signal, expressed as f Im Tone represents the image frequency corresponding to the first single tone signal, where f TX Tone =f TX LO +f Tone , f Im Tone =f TX LO -f Tone , expressed as RX Offset f TX Tone -f RX LO .

[0248] See also Fig.10 , f RX LO is 2546MHz, and BW = 20MHz, f TX Tone With f RX LO The difference is less than half of BW, that is, the value range of RX Ton Offset is within (-10MHz, 10MHz), and the corresponding f TX Tone The frequency range is within (2536MHz,2556MHz).

[0249] Assuming RX Ton Offset = -8MHz, then f TX Tone =2538MHz. Im ToneNeed to fall outside (2536MHz,2556MHz), the corresponding f TX LO and f Tone There can be a number of scenarios, as follows:

[0250] Scenario 1: If Fig.10 As shown, the image frequency f corresponding to the first single tone signal Im Tone Can be located on the left side of the bandwidth, i.e. f Im Tone <2536MHz, corresponding to f TX LO -f Tone <2536MHz. Considering f TX LO +f Tone =2538MHz, then f TX LO <2537MHz. TX LO It can be set to 2536MHz, 2534MHz, 2531MHz, etc., for example, TX LO =2534MHz, then f Tone =f TX Tone -f TX LO =4MHz, and f Im Tone =f TX LO -f Tone =2530MHz, f Im Tone Falls outside (2536MHz,2556MHz).

[0251] Scenario 2: If Fig.11 As shown, the image frequency f corresponding to the first single tone signal Im Tone Can be located on the right side of the bandwidth, i.e. f Im Tone >2556MHz, corresponding to f TX LO -f Tone >2556MHz. Considering f TX LO +f Tone =2538MHz, then f TX LO >2547MHz. TX LO It can be set to 2548MHz, 2550MHz, 2554MHz, etc. For example, f TX LO =2550MHz, then f Tone =f TX Tone -f TX LO = -12MHz, and f Im Tone =f TX LO -f Tone =2562MHz, f Im Tone Falls outside (2536MHz,2556MHz).

[0252] Assuming RX Ton Offset = 8MHz, then f TX Tone =2554MHz.Im Tone Need to fall outside (2536MHz,2556MHz), the corresponding f TX LO and f Tone There can be a number of scenarios, as follows:

[0253] Scenario 3: If Fig.12 As shown, the image frequency f corresponding to the first single tone signal Im Tone Can be located on the right side of the bandwidth, i.e. f Im Tone >2556MHz, corresponding to f TX LO -f Tone >2556MHz. Considering f TX LO +f Tone =2554MHz, then f TX LO >2555MHz. TX LO It can be set to 2556MHz, 2558MHz, 2561MHz, etc. For example, f TX LO =2558MHz, then f Tone =f TX Tone -f TX LO = -4MHz, and f Im Tone =f TX LO -f Tone =2562MHz, f Im Tone Falls outside (2536MHz,2556MHz).

[0254] Scenario 4: If Fig.13 As shown, the image frequency f corresponding to the first single tone signal Im Tone Can be located on the left side of the bandwidth, i.e. f Im Tone <2536MHz, corresponding to f TX LO -f Tone <2536MHz. Considering f TX LO +f Tone =2554MHz, then f TX LO <2545MHz. TX LO It can be set to 2544MHz, 2542MHz, 2538MHz, etc. For example, f TX LO =2542MHz, then f Tone =f TX Tone -f TX LO =12MHz, and f Im Tone =f TX LO -f Tone =2530MHz, f Im Tone Falls outside (2536MHz,2556MHz).

[0255] Through the above examples, in cases 1 and 2, within the calibration bandwidth, it can be ensured that there is a -8 MHz clean, interference-free frequency band on the receiving path; in cases 3 and 4, within the calibration bandwidth, it can be ensured that there is an 8 MHz clean, interference-free frequency band on the receiving path.

[0256] The calculation of the gain mismatch calibration coefficient and the phase mismatch calibration coefficient in step 703 is further described below.

[0257] Considering that at the same calibration frequency, when the calibration bandwidth is small, for example, when the calibration bandwidth is less than 50 MHz, the impact of some components of the electronic device 500 on the IQ imbalance may be frequency-independent, for example Figure 8 and Fig. 9 In this case, the calculation of the gain mismatch calibration coefficient and the phase mismatch calibration coefficient can refer to Fig.14 The embodiment shown.

[0258] like Fig.14 As shown, the gain mismatch calibration coefficient may include a first gain mismatch calibration coefficient, and the phase mismatch calibration coefficient may include a first phase mismatch calibration coefficient. Step 703 calculates the gain mismatch calibration coefficient and the phase mismatch calibration coefficient according to the first baseband data and the second baseband data, including:

[0259] In step 1401, the electronic device 500 calculates a first gain mismatch calibration coefficient and a first phase mismatch calibration coefficient according to the first baseband data and the second baseband data.

[0260] The following specifically describes the calculation process by placing both the gain mismatch value and the phase mismatch value in the Q path and combining the above formula (3).

[0261] Formula (3) can be expressed in a matrix as follows:

[0262]

[0263] By inverting the matrix in formula (5), we can obtain:

[0264]

[0265] Among them, in formula (6) In formula (6) That is, G is the first gain mismatch calibration coefficient, and P is the first phase mismatch calibration coefficient.

[0266] G and P can be solved based on the statistical characteristics of the signal, as shown below:

[0267] Combined with formula (3), we can get:

[0268]

[0269]

[0270] Since the statistical power of the I receiving path and the Q receiving path are equal, that is, E{I 2 (n)}=E{Q 2 (n)}, and I(n) and Q(n) without RXLO are orthogonal to each other, that is, E{I(n)Q(n)}=0, so formula (7) can be simplified to:

[0271] E{Q' 2 (n)}=(1+ε) 2 E{I 2 (n)}=(1+ε) 2 E{I' 2 (n)} (8)

[0272] Furthermore, according to formula (8), we can get:

[0273]

[0274] According to I' 2 (n) and Q' 2 The statistical characteristics of (n) can be obtained:

[0275]

[0276] According to formula (10), we can get:

[0277]

[0278] According to formula (11), the calculation formula (12) for G and the calculation formula (13) for P can be obtained as shown below:

[0279]

[0280]

[0281] It can be understood that since the first baseband data can be the data of the digital baseband signal I(n) transmitted by the I receiving path, and the second baseband data can be the data of the digital baseband signal Q(n) transmitted by the Q receiving path, E{I' 2 (n)}, E{Q' 2 (n)} and E{I'(n)Q'(n)}, and then the first gain mismatch calibration coefficient G and the first phase mismatch calibration coefficient P can be calculated by formula (12) and formula (13).

[0282] The above calibration method provided in the embodiment of the present application can calculate the gain mismatch calibration coefficient and the phase mismatch calibration coefficient when the calibration bandwidth is small through formulas (12) and (13).

[0283] Figure 7 In step 704, the electronic device 500 compensates the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient, which may include:

[0284] In step 1402, the electronic device 500 compensates for a baseband signal of an I receiving path and / or a baseband signal of a Q receiving path according to a first gain mismatch calibration coefficient and a first phase mismatch calibration coefficient.

[0285] It should be understood that Fig.14 The embodiment shown may be used Fig. 9 The compensation method described in the illustrated embodiment compensates for the baseband signal of the I receiving path and the baseband signal of the Q receiving path to complete calibration.

[0286] It should be noted that, when the calibration bandwidth is large, for example, when the calibration bandwidth is greater than 50 MHz, the impact of the related components of the receiving device 520 of the electronic device 500 on the IQ imbalance may be frequency-dependent, for example Figure 8 and Fig. 9 The first filter 304 and the second filter 305 in the embodiment are different in phase frequency response. In this case, the calculation of the gain mismatch calibration coefficient and the phase mismatch calibration coefficient can be referred to. Fig.15 The embodiment shown.

[0287] like Fig.15 As shown, the gain mismatch calibration coefficient includes a second gain mismatch calibration coefficient, and the phase mismatch calibration coefficient includes a second phase mismatch calibration coefficient. Figure 7 In step 701 shown, after transmitting the first radio frequency single tone signal, the following steps are performed:

[0288] Step 1501: the electronic device 500 transmits a second radio frequency single tone signal.

[0289] Optionally, the frequency of the second single-tone radio frequency signal is different from the frequency of the first single-tone radio frequency signal.

[0290] Optionally, the difference between the frequency of the second RF tone signal and the calibration frequency is less than half of the calibration bandwidth. Fig.10 Taking the illustrated situation 1 as an example, the frequency of the second RF single tone signal may be in the range of (2536MHz, 2538MHz), or the frequency of the second RF single tone signal may be in the range of (2538MHz, 2546MHz). Fig.12 Taking the illustrated scenario 3 as an example, the frequency of the second RF single-tone signal may be within the range of (2554 MHz, 2556 MHz), or the frequency of the second RF single-tone signal may be within the range of (2546 MHz, 2554 MHz).

[0291] Figure 7 In step 702, obtaining first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path includes:

[0292] Step 1502: The electronic device 500 obtains first baseband data and second baseband data corresponding to a first radio frequency single tone signal, and obtains first baseband data and second baseband data corresponding to a second radio frequency single tone signal.

[0293] It should be noted that since the first RF single-tone signal comes first and the second RF single-tone signal comes later, the receiver 520 of the electronic device 500 first receives the first baseband data and the second baseband data corresponding to the first RF single-tone signal, and then receives the first baseband data and the second baseband data corresponding to the second RF single-tone signal.

[0294] Figure 7 In step 703 shown, the gain mismatch calibration coefficient and the phase mismatch calibration coefficient are calculated according to the first baseband data and the second baseband data, including:

[0295] Step 1503: The electronic device 500 calculates a first amplitude difference and a first phase difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path according to the first baseband data and the second baseband data corresponding to the first RF tone signal.

[0296] In step 1504, the electronic device 500 calculates a second amplitude difference and a second phase difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path according to the first baseband data and the second baseband data corresponding to the second RF tone signal.

[0297] It can be understood that the amplitude difference ε and phase difference between the IQ baseband signals can be calculated using formulas (12) and (13): Thus, the first amplitude difference and the first phase difference between the IQ two-path baseband signals corresponding to the first RF single tone signal can be calculated, and the second amplitude difference and the second phase difference between the IQ two-path baseband signals corresponding to the second RF single tone signal can be calculated.

[0298] In step 1505, the electronic device 500 calculates a second gain mismatch calibration coefficient according to the first amplitude difference and the second amplitude difference, and calculates a second phase mismatch calibration coefficient according to the first phase difference and the second phase difference.

[0299] Optionally, the electronic device 500 calculates a second gain mismatch calibration coefficient according to the first amplitude difference and the second amplitude difference, including:

[0300] In step 1505-1, the electronic device 500 performs fitting according to the first amplitude difference and the second amplitude difference to calculate the amplitude difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path at each frequency within the calibration bandwidth.

[0301] Optionally, the electronic device 500 may use a polynomial interpolation method for fitting, such as a Lagrange interpolation method or a Newton interpolation method.

[0302] It should be noted that the electronic device 500 may also use other interpolation methods for fitting, and the embodiment of the present application does not specifically limit this.

[0303] For example, the amplitude difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path at each frequency within the calibration bandwidth can be calculated using the function ε=H G (f BW ) indicates that f BW The value is within the calibration bandwidth.

[0304] Step 1505-2: the electronic device 500 calculates a first gain mismatch calibration coefficient at each frequency within the calibration bandwidth according to the amplitude difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path at each frequency within the calibration bandwidth.

[0305] Taking the gain mismatch value placed in the Q path as an example, the implementation method of calculating the first gain mismatch calibration coefficient at each frequency within the calibration bandwidth is described.

[0306] Exemplarily, according to formula (12), the first gain mismatch calibration coefficient can be obtained: Accordingly, the first gain mismatch calibration coefficient at each frequency within the calibration bandwidth can be expressed as a function:

[0307] In step 1505 - 3 , the electronic device 500 performs an inverse Fourier transform (IFT) according to the first gain mismatch calibration coefficient at each frequency within the calibration bandwidth to obtain a first frequency response function.

[0308] Optionally, the electronic device 500 may also obtain the first frequency response function through an inverse fast Fourier transform (IFFT).

[0309] Optionally, the first frequency response function is a second gain mismatch calibration coefficient.

[0310] Step 1505 - 4 : The electronic device 500 performs fitting according to the first phase difference value and the second phase difference value, and calculates the phase difference value between the baseband signal of the I receiving path and the baseband signal of the Q receiving path at each frequency within the calibration bandwidth.

[0311] For example, the phase difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path at each frequency within the calibration bandwidth can be calculated using the function Indicates that, where f BW The value is within the calibration bandwidth.

[0312] In step 1505-5, the electronic device 500 calculates a first phase mismatch calibration coefficient at each frequency within the calibration bandwidth according to the phase difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path at each frequency within the calibration bandwidth.

[0313] Taking the case where all phase mismatch values ​​are in the Q path as an example, the implementation method of calculating the first phase mismatch calibration coefficient at each frequency within the calibration bandwidth range is described.

[0314] Exemplarily, according to formula (13), the first phase mismatch calibration coefficient can be obtained: Accordingly, the first phase mismatch calibration coefficient at each frequency within the calibration bandwidth can be expressed as a function P(f BW )=arcsin[H P (f BW )].

[0315] In step 1505 - 6 , the electronic device 500 performs an inverse Fourier transform (IFT) according to the first phase mismatch calibration coefficient at each frequency within the calibration bandwidth to obtain a second frequency response function.

[0316] Optionally, the electronic device 500 may obtain the second frequency response function through an Inverse Fast Fourier Transform (IFFT).

[0317] Optionally, the second frequency response function is a second phase mismatch calibration coefficient.

[0318] Figure 7 In step 704, the electronic device 500 compensates the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient, which may include:

[0319] Step 1506: The electronic device 500 compensates for the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the second gain mismatch calibration coefficient and the second phase mismatch calibration coefficient.

[0320] The compensation method of the electronic device 500 is described below by taking an example in which the electronic device 500 compensates all the gain mismatch calibration coefficients and the phase mismatch calibration coefficients in the baseband signal of the Q receiving path.

[0321] See also Fig.16 The electronic device 500 may set a fourth filter 1601 after the second ADC 307 of the Q receiving path, and a delay device 1602 after the first ADC 306 of the I receiving path. The fourth filter 1601 may be a finite impulse response filter (FIR).

[0322] Optionally, the fourth filter 1601 may be obtained by convolving the first FIR and the second FIR in the time domain, wherein the impulse response coefficient of the first FIR is the second gain mismatch calibration coefficient, and the impulse response coefficient of the second FIR is the second phase mismatch calibration coefficient.

[0323] Optionally, the delay device 1602 is used to compensate for the delay of the fourth filter 1601 .

[0324] Optionally, the delay value compensated by the delay device 1602 may be an integer multiple of the sampling period of the fourth filter 1601 .

[0325] In a possible implementation, the gain mismatch calibration coefficient may include a first gain mismatch calibration coefficient and a second gain mismatch calibration coefficient, and the phase mismatch calibration coefficient may include a first phase mismatch calibration coefficient and a second phase mismatch calibration coefficient. In this implementation, the electronic device 500 may compensate for the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the first gain mismatch calibration coefficient, the second gain mismatch calibration coefficient, the first phase mismatch calibration coefficient, and the second phase mismatch calibration coefficient.

[0326] Exemplarily, the electronic device 100 can also solve G in formula (12) and P in formula (13) according to the first baseband data and the second baseband data corresponding to the first RF single tone signal, and the first baseband data and the second baseband data corresponding to the second RF single tone signal.

[0327] The compensation method of the electronic device 500 is described below by taking an example in which the electronic device 500 compensates all the gain mismatch calibration coefficients and the phase mismatch calibration coefficients in the baseband signal of the Q receiving path.

[0328] See also Fig.17 ,exist Fig.16 Based on the compensation method shown, Fig. 9 In combination with the compensation method shown in FIG. 1 , the electronic device 500 can set a first multiplier 901 after the fourth filter 1601 of the Q receiving path, and set a second multiplier 903 between the first adder 902 and the I receiving path. The gain mismatch calibration coefficient can be The phase mismatch calibration coefficient can be Furthermore, the coefficient of the first multiplier 901 can be set to G, and the coefficient of the second multiplier 903 can be set to P.

[0329] It should be noted that the above Fig. 9 , Fig.16 , Fig.17 The compensation method shown is only an exemplary description. Through the above exemplary description, it is easy to obtain a technical solution for setting the relevant compensation device on the I receiving path to realize the compensation of the gain mismatch calibration coefficient and the phase mismatch calibration coefficient in the baseband signal of the I receiving path. Fig. 9 , Fig.16 , Fig.17 The compensation method shown can also be easily obtained by compensating the gain mismatch calibration coefficient and the phase mismatch calibration coefficient in the I receiving path and the Q receiving path respectively.

[0330] In the above Figures 15 to 17 In the illustrated embodiment, in order to prevent the non-ideal characteristics of the transmission path of the electronic device 500 from affecting the calibration, the second radio frequency signal may be set.

[0331] Optionally, a difference between a frequency of the second RF single-tone signal and a calibration frequency is greater than or equal to a first threshold.

[0332] Exemplarily, the first threshold value may be greater than or equal to one fifth of the calibration bandwidth, for example, the first threshold value may be one quarter of the calibration bandwidth, or the first threshold value may be one third of the calibration bandwidth. In this way, the frequency of the baseband signal after down-conversion can be prevented from being too close to the zero frequency, thereby preventing the calibration effect from being affected.

[0333] Optionally, the frequency of the second radio frequency single tone signal may be between the frequency of the first radio frequency single tone signal and the calibration frequency.

[0334] Exemplarily, the transmitting of the second radio frequency single tone signal in step 1501 includes:

[0335] A second single-tone signal is generated, the second single-tone signal is up-converted into a second radio frequency single-tone signal by using a third local frequency, and the second radio frequency single-tone signal is transmitted.

[0336] Optionally, the third local frequency may be the same as the first local frequency. In this way, the electronic device 500 may change the frequency of the second radio frequency single tone signal by adjusting the frequency of the second single tone signal.

[0337] In a possible implementation, the third local frequency may be different from the first local frequency, and the first tone signal has the same frequency as the second tone signal. In this way, the electronic device 500 may change the frequency of the second RF tone signal by adjusting the frequency of the TX LO.

[0338] Optionally, the difference between the image frequency corresponding to the second single-tone signal and the calibration frequency is greater than half of the calibration bandwidth, and the image frequency corresponding to the second single-tone signal is the third local frequency minus the frequency of the second single-tone signal. Figures 10 to 13 The corresponding situations 1 to 4 are not described in detail here. In this way, the image signal corresponding to the second single tone signal falls outside the calibration bandwidth, which can avoid the influence caused by the non-ideal characteristics of the transmission path.

[0339] It can be understood that after transmitting the second RF single-tone signal, the electronic device 500 can continue to transmit the third RF single-tone signal or the fourth RF single-tone signal. The embodiment of the present application does not specifically limit the number of times the electronic device 500 transmits the RF single-tone signal during calibration.

[0340] In a possible implementation, the electronic device 500 may transmit a plurality of RF single-tone signals in an equal frequency interval manner. For example, the frequency interval between the first RF single-tone signal and the second RF single-tone signal is a first interval value, and the frequency interval between the third RF single-tone signal and the second RF single-tone signal is the first interval value.

[0341] Optionally, the first frequency may be one tenth of the calibration bandwidth, or the first frequency may be one fifth of the calibration bandwidth. The embodiment of the present application does not specifically limit the first interval value.

[0342] In a possible implementation, the electronic device 500 may schedule the execution of the above Figures 7 to 17 The steps of the IQ imbalance calibration process provided by any embodiment.

[0343] Optionally, the time unit may be a time slot (Slot), a subframe (Sub Frame), a frame (Frame), etc., and the embodiments of the present application do not specifically limit this.

[0344] See again Figure 8 and Fig. 9An amplifier 301 is provided on the receiving path of the electronic device 500. Different amplification gains of the amplifier 301 will have different effects on the performance of the first mixer 302 and the second mixer 303. Therefore, calibration is required at different gain levels of the amplifier 301.

[0345] In a possible implementation, the electronic device 500 may use Figure 7 to Figure 17 Any of the IQ imbalance calibration methods shown calculates the gain mismatch calibration coefficient and the phase mismatch calibration coefficient at different gain levels of the amplifier 301 .

[0346] In another possible implementation, the electronic device 500 may use Figure 7 to Figure 17 Any of the IQ imbalance calibration methods shown herein calculates a gain mismatch calibration coefficient and a phase mismatch calibration coefficient at each target power within a first power range, as shown below:

[0347] The transmitting of the first radio frequency single tone signal in step 701 may include:

[0348] The first radio frequency single tone signal is transmitted respectively using each target power within the first power range.

[0349] Optionally, the first power range may be set according to the performance of the transmitter 510, which is not specifically limited in the embodiment of the present application.

[0350] Optionally, the target power may also be the power of the second radio frequency single tone signal.

[0351] After transmitting the first radio frequency single tone signal in step 702, obtaining first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path includes:

[0352] The first baseband data and the second baseband data corresponding to each target power are respectively obtained.

[0353] Step 703 of calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data includes:

[0354] A gain mismatch calibration coefficient and a phase mismatch calibration coefficient corresponding to each target power are calculated respectively according to the first baseband data and the second baseband data corresponding to each target power.

[0355] It can be understood that although the attenuation caused by the leakage of the RF single-tone signal to the receiving path through the duplexer, switch and other devices, or the attenuation of the RF single-tone signal reaching the receiving path is different in different frequency bands, the attenuation is a fixed value that can be measured. Therefore, by adjusting different transmit powers, the gain of the signal entering the receiving path can be different, thereby covering the characteristics of different gain gears of the amplifier 301. Moreover, by adjusting the transmit power to simulate the different gain gears of the amplifier 301, the problem of the single-tone signal transmitted on the two IQ receiving paths being too small or oversaturated can be avoided.

[0356] The embodiment of the present application uses the above IQ imbalance calibration method to obtain a gain mismatch calibration coefficient and a phase mismatch calibration coefficient under different transmit powers at the same calibration frequency and calibration bandwidth.

[0357] It should be understood that at the same calibration frequency and the same target power, the gain mismatch calibration coefficient and the phase mismatch calibration coefficient are different at different calibration bandwidths.

[0358] In a possible implementation, the electronic device 500 may use the IQ imbalance calibration method provided in any one of the above embodiments to calculate the gain mismatch calibration coefficient and the phase mismatch calibration coefficient at each target bandwidth value within the first bandwidth value range, where the target bandwidth value is the bandwidth value of the calibration bandwidth. Examples are as follows:

[0359] The step 702 of acquiring the first baseband data transmitted by the I receiving path and the second baseband data transmitted by the Q receiving path includes:

[0360] Each target bandwidth value within the first bandwidth value range is used as a bandwidth value for the calibration bandwidth.

[0361] According to the calibration bandwidth corresponding to each target bandwidth value, respectively obtain the first baseband data and the second baseband data corresponding to each target bandwidth value;

[0362] Step 703 of calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data includes:

[0363] A gain mismatch calibration coefficient and a phase mismatch calibration coefficient corresponding to each target bandwidth value are calculated respectively according to the first baseband data and the second baseband data corresponding to each target bandwidth value.

[0364] Through this implementation, the gain mismatch calibration coefficient and the phase mismatch calibration coefficient under different calibration bandwidths at the same calibration frequency can be obtained. Moreover, the gain mismatch calibration coefficient and the phase mismatch calibration coefficient under each target power at different calibration bandwidths at the same calibration frequency can also be obtained.

[0365] Optionally, the first bandwidth value may range from 5 MHz to 100 MHz, which is not specifically limited in the embodiment of the present application.

[0366] Optionally, the target bandwidth value may be 5 MHz, 20 MHz, 50 MHz, 60 MHz, etc. This application does not limit the specific value of the target bandwidth value.

[0367] It can be understood that, since the IQ imbalance is frequency-dependent, the gain mismatch calibration coefficient and the phase mismatch calibration coefficient are different at different calibration frequencies even with the same calibration bandwidth and target power.

[0368] In a possible implementation, the electronic device 500 can use the IQ imbalance calibration method provided in any one of the above embodiments to calculate the gain mismatch calibration coefficient and the phase mismatch calibration coefficient at each target frequency value within the first calibration frequency value range, and the target frequency value is the frequency value of the calibration frequency. Through this implementation, the gain mismatch calibration coefficient and the phase mismatch calibration coefficient at different calibration frequencies, different calibration bandwidths, and different target powers can be obtained. Examples are as follows:

[0369] The step 702 of acquiring the first baseband data transmitted by the I receiving path and the second baseband data transmitted by the Q receiving path includes:

[0370] using each target frequency value within the first calibration frequency value range as a frequency value of the calibration frequency;

[0371] According to the calibration frequency corresponding to each target frequency value, respectively obtain the first baseband data and the second baseband data corresponding to each target frequency value;

[0372] Step 703 of calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data includes:

[0373] A gain mismatch calibration coefficient and a phase mismatch calibration coefficient corresponding to each target frequency value are calculated respectively according to the first baseband data and the second baseband data corresponding to each target frequency value.

[0374] Optionally, the first calibration frequency value range may be the operating frequency band of the electronic device 500. For example, the operating frequency band of the electronic device 500 is N41, the first calibration frequency value range is 2496 to 2690 MHz, and the target frequency value may be 2496 MHz, 2546 MHz, 2596 MHz, 2646 MHz, etc.

[0375] In a possible implementation, an interval between two adjacent target frequency values ​​within the first calibration frequency value range may be half of a channel bandwidth (CBW).

[0376] It can be understood that through the IQ imbalance calibration method provided in the above embodiment, the electronic device 500 can obtain gain mismatch calibration coefficients and phase mismatch calibration coefficients under different calibration frequencies, different calibration bandwidths, and different gain levels, as shown in Table 1.

[0377] For ease of understanding, A1~L1 are used to represent the gain mismatch calibration coefficients corresponding to different gain gears, calibration frequencies, and calibration bandwidths, and A2~L2 are used to represent the phase mismatch calibration coefficients corresponding to different gain gears, calibration frequencies, and calibration bandwidths.

[0378] Table 1

[0379]

[0380]

[0381] It should be noted that Table 1 is only an illustrative example of the correspondence between the gain mismatch calibration coefficient and the phase mismatch calibration coefficient under different calibration frequencies, calibration bandwidths, and gain levels. In practice, more or fewer gain levels, calibration frequencies, or calibration bandwidths may be included. Table 1 does not constitute a specific limitation on the IQ imbalance calibration of the electronic device 500.

[0382] After the electronic device 500 calculates the gain mismatch calibration coefficient and the phase mismatch calibration coefficient, the obtained calibration coefficients may be verified, such as Fig.18 As shown, before compensating the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient in step 704, the following may be included:

[0383] In step 1801, the electronic device 500 transmits a first verification signal, and compensates for a baseband signal of an I receiving path and / or a baseband signal of a Q receiving path according to a gain mismatch calibration coefficient and a phase mismatch calibration coefficient.

[0384] Optionally, the first verification signal may be a radio frequency single tone signal.

[0385] Optionally, the electronic device 500 may up-convert the first verification tone signal into the first verification signal.

[0386] Optionally, the process of transmitting the first verification signal may be the same as the process of transmitting the first radio frequency single tone signal provided in the above embodiment, which is not described again.

[0387] It can be understood that the electronic device 500 can verify the corresponding gain mismatch calibration coefficient and phase mismatch calibration coefficient in sequence according to the gain gear, calibration frequency, and calibration bandwidth.

[0388] For example, in the order shown in Table 1, first verify the calibration coefficients A1 and A2 when the gain gear is 1, the calibration frequency is 2496 MHz, and the calibration bandwidth is 20 MHz, then verify the calibration coefficients B1 and B2 when the gain gear is 1, the calibration frequency is 2496 MHz, and the calibration bandwidth is 50 MHz, and so on, and verify in sequence.

[0389] It is understandable that the electronic device 500 can also select any gain level, calibration frequency, and calibration coefficient under the calibration bandwidth for verification, and the embodiment of the present application does not specifically limit this.

[0390] In step 1802, the electronic device 500 obtains the third baseband data transmitted by the compensated I receiving path and the fourth baseband data transmitted by the compensated Q receiving path.

[0391] Optionally, this step may refer to step 702, which will not be described in detail here.

[0392] Step 1803: The electronic device 500 calculates an image rejection ratio IRR according to the third baseband data and the fourth baseband data.

[0393] In a possible implementation, the electronic device 500 may use a frequency domain method to calculate the IRR. Exemplarily, the electronic device 500 converts the third baseband data and the fourth baseband data into a frequency domain by using a fast Fourier transform (FFT), and then the electronic device 500 may calculate the amplitude of the first verification tone signal and the amplitude of the mirror position corresponding to the first verification tone signal according to the sampling rate, the number of FFT points, and the frequency of the first verification tone signal that sends the first verification signal, and then calculate the IRR of the first verification tone signal.

[0394] Step 1804: when the IRR is greater than the second threshold, the electronic device 500 saves the gain mismatch calibration coefficient and the phase mismatch calibration coefficient.

[0395] It can be understood that the embodiment of the present application does not limit the specific value of the second threshold.

[0396] In order to improve the efficiency of the electronic device 500 in calculating the IRR, the embodiment of the present application provides a method for calculating the IRR in the time domain. Fig.18 The step 1803 may include:

[0397] Step 1803-1, the electronic device 500 calculates the amplitude difference and phase difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path according to the third baseband data and the fourth baseband data;

[0398] Step 1803 - 2 : The electronic device 500 calculates the IRR according to the amplitude difference and the phase difference.

[0399] For ease of understanding, the relevant mathematical principles are introduced first. Assuming that the received signal x(n)=I(n)+jQ(n) is obtained by the electronic device 500 after compensation, the statistical average value E{I 2}、E{Q 2} and E{IQ}, the relationship between the amplitude difference ε and the gain mismatch calibration coefficient G can be obtained according to formula (12), and then the gain mismatch value delta_gain_meas of the IQ path can be obtained, as shown in formula (14):

[0400]

[0401] According to formula (13), the phase difference can be obtained The relationship between the gain mismatch calibration coefficient P and the phase mismatch value delta_phase_meas can be obtained, as shown in formula (15):

[0402]

[0403] According to the following formula:

[0404]

[0405] You can get:

[0406]

[0407] Then, the IRR calculation formula (18) in the time domain can be obtained as follows:

[0408] IRR meas =-10*log 10 (delta_phase_meas 2 +elision_meas 2 ) (18)

[0409] Among them, elision_meas can be used to represent the amplitude difference ε between the IQ two-way signal.

[0410] In the embodiment of the present application, by calculating the IRR in the time domain, the time overhead can be reduced compared to calculating the IRR in the frequency domain.

[0411] In a possible implementation, when the calibration bandwidth is small, the electronic device 500 may send a first verification signal for verification.

[0412] In another possible implementation, when the calibration bandwidth is large, for example, the calibration bandwidth is greater than or equal to 50 MHz, the electronic device 500 may send multiple verification signals to verify the gain mismatch calibration coefficient and the phase mismatch calibration coefficient calculated by the electronic device 500.

[0413] For example, Fig.19 As shown, the process of the electronic device 500 sending multiple verification signals is as follows:

[0414] Step 1901: The electronic device 500 sends a first verification signal, and the frequency of the first verification signal minus the calibration frequency is a first frequency value.

[0415] Optionally, the first frequency value may be 0.4 times the calibration bandwidth, or the first frequency value may be 0.45 times the calibration bandwidth, etc. In this way, the compensation effect of the calibration coefficient at the edge of the calibration bandwidth may be verified.

[0416] Step 1902: The electronic device 500 calculates the IRR corresponding to the first verification signal.

[0417] In step 1903, the electronic device 500 determines whether the IRR corresponding to the first verification signal is greater than the second threshold. If so, execute step 1904; if not, enter step 701 to recalculate the gain mismatch calibration coefficient and phase mismatch calibration coefficient under the calibration frequency, calibration bandwidth and gain gear.

[0418] Step 1904: The electronic device 500 sends a second verification signal, and the calibration frequency minus the frequency of the second verification signal is equal to the first frequency value.

[0419] In this way, the compensation effect of the calibration coefficient at the edge of the other side of the calibration bandwidth can be verified.

[0420] Step 1905 : The electronic device 500 calculates the IRR corresponding to the second verification signal.

[0421] In step 1906, the electronic device 500 determines whether the IRR corresponding to the second verification signal is greater than the second threshold. If so, execute step 1907; if not, enter step 701 to recalculate the gain mismatch calibration coefficient and phase mismatch calibration coefficient under the calibration frequency, calibration bandwidth and gain gear.

[0422] Step 1907: The electronic device 500 sends a third verification signal, and the frequency of the third verification signal minus the calibration frequency is equal to the second frequency value.

[0423] Optionally, the second frequency value may be 0.1 times the calibration bandwidth, or the second frequency value may be 0.5 times the calibration bandwidth, or the second frequency value may be 0.15 times the calibration bandwidth, etc. By this setting, the compensation effect of the calibration coefficient at the center of the calibration bandwidth can be verified.

[0424] Step 1908: The electronic device 500 calculates the IRR corresponding to the third verification signal.

[0425] In step 1909, the electronic device 500 determines whether the IRR corresponding to the third verification signal is greater than the second threshold. If so, execute step 1910; if not, enter step 701 to recalculate the gain mismatch calibration coefficient and phase mismatch calibration coefficient under the calibration frequency, calibration bandwidth and gain gear.

[0426] Step 1910: The electronic device 500 sends a fourth verification signal, and the calibration frequency minus the frequency of the fourth verification signal is equal to a second frequency value.

[0427] In this way, the compensation effect of the calibration coefficients at the center of the calibration bandwidth can be verified.

[0428] Step 1911: The electronic device 500 calculates the IRR corresponding to the fourth verification signal.

[0429] In step 1912, the electronic device 500 determines whether the IRR corresponding to the fourth verification signal is greater than the second threshold value. If so, it is saved; if not, it enters step 701 to recalculate the gain mismatch calibration coefficient and phase mismatch calibration coefficient under the calibration frequency, calibration bandwidth and gain gear.

[0430] It should be understood that when the IRRs under the first verification signal to the fourth verification signal are all greater than the second threshold, the electronic device 500 saves the gain mismatch calibration coefficient and the phase mismatch calibration coefficient.

[0431] In order to improve the compensation effect of the electronic device 500 according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient, it is possible to use Fig. 20 The embodiment shown, as Fig. 20 As shown, Figures 7 to 19 Any of the IQ imbalance calibration methods shown, further comprising:

[0432] Step 2001: the electronic device 500 obtains current environmental parameters.

[0433] Optionally, the current environmental parameter includes one of temperature, standing wave ratio, adjacent channel leakage ratio, spurious radiation, receiving sensitivity, spurious response, return loss, and attenuation coefficient.

[0434] Step 2002, if the current environmental parameters meet the preset conditions, enter the step of generating the first single tone signal, that is, step 701.

[0435] The embodiment of the present application can detect changes in external conditions, such as temperature, after the electronic device 500 is calibrated through the above-mentioned IQ imbalance calibration method. When the temperature changes greatly, the calibration process can be automatically triggered in the non-business state of the electronic device 500 to ensure the compensation effect. In addition, the electronic device 500 can also track the aging of the internal components of the electronic device 500. When the communication performance indicators, such as the standing wave ratio, adjacent channel leakage ratio, and stray radiation, deteriorate, the calibration process can be triggered again to improve the compensation effect.

[0436] Optionally, the preset conditions may include one or more of the following:

[0437] Condition 1: The current environmental parameter is greater than or equal to the third threshold.

[0438] Optionally, the third threshold value may be a temperature value set by the electronic device 500, such as a temperature threshold value of a radio frequency circuit or a chip. The specific value of the third threshold value is not limited in the embodiment of the present application. In this way, when the electronic device 500 works for too long or consumes too much power, resulting in excessive temperature affecting the working performance of the device, step 701 can be re-entered to calculate the calibration coefficient to better ensure the compensation effect.

[0439] Condition 2: The current environmental parameter is less than or equal to the fourth threshold.

[0440] Optionally, the fourth threshold may be a low temperature threshold of a radio frequency circuit or chip inside the electronic device 500, or a low temperature threshold of a component inside the electronic device 500 that affects communication performance. The embodiment of the present application does not specifically limit the specific value of the fourth threshold.

[0441] Condition three: the difference between the current environmental parameter and the historical environmental parameter is greater than or equal to the fifth threshold, and the historical environmental parameter is the environmental parameter obtained by the electronic device 500 after generating the first single tone signal and before calculating the gain mismatch calibration coefficient and the phase mismatch calibration coefficient.

[0442] Optionally, the historical environmental parameters may be environmental parameters acquired between step 701 and step 703 .

[0443] The above is a description of a specific embodiment of the specification. The processes depicted in the drawings do not necessarily require the specific order or sequential order shown to achieve the desired results. In some embodiments, multitasking and parallel processing are also possible or may be advantageous.

[0444] Fig.21 The structural diagram of an electronic device provided by an embodiment of the present invention is shown in FIG. Fig.21As shown, the electronic device 500 includes: a first transmitting module 2101 , a first acquiring module 2102 , a first calculating module 2103 and a compensating module 2104 .

[0445] The first transmitting module 2101 is configured to generate a first single-tone signal, convert the first single-tone signal into a first radio frequency single-tone signal, and transmit the first radio frequency single-tone signal.

[0446] Optionally, the first transmitting module 201 may include a signal generating unit. The signal generating unit is configured to generate a first single tone signal.

[0447] The first acquisition module 2102 is used to acquire first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path.

[0448] The first calculation module 2103 is used to calculate a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data.

[0449] The compensation module 2104 is used to compensate the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient.

[0450] In one possible implementation, the difference between the frequency of the first RF single-tone signal and the calibration frequency is less than half of the calibration bandwidth, the calibration frequency is the frequency used by the electronic device when down-converting the received signal, and the calibration bandwidth is the bandwidth of the filter used by the electronic device when filtering the received signal after down-conversion.

[0451] In one possible implementation, the first transmitting module 2101 includes:

[0452] The first transmitting unit is used to up-convert the first single-tone signal into a first RF single-tone signal using a first local frequency, the difference between the image frequency corresponding to the first single-tone signal and the calibration frequency is greater than half of the calibration bandwidth, and the image frequency corresponding to the first single-tone signal is the first local frequency minus the frequency of the first single-tone signal.

[0453] In one possible implementation manner, a difference between a frequency of the first RF single-tone signal and a calibration frequency is greater than or equal to a first threshold.

[0454] In one possible implementation manner, the gain mismatch calibration coefficient includes a first gain mismatch calibration coefficient, and the phase mismatch calibration coefficient includes a first phase mismatch calibration coefficient.

[0455] Optionally, the first calculation module 2103 includes:

[0456] The first calculation unit is used to calculate a first gain mismatch calibration coefficient and a first phase mismatch calibration coefficient according to the first baseband data and the second baseband data.

[0457] In one possible implementation manner, the gain mismatch calibration coefficient includes a second gain mismatch calibration coefficient, and the phase mismatch calibration coefficient includes a second phase mismatch calibration coefficient.

[0458] Optionally, the electronic device 500 further includes:

[0459] The second transmitting module is used to transmit a second RF single-tone signal after transmitting the first RF single-tone signal, wherein the difference between the frequency of the second RF single-tone signal and the calibration frequency is less than half of the calibration bandwidth and is different from the frequency of the first RF single-tone signal.

[0460] Optionally, the first acquisition module 2102 includes:

[0461] The first acquisition unit is used to acquire the first baseband data and the second baseband data corresponding to the first radio frequency single tone signal, and to acquire the first baseband data and the second baseband data corresponding to the second radio frequency single tone signal.

[0462] Optionally, the first calculation module 2103 further includes:

[0463] A second calculation unit is used to calculate a first amplitude difference and a first phase difference between a baseband signal of an I receiving path and a baseband signal of a Q receiving path according to first baseband data and second baseband data corresponding to the first radio frequency single tone signal;

[0464] A third calculation unit, configured to calculate a second amplitude difference and a second phase difference between a baseband signal of an I receiving path and a baseband signal of a Q receiving path according to the first baseband data and the second baseband data corresponding to the second radio frequency single tone signal;

[0465] The fourth calculation unit is used to calculate a second gain mismatch calibration coefficient according to the first amplitude difference and the second amplitude difference, and to calculate a second phase mismatch calibration coefficient according to the first phase difference and the second phase difference.

[0466] In one possible implementation manner, a difference between a frequency of the second RF single-tone signal and a calibration frequency is greater than or equal to a first threshold.

[0467] In one possible implementation, the second transmitting module includes:

[0468] The second transmitting unit is used to generate a second single-tone signal, up-convert the second single-tone signal into a second radio frequency single-tone signal using a third local frequency, and transmit the second radio frequency single-tone signal, wherein:

[0469] The difference between the image frequency corresponding to the second single-tone signal and the calibration frequency is greater than half of the calibration bandwidth, and the image frequency corresponding to the second single-tone signal is the third local frequency minus the frequency of the second single-tone signal.

[0470] In one possible implementation, the electronic device 500 further includes:

[0471] a third transmitting module, configured to transmit a first verification signal before compensating the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient, and to compensate the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient;

[0472] A second acquisition module, used to acquire the third baseband data transmitted by the compensated I receiving path and the fourth baseband data transmitted by the compensated Q receiving path;

[0473] A second calculation module, used for calculating an image suppression ratio IRR according to the third baseband data and the fourth baseband data;

[0474] The storage module is used to store the gain mismatch calibration coefficient and the phase mismatch calibration coefficient when the IRR is greater than the second threshold.

[0475] In one possible implementation, the second computing module includes:

[0476] a fifth calculation unit, configured to calculate an amplitude difference and a phase difference between a baseband signal of an I receiving path and a baseband signal of a Q receiving path according to the third baseband data and the fourth baseband data;

[0477] The sixth calculation unit is used to calculate the IRR according to the amplitude difference and the phase difference.

[0478] In one possible implementation, the electronic device 500 further includes:

[0479] A third acquisition module is used to obtain current environmental parameters, where the current environmental parameters include one of temperature, standing wave ratio, adjacent channel leakage ratio, spurious radiation, receiving sensitivity, spurious response, return loss, and attenuation coefficient;

[0480] The trigger module is used to enter the step of generating a first single tone signal if the current environmental parameters meet the preset conditions.

[0481] In one possible implementation, the preset conditions include one or more of the following:

[0482] The current environmental parameter is greater than or equal to a third threshold;

[0483] The current environmental parameter is less than or equal to the fourth threshold;

[0484] The difference between the current environment parameter and the historical environment parameter is greater than or equal to a fifth threshold, and the historical environment parameter is an environment parameter acquired by the electronic device after generating the first single tone signal and before calculating the gain mismatch calibration coefficient and the phase mismatch calibration coefficient.

[0485] Fig.21 The electronic device 500 provided in the illustrated embodiment can be used to execute the present specification Figures 7 to 20 The technical solution of the method embodiment shown, its implementation principle and technical effects can be further referred to the relevant description in the method embodiment.

[0486] The IQ imbalance calibration method provided in the embodiment of the present application can be performed by the following devices: a chip or a chip module. Regarding the various modules / units contained in the various devices and products described in the above embodiments, they can be software modules / units, or hardware modules / units, or they can be partially software modules / units and partially hardware modules / units. For example, for various devices and products applied to or integrated in a chip, the various modules / units contained therein can all be implemented in the form of hardware such as circuits, or at least some of the modules / units can be implemented in the form of software programs, which run on a processor integrated inside the chip. Different modules / units may be located in the same component (e.g., chip, circuit module, etc.) or in different components of the chip module, or at least some of the modules / units may be implemented in the form of a software program that runs on a processor integrated inside the chip module, and the remaining (if any) modules / units may be implemented in hardware such as circuits; for various devices and products applied to or integrated in a terminal, the various modules / units contained therein may all be implemented in hardware such as circuits, and different modules / units may be located in the same component (e.g., chip, circuit module, etc.) or in different components in the terminal, or at least some of the modules / units may be implemented in the form of a software program that runs on a processor integrated inside the terminal, and the remaining (if any) modules / units may be implemented in hardware such as circuits.

[0487] The embodiment of the present application also provides a chip system, including: a communication interface for inputting and / or outputting information; a processor for calling a computer program so that a device equipped with the chip system executes the instructions in the present specification. Figures 7 to 20 The illustrated embodiment provides an IQ imbalance calibration method.

[0488] Fig. 22 FIG. 5 is a schematic diagram of the structure of an electronic device 500 provided in another embodiment of the present specification. Fig. 22As shown, the electronic device 500 may include at least one processor; and at least one memory in communication with the processor, wherein: the memory stores a computer program executable by the processor, and the processor calls the computer program to execute the instructions of this specification. Figures 7 to 20 The illustrated embodiment provides an IQ imbalance calibration method.

[0489] The electronic device 500 may be a smart electronic device such as a smart phone or a tablet computer, and the present embodiment does not limit the form of the electronic device 500.

[0490] For example, Fig. 22 A schematic diagram of the structure of the electronic device 500 is shown by taking a smart phone as an example. Fig. 22 As shown, the electronic device 500 may include a processor 110, an internal memory 121, an antenna 1, an antenna 2, a mobile communication module 150, a wireless communication module 160, and a subscriber identification module (SIM) card interface 195, etc.

[0491] It is to be understood that the structure illustrated in the embodiment of the present application does not constitute a specific limitation on the electronic device 500. In other embodiments of the present application, the electronic device 500 may include more or fewer components than shown in the figure, or combine some components, or split some components, or arrange the components differently. The components shown in the figure may be implemented in hardware, software, or a combination of software and hardware.

[0492] The processor 110 may include one or more processing units, for example, the processor 110 may include an application processor (AP), a modem processor, a graphics processor (GPU), an image signal processor (ISP), a controller, a video codec, a digital signal processor (DSP), a baseband processor, and / or a neural-network processing unit (NPU), etc. Different processing units may be independent devices or integrated into one or more processors.

[0493] The controller can generate operation control signals according to the instruction operation code and timing signal to complete the control of instruction fetching and execution.

[0494] The processor 110 may also be provided with a memory for storing instructions and data. In some embodiments, the memory in the processor 110 is a cache memory. The memory may store instructions or data that the processor 110 has just used or circulated. If the processor 110 needs to use the instruction or data again, it may be directly called from the memory.

[0495] The processor 110 executes various functional applications and data processing by running the programs stored in the internal memory 121, such as implementing the present application Figures 3 to 10 The method provided by the illustrated embodiment.

[0496] In some embodiments, the processor 110 may include one or more interfaces. The interface may include an inter-integrated circuit (I2C) interface, an inter-integrated circuit sound (I2S) interface, a pulse code modulation (PCM) interface, a universal asynchronous receiver / transmitter (UART) interface, a mobile industry processor interface (MIPI), a general-purpose input / output (GPIO) interface, a subscriber identity module (SIM) interface, and / or a universal serial bus (USB) interface, etc.

[0497] The wireless communication function of the electronic device 500 can be implemented through the antenna 1, the antenna 2, the mobile communication module 150, the wireless communication module 160, the modem processor and the baseband processor.

[0498] Antenna 1 and antenna 2 are used to transmit and receive electromagnetic wave signals. Each antenna in electronic device 500 can be used to cover a single or multiple communication frequency bands. Different antennas can also be reused to improve the utilization of antennas. For example, antenna 1 can be reused as a diversity antenna for a wireless local area network. In some other embodiments, the antenna can be used in combination with a tuning switch.

[0499] The mobile communication module 150 can provide solutions for wireless communications including 2G / 3G / 4G / 5G, etc., applied to the electronic device 500. The mobile communication module 150 may include at least one filter, a switch, a power amplifier, a low noise amplifier (LNA), etc. The mobile communication module 150 can receive electromagnetic waves from the antenna 1, and filter, amplify, and process the received electromagnetic waves, and transmit them to the modulation and demodulation processor for demodulation. The mobile communication module 150 can also amplify the signal modulated by the modulation and demodulation processor, and convert it into electromagnetic waves for radiation through the antenna 1. In some embodiments, at least some of the functional modules of the mobile communication module 150 can be set in the processor 110. In some embodiments, at least some of the functional modules of the mobile communication module 150 can be set in the same device as at least some of the modules of the processor 110.

[0500] The modem processor may include a modulator and a demodulator. The modulator is used to modulate the low-frequency baseband signal to be sent into a medium-high frequency signal. The demodulator is used to demodulate the received electromagnetic wave signal into a low-frequency baseband signal. The demodulator then transmits the demodulated low-frequency baseband signal to the baseband processor for processing. After the low-frequency baseband signal is processed by the baseband processor, it is passed to the application processor. In some embodiments, the modem processor may be an independent device. In other embodiments, the modem processor may be independent of the processor 110 and be set in the same device as the mobile communication module 150 or other functional modules.

[0501] The wireless communication module 160 can provide wireless communication solutions including wireless local area networks (WLAN) (such as wireless fidelity (Wi-Fi) networks), bluetooth (BT), global navigation satellite system (GNSS), frequency modulation (FM), near field communication (NFC), infrared (IR), etc., which are applied to the electronic device 500. The wireless communication module 160 can be one or more devices integrating at least one communication processing module. The wireless communication module 160 receives electromagnetic waves via the antenna 2, modulates the frequency of the electromagnetic wave signal and performs filtering, and sends the processed signal to the processor 110. The wireless communication module 160 can also receive the signal to be sent from the processor 110, modulate the frequency of it, amplify it, and convert it into electromagnetic waves for radiation through the antenna 2.

[0502] In some embodiments, the antenna 1 of the electronic device 500 is coupled to the mobile communication module 150, and the antenna 2 is coupled to the wireless communication module 160, so that the electronic device 500 can communicate with the network and other devices through wireless communication technology. The wireless communication technology may include global system for mobile communications (GSM), general packet radio service (GPRS), code division multiple access (CDMA), wideband code division multiple access (WCDMA), time-division code division multiple access (TD-SCDMA), long term evolution (LTE), BT, GNSS, WLAN, NFC, FM, and / or IR technology, etc.

[0503] NPU is a neural network (NN) computing processor, which can quickly process input information and continuously self-learn by drawing on the structure of biological neural networks, such as the transmission mode between neurons in the human brain. NPU can realize applications such as intelligent cognition of electronic device 500, such as image recognition, face recognition, voice recognition, text understanding, etc.

[0504] The internal memory 121 can be used to store computer executable program codes, which include instructions. The internal memory 121 may include a program storage area and a data storage area. Among them, the program storage area may store an operating system, at least one application required for a function, etc. The data storage area may store data created during the use of the electronic device 500, etc. In addition, the internal memory 121 may include a high-speed random access memory, and may also include a non-volatile memory, such as at least one disk storage device, a flash memory device, a universal flash storage (UFS), etc. The processor 110 executes various functional applications and data processing of the electronic device 500 by running instructions stored in the internal memory 121 and / or instructions stored in a memory provided in the processor.

[0505] The SIM card interface 195 is used to connect a SIM card. The SIM card can be connected to and separated from the electronic device 500 by inserting it into the SIM card interface 195 or pulling it out from the SIM card interface 195. The electronic device 500 can support 1 or N SIM card interfaces, where N is a positive integer greater than 1. The SIM card interface 195 can support Nano SIM cards, Micro SIM cards, SIM cards, and the like. Multiple cards can be inserted into the same SIM card interface 195 at the same time. The types of the multiple cards can be the same or different. The SIM card interface 195 can also be compatible with different types of SIM cards. The SIM card interface 195 can also be compatible with external memory cards. The electronic device 500 interacts with the network through the SIM card to implement functions such as calls and data communications. In some embodiments, the electronic device 500 uses an eSIM, i.e., an embedded SIM card. The eSIM card can be embedded in the electronic device 500 and cannot be separated from the electronic device 500.

[0506] The embodiment of the present application also provides a computer storage medium, including a computer program. When the computer program is run on the electronic device 500, the electronic device 500 executes the following Figure 7 to Figure 20 The steps performed by the electronic device 500 in the method shown in any embodiment.

[0507] It should be understood that the term "unit" in the embodiments of the present application can be implemented in the form of software and / or hardware, and is not specifically limited to this. For example, a "unit" can be a software program, a hardware circuit, or a combination of the two that implements the above functions. The hardware circuit may include an application specific integrated circuit (ASIC), an electronic circuit, a processor (such as a shared processor, a dedicated processor, or a group processor, etc.) and a memory for executing one or more software or firmware programs, a merged logic circuit, and / or other suitable components that support the described functions.

[0508] Therefore, the units of each example described in the embodiments of the present application can be implemented by electronic hardware, or a combination of computer software and electronic hardware. Whether these functions are performed in hardware or software depends on the specific application and design constraints of the technical solution. Professional and technical personnel can use different methods to implement the described functions for each specific application, but such implementation should not be considered to be beyond the scope of the present application.

[0509] In several embodiments provided in the present application, any function can be stored in a computer-readable storage medium if it is implemented in the form of a software functional unit and sold or used as an independent product. Based on this understanding, the technical solution of the present application can be embodied in the form of a software product in essence or in other words, the part that contributes to the prior art or the part of the technical solution. The computer software product is stored in a storage medium, including several instructions for a computer device (which can be a personal computer, a server, or a network device, etc.) to perform all or part of the steps of the method described in each embodiment of the present application. The aforementioned storage medium includes: various media that can store program codes, such as a USB flash drive, a mobile hard disk, a read-only memory (ROM), a random access memory (RAM), a magnetic disk or an optical disk.

Claims

1. A method for calibrating an IQ imbalance, characterized in that: Applied to wireless transceiver components, including: Generate a first single-tone signal, convert the first single-tone signal into a first radio frequency single-tone signal, and transmit the first radio frequency single-tone signal, wherein the difference between the frequency of the first radio frequency single-tone signal and the calibration frequency is less than half of the calibration bandwidth, the difference between the frequency of the first radio frequency single-tone signal and the calibration frequency is greater than or equal to a first threshold, the calibration frequency is the frequency used by the wireless transceiver component when down-converting the received signal, and the calibration bandwidth is the bandwidth of the filter used by the wireless transceiver component when filtering the received signal after down-conversion; After transmitting the first radio frequency single tone signal, obtaining first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path; Calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data; Compensating the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient; The method further includes: during the process of transmitting the first RF single tone signal, receiving the leaked first RF single tone signal through a receiving path, and converting the first RF single tone into a baseband signal of an I receiving path and / or a baseband signal of the Q receiving path.

2. The method according to claim 1, characterized in that: The converting the first single tone signal into a first radio frequency single tone signal comprises: The first single-tone signal is up-converted into the first RF single-tone signal using a first local frequency, the difference between an image frequency corresponding to the first single-tone signal and a calibration frequency is greater than half of a calibration bandwidth, and the image frequency corresponding to the first single-tone signal is the first local frequency minus the frequency of the first single-tone signal.

3. The method according to any one of claims 1 to 2, characterized in that: The gain mismatch calibration coefficient includes a first gain mismatch calibration coefficient, and the phase mismatch calibration coefficient includes a first phase mismatch calibration coefficient. The calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data comprises: The first gain mismatch calibration coefficient and the first phase mismatch calibration coefficient are calculated according to the first baseband data and the second baseband data.

4. The method according to any one of claims 1 to 2, characterized in that: The gain mismatch calibration coefficient includes a second gain mismatch calibration coefficient, and the phase mismatch calibration coefficient includes a second phase mismatch calibration coefficient, After transmitting the first radio frequency single tone signal, the method further comprises: Transmitting a second radio frequency single tone signal, wherein the difference between the frequency of the second radio frequency single tone signal and the calibration frequency is less than half of the calibration bandwidth, and the frequency of the second radio frequency single tone signal is different from the frequency of the first radio frequency single tone signal; The obtaining of first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path comprises: Acquire first baseband data and second baseband data corresponding to the first radio frequency single tone signal, and acquire first baseband data and second baseband data corresponding to the second radio frequency single tone signal; The calculating, according to the first baseband data and the second baseband data, a gain mismatch calibration coefficient and a phase mismatch calibration coefficient comprises: Calculate a first amplitude difference and a first phase difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path according to the first baseband data and the second baseband data corresponding to the first radio frequency single tone signal; Calculating a second amplitude difference and a second phase difference between the baseband signal of the I receiving path and the baseband signal of the Q receiving path according to the first baseband data and the second baseband data corresponding to the second RF single tone signal; A second gain mismatch calibration coefficient is calculated according to the first amplitude difference value and the second amplitude difference value, and a second phase mismatch calibration coefficient is calculated according to the first phase difference value and the second phase difference value.

5. The method according to claim 4, characterized in that A difference between a frequency of the second radio frequency single tone signal and the calibration frequency is greater than or equal to a first threshold.

6. The method according to claim 4, characterized in that The transmitting of the second radio frequency single tone signal comprises: Generate a second single-tone signal, up-convert the second single-tone signal into the second radio frequency single-tone signal using a third local frequency, and transmit the second radio frequency single-tone signal, wherein: The difference between the image frequency corresponding to the second single-tone signal and the calibration frequency is greater than half of the calibration bandwidth, and the image frequency corresponding to the second single-tone signal is the third local frequency minus the frequency of the second single-tone signal.

7. The method according to claim 1, characterized in that Before compensating the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient, the method includes: Transmitting a first verification signal, and compensating a baseband signal of the I receiving path and / or a baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient; Acquire the third baseband data transmitted by the I receiving path after compensation, and the fourth baseband data transmitted by the Q receiving path after compensation; Calculating an image rejection ratio IRR according to the third baseband data and the fourth baseband data; When the IRR is greater than a second threshold, the gain mismatch calibration coefficient and the phase mismatch calibration coefficient are saved.

8. The method according to claim 7, characterized in that The calculating the image rejection ratio IRR according to the third baseband data and the fourth baseband data comprises: Calculating an amplitude difference and a phase difference between a baseband signal of the I receiving path and a baseband signal of the Q receiving path according to the third baseband data and the fourth baseband data; The IRR is calculated according to the amplitude difference and the phase difference.

9. The method according to claim 1, characterized in that: The method further comprises: Acquire current environmental parameters, where the current environmental parameters include one of temperature, standing wave ratio, adjacent channel leakage ratio, spurious radiation, receiving sensitivity, spurious response, return loss, and attenuation coefficient; If the current environmental parameters meet the preset conditions, the step of generating the first single tone signal is entered.

10. The method according to claim 9, characterized in that The preset conditions include one or more of the following: The current environmental parameter is greater than or equal to a third threshold; The current environmental parameter is less than or equal to a fourth threshold; The difference between the current environmental parameter and the historical environmental parameter is greater than or equal to a fifth threshold, and the historical environmental parameter is the environmental parameter obtained by the wireless transceiver component after generating the first single-tone signal and before calculating the gain mismatch calibration coefficient and the phase mismatch calibration coefficient.

11. The method according to claim 1, characterized in that: The transmitting the first radio frequency single tone signal comprises: Transmit the first radio frequency single tone signal respectively using each target power within a first power range; After transmitting the first radio frequency single tone signal, obtaining first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path includes: Respectively acquiring first baseband data and second baseband data corresponding to each of the target powers; The calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data comprises: According to the first baseband data and the second baseband data corresponding to each of the target powers, a gain mismatch calibration coefficient and a phase mismatch calibration coefficient corresponding to each of the target powers are calculated respectively.

12. The method according to claim 1, characterized in that The obtaining of first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path comprises: using each target bandwidth value within the first bandwidth value range as a bandwidth value of the calibration bandwidth; According to the calibration bandwidth corresponding to each target bandwidth value, respectively acquiring the first baseband data and the second baseband data corresponding to each target bandwidth value; The calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data comprises: A gain mismatch calibration coefficient and a phase mismatch calibration coefficient corresponding to each target bandwidth value are calculated respectively according to the first baseband data and the second baseband data corresponding to each target bandwidth value.

13. The method according to claim 1, characterized in that The obtaining of first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path comprises: Using each target frequency value within the first calibration frequency value range as the frequency value of the calibration frequency; According to the calibration frequency corresponding to each target frequency value, respectively acquiring the first baseband data and the second baseband data corresponding to each target frequency value; The calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data comprises: A gain mismatch calibration coefficient and a phase mismatch calibration coefficient corresponding to each target frequency value are calculated respectively according to the first baseband data and the second baseband data corresponding to each target frequency value.

14. A wireless transceiver component, characterized in that: include: A signal generating module, used for generating a first single tone signal; A first transmitting module, used for converting the first single-tone signal into a first radio frequency single-tone signal, and transmitting the first radio frequency single-tone signal, wherein the difference between the frequency of the first radio frequency single-tone signal and the calibration frequency is less than half of the calibration bandwidth, the difference between the frequency of the first radio frequency single-tone signal and the calibration frequency is greater than or equal to a first threshold, the calibration frequency is the frequency used by the wireless transceiver component when down-converting the received signal, and the calibration bandwidth is the bandwidth of the filter used by the wireless transceiver component when filtering the received signal after down-conversion; A first acquisition module, used for acquiring first baseband data transmitted by the I receiving path and second baseband data transmitted by the Q receiving path; A first calculation module, used for calculating a gain mismatch calibration coefficient and a phase mismatch calibration coefficient according to the first baseband data and the second baseband data; A compensation module, configured to compensate the baseband signal of the I receiving path and / or the baseband signal of the Q receiving path according to the gain mismatch calibration coefficient and the phase mismatch calibration coefficient; The conversion module is used to receive the leaked first RF single-tone signal through the receiving path during the process of transmitting the first RF single-tone signal, and convert the first RF single-tone into a baseband signal of the I receiving path and / or a baseband signal of the Q receiving path.

15. A chip system, characterized in that: include: A communication interface for inputting and / or outputting information; A processor, configured to call a computer program so that a device equipped with the chip system executes the method according to any one of claims 1 to 13.

16. An electronic device, characterized in that: include: at least one processor; as well as at least one memory in communication with the processor, wherein: The memory stores a computer program executable by the processor, and the processor runs the computer program to implement the method according to any one of claims 1 to 13.

17. A computer-readable storage medium, characterized in that: The computer-readable storage medium stores a computer program, and the computer program implements the method according to any one of claims 1 to 13 when executed by a computer.

Images