Radiation coupling antenna with network

Abstract

The invention relates to an antenna system having two identical, mutually radiation-coupled antennas arranged symmetrically to one another, which antennas each have an antenna connection port; and a network connected on the input side to the antenna connection ports and composed of reactive elements, which network has, for each antenna, an antenna path to a network connection port which is individually assigned to the antenna and is identical to the other antenna path.

Description

Technical Field

[0001] The present invention relates to an antenna system 1 having two identical, mutually radiating and coupled, symmetrically arranged antennas 1a and 1b, each having antenna connection ports 2a and 2b; and a network 3 connected to the antenna connection ports 2a and 2b on the input side and composed of blend elements, the network having antenna paths 15a and 15b separately assigned to the antennas 1a and 1b to the network connection ports 4a and 4b. Background Technology

[0002] For antennas used on vehicles, the required surface area, structural height, aerodynamic shape, and drag coefficient are particularly important. However, due to the typically large production volumes in vehicle manufacturing, the economics of manufacturing such antennas are especially crucial.

[0003] Many modern mobile radio networks, such as those designed or still under development according to the LTE (Long Term Evolution) mobile radio standard, typically require multiple identical antennas that must be arranged close together due to limited space available in vehicle technology. The LTE mobile radio standard, for example, provides a frequency range between 698 and 960 MHz. To design a pair of identical LTE array antennas, these array antennas can, as shown in patent document EP 3 734 755 A1, be advantageously, for example, mechanically and electrically connected to the network on a circuit board and connected to separate transmitting and receiving devices via separately designed antenna paths on that circuit board.

[0004] In this case, for very dense structures and those in free space with electrical wavelengths For example, with antennas arranged perpendicularly and parallel to each other over a small distance, electromagnetic coupling between the two antennas can cause interference from the strong transmitted signal of the other transmitter-receiver in a transmitter-receiver configured to receive. This is especially problematic in the case of digital signal transmission, as it significantly increases the bit error rate during reception.

[0005] Of course, the applications of the antennas and antenna arrangements presented in this document are by no means limited to the LTE systems exemplified herein. Furthermore, these antennas and network arrangements are particularly advantageous for use in all communication systems where the antenna system operates within the described frequency range, such as in communication systems like 5G, WLAN, and vehicle-to-vehicle (Car-to-Car) communication, for example, according to standards such as IEEE 802.11p. For these applications, antennas are needed that, in addition to their electrical functions, are suitable for vehicles due to their compactness and stylistic characteristics, where manufacturing economics are particularly important. Summary of the Invention

[0006] Therefore, the object of the present invention is to provide a network 3 in the case of a networked radiating coupled antenna according to the preamble of the present invention, by means of which the electromagnetic radiating coupling between the antennas is significantly reduced to the residual coupling that occurs between the two outputs of the network 3, and the antenna system with network 3 is compact in structure for mobile radio services, simple in manufacturing process and inexpensive.

[0007] The solution to this task is achieved through the features of the invention. Advantageous embodiments of the invention are described in the specification.

[0008] An antenna system 1 has two identical, mutually radiating and symmetrically arranged antennas 1a and 1b, each having antenna connection ports 2a and 2b, and a network 3 connected to the antenna connection ports 2a and 2b on the input side and composed of reactive components. This network has identical antenna paths 15a and 15b, individually assigned to the antennas 1a and 1b, leading to the network connection ports 4a and 4b. The system includes the following features:

[0009] - Each antenna connection port 2a, 2b consists of a connection point 22a, 22b that is energized relative to ground and a ground connection point 6, which is connected to the common ground 6 for all connection ports 2a, 2b, 4a, 4b;

[0010] - Each charged connection point 22a, 22b of antenna connection ports 2a, 2b is connected to the charged connection points 39a, 39b of the corresponding network connection ports 4a, 4b of antenna paths 5a, 5b via linear conductors 8a, 8b.

[0011] - At least along two linear electrical conductors 8a, 8b, there exists a grounding surface 5 designed to be flat at a grounding distance of 12;

[0012] - Two identical linear conductors 8a and 8b are guided parallel to each other not only with a conductor distance 11 but also with a ground distance 12 parallel to the ground surface 9, at least along a common length l 10, such that wires 13 symmetrical about ground 6 are formed through the two linear conductors 8a and 8b respectively, and antisymmetric wires 14 related to each other are formed between the two linear conductors 8a and 8b, and thus a coupling wire 15 is formed;

[0013] - The wave impedance ZLp and electrical angle αp present when the coupling wire 15 is excited in the same direction, and the wave impedance ZLn and electrical angle αn present when the coupling wire 15 is excited in the opposite direction, are designed such that the coupling S4a4b between network connection ports 4a and 4b at at least one frequency f1 is significantly less than the coupling S2a2b between the corresponding antenna connection ports 2a and 2b by using the network 3.

[0014] In order to form a match with the coupling wire, the antenna system 1 can detect the admittance YAp = GAp + jBAp at one of the antenna connection ports 2a and 2b, which occurs when the antenna connection port is excited with a measurement signal and when the other antenna connection port 2a and 2b is excited with the same co-polarized measurement signal.

[0015] Similarly, the admittance YAn = GAn + jBAn at one of these connection ports can be detected when the connection port is excited with a measurement signal and when the other antenna connection ports 2a and 2b are excited simultaneously with the same but reverse-polarized measurement signal.

[0016] In order to form a connection with antenna system 1, the electrical characteristics of the coupling wire 15, which is detached from antenna connection ports 2a and 2b at one end of the geometric length l 10, can be determined based on the following susceptance present at the other end of the wire 15.

[0017] On the one hand, it is detected and set when one end of the two conductors is short-circuited to ground 6, and these susceptances are:

[0018] • The susceptance Bkp at one of the network connection ports 4a and 4b when the network connection port is excited by a measurement signal and simultaneously excited by the other decoupled network connection port 4a and 4b with the same same polarized measurement signal,

[0019] • The susceptance Bkn at one of the network connection ports 4a and 4b when the network connection port is excited by a measurement signal and simultaneously excited by the other decoupled network connection port 4a or 4b with the same but reverse polarized measurement signal.

[0020] Furthermore, these electrical characteristics can be detected and set based on the following susceptance, which occurs when both conductors are open-circuited at one end of wire 15—and these susceptances are:

[0021] • The susceptance Blp at one of the network connection ports 4a and 4b under the condition of the same polarized measurement signal.

[0022] • The susceptance Bln at one of the network connection ports 4a and 4b when the network connection port is excited by a measurement signal and simultaneously excited by the other decoupled network connection port 4a or 4b by the same but reverse polarized measurement signal.

[0023] Therefore, the solid waveguide admittance YLp and its electrical angle αp of the symmetrical wire 13, and the solid waveguide admittance YLn and its electrical angles αp and αn of the antisymmetric wire 14 can be determined and set separately according to the following relationship.

[0024]

[0025] At the two network connection ports 4a and 4b, impedance matching and decoupling with the technically standardized real resistance Z0 = 1 / Y0 can exist at at least one frequency f1, in such a way that the structure and length l10 of the coupling wire 15 are designed such that not only the Y0-related line waveguide admittance yLp = YLp / Y0 and its electrical angle αp of the symmetrical wire 13, but also the corresponding Y0-related line waveguide admittance yLn = YLn / Y0 and its electrical angle αn of the antisymmetric wire 14 are formed according to the following relationships, which are specified by the Y0-related admittances YAp / Y0 and YAn / Y0 of the antenna system 1 at the antenna connection ports 2a and 2b, wherein applicable:

[0026]

[0027]

[0028] in .

[0029] Antenna system 1 can have real admittances GAp and GAn at at least one frequency f1 at these antenna connection ports 2a and 2b, such that YAp = GAp and YAn = GAn, and the decoupling of the two decoupling network connection ports 4a and 4b and the admittance adaptation of the two decoupling network connection ports 4a and 4b to the reference conductance Y0 are S. 4a4a =0、S 4b4b =0 is generated by the electrical angle of the symmetrical wire 13. and the electrical angle of the antisymmetric wire 14 The linear conductances of the symmetrical wire 13, with αp = αn = 90°, are selected as follows: Furthermore, it is argued that the linear wave conductance of wire 14 was chosen as... ,in The wavelengths of symmetric and antisymmetric waves are described and set to be the same size.

[0030] It is possible that, when one of the two decoupling connection ports 4a and 4b is at at least one frequency f1, the same specified admittance related to Y0 exists not only when excited in the same direction at the corresponding other decoupling network connection port 4a or 4b, but also when excited in the opposite direction at the corresponding other decoupling network connection port.

[0031] ,

[0032] The method is that, for the symmetrical wire 13, the linear waveguide admittance related to Y0 is... according to

[0033]

[0034] The electric angle, measured in degrees, is selected based on (3).

[0035]

[0036] To be chosen.

[0037] Furthermore, it is possible that for the antisymmetric wire 14, the linear waveguide admittance related to Y0 is based on...

[0038] The electric angle is selected based on (4).

[0039]

[0040] The two network connection ports 4a and 4b are selected, thereby achieving decoupling of the two network connection ports 4a and 4b at the end of the coupling wire 15.

[0041] It is possible that, in order to achieve impedance matching with the normalized admittance Y0 at the decoupling network connection ports 4a and 4b for at least one frequency f1, each of the two decoupled ends of the coupling wire, i.e., line connection ports 4a' and 4b', is embedded with the same output adapter networks 25a and 25b between them and the corresponding decoupling network connection ports 4a and 4b, for transforming the admittance present at the line connection ports 4a' and 4b' of the coupling wire 15 into the normalized admittance Y0.

[0042] It is possible that: Identical antenna adapter networks 24a, 24b are connected between each of the two antenna connection ports 4a, 4b and the corresponding conductors 8a, 8b of the coupling wire 15. Through these antenna adapter networks, the antenna admittances YAp or YAn present at connection ports 4a, 4b at at least one frequency f1 will be transformed into real admittances GAP or GAn, respectively. Furthermore, the coupling wire 15 is designed such that the linear admittance of the symmetrical wire 13 is...

[0043] And the electrical angle of the symmetrical wire is Furthermore, it opposes the claim that the waveguide admittance of wire 14 is... And the electrical angle of the antisymmetric wire is .

[0044] It is possible that, in order to achieve real admittances GAp and GAn, each antenna adapter network 24 consists of a parallel dipole with susceptance Bp from at least one reactive element connected in parallel with antenna connection ports 2a, 2b, and a series dipole with reactance Xs from at least one reactive element connected in series, wherein, in order to satisfy the requirement YAp = GAp or YAn = GAn, Bp / Y0 or Xs / Z0 has the following value at at least one frequency f1:

[0045] (5)

[0046] Applicable to:

[0047] .

[0048] It is possible that: in the first combination of the alternative combinations, in each antenna adapter network 24a, 24b, the parallel dipole with susceptance Bp is implemented as a capacitor with capacitance Cp and the serial dipole with reactance Xs is implemented as an inductor with inductance Ls; and it is possible that: in the second combination of the alternative combinations, the parallel dipole with susceptance Bp is implemented as an inductor with inductance Lp and the serial dipole with reactance Xs is implemented as a capacitor with capacitance Cs and inductance Ls.

[0049] It is possible that, in order to extend the frequency range around at least one frequency f1, where at least one frequency f1 is the center frequency in which not only the decoupling between network connection ports 4a and 4b is improved but also the impedance is adapted to the improved standard resistor Z0, the parallel dipole is implemented as a parallel resonant circuit and the serial dipole is implemented as a serial resonant circuit.

[0050] It is possible that the antenna adaptation network 24 is designed such that the frequency-dependent transformation of the antenna admittance ZAp = 1 / YAp or ZAn = 1 / YAn, which is transformed into real admittance GAp or GAn, at an increasing frequency within the complex impedance plane, at least partially around the real resistance value RAp = 1 / GAp or RAn = 1 / GAn, and the resistance value RAp or RAn is transformed into a reference resistance Z0 by the coupling wire 15, thereby improving not only decoupling but also impedance matching in a larger frequency range near frequency f1 at the decoupling network connection ports 4a, 4b.

[0051] It is possible that, in order to increase the frequency range near the at least one frequency f1 in the case of improved decoupling between decoupling network connection ports 4a, 4b, the parallel dipole and the serial dipole are supplemented according to the parameters described above for the parallel dipole with susceptance Bp and the serial dipole with reactance Xs, starting from the antenna connection networks 24a, 24b that are identical to each other, so as to form a parallel resonant circuit 7a in parallel and a serial resonant circuit 7b in series, respectively.

[0052] It is possible that, in order to increase the frequency range near at least one frequency f1 with improved coupling and improved impedance matching with the decoupling network connection ports 4a, 4b in the frequency range near the frequency f1, the output matching networks 24a, 25b respectively include at least one parallel resonant circuit 7a and at least one serial resonant circuit 7b, which are designed such that the impedances ZNp and ZNn at the decoupling network connection ports 4a, 4b, surround the normalized impedance Z0 at a small distance with increasing frequency in the complex impedance plane during their variation.

[0053] It is possible that the electrical characteristics of the coupling wire 15 are individually designed such that the effective distributed capacitance Cm` = dCm / dl relative to ground on the conductor length l 10 of each conductor 8a, 8b, the distributed capacitance Cb` = dCb / dl between each conductor, the distributed inductance Lp` = dLp / dl for the same-direction current in the two conductors, and the distributed inductance Ln` = dLn / dl for the opposite-direction current in the two conductors are set such that the required linear conductance of the symmetrical wire 13 is achieved for this frequency f1.

[0054] (6)

[0055] The electrical angle per unit length of the symmetrical wire as a phase constant.

[0056] (7)

[0057] Furthermore, the required linear conductance of the antisymmetric wire 14 was achieved.

[0058] (8)

[0059] and the electrical angle per unit length of the phase constant of the antisymmetric wire.

[0060] (9).

[0061] To design a coupled wire 15 having the required values ​​of YLp and YLn according to the invention and the required electrical angles αp and αn, it may be advantageous to first determine the wire structure, which provides the required values ​​for YLp and YLn and provides values ​​for the phase constants βp and βn independent of the line length l 10, such that the ratio βp / βn = αp / αn is satisfied and the geometric line length l 10 is selected according to l = αp / βp = αn / βn.

[0062] It is possible that the coupling wire 15 is implemented as a microstrip line with a conductor strip 17 on a dielectric substrate 27, and the conductor width 26 is correspondingly selected taking into account the dielectric constant of the substrate 27, to achieve the desired capacitance distribution Cm` and inductance distribution Lp` to form the line waveguide admittance YLp and phase constant βp, and

[0063] It is possible that: the conductor distance 11 is to form the capacitance distribution Cb` and—under the influence of the distribution of the coupling inductance M`—the inductance distribution Ln` is selected such that the required line waveguide admittance YLn and phase constant βn are set.

[0064] It is possible that each of the identical antennas 1a and 1b is formed as a monopole antenna 1a and 1b connected to ground 6 by their foot connection terminals 2a' and 2b', which is connected to its respective antenna connection ports 2a and 2b via correspondingly identical short electrically grounded asymmetrical wire segments 35 in order to bridge the antenna distance d.

[0065] It is possible that each of the identical antennas 1a and 1b is formed as a dipole antenna 1a' and 1b' designed symmetrically with respect to ground 6. The ground symmetrical connection ports 2a' and 2b' of the dipole antenna are connected to the assigned antenna connection ports 2a and 2b respectively via identical resymmetry devices 34. The resymmetry devices have short ground symmetrical input lines and ground asymmetrical output lines, which are used to bridge the antenna distance d and to resymmetric the ground symmetrical dipole connection ports 2a' and 2b' toward the ground asymmetrical antenna connection ports 2a and 2b.

[0066] The two linearly electromagnetically coupled conductors 8a and 8b of the decoupling network 3, which consist of a continuous infinitesimal line segment of symmetrically coupled wires 15 composed of infinitesimal reactive elements dLp = dLn, dCm, dCb, with length dl, can be replaced by a finite number N circuit stages S1, S2,... corresponding to the line segment but composed of discrete reactive elements. Instead of the infinitesimal reactive elements dLp, dLn, dCm, and dCb, discrete devices corresponding to these reactive elements, having a series inductance ΔLp = ΔLs 36, a bridge capacitance ΔCb 38, and a grounding capacitance ΔCm 37, can be used, and the decoupling of the decoupling network connection ports 4a and 4b can be set at a frequency f1 sufficiently lower than the threshold frequency ftp.

[0067] Reactive components 36, 37, and 38 can be replaced by dipole series reactive component circuits 43, 44, and 45, respectively, making them suitable for use at decoupling frequency f1.

[0068] The series reactive component circuits 43 and 43' used to replace inductors ΔLp and ΔLs 36 are inductive and have the same reactance as inductors ΔLp and ΔLs 36.

[0069] The bridge reactive power component circuits 45 and 45' used to replace the bridge capacitor ΔCb 38 are capacitive and have the same reactance value as the bridge capacitor ΔCb 38. The ground reactive power component circuits 44 and 44' used to replace the ground capacitor ΔCm 37 are capacitive and have the same reactance value as the ground capacitor ΔCm 37.

[0070] In the decoupling network 3, at least two stages S1, S2, etc., can be formed. Through the cooperation of the first stage S1 with the inductive series reactive element circuit 43 and the dipole bridge reactive element circuit 45, decoupling can also exist at the decoupling network connection ports 4a and 4b for frequency f1. This is achieved by having a series resonant circuit 46, which is permissible at frequency f1, in the series reactive element circuit 43' of the second stage S2, and a series parallel resonant circuit, which is blocked at frequency f1, in the bridge reactive element circuit 45'. At another frequency f2, decoupling at the decoupling network connection ports 4a and 4b can be achieved through the matching of the series reactive element circuit 43' and the bridge reactive element circuit 45' according to the present invention.

[0071] However, the decoupling network 3, designed as a low-pass structure, can be dually implemented as a high-pass structure. For this purpose, the inductor ΔLp = ΔLs 36 is replaced by the series capacitor ΔCs 42, the bridge capacitor ΔCb 38 is replaced by the bridge inductor ΔLb 41, and the ground capacitor ΔCm 37 is replaced by the ground inductor ΔLm 40. The values ​​of the series capacitor ΔCs 42, the ground capacitor ΔCm 37, and the ground inductor ΔLm 40 can be selected based on the high-pass structure elements obtained by dual transformation of the low-pass structure elements under the application of the duality formation rule, and the decoupling of the decoupling network connection ports 4a and 4b can be set if the frequency f1 is sufficiently higher than the threshold frequency fhp. Attached Figure Description

[0072] The invention will now be described in more detail with reference to embodiments. The accompanying drawings are shown in detail:

[0073] Figure 1 :

[0074] a) An example of antenna system 1, which has two identical, mutually radiating and symmetrically coupled antennas 1a and 1b arranged at a distance d, each antenna having antenna connection ports 2a and 2b according to the prior art. Each antenna is individually connected to transmitting-receiving devices 16a and 16b.

[0075] b) The four-pole or two-port equivalent circuit of antenna system 1, which has a complex antenna impedance. and complex coupling impedance It is used to detect the radiative coupling between antenna connection ports 2a and 2b.

[0076] c) Taking two parallel rod antennas with relative height h / λ=0.25 as an example according to a), the process of the change of the real part R12A and the imaginary part X12A of the coupling impedance Z12A as a function of the antenna distance d / λ0 related to the free space wavelength λ0.

[0077] Figure 2 :

[0078] a) Taking the antenna system in a) as an example, the real part R12A and the imaginary part X12A of the coupling impedance Z12A change as a function of a frequency of 750MHz at a relative antenna distance of d / λ = 0.2.

[0079] b) The divergence parameters of the transmission factor S2a2b between antenna connection ports 2a and 2b and the reflection factor S2a2a or S2b2b at antenna connection ports 2a or 2b, respectively, when the other port terminates at Z0.

[0080] Figure 3 :

[0081] a) Antenna system 1 having two identical radiating coupled antennas 1a, 1b according to the prior art, wherein each antenna has an antenna path 15a, 15b individually assigned to the transmitting-receiving device 16a, 16b.

[0082] b) A schematic diagram of antenna system 1 from a top view, the antenna system having a decoupling network 3 with coupling wires 15 for decoupling network connection ports 4a, 4b according to the present invention, and transmitting and receiving devices 16a, 16b connected thereto.

[0083] Figure 4 :

[0084] a) Antenna system 1 having two identical radiating coupled antennas 1a and 1b, each antenna having two linear conductors 8a and 8b of a formed coupling wire 15, serving as a decoupling network 3 according to the invention with decoupling network connection ports 4a and 4b. These two linear conductors 8a and 8b are guided along a common length l 10, parallel to each other not only with a conductor distance 11 but also parallel to the ground surface 9 with a ground distance 12. Through these two linear conductors 8a and 8b, not only a symmetrical wire 13 associated with ground 6 is formed but also an antisymmetric wire 14 associated with each other is formed between these two linear conductors 8a and 8b. The wave impedance and electrical angle αp or αn of the symmetrical and antisymmetric wires 13 and 14 in the coupling wire 15 are designed such that, at at least one frequency f1, the coupling S between the decoupling network connection ports 4a and 4b is... 4a4b The coupling S between the corresponding antenna connection ports 2a and 2b is significantly smaller due to the use of network 3. 2a2b .

[0085] b) Top view of antenna system 1 with decoupling network 3, as shown in a).

[0086] Figure 5 :

[0087] a) to b) Explanation of how to detect the characteristics of the antenna system by determining the following impedance based on the symmetrical impedance-equivalent circuit diagram (ZA) of antenna system 1 with respect to antenna connection ports 2a and 2b.

[0088] a) Detect, for example, the impedance ZAp = RAp + jXAp = 1 / YAp at antenna connection port 2b, which occurs when the antenna connection port is excited with the measurement signal U and when the other antenna connection port 2a is excited with the same co-polarized measurement signal U.

[0089] b) Detect the impedance ZAn=RAn + jXAn=1 / YAn at antenna connection port 2b, which appears when the antenna connection port is excited with the measurement signal U and when the other antenna connection port 2a is excited with the same but reverse polarized measurement signal -U.

[0090] c) to d) are descriptions of how the characteristics of the antenna system are detected by ascertaining the following admittances based on the admittance-equivalent circuit diagram (YA) of antenna system 1 with respect to antenna connection ports 2a and 2b.

[0091] c) Detect the admittance YAp=GAp + jBAp = 1 / ZAp at antenna connection port 2b, which occurs when the antenna connection port is excited with the measurement signal U and when the other antenna connection port 2a is excited with the same co-polarized measurement signal U.

[0092] d) Detect the admittance YAn=GAn + jBAn=1 / ZAn at antenna connection port 2b, which occurs when the antenna connection port is excited with the measurement signal U and when the other decoupling network connection port 2a is excited with the same but reverse polarized measurement signal -U.

[0093] Given the relationship YAp = 1 / ZAp and YAn = 1 / ZAn,

[0094] Similarly, the characteristics of the antenna system 1 with network 3 according to the present invention are detected by determining the impedances ZNp=RNp + jXNp =1 / YNp and ZNn=RNn + jXNn = 1 / YNn based on its symmetrical impedance-equivalent circuit diagram (ZN) about the mutually symmetrical decoupling network connection ports 4a, 4b, and by determining the admittances YNp=GNp + jBNp = 1 / ZNp and YNn=GNn + jBNn = 1 / ZNn based on its symmetrical admittance equivalent circuit diagram (YN).

[0095] Figures e) and f)

[0096] The electrical characteristics are detected based on the susceptance present at the other end of the coupling wire 15, which is disconnected from the antenna connection ports 2a and 2b at one end, as follows: the susceptance is expressed by the assumption that the symmetrical and antisymmetric wires 13 and 14 of the wire 15 are lossless real wave impedances ZLp = 1 / YLp and ZLn = 1 / YLn, and electrical angles αp and αn.

[0097] When one end of two conductors is short-circuited to ground 6, the following susceptance is obtained at the other end.

[0098] • The susceptance Bkp at one of the network connection ports 4a and 4b when the network connection port is excited by the measurement signal U and simultaneously excited by the other decoupled network connection port 4a and 4b with the same same polarized measurement signal U.

[0099] • The susceptance Bkn when the network connection port 4a or 4b is excited by the measurement signal and simultaneously excited by the same but reverse polarized measurement signal -U at the other decoupled network connection port 4a or 4b.

[0100] With two conductors open at one end to ground 6, the following susceptance is obtained at the other end accordingly.

[0101] • The susceptance Blp at one of the network connection ports 4a and 4b when the network connection port is excited by the measurement signal U and simultaneously excited by the other decoupled network connection port 4a and 4b with the same same polarized measurement signal U.

[0102] • The susceptance Bln when the network connection port 4a or 4b is excited by the measurement signal and simultaneously excited by the same but reverse polarized measurement signal -U at the other decoupled network connection port 4a or 4b.

[0103] Figure 6 :

[0104] The wave resistances ZLp and Zln of symmetrical and antisymmetrical wires 13 and 14, as well as electrical angles αp and αn, are formed by finite element method based on the representation of coupled wire 15. The finite element method is used to individually design the distributed capacitance Cm` = DCm / dl of each conductor 8a and 8b relative to each other over the conductor length l 10, the distributed capacitance Cb` = dCb / dl between each conductor 8a and 8b, and the distributed inductance Lp` = dLp / dl for currents flowing in the same direction in the two conductors and the distributed inductance Ln` = dLn / dl for currents flowing in opposite directions in the two conductors, which are affected by the distributed magnetic coupling M` = dM / dl of the two conductors.

[0105] Figure 7 :

[0106] A top view of the antenna system 1 with coupling wire 15 according to the invention is particularly notable in that: the symmetric admittance YAp and the antisymmetric admittance Yn present at the antenna connection ports 2a and 2b arranged symmetrically to each other are both real numbers at at least one frequency f1, such that:

[0107] Y1 = GAp and Yn = GAn.

[0108] The admittance adaptation of decoupling S4a4b and the two output connection ports 4a, 4b with the reference susceptance Y0, S4a4a=0, S4b4b=0, is established as follows: the electrical angle of the symmetrical wire 13 and the electrical angle of the antisymmetric wire 14 Given αp = αn = 90°, and the linear conductance of the symmetrical wire 13 being chosen as... Furthermore, it is argued that the linear wave conductance of wire 14 was chosen as... .

[0109] Figure 8 :

[0110] a) Frequency dependence of the divergence parameter S4a4b in dB around frequency f1 = 700MHz, as a basis Figure 4 The measure of decoupling between ports 4a and 4b of the decoupling network of the antenna system 1 with coupling wire 15. In the frequency range near deep decoupling at f1 = 700 MHz, coupling of -15 dB over a frequency bandwidth of 30 MHz and -20 dB over a frequency bandwidth of 16 MHz are obtained.

[0111] b) In comparison, the divergence parameters S2a2b show the radiative coupling of the two antennas at frequency f1 with a distance of d / λ0 = 0.2 between them at antenna connection ports 2a and 2b.

[0112] c) The frequency dependence of the divergence parameter S4a4b in dB around frequency f1 = 700MHz, as a measure of the reflection factor at the corresponding decoupling network connection port 4a or 4b when the other decoupling network connection port 4b or 4a terminates with standard conductance Y0.

[0113] In comparison, the divergence parameters S2a and S2b show the reflection factor at the corresponding antenna connection port 2a or 2b when the other decoupling network connection port 2b or 2a terminates with the standard conductance Y0.

[0114] Figure 9 :

[0115] a) A top view of the antenna system 1 with coupling wire 15, as shown below. Figure 7As shown, however, the difference is that the admittances YAp and YAn at antenna connection ports 2a and 2b are not real numbers at frequency f1, and the coupling wire 15 is implemented such that the linear wave admittance YLp and electrical angle αp of the symmetrical wire 13 and the linear wave admittance YLn and electrical angle αn of the antisymmetric wire 14 are designed according to the specified relationship at frequency f1 such that the symmetrical admittance and the antisymmetric admittance are the same at the two decoupling network connection ports 4a' and 4b', respectively, so that YNp = YNn applies and thereby the two decoupling network connection ports 4a' and 4b' are decoupled from each other;

[0116] b) As shown in Figure a), however, the difference is that: in order to achieve impedance matching with the normalized admittance Y0 at the decoupling network connection ports 4a, 4b for at least one frequency f1, an identical output adapter network 25 is embedded between each of the two decoupled ends of the coupling wire 15—line connection ports 4a', 4b'—and the corresponding decoupling network connection ports 4a, 4b, for transforming the admittance present at both ends of the coupling wire 15 into the normalized admittance Y0.

[0117] Figure 10 :

[0118] a) such as Figure 4 The antenna system 1 shown has a coupling wire 15, but differs in that: identical antenna adapter networks 24a, 24b are connected between each of the two antenna connection ports 2a, 2b and the corresponding conductors 8a, 8b of the coupling wire 15. Through these antenna adapter networks, the antenna admittances YAp or YAn present at the antenna connection ports 2a, 2b are transformed into real admittances GAP or GAn at at least one frequency f1, respectively. Furthermore, the coupling wire 15 is designed such that the linear admittance of the symmetrical wire 13 is...

[0119] And its electrical angle is Furthermore, it is opposed to claim that the waveguide admittance of wire 13 is... And its electrical angle is Furthermore, the decoupled network connection ports 4a and 4b are compatible with the standard admittance Y0;

[0120] b) A schematic top view of the arrangement in Figure a.

[0121] Figure 11 :

[0122] a) such as Figure 10The schematic top view shown in b, however, is notable in that each antenna adapter network 24a, 24b consists of a parallel dipole with a susceptance Bp from at least one reactive element connected in parallel to the antenna connection ports 2a, 2b, and a series dipole with a reactance Xs from at least one reactive element connected in series. The values ​​of the susceptance and the reactance, Bp and Xs, are selected according to functions of ZAn and ZAp, as described below. To form an improved admittance fit with the standard conductance Y0 over a wider bandwidth, at the line connection ports 4a', 4b', as... Figure 9 As shown in Figure b, output adapter networks 25a and 25b are connected downstream of coupling wire 15, respectively. The output terminals of these output adapter networks now form one of the decoupling network connection ports 4a and 4b, respectively.

[0123] b) As in a), however Bp is implemented as positive parallel conductance 28 and Xs is implemented as positive series reactance 29;

[0124] c) As in a), however, Bp is implemented as negative parallel conductance 30 and Xs is implemented as negative serial reactance 31.

[0125] Figure 12 :

[0126] One of the curves presented in the Cartesian reflectivity graph related to Z0 shows the change of the symmetrical antenna impedance ZAp at one of the antenna connection ports 2a and 2b when the other antenna connection port in 2a and 2b is excited in the same direction with an increasing frequency, and the impedance point Zp at frequency f1. Correspondingly, another curve shows the change of the antisymmetric antenna impedance ZAn at one of the antenna connection ports 2a and 2b, however, when the other antenna connection port in 2a and 2b is excited in the opposite direction with an increasing frequency, and the impedance point Zn at frequency f1.

[0127] Starting from impedance points Zp and Zn, each antenna connection network 24a, 24b utilizes... Figure 11 The values ​​of Bp—implemented via Lp—and Xs—implemented via Cs—described in section d for one of the two possible solutions, transform the symmetrical impedance Zp into a real resistance RAP and similarly transform the antisymmetric impedance Zn into a real resistance RAN. Then, according to Figure 10The designed coupling wire 15 utilizes its symmetrical wire 13, which has line waveguide admittance YLp and l / λp = ¼, to transform the real resistance RAp into the adaptation point Z0. Correspondingly, the antisymmetric wire 14, which has line waveguide admittance YLn and l / λn = ¼, transforms the real resistance RAn into the adaptation point Z0. Thus, at frequency f1, complete decoupling between the decoupling network connection ports 4a and 4b is achieved through the coupling wire 15.

[0128] Figure 13 :

[0129] a) Frequency dependence of the divergence parameter S4a4b in dB around frequency f1 = 700MHz, as a basis Figures 10 to 12 The measure of decoupling between ports 4a and 4b of the decoupling network of the antenna system 1 with coupling wire 15. In the frequency range near deep decoupling at f1 = 700MHz, coupling of <-15dB is obtained in a frequency bandwidth of 28 MHz and <-20dB in a frequency bandwidth of 15 MHz.

[0130] In comparison, the divergence parameters S2a and S2b show the radiative coupling of two antennas at a frequency f1 with a distance of d / λ0 = 0.2 between them.

[0131] b) As shown in Figure a), however, it has a higher frequency resolution.

[0132] Figure 14 :

[0133] a) Corresponding to Figure 13 a:

[0134] A graph of the frequency dependence of the divergence parameter S4a4b in dB around frequency f1 = 700MHz, serving as a measure of the reflection factor at the corresponding decoupling network connection port 4a or 4b when the corresponding other decoupling network connection port 4b or 4a terminates with standard conductance Y0 = 1 / Z0.

[0135] In comparison, the divergence parameters S2a and S2b show the reflection factor at the corresponding antenna connection port 2a or 2b when the corresponding other decoupling network connection port 2b or 2a terminates with the standard conductance Y0 = 1 / Z0.

[0136] b) As shown in Figure a), however, it has a higher frequency resolution.

[0137] Figure 15 :

[0138] a) In order to Figure 9In the device in b, two output adapter networks 25a and 25b are used at the decoupling network connection ports 4a and 4b within a larger bandwidth to form an improved admittance fit with the standard conductance Y0. Parallel resonant circuits 7a and series resonant circuits 7b are used in these output adapter networks 25a and 25b, respectively.

[0139] b) In order to increase the frequency range near at least one frequency f1 under the condition of improved decoupling between decoupling network connection ports 4a and 4b, starting from antenna connection networks 24a and 24b that are identical to each other, according to... Figure 11 The parameters of the two dipoles composed of reactive elements shown in Figure a are determined to supplement the two dipoles to form a parallel resonant circuit 7a and a series resonant circuit 7b, respectively.

[0140] Figure 16 :

[0141] a) In order to increase the frequency range near frequency f1 while improving the impedance matching of the connection ports 4a and 4b with the decoupling network, Figure 16 In the output adapter networks 24a and 25b of a, there are at least one parallel resonant circuit 7a and at least one serial resonant circuit 7b, respectively. The parallel resonant circuit and the serial resonant circuit are designed such that the impedances ZNp and ZNn at the decoupling network connection ports 4a and 4b, during their variation, surround the normalized impedance Z0 at a small distance with an increasing frequency in the complex impedance plane.

[0142] b) Typical variations of the symmetrical antenna impedance ZAp and the antisymmetric antenna impedance ZAn at increasing frequency f within the complex reflection factor plane relative to the normalized impedance Z0 used for impedance matching. Both variations partially revolve around the fitting point Z0. The distance between these variations is caused by the radiative coupling of the two antennas 1a and 1b.

[0143] c) In conjunction with antenna adapter networks 24a, 24a and coupling wire 15 according to the invention, antenna adapter networks 25a, 25b are designed such that the impedance changes of ZAp and ZAn in Figure b) are transformed such that these impedance changes are spaced apart from each other at an increasing frequency f with a small deviation from the impedance point Z0 used for impedance matching, thereby achieving sufficient decoupling between the two decoupling network connection ports 4a, 4b according to the invention.

[0144] Figure 17:

[0145] The exemplary frequency dependence of the divergence parameter near frequency f1 = 700 MHz serves as a measure of broadband decoupling between ports 4a and 4b of the decoupling network of the antenna system 1 with coupling wire 15. Improved decoupling is achieved over a wider frequency range in the frequency range near f1 = 700 MHz, without the need for deep decoupling.

[0146] a) To achieve an improved admittance fit with the standard conductance Y0 over a wider bandwidth, in this example, at line connection ports 4a' and 4b', as... Figure 9 As shown in b, output adapter networks 25a and 25b are connected downstream of the coupling wire 15, respectively, and the output ends of the output adapter networks now form one of the decoupling network connection ports 4a and 4b.

[0147] b) For the change in antenna impedance, in this example— Figure 10 As shown, identical antenna adapter networks 24a and 24b are connected between the two antenna connection ports 2a and 2b and the corresponding connection terminals on the coupling wire 15. Through these antenna adapter networks, the antenna admittances present at the antenna connection ports 2a and 2b are transformed into identical real admittances GAp or GAn.

[0148] In a) and b), the coupling or reflection factor is shown as a divergence parameter in dB related to Z0 = 1 / Y0 = 50 ohms, respectively.

[0149] They are shown separately:

[0150] Curve 1: The frequency change process of the coupling at antenna connection ports 2a and 2b based on the change process of the divergence parameter S2a2b in dB.

[0151] Curve 2: The frequency change of coupling between network connection ports 4a and 4b based on the change of divergence parameters S4a and 4b in dB.

[0152] Curve 3: The frequency variation of the reflection factor S2a2a = S2b2b in dB at one of the antenna connection ports 2a and 2b respectively.

[0153] Curve 4: The frequency variation of the reflection factor S4a4a = S4b4b in dB at one of the network connection ports 4a and 4b respectively.

[0154] Figure 18 :

[0155] a) The coupling wire 15 is implemented as a microstrip line with conductor strips 17 on a dielectric substrate 27. The conductor width 26 is selected accordingly, taking into account the dielectric constant of the substrate 27, to achieve the desired capacitance distribution Cm` and inductance distribution Lp` to form the line waveguide admittance YLp. Similarly, the conductor distance 11 is selected to form the distribution of capacitance distribution Cb` and the coupled inductance M` contained in the inductance distribution Ln`, such that the required line waveguide admittance YLp and phase constant βp, and YLp and phase constant βn are set. The characteristic parameters can be preferably and efficiently set by a simulation model of the coupling wire 15. For this purpose, the coupling wire is divided into two halves in its cross-section, as shown in Figure b), along its plane of symmetry, each half containing only one conductor 8a, 8b. This is achieved in the symmetrical case by terminating the plane of symmetry with a magnetic wall (positive) and in the antisymmetric case by terminating the plane of symmetry with an electric wall (negative).

[0156] b) The simulation model of the cross-section of the symmetrical microstrip line 13 in Figure a) is used to determine the wave impedance ZLp.

[0157] c) The simulation model of the cross-section of the antisymmetric microstrip line 14 in Figure a) is used to determine the wave resistance ZLn.

[0158] d) Coupled wire 15, which is implemented as a two-conductor line having a circular conductor, a conductor diameter of 23 and a conductor distance of 11 on the grounding surface at a grounding distance of 19.

[0159] e) As shown in Figure d), however, a two-wire conductor with a circular tubular shield.

[0160] Figure 19 :

[0161] a) According to the antenna system 1 of the present invention, it has monopole antennas 1a and 1b, each having the same electrically grounded asymmetrical line segment 35 for bridging antenna distance d.

[0162] b) According to the antenna system 1 of the present invention, it has dipole antennas 1a' and 1b', which have identical resymmetry devices 34, having short ground symmetry input lines and ground asymmetry output lines for bridging antenna distance d and for resymmetry of ground symmetry dipole connection ports 2a' and 2b' toward ground asymmetry antenna connection ports 2a and 2b.

[0163] Figure 20 :

[0164] It shows the relationship with Figure 6A similar equivalent circuit diagram is used for the two linear electromagnetically coupled conductors 8a and 8b of the decoupling network 3 according to the invention, which have concentrated electrical components. For this purpose, a continuous infinitesimal line segment of length dl, consisting of infinitesimal reactive elements dLp = dLn, dCm, dCb of symmetrically coupled wires 15, is replaced by a sufficient number N of continuous circuit stages S1, S2,... consisting of discrete reactive elements. Instead of the infinitesimal reactive elements dLp, dLn, dCm, and dCb, discrete components corresponding to these reactive elements are used, having a series inductance ΔLp = ΔLs 36, a bridge capacitance ΔCb 38, and a grounding capacitance ΔCm 37. The number of circuit stages S1, S2,... and the values ​​of the discrete components are selected such that decoupling of the decoupling network connection ports 4a and 4b is set at a frequency f1 sufficiently lower than the threshold frequency ftn. To adapt to the standardized wave conductance Y0 = 1 / Z0 of the decoupling network 3 at decoupling network connection ports 4a and 4b, the associated wave conductances YLp / Y0 and YLn / Y0 and the electrical angles αp and αn of the analog wire are required according to the complex antenna admittances YAp / Y0 and YAn / Y0 according to equation (2). Completely similar to a wire with distributed reactive components, the wave conductances YLp and YLn, and the electrical angles αp and αn are determined by the same-direction or opposite-direction short-circuit and open-circuit admittances Blp, Bkp, Bln, and Bkn according to equation (1).

[0165] The equivalent line surge resistance ZLp = 1 / YLp for co-directional excitation is...

[0166] (18a)

[0167] And for the electrical angle of the circuit, we obtain

[0168] (18b)

[0169] The equivalent line surge resistance ZLn = 1 / YLn for the reverse excitation is...

[0170] (18c)

[0171] And for the electrical angle of the circuit, we obtain

[0172] (19)

[0173] In order to achieve the same result as at a decoupling frequency of f1. Figure 20 The adaptation of the standardized resistor Z0=1 / Y0 at the connection ports 4a and 4b of the decoupling network requires setting the surge resistance ZLp and ZLn of the decoupling network 3, as well as setting the electrical angle of the decoupling network. and These parameters are obtained from the antenna admittances YAp and YAn according to equation (2).

[0174] For the current considerations, it is necessary to select a decoupling frequency that is sufficiently lower than the threshold frequency ftn of the circuit stage with a low-pass structure. The following applies to this decoupling frequency:

[0175] (20)

[0176] Therefore, the electrical angle of the circuit comes from equation (19). The minimum necessary level N is given by the following formula.

[0177]

[0178] Regarding electrical angle The larger the series N is chosen, the more accurately the electrical properties of the continuous wires from the linear conductors 8a and 8b—composed of their infinitesimally small components—can be realized using a simulated wire constructed from lumped components. Therefore, the scale should be chosen accordingly. .

[0179] In this way, Figure 7 , 9 The linear conductors 8a and 8b in 11, 15 and 16 can be completely and similarly replaced by analog coupled wires composed of centralized devices to achieve the same decoupling and adaptation to the decoupling network connection ports 4a and 4b.

[0180] Figure 21 :

[0181] As shown Figure 20 The decoupling network 3 according to the invention shown is, however, in all stages S1, S2..., the discrete inductors ΔLp, ΔLs 36 are replaced by discrete dipole series reactive element circuits 43, the discrete capacitors with bridge capacitance ΔCb 38 are replaced by discrete dipole bridge reactive element circuits 45, and the discrete capacitors with ground capacitance ΔCm 37 are replaced by discrete dipole ground reactive element circuits 44. For all reactive element circuits used to form decoupling at decoupling frequency f1, the replacing reactive element circuits 43, 44, 45 are designed such that they have sufficiently similar reactance values ​​to the discrete reactive elements to be replaced at decoupling frequency f1 and in their frequency environment.

[0182] Figure 22 :

[0183] As shown Figure 21 The decoupling network 3 according to the invention shown in the diagram uses only two distinct stages, S1 and S2, to achieve decoupling between network connection ports 4a and 4b at two frequencies f1 and f2. With the cooperation of the inductive series varistor circuit 43 and the dipole bridge varistor circuit 45 of the first stage S1, decoupling at network connection ports 4a and 4b exists at frequency f1, in such a way that the second stage S2 is designed to be permissible at the first frequency f1. This permissibility is achieved by including a series resonant circuit 46 in the series varistor circuit 43' of the second stage S2 that is permissible at frequency f1, and a series parallel resonant circuit 47 in the bridge varistor circuit 45' that is blocked at frequency f1, so as not to compromise the achieved decoupling. At another frequency f2, decoupling occurs at the decoupling network connection ports 4a and 4b, which are provided by the matching of the serial reactive element circuit 43' and the bridge reactive element circuit 45' according to the present invention. The downstream output adapter networks 25a and 25b do not affect decoupling and exist only for impedance matching. Similarly, the two antenna adapter networks 24a and 24b are only used to transform the antenna impedance at their outputs, which are also coupled due to the radiation coupling of the antennas.

[0184] Figure 23 :

[0185] exist Figure 22 The frequency variation process in dB of the decoupling network connection ports S4a4b of the decoupling network 3 for deep decoupling at frequencies f1 and f2.

[0186] Figure 24 :

[0187] like Figure 21 The antenna system 1 shown according to the invention, however, has only a decoupling network 3 with stage S1, which consists of two discrete dipole series reactive element circuits 43 that are inductive at frequency f1 and have reactance X1, a discrete dipole bridge reactive element circuit 45 that causes decoupling at frequency f1 and has admittance B1, and a discrete ground reactive element circuit 44, for impedance matching.

[0188] For frequency f1, using identical radiating coupled antennas 1a and 1b, the direction of radiation is determined based on whether the radiation is in the same or opposite direction.

[0189] The antenna admittances YAp = GAp + jBAp and YAn = GAn + jBAn determine the reactance X1. The reactance X1 required for decoupling at frequency f1 in terms of pure susceptance B1 is as follows. Therefore, two values ​​for X1 are possible:

[0190]

[0191] and (21)

[0192] .

[0193] Therefore, we choose either X1 = X11 or X1 = X12.

[0194] Regarding the value of X1 chosen accordingly, using

[0195] (twenty two)

[0196] Utilization value

[0197]

[0198] and

[0199]

[0200] Decoupling between decoupling network connection ports 4a and 4b is achieved independently of the discrete grounding reactive component circuit 44.

[0201] Figure 25 :

[0202] With respect to the present invention Figure 20 The circuit simulation with low-pass structure characteristics presented in the figure is completely dualized by the discrete components ΔLp = ΔLs, ΔCb, and ΔCm, forming the circuit simulation with high-pass structure presented in the current figure. It consists of stages S1, S2, ... connected in series with two series capacitors ΔCs, a bridge inductor ΔLb, and two ground inductors ΔLm, respectively.

[0203] Through Figure 20 The circuit in the middle undergoes a double transformation, which is obtained under the condition of unidirectional excitation. Figure 25 The wave impedance of the equivalent circuit in the diagram is

[0204] And in the case of reverse excitation, .

[0205] Correspondingly, the threshold frequency of the stage with high-pass structure regarding the reverse excitation is:

[0206] To achieve a sufficiently good fit at frequencies within the decoupling frequency f1 and at frequencies outside that range, the ratio f1 / fhn > 2 should be followed.

[0207] Figure 26 :

[0208] It shows Figure 20 The decoupling network 3 presented in the paper has a low-pass structure formed by concentrated reactive components. Figure 25 The diagram presents a combination of decoupling networks 3 with a high-pass structure formed by concentrated reactive components. Using this combined structure, decoupling network 3 is provided, achieving decoupling at low frequency f1 through its low-pass structure characteristics and at higher frequency f2 through its high-pass structure characteristics.

[0209] Figure 20 and 25 The individual serial elements 36 and 42 are replaced by parallel resonant circuits 36 and 42, respectively, in the combined structure. These parallel resonant circuits have a positive reactance value ΔLs of a low-pass structure at a low frequency f1 and a negative reactance value ΔCs of a high-pass structure at a higher frequency f2. Figure 20 and 25 In the combined structure, the individual grounding elements 37 and 40 are replaced by series resonant circuits 37 and 40, respectively. These series resonant circuits have a negative grounding reactance value ΔCm in a low-pass structure at low frequency f1 and correspondingly a positive grounding reactance value ΔLm in a high-pass structure at higher frequency f2. Similarly, Figure 20 and 25 In the combined structure, the individual bridge components 38 and 41 are replaced by series resonant circuits 38 and 41, respectively. These series resonant circuits have a negative bridge reactance value ΔCb at low frequency f1 and a positive reactance value ΔLb at higher frequency f2. Using this decoupling network 3, a similar system can be achieved. Figure 23 The frequency change process of decoupling in the process. Detailed Implementation

[0210] Other advantages of the invention are described in detail below:

[0211] A particular advantage of the antenna system 1 with network 3 according to the invention is that each of the two separate antennas 1a and 1b, which are radiatively coupled due to their small distance from each other, can be connected to the transmitting-receiving device via a coupling wire 15. This coupling wire avoids mutual interference caused by coupling in the transmitting-receiving devices 16a and 16b and can be designed separately at a particularly low cost. Furthermore, the coupling wire can simultaneously achieve decoupling and impedance matching with a standardized resistance Z0 = 1 / Y0. This resistance is mostly Z0 = 50 ohms in the present technology.

[0212] In particular, antennas 1a and 1b, together with the coupling wires, can be advantageously arranged as a decoupling network 3 on a common mechanical carrier. The direct mechanical connection between the antennas and the coupling wires 15 implemented on the circuit board serving as the mechanical carrier saves a significant amount of space, which is often crucial in motor vehicle manufacturing.

[0213] According to the present invention Figure 4 and 7 The device, which incorporates the coupling wire 15 as a decoupling network 3 according to the present invention, can be implemented independently and at very low cost, and can be flexibly adapted to the antenna system 1—depending on the parameters described. This achieves [the following]: Figure 8 Decoupling in a and 8b. Here, it is particularly advantageous to implement the coupling wire 15 as a microstrip line 26 on the circuit board, as exemplarily in… Figure 18 As presented in a.

[0214] With only a few electrical components, it is possible to give Figure 7 The device in the middle supplements antenna adapter networks 24a, 24b and output adapter networks 25a, 25b—such as Figure 16 As shown in a—and can generate according to a wider frequency range. Figure 17a , 17b The broadband and generally sufficient decoupling. The coupling wire 15 is established based on the parameters for the line impedance ZLp and Zln and for the electrical angles αp and αn mentioned above, which are obtained from the antenna admittances ZAp = 1 / Yap and ZAn = 1 / Yan according to the mathematical relationships described above. Suitablely, the geometry of the coupling wire 15 as the microstrip line 26 is transformed according to the simulation model. For this purpose, the simulation model can be suitably divided into two halves along its plane of symmetry in the cross section, each half containing only one wire. In the case of the symmetrical wire 13, this is accomplished by terminating the plane of symmetry with a magnetic wall as shown in Figure 18b, while in the case of the antisymmetric wire, this is accomplished by terminating the plane of symmetry with an electric wall, as shown in Figure 18b. Figure 18 As shown in c.

[0215] Then, half of the conductor width 11 and conductor distance 11 can be varied given the substrate height 19 and the specified dielectric material until the desired line impedance ZLp and phase constant βp for the left side of the cross-section and the desired line impedance ZLn and phase constant βn for the right side of the cross-section are achieved. Additionally, for example, the opposing edges of the conductors can be toothed together at a distance, thereby allowing ZLp and βp, ​​and ZLn and βn, to be better set individually. For example, specialized, high-performance digital computation programs based on the finite element method are used for this simulation, which can specifically determine the printed conductor structure suitable for having the required coupled wires 15 in microstrip line technology.

[0216] The coupling wire 15 is presented as a microstrip line 26 and Figure 18 The comparison of the coupling wire 15 in d and 18e illustrates the advantageous simplicity and flexible designability of the microstrip line 26 compared to being implemented as a double conductor on a conductive plane or as a double conductor with circular shielding.

[0217] In the following text, Figure 5 The symmetrical impedances (Z..) or admittances (Y..) present at port 2b, and therefore also at antenna connection port 2a, in 5d—the impedances or admittances corresponding to each other—are defined as follows:

[0218] Figure a) U / I = ZAp = Z11A+Z12A; Applicable to Z0: ;

[0219] Figure b) U / I = ZAn = Z11A -Z12A; Applicable to Z0: ;

[0220] Figure c) I / U = YAp = Y11A + Y12A; Applicable to Y0: ;

[0221] Figure d) I / U = YAn = Y11A -Y12A; Applicable to Y0: (10)

[0222] The output ports of antenna adapter networks 24a and 24b are denoted by 2a' and 2b'. Correspondingly, the above relationship applies to impedances or admittances marked with an apostrophe ` to describe the two-port or four-pole relationship between ports 2a' and 2b'.

[0223] and Figure 5Figures a through 5d correspond completely, and the electrical characteristic parameters are excited as a quadrupole with impedance or admittance at the output of network 3 between antenna connection ports 4a and 4b. Similar to these figures, the following applies between the outputs of network 3:

[0224] Figure a) U / I = ZNp = Z11N+Z12N; Applicable to Z0:

[0225] Figure b) U / I = ZNn = Z11N -Z12N; Applicable to Z0:

[0226] Figure c) I / U = YNp = Y11N + Y12N; Applicable to Y0:

[0227] Figure d) I / U = YNn = Y11N -Y12N; Applicable to Y0: (11)

[0228] Based on the constraints made here regarding the impedance or admittance at the connection ports, the impedance matrices related to Z0 or the admittance matrices related to Y0 for antenna connection ports 2a and 2b are as follows:

[0229] (12)

[0230] Correspondingly, the relevant impedance matrix zNM or admittance matrix yNM at the decoupling network connection ports 4a and 4b are as follows:

[0231] (13)

[0232] Applicable to: and

[0233] Based on symmetrical and asymmetrical impedance or admittance, the coupling between the two antenna connection ports 2a and 2b, or between the two decoupling network connection ports 4a and 4b, can be expressed as the divergence parameters S2a2b or S4a4b when each of these ports is terminated by a standard resistor Z0 = 1 / Y0, according to the following relationship:

[0234] in (14)

[0235] in

[0236] When the difference between the symmetrical impedance or admittance ending in "p" and the asymmetrical impedance or admittance ending in "n" disappears at that frequency, that is, when yNn = yNp is equivalent to zNp = zNn, complete, that is, deep decoupling is achieved between the decoupling network connection ports 4a and 4b at frequency f1.

[0237] Consider the reflection factor as a measure of impedance matching with a standard resistor Z0 at the connection port, which occurs when another connection port terminates with a standard resistor Z0.

[0238] Therefore, this applies to antenna connection ports 2a and 2b:

[0239] in (15)

[0240] And it is correspondingly applicable to decoupling network connection ports 4a and 4b:

[0241] in (16)

[0242] This demonstrates that when the impedance or admittance characterized by symmetry ending in "p" is the same as the impedance or admittance characterized by asymmetry ending in "n" at that frequency, that is, when yNn = yNp is equivalent to zNp = zNn being satisfied, the reflection at the decoupling network connection ports 4a and 4b... disappear.

[0243] Therefore, according to the present invention, the present invention is in Figure 4 a, 4b and in Figure 7 The particularly economically advantageous basic form of the coupling wire 15 not only achieves deep decoupling at frequency f1—that is, complete decoupling between network connection ports 4a and 4b—but also transforms the impedance present at antenna connection ports 2a and 2b, such that when the other connection port terminates with Z0 respectively, there is an impedance matching with Z0 = 1 / Y0 at each of the decoupling network connection ports 4a and 4b. According to the invention, this is achieved by the geometry and implementation of the coupling wire 15 being...

[0244] The design is such that not only the Y0-related linear waveguide admittance yLp = YLp / Y0 and its electrical angle αp of the symmetrical wire 13, but also the corresponding Y0-related linear waveguide admittance yLn = YLn / Y0 and its electrical angle αn of the antisymmetric wire 14 are formed according to the following relationships, which are specified by the Y0-related admittances YAp / Y0 and YAn / Y0 of the antenna system 1, wherein the following requirements apply:

[0245] ;

[0246] ;

[0247] And the following provisions apply:

[0248] and

[0249] in: .

[0250] In order to design a coupling wire 15 having the values ​​required by the present invention for YLp and YLn and the required electrical angles αp and αn, it is advantageous to first determine the electrical structure of the coupling wire 15 independently of the line length l 10, which provides values ​​for YLp and YLn and for the phase constants βp and βn such that the ratio βp / βn = αp / αn is satisfied, and the geometric line length l 10 is selected according to l = αp / βp being equivalent to αn / βn for the frequency f1.

[0251] As a guide for achieving this goal, , and The design should be based on the distributed capacitance and inductance as described above, according to the relationships explained above:

[0252] ; , (17)

[0253] Decoupling and reflection attenuation at antenna connection ports 4a and 4b Figure 8 As shown in 8a and 8b, and discussed above in conjunction with these figures.

[0254] In cases where determining the parameters of the coupling wire 15 is difficult or results in inconvenient dimensions due to the special symmetrical and antisymmetrical antenna impedances ZAp and ZAn of the antenna system 1 at frequency f1, the present invention... Figure 9In an advantageous implementation of a, the antenna impedances ZAp and ZAn are transformed at their ends by a coupling wire 15 into two identical admittances YNp = YNn, thereby enabling... Figure 9 In section a, the decoupling of two network connection ports 4a and 4b is implemented at the end of coupling 15.

[0255] To achieve the same specified admittance related to Y0 At frequency f1, for the symmetrical wire 13, the line waveguide admittance related to Y0 is... According to

[0256]

[0257] The selected electrical angle, measured in degrees, should be based on...

[0258]

[0259] To be selected, and for antisymmetric wire 14, the linear waveguide admittance YLn associated with Y0 should be based on

[0260]

[0261] To be set and the electrical angle should be according to

[0262]

[0263] It is set.

[0264] To perform impedance matching with the standardized admittance Y0 at decoupling network connection ports 4a and 4b for frequency f1, in Figure 9 In b, at each of the two decoupled ends of the coupling wire 15—line connection ports 4a', 4b'—and the corresponding decoupling network connection ports 4a, 4b, the same output adapter networks 25a, 25b are embedded to transform the admittances present at the two ends of the coupling wire 15 into normalized admittances Y0.

[0265] For antenna system 1, when these antenna connection ports 2a and 2b are at frequency f1, it has the following characteristics: Figure 7 In the special case shown where the real admittance YAp = GAp and YAn = GAn, the decoupling S of the two decoupling network connection terminals 4a and 4b... 4a4b The admittance adaptation (S) of the two decoupling network connection ports with the reference conductance Y0 4a4a =0、S 4b4b =0) is generated in the manner of the electrical angle of the symmetrical wire 13. and the electrical angle of the antisymmetric wire 14 The linear conductance of the symmetrical wire 13, with αp = αn = 90°, is selected as follows: Furthermore, it is argued that the linear wave conductance of wire 14 was chosen as... .

[0266] use To describe the wavelength of symmetrical and antisymmetrical waves.

[0267] Normally, there is no such thing as Figure 7 The special case shown is YAp = GAp and YAn = GAn. In this case, according to the invention... Figure 10 The specification stipulates that identical antenna adapter networks 24a and 24b, each with an output port 2a' or 2b', are connected between each of the two antenna connection ports 4a and 4b and the corresponding conductors 8a and 8b of the coupling wire 15. Through these antenna adapter networks, the antenna admittances YAp or YAn present at the antenna connection ports 2a and 2b are transformed into real admittances GAP or GAn, respectively, at the frequency f1. The coupling wire 15, as described above, is designed such that the linear wave admittance of the symmetrical wire 13 is...

[0268] And its electrical angle is Furthermore, the opposing statement is that the linear waveguide admittance of wire 14 is And its electrical angle is .

[0269] To achieve real admittances GAP and GAn, each antenna adapter network 24a, 24b, such as Figure 11 As shown in b and 11c, it consists of parallel dipoles 28 and 30 connected in parallel with antenna connection ports 2a and 2b, each having a susceptance Bp from at least one reactive element, and series dipoles 29 and 31 connected in series, each having a reactance Xs from at least one reactive element. To satisfy the requirement YAp = GAP or YAn = GAn, Bp / Y0 or Xs / Z0 has the following values ​​at this frequency f1, based on the following relationship between the two... Figure 11 b or Figure 11 Regarding the solution presented in C:

[0270]

[0271] and

[0272]

[0273] in

[0274]

[0275] as well as .

[0276] The choice of this combination of parallel dipole Bp and serial dipole Xs is determined based on one of the parameters specified for frequency f1. Figure 12 The complex reflection factor in the plane causes the impedance point Zp to be transformed into a real resistance RAP, and similarly, the impedance point Zn is transformed into a real resistance RAN. This allows for complete decoupling between the decoupling network connection ports 4a and 4b at the frequency f1 through a quarter-wave transformation of the symmetrical wire 13 or the anti-symmetrical wire 14 of the coupling wire 15. As a measure of decoupling between the decoupling network connection ports 4a and 4b... Figure 13 a and 13b show that the above has been combined Figure 14 The frequency dependence of the divergence parameters S4a4b, described in dB, is compared to the divergence parameters S2a2b of antenna system 1. Similar to this, Figure 14 a and 14b show the divergence parameters S4a4a = S4b4b in dB, as the reflection factor at one of the decoupling network connection ports 4a and 4b, or at one of the antenna connection ports 2a and 2b—at a frequency near f1 = 700 MHz—with the other connection port terminating at Z0 = 1 / Y0.

[0277] In further developments of the invention, it is generally advantageous to omit deep decoupling at frequency f1 to facilitate wider, but sufficient, decoupling of connection ports 4a and 4b at that frequency and to induce greater bandwidth by: Figure 12 Neutralization Figure 16 The frequency changes of the antenna impedances ZAp and ZAn in b are very close to each other after being transformed by the antenna adapter networks 24a and 24b and subsequently by the coupling wire 15. According to the invention, this is achieved by starting from the implementation of the antenna connection networks 24a and 24b, and according to... Figure 11 The parameters describing the two dipoles composed of reactive elements in 11b are determined to complement them, so as to form at least one parallel resonant circuit 7a and at least one series resonant circuit 7b, respectively. Figure 16 As shown in a.

[0278] To achieve sufficient impedance matching at connection ports 4a and 4b for complete decoupling, according to the present invention, the output adapter networks 25a and 25b are designed such that the frequency changes of the impedances ZNp and ZNn at the decoupling network connection ports 4a and 4b occur within the complex impedance plane near the center frequency fm, at an increasing frequency between the lower frequency fu and the upper frequency fo, around the standardized impedance Z0 at a small distance. Figure 16 As shown in c. This output is adapted to... Figure 16 The output adapter networks 25a and 25b in a are implemented using at least one parallel resonant circuit 7a and at least one serial resonant circuit 7b, respectively. The broadband decoupling achieved thereby is exemplarily demonstrated in… Figure 17a The frequency variation of the divergence parameters S4a4b near the center frequency of 750 MHz is presented. The improvement achieved by the broadband decoupling by the decoupling network 3 according to the invention is derived from the distance from the curve S4a4b of the divergence parameters S2a2b presented for comparison between the antenna connection ports 2a and 2b. Figure 10 The device shown performs deep decoupling in Figure 17b The frequency variation process presented in the figure demonstrates the advantage of broadband decoupling, which is sufficient in many applications to avoid the mutual interference discussed at the beginning in the transmitting-receiving devices 16a, 16b.

[0279] exist Figure 1-18 The example shown suggests applying the teachings of the invention to identical rod-shaped monopoles 1a, 1b. However, the invention is by no means limited to this antenna configuration. Rather, the invention applies to all identical antenna pairs (antenna double peaks) with mutual radiative coupling. In this case, only the radiative coupling according to the divergence parameter S2a2b between the antenna connection ports 2a, 2b is important, which is independent of the radiative coupling related to the direction of the antenna double peaks having an antenna in the far field—and therefore independent of the radiation pattern of the antenna double peaks. Therefore, the invention is not limited to paired rod-shaped monopoles, but also examines all paired ground-asymmetric monopole structures whose monopole connection ports 2a', 2b' are given with one pole through ground 6. This is similar to paired antennas with ground-symmetric dipole structures, which respectively have ground-symmetric connection ports 2a', 2b'. To maintain the symmetry of the device, a re-symmetric device 34 is connected downstream in the transition of the assigned ground-asymmetric connection ports 2a, 2b. Therefore, the present invention is not limited to paired rod dipoles, but also examines all paired grounded symmetrical dipole structures, whose dipole connection ports 2a' and 2b' are designed through grounded symmetrical connection terminals.

[0280] Figure 19 A diagram illustrates an apparatus in which each of the identical antennas 1a and 1b is configured as a monopole antenna 1a and 1b connected to ground 6 via its foot connection ports 2a' and 2b'. To bridge the antenna distance d via their respective antenna connection ports 2a and 2b, these ports are connected via separate, identical short grounding asymmetric line segments 35.

[0281] Unlike this, in Figure 19 In b, each of the identical antennas 1a and 1b is formed as a dipole antenna 1a' and 1b' designed to be symmetrical with respect to ground 6. Each ground symmetrical connection port 2a' and 2b' is connected to the assigned antenna connection port 2a and 2b via an identical re-symmetry device 34 having a short ground symmetrical input line and a ground asymmetrical output line, for bridging the antenna distance d and for re-symmetry of the ground symmetrical dipole connection ports 2a' and 2b' toward the ground asymmetrical antenna connection ports 2a' and 2b.

[0282] List of reference numerals

[0283] Antenna System 1

[0284] Radiating Coupled Antennas 1a and 1b

[0285] Antenna connection ports 2a, 2b

[0286] Antenna - Adapter network connection ports 2a', 2b'

[0287] Network 3

[0288] Network connection ports 4a and 4b

[0289] Line connection ports 4a' and 4b'

[0290] Antenna paths 5a, 5b

[0291] Common Land 6

[0292] Parallel resonant circuits, series resonant circuits 7a and 7b

[0293] Linear conductors 8a and 8b

[0294] Grounding connection terminal 9

[0295] Common length l 10

[0296] Conductor distance 11

[0297] Grounding distance 12

[0298] Symmetrical (same-direction excitation) wire 13

[0299] Antisymmetric (reverse excitation) wire 14

[0300] 15 coupling wires

[0301] Transmitter-receiver device 16

[0302] With conductor 17

[0303] Conductor distance 18

[0304] Grounding distance 19

[0305] Grounding surface 20

[0306] Coupled inductor M21,

[0307] live connection points 22a, 22b

[0308] Conductor diameter 23

[0309] Antenna adapter networks 24a, 24b

[0310] Output adapter networks 25a and 25b

[0311] Conductor width 26

[0312] Dielectric substrate 27

[0313] Positive parallel susceptance 28

[0314] Positive series susceptance 29

[0315] negative parallel susceptance 30

[0316] Negative series susceptance 31

[0317] Microstrip line 32

[0318] Circular conductor 33

[0319] Resymmetric device 34

[0320] Grounding asymmetry line segment 35

[0321] Discrete inductor devices with inductances ΔLp and ΔLs 36

[0322] Discrete capacitor unit 37 with grounding capacitance ΔCm

[0323] 38 discrete capacitors with bridge capacitance ΔCb

[0324] live connection points 39a and 39b

[0325] Discrete inductor with grounding inductance ΔLm 40

[0326] Discrete inductor with bridge inductance ΔLb 41

[0327] Discrete capacitor 42 with series capacitance ΔCs

[0328] Discrete dipole series reactive power component circuit; in the case of f1, it is inductive 43.

[0329] Discrete dipole grounded reactive power component circuit; in cases f1 and f2, it is capacitive 44, 44`

[0330] Discrete dipole bridge reactive power component circuit; capacitive 45° under f1 and f2 conditions.

[0331] 46 Isolated Series Resonant Circuit

[0332] Isolated parallel resonant circuit 47

[0333] Antenna admittance Y

[0334]

[0335] The standardized real reference resistance Z0 = 1 / Y0

[0336] Antenna distance d

[0337] Coupling: Antenna connection port, divergence parameter S2a2b_dB

[0338] Coupling: Decoupling network connection ports, divergence parameter S4a4b_dB

[0339] Reflection factor: Antenna connection port, divergence parameters S2a2a_dB, S2b2b_dB

[0340] Reflection factors: Decoupled network connection ports, divergence parameters S4a4a_dB, S4b4b_dB

[0341] With the ground's susceptance Bp

[0342] Infinitely small series inductances dLp, dLn

[0343] Infinitely small grounding capacitance dCm

[0344] Infinitely small bridge capacitance dCb

[0345] The distributed capacitance relative to ground is Cm` = dCm / dl

[0346] The distributed capacitance between conductors is Cb` = dCb / dl

[0347] Symmetrical linear wave conductance

[0348] Anti-symmetric linear conductance

[0349] Phase constant of symmetrical line

[0350] Antisymmetric line phase constant

[0351] Free space wavelength

[0352] Symmetrical wavelength

[0353] antisymmetric wavelength

[0354] electric angle

[0355] electric angle

Claims

1. An antenna system (1) having two identical, mutually radiatingly coupled, symmetrically arranged antennas (1a, 1b), each antenna (1a, 1b) having an antenna connection port (2a, 2b), and a network (3) connected to the antenna connection ports (2a, 2b) on the input side and composed of reactive elements, the network (3) having identical antenna paths (5a, 5b) to the network connection ports (4a, 4b) for each of the antennas (1a, 1b), the antenna system (1) comprising the following features: - Each antenna connection port (2a, 2b) consists of a connection point (22a, 22b) that is charged relative to ground and a ground connection point, the ground connection point being connected to a common ground (6) for all the antenna connection ports (2a, 2b) and the network connection ports (4a, 4b); - Each charged connection point (22a, 22b) of the antenna connection ports (2a, 2b) is connected to the charged connection point (39a, 39b) of the corresponding network connection port (4a, 4b) of the antenna path (5a, 5b) via a linear conductor (8a, 8b). - A flat grounding surface (5) exists at least along two linear electrical conductors (8a, 8b) at a grounding distance (12); - Two identical linear conductors (8a, 8b) are guided parallel to each other not only with a conductor distance (11) but also with a ground distance (12) parallel to the ground surface (9) at least along a common length l (10), such that wires (13) symmetrical about ground (6) are formed through the two linear conductors (8a, 8b) respectively, and antisymmetric wires (14) related to each other are formed between the two linear conductors (8a, 8b) and thus a coupling wire (15) is formed. - The wave impedance ZLp and electrical angle αp present when the coupling wire (15) is excited in the same direction, and the wave impedance ZLn and electrical angle αn present when the coupling wire (15) is excited in the opposite direction, are designed such that the coupling (S4a4b) between the network connection ports (4a, 4b) at at least one frequency f1 is significantly less than the coupling (S2a2b) between the corresponding antenna connection ports (2a, 2b) by using the network (3).

2. The antenna system (1) according to claim 1. Its features are, In order to form a connection with the coupling wire, the antenna system (1) detects... - The admittance YAp = GAp + jBAp at one of these antenna connection ports (2a, 2b), which appears when that antenna connection port is excited with a measurement signal and simultaneously when the other antenna connection port (2a, 2b) is excited with the same co-polarized measurement signal, and - The admittance YAn = GAn + jBAn at one of these antenna connection ports occurs when the connection port is excited with a measurement signal and when the other antenna connection port (2a, 2b) is excited simultaneously with the same but reverse-polarized measurement signal.

3. The antenna system (1) according to claim 2. Its features are, In order to form a fit in the antenna system (1), the electrical characteristics of the coupling wire (15) of geometric length l (10) that is detached from the antenna connection port (2a, 2b) at one end are based on the following susceptance present at the other end of the coupling wire (15), i.e. at the network connection port (4a, 4b). On the one hand, these susceptances are detected and set when one end of the two linear conductors (8a, 8b) is short-circuited to ground (6), and these susceptances are: - The susceptance Bkp at one of the network connection ports (4a, 4b) when the network connection port is excited by a measurement signal and simultaneously excited by the other network connection port (4a, 4b) with the same co-polarized measurement signal. - The susceptance Bkn at one of the network connection ports (4a, 4b) when the network connection port is excited by a measurement signal and simultaneously excited by the other network connection port (4a, 4b) with the same but reverse-polarized measurement signal. Furthermore, the electrical characteristics are detected and set based on the following susceptance, which is detected and set when one end of the two linear conductors (8a, 8b) is open-circuited in the coupling wire (15). These susceptances are: - The susceptance Blp at one of the network connection ports (4a, 4b) under the condition of the same polarized measurement signal. - The susceptance Bln at one of the network connection ports (4a, 4b) when the network connection port is excited by a measurement signal and simultaneously excited by the other network connection port (4a, 4b) with the same but reverse-polarized measurement signal. Furthermore, the solid waveguide admittance YLp of the co-directional excitation wire (13) and the electrical angle αp of the co-directional excitation wire (13), as well as the solid waveguide admittance YLn of the reverse excitation wire (14) and the electrical angle αn of the reverse excitation wire (14), are individually determined and set according to the following relationship. and ； 。 4. The antenna system (1) according to claim 3. Its features are, At the two network connection ports (4a, 4b), at at least one frequency f1, there exists an impedance matching with the technically standardized real resistance Z0 = 1 / Y0, in such a way that the structure and length l (10) of the coupling wire (15) are designed such that not only the Y0-related line waveguide admittance yLp = YLp / Y0 and the electrical angle αp of the symmetrical wire (13), but also the Y0-related line waveguide admittance yLn = YLn / Y0 and the electrical angle αn of the antisymmetric wire (14) are formed according to the following relationships, which are given by the Y0-related admittances YAp / Y0 and YAn / Y0 of the antenna system (1) at the antenna connection ports (2a, 2b), wherein applicable: and ; ； in: .

5. The antenna system (1) according to claim 4. Its features are, The antenna system (1) has real admittances GAp and GAn at the antenna connection ports (2a, 2b) at at least one frequency f1, such that YAp = GAp and YAn = GAn, and the decoupling of the two network connection ports (4a, 4b) and the admittance matching of the two network connection ports (4a, 4b) with reference conductance Y0 are generated by the electrical angle of the symmetrical wire (13). Electrical angle of antisymmetric wire (14) αp = αn = 90° respectively, and the linear conductance of the symmetrical wire (13) is selected as Furthermore, the line wave conductance of the antisymmetric wire (14) was selected as ,in The wavelengths of symmetric and antisymmetric waves are described and set to be the same size.

6. The antenna system (1) according to claim 1, Its features are, The coupling wire (15) is implemented as a microstrip line with a conductor strip (17) on a dielectric substrate (27), and the conductor width (26) is selected accordingly, taking into account the dielectric constant of the substrate (27), to achieve the desired capacitance distribution Cm` and inductance distribution Lp` to form line waveguide admittance YLp and phase constant βp, and the conductor distance (11) for forming capacitance distribution Cb` and the inductance distribution Ln` under the influence of the distribution of coupling inductance M` are selected such that the required line waveguide admittance YLn and phase constant βn are set.

7. The antenna system (1) according to claim 1. Its features are, Each of the identical antennas (1a, 1b) is configured as a monopole antenna (1a, 1b) connected to ground (6) via its foot connection terminals (2a', 2b'), the monopole antennas (1a, 1b) being connected to the associated antenna connection ports (2a, 2b) via the same, short, electrically grounded asymmetrical line segment (35) to bridge the antenna distance d.

8. The antenna system (1) according to claim 1. Its features are, Each of the identical antennas (1a, 1b) is formed as a dipole antenna (1a', 1b') designed symmetrically with respect to ground (6). The ground symmetrical connection ports (2a', 2b') of the dipole antennas (1a', 1b') are connected to the assigned antenna connection ports (2a, 2b) via the same resymmetry device (34), which has short ground symmetrical input lines and ground asymmetrical output lines for bridging the antenna distance d and for resymmetry of the ground symmetrical dipole connection ports (2a', 2b') toward the ground asymmetrical antenna connection ports (2a, 2b).

9. The antenna system (1) according to claim 1. Its features are, The two linearly electromagnetically coupled linear conductors (8a, 8b) of the network (3) consisting of infinitesimal reactive elements dLp = dLn, dCm, dCb of the symmetrically coupled wire (15) with a length dl are replaced by a finite number of circuit stages corresponding to the line segments, but consisting of discrete reactive elements, wherein the infinitesimal reactive elements dLp, dLn, dCm and dCb are replaced by discrete devices corresponding to these reactive elements with inductance ΔLp = ΔLs (36), bridge capacitance ΔCb (38) and grounding capacitance ΔCm (37), and the values ​​of these discrete devices are selected such that the decoupling of the network connection ports (4a, 4b) is set at a frequency f1 that is sufficiently lower than the threshold frequency ftp.

10. The antenna system (1) according to claim 9. Its features are, The discrete reactive power components (36, 37, 38) are replaced by dipole discrete reactive power component circuits (43, 44, 45), respectively, making them suitable for use at the decoupling frequency f1: The serial reactive element circuit (43) used in series in place of the inductors ΔLp and ΔLs (36) is inductive and has the same reactance value as the inductors ΔLp and ΔLs (36). The bridge reactive component circuit (45) used to replace the bridge capacitor ΔCb (38) is capacitive and has the same reactance value as the bridge capacitor ΔCb (38). The grounding reactive element circuit (44) used to replace the grounding capacitor ΔCm (37) is capacitive and has the same reactance value as the grounding capacitor ΔCm (37).

11. The antenna system (1) according to claim 8. Its features are, Two stages are formed in the network (3). With the cooperation of the inductive series reactive element circuit (43) and the dipole bridge reactive element circuit (45) in the first stage (S1) of the two stages, there is also decoupling at the network connection ports (4a, 4b) for the frequency f1. The way is that in the second stage (S2) of the two stages, there is a series resonant circuit (46) that can be passed for the frequency f1 in the series reactive element circuit (43'), and there is a series parallel resonant circuit (47) that blocks the frequency f1 in the bridge reactive element circuit (45'). Moreover, in the case of another frequency f2, there is decoupling at the network connection ports (4a, 4b) through the matching of the series reactive element circuit (43') and the bridge reactive element circuit (45').

12. The antenna system (1) according to claim 9. Its features are, The network (3) designed as a low-pass structure is also implemented as a high-pass structure, for this reason, The inductance ΔLp = ΔLs (36) is replaced by the series capacitance ΔCs (42). The bridge capacitor ΔCb(38) is replaced by the bridge inductance ΔLb(41), and The grounding capacitance ΔCm(37) is replaced by the grounding inductance ΔLm(40), and The values ​​of the serial capacitor ΔCs (42), the ground capacitor ΔCm (37), and the ground inductance ΔLm (40) are selected based on the high-pass structural element obtained by the dual transformation of the low-pass structural element under the application of the duality formation rule. Furthermore, the values ​​of these discrete components are selected such that the decoupling of the network connection ports (4a, 4b) is set at a frequency f1 that is sufficiently higher than the limiting frequency fhp.

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