A noise-resistant integrated satellite communication ranging and phase measurement method

Abstract

A noise-resistant integrated communication ranging and phase measurement method comprises the following steps: generating two squared signals from the reference and measurement electrical signals of the heterodyne interferometric communication ranging integrated optical system; mixing the two squared signals with sine and cosine signals generated by an FPGA to obtain four sine and cosine mixed signals; calculating the phase of the difference frequency signal between the two complex exponential signals generated by the mixed signals using a full-phase FFT algorithm; and subtracting the phases of the two difference frequency signals to complete the phase measurement. This invention solves the problems of excessive logic resource consumption and ineffective suppression of spatial noise in FPGAs. It can suppress the severe interference of communication signal modulation on phase measurement in displacement measurement while reducing the use of logic resources in the FPGA, achieving a 2×10⁻⁶ phase ratio at a 50dB signal-to-noise ratio. ‑ 4 Phase measurement accuracy in rad.

Description

Technical Field

[0001] This invention belongs to the field of communication technology, and more specifically relates to a noise-resistant integrated phase measurement method for satellite communication ranging within the field of satellite communication technology. This invention can be used for phase measurement when inter-satellite communication and displacement measurement are combined. Background Technology

[0002] In the integration of satellite communication and displacement measurement, the shared signal beam causes severe interference from communication signal modulation on phase measurement during displacement measurement. Furthermore, the influence of space noise further reduces phase measurement accuracy, making high-precision displacement measurement impossible. Current phase measurement methods in integrated communication and ranging systems have addressed the problem of severe interference from communication signal modulation on phase measurement during displacement measurement. However, they still have two shortcomings: first, they consume excessive logic resources in programmable gate arrays; second, the presence of significant space noise degrades phase measurement accuracy, and current phase measurement methods cannot effectively suppress this space noise.

[0003] Xi'an University of Electronic Science and Technology proposed an integrated satellite communication ranging and phase measurement method in its patent application "Integrated Communication and Ranging Target Displacement Measurement Method Based on Heterodyne Interferometry System" (application number: 202211032874.6, application date: 2022.08.26). This method mixes the reference and measurement electrical signals obtained from the integrated optical system of heterodyne interferometry with pre-set sine and cosine signals within a programmable gate array to obtain four mixed signals. These four mixed signals are then filtered by a low-pass filter to remove the sum-frequency term, resulting in four difference-frequency signals. These four sine and cosine signals are squared, subtracted, and then mixed again to obtain two sine and cosine signals. The phase can then be calculated using arctangent operations on these two signals. This method can suppress the interference of communication signal modulation on the phase measurement during displacement measurement. However, this method still has some shortcomings. First, when dealing with the interference of communication signal modulation on phase measurement during displacement measurement, the subtraction and mixing of the four sine and cosine signals after squaring them consume too many resources in the programmable gate array. Second, this method cannot effectively suppress the impact of noise on the phase measurement accuracy. Compared with the noise-free case, the phase measurement accuracy will drop by an order of magnitude when the signal-to-noise ratio is 50dB. The lower the signal-to-noise ratio, the more serious the accuracy drop. When the signal-to-noise ratio is too low, phase measurement may not even be possible. Summary of the Invention

[0004] The purpose of this invention is to address the shortcomings of the existing technologies by providing a noise-resistant integrated satellite communication ranging and phase measurement method, thereby solving the problems of excessive logic resource consumption in programmable gate arrays and the inability to effectively suppress noise in space.

[0005] The specific approach to achieving the objective of this invention is as follows: This invention squares the two electrical signals obtained from the reference and measurement ends of the heterodyne interferometric communication ranging integrated optical system, resulting in two squared signals. Because the communication uses binary phase shift keying (BPSK) modulation, the signal undergoes a 180-degree phase transition. The squaring operation eliminates the influence of this 180-degree phase transition, thus suppressing the interference of communication signal modulation on phase measurement during displacement measurement. The two squared signals are then compared with sine and cosine signals generated internally by a programmable gate array. The signals are mixed to obtain four sine and cosine mixed signals. Two sine mixed signals are used as the imaginary parts, and two cosine mixed signals are used as the real parts to form two complex exponential signals. The phase of the difference frequency signal in the two complex exponential signals is calculated by the full-phase FFT algorithm. Because the full-phase FFT algorithm transforms the signal from the time domain to the frequency domain, the amplitude of the useful signal is significantly higher than the amplitude of the noise in the frequency domain, so it can effectively distinguish between noise and useful signals, thereby effectively suppressing noise in the space. Finally, the phase of the two difference frequency signals is subtracted to complete the phase measurement.

[0006] The method for achieving the objective of this invention includes the following steps:

[0007] Step 1: The reference end electrical signal and the measurement end electrical signal output by the heterodyne interferometric communication ranging integrated optical system are squared inside the FPGA to obtain two squared signals.

[0008] Step 2: Generate two orthogonal signals using the FPGA, and then mix these two orthogonal signals with two squared signals to obtain four mixed signals sinA, cosA, sinB and cosB.

[0009] Step 3: Generate two complex exponential signals e A and e B ;

[0010] Step 4: Calculate the complex exponential signal e using the full-phase FFT algorithm. A and e B Instantaneous phase of the difference frequency signal in and

[0011] Step 5, instantaneous phase and The difference is calculated to obtain the phase containing the target displacement information.

[0012] Compared with the prior art, the present invention has the following advantages:

[0013] First, because this invention calculates the instantaneous phase of the difference frequency signal using a full-phase FFT algorithm, it overcomes the problem of ineffective noise suppression in existing technologies. This allows for the achievement of 2×10-1 signals at a signal-to-noise ratio of 50dB. -4 Phase measurement accuracy in rad.

[0014] Secondly, because this invention performs a squaring operation on the reference and measurement electrical signals output by the heterodyne interferometric communication ranging integrated optical system within a programmable gate array (PGA), it overcomes the excessive consumption of logic resources in the PGA in existing technologies. This allows the invention to suppress severe interference from communication signal modulation on phase measurement in displacement measurement while reducing the use of logic resources in the PGA. Attached Figure Description

[0015] Figure 1 This is a schematic diagram of the heterodyne interferometric communication ranging integrated system according to an embodiment of the present invention;

[0016] Figure 2 This is a flowchart of the method of the present invention;

[0017] Figure 3 This is a phase measurement accuracy diagram of the integrated communication ranging method and existing methods, wherein, Figure 3 (a) A phase measurement accuracy diagram for achieving integrated communication ranging using existing methods. Figure 3 (b) is a phase measurement accuracy diagram of the method of the present invention for achieving integrated communication ranging. Detailed Implementation

[0018] The steps for implementing the present invention will be further described below with reference to the accompanying drawings and embodiments.

[0019] Reference Figure 1 The heterodyne interferometric communication and ranging integrated optical system in the embodiments of the present invention will be further described.

[0020] The heterodyne interferometric communication and ranging integrated optical system in this embodiment of the invention includes a frequency-stabilized laser LASER1, a modulator Mondulator, a beam splitter BS, a reference polarizer PA1, a polarizing beam splitter PBS, and a reference corner mirror P. r Measuring the angle reflector P m The measurement includes a polarizer PA2, a photoelectric conversion device PT, and a programmable gate array (FPGA), among which:

[0021] The frequency-stabilized laser LASER1 is used to simultaneously generate orthogonally polarized light E1(t) and E2(t) with frequencies f1 and f2;

[0022] The modulator, located to the right of the frequency-stabilized laser LASER1, is used to modulate the baseband data d(t) using polarized light E1(t) and E2(t) as carriers via binary phase shift keying (BPSK).

[0023] The beam splitter prism BS, located to the right of the frequency-stabilized laser LASER1, splits the beam emitted from LASER1 into two beams with mutually perpendicular propagation directions, namely the reference beam E. r and the measuring end beam E m ;

[0024] Reference polarizer PA1, located directly below beam splitter BS, is used to change the reference beam E. r The polarization directions of two mutually orthogonal polarized lights are reversed to cause them to interfere.

[0025] The polarizing beam splitter PBS, located to the right of the beam splitter BS, is used to divert the measurement beam E. m It is divided into two polarized beams, E1(t) and E2(t), whose propagation directions are perpendicular to each other and whose polarization directions are perpendicular to each other;

[0026] Reference corner mirror P r It is located directly above the polarizing beam splitter PBS and is used to reflect the polarized beam E1(t) back to the polarizing beam splitter PBS.

[0027] Measuring angle reflector P m It is located to the right of the polarizing beam splitter PBS and maintains a horizontal motion state to reflect the polarized beam E2(t) back to the polarizing beam splitter PBS.

[0028] The measuring polarizer PA2, located to the right of the reference polarizer PA1, is used to change the polarization direction of the two beams reflected back to the polarizing beam splitter PBS, causing them to interfere.

[0029] The photoelectric conversion device PT, located below the reference polarizer PA1 and the measurement polarizer PA2, is used to convert the two beams after interference at the reference end and the measurement end into electrical signals.

[0030] A programmable gate array (FPGA) is connected to the output signal of a photoelectric conversion device (PT) to perform phase calculations on the electrical signals converted from photoelectric signals within the FPGA.

[0031] Reference Figure 2 The steps for implementing the present invention will be further described below.

[0032] Step 1: Generate a mixed beam E.

[0033] Step 1.1: Simultaneously generate orthogonally polarized light E1(t) and E2(t) with frequencies f1 and f2 = f1 + 10 MHz respectively using the frequency-stabilized laser LASER1:

[0034] E1(t)=cos(2πf1t)

[0035] E2(t)=cos(2πf2t)

[0036] Where π represents the mathematical constant pi, and t represents the time in the time domain of the electrical signal output by the heterodyne interferometric communication ranging integrated optical system;

[0037] Step 1.2: Using a modulator, orthogonally polarized beams E1(t) and E2(t) are used as carriers to modulate the baseband data d(t) via binary phase shift keying (BPSK), resulting in modulated beams E1′(t) and E2′(t).

[0038] E1′(t)=D r (t)cos(2πf1t)

[0039] E2′(t)=D r (t)cos(2πf2t)

[0040] Among them, D r (t) represents the baseband data transmitted by the heterodyne interferometric communication ranging integrated optical system.

[0041] In step 1.3, beams E1′(t) and E2′(t) will mix to form a mixed beam E because they have the same transmission direction.

[0042] Step 2, generate reference terminal electrical signal I r .

[0043] Step 2.1: The path of beam E transmitted through the beam splitter BS is divided into a measurement end and a reference end, that is, the mixed beam E that continues to be transmitted forward through the beam splitter BS is split into a measurement end and a reference end. r As a reference end, the mixed beam E, which is reflected by the beam splitter BS and then transmitted vertically downwards, is used as a reference. m As a measuring end;

[0044] Step 2.2, mix the reference end beam E r A reference-end interference beam is obtained by directly interfering through the reference polarizer PA1: E R (t)=D r (t)cos(2πf1t-2πf2t);

[0045] Step 2.3, the obtained interference beam E R(t) is then converted into a reference terminal electrical signal I by a photoelectric conversion device PT. r , means as follows:

[0046] I r =D r (t)cos(w r t)

[0047] Among them, w r =2πf1-2πf2 is the reference terminal electrical signal I r angular frequency.

[0048] Step 3, generate the measurement terminal electrical signal I m .

[0049] Step 3.1, mix the beam E at the measurement end. m The beams are directly incident on the polarizing beam splitter PBS, which separates the orthogonal beams E1(t) and E2′(t) with frequencies of f1 and f2, respectively.

[0050] Step 3.2: Transmit the beam E1(t) with frequency f1 to the stationary reference corner mirror P. r It is then reflected back to the polarizing beam splitter PBS;

[0051] Step 3.3: Transmit the beam E2′(t) with frequency f2 to the movable measuring angle mirror P. m The resulting beam E3(t) has a frequency of f2±Δf:

[0052] E3(t)=D m (t)cos(2π(f2±Δf)t)

[0053] in, The frequency difference caused by the Doppler frequency shift is v, and the value of the measuring corner mirror P is v. m The moving speed, λ is the wavelength of the laser emitted by the frequency-stabilized laser, and the two frequency-stabilized lasers emit lasers with the same wavelength, D m (t) represents the baseband data at the receiver of the heterodyne interferometric communication ranging integrated optical system.

[0054] Step 3.4: Reflect beam E3(t) back to the polarizing beam splitter PBS, and then interfere with the beam E1(t) returned to the polarizing beam splitter PBS in step 3.2 through the measuring polarizer PA2 to obtain a measuring end interference beam: E M (t)=D m (t)cos(2πf1t-2π(f2±Δf)t);

[0055] Step 3.5, the interference beam E at the measurement end M(t) The electrical signal I at the measurement end is obtained by conversion through a photoelectric conversion device PT. m :

[0056] I m =D m (t)cos(w m t)

[0057] Among them, w m =2πf1-2π(f2±Δf) is the electrical signal I at the measurement end. m angular frequency.

[0058] Step 4: Obtain the four mixing signals sinA, cosA, sinB, and cosB.

[0059] Step 4.1, set the reference signal I r and measurement signal I m After performing squaring operations to obtain two squared signals,

[0060] Number f r and f r They are represented as follows:

[0061]

[0062]

[0063] Step 4.2: Generate two orthogonal signals sin = sin(w3t) and cos = cos(w3t) using the FPGA, and then combine these two orthogonal signals with signal f respectively. r and f r After frequency mixing, four mixed signals sinA, cosA, sinB, and cosB are obtained, which are represented as follows:

[0064]

[0065]

[0066]

[0067]

[0068] Where w3 is the angular frequency of the two orthogonal signals sin and cos generated inside the FPGA, and its value is 20MHz;

[0069] Step 5: Calculate the instantaneous phase of the reference end using the full-phase FFT algorithm.

[0070] Step 5.1: Generate two complex exponential signals e A and e BThey are represented as follows:

[0071] e A =cosA +jsinA

[0072] e B =cosB + jsinB

[0073] Where j represents the imaginary part identifier in the complex number;

[0074] Step 5.2, apply the convolution window of length 2N-1 to the complex exponential signal e. A and e B The nth sample point, along with the N-1 samples before and N-1 samples after the nth sample point, totaling 2N-1 samples, are weighted, where N represents the number of sampling points in the full-phase FFT algorithm;

[0075] Step 5.3: Add the 2N-1 weighted samples in pairs with a spacing of N to obtain N summed samples;

[0076] Step 5.4: Perform a Fast Fourier Transform on the N summed samples to obtain the complex exponential signal e. A and e B The results of the full-phase spectrum analysis;

[0077] Step 5.5, complex exponential signal e A and e B In the full phase spectrum, there will be three peak points. The phase value corresponding to the peak point with the smallest frequency is the complex exponential signal e. A and e B Instantaneous phase value of the nth sample point of the intermediate frequency signal and They are represented as follows:

[0078]

[0079]

[0080] Step 6, Calculate the phase

[0081] Instantaneous phase phase and instantaneous phase phase The difference is calculated to obtain the phase containing the target displacement information. It is expressed as follows:

[0082]

[0083] The effectiveness of this invention can be further demonstrated through the following simulation.

[0084] 1. Simulation experimental conditions.

[0085] The software platform for the simulation experiments of this invention is: Windows 11 operating system and Matlab R2019b.

[0086] 2. Simulation content and result analysis.

[0087] There are two simulation experiments for this invention.

[0088] 2.1 Simulation Experiment 1 is a simulation of the phase measurement error of the heterodyne interferometric communication ranging integrated system.

[0089] Simulation Experiment 1 of this invention sets up a reference terminal electrical signal f r The frequency is 10MHz, and the electrical signal f at the measurement end is... m The frequency of 10.005MHz corresponds to the measurement of the corner reflector P. m The movement speed is 1.5 mm / s, and the sin and cosine frequencies generated inside the FPGA are set to 20 MHz; the sampling rate is set to 100 MHz, the sampling resolution is 10 bits, and the signal-to-noise ratio is 50 dB.

[0090] Simulation Experiment 1 of this invention uses an existing technique to calculate the phase measurement error when different numbers of sampling points are taken under a signal-to-noise ratio of 50dB. The relationship between the obtained phase measurement error and the different numbers of sampling points is then plotted as follows. Figure 3 The curve shown in (a).

[0091] In simulation experiment 1, the existing technology used refers to the satellite communication ranging integrated phase measurement method proposed by Xi'an University of Electronic Science and Technology in its patent application document "Integrated Target Displacement Measurement Method Based on Heterodyne Interferometric Measurement System" (application number: 202211032874.6, application date: 2022.08.26).

[0092] 2.2 Simulation Experiment 2 is a simulation of the phase measurement error of the heterodyne interferometric communication ranging integrated system.

[0093] The parameters of the heterodyne interferometric communication ranging integrated system used in simulation experiment 2 of this invention are the same as those in simulation experiment 1.

[0094] Simulation Experiment 2 of this invention uses the method of this invention to obtain the phase measurement error when calculating the phase difference with different numbers of sampling points under a signal-to-noise ratio of 50dB. The relationship between the obtained phase measurement error and the different numbers of sampling points is then plotted as follows: Figure 3 The curve shown in (b).

[0095] The following is combined with Figure 2 The simulation diagrams further illustrate the effects of the present invention.

[0096] Figure 3 In (a), the horizontal axis represents the number of different sampling points, in units of points, and the vertical axis represents the phase measurement error, in radians. Figure 3 The line in (a) represents the relationship between the phase measurement error obtained by simulation using existing technology and the number of different sampling points.

[0097] from Figure 3 (a) It can be seen that the phase measurement error obtained by the existing method is unrelated to the number of sampling points and is a random error. Under a signal-to-noise ratio of 50dB, the phase measurement accuracy of the existing method for integrated communication ranging is 2×10⁻⁶. -3 rad.

[0098] Figure 3 In (b), the horizontal axis represents the number of different sampling points, in units of points, and the vertical axis represents the phase measurement error, in radians. The line in the figure represents the relationship between the phase measurement error obtained by simulation using the method of this invention and the number of different sampling points.

[0099] from Figure 3 As can be seen from (b) in the figure, the phase measurement error obtained by the method of the present invention is not related to the number of sampling points and is a random error. Under the condition of 50dB signal-to-noise ratio, the phase measurement accuracy of the existing method for achieving integrated communication ranging is 2×10⁻⁶. -4 rad.

Claims

1. A noise-resistant satellite communication ranging and phase measurement integrated method, characterized in that, The reference and measurement electrical signals output by the heterodyne interferometric communication and ranging integrated optical system are squared respectively, and then subjected to full-phase operation. The algorithm calculates the instantaneous phase of the difference frequency signal; the steps of this method include the following: Step 1: Squaring the reference-end electrical signal and the measurement-end electrical signal output from the heterodyne interferometric communication ranging integrated optical system within the FPGA to obtain two squared signals. and ; Step 2: Generate two orthogonal signals using the FPGA, and then mix these two orthogonal signals with two squared signals respectively to obtain four mixed signals. , , and ; Step 3: Generate two complex exponential signals. and ; Step 4, through full phase Algorithms, respectively, calculate complex exponential signals and Instantaneous phase of the difference frequency signal in and ; Step 5, instantaneous phase and The difference is calculated to obtain the phase containing the target displacement information. .

2. The noise-resistant satellite communication ranging and phase measurement integrated method according to claim 1, characterized in that, The two squared signals mentioned in step 1 and It is expressed as follows: ； in, and These are the angular frequencies of the electrical signals at the reference and measurement ends, respectively. This indicates the time in the time domain of the electrical signal output by the heterodyne interferometric communication and ranging integrated optical system. and These represent the baseband data at the transmitting and receiving ends of the heterodyne interferometric communication and ranging integrated optical system, respectively. , ;in, This represents baseband data.

3. The noise-resistant satellite communication ranging and phase measurement integrated method according to claim 2, characterized in that, The four-channel mixing signals mentioned in step 2 are represented as follows: ； in, for The angular frequencies of the two orthogonal signals generated internally.

4. The noise-resistant satellite communication ranging and phase measurement integrated method according to claim 1, characterized in that, The two complex exponential signals mentioned in step 3 and They are represented as follows: ； in, The symbol representing the imaginary part of a complex number.

5. The noise-resistant satellite communication ranging and phase measurement integrated method according to claim 3, characterized in that, The full phase described in step 4 The algorithm steps are as follows: The first step is to make the length of The convolution windows are respectively for complex exponential signals and The Each sample point and the first Before each sample point individual samples and subsequent 100 sample points, total Weighting of each sample point Indicates full phase The number of sampling points in the algorithm; The second step is to The median of the weighted samples is... The sample points are added pairwise to obtain The sample points after addition; The third step is to The complex exponential signal is obtained by performing a fast Fourier transform on the summed samples. and The results of the full-phase spectrum analysis; Step 4, Complex Exponential Signals and In the full phase spectrum, there will be three peak points. Find the phase value corresponding to the peak point with the smallest frequency, which is the complex exponential signal. and The first of the intermediate frequency signal Instantaneous phase value of the sample point and as follows: ； in, and These are the angular frequencies of the electrical signals at the reference and measurement ends, respectively. for The angular frequencies of the two orthogonal signals generated internally. This indicates the time in the time domain of the electrical signal output by the heterodyne interferometric communication ranging integrated optical system.

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