A modulation method for VIENNA rectifiers to suppress current zero-crossing distortion

Abstract

This application belongs to the field of rectifier modulation, specifically a modulation method for a VIENNA rectifier to suppress current zero-crossing distortion. The method includes a VIENNA rectifier, characterized by the following modulation process between its three phases: When the current in phase A crosses zero, the modulation wave generated by SVPWM in phase A is clamped to 0, while the amplitude of the current modulation wave in phases B and C is reduced by the amplitude of the current modulation wave in phase A; when the current in phase B crosses zero, the modulation wave generated by SVPWM in phase B is clamped to 0, while the amplitude of the current modulation wave in phases A and C is reduced by the amplitude of the current modulation wave in phase B; when the current in phase C crosses zero, the modulation wave generated by SVPWM in phase C is clamped to 0, while the amplitude of the current modulation wave in phases A and B is reduced by the amplitude of the current modulation wave in phase C. Furthermore, in the first few switching cycles, the duty cycle is gradually clamped to 0, while ensuring that the zero-crossing switching cycle is long enough to eliminate the current zero-crossing distortion. Its advantage lies in eliminating the distortion generated by the VIENNA rectifier near the current zero-crossing point, thus improving current quality.

Description

Technical Field

[0001] This application belongs to the field of rectifier modulation, specifically a VIENNA rectifier modulation method for suppressing current zero-crossing distortion. Background Technology

[0002] As the global energy crisis intensifies, the efficient utilization of non-renewable energy sources such as coal, oil, and natural gas has become a critical issue that humanity urgently needs to address. In recent years, with the widespread application of new energy vehicles, VIENNA rectifiers have attracted considerable attention due to their low voltage stress, fewer switching devices, and high power density. However, due to the nonlinearity of the controlled object and the converter itself, they can easily introduce high-order harmonics and reactive power into the power grid, causing pollution and degrading power quality. Furthermore, due to inherent characteristics, when the current is near zero crossing, the reference voltage vector and the input current vector cannot be guaranteed to have identical signs, leading to current zero-crossing distortion. Summary of the Invention

[0003] To address the aforementioned problems, this application provides an improved SVPWM modulation method that eliminates the distortion generated by the VIENNA rectifier near the current zero-crossing point, thereby improving current quality. The technical solution is as follows: a VIENNA rectifier modulation method for suppressing current zero-crossing distortion, comprising a VIENNA rectifier, wherein the coupling process between its three phases is modulated as follows:

[0004] When the current of phase A crosses zero, the modulation wave generated by SVPWM in phase A is clamped to 0 in stages. At the same time, the modulation waves of phase B and phase C are reduced by the amplitude of the current modulation wave of phase A. Meanwhile, the 0 switching cycle is long enough to eliminate the phenomenon of current zero-crossing distortion.

[0005] When the current of phase B crosses zero, the modulation wave generated by SVPWM in phase B is clamped to 0 in stages. At the same time, the amplitude of the current modulation wave of phase B is reduced by the modulation waves of phases A and C. Meanwhile, the 0 switching cycle is long enough to eliminate the phenomenon of current zero-crossing distortion.

[0006] When the current of phase C crosses zero, the modulation wave of phase C generated by SVPWM is clamped to 0 in stages. At the same time, the amplitude of the current modulation wave of phase C is reduced by the modulation waves of phases A and B. Meanwhile, the 0 switching cycle is long enough to eliminate the phenomenon of current zero-crossing distortion.

[0007] During the process of clamping the modulation wave to 0 in each phase, the number of 0 switching cycles should be much larger than the number of switching cycles that gradually clamp the duty cycle to 0, in order to ensure the elimination of current zero-crossing distortion.

[0008] Preferably, the inductor voltage of the VIENNA rectifier is expressed according to the sign of the current and the sign of the modulation wave, respectively.

[0009]

[0010]

[0011] When the sign of the current is different from the sign of the modulating wave, the inductor voltage is

[0012]

[0013] Because near the zero-crossing point, the three-phase input voltage E k V k Since both are close to 0, the above equation can be simplified to:

[0014]

[0015] U dc For output voltage, i k Injecting current into three phases, V k For the sinusoidal component, V com As a common-mode component, near the zero-crossing point, since the signs of the modulated wave and the current are not the same, the inductor voltage will cause the reference voltage vector to lag behind the input current vector. Based on the circuit relationship, the lag angle θ can be solved.

[0016] Preferably, the VIENNA rectifier uses a space vector distribution diagram when employing SVPWM. The entire space vector distribution diagram can be divided into 6 large sectors, and each large sector is further divided into 6 small sectors. Within one power frequency cycle, the modulation waves within 6 lag angles θ are clamped to 0, and the 0 switching cycle is ensured to be sufficiently long. The calculation steps are as follows:

[0017] Since the lag angle is caused by the inductor voltage, the lag angle θ can be obtained based on the circuit relationship.

[0018]

[0019]

[0020] Since the rotation angle of the rotating vector within one cycle is 2π, the angle θ0 occupied by each switching cycle can be calculated.

[0021]

[0022] Among them, T S T is the carrier frequency. b The fundamental frequency;

[0023] Then the number of adjustment cycles N within each θ can be calculated as follows:

[0024]

[0025] In the formula, ω = 2πf, and R is the resistance.

[0026] Preferably, due to the symmetry of the three-phase currents, it is only necessary to analyze the changes within half a power frequency cycle to represent the zero-crossing changes of all currents within one cycle.

[0027] Within the range of π / 6+θ to π / 6, the current of phase B is near the zero-crossing point. In the first few switching cycles within this range, the modulation wave of phase B is reduced to 0 proportionally, and the modulation waves of phases A and C are also compensated proportionally. The compensation amount is the amplitude of the modulation wave of phase B divided by the number of switching cycles in which it is reduced to 0. In the remaining switching cycles, it remains at 0.

[0028] Within the range of π / 2+θ to π / 2, the current of phase A is near its zero-crossing point. In the first few switching cycles within this range, the modulation wave of phase A is proportionally reduced to 0, and the modulation waves of phases B and C are also proportionally compensated. The compensation amount is the amplitude of the modulation wave of phase A divided by the number of switching cycles in which it is reduced to 0. In the remaining switching cycles, it remains at 0.

[0029] Within the range of 5π / 6+θ to 5π / 6, the current of phase C is near its zero-crossing point. In the first few switching cycles within this range, the modulation wave of phase C is proportionally reduced to 0, and the modulation waves of phases A and B are also proportionally compensated. The compensation amount is the amplitude of the modulation wave of phase C divided by the number of switching cycles in which it is reduced to 0. In the remaining switching cycles, it remains at 0.

[0030] Preferably, the amplitude and precise range of the three-phase modulated wave generated by the VIENNA rectifier using conventional SVPWM are adjusted within the zero-crossing range as follows:

[0031]

[0032] Where K represents the number of switching cycles in which the modulation wave generated by the conventional SVPWM near the zero crossing point is gradually clamped to 0, and the size of NK represents the number of switching cycles in which the modulation wave is clamped to 0, and n is a natural number whose value ranges from [0, K].

[0033] Beneficial effects

[0034] The purpose of this invention is to propose a modulation method for VIENNA rectifiers that suppresses current distortion at zero crossings. When using traditional space vector pulse width modulation (SVPWM) in a VIENNA rectifier, the current undergoes distortion near the zero crossing, causing hysteresis in the entire current loop and introducing harmonic pollution to the power grid. Through a detailed analysis of the causes of zero-crossing distortion in VIENNA rectifiers, an improved SVPWM method is proposed to eliminate the distortion at the zero crossing. Attached Figure Description

[0035] Figure 1 This is the circuit diagram for VIENNA.

[0036] Figure 2 This is a spatial vector distribution diagram for traditional SVPWM.

[0037] Figure 3 This is the three-phase equivalent circuit of the VIENNA rectifier.

[0038] Figure 4 This refers to the actual synthetic vector route and the expected synthetic vector route.

[0039] Figure 5 This is a diagram showing the phase relationship between the reference voltage vector and the input current vector.

[0040] Figure 6 The following are the modulation and carrier waveforms of phase A of the VIENNA rectifier: (a) Modulation waveform of phase A in one power frequency cycle using conventional SVPWM; (b) Modulation waveform of phase A near the zero crossing point using conventional SVPWM; (c) Modulation waveform of phase A in one power frequency cycle using improved SVPWM; (d) Modulation waveform of phase A near the zero crossing point using improved SVPWM.

[0041] Figure 7 (a) shows the traditional SVPWM switching transistor trigger waveform and the A-phase input current waveform, while (b) shows the improved SVPWM switching transistor trigger waveform and the A-phase input current waveform.

[0042] Figure 8 (a) shows the traditional SVPWM three-phase current input waveform, while (b) shows the improved SVPWM three-phase current input waveform.

[0043] Figure 9 The waveforms show the input current and output voltage when the VIENNA rectifier starts up.

[0044] Figure 10 The input current and output voltage waveforms when a voltage change occurs in the VIENNA rectifier.

[0045] Figure 11 The waveforms of the input current and output voltage of the VIENNA rectifier when the load changes abruptly.

[0046] Figure 12 Overall flowchart of the improved SVPWM modulation method. Detailed Implementation

[0047] The following detailed descriptions are illustrative and intended to provide further explanation of this application. Unless otherwise specified, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this application pertains. It should be noted that the terminology used herein is for the purpose of describing particular embodiments only and is not intended to limit the exemplary embodiments according to this application.

[0048] A modulation method for VIENNA rectifiers to suppress current zero-crossing distortion, wherein the main circuit of the converter is as follows: Figure 1 As shown, it mainly includes: AC side inductor L = 1.8mH, diodes D1-D6, insulated gate bipolar transistors S1-S6, DC bus capacitors C1 = C2 = 1500uF, and power load R. L =75Ω, E k Including E a E b E c For three-phase input voltage, i k Including i a i b i c For three-phase injection current, U dc To output DC voltage, the switching frequency is 10kHz and the fundamental frequency is 50Hz.

[0049] The space vector distribution diagram of the VIENNA rectifier using conventional SVPWM is as follows: Figure 2 As shown. The entire spatial vector distribution diagram can be divided into 6 large sectors, and each large sector is further divided into 6 smaller sectors. The dashed lines represent the zero-crossing positions of the current in each phase. The three-phase equivalent circuit of the VIENNA rectifier is shown below. Figure 3 As shown. Based on the equivalent circuit, its three-phase equivalent model in the abc coordinate system can be established.

[0050]

[0051] When using the traditional three-level SVPWM modulation strategy, the modulation wave can be divided into a sine wave and a common-mode component. This can be expressed as:

[0052] V rk =V k +[-rV a +(r-1)V c +2r-1]=V k +V com k = a, b, c

[0053] In the formula, U on V is the voltage between the neutral point of the capacitor and the neutral point of the power grid. rk It is a three-phase modulated wave, V k For the sinusoidal component (V)k Including V a V b V c (sine component in a three-phase modulated wave), r is the redundancy vector allocation coefficient, V com This is the common-mode component.

[0054] According to the state-space averaging model, U ko U on It can be expressed as

[0055]

[0056]

[0057] Combining the above equations, when the sign of the current is the same as the sign of the modulating wave, the inductor voltage is:

[0058]

[0059] When the sign of the current is different from the sign of the modulating wave, the inductor voltage is

[0060]

[0061] Because it is near the zero point, E k V k Since both are close to 0, the above equation can be simplified to:

[0062]

[0063] It is evident that near the zero-crossing point, due to the difference in the signs of the modulation wave and the current, the influence of the inductor voltage causes the reference voltage vector to lag behind the input current vector. Since traditional SVPWM divides sectors based on the reference voltage vector, this leads to errors in the synthesized vector near the zero-crossing point when using traditional SVPWM. The actual synthesized vector path differs from the expected synthesized vector path as follows... Figure 4 As shown.

[0064] The phase relationship between its reference voltage vector and input current vector is as follows: Figure 5 As shown. Since the lag angle is caused by the inductor voltage analyzed above, the lag angle θ can be calculated based on the circuit relationship.

[0065]

[0066]

[0067] In the formula, ω = 2πf and R is the resistance.

[0068] Each phase current has two zero-crossing points within one power frequency cycle, resulting in six zero-crossing points for the three-phase current within one power frequency cycle. Furthermore, due to the coupling phenomenon between the three phases, each phase current experiences six distortions within one power frequency cycle. Improvements to the traditional SVPWM are needed at these six points.

[0069] To eliminate the current zero-crossing distortion caused by the sudden change in inductor voltage near the current zero-crossing point due to the difference in the sign of the modulating wave and the input current, based on the formula for inductor voltage when the modulating wave is different, clamping the modulating wave to 0 means that the inductor voltage is considered to be 0 near the current zero-crossing point. This ensures that the vector on the inductor voltage will not cause current distortion regardless of whether the signs of the modulating wave and the input current are the same. Based on the above analysis, within one power frequency cycle, the modulating wave needs to be clamped to 0 within 6 θ intervals. However, clamping the modulating wave to 0 will cause a sudden change in the duty cycle, which is detrimental to the stability of the entire VIENNA rectifier system. Therefore, it is necessary to gradually clamp the modulating wave to 0 and ensure that the 0-switching period is long enough.

[0070] Since the rotation angle of the rotating vector within one cycle is 2π, the angle θ0 occupied by each switching cycle can be calculated.

[0071]

[0072] Among them, T S T is the carrier frequency. b This is the fundamental frequency.

[0073] Then the number of adjustment cycles N within each θ can be calculated as follows:

[0074]

[0075] Due to the coupling between the three phases, when the current in phase A crosses zero, the current in phases B and C will also be distorted; when the current in phase B crosses zero, the current in phases A and C will also be distorted; and when the current in phase C crosses zero, the current in phases A and B will also be distorted. Therefore, when the current in phase A crosses zero, the modulation wave generated by SVPWM in phase A is clamped to 0 in stages, while the amplitude of the current modulation wave in phases B and C is reduced by the amplitude of the current modulation wave in phase A. At the same time, the 0-switching period is ensured to be long enough to eliminate the current zero-crossing distortion. When the current in phase B crosses zero, the modulation wave generated by SVPWM in phase B is clamped to 0 in stages, while the amplitude of the current modulation wave in phases A and C is reduced by the amplitude of the current modulation wave in phase B. At the same time, the 0-switching period is ensured to be long enough to eliminate the current zero-crossing distortion. When the current in phase C crosses zero, the modulation wave generated by SVPWM in phase C is clamped to 0 in stages, while the amplitude of the current modulation wave in phases A and B is reduced by the amplitude of the current modulation wave in phase C. At the same time, the 0-switching period is ensured to be long enough to eliminate the current zero-crossing distortion. During the process of clamping the modulation wave to 0 in each phase, the number of 0 switching cycles should be much larger than the number of switching cycles that gradually clamp the duty cycle to 0, in order to ensure the elimination of current zero-crossing distortion.

[0076] Based on the traditional SVPWM space voltage vector distribution diagram, the zero-crossing points of the current in each phase are approximately π / 6, π / 2, 5π / 6, 7π / 6, 3π / 2, and 11π / 6, respectively. Improvements to the traditional SVPWM are made within these six ranges. Due to the symmetry of the three-phase currents, analyzing the changes within half a power frequency cycle is sufficient to represent the zero-crossing point changes of all currents within one cycle.

[0077] 1. Within the range of π / 6+θ to π / 6, the current of phase B is near its zero-crossing point. In the first few switching cycles within this range, the modulation wave of phase B is proportionally reduced to 0, and the modulation waves of phases A and C are also proportionally compensated. The compensation amount is the amplitude of the modulation wave of phase B divided by the number of switching cycles in which it is reduced to 0. In the remaining switching cycles, it remains at 0.

[0078] 2. Within the range of π / 2+θ to π / 2, this is near the zero-crossing point of the phase A current. In the first few switching cycles within this range, the modulation wave of phase A is proportionally reduced to 0, and the modulation waves of phases B and C are also proportionally compensated. The compensation amount is the amplitude of the phase A modulation wave divided by the number of switching cycles in which it is reduced to 0. In the remaining switching cycles, it remains at 0.

[0079] 3. Within the range of 5π / 6+θ to 5π / 6, the current of phase C is near its zero-crossing point. In the first few switching cycles within this range, the modulation wave of phase C is proportionally reduced to 0, and the modulation waves of phases A and B are also proportionally compensated. The compensation amount is the amplitude of the modulation wave of phase C divided by the number of switching cycles in which it drops to 0. In the remaining switching cycles, it remains at 0.

[0080] This table lists the amplitude and precise range of the three-phase modulated wave generated by the VIENNA rectifier using conventional SVPWM within the zero-crossing range. In this table, K represents the number of switching cycles required to gradually clamp the modulated wave generated by conventional SVPWM to 0 near the zero-crossing point, and NK represents the number of switching cycles required to clamp the modulated wave to 0. n is a natural number, ranging from [0, K]. It is worth noting that the value of K is closely related to the value of N, and the carrier frequency, switching frequency, and hysteresis angle θ directly determine N. The range of K can be adjusted appropriately. When N is large, the modulated wave can be clamped to 0 more precisely; when the value of K is small, the zero-state switching cycle should be ensured to have a sufficiently long duration. Figure 6 The waveforms of the A-phase voltage modulation wave are shown respectively when using conventional SVPWM and the improved SVPWM modulation strategy of this invention. It can be clearly seen that when using the improved SVPWM modulation strategy of this invention, the modulation wave is gradually clamped to 0.

[0081]

[0082] The overall flowchart of the improved SVPWM software control is as follows: Figure 12 As shown, Hall effect sensors are used, and an AD acquisition chip and DSP are employed to acquire grid voltage, three-phase input current, and DC output voltage information. Based on the grid voltage and input current, the lag angle θ between the reference voltage and input current is calculated. The given voltage is compared with the feedback voltage, and a suitable control algorithm is selected to obtain the voltage reference vector. The voltage reference vector is then fed into a traditional SVPWM modulation algorithm to obtain the three-phase modulation signal. The three-phase modulation signal is improved within the aforementioned six θ ranges to obtain the improved SVPWM three-phase modulation signal. The improved three-phase modulation signal is then fed into a comparator to obtain the trigger signals for the six switching transistors.

[0083] Figure 7 The image shows the input current waveforms of phase A using both traditional SVPWM and improved SVPWM, as well as the trigger signal of the switching transistor. It is clearly visible that with the improved SVPWM, the switching transistor trigger signal is clamped to 0 near the current zero-crossing point. Figure 8 The waveforms of the three-phase input currents are shown after applying traditional SVPWM and improved SVPWM respectively. It can be seen that when the improved SVPWM modulation method is used, the zero-crossing distortion of the three-phase currents is eliminated. Figure 9 Figure 10 Figure 11 The current waveforms of the VIENNA rectifier during startup, sudden change of given voltage, and sudden change of load are shown respectively. It can be seen that there is no zero-crossing distortion phenomenon in any of them.

[0084] The above description is merely a preferred embodiment of this application and is not intended to limit this application. Various modifications and variations can be made to this application by those skilled in the art. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of this application should be included within the protection scope of this application.

Claims

1. A modulation method for a VIENNA rectifier to suppress current zero-crossing distortion, comprising a VIENNA rectifier, characterized in that, The coupling process between the three phases is modulated as follows: When the current of phase A crosses zero, the modulation wave generated by SVPWM in phase A is clamped to 0 in stages, while the amplitude of the current modulation wave of phase A is reduced by the modulation waves of phases B and C. At the same time, the 0 switching period is ensured to be long enough to eliminate the phenomenon of current zero-crossing distortion. When the current of phase B crosses zero, the modulation wave generated by SVPWM in phase B is clamped to 0 in stages, while the amplitude of the current modulation wave of phase B is reduced by the modulation waves of phases A and C. At the same time, the 0 switching period is ensured to be long enough to eliminate the phenomenon of current zero-crossing distortion. When the current of phase C crosses zero, the modulation wave generated by SVPWM in phase C is clamped to 0 in stages, while the amplitude of the current modulation wave of phase C is reduced by the modulation waves of phases A and B. At the same time, the 0 switching period is ensured to be long enough to eliminate the phenomenon of current zero-crossing distortion. During the clamping of each phase modulation wave to 0, the number of 0 switching cycles should be much larger than the number of switching cycles used to gradually clamp the duty cycle to 0, in order to ensure the elimination of current zero-crossing distortion; The inductor voltage of a VIENNA rectifier is expressed according to the sign of the current and the sign of the modulating wave. When the signs of the current and the modulating wave are the same, the inductor voltage is... for: ； When the sign of the current is different from the sign of the modulating wave, the inductor voltage is: ； Because it is near the zero point, , Since both are close to 0, the above equation can be simplified to: ; For three-phase input voltage, For output voltage, Injecting current into three phases For sine wave components, As the common-mode component, near the zero-crossing point, since the signs of the modulated wave and the current are not the same, the inductor voltage will cause the reference voltage vector to lag behind the input current vector. Therefore, the lag angle θ can be calculated based on the circuit relationships. The space vector distribution diagram of the VIENNA rectifier using SVPWM is divided into 6 large sectors, each of which is further divided into 6 smaller sectors. Within one power frequency cycle, the modulation wave within the 6 lag angles θ is clamped to 0, ensuring that the 0 switching cycle is long enough. The calculation steps are as follows: Since the lag angle is caused by the inductor voltage, the lag angle θ is obtained according to the circuit relationship: ； ； Since the rotation angle of the rotating vector within one cycle is 2π, the angle θ0 occupied by each switching cycle can be calculated. ; Among them, T S T is the carrier frequency. b Let θ be the fundamental frequency; then the number of cycles N adjusted within each θ can be calculated as: ； In the formula, ω = 2πf, and R is the resistance.

2. The VIENNA rectifier modulation method for suppressing current zero-crossing distortion according to claim 1, characterized in that, Due to the symmetry of the three-phase currents, analyzing the changes within half a power frequency cycle is sufficient to represent the zero-crossing changes of all currents within one cycle. Within the range of π / 6 + θ to π / 6, the B-phase current is near its zero-crossing point. In the first few switching cycles within this range, the B-phase modulation wave is proportionally reduced to 0, and the modulation waves of phases A and C are also proportionally compensated. The compensation amount is the amplitude of the B-phase modulation wave divided by the number of switching cycles in which it drops to 0. In the remaining switching cycles, it remains at 0. Within the range of π / 2 + θ to π / 2, the A-phase current is near its zero-crossing point. In the first few cycles within this range... In each switching cycle, the modulation wave of phase A is proportionally reduced to 0, and the modulation waves of phases B and C are also proportionally compensated. The compensation amount is the amplitude of the modulation wave of phase A divided by the number of switching cycles in which it drops to 0. In the remaining switching cycles, it remains 0. In the range of 5π / 6+θ to 5π / 6, it is near the zero-crossing point of the current of phase C. In the first few switching cycles within this range, the modulation wave of phase C is proportionally reduced to 0, and the modulation waves of phases A and B are also proportionally compensated. The compensation amount is the amplitude of the modulation wave of phase C divided by the number of switching cycles in which it drops to 0. In the remaining switching cycles, it remains 0.

3. The VIENNA rectifier modulation method for suppressing current zero-crossing distortion according to claim 2, characterized in that, The amplitude and precision range of the three-phase modulated wave generated by the VIENNA rectifier using conventional SVPWM are adjusted within the zero-crossing range as follows. ； It is a three-phase modulated wave. k = a, b, c Where K represents the number of switching cycles in which the modulation wave generated by the conventional SVPWM near the zero crossing point is gradually clamped to 0, and the size of NK represents the number of switching cycles in which the modulation wave is clamped to 0, and n is a natural number whose value range is [0, K].

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