A high-resolution wide-band imaging method for airborne bistatic MIMO radar

Abstract

The present invention discloses a high-resolution, wide-band imaging method for an airborne bistatic MIMO radar, comprising the following steps: obtaining an optimal pulse coding coefficient; encoding a transmitted signal to obtain a coded signal; performing matched filtering on an aliased signal received from a transmitting array element to obtain an echo signal; vectorizing the received signal to obtain a vectorized aliased signal; performing pulse coding compensation on the aliased signal to obtain a compensated echo signal; solving for the common weights corresponding to the echo data matrix to obtain the optimal weights; performing fuzzy echo suppression on the echo data matrix using the optimal weights to obtain a fuzzy-suppressed echo signal; removing the Doppler center fuzziness of the fuzzy-suppressed echo signal from the first receiving array element to obtain a compensated signal; and performing azimuth focusing on the compensated signal to obtain a final unambiguous image. The present invention can effectively suppress range fuzziness in a bistatic system, reduce the impact of fuzzy echoes, eliminate Doppler center fuzziness, and improve imaging quality.

Description

Technical Field

[0001] The present invention belongs to the technical field of radar signal processing, and in particular relates to an airborne bistatic MIMO radar high-resolution wide-band imaging method. Background Art

[0002] Due to its all-weather imaging capabilities, synthetic aperture radar (SAR) has gained widespread military application. However, due to the complex electromagnetic environment, existing monostatic radars are highly susceptible to detection and jamming. Unlike traditional monostatic radars, bistatic radars employ separate transmitters, with transmitters transmitting signals and receivers passively receiving echoes. This separation of the two provides strong anti-reconnaissance and anti-jamming capabilities. However, existing airborne bistatic imaging radars often employ single-transmitter-single-receiver (SISO) or single-transmitter-multiple-receiver (SIMO) configurations with low system degrees of freedom. Furthermore, the modeling process ignores the impact of range ambiguity on imaging and moving target detection, making these methods primarily applicable to small imaging scenarios. In practice, however, range-ambiguous echoes are unavoidable and cannot be ignored. The aliasing of ambiguous echoes with the desired echo can drown out weak target echoes, resulting in degraded imaging quality. While a low PRF (Pulse Repetition Frequency) can theoretically ensure ambiguity-free imaging of large scenes, the introduced Doppler ambiguity can hinder imaging operations. Therefore, in the dual-base system, suppressing range ambiguity or separating range-ambiguous echoes is of great significance for refining high-quality wide-band imaging.

[0003] Among existing waveform diversity technology solutions, the FDA (Frequency Diversity Array) radar system introduces a tiny frequency difference between each transmitting unit. The echo is coupled with distance and angle, giving it three-dimensional freedom in space and time. Therefore, applying the FDA radar in imaging mode can effectively suppress range ambiguity and achieve high-resolution wide-band imaging. The array element pulse coding method introduces specific phase differences between pulses, so that signals in different areas have different spatial frequencies, and then uses spatial filtering methods to achieve ambiguity and clutter separation.

[0004] However, the FDA radar uses a frequency difference between array elements to obtain the range-dimensional degree of freedom. The frequency domain implementation method has high engineering complexity and uses many approximate conditions in the imaging processing, resulting in a certain loss of imaging accuracy. In addition, the existing array element pulse coding form is considered to be a single-base front-side-view airborne model. Its coding design and characteristic analysis are only applicable to the single-base front-side-view system, which has great limitations. Moreover, the two existing technologies are both applied to the background of single-base radar. Bistatic radar has problems such as echo non-uniformity, severe spatial expansion, and echo Doppler center aliasing. Therefore, it will be difficult to achieve fuzzy echo suppression using the above methods, resulting in reduced imaging quality. Summary of the Invention

[0005] To address the above-mentioned problems in the prior art, the present invention provides an airborne bistatic MIMO radar high-resolution wide-band imaging method. The technical problem to be solved by the present invention is achieved through the following technical solutions:

[0006] An airborne bistatic MIMO radar high-resolution wide-band imaging method, the imaging method comprising:

[0007] Step 1: Minimizing a pulse coding coefficient according to a coding optimization constraint function constructed based on a desired imaging area and transceiver dual-station configuration information to obtain an optimal pulse coding coefficient, wherein the pulse coding coefficient is used to encode a transmit signal, and the transceiver dual-station includes a radar transmitting station and a radar receiving station, the radar transmitting station includes M transmitting array elements, and the radar receiving station includes N receiving array elements;

[0008] Step 2: Encode the transmission signal transmitted by the mth transmitting array element using the optimal pulse coding coefficient to obtain the mth coded signal, 1≤m≤M;

[0009] Step 3: Perform orthogonal matched filtering on the aliased signals of the M transmitting array elements received by the n-th receiving array element to obtain an echo signal received by the n-th receiving array element, wherein the aliased signals of the M transmitting array elements are signals obtained by spatially aliasing the M coded signals, and 1≤n≤N;

[0010] Step 4: vectorize all the echo signals to obtain an aliased signal received by the k-th pulse based on the vectorized echo signals;

[0011] Step 5: Perform pulse coding compensation on the aliased signal using a compensation vector to obtain a compensated echo signal;

[0012] Step 6: Rearrange the compensated echo signals to obtain an echo data matrix for each receiving element, and use a linear constrained minimum variance criterion to solve the common weights corresponding to the echo data matrix to obtain the optimal weights, wherein the common weights are used to filter out echo signals in fuzzy areas;

[0013] Step 7: Using the optimal weight, perform fuzzy echo suppression on the echo data matrix of the nth receiving element to obtain a fuzzy suppressed echo signal of the nth receiving element;

[0014] Step 8: Use a Doppler center compensation function to remove the ambiguity of the Doppler center of the ambiguity-suppressed echo signal of the first receiving array element to obtain a compensated signal, and perform azimuth focusing processing on the compensated signal to obtain a final unambiguous image, wherein the Doppler center compensation function is a function obtained based on the range frequency domain, the signal carrier frequency, and the center slant distance of the current imaging area.

[0015] In one embodiment of the present invention, the expression of the coding optimization constraint function is: min M0

[0016] stN r ≤M

[0017]

[0018]

[0019]

[0020] Among them, min(·) is minimization, M0 is the pulse coding coefficient, N r is the maximum fuzzy number, θ Tp0 is the oblique angle of the pth desired imaging area, θ Tq0 is the oblique angle of the qth desired imaging area, θ Tpa1 is half of the width of the main lobe of the transmit beam in the pth desired imaging area, θ Tqa2 is half of the main lobe coverage width of the qth region, λ is the wavelength of the transmitted signal, d T is the distance between two adjacent transmitting array elements.

[0021] In one embodiment of the present invention, the transmit signal transmitted by the mth transmit array element is encoded using a coding function, and the expression of the coding function is:

[0022]

[0023] Among them, c m (t k ) is the mth coded signal, M 0best is the optimal pulse coding coefficient, and k is the azimuth pulse index.

[0024] In one embodiment of the present invention, the expression of the aliased signal of M transmitting array elements received by the nth receiving array element is:

[0025]

[0026] Where A is the target scattering coefficient, is the quadrature signal envelope, w r (·) is the distance envelope, t r is the distance fast time, τ mn is the round-trip delay of the signal, R Tm is the slant distance between the mth transmitting array element and the scattering target, R Rn is the slant distance between the nth receiving array element and the scattering target, w a (·) is the azimuth window function, tk is the azimuth slow time, x is the x-axis coordinate of the ground scattering target, d Tm is the distance from the mth transmitting array element to the first transmitting array element, d Rn is the distance from the nth receiving element to the first receiving element, v R is the velocity component of the receiver along the x-axis, λ is the wavelength of the transmitted signal, j is the imaginary number sign, c m (t k ) is the mth coded signal.

[0027] In one embodiment of the present invention, the expression of the echo signal received by the nth receiving element is:

[0028]

[0029] Among them, φ ref is a parameter set, which includes the target scattering coefficient, the signal range envelope, the signal azimuth window function, θ′ T is the instantaneous slant angle of the transmitter, θ′ R is the instantaneous squint angle at the receiving end.

[0030] In one embodiment of the present invention, the expression of the vectored echo signal is:

[0031]

[0032]

[0033] Among them, Y(t k θ′ T ,θ′ R ) is the vectorized echo signal, a(θ′ T ,t k ) is the signal transmission steering vector, b(θ′ R ) is the signal receiving steering vector, θ′ R is the instantaneous slant angle of the receiving end, is the Kronecker product, ζ ref is the receiving signal of the reference array element, A is the target scattering coefficient, w r (·) is the distance envelope, t r is the distance fast time, τ ref is the two-way slant range at the center of the ground scattering target beam, w a (·) is the azimuth window function, t k is the azimuth slow time, x is the x-axis coordinate of the ground scattering target, v R is the velocity of the receiver along the x-axis, R T1 is the slant range from the first transmitting array element to the ground scattering target, R R1is the slant range from the first receiving array element to the ground scattering target, and λ is the wavelength of the transmitted signal;

[0034] The expression of the aliased signal received by the k-th pulse from each imaging area is:

[0035]

[0036] Among them, Θ k is the set of all targets within the coverage of the kth pulse beam, Θ k ∈{i|θ′ pR,il ∈θ R_beamform}, θ R_beamform is the beam coverage of the receiving end, L is the number of range gates, θ′ pT,il is the instantaneous azimuth angle between the i-th target in the p-th desired imaging area at the l-th range gate and the reference transmitting array element, θ′ pR,il N represents the instantaneous azimuth angle between the i-th target in the p-th desired imaging area and the reference receiving array element at the l-th range gate, r is the maximum fuzzy number.

[0037] In one embodiment of the present invention, the expression of the compensated echo signal is:

[0038]

[0039] Among them, x * (k) is the echo signal after compensation, C(k) is the compensation vector, c * (k) is the coding steering vector before compensation, 1 N is the N×1 dimensional identity matrix, is the Kronecker product, ⊙ is the Hadamard product, and x(k) is the aliased signal received by the k-th pulse.

[0040] In one embodiment of the present invention, the expression of the common weight is:

[0041] w=v(v H v) -1 F

[0042] Among them, w is the common weight, v is the final constraint matrix, v = [v1 v′], v1 is the constraint guidance matrix, v′ is the zero-point expansion matrix, F is the region selection matrix, F = [F1 T F′ T ] T , F1=[1 0…0] T , F′ is an all-zero matrix, [·] T is the conjugate transpose operation;

[0043] The solution method for the optimal weight is:

[0044]

[0045] stv H w=F

[0046] in, is the echo data matrix of the nth receiving array element, w(x * (k)) is the common weight of the fuzzy suppressed echo signal of the nth receiving array element, and H is the conjugate transpose.

[0047] In one embodiment of the present invention, the expression of the ambiguity suppressed echo signal of the nth receiving element of the kth pulse is:

[0048]

[0049] in, is the fuzzy suppressed echo signal of the nth receiving element of the kth pulse, w best is the optimal weight, is the echo data matrix of the nth receiving element, and H is the conjugate transpose.

[0050] In one embodiment of the present invention, the expression of the ambiguity suppressed echo signal of the first receiving array element is:

[0051]

[0052] Among them, w r (·) is the distance dimension envelope, t r is the distance time, τ ref is the two-way slant range of the ground scattering target beam center at the moment, w a (·) is the echo azimuth envelope, t k is the azimuth slow time, x is the x-axis coordinate of the ground scattering target, v R is the velocity of the receiver along the x-axis, R T1 is the slant range from the first transmitting array element to the ground scattering target, R R1 is the slant range from the first receiving array element to the ground scattering target, λ is the wavelength of the transmitted signal, d R1 is the receiving array element spacing, θ′ R1 The receiving oblique angle corresponding to the center of the current imaging area;

[0053] The expression of the Doppler center compensation function is:

[0054]

[0055] Among them, R′(0;0,0) is the first-order coefficient of the slant range expansion of the current imaging area center, f r is the distance frequency domain, fc is the signal carrier frequency, c is the speed of light;

[0056] The expression of the compensated signal is:

[0057]

[0058] in, is the compensated signal.

[0059] Beneficial effects of the present invention:

[0060] Aiming at the problem that the existing dual-base imaging system does not consider the distance ambiguity, the present invention establishes a corresponding dual-base forward-looking imaging model, analyzes the fuzzy echo characteristics under the dual-base system, and solves the problem of establishing the dual-base system fuzzy echo model.

[0061] In view of the problem that existing waveform grouping systems mostly adopt a single-base model and suffer from serious mismatch in a dual-base system, the present invention expands the array element pulse coding form, optimizes the coding design and zero-point extension method, and adopts a specific optimized filter to achieve fuzzy echo separation, thereby solving the problems of non-uniform distribution of dual-base echoes and severe spatial expansion leading to reduced separation effect, and effectively reducing the impact of range fuzzy echoes on imaging.

[0062] The present invention aims to solve the problem that range ambiguity and Doppler center ambiguity affect imaging quality in a dual-base system. By separating fuzzy echoes and constructing a specific Doppler compensation function, the Doppler center ambiguity can be effectively improved. BRIEF DESCRIPTION OF THE DRAWINGS

[0063] Figure 1 This is a flow chart of a method for high-resolution wide-band imaging using an airborne bistatic MIMO radar, provided by an embodiment of the present invention;

[0064] Figure 2 A schematic diagram of the fuzzy area echo imaging result provided by an embodiment of the present invention;

[0065] Figure 3 A schematic diagram of the imaging result of range gate No. 514 provided in an embodiment of the present invention. DETAILED DESCRIPTION

[0066] The present invention will be further described in detail below with reference to specific examples, but the embodiments of the present invention are not limited thereto.

[0067] Example 1

[0068] See Figure 1 , Figure 1The flowchart of an airborne bistatic MIMO radar high-resolution wide-width imaging method provided by an embodiment of the present invention is shown. The airborne bistatic MIMO (Multiple Input Multiple Output) radar high-resolution wide-width imaging method provided by an embodiment of the present invention includes steps 1 to 8, wherein:

[0069] Step 1: Minimize the pulse coding coefficient according to the coding optimization constraint function constructed based on the desired imaging area and the transceiver dual-station configuration information to obtain the optimal pulse coding coefficient, wherein the pulse coding coefficient is used to encode the transmitted signal. The transceiver dual-station includes a radar transmitting station and a radar receiving station. The radar transmitting station includes M transmitting array elements, and the radar receiving station includes N receiving array elements.

[0070] Specifically, the radar transmitting station and radar receiving station of the dual-base are separated. The radar transmitting station has M transmitting array elements, and the radar receiving station has N receiving array elements. The radar transmitting station works in the front view and the radar receiving station works in the front view. The transmitting antenna consists of a linear array of M array elements. The first transmitting array element is the reference transmitting array element. The distance between the mth transmitting array element and the first transmitting array element is d. Tm =(m-1)d T ,(m=1,…,M), where d T is the distance between two adjacent transmitting array elements. Similarly, the distance between the nth transmitting array element and the first transmitting array element is d Rn =(n-1)d R , (n=1,…,N). Using the existing transceiver dual-station configuration information (the transceiver dual-station configuration information is the position, speed, direction information of the receiving platform and the transmitting platform, as well as the angular distance information relative to the desired imaging area) and the desired imaging area information, a coding optimization constraint function is constructed. The expression of the coding optimization constraint function is as follows:

[0071] min M0

[0072] stN r ≤M

[0073]

[0074]

[0075]

[0076] Among them, min(·) is minimization, M0 is the pulse coding coefficient, N r is the maximum fuzzy number, θ Tp0 is the oblique angle of the pth desired imaging area, θ Tq0is the oblique angle of the qth desired imaging area, θ Tpa1 is half of the width of the main lobe of the transmit beam in the pth desired imaging area, θ Tqa2 is half of the main lobe coverage width of the beam in the qth region, and λ is the wavelength of the transmitted signal.

[0077] Step 2: Encode the transmit signal transmitted by the mth transmit array element using the optimal pulse coding coefficient to obtain the mth coded signal, and direct the transmit main lobe toward the desired imaging area.

[0078] Specifically, by solving the optimization problem in step 1, the applicable optimal pulse coding coefficient can be obtained, so that the transmitted signal is encoded using a coding function. The expression of the coding function is:

[0079]

[0080] Among them, c m (t k ) is the mth coded signal, t k is the azimuth slow time, M 0best is the optimal pulse coding coefficient, and k is the azimuth pulse index.

[0081] Step 3: Perform orthogonal matched filtering on the aliased signals of the M transmitting array elements received by the n-th receiving array element to obtain an echo signal received by the n-th receiving array element, where the aliased signals of the M transmitting array elements are signals obtained by spatially aliasing the M coded signals.

[0082] Specifically, the expression of the aliased signal of M transmitting array elements received by the nth receiving array element is:

[0083]

[0084] Where A is the target scattering coefficient, is the quadrature signal envelope, w r (·) is the distance envelope, t r is the distance fast time, τ mn is the round-trip delay of the signal, R Tm is the slant distance between the mth transmitting array element and the scattering target, R Rn is the slant distance between the nth receiving array element and the scattering target, w a (·) is the azimuth window function, x is the x-axis coordinate of the ground scattering target, v R is the velocity component of the receiver along the x-axis, and j is the imaginary number sign.

[0085] After matched filtering at the receiving end, the signal of each transmitting array element can be obtained. At this time, the expression of the echo signal received by the nth receiving array element is:

[0086]

[0087] Among them, φ ref is a parameter set, which includes the target scattering coefficient, the signal range envelope, the signal azimuth window function, θ′ T is the instantaneous slant angle of the transmitter, θ′ R is the instantaneous squint angle at the receiving end.

[0088] Similarly, by performing M times of matched filtering on each receiving array element, the echo of each transmitting array element at each receiving array element can be obtained.

[0089] Step 4: vectorize all echo signals to obtain an aliased signal received by the k-th pulse based on the vectorized echo signals.

[0090] Specifically, the expression of the vectored echo signal is:

[0091]

[0092]

[0093] Among them, Y(t k θ′ T ,θ′ R ) is the vectorized echo signal, a(θ′ T ,t k ) and b(θ′ R ) are the signal transmission steering vector and the signal reception steering vector, is the Kronecker product, ζ ref is the reference array element receiving signal, τ ref is the two-way slant range at the center of the ground scattering target beam, R T1 is the slant range from the first transmitting array element to the ground scattering target, R R1 is the slant range from the first receiving array element to the ground scattering target.

[0094] The expression of the signal transmission steering vector is:

[0095] a(θ′ T ,t k )=g(θ′ T )⊙c(k)

[0096] The expression of the signal reception steering vector is:

[0097]

[0098] Among them, g(θ′ T ) is the echo receiving angle steering vector, c(k) is the coding steering vector, ⊙ is the Hadamard product, dRN is the distance between the Nth receiving element and the reference receiving element, [·] T Represents the conjugate transpose operation.

[0099] The expression of the echo reception angle steering vector is:

[0100]

[0101] The expression of the coded steering vector is:

[0102]

[0103] Among them, d TM is the distance between the Mth transmitting array element and the reference transmitting array element.

[0104] Therefore, the expression of the aliased signal received by the kth pulse from each imaging area is:

[0105]

[0106] Among them, Θ k is the set of all targets within the coverage of the kth pulse beam, Θ k ∈{i|θ′ pR,il ∈θ R_beamform}, θ R_beamform is the beam coverage of the receiving end, L is the number of range gates, θ′ pT,il is the instantaneous azimuth angle between the i-th target in the p-th desired imaging area at the l-th range gate and the reference transmitting array element, θ′ pR,il It represents the instantaneous azimuth between the i-th target in the p-th desired imaging area at the l-th range gate and the reference receiving array element, where the reference transmitting array element is the first receiving array element, N r is the maximum fuzzy number.

[0107] Step 5: Use the compensation vector to perform pulse coding compensation on the aliased signal to obtain a compensated echo signal.

[0108] Specifically, in order to solve the coding dependency, a compensation vector is constructed, and the expression of the compensation vector is:

[0109]

[0110] Among them, C(k) is the compensation vector, c * (k) is the coding steering vector before compensation, 1 N is the N×1 dimensional identity matrix.

[0111] Afterwards, the aliasing signal is pulse coded compensated using the compensation vector. The expression of the compensated echo signal is:

[0112]

[0113] Among them, x * (k) is the compensated echo signal, and x(k) is the aliased signal received by the k-th pulse.

[0114] Therefore, the compensated code steering vector is:

[0115] Step 6: Rearrange the compensated echo signals to obtain the echo data matrix of each receiving array element, and use the linear constrained minimum variance criterion to solve the common weights corresponding to the echo data matrix to obtain the optimal weights, where the common weights are used to filter out the echo signals in the fuzzy area.

[0116] The rearrangement method is: for the original MN*1 data matrix (i.e. the compensated echo signal), extract the M data of the corresponding n-th receiving array element, and arrange the extracted data into an M*1 echo data matrix in the order of the transmitting array elements. The echo data matrix of the n-th receiving array element is

[0117] The expression of the common weight corresponding to the echo data matrix is solved using the linear constrained minimum variance criterion:

[0118]

[0119] stv H w=F

[0120] The expression of ordinary weight is:

[0121] w=v(v H v) -1 F

[0122] Among them, w is the common weight, v is the final constraint matrix, v = [v1 v′], v1 is the constraint steering matrix, is the echo transmitting end steering vector, is the emission squint angle of the center of the Nrth region, v′ is the zero-point expansion matrix, θ T1 The first angle near the oblique angle of the blurred area center, θ TJ is the J-th angle near the oblique angle of the fuzzy area center, F is the area selection matrix, F=[F1 T F′ T ] T , F1=[1 0…0] T , F′ is an all-zero matrix, w(x * (k)) is the common weight of the fuzzy suppressed echo signal of the nth receiving array element, and H is the conjugate transpose.

[0123] In summary, the constraint steering matrix v1 and the extraction matrix F1 are constructed by utilizing the prior angle information of each fuzzy area. In order to improve the suppression effect, the zero-point extension matrix v′ and the suppression constraint matrix F′ are constructed in the specific fuzzy echo direction, and the final constraint matrix v = [v1 v′] is obtained. Subsequently, the optimal weight is solved using the linear constrained minimum variance criterion.

[0124] Step 7: Use the optimal weight to perform fuzzy echo suppression on the echo data matrix of the nth receiving array element to obtain the fuzzy suppressed echo signal of the nth receiving array element.

[0125] Specifically, the fuzzy echo filter is designed using the optimal weights in step 6 to extract the echoes in each desired area and achieve fuzzy echo separation. Specifically, the fuzzy echo suppression is performed on the echo data matrix of each receiving element using the optimal weights. The expression of the fuzzy suppressed echo signal of the nth receiving element of the kth pulse is:

[0126]

[0127] in, is the fuzzy suppressed echo signal of the nth receiving element of the kth pulse, w best For the optimal right.

[0128] Step 8. Use the Doppler center compensation function to remove the ambiguity of the Doppler center of the ambiguity-suppressed echo signal of the first receiving array element to obtain a compensated signal, and perform azimuth focusing processing on the compensated signal to obtain the final unambiguous image, where the compensation function is a function obtained based on the range frequency domain, the signal carrier frequency, and the center slant distance of the current imaging area.

[0129] Specifically, the expression of the suppressed signal in the first receiving element is:

[0130]

[0131] Among them, w r (·) is the distance dimension envelope, w a (·) is the echo azimuth envelope, d R1 is the receiving array element spacing, θ′ R1 The receiving oblique angle corresponding to the center of the current imaging area.

[0132] The expression of the Doppler center compensation function is:

[0133]

[0134] Among them, R′(0;0,0) is the first-order coefficient of the slant range expansion of the current imaging area center, f r is the distance frequency domain, f cis the signal carrier frequency, and c is the speed of light.

[0135] The expression of the compensated signal is:

[0136]

[0137] in, is the compensated signal.

[0138] Because traditional waveform diversity coding design and processing procedures are not applicable to airborne bistatic radars, the present invention proposes a high-resolution wide-band imaging method for airborne bistatic MIMO radars. This method is used for MIMO radars. By optimizing the design of coding coefficients and transmitting pulse-coded signals, the present invention increases the spatial distinction of echoes in different regions. By optimizing the filter, the present invention effectively separates echoes in fuzzy regions and suppresses the impact of echoes in fuzzy regions on imaging. The present invention also constructs a Doppler compensation function using prior configuration information to remove Doppler center blur, thereby improving imaging quality.

[0139] Simulation experiments can further demonstrate the beneficial effects of the present invention.

[0140] In the simulation experiment, equidistant linear arrays are used for both transmission and reception. The number of transmitting elements is set to 6, the number of receiving elements is set to 4, and the element spacing is 0.0156m. The transmitting station operates in the front-view mode at a speed of 200m / s, and the receiving station operates in the front-view mode at a speed of 250m / s. The carrier frequency of the transmitted signal is 9.6GHz, the bandwidth is 100MHz, and the pulse repetition frequency is 4000Hz, corresponding to a maximum unambiguous distance of 75km and a distance ambiguity number of 2.

[0141] Simulation content: Under the above simulation parameters, the results of the airborne dual-base system without range ambiguity separation and the imaging processing using the present invention are compared, and the imaging quality is compared.

[0142] Figure 2 The imaging processing results of the second imaging area before and after suppression are given. Figure 2 The region in represents the imaging area, the Azimuth cell represents the azimuth unit, and the Range cell represents the distance unit. Figure 2 (a) It can be seen that although the target echo in the first imaging area enters from the beam sidelobe and is mismatched in azimuth, making it impossible to focus well, it is still strong due to its small attenuation due to its short distance. However, in the second imaging area, due to the serious attenuation of distance, the difference between the corresponding echo and the blurred energy is not large, which seriously affects the imaging. Figure 2 (b) It can be seen that when the coding coefficient is set to 4, the normalized spatial frequency difference between the two imaging areas increases from 0.062 in the traditional MIMO system to 0.312, and the distance between the two is significantly increased, thereby effectively suppressing the blurred echo in the first imaging area.

[0143] Figure 3 The imaging results of the 514th range gate for fuzzy echo separation are given. Figure 3 (a) is the result before fuzzy signal separation, Figure 3 (b) is the result after fuzzy signal separation. Figure 3 The Power in represents the signal energy. Figure 3 It can be seen that before the fuzzy signal separation, the echo of the first imaging area occupies multiple azimuth units due to defocus. Although the second imaging area is well focused, its amplitude is reduced by about 8.5dB compared with the first imaging area. After using the distance fuzzy separation method proposed in the present invention, the echo peak of the first imaging area is suppressed by about 28dB, and the echo of the second imaging area is effectively retained.

[0144] It can be seen that the method proposed in the present invention can effectively achieve range ambiguity suppression under the dual-base system, reduce the influence of blurred echoes, eliminate Doppler center ambiguity, and improve imaging quality.

[0145] In summary, the present invention addresses the issue of range ambiguity in bistatic radars leading to degraded imaging quality by proposing a method for separating fuzzy echoes with optimized coding and filtering designs, effectively suppressing range-ambiguous echoes and improving imaging. To address the shortcomings of traditional pulse coding in bistatic systems, an optimization function is established by considering the non-uniform spatial distribution and scalability of echoes from different regions of an airborne bistatic radar, optimizing the pulse coding design and improving echo separation. To address the problem of traditional filters with poor echo suppression in fuzzy regions due to overly narrow nulls, a zero-point expansion method is employed to effectively improve clutter suppression performance.

[0146] In the description of this specification, the reference terms "one embodiment", "some embodiments", "example", "specific example", or "some examples" mean that the specific features, structures, materials, or characteristics described in conjunction with the embodiment or example are included in at least one embodiment or example of the present invention. In this specification, the schematic representations of the above terms do not necessarily refer to the same embodiment or example. Moreover, the specific features, structures, materials, or characteristics described can be combined in any appropriate manner in any one or more embodiments or examples. In addition, those skilled in the art can combine and combine different embodiments or examples described in this specification.

[0147] Although the present application is described herein in conjunction with various embodiments, in the process of implementing the claimed application, those skilled in the art can understand and implement other changes to the disclosed embodiments by reviewing the drawings, the disclosure, and the appended claims. In the claims, the word "comprising" does not exclude other components or steps, and "a" or "an" does not exclude multiple situations. A single processor or other unit can implement several functions listed in the claims. Certain measures are recorded in different dependent claims, but this does not mean that these measures cannot be combined to produce good results.

[0148] The above is a further detailed description of the present invention in conjunction with specific preferred embodiments, and the specific implementation of the present invention should not be considered to be limited to these descriptions. For those skilled in the art of the present invention, without departing from the concept of the present invention, several simple deductions or substitutions can be made, which should be considered to fall within the scope of protection of the present invention.

Claims

1. A high-resolution wide-band imaging method for airborne bistatic MIMO radar, characterized in that: The imaging method comprises: Step 1: Minimizing a pulse coding coefficient according to a coding optimization constraint function constructed based on a desired imaging area and transceiver dual-station configuration information to obtain an optimal pulse coding coefficient, wherein the pulse coding coefficient is used to encode a transmit signal, and the transceiver dual-station includes a radar transmitting station and a radar receiving station, the radar transmitting station includes M transmitting array elements, and the radar receiving station includes N receiving array elements; Step 2: Encode the transmission signal transmitted by the mth transmitting array element using the optimal pulse coding coefficient to obtain the mth coded signal, 1≤m≤M; Step 3: Perform orthogonal matched filtering on the aliased signals of the M transmitting array elements received by the n-th receiving array element to obtain an echo signal received by the n-th receiving array element, wherein the aliased signals of the M transmitting array elements are signals obtained by spatially aliasing the M coded signals, and 1≤n≤N; Step 4: vectorize all the echo signals to obtain an aliased signal received by the k-th pulse based on the vectorized echo signals; Step 5: Perform pulse coding compensation on the aliased signal using a compensation vector to obtain a compensated echo signal; Step 6: Rearrange the compensated echo signals to obtain an echo data matrix for each receiving element, and use a linear constrained minimum variance criterion to solve the common weights corresponding to the echo data matrix to obtain the optimal weights, wherein the common weights are used to filter out echo signals in fuzzy areas; Step 7: Using the optimal weight, perform fuzzy echo suppression on the echo data matrix of the nth receiving element to obtain a fuzzy suppressed echo signal of the nth receiving element; Step 8: Use a Doppler center compensation function to remove the ambiguity of the Doppler center of the ambiguity-suppressed echo signal of the first receiving array element to obtain a compensated signal, and perform azimuth focusing processing on the compensated signal to obtain a final unambiguous image, wherein the Doppler center compensation function is a function obtained based on the range frequency domain, the signal carrier frequency, and the center slant distance of the current imaging area.

2. The airborne bistatic MIMO radar high-resolution wide-band imaging method according to claim 1, characterized in that: The expression of the encoding optimization constraint function is: min M0 s.t.N r ≤M Among them, min(·) is minimization, M0 is the pulse coding coefficient, N r is the maximum fuzzy number, θ Tp0 is the oblique angle of the pth desired imaging area, θ Tq0 is the oblique angle of the qth desired imaging area, θ Tpa1 is half of the width of the main lobe of the transmit beam in the pth desired imaging area, θ Tqa2 is half of the main lobe coverage width of the qth region, λ is the wavelength of the transmitted signal, d T is the distance between two adjacent transmitting array elements.

3. The airborne bistatic MIMO radar high-resolution wide-band imaging method according to claim 1, characterized in that: The transmission signal transmitted by the mth transmitting array element is encoded by a coding function, and the expression of the coding function is: Among them, c m (t k ) is the mth coded signal, M 0best is the optimal pulse coding coefficient, and k is the azimuth pulse index.

4. The airborne bistatic MIMO radar high-resolution wide-band imaging method according to claim 1, wherein: The expression of the aliased signal of the M transmitting array elements received by the nth receiving array element is: Where A is the target scattering coefficient, is the quadrature signal envelope, w r (·) is the distance envelope, t r is the distance time, τ mn is the round-trip delay of the signal, R Tm is the slant distance between the mth transmitting array element and the scattering target, R Rn is the slant distance between the nth receiving array element and the scattering target, w a (·) is the azimuth window function, t k is the azimuth slow time, x is the x-axis coordinate of the ground scattering target, d Tm is the distance from the mth transmitting array element to the first transmitting array element, d Rn is the distance from the nth receiving element to the first receiving element, v R is the velocity component of the receiver along the x-axis, λ is the wavelength of the transmitted signal, j is the imaginary number sign, c m (t k ) is the mth coded signal.

5. The airborne bistatic MIMO radar high-resolution wide-band imaging method according to claim 4, characterized in that: The expression of the echo signal received by the nth receiving array element is: Among them, φ ref is a parameter set, which includes the target scattering coefficient, the signal range envelope, the signal azimuth window function, θ′ T is the instantaneous slant angle of the transmitter, θ′ R is the instantaneous squint angle at the receiving end.

6. The airborne bistatic MIMO radar high-resolution wide-band imaging method according to claim 1, characterized in that: The expression of the vectored echo signal is: Among them, Y(t k θ′ T ,θ′ R ) is the vectorized echo signal, a(θ′ T ,t k ) is the signal transmission steering vector, b(θ′ R ) is the signal receiving steering vector, θ′ R is the instantaneous slant angle of the receiving end, is the Kronecker product, ζ ref is the receiving signal of the reference array element, A is the target scattering coefficient, w r (·) is the distance envelope, t r is the distance time, τ ref is the two-way slant range at the center of the ground scattering target beam, w a (·) is the azimuth window function, t k is the azimuth slow time, x is the x-axis coordinate of the ground scattering target, v R is the receiver speed along the x-axis, R T1 is the slant range from the first transmitting array element to the ground scattering target, R R1 is the slant range from the first receiving array element to the ground scattering target, and λ is the wavelength of the transmitted signal; The expression of the aliased signal received by the k-th pulse from each imaging area is: Among them, Θ k is the set of all targets within the coverage of the k-th pulse beam, θ R_beamform is the beam coverage of the receiving end, L is the number of range gates, θ′ pT,il is the instantaneous azimuth angle between the i-th target in the p-th desired imaging area and the reference transmitting array element at the i-th range gate, θ′ pR,il N represents the instantaneous azimuth angle between the i-th target in the p-th desired imaging area and the reference receiving array element at the l-th range gate, r is the maximum fuzzy number.

7. The airborne bistatic MIMO radar high-resolution wide-band imaging method according to claim 1, characterized in that: The expression of the compensated echo signal is: Among them, x * (k) is the echo signal after compensation, C(k) is the compensation vector, c * (k) is the coding steering vector before compensation, 1 N is the N×1 dimensional identity matrix, is the Kronecker product, ⊙ is the Hadamard product, and x(k) is the aliased signal received by the k-th pulse.

8. The airborne bistatic MIMO radar high-resolution wide-band imaging method according to claim 1, characterized in that: The expression of the common right is: w=v(v H v) -1 F Among them, w is the common weight, v is the final constraint matrix, v = [v1 v′], v1 is the constraint guidance matrix, v′ is the zero-point expansion matrix, F is the region selection matrix, F = [F1 T F′ T ] T , F1=[1 0 … 0] T , F′ is an all-zero matrix, [·] T is the conjugate transpose operation; The solution method for the optimal weight is: s.t.v H w=F in, is the echo data matrix of the nth receiving array element, w(x * (k)) is the common weight of the fuzzy suppressed echo signal of the nth receiving array element, and H is the conjugate transpose.

9. The airborne bistatic MIMO radar high-resolution wide-band imaging method according to claim 1, characterized in that: The expression of the fuzzy suppressed echo signal of the nth receiving element of the kth pulse is: in, is the fuzzy suppressed echo signal of the nth receiving element of the kth pulse, w best is the optimal weight, is the echo data matrix of the nth receiving array element, and H is the conjugate transpose.

10. The airborne bistatic MIMO radar high-resolution wide-band imaging method according to claim 1, characterized in that: The expression of the fuzzy suppressed echo signal of the first receiving array element is: Among them, w r (·) is the distance dimension envelope, t r is the distance fast time, τ ref is the two-way slant range of the ground scattering target beam center at the moment, w a (·) is the echo azimuth envelope, t k is the azimuth slow time, x is the x-axis coordinate of the ground scattering target, v R is the velocity of the receiver along the x-axis, R T1 is the slant range from the first transmitting array element to the ground scattering target, R R1 is the slant range from the first receiving array element to the ground scattering target, λ is the wavelength of the transmitted signal, d R1 is the receiving array element spacing, θ′ R1 The receiving oblique angle corresponding to the center of the current imaging area; The expression of the Doppler center compensation function is: Among them, R′(0;0,0) is the first-order coefficient of the slant range expansion of the current imaging area center, f r is the distance frequency domain, f c is the signal carrier frequency, c is the speed of light; The expression of the compensated signal is: in, is the compensated signal.

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