A method and system for hierarchical synchronous reception of weak spread spectrum signals

Abstract

The present invention discloses a hierarchical synchronous reception method for weak spread spectrum signals. Based on the long coherent accumulation mechanism, the method utilizes the difference in sensitivity of different spread spectrum code lengths to frequency deviation, and realizes efficient synchronous reception of signals by hierarchically capturing the code phase and compensating for the frequency deviation in steps. Compared with the traditional two-dimensional joint search (code phase and frequency) method, the present invention optimizes the search process into a one-dimensional search, avoiding the exhaustive search in the frequency dimension, thereby significantly reducing the computational complexity and improving the processing efficiency. The present invention adopts a strategy of capturing the code phase step by step, combined with a coarse estimation and fine estimation scheme of the frequency deviation, to ensure high capture sensitivity and synchronization accuracy. Especially in low signal-to-noise ratio environments and complex channel conditions, the present invention can effectively improve the reception reliability of spread spectrum signals. The method and system have the advantages of low computational complexity, simple hardware implementation, and strong adaptability, and provide an innovative solution for the reception and processing of weak spread spectrum signals.

Description

Technical Field

[0001] The present invention belongs to the technical field of wireless spread spectrum communication, and in particular relates to a hierarchical synchronous receiving method and system for weak spread spectrum signals. Background Art

[0002] Direct sequence spread spectrum systems, with their superior anti-interference, anti-interception, and anti-multipath capabilities, have secured an irreplaceable position in modern communications technology. With the continuous advancement of satellite communications, mobile communications, new communication data services, and navigation, positioning, and measurement and control technologies, higher requirements are being placed on receiver demodulation capabilities, especially in low signal-to-noise ratio environments and with significant Doppler shift, placing increasing demands on device complexity and performance.

[0003] Satellite communications and complex mobile channel environments often experience wide frequency offsets, weak signal power, and low signal-to-noise ratios. To address this challenge, acquisition, the first step in signal synchronization, not only detects weak signals but also synchronizes the pseudo-code phase and carrier frequency offset, providing a reliable foundation for subsequent despreading and demodulation.

[0004] Existing weak signal acquisition algorithms typically increase the signal-to-noise ratio gain by extending the integration time. However, because data bits in pseudo-code correlation operations may transition during the integration time, this can affect signal peak detection and limit the integration time. To address this issue, a half-bit integration algorithm was proposed. This algorithm divides the data into odd and even parts for processing, ensuring that at least a portion of the data is unaffected by transitions. However, this method cuts the integration time in half, resulting in low data utilization. A further improvement, the full-bit algorithm, generates multiple sets of correlation values ​​through data shifting and selects the set that does not contain transition points. However, this method is computationally intensive and complex. Furthermore, incoherent integration effectively overcomes the impact of data symbol transitions by squaring the coherent results, but this method introduces squaring loss. The differential coherence algorithm reduces interference from symbol transitions by conjugate multiplying the preceding and subsequent coherent results, thus improving the squaring loss problem. In recent years, improved methods based on the above algorithms have emerged to further enhance weak signal acquisition performance.

[0005] However, under low signal-to-noise ratio (SNR) communication conditions, traditional direct sequence spread spectrum systems are extremely sensitive to subtle variations in frequency offset. When there is a frequency offset between the received signal and the local carrier, the height of the correlation peak decreases sharply as the frequency offset increases, following the squared envelope of the sinc function. Severely attenuated correlation peaks are likely to be submerged below the noise floor, posing significant challenges for receiver signal acquisition. Although measures such as incoherent integration have been introduced at the receiver end to enhance weak signal detection, fundamental improvements in acquisition performance are difficult to achieve. Therefore, achieving synchronous signal acquisition requires a joint search in both the pseudo-code phase and Doppler frequency offset to improve acquisition success rates. Common acquisition methods often combine serial and parallel approaches, including serial and parallel pseudo-code search and serial and parallel frequency search. Serial acquisition algorithms are simple in principle and easy to implement, but they are time-consuming and lack real-time performance. In contrast, parallel acquisition algorithms improve acquisition efficiency and shorten acquisition time, but also require more hardware resources. With the continuous advancement of Fast Fourier Transform (FFT) technology, many FFT-based acquisition algorithms have been proposed. These methods significantly improve search speed and effectively reduce acquisition time. However, such algorithms also bring challenges such as increased computational complexity and increased hardware implementation difficulty.

[0006] Traditional direct sequence spread spectrum (DSSS) system acquisition algorithms face a trade-off between acquisition sensitivity and search complexity. This trade-off is exacerbated by their high sensitivity to frequency offset, particularly in communication environments with wide frequency offsets and low signal-to-noise ratios, limiting the system's practical application. Maintaining signal acquisition sensitivity under low SNR conditions often requires significant hardware resources and time to complete a joint search of the pseudo-code phase and frequency offset at the receiver. However, this costly implementation approach is difficult to meet practical requirements. Therefore, designing an efficient, fast, and computationally low-complexity weak signal synchronous reception algorithm is crucial for alleviating this trade-off.

[0007] The invention patent application with publication number CN202410943072.3 discloses a capture device for direct sequence spread spectrum signals under low signal-to-noise ratio. The implementation steps are as follows: (1) the signal preprocessing module performs orthogonal up-down conversion, sampling rate conversion and matched filtering on the received signal to generate a digital baseband signal; (2) the local spread spectrum code module generates a local spread spectrum code that matches the transmitting end according to the protocol; and performs correlation matching with the digital baseband signal to generate a correlation sequence; (3) the differential transformation module performs first-order differential processing on the correlation sequence, the coherent accumulation module accumulates the differential sequence to improve the signal-to-noise ratio, and the dynamic threshold calculation module calculates the dynamic threshold value; (4) the capture decision module combines the cumulative maximum value, dynamic threshold and fixed threshold to make a decision, and the K-time confirmation module verifies the continuity and consistency of the decision result, confirming that the capture is successful or continuing the search. Compared with the parallel processing method, this method requires multiple steps of accumulation and judgment operations, the capture time is longer, the real-time performance is relatively poor, and the search complexity in the frequency dimension is still high, which is not suitable for scenarios with large frequency deviations.

[0008] A high-precision spread spectrum signal capture method is disclosed in the patent application document with publication number CN202310024540.2. Its implementation steps are as follows: (1) sampling the received signal and dividing it into several frequency sub-slots according to the carrier frequency deviation range, and performing carrier frequency compensation correction on the sampled data; (2) using fast Fourier transform (FFT) to correlate and despread the local pseudo code with the received signal, generate correlation integral data, and calculate the adaptive detection threshold of the correlation integral, while obtaining the correlation integral peak value and its corresponding pseudo code phase and carrier frequency; (3) using trigonometric relationship to correct the pseudo code phase and using quadratic function to interpolate and correct the carrier frequency to improve the capture accuracy; (4) by comparing the corrected correlation integral peak value with the detection threshold, determine whether the capture is successful, and further solve the received signal-to-noise ratio, pseudo code phase and carrier Doppler shift. This method is based on interpolation correction instead of directly reducing the search step size. Although it improves the capture accuracy, it increases the computational complexity and puts forward higher design requirements and resource requirements for hardware implementation. Summary of the Invention

[0009] In order to overcome the above-mentioned shortcomings of the prior art, the purpose of the present invention is to provide a method and system for hierarchical synchronous reception of weak spread spectrum signals, aiming to solve the problems of high synchronization complexity and low capture sensitivity existing in the prior art.

[0010] In order to achieve the above-mentioned purpose, the present invention utilizes a flexible long coherent accumulation mechanism, fully considers the difference in sensitivity of different spread spectrum code lengths to frequency deviation, and realizes synchronous reception by hierarchical capture of code phase and step-by-step compensation of frequency deviation. It also converts the traditional two-dimensional joint search (code phase and frequency) into a one-dimensional search, avoiding the exhaustive search in the frequency dimension and significantly reducing the computational complexity. In addition, the step-by-step capture of code phase further improves the hardware processing speed, while the combination of coarse estimation of frequency deviation and fine correction ensures a higher capture sensitivity. The present invention can provide a new solution for the efficient reception of weak spread spectrum signals and reliable processing under low signal-to-noise ratio conditions.

[0011] The technical solution adopted by the present invention can be described as follows:

[0012] A method for hierarchical synchronous reception of a weak spread spectrum signal comprises the following steps:

[0013] Step 1: At the transmitter, two signals with different information transmission rates are assigned spreading codes of different lengths, which are then spread to the same bandwidth and loaded onto the I and Q orthogonal channels, respectively. The spreading code length of the Q channel signal is smaller than that of the I channel signal.

[0014] Step 2: At the receiving end, the two signals are received and processed respectively, and the tolerance range of the differential coherent integration results under different spreading code lengths to the Doppler frequency offset is determined;

[0015] Step 3: Within the maximum acceptable Doppler frequency deviation range of the Q-channel short spreading code determined in step 2, set the pseudo code phase search step size of the local receiver, and perform an FFT-based parallel code phase search on the Q-channel received signal to complete the synchronization of the Q-channel short spreading code and achieve the first-level capture of the pseudo code phase;

[0016] Step 4: Based on the Q-path short spreading code synchronization completed in step 3, the Q-path received signal is despread and dedifferentiated, and an FFT-based parallel code phase search is performed using the Q-path synchronization header to complete the second-level capture of the pseudo code phase, ultimately achieving frame synchronization of the Q-path received signal.

[0017] Step 5: Combined with the wide frequency offset tolerance of the Q-path short spreading code, the second-level captured position of the pseudo code phase obtained in step 4 is directly used to perform coarse frequency offset estimation and perform preliminary frequency offset compensation on the received signal.

[0018] In step 6, based on the second-level capture position of the pseudo code phase obtained in step 4, the I-channel received signal is despread and dedifferentiated, and a correlation operation is performed on the I-channel received signal in combination with the I-channel synchronization header. Fine frequency offset estimation is achieved through the integration result. Subsequently, fine frequency offset correction is performed on the received signal to complete high-precision synchronization of the received signal.

[0019] In one embodiment, the step 1 of assigning spreading codes of different lengths to two signals with different information transmission rates is implemented as follows:

[0020] The signal with less information is called I signal, which has a low information transmission rate. It is spread twice, i.e., a long spreading code is assigned to it.

[0021] The Q-channel signal contains more information and has a higher information transmission rate. It is spread once, i.e., a short spreading code is assigned to it.

[0022] The spreading code for the second spread of the I-channel signal and the spreading code for the Q-channel signal are orthogonal to each other, and the two signals pass through the I and Q orthogonal channels.

[0023] In one embodiment, in step 1, the two signals expanded to the same bandwidth are loaded onto the I and Q orthogonal channels respectively using 2DPSK modulation, and the orthogonal characteristics of the spread spectrum code are utilized to achieve independent transmission of the I and Q signals, and have variable rate transmission capability; in step 2, the receiving end uses two independent receiving channels to process the input signals as two 2DPSK modulated spread spectrum signals respectively.

[0024] In one embodiment, the step 2 of determining the tolerance range of the differential dry integration results to Doppler frequency offset under different length spreading codes is implemented as follows:

[0025] Step 2.1, ignore the pseudo code, only consider the Doppler frequency error of the signal, and express it in the form of a complex number;

[0026] Step 2.2, using a single spreading code period of length N as the coherence time, coherently integrate all chips of the kth spreading code;

[0027] Step 2.3, perform forward and backward conjugate multiplication on the two adjacent coherent integration results to obtain the differential coherent integration result;

[0028] In step 2.4, the differential coherent integration results are analyzed to obtain the system's tolerance range for Doppler frequency offset when the spreading code lengths are different. That is, the smaller N is, the greater the system's tolerance range for Doppler frequency offset is.

[0029] In one embodiment, in step 2.1, for the kth spreading code, the starting carrier phase difference including the carrier frequency offset corresponding to the nth chip in the spreading code period is:

[0030]

[0031] In step 2.2, the phase information after coherent integration is expressed as follows:

[0032]

[0033] The result of the forward and backward conjugate multiplication in step 2.3 is expressed as follows:

[0034]

[0035] Among them, f s is the chip rate, That is, 1 chip width; t k is the starting time of the kth spreading code; N is the length of the spreading code cycle, that is, the number of chips contained in a single spreading code, is the Doppler frequency shift estimate; is the initial carrier phase; φ coh,k is the phase angle of the kth spreading code after coherent integration, and the time interval between two adjacent spreading codes is Right now D k 、D k+1 They are different data bits corresponding to the kth and k+1th spreading codes respectively.

[0036] In one embodiment, the first-level capture of the pseudo code phase in step 3 is implemented as follows:

[0037] Step 3.1: Ensure that the frequency deviation between the Q-channel received signal and the local carrier is within the tolerance range of the Q-channel short spreading code.

[0038] Step 3.2, perform fast Fourier transform on the Q-channel received signal to convert the signal from the time domain to the frequency domain;

[0039] Step 3.3: Set the pseudo code phase search step size of the local receiver, perform fast Fourier transform on the locally generated Q-path short spreading code, and perform conjugate processing on the transformed frequency domain signal;

[0040] Step 3.4: multiply the frequency domain result obtained in step 3.2 by the conjugate frequency domain signal obtained in step 3.3, and then perform an inverse fast Fourier transform to restore the result to the time domain;

[0041] Step 3.5: Perform multiple non-coherent integrations on the time domain result obtained in step 3.4 to increase the signal accumulation gain, and perform peak detection on the non-coherent accumulation result to determine the synchronization position of the Q-path short spread spectrum code.

[0042] In one embodiment, the second-stage capture of the pseudo code phase in step 4 is implemented as follows:

[0043] Step 4.1: Despread the Q-channel received signal based on the code synchronization position of the Q-channel short spreading code obtained in step 3, and perform differential decoding of the adjacent code elements before and after the despread signal to eliminate the initial carrier frequency offset;

[0044] Step 4.2, performing a fast Fourier transform on the signal obtained after differential decoding in step 4.1 to convert the signal from the time domain to the frequency domain;

[0045] Step 4.3: Perform fast Fourier transform on the Q-channel synchronization header and perform conjugate processing on the signal after transforming into the frequency domain;

[0046] Step 4.4: multiply the frequency domain result obtained in step 4.2 by the conjugate frequency domain signal obtained in step 4.3, and then perform an inverse fast Fourier transform to restore the result to the time domain;

[0047] In step 4.5, the time domain results obtained in step 4.4 are directly accumulated several times, and then the peak value of the accumulated results is detected to determine the pseudo code phase position corresponding to the Q-channel synchronization header, thereby achieving frame synchronization of the Q-channel received signal.

[0048] In one embodiment, the step 5 of performing preliminary frequency offset compensation on the received signal is implemented as follows:

[0049] Step 5.1: directly use the second-level capture position of the pseudo code phase obtained in step 4 to extract the frequency offset information before modulo-taking the peak at the correlation peak position to perform coarse frequency offset estimation;

[0050] Step 5.2: Based on the frequency offset information obtained in step 5.1, perform preliminary frequency offset compensation on the Q-channel received signal to reduce the range of the remaining frequency offset.

[0051] In one embodiment, the step 6, performing fine frequency offset correction on the received signal, is implemented as follows:

[0052] Step 6.1: Based on the second-stage capture position of the pseudo code phase obtained in step 4, i.e., the frame synchronization position, frame synchronization is performed on the I-channel received signal. Combined with the locally generated I-channel long spreading code, the synchronized I-channel received signal is despread.

[0053] Step 6.2: differentially decode the adjacent symbols before and after the despread I-channel received signal to eliminate the initial carrier frequency offset;

[0054] Step 6.3: Use the I-channel synchronization header to perform a correlation operation on the I-channel received signal, perform peak detection on the obtained integration result, extract the frequency offset information from the correlation peak, and achieve fine frequency offset estimation;

[0055] In step 6.4, based on the result of the fine frequency offset estimation, the received signal is finely frequency offset corrected to correct the residual frequency offset error and achieve high-precision synchronization of the received signal.

[0056] The present invention also provides a hierarchical synchronous receiving system for weak spread spectrum signals, comprising:

[0057] The spread spectrum signal frame structure design module designs a spread spectrum signal frame structure based on a dual information rate transmission system. By loading two signals with different spreading code lengths onto the I and Q orthogonal channels, the independent transmission of the two signals is ensured. The spreading code length of the Q channel signal is smaller than that of the I channel signal.

[0058] The frequency error range determination module analyzes and determines the tolerance range of the differential coherent integration results of the I and Q channel signals under different spread spectrum code lengths to the Doppler frequency offset;

[0059] The pseudo code phase first-level capture module performs parallel code phase search based on fast Fourier transform on the Q-path short spread spectrum code, realizing the first-level capture of the pseudo code phase of the Q-path short spread spectrum code;

[0060] The pseudo code phase secondary capture module despreads and dedifferentiates the Q-channel received signal based on the completion of the primary capture. It also uses the Q-channel synchronization header to perform FFT-based parallel code phase search to complete the secondary capture of the pseudo code phase and achieve frame synchronization of the Q-channel received signal.

[0061] The coarse frequency offset estimation and compensation module uses the correlation peak position obtained by the second-stage capture of the Q-path pseudo-code phase to extract the frequency offset information before the peak is modulo-ed, i.e., the residual frequency offset information. It performs coarse frequency offset estimation through frequency domain analysis and performs preliminary frequency offset compensation on the received signal.

[0062] The precise frequency offset estimation and compensation module despreads and dedifferentiates the I-channel received signal, performs correlation operations on the I-channel received signal in conjunction with the I-channel synchronization header, achieves precise frequency offset estimation through integration results, and performs fine frequency offset correction on the received signal to ensure high-precision signal synchronization.

[0063] Compared with the prior art, the present invention has the following beneficial effects:

[0064] First, this invention significantly reduces the search dimension and the algorithm's computational complexity by optimizing the traditional combined code phase and frequency search method into a one-dimensional search. This improvement not only avoids exhaustive searches in the frequency dimension, reducing computational time and resources, but also significantly improves acquisition efficiency, resulting in superior system performance in complex channel environments, particularly in scenarios with large frequency ranges.

[0065] Second, the present invention employs a hierarchical acquisition strategy, combining step-by-step code phase adjustment with coarse frequency offset estimation and fine compensation. This layer-by-layer optimization approach significantly improves the hardware system's processing speed while maintaining acquisition sensitivity. Coarse frequency offset estimation rapidly reduces the frequency uncertainty range, while fine compensation provides more precise frequency offset correction, ensuring efficient synchronization in low signal-to-noise ratio environments.

[0066] Third, the present invention utilizes a flexible long coherent accumulation method, fully considering the differences in frequency offset sensitivity of different spread spectrum code lengths, and designs a synchronous reception process that is more adaptable to complex communication scenarios. Long coherent accumulation not only improves the signal's anti-interference capability but also, through synergy with the hierarchical capture strategy, optimizes the pseudo-code synchronization and frequency compensation process. This enables the system to maintain a high synchronization success rate even in the presence of significant Doppler shift or low signal power, while also enhancing its adaptability to dynamic channel environments. BRIEF DESCRIPTION OF THE DRAWINGS

[0067] Figure 1 It is a flow chart of the method implementation of the present invention.

[0068] Figure 2 This is a schematic diagram of the design of the I and Q channel spread spectrum signal frame structure at the transmitting end provided by the present invention.

[0069] Figure 3 It is a system block diagram of the receiving end hierarchical synchronization implementation method provided by the present invention.

[0070] Figure 4 It is a block diagram of a pseudo code phase first-level capture module in the hierarchical synchronous receiving method provided by an embodiment of the present invention.

[0071] Figure 5 This is a diagram of the pseudo code phase first-level capture simulation results provided by an embodiment of the present invention.

[0072] Figure 6 It is a block diagram of a pseudo code phase secondary capture module in the hierarchical synchronous receiving method provided by an embodiment of the present invention.

[0073] Figure 7 This is a diagram of the simulation results of the pseudo code phase two-stage capture provided by an embodiment of the present invention.

[0074] Figure 8 This is a block diagram of a precise frequency offset estimation and compensation module in a hierarchical synchronous reception method provided by an embodiment of the present invention. DETAILED DESCRIPTION

[0075] In order to better understand the technical solution of the present invention, the present invention is further described in detail below with reference to the accompanying drawings and embodiments. It is also stated that the embodiments described below with reference to the accompanying drawings are exemplary and are intended to be used to explain the present invention, but should not be understood as limiting the present invention.

[0076] Experimental condition parameter settings:

[0077] First, a set of raw data to be transmitted was generated. The I-channel signal frame header length was set to 13, the primary spreading code length was 15, and the secondary spreading code length was 1023. The Q-channel signal frame header length was set to 127, and the spreading code length was 1023. Next, to simulate the dynamic characteristics of a real communication environment, the code phase offset was set to 2230 sampling points, and a Doppler frequency shift of 3833 Hz was introduced. Gaussian white noise was then added to the signal to simulate a low signal-to-noise ratio (SNR) reception signal for subsequent reception experiments.

[0078] Reference Figure 1 The specific implementation of the hierarchical synchronous reception system for weak spread spectrum signals in this embodiment relies on the following modules:

[0079] 1. Spread spectrum signal frame structure design module: Design a spread spectrum signal frame structure based on a dual information rate signal transmission system. By loading two signals with different spread spectrum code lengths onto the I and Q orthogonal channels, the independent transmission of the two signals is ensured, while providing basic support for subsequent hierarchical synchronous reception.

[0080] 2. Frequency Error Range Determination Module: This module analyzes and determines the tolerance of the differential coherent integration results of the I and Q signals for Doppler frequency offset at different spreading code lengths. This module improves the system's adaptability to frequency offset errors, ensures the accuracy and stability of subsequent synchronization and compensation processes, and provides parameter basis for subsequent acquisition steps.

[0081] 3. Pseudo-code phase first-level capture module: This module performs parallel code phase search based on fast Fourier transform (FFT) on the Q-path short spreading code, quickly completing the first-level capture of the pseudo-code phase of the Q-path short spreading code, laying the foundation for frame synchronization and frequency offset estimation.

[0082] 4. Pseudo-code phase secondary acquisition module: Based on the completion of the first-stage acquisition, it despreads and dedifferentiates the Q-channel received signal and uses the Q-channel synchronization header to perform an FFT-based parallel code phase search to complete the second-stage acquisition of the pseudo-code phase. This module accurately determines the location of the pseudo-code phase synchronization header and achieves frame synchronization of the Q-channel received signal.

[0083] 5. Coarse Frequency Offset Estimation and Compensation Module: This module uses the Q-channel frame synchronization capture results to extract residual frequency offset information, performs coarse frequency offset estimation through frequency domain analysis, and performs preliminary frequency offset compensation on the two received signals to reduce the frequency offset range in subsequent processing. The Q-channel frame synchronization capture results here refer to the correlation peak position obtained by the second-stage capture of the Q-channel pseudo-code phase. The frequency offset information before modulo-taking of this peak, i.e., the residual frequency offset information, is extracted.

[0084] 6. Precision Frequency Offset Estimation and Compensation Module: Despreads and dedifferentiates the I-channel received signal, performs correlation operations on the I-channel received signal in conjunction with the I-channel synchronization header, achieves precise frequency offset estimation through integration results, and performs fine frequency offset correction on the two received signals to ensure high-precision signal synchronization.

[0085] The spread spectrum signal frame structure design module needs to be deployed at the transmitter, refer to Figure 2 The specific implementation of the dual information rate signal transmission system is as follows: of the two signals, one has short data bits, contains less information, and has a low information transmission rate. To ensure the reliability of signal reception, this signal is set as the I channel and subjected to two-stage spread spectrum processing, that is, assigned a long spreading code to achieve the long spreading code spreading effect. The other signal has long data bits, contains more information, and has a high information transmission rate. It is set as the Q channel and only undergoes single spread spectrum processing, that is, assigned a short spreading code to achieve the short spreading code spreading effect. Through the above operation, the two signals with different information transmission rates are respectively assigned spreading codes of different lengths. The difference in information bit length ensures that the total frame length of the transmitted signal after spread spectrum is consistent.

[0086] For example, the data bit length of the I signal is set to 128 bits, and a two-stage spreading process is performed: the first stage uses a 15-bit m-sequence for primary spreading; the second stage uses a 1023-bit Gold code for secondary spreading, achieving long-code transmission. The data bit length of the Q signal is set to 2002 bits, and a 1023-bit Gold code, orthogonal to the secondary spreading code of the I signal, is directly used for primary spreading. By using the information bit length difference operation, the two signals ensure that the total frame length of the transmitted signal after spreading is consistent, both being 2130 × 1023.

[0087] Subsequently, the two signals are expanded to the same bandwidth and modulated using a 2DPSK or other modulation scheme, respectively, onto the I and Q orthogonal channels. Leveraging the orthogonal nature of the spreading codes, the I and Q signals are independently transmitted, with variable-rate transmission capabilities. Because the spreading code used for the second spread of the I signal and the Q signal are orthogonal, and the two signals pass through the I and Q orthogonal channels, complete orthogonality is achieved through carrier and code orthogonality, enabling independent transmission of two signals with different information rates and noise immunity.

[0088] The frequency error range determination module is deployed at the receiving end. Based on the transmitting end frame structure, the receiving end adopts two independent receiving channels to receive two signals respectively, and treats the input signals as two 2DPSK modulated spread spectrum signals for processing. The tolerance range of the differential coherent integration results under different spread spectrum code lengths to the Doppler frequency deviation is analyzed as follows: First, the formula of the capture scheme is simplified. It is assumed that the correlator can completely strip off the spread spectrum code of the received signal. Therefore, the pseudo code is ignored in the signals involved subsequently. Only the Doppler frequency error of the signal is considered and expressed in the form of a complex number. For the kth spread spectrum code, the starting carrier phase difference including the carrier frequency deviation corresponding to the nth code chip in its spread spectrum code period is:

[0089]

[0090] Then, based on the implementation principle of coherent integration, a mathematical formula is derived. A single spreading code period of length N is used as the coherence time. All chips of the kth spreading code are coherently integrated. The phase information obtained by integration is expressed as follows:

[0091]

[0092] The conjugate multiplication of the two adjacent coherent integration results is performed to derive the differential coherent integration result formula:

[0093]

[0094] Among them, t k is the starting time of the kth spreading code, is the initial carrier phase, D k 、D k+1 are the different data bits corresponding to the kth and k+1th spreading codes, is the Doppler frequency offset estimate, N is the length of the spreading code period, that is, the number of chips contained in a single spreading code, f s is the chip rate, That is, 1 chip width, φ coh,k is the phase angle of the kth spreading code after coherent integration, which reflects the phase difference including Doppler frequency offset estimation. The time interval between the two adjacent spreading codes is Right now

[0095] Analyzing the differential coherent integration formula reveals that the correlation peak after differential coherent integration is affected by the spreading code length N. Specifically, as N decreases, the attenuation of the correlation peak due to Doppler offset during the coherent integration process decreases, thereby extending the system's tolerance to Doppler offset. This indicates that a shorter spreading code length allows the system to better adapt to larger Doppler offsets, meaning that the system's tolerance to Doppler offset is greater.

[0096] In the embodiment of the present invention, the signal transmission bandwidth is 20.46 MHz. If a 1023-bit spreading code is selected as the coherent integration time, the bandwidth that affects the coherent peak is: That is, the tolerable Doppler frequency deviation range is: [-10KHZ, 10KHZ];

[0097] If a spreading code of 1023×15 is selected as the coherent integration time, the bandwidth that affects the coherent peak is: That is, the tolerable Doppler frequency deviation range is: [-0.667KHZ, 0.667KHZ].

[0098] Through the above analysis, in the embodiment of the present invention, under the Doppler frequency offset range of [-10KHZ, 10KHZ], the error loss caused by the frequency offset on the coherent peak of the short spreading code is within an acceptable range. Therefore, the superior characteristics of the short spreading code, which are strong in resisting the interference of the long frequency offset, can be utilized to directly search the one-dimensional code phase dimension, avoiding the exhaustive search in the frequency dimension. After compensating for the coarse frequency offset, the range of the remaining frequency offset is reduced. Then, the characteristics of the long spreading code that are sensitive to frequency offset and have high frequency offset estimation accuracy can be utilized to further obtain a more refined frequency offset estimate.

[0099] Reference Figure 3 The system implementation of the receiving end hierarchical synchronous receiving method provided by the present invention mainly includes a pseudo code phase first-level capture module, a pseudo code phase second-level capture module, a coarse frequency offset estimation and compensation module, and a fine frequency offset estimation and compensation module.

[0100] Reference Figure 4 The pseudo code phase first-level acquisition module sets the pseudo code phase search step size of the local receiver within the maximum acceptable Doppler frequency deviation range of the determined Q-channel short spreading code, and performs FFT-based parallel code phase search on the Q-channel received signal to complete the synchronization of the Q-channel short spreading code and achieve the first-level acquisition of the pseudo code phase. The specific implementation process is as follows:

[0101] Step 1: Ensure that the frequency deviation between the Q-channel received signal and the local carrier is controlled within the tolerance range of the Q-channel short spreading code to ensure the effectiveness of subsequent processing. In this embodiment, the tolerance range is [-10 kHz, 10 kHz].

[0102] Step 2: Perform a fast Fourier transform on the Q-channel received signal. In this embodiment, data longer than a short code period is taken and a 2048-point FFT is performed to convert the signal from the time domain to the frequency domain.

[0103] Step 3: Set the pseudo code phase search step size of the local receiver to, for example, 1 chip, perform the same FFT as in step 2 on the locally generated Q-path short spreading code, and perform conjugate processing on the transformed frequency domain signal.

[0104] Step 4: Multiply the frequency domain result obtained in step 2 by the conjugate frequency domain signal obtained in step 3, and then perform an inverse fast Fourier transform (IFFT) to restore the result to the time domain.

[0105] Step 5: Perform multiple (1500 times in this embodiment) non-coherent integrations on the time domain result obtained in step 4 to increase the signal accumulation gain, and perform peak detection on the non-coherent accumulation result to obtain the synchronization position of the Q-path short spread spectrum code, such as Figure 5 As shown in the figure, it can be seen that the captured code phase position is 184 sampling points, which is an integer (2) short spread spectrum code elements different from the set value 2230 (2×1023+184) sampling points. Therefore, Q-path short spread spectrum code synchronization can be performed correctly.

[0106] Reference Figure 6 The pseudo code phase secondary capture module, based on the Q-channel short spread spectrum code synchronization, despreads and dedifferentiates the Q-channel received signal, and uses the Q-channel synchronization header to perform FFT-based parallel code phase search to complete the second-level capture of the pseudo code phase, ultimately achieving frame synchronization of the Q-channel received signal. The specific implementation process is as follows:

[0107] Step 1: Despread the Q-channel received signal based on the code synchronization position of the Q-channel short spreading code obtained in the pseudo-code phase first-level capture module, and perform differential decoding of the adjacent code elements before and after the despread signal to eliminate the initial carrier frequency offset;

[0108] Step 2: Perform a fast Fourier transform on the signal obtained after differential decoding. In this embodiment, data longer than one frame is taken and a 4096-point FFT is performed to convert the signal from the time domain to the frequency domain.

[0109] Step 3: Perform the same FFT as in step 2 on the Q-channel synchronization header, and perform conjugate processing on the signal after transforming into the frequency domain;

[0110] Step 4: multiply the frequency domain result obtained in step 2 by the conjugate frequency domain signal obtained in step 3, and then perform an inverse fast Fourier transform to restore the result to the time domain to obtain a time domain correlation signal;

[0111] Step 5: Since the polarity of the synchronization header is unique, the time domain results obtained in step 4 are directly accumulated a certain number of times, and then the peak value of the accumulated results is detected to determine the pseudo code phase position corresponding to the Q-channel synchronization header, such as Figure 7 As shown in the figure, it can be seen that the captured code phase is 3 or 2133 (2130+3) (since the searched data length is greater than one frame, it may be possible that both frame signals are synchronized. The two positions with a difference of an integer frame length are both correct frame synchronization positions, and the maximum of the two values ​​is sufficient), which is the same as the actual position of the synchronization head. Therefore, the frame synchronization of the Q-channel received signal can be correctly performed.

[0112] The coarse frequency offset estimation and compensation module, combined with the wide frequency offset tolerance of the Q-path short spreading code, directly uses the Q-path frame synchronization capture results to perform coarse frequency offset estimation and perform preliminary frequency offset compensation on the two received signals. The specific implementation process is as follows:

[0113] Step 1: Since the receiving end offsets the initial carrier phase error through conjugate differential decoding and demodulation, the pseudo-code phase capture result obtained in the pseudo-code phase secondary capture module can be directly used to extract the frequency deviation information of the Q-channel received signal as a coarse frequency deviation estimate, without the need for additional frequency deviation search or channel state estimation. Since the Q-channel short spread spectrum code has a shorter code length and a wider tolerance range for frequency deviation, the extracted frequency deviation value is used as the basis for coarse frequency deviation estimation. In this embodiment, the frequency deviation estimate extracted from the correlation peak position obtained from the code phase secondary capture is 3876.1036Hz, which is relatively close to the actual frequency deviation setting value of 3833Hz.

[0114] Step 2: Based on the coarse frequency offset information obtained in Step 1, perform preliminary frequency offset compensation on the Q-channel received signal, reducing the remaining frequency offset to approximately 34 Hz, which is well within the Doppler frequency offset tolerance of the I-channel long spreading code. This provides more reliable input conditions for subsequent fine frequency offset correction and ensures the reliability of the subsequent fine frequency offset correction process.

[0115] Reference Figure 8 The precise frequency offset estimation and compensation module despreads and dedifferentiates the I-channel received signal based on the pseudo-code phase capture result, and performs correlation operations on the I-channel received signal in combination with the I-channel synchronization header. The module then implements precise frequency offset estimation based on the integration result. Subsequently, it performs precise frequency offset correction on the two received signals to achieve high-precision synchronization of the received signals. The specific implementation process is as follows:

[0116] Step 1: Despread the synchronized I-channel received signal based on the obtained pseudo code phase synchronization position and the locally generated I-channel long spreading code;

[0117] Step 2: Perform differential decoding on the adjacent symbols of the despread I-channel received signal to eliminate the initial carrier frequency offset.

[0118] Step 3: Use the I-channel synchronization header to perform a correlation operation on the I-channel received signal. Peak detection is performed on the resulting integration result to extract the frequency offset information from the correlation peak, achieving a precise frequency offset estimation. In this embodiment, the frequency offset estimate extracted from the correlation peak obtained from the I-channel synchronization header coherent integration result is -38.5934 Hz, further reducing the difference between the actual residual frequency offset and the frequency offset.

[0119] Step 4: Based on the fine frequency offset estimation result obtained in step 3, the received signal is finely frequency-offset corrected. The corrected frequency offset error differs from the actual value by only a few Hz, thus achieving high-precision synchronization of the received signal.

[0120] In summary, the present invention is based on a flexible long coherent accumulation mechanism, fully utilizing the differences in the sensitivity of different spread spectrum code lengths to frequency deviation, and realizes efficient synchronous reception of signals by hierarchically capturing the code phase and compensating for the frequency deviation in steps. Compared with the traditional two-dimensional joint search (code phase and frequency) method, the present invention optimizes the search process to a one-dimensional search, avoiding the exhaustive search in the frequency dimension, thereby significantly reducing the computational complexity and improving processing efficiency. The system adopts a strategy of capturing the code phase step by step, combined with a coarse estimation and fine estimation scheme of the frequency deviation, to ensure high capture sensitivity and synchronization accuracy. Especially in low signal-to-noise ratio environments and complex channel conditions, the present invention can effectively improve the reception reliability of spread spectrum signals. The method and system have the advantages of low computational complexity, simple hardware implementation, and strong adaptability, and provide an innovative solution for the reception and processing of weak spread spectrum signals.

Claims

1. A method for hierarchical synchronous reception of weak spread spectrum signals, characterized in that: The steps include: Step 1: At the transmitter, two signals with different information transmission rates are assigned spreading codes of different lengths, which are then spread to the same bandwidth and loaded onto the I and Q orthogonal channels, respectively. The spreading code length of the Q channel signal is smaller than that of the I channel signal. Step 2: At the receiving end, the two signals are received and processed respectively, and the tolerance range of the differential coherent integration results under different spreading code lengths to the Doppler frequency offset is determined; Step 3: Within the maximum acceptable Doppler frequency deviation range of the Q-channel short spreading code determined in step 2, set the pseudo code phase search step size of the local receiver, and perform an FFT-based parallel code phase search on the Q-channel received signal to complete the synchronization of the Q-channel short spreading code and achieve the first-level capture of the pseudo code phase; Step 4: Based on the Q-path short spreading code synchronization completed in step 3, the Q-path received signal is despread and dedifferentiated, and an FFT-based parallel code phase search is performed using the Q-path synchronization header to complete the second-level capture of the pseudo code phase, ultimately achieving frame synchronization of the Q-path received signal. Step 5: Combined with the wide frequency offset tolerance of the Q-path short spreading code, the second-level captured position of the pseudo code phase obtained in step 4 is directly used to perform coarse frequency offset estimation and perform preliminary frequency offset compensation on the received signal. In step 6, based on the second-level capture position of the pseudo code phase obtained in step 4, the I-channel received signal is despread and dedifferentiated, and a correlation operation is performed on the I-channel received signal in combination with the I-channel synchronization header. Fine frequency offset estimation is achieved through the integration result. Subsequently, fine frequency offset correction is performed on the received signal to complete high-precision synchronization of the received signal.

2. The method for hierarchical synchronous reception of weak spread spectrum signals according to claim 1, wherein: The step 1 is to allocate spreading codes of different lengths to two signals with different information transmission rates, and the implementation method is as follows: The signal with less information is called I signal, which has a low information transmission rate. It is spread twice, i.e., a long spreading code is assigned to it. The Q-channel signal contains more information and has a higher information transmission rate. It is spread once, i.e., a short spreading code is assigned to it. The spreading code for the second spread of the I-channel signal and the spreading code for the Q-channel signal are orthogonal to each other, and the two signals pass through the I and Q orthogonal channels.

3. The method for hierarchical synchronous reception of weak spread spectrum signals according to claim 1 or 2, characterized in that: In step 1, the two signals expanded to the same bandwidth are loaded onto the I and Q orthogonal channels respectively using 2DPSK modulation, and the orthogonal characteristics of the spread spectrum code are utilized to realize independent transmission of the I and Q signals, and have variable rate transmission capability; in step 2, the receiving end uses two independent receiving channels and processes the input signals as two 2DPSK modulated spread spectrum signals respectively.

4. The method for hierarchical synchronous reception of weak spread spectrum signals according to claim 1, wherein: The step 2 is to determine the tolerance range of the differential dry integration results under different length spreading codes to the Doppler frequency offset, and the implementation method is as follows: Step 2.1, ignore the pseudo code, only consider the Doppler frequency error of the signal, and express it in the form of a complex number; Step 2.2, using a single spreading code period of length N as the coherence time, coherently integrate all chips of the kth spreading code; Step 2.3, perform forward and backward conjugate multiplication on the two adjacent coherent integration results to obtain the differential coherent integration result; In step 2.4, the differential coherent integration results are analyzed to obtain the system's tolerance range for Doppler frequency offset when the spreading code lengths are different. That is, the smaller N is, the greater the system's tolerance range for Doppler frequency offset is.

5. The method for hierarchical synchronous reception of weak spread spectrum signals according to claim 4, characterized in that: In step 2.1, for the kth spreading code, the starting carrier phase difference including the carrier frequency offset corresponding to the nth chip in the spreading code period is: In step 2.2, the phase information after coherent integration is expressed as follows: The result of the forward and backward conjugate multiplication in step 2.3 is expressed as follows: Among them, f s is the chip rate, That is, 1 chip width; t k is the starting time of the kth spreading code; N is the length of the spreading code cycle, that is, the number of chips contained in a single spreading code, is the Doppler frequency shift estimate; is the initial carrier phase; φ coh,k is the phase angle of the kth spreading code after coherent integration, and the time interval between two adjacent spreading codes is Right now D k 、D k+1 They are different data bits corresponding to the kth and k+1th spreading codes respectively.

6. The method for hierarchical synchronous reception of weak spread spectrum signals according to claim 1, characterized in that: The first-level capture implementation process of the pseudo code phase in step 3 is as follows: Step 3.1: Ensure that the frequency deviation between the Q-channel received signal and the local carrier is within the tolerance range of the Q-channel short spreading code. Step 3.2, perform fast Fourier transform on the Q-channel received signal to convert the signal from the time domain to the frequency domain; Step 3.3: Set the pseudo code phase search step size of the local receiver, perform fast Fourier transform on the locally generated Q-path short spreading code, and perform conjugate processing on the transformed frequency domain signal; Step 3.4: multiply the frequency domain result obtained in step 3.2 by the conjugate frequency domain signal obtained in step 3.3, and then perform an inverse fast Fourier transform to restore the result to the time domain; Step 3.5: Perform multiple non-coherent integrations on the time domain result obtained in step 3.4 to increase the signal accumulation gain, and perform peak detection on the non-coherent accumulation result to determine the synchronization position of the Q-path short spread spectrum code.

7. The method for hierarchical synchronous reception of weak spread spectrum signals according to claim 6, characterized in that: The second-level capture implementation process of the pseudo code phase in step 4 is as follows: Step 4.1: Despread the Q-channel received signal based on the code synchronization position of the Q-channel short spreading code obtained in step 3, and perform differential decoding of the adjacent code elements before and after the despread signal to eliminate the initial carrier frequency offset; Step 4.2, performing a fast Fourier transform on the signal obtained after differential decoding in step 4.1 to convert the signal from the time domain to the frequency domain; Step 4.3: Perform fast Fourier transform on the Q-channel synchronization header and perform conjugate processing on the signal after transforming into the frequency domain; Step 4.4: multiply the frequency domain result obtained in step 4.2 by the conjugate frequency domain signal obtained in step 4.3, and then perform an inverse fast Fourier transform to restore the result to the time domain; In step 4.5, the time domain results obtained in step 4.4 are directly accumulated several times, and then the peak value of the accumulated results is detected to determine the pseudo code phase position corresponding to the Q-channel synchronization header, thereby achieving frame synchronization of the Q-channel received signal.

8. The method for hierarchical synchronous reception of weak spread spectrum signals according to claim 7, characterized in that: The step 5 is to perform preliminary frequency offset compensation on the received signal, and the implementation process is as follows: Step 5.1: directly use the second-level capture position of the pseudo code phase obtained in step 4 to extract the frequency offset information before modulo-taking the peak at the correlation peak position to perform coarse frequency offset estimation; Step 5.2: Based on the frequency offset information obtained in step 5.1, perform preliminary frequency offset compensation on the Q-channel received signal to reduce the range of the remaining frequency offset.

9. The method for hierarchical synchronous reception of weak spread spectrum signals according to claim 7, characterized in that: The step 6 is to perform fine frequency offset correction on the received signal, and the implementation process is as follows: Step 6.1: Based on the second-stage capture position of the pseudo code phase obtained in step 4, i.e., the frame synchronization position, frame synchronization is performed on the I-channel received signal. Combined with the locally generated I-channel long spreading code, the synchronized I-channel received signal is despread. Step 6.2: differentially decode the adjacent symbols before and after the despread I-channel received signal to eliminate the initial carrier frequency offset; Step 6.3: Use the I-channel synchronization header to perform a correlation operation on the I-channel received signal, perform peak detection on the obtained integration result, extract the frequency offset information from the correlation peak, and achieve fine frequency offset estimation; In step 6.4, based on the result of the fine frequency offset estimation, the received signal is finely frequency offset corrected to correct the residual frequency offset error and achieve high-precision synchronization of the received signal.

10. A hierarchical synchronous reception system for weak spread spectrum signals, comprising: The spread spectrum signal frame structure design module designs a spread spectrum signal frame structure based on a dual information rate transmission system. By loading two signals with different spreading code lengths onto the I and Q orthogonal channels, the independent transmission of the two signals is ensured. The spreading code length of the Q channel signal is smaller than that of the I channel signal. The frequency error range determination module analyzes and determines the tolerance range of the differential coherent integration results of the I and Q channel signals under different spread spectrum code lengths to the Doppler frequency offset; The pseudo code phase first-level capture module performs parallel code phase search based on fast Fourier transform on the Q-path short spread spectrum code, realizing the first-level capture of the pseudo code phase of the Q-path short spread spectrum code; The pseudo code phase secondary capture module despreads and dedifferentiates the Q-channel received signal based on the completion of the primary capture. It also uses the Q-channel synchronization header to perform FFT-based parallel code phase search to complete the secondary capture of the pseudo code phase and achieve frame synchronization of the Q-channel received signal. The coarse frequency offset estimation and compensation module uses the correlation peak position obtained by the second-stage capture of the Q-path pseudo-code phase to extract the frequency offset information before the peak is modulo-ed, i.e., the residual frequency offset information. It performs coarse frequency offset estimation through frequency domain analysis and performs preliminary frequency offset compensation on the received signal. The precise frequency offset estimation and compensation module despreads and dedifferentiates the I-channel received signal, performs correlation operations on the I-channel received signal in conjunction with the I-channel synchronization header, achieves precise frequency offset estimation through integration results, and performs fine frequency offset correction on the received signal to ensure high-precision signal synchronization.

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