A Coupled Channel Estimation Method for Aperture-Level Simultaneous Transceiver Systems

Abstract

This invention discloses a method for estimating the coupled channel of an aperture-level simultaneous transmit / receive system, comprising the following steps: S1. In a full-duplex simulated phased array system, given an azimuth scanning range of (-a, a), where a represents the maximum scanning angle, determine the variation law of self-interference combined power under different transmit beam directions; S2. Let the transmit beam direction be -a degrees and the receive beam direction be the normal direction, perform signal transmission and reception to obtain the combined signal; S3. Calculate the power of the combined self-interference signal; change the receive beamforming coefficient multiple times to obtain multiple self-interference combined power values; S4. Estimate the equivalent channel matrix including the influence of transmit beamforming coefficient, self-interference channel, and transmit / receive noise; S5. Based on the variation law of self-interference combined power under different beam directions, quickly calculate the equivalent channel for different directions. This invention considers the influence of system transmit and receive noise, and provides a method for quickly calculating the equivalent channel under different beam positions for beam scanning scenarios.

Description

Technical Field

[0001] This invention relates to the field of wireless communication, and in particular to a method for estimating coupled channel in an aperture-level simultaneous transmit / receive system. Background Technology

[0002] Simultaneous full-duplex technology allows for signal transmission and reception on the same time-frequency resources. Compared to half-duplex communication, it theoretically doubles spectral efficiency, effectively alleviating issues such as spectrum resource scarcity and electromagnetic equipment compatibility. With the rapid development of wireless communication technology, integrated phased arrays are widely used in civilian and military communications. In multi-antenna array transceiver scenarios, simultaneous full-duplex operation on the same frequency faces strong self-interference problems. Due to intensified near-field cross-coupling effects, the self-interference power generated by the transmitting array near the receiving array is significantly enhanced, reducing the spatial isolation between antennas. Existing research often utilizes known self-interference channel state information to adaptively weighted beamforming the transceiver arrays, thereby actively reducing the coupled self-interference power between the transceiver arrays. How to obtain self-interference channel state information has become the primary issue in spatial domain self-interference suppression. Furthermore, to cover a certain spatial range, the transceiver beam pointing is usually dynamically switched within a given angle range. How to suppress spatial domain self-interference in scenarios with dynamic beam scanning is also a problem that urgently needs to be solved. Summary of the Invention

[0003] The purpose of this invention is to overcome the shortcomings of the prior art and provide a method for estimating the coupled channel of an aperture-level simultaneous transmit and receive system. It takes into account the influence of system transmit noise and receive noise, and provides a method for quickly calculating the equivalent channel under different beam positions for beam scanning scenarios.

[0004] The objective of this invention is achieved through the following technical solution: a coupled channel estimation method for an aperture-level simultaneous transmit / receive system, comprising the following steps:

[0005] S1. In a full-duplex analog phased array system, assuming that the transmitting and receiving beams are scanning in the same horizontal plane, and given the azimuth scanning range as (-a, a), where a represents the maximum scanning angle, determine the variation law of self-interference combined power under different transmitting beam directions.

[0006] S2. Let the direction of the transmit beam be -a degrees and the direction of the receive beam be the normal. The desired signal is transmitted through the analog transmit array after being beamformed by the transmit beam. After being coupled to the analog receive array through the self-interference channel, the received signal of the analog receive array is combined after being beamformed by the receive beam to obtain the combined signal.

[0007] S3. The combined signal passes through the coupler and enters the power measurement module, which calculates the power of the self-interference signal after combining. The parameter control module changes the receiving beamforming coefficient multiple times to obtain multiple self-interference combined power values.

[0008] S4. Estimate the equivalent channel matrix, including the effects of transmit beamforming coefficients, self-interference channels, and transmit / receive noise;

[0009] S5. Based on the self-interference combining power variation law under different beam directions, quickly calculate the equivalent channel for different directions.

[0010] The beneficial effects of this invention are: this invention takes into account the influence of system transmit noise and receive noise, and at the same time, for beam scanning scenarios, it provides a method for quickly calculating the equivalent channel under different beam positions. Attached Figure Description

[0011] Figure 1 This is a schematic diagram of a beam scanning scenario for a full-duplex array transceiver system.

[0012] Figure 2 This is a schematic diagram of an analog array simultaneous transmit / receive system in power feedback mode;

[0013] Figure 3 This is a schematic diagram showing the relationship between the transmitted beam direction and the normalized power value. Detailed Implementation

[0014] The technical solution of the present invention will be further described in detail below with reference to the accompanying drawings, but the scope of protection of the present invention is not limited to the following description.

[0015] like Figures 1-2 The diagram shows a full-duplex simulated phased array system, assuming the simulated transmit array and simulated receive array are equipped with J antennas and K antennas, respectively. The signal emitted from the transmit array will interfere with the adjacent receive array. Throughout the beam scanning process of the transmit array, self-interference exists, but the interference power changes. Experiments show that when the transmit and receive beams point towards each other, the interference power increases; when the transmit and receive beams point away from each other, the interference power decreases.

[0016] This patent first proposes an algorithm based on received combining self-interference power feedback to estimate the equivalent self-interference channel. Then, utilizing experimentally derived data on the variation of feedback power with beam pointing, it presents a fast algorithm for calculating the equivalent channel at different beam positions, effectively reducing system complexity. If a power feedback-based channel estimation algorithm is used to estimate the equivalent self-interference channel after each beam switch, the required measurement complexity is K·J times, while the fast algorithm in this application has a measurement complexity of only K times.

[0017] A method for estimating the coupled channel of an aperture-level simultaneous transmit / receive system includes the following steps:

[0018] S1. In a full-duplex analog phased array system, assuming that the transmitting and receiving beams are scanning in the same horizontal plane, and given the azimuth scanning range as (-a, a), where a represents the maximum scanning angle, determine the variation law of self-interference combined power under different transmitting beam directions.

[0019] Assuming the transmitting and receiving beams scan in the same horizontal plane, the azimuth scanning range is (-a, a), where a represents the maximum scanning angle in degrees. The azimuth angle corresponding to the maximum coupling self-interference power is determined, typically by the relative angle between the transmitting and receiving beams.

[0020] The transmit beam is pointed in 1-degree increments from -a degrees to a degrees using conventional beamforming; the receive array is pointed in the normal direction using conventional beamforming. The normalized power curve as a function of the transmit beam pointing direction is measured and plotted. The curve reaches its maximum value of 1 at -a degrees transmit beam and 0 degrees receive beam. As the transmit beam scans from -a degrees to a degrees, the normalized power gradually decreases. The fluctuations in the curve are related to the transmit and receive beamwidths. This curve shows a strong correlation between the self-interference combined power and the transmit and receive beam pointing directions. If the transmit and receive beam pointing directions and gains remain constant, the coupled self-interference power will be relatively stable.

[0021] When the transmitting and receiving beams are close together, i.e., the transmitting beam points to -a degrees and the receiving beam points to 0 degrees, the self-interference combining power is at its maximum. This value is used as a reference to normalize the power values ​​at other angles.

[0022] Based on the test results, the variation law of self-interference combined power under different transmit beam directions is as follows: Figure 3 As shown:

[0023] As can be seen, when the transmit beam is scanned, the power of the self-interference combiner will follow... Figure 3 The pattern shown changes accordingly. Therefore, it is advisable to select a time when the transmitted beam direction is -a degrees and the received beam direction is normal to perform the equivalent channel measurement, corresponding to steps 3-19. Furthermore, using the measured channel and... Figure 3 Curve conversion of equivalent channel at any angle.

[0024] S2. Let the direction of the transmit beam be -a degrees and the direction of the receive beam be the normal. The desired signal is transmitted through the analog transmit array after being beamformed by the transmit beam. After being coupled to the analog receive array through the self-interference channel, the received signal of the analog receive array is combined after being beamformed by the receive beam to obtain the combined signal.

[0025] S201. In a full-duplex analog phased array system, the analog transmitting array and the analog receiving array are equipped with J antennas and K antennas, respectively. Let the transmitting beam point to -a degrees and the receiving beam point to the normal direction.

[0026] The desired signal is divided into J paths by a power divider. Each signal corresponds to an antenna in the analog transmission array. Each signal is weighted by different adjustable phase shifters and adjustable attenuators, then amplified by a power amplifier, and then transmitted to the analog transmission array for transmission.

[0027] The transmitted signal is represented as:

[0028] d(t)=w t x(t)+n t (t) (1)

[0029] Where x(t) represents the desired signal, n represents the transmitted beamforming factor. t (t) represents the emitted noise, and the covariance matrix of the noise is expressed as: η t Indicates the transmit signal-to-noise ratio;

[0030] S202. The transmitted signal is coupled to a nearby receiving array through a near-field self-interference channel, and the received signal is represented as follows:

[0031] y(t)=Hd(t)+s(t) (2)

[0032] in, Let represent the self-interference coupling channel, and s(t) represent the desired far-field received signal;

[0033] The signals received by each antenna in the analog receiver array are amplified by a low-noise amplifier, then weighted by different adjustable phase shifters and adjustable attenuators before being combined. The resulting combined signal is represented as:

[0034]

[0035] in, n represents the received beamforming coefficient. r (t) represents the received noise, and the covariance matrix of the noise is expressed as: Where, η r Indicates the received signal-to-noise ratio. I represents the variance of thermal noise. K This represents a K×K identity matrix.

[0036] S3. The combined signal passes through the coupler and enters the power measurement module, which calculates the power of the self-interference signal after combining. The parameter control module changes the receiving beamforming coefficient multiple times to obtain multiple self-interference combined power values.

[0037] S301. The combined signal passes through the coupler and enters the power measurement module. The power measurement module calculates the power of the self-interference signal after combining, which is expressed as:

[0038]

[0039] in,

[0040]

[0041] M r That is, the equivalent channel matrix, M r This includes the effects of transmit beamforming coefficients, self-interference channels, and transmit / receive noise. Estimation M r During this process, the self-interference channel needs to be static and the transmit beamforming coefficient needs to remain unchanged;

[0042] S302. By repeatedly changing the receiving beamforming coefficient through the parameter control module, multiple self-interference combining power values ​​are obtained, forming a power feedback matrix:

[0043] W H M r W = P (6)

[0044] Where P is the power feedback matrix, and W represents a matrix composed of K different receiving beamforming coefficients, i.e.:

[0045] W = [w r1 ,w r2 ,...,w rK (7)

[0046] Among them, w rk (k = 1, 2, ..., K) represents the k-th receiving beamforming coefficient, which will shape different w rk As w r When coupled into a receiver array, different combining power is obtained, w r The magnitude and phase of each complex number in the vector are adjusted by each adjustable attenuator and adjustable phase shifter; P is obtained by calculating the power of multiple combined paths. When W is invertible, the estimated value of the equivalent self-interference channel matrix is ​​expressed as:

[0047] M r =(W H ) -1 PW -1 (8).

[0048] S4. Estimate the equivalent channel matrix, including the effects of transmit beamforming coefficients, self-interference channels, and transmit / receive noise;

[0049] S401. Assume the receiving beam pointing steering vector is q. r ,right Perform null space decomposition to obtain X, where (·) H Let X denote the conjugate transpose of a matrix, and let X denote the null space spanned by a set of orthonormal bases, i.e., X = span{x1, x2, ..., xn}.K-1}

[0050] Let W = [q] r [X], obviously W is invertible;

[0051] W = [q] r Substituting X] into equation (8), we obtain the following using block matrix multiplication and matrix inversion:

[0052]

[0053] in,

[0054]

[0055] H A =X H M r X (11)

[0056] H b =X H M r q r (12)

[0057] This represents the matrix obtained by multiplying the equivalent self-interference channel matrix by the null space of the steering vector. H represents the vector obtained by multiplying the equivalent self-interference channel matrix, the null space of the steering vector, and the steering vector. A and H b This will be used as an intermediate variable in the subsequent calculation of the receiving beamforming coefficient; P0 represents the feedback self-interference combining power;

[0058] S402. Configure the transmit beamforming factor as w t The transmit beamforming factor remains constant during the measurement process, and the receive beamforming factor is configured as q. r The feedback self-interference combined power P0 is obtained; due to x k Taken from the null space, therefore q r Add any x k Or x i and x k All combinations can ensure that the gain of the desired beam pointing remains unchanged;

[0059] S403. Configure the receiving beamforming factor as x k Given k = 1, 2, ..., K-1, the feedback self-interference combining power P1 is obtained. k ;

[0060] S404. Configure the receiver beamforming factor as q. r +x kGiven k = 1, 2, ..., K-1, the feedback self-interference combining power is obtained.

[0061] S405. Configure the receiving beamforming factor as q. r +jx k j is the imaginary unit, k = 1, 2, ..., K-1, to obtain the feedback self-interference combining power P3. k ;

[0062] S406. Calculation and

[0063] S407. Configure the receiving beamforming factor as q. r +x i +x k Given k = 1, 2, ..., K-1, i = 1, 2, ..., K-1, i ≠ k, the feedback self-interference combining power is obtained.

[0064] S408. Configure the receiving beamforming factor as q. r +x i +jx k Given k = 1, 2, ..., K-1, i = 1, 2, ..., K-1, i ≠ k, the total power of the feedback self-interference is obtained.

[0065] S409. Calculation and

[0066] S410.H b It is a column vector, where the real and imaginary parts of the k-th element are respectively... and Therefore, H is calculated. b ;

[0067] S411. Matrix H A The real and imaginary parts of the element in the k-th row and i-th column (i≠k) are respectively and The diagonal element is P1 k Therefore, H is calculated. A ;

[0068] S412. Obtain P0, H A and H b Then, the equivalent self-interference channel M is estimated according to equation (9). r .

[0069] S5. Based on the self-interference combining power variation law under different beam directions, quickly calculate the equivalent channel for different directions.

[0070] Observation of equation (5) reveals that the equivalent self-interference channel M r It includes the transmit beamforming coefficient. When the transmit beam dynamically switches, M... r This will change accordingly. If the equivalent channel is measured once according to steps S403 to S412 every time the beam is switched, the measurement complexity is K·J times. This patent utilizes the self-interference combining power variation law under different beam directions obtained in step 2, and proposes a method to quickly calculate the equivalent channel under different directions according to the following steps. Only K measurements are needed to calculate the channel under any scanning angle.

[0071] S501. If the transmit beam direction is switched to φ and the transmit beamforming factor is w φ According to equation (5), the equivalent self-interference channel matrix will change;

[0072]

[0073] Use a tilde superscript to indicate the variables after switching the transmit beam;

[0074] Configure the receive beamforming factor as x k For k = 1, 2, ..., K-1, remeasure to obtain the feedback self-interference combined power.

[0075] S502. For any transmitted beam pointing φ, determine the corresponding normalized power factor β based on the measurement results in step S1;

[0076] For any transmitted beam pointing φ, based on the curve of the measured combined interference power changing with the transmitted beam angle, if φ is one of the measured angles, the corresponding normalized power factor β is determined directly based on the measurement result; if φ is between two measured angles, the curve is linearly interpolated to determine the corresponding normalized power factor β.

[0077] During the measurement process, except for step S501, the transmit and receive beam pointing and gain are maintained in all other steps. According to the curve of the measured combined interference power changing with the transmit beam angle, when the transmit and receive beam pointing and gain remain unchanged, the coupled self-interference power will remain stable. Therefore, in the measurement after the transmit beam is switched, only the receive beamforming factor needs to be remeasured. Feedback power at time The remaining power values ​​can be calculated by multiplying by the normalized power factor β;

[0078] After beam switching and The calculation is as follows:

[0079]

[0080] S503. It is a column vector, where the real and imaginary parts of the k-th element are respectively... and Therefore, the calculation yields...

[0081] S504. Matrix The real and imaginary parts of the element in the k-th row and i-th column (i≠k) are respectively and diagonal elements are Therefore, the calculation yields...

[0082] S505. Substituting the result into equation (8), and using block matrix multiplication and matrix inversion, we obtain:

[0083]

[0084] That is, the equivalent self-interference channel matrix pointing downwards for any transmitted beam is obtained.

[0085] In a complete feedback power measurement process S402~S409, a total of K needs to be measured. 2 Next, that is:

[0086] O = (K-1)(K-2) + 2(K-1) + K = K 2 (19)

[0087] In measurements after the transmit beam is switched, only the receive beamforming factor needs to be remeasured. Feedback power at time The remaining power values ​​can be calculated by multiplying by the normalized power factor β, thereby reducing the measurement complexity to K-1 times.

[0088] The above description represents preferred embodiments of the present invention. It should be understood that the present invention is not limited to the forms disclosed herein and should not be construed as excluding other embodiments. It can be used in other combinations, modifications, and environments, and can be altered within the scope of the concept described herein through the above teachings or related technical or knowledge. Modifications and variations made by those skilled in the art that do not depart from the spirit and scope of the present invention should be within the protection scope of the appended claims.

Claims

1. A coupled channel estimation method for an aperture-level simultaneous transmit / receive system, characterized in that: Includes the following steps: S1. In a full-duplex analog phased array system, assuming that the transmitting and receiving beams are scanning in the same horizontal plane, and given the azimuth scanning range as (-a, a), where a represents the maximum scanning angle, determine the variation law of self-interference combined power under different transmitting beam directions. S2. Let the direction of the transmit beam be -a degrees and the direction of the receive beam be the normal. The desired signal is transmitted through the analog transmit array after being beamformed by the transmit beam. After being coupled to the analog receive array through the self-interference channel, the received signal of the analog receive array is combined after being beamformed by the receive beam to obtain the combined signal. S3. The combined signal passes through the coupler and enters the power measurement module, which calculates the power of the self-interference signal after combining. By repeatedly changing the receiving beamforming coefficient through the parameter control module, multiple self-interference combining power values ​​are obtained. Step S3 includes: S301. The combined signal passes through the coupler and enters the power measurement module. The power measurement module calculates the power of the self-interference signal after combining, which is expressed as: Among them, w r M represents the received beamforming factor. r This is the equivalent channel matrix: M r This includes the effects of transmit beamforming coefficients, self-interference channels, and transmit / receive noise; M is estimated. r During this process, the self-interference channel needs to be static and the transmit beamforming coefficient needs to remain unchanged; H represents the self-interference coupling channel, w t η represents the transmitted beamforming coefficient. t η represents the transmit signal-to-noise ratio. r Indicates the received signal-to-noise ratio. I represents the variance of thermal noise. K Represents a K×K identity matrix; S302. By repeatedly changing the receiving beamforming coefficient through the parameter control module, multiple self-interference combining power values ​​are obtained, forming a power feedback matrix: W H M r W=P (3) Where P is the power feedback matrix, and W represents a matrix composed of K different receiving beamforming coefficients, i.e.: In=[in r1 ,In r2 ,...,In rK ] (4) Among them, w rk (k = 1, 2, ..., K) represents the k-th receiving beamforming coefficient, which will shape different w rk As w r When coupled into a receiver array, different combining power is obtained, w r The magnitude and phase of each complex number in the vector are adjusted by each adjustable attenuator and adjustable phase shifter; P is obtained by calculating the power of multiple combined paths. When W is invertible, the estimated value of the equivalent self-interference channel matrix is ​​expressed as: M r =(W H ) -1 PW -1 (5) S4. Estimate the equivalent channel matrix, including the effects of transmit beamforming coefficients, self-interference channels, and transmit / receive noise; Step S4 includes: S401. Assume the receiving beam pointing steering vector is q. r ,right Perform null space decomposition to obtain X, where (·) H Let X denote the conjugate transpose of a matrix, and let X denote the null space spanned by a set of orthonormal bases, i.e., X = span{x1, x2, ..., xn}. K-1 }, Let W = [q] r [X], W is reversible; W = [q] r Substituting X] into equation (8), we obtain the following using block matrix multiplication and matrix inversion: in, H A =X H M r X (8) H b =X H M r q r (9) This represents the matrix obtained by multiplying the equivalent self-interference channel matrix by the null space of the steering vector. H represents the vector obtained by multiplying the equivalent self-interference channel matrix, the null space of the steering vector, and the steering vector. A and H b This will be used as an intermediate variable in the subsequent calculation of the receiving beamforming coefficient; P0 represents the feedback self-interference combining power; S402. Configure the transmit beamforming factor as w t The transmit beamforming factor remains constant during the measurement process, and the receive beamforming factor is configured as q. r The feedback self-interference combined power P0 is obtained; due to x k Taken from the null space, therefore q r Add any x k Or x i and x k All combinations can ensure that the gain of the desired beam pointing remains unchanged; S403. Configure the receiving beamforming factor as x k Given k = 1, 2, ..., K-1, the feedback self-interference combining power P1 is obtained. k ; S404. Configure the receiving beamforming factor as q. r +x k Given k = 1, 2, ..., K-1, the feedback self-interference combining power is obtained. S405. Configure the receiving beamforming factor as q. r +jx k Where j is the imaginary unit, k = 1, 2, ..., K-1, the feedback self-interference combining power is obtained. S406. Calculation and S407. Configure the receiving beamforming factor as q. r +x i +x k Given k = 1, 2, ..., K-1, i = 1, 2, ..., K-1, i ≠ k, the feedback self-interference combining power is obtained. S408. Configure the receiving beamforming factor as q. r +x i +jx k Given k = 1, 2, ..., K-1, i = 1, 2, ..., K-1, i ≠ k, the total power of the feedback self-interference is obtained. S409. Calculation and S410.H b It is a column vector, where the real and imaginary parts of the k-th element are respectively... and Therefore, H is calculated. b ; S411. Matrix H A The real and imaginary parts of the element in the k-th row and i-th column (i≠k) are respectively and The diagonal element is P1 k Therefore, H is calculated. A ; S412. Obtain P0, H A and H b Then, the equivalent self-interference channel M is estimated according to equation (9). r ; S5. Based on the variation law of self-interference combining power under different beam directions, calculate the equivalent channel for different directions; Step S5 includes: S501. If the transmit beam direction is switched to φ and the transmit beamforming factor is w φ According to equation (5), the equivalent self-interference channel matrix will change; Use a tilde superscript to indicate the variables after switching the transmit beam; Configure the receive beamforming factor as x k For k = 1, 2, ..., K-1, remeasure to obtain the feedback self-interference combined power. S502. For any transmitted beam pointing φ, determine the corresponding normalized power factor β based on the measurement results in step S1; For any transmitted beam pointing φ, based on the curve of the measured combined interference power changing with the transmitted beam angle, if φ is one of the measured angles, the corresponding normalized power factor β is determined directly based on the measurement result; if φ is between two measured angles, the curve is linearly interpolated to determine the corresponding normalized power factor β. During the measurement process, except for step S501, the transmit and receive beam pointing and gain are maintained in all other steps. According to the curve of the measured combined interference power changing with the transmit beam angle, when the transmit and receive beam pointing and gain remain unchanged, the coupled self-interference power will remain stable. Therefore, in the measurement after the transmit beam is switched, only the receive beamforming factor needs to be remeasured. Feedback power at time The remaining power values ​​can be calculated by multiplying by the normalized power factor β; After beam switching and The calculation is as follows: S503. It is a column vector, where the real and imaginary parts of the k-th element are respectively... and Therefore, the calculation yields... S504. Matrix The real and imaginary parts of the element in the k-th row and i-th column (i≠k) are respectively and diagonal elements are Therefore, the calculation yields... S505. Substituting the result into equation (8), and using block matrix multiplication and matrix inversion, we obtain: That is, the equivalent self-interference channel matrix pointing downwards for any transmitted beam is obtained.

2. The coupled channel estimation method for an aperture-level simultaneous transmit / receive system according to claim 1, characterized in that: In step S1, the transmitting beam is scanned from -a degrees to a degrees in 1-degree increments; the receiving array is pointed in the normal direction, and a normalized power curve is measured and plotted as the transmitting beam direction changes. The curve reaches a maximum value of 1 when the transmitting beam is pointed at -a degrees and the receiving beam is pointed at 0 degrees. As the transmitting beam scans from -a degrees to a degrees, the normalized power gradually decreases. That is, when the transmitting beam is pointed at -a degrees and the receiving beam is pointed at 0 degrees, the self-interference combining power is the maximum. This value is used as a reference to normalize the power values ​​at other angles.

3. The coupled channel estimation method for an aperture-level simultaneous transmit / receive system according to claim 1, characterized in that: Step S2 includes: S201. In a full-duplex analog phased array system, the analog transmitting array and the analog receiving array are equipped with J antennas and K antennas, respectively. Let the transmitting beam point to -a degrees and the receiving beam point to the normal direction. The desired signal is divided into J paths by a power divider. Each signal corresponds to an antenna in the analog transmission array. Each signal is weighted by different adjustable phase shifters and adjustable attenuators, then amplified by a power amplifier, and then transmitted to the analog transmission array for transmission. The transmitted signal is represented as: d(t)=w t x(t)+n t (t) (16) Where x(t) represents the desired signal, n represents the transmitted beamforming factor. t (t) represents the emitted noise, and the covariance matrix of the noise is expressed as: η t Indicates the transmit signal-to-noise ratio; S202. The transmitted signal is coupled to a nearby receiving array through a near-field self-interference channel, and the received signal is represented as follows: y(t)=Hd(t)+s(t) (17) in, Let represent the self-interference coupling channel, and s(t) represent the desired far-field received signal; The signals received by each antenna in the analog receiver array are amplified by a low-noise amplifier, then weighted by different adjustable phase shifters and adjustable attenuators before being combined. The resulting combined signal is represented as: in, n represents the received beamforming coefficient. r (t) represents the received noise, and the covariance matrix of the noise is expressed as: Where, η r Indicates the received signal-to-noise ratio. I represents the variance of thermal noise. K This represents a K×K identity matrix.

4. The coupled channel estimation method for an aperture-level simultaneous transmit / receive system according to claim 1, characterized in that: The desired signal is obtained by passing the baseband digital transmission signal through a DAC and an up-conversion mixer; When a full-duplex analog phased array system receives signals, the signal output from the combiner also enters a down-conversion mixer and ADC through a coupler to undergo down-conversion processing and analog-to-digital conversion to obtain a baseband digital received signal.

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