A hybrid transmission line 3dB coupler and a design method thereof

Abstract

This invention discloses a hybrid transmission line 3dB coupler and its design method, comprising a first conductor layer, a first dielectric layer, a second conductor layer, a second dielectric layer, and a third conductor layer stacked sequentially. A first coupling line is disposed within the first conductor layer, a second coupling line is disposed within the second conductor layer, and a reference ground plane is disposed within the third conductor layer. The first and second coupling lines share the reference ground plane as their reference ground. Perpendicular to the stacking direction, the projections of the first and second coupling lines overlap, and the orthographic projection of the second coupling line completely falls within the orthographic projection of the first coupling line. This invention employs a shared reference ground and a full projection coverage structure, ensuring that the area of ​​the coupling lines facing each other is largely unaffected by PCB layer alignment errors, improving the consistency of mass production performance, while simplifying the stacking structure and reducing costs.

Description

Technical Field

[0001] This invention relates to the field of radio frequency and microwave technology, and in particular to a hybrid transmission line 3dB coupler and its design method. Background Technology

[0002] A 3dB bridge (90° quadrature coupler) is a core component in radio frequency circuits used to divide signal power equally (3dB) and generate a 90° phase difference. Currently, the main design schemes of 3dB bridges used on PCBs fall into two categories: branch-line couplers and electromagnetic field induction couplers. Branch-line couplers consist of multiple microstrip lines or striplines with different characteristic impedances. They offer advantages such as simple structure and good tolerance performance, but their bandwidth is relatively narrow (typically <20%) and their size is relatively large, often failing to meet the requirements of circuit operating bandwidth and PCB routing space in many applications.

[0003] Electromagnetic field induction couplers are divided into side couplers and wide-side couplers. Side couplers consist of two parallel coupling lines in the same plane. However, under conventional PCB manufacturing processes, the minimum conductor spacing needs to be greater than 0.07mm, making it difficult to achieve strong coupling. Even if mutual capacitance is enhanced by adding floating conductors, the two coupling lines still need to be as close as possible. Due to manufacturing errors, the coupling degree will vary greatly from batch to batch.

[0004] Wide-side couplers consist of two striplines located on different conductor planes, offering a large facing area and facilitating strong coupling. However, existing wide-side couplers have two drawbacks: first, they require four conductor layers, leading to complex PCB stack-up and increased costs; second, they are susceptible to interlayer alignment errors, resulting in large deviations in the facing area and significant batch-to-batch fluctuations in coupling strength. To mitigate the impact of alignment errors, current technologies attempt to stabilize the facing area by altering the shape of the coupling lines (such as cross traces). However, this does not completely eliminate the error's influence. The increased coupling energy from the larger area portion is not entirely equal to the decreased coupling energy from the smaller area portion, and the uneven phase distribution of the coupling energy leads to deterioration in port standing waves and other performance indicators such as directivity.

[0005] Especially in phased array antenna applications such as satellite communications, thousands of 3dB bridges need to be integrated. Currently, most solutions use narrow-bandwidth branch-line couplers integrated inside the PCB, leading to degraded performance at the bandwidth edges; while using wide-bandwidth couplers as separate components is too expensive. Therefore, there is an urgent need for a PCB-integrated coupler solution that can balance wide bandwidth, low cost, and high tolerance for alignment errors. Summary of the Invention

[0006] Therefore, the present invention provides a hybrid transmission line 3dB coupler and its design method to solve the aforementioned problems existing in the prior art.

[0007] To achieve the above objectives, in one aspect, the present invention provides a hybrid transmission line 3dB coupler, comprising:

[0008] A first conductor layer, a first dielectric layer, a second conductor layer, a second dielectric layer, and a third conductor layer are stacked sequentially.

[0009] A first coupling line is provided within the first conductor layer;

[0010] A second coupling line is provided within the second conductor layer;

[0011] A reference ground plane is provided within the third conductor layer;

[0012] The projection areas of the first coupling line and the second coupling line overlap in the direction perpendicular to the stacking arrangement, and the orthogonal projection area of ​​the second coupling line falls completely into the orthogonal projection area of ​​the first coupling line.

[0013] The first coupling line and the second coupling line both use the reference ground plane as the reference ground.

[0014] Furthermore, the first vertical distance from the plane containing the first coupling line to the reference ground plane is greater than the second vertical distance from the plane containing the second coupling line to the reference ground plane; and under the constraint that the 3dB directional coupler satisfies the preset odd-even mode impedance matching condition, the linewidth of the first coupling line is greater than the linewidth of the second coupling line.

[0015] Furthermore, the first coupling line is a microstrip line, the upper surface of which is exposed to air or covered with a solder resist layer, and the lower surface is in close contact with the first dielectric layer; the second coupling line is a stripline, the upper and lower surfaces of which are physically wrapped by the first dielectric layer and the second dielectric layer, respectively.

[0016] Furthermore, the physical dimension of the first vertical distance is equal to the sum of the thickness of the first dielectric layer and the thickness of the second dielectric layer; the physical dimension of the second vertical distance is equal to the thickness of the second dielectric layer.

[0017] Furthermore, the distributed parameters of the first coupling line and the distributed parameters of the second coupling line, under non-independent adjustment conditions, satisfy the target value of 3dB coupling odd-mode impedance; wherein, the odd-mode impedance of the first coupling line is equal to the odd-mode impedance of the second coupling line, and the even-mode impedance of the first coupling line is equal to the even-mode impedance of the second coupling line.

[0018] On the other hand, the present invention also provides a design method for a hybrid transmission line 3dB coupler, comprising:

[0019] Step S1: Based on the stacked structure of the outer microstrip line and the inner stripline, extract the linewidth of the outer microstrip line, the linewidth of the inner stripline, the thickness of the outer dielectric, and the thickness of the inner dielectric as multidimensional physical size variables.

[0020] Step S2: Apply step perturbations to the linewidth of the outer microstrip line and the linewidth of the inner stripline respectively, and extract the changes in the electromagnetic characteristic response of the inner stripline; when it is determined that the absolute value of the change in the electromagnetic characteristic response caused by the perturbation of the outer microstrip linewidth is greater than the absolute value of the change in the electromagnetic characteristic response caused by the perturbation of the inner stripline linewidth, a cross-layer dominant marker is generated.

[0021] Step S3: Replace the predefined odd-even mode impedance constraint target with a comprehensive evaluation function according to the cross-layer dominant marker. The comprehensive evaluation function is composed of the power distribution difference between the through end and the coupled end, the phase error value, and the port reflection coefficient weighted sum.

[0022] Step S4: Perform parameter traversal and calculate the comprehensive evaluation function value within the manufacturing tolerance boundary of the multidimensional physical dimension variable; when it is detected that the absolute value of the rate of change of the comprehensive evaluation function value of two adjacent traversals is less than the preset gradient threshold, calculate the first partial derivative of the outer layer medium thickness with respect to the phase error value and the second partial derivative of the inner layer medium thickness with respect to the phase error value respectively.

[0023] Step S5: When it is determined that the signs of the first partial derivative and the second partial derivative are opposite, the traversal step size of the outer layer medium thickness and the inner layer medium thickness is differentially reduced according to the proportional relationship between the absolute values ​​of the first partial derivative and the absolute values ​​of the second partial derivative. The parameter traversal after the reduction step size is executed until the value of the comprehensive evaluation function is lower than the preset target threshold, and the current combination of multidimensional physical size variables is output.

[0024] Furthermore, the process of step S3 includes:

[0025] Step S31, in response to the cross-layer dominant marker confirming that the independent decoupling space of the even and odd mode impedances has been distorted;

[0026] Step S32: Construct the comprehensive evaluation function including the first weighting coefficient, the second weighting coefficient, and the third weighting coefficient; wherein, the difference between the absolute values ​​of the transmission coefficients of the through end and the coupled end is used as the power allocation difference, the absolute value of the phase difference between the transmission of the through end and the coupled end deviating from the 90-degree orthogonal state is used as the phase error value, and the absolute value of the reflection coefficient of the input port is used as the port reflection coefficient;

[0027] Step S33: Based on the degree of interference of the outer edge electric field to the inner phase propagation constant represented by the cross-layer dominant marker, the first weighting coefficient corresponding to the phase error value is set to be greater than the second weighting coefficient corresponding to the power distribution difference, and greater than the third weighting coefficient corresponding to the port reflection coefficient.

[0028] Furthermore, the process of step S33 includes:

[0029] Under the current parameter combination for generating the cross-layer dominant marker, the first relative sensitivity of the outer microstrip line width to the inner stripline phase propagation constant and the second relative sensitivity of the inner stripline line width to the inner stripline phase propagation constant are calculated respectively, and the ratio of the two is used as the phase interference index.

[0030] The phase interference index is multiplied by a preset phase base weight to obtain the first relative weight requirement value; at the same time, preset base power weight values ​​and base reflection weight values ​​are respectively configured for the second weight coefficient and the third weight coefficient;

[0031] The first relative weight requirement value, the basic power weight value, and the basic reflection weight value are normalized to obtain the first weight coefficient, the second weight coefficient, and the third weight coefficient.

[0032] Compared with the prior art, the beneficial effects of the present invention are as follows: by sharing a reference ground structure, the vertical distance from the first coupling line to the reference ground is necessarily greater than that of the second coupling line. In order to meet the odd-even mode impedance matching conditions required for 3dB coupling, the first coupling line, which is farther away, must use a larger line width to compensate for the distributed capacitance to ground. This naturally forms a physical size difference where the width of the first coupling line is significantly greater than that of the second coupling line, thereby achieving a full-coverage structure where the projection of the second coupling line falls completely within the projection of the first coupling line. This causal relationship, from distance difference to line width compensation and then to projection coverage, ensures that the facing area of ​​the two coupling lines remains constant when there is an alignment deviation between PCB layers. This effectively eliminates the influence of alignment error on coupling degree. While achieving low-cost simplification with only three conductor layers, it greatly improves the performance consistency of mass manufacturing. Attached Figure Description

[0033] Figure 1 A three-dimensional exploded view of a hybrid transmission line 3dB coupler device provided by the present invention;

[0034] Figure 2 An XY plane perspective view of a hybrid transmission line 3dB coupler device provided by the present invention;

[0035] Figure 3An XZ-plane cross-sectional view of a hybrid transmission line 3dB coupler device provided by the present invention;

[0036] Figure 4 A flowchart illustrating a design method for a hybrid transmission line 3dB coupler provided by the present invention;

[0037] Reference numerals: 101, first conductor layer; 102, second conductor layer; 103, third conductor layer; 201, first dielectric layer; 202, second dielectric layer; 301, first coupling line; 302, second coupling line; 303, reference ground plane. Detailed Implementation

[0038] To make the objectives and advantages of the present invention clearer, the present invention will be further described below with reference to embodiments; it should be understood that the specific embodiments described herein are merely for explaining the present invention and are not intended to limit the present invention.

[0039] Preferred embodiments of the present invention will now be described with reference to the accompanying drawings. Those skilled in the art should understand that these embodiments are merely illustrative of the technical principles of the present invention and are not intended to limit the scope of protection of the present invention.

[0040] It should be noted that in the description of this invention, the terms "upper", "lower", "left", "right", "inner", "outer", etc., which indicate directions or positional relationships, are based on the directions or positional relationships shown in the accompanying drawings. This is only for the convenience of description and is not intended to indicate or imply that the device or element must have a specific orientation, or be constructed and operated in a specific orientation. Therefore, it should not be construed as a limitation of this invention.

[0041] Furthermore, it should be noted that, in the description of this invention, unless otherwise explicitly specified and limited, the terms "installation," "connection," and "linking" should be interpreted broadly. For example, they can refer to a fixed connection, a detachable connection, or an integral connection; they can refer to a mechanical connection or an electrical connection; they can refer to a direct connection or an indirect connection through an intermediate medium; and they can refer to the internal connection of two components. Those skilled in the art can understand the specific meaning of the above terms in this invention according to the specific circumstances.

[0042] Please see Figure 1 , Figure 2 and Figure 3 As shown, the present invention provides a hybrid transmission line 3dB coupler, comprising:

[0043] It includes a first conductor layer 101, a first dielectric layer 201, a second conductor layer 102, a second dielectric layer 202, and a third conductor layer 103, which are stacked sequentially.

[0044] A first coupling line 301 is provided within the first conductor layer 101;

[0045] A second coupling line 302 is provided within the second conductor layer 102;

[0046] A reference ground plane is provided within the third conductor layer 103;

[0047] The projection areas of the first coupling line 301 and the second coupling line 302 in the direction perpendicular to the stacking arrangement overlap, and the orthographic projection area of ​​the second coupling line 302 falls completely into the orthographic projection area of ​​the first coupling line 301.

[0048] The first coupling line 301 and the second coupling line 302 both use the reference ground plane 303 as the reference ground.

[0049] In this embodiment, the hybrid transmission line 3dB coupler is manufactured based on conventional PCB technology, and a first conductor layer, a first dielectric layer, a second conductor layer, a second dielectric layer, and a third conductor layer are sequentially stacked along a direction perpendicular to the board surface.

[0050] Specifically, a first coupling line is provided within the first conductor layer. Since the first conductor layer is located on the outermost layer of the PCB, the first coupling line is exposed to air or covered with a solder mask layer above it, while it is in close contact with the first dielectric layer below it. Therefore, the first coupling line constitutes a microstrip line structure.

[0051] A second coupling line is disposed within the second conductor layer. The second conductor layer is located between the first dielectric layer and the second dielectric layer, such that the upper surface of the second coupling line is wrapped by the first dielectric layer and the lower surface is wrapped by the second dielectric layer, thus forming a stripline structure. Therefore, the present invention employs a "hybrid transmission line structure" combining an outer microstrip line and a sub-outer stripline.

[0052] A reference ground plane is provided within the third conductor layer. Unlike traditional wide-side couplers that require separate ground planes above and below the two coupling lines, this invention eliminates the upper ground plane, allowing the first and second coupling lines to share the reference ground plane within the third conductor layer as their unique and common reference ground. This three-layer conductor structure reduces the number of conductor layers required compared to traditional four-layer wide-side couplers, effectively lowering the complexity of the PCB stack-up structure and manufacturing costs.

[0053] In terms of physical dimensions and spatial position, since the vertical distance of the first coupling line from the reference ground plane (i.e., the sum of the thicknesses of the first and second dielectric layers) is significantly greater than the vertical distance of the second coupling line from the reference ground plane (i.e., the thickness of the second dielectric layer), in order for the first and second coupling lines to meet the specific odd-mode impedance matching conditions required by the 3dB coupler (e.g., odd-mode impedance approximately 20.7 ohms, even-mode impedance approximately 120.7 ohms) in the wide-side coupling state, it is necessary to design the linewidth of the first coupling line to be greater than that of the second coupling line. Based on the inevitable result that the linewidth of the first coupling line is significantly greater than that of the second coupling line, further improvements can be achieved by reasonably designing the thicknesses of the first and second dielectric layers:

[0054] The first coupling line and the second coupling line overlap in their projection areas on a horizontal plane perpendicular to the stacking direction, forming a wide-edge coupling area. The orthographic projection area of ​​the second coupling line completely falls within the orthographic projection area of ​​the first coupling line, creating a "top-and-bottom" enveloping structure. In conventional PCB manufacturing, an interlayer alignment error of 2 mil to 3 mil (approximately 0.05 mm to 0.08 mm) is unavoidable. Since the projection of the second coupling line is completely covered by the first coupling line, the facing area of ​​the two coupling lines remains essentially unchanged when horizontal misalignment occurs. This significantly reduces the impact of interlayer alignment error on performance indicators such as coupling degree, improves the performance consistency during coupler mass production, and reduces the application risks of directly integrating a 3dB bridge into the PCB.

[0055] Furthermore, since the hybrid transmission line structure described in this invention breaks the characteristic that the physical parameters of traditional symmetrical wide-side couplers can be adjusted relatively independently, the distributed parameters of its transmission lines (such as self-capacitance, mutual capacitance, self-inductance, etc.) are highly correlated. Adjusting one part affects the whole. Therefore, the specific values ​​of the linewidth of the first coupling line, the linewidth of the second coupling line, the thickness of the first dielectric layer, and the thickness of the second dielectric layer need to be determined by setting a comprehensive error function that includes multiple dimensions such as port standing wave ratio, isolation, amplitude balance, and phase balance. The final result is obtained by using electromagnetic field numerical simulation software within the physical boundary to explore and optimize the parameters, so as to obtain the optimal size combination that can take into account both bandwidth and processing tolerance.

[0056] Specifically, the first vertical distance from the plane where the first coupling line 301 is located to the reference ground plane 303 is greater than the second vertical distance from the plane where the second coupling line 302 is located to the reference ground plane 303; and under the constraint that the 3dB directional coupler satisfies the preset odd-even mode impedance matching condition, the line width of the first coupling line 301 is greater than the line width of the second coupling line 302.

[0057] Specifically, according to the design theory of 3dB directional couplers, to ensure the power division ratio and phase quadrature characteristics of the coupler, a preset odd-mode impedance matching condition must be met. That is, the odd-mode impedance of the first coupling line must be equal to the odd-mode impedance of the second coupling line (for example, in a 50-ohm system, the odd-mode impedance should be approximately 20.7 ohms and the even-mode impedance should be approximately 120.7 ohms). However, since the first vertical distance (H1) is much larger than the second vertical distance (H2), and the first coupling line is a microstrip line (the electromagnetic field is partially exposed to the air, resulting in a lower equivalent dielectric constant), while the second coupling line is a stripline line (the electromagnetic field is completely enclosed by a dielectric, resulting in a higher equivalent dielectric constant), these two factors cause the distributed capacitance of the first coupling line to the reference ground to be naturally much smaller than that of the second coupling line. If the linewidths of the two are the same, the characteristic impedance of the first coupling line will be much higher than that of the second coupling line, failing to meet the matching condition of equal odd-mode impedance. To enforce impedance matching constraints under the asymmetric structure H1 >> H2, it is necessary to compensate for the high impedance caused by the large distance to ground and low dielectric constant of the first coupling line by increasing its edge capacitance and capacitance to ground. Since the linewidth of the transmission line is positively correlated with the distributed capacitance to ground and negatively correlated with the distributed inductance, the linewidth (W1) of the first coupling line must be designed to be significantly larger than that of the second coupling line (W2). This wider conductor area lowers the impedance of the first coupling line, making it consistent with the impedance of the narrower, closer-distance second coupling line. In practical design, due to the strong coupling relationship between H1, H2, W1, W2, and the dielectric distribution parameters, the qualitative principle of "large linewidth compensating for large distance" needs to be transformed into a quantitative combination of dimensions. By setting the target odd-even mode impedance value of 3dB coupling as the constraint boundary, the global collaborative solution of multi-dimensional parameters of the first dielectric layer thickness, the second dielectric layer thickness, W1 and W2 is performed (for example, by using electromagnetic field numerical simulation to explore and optimize). Under the geometric premise that H1 > H2, the unique combination of W1 and W2 that makes Zo1=Zo2 and Ze1=Ze2 is found, thereby ensuring that the linewidth of the first coupling line is strictly greater than the linewidth of the second coupling line, and providing a dimensional basis for further realizing the "sky-over-earth" covering structure of the projection.

[0058] Specifically, the first coupling line 301 is a microstrip line, the upper surface of which is exposed to air or covered with a solder resist layer, and the lower surface is in close contact with the first dielectric layer 201; the second coupling line 302 is a stripline, the upper and lower surfaces of which are physically wrapped by the first dielectric layer 201 and the second dielectric layer 202, respectively.

[0059] Specifically, in conventional PCB manufacturing processes, the first conductor layer is located on the outermost side of the PCB (e.g., the top layer), thus forming a microstrip line structure. The electromagnetic field on its upper surface partially leaks into the air or solder mask (dielectric constant typically between 3 and 4), while the electromagnetic field on its lower surface is concentrated within the first dielectric layer (dielectric constant typically between 4 and 5). This semi-open field distribution results in a relatively low equivalent dielectric constant. In contrast, the second conductor layer is located between the first and second dielectric layers, making the second coupling line an inner-layer stripline structure. The electromagnetic fields on both its upper and lower surfaces are completely physically encapsulated by the dielectric material, with no air interface. Therefore, its equivalent dielectric constant is significantly higher than that of the first coupling line, and the electromagnetic field distribution is more concentrated and uniform.

[0060] Specifically, the physical dimension of the first vertical distance is equal to the sum of the thickness of the first dielectric layer 201 and the thickness of the second dielectric layer 202; the physical dimension of the second vertical distance is equal to the thickness of the second dielectric layer 202.

[0061] Specifically, the vertical distance from the plane containing the first coupling line to the reference ground plane is defined as the first vertical distance (H1), and the vertical distance from the plane containing the second coupling line to the reference ground plane is defined as the second vertical distance (H2). In the three-layer conductor stack-up architecture of the present invention, since the first coupling line is disposed above the first dielectric layer, the second coupling line is disposed between the first dielectric layer and the second dielectric layer, and the reference ground plane is disposed below the second dielectric layer, there is a strict dimensional mapping relationship in geometric position: the physical dimension of the first vertical distance H1 is equal to the sum of the thickness of the first dielectric layer and the thickness of the second dielectric layer; the physical dimension of the second vertical distance H2 is equal to the thickness of the second dielectric layer. Based on the above geometric mapping relationship, a key physical feature of the structure of the present invention can be further derived: the coupling spacing (denoted as Hc) between the first coupling line and the second coupling line in the vertical direction is calculated as Hc = H1 - H2, that is, the physical dimension of the coupling spacing Hc is exactly equal to the thickness of the first dielectric layer.

[0062] In a traditional four-layer wide-side coupler, the coupling spacing and the distance from each coupled line to the reference ground are relatively independently adjustable variables. However, in the hybrid transmission line structure of this invention, attempting to adjust the mutual capacitance between two coupled lines by changing the coupling spacing Hc (i.e., the thickness of the first dielectric layer) will inevitably lead to an equal change in the first vertical distance H1, thereby causing a drastic change in the self-capacitance of the first coupled line. This deep coupling in the vertical geometric dimensions means that adjusting the dielectric layer thickness has a ripple effect. Therefore, this invention uses electromagnetic field numerical simulation software to perform global joint exploration and optimization of all dimensional parameters within the set physical parameter boundaries, rather than iteratively designing using analytical formulas.

[0063] Specifically, the distributed parameters of the first coupling line 301 and the distributed parameters of the second coupling line 302, under non-independent adjustment conditions, satisfy the target value of 3dB coupling odd-mode impedance; wherein, the odd-mode impedance of the first coupling line 301 is equal to the odd-mode impedance of the second coupling line 302, and the even-mode impedance of the first coupling line 301 is equal to the even-mode impedance of the second coupling line 302.

[0064] Specifically, based on the design theory of the 3dB directional coupler with coupling coefficient k=0.707 and port output impedance Z0=50 ohms, the target values ​​of the odd-mode impedances must satisfy the following: the odd-mode impedance Zo1 of the first coupling line and the odd-mode impedance Zo2 of the second coupling line are both approximately 20.7 ohms, and the even-mode impedance Ze1 of the first coupling line and the even-mode impedance Ze2 of the second coupling line are both approximately 120.7 ohms. Since this invention employs a hybrid transmission line and common ground structure, the distance (H1) of the first coupling line from the ground is significantly greater than the distance (H2) of the second coupling line from the ground, and the effective dielectric constant of the first coupling line (microstrip line) is lower than that of the second coupling line (strip line). This results in the first coupling line having a naturally much smaller distributed capacitance to ground than the second coupling line. In a non-independent state where distributed parameters cannot be independently adjusted, to achieve the same odd-mode impedance, compensation must be made through extreme asymmetry in physical dimensions: that is, the linewidth (W1) of the first coupling line is set to be significantly larger than the linewidth (W2) of the second coupling line, thereby increasing the edge capacitance and ground capacitance of the first coupling line, so that the two can rebalance in terms of distributed parameters to ground. In a non-independent adjustment state, adjusting any physical parameter (W1, W2, H1, H2) will trigger a chain of nonlinear changes in self-capacitance, mutual capacitance, and self-inductance, making it impossible to simultaneously achieve Zo1=Zo2 and Ze1=Ze2 using conventional analytical formulas or iterative methods. Therefore, satisfying the above-mentioned equal odd-mode impedance relationship requires transforming the impedance consistency target (or its equivalent electrical performance indicators, such as port standing wave ratio, isolation, amplitude and phase balance) into a comprehensive error function, and using electromagnetic field numerical simulation to conduct multi-dimensional global trial and optimization within the manufacturing tolerance boundaries of the physical parameters. By continuously adjusting parameter boundaries and optimization strategies, we can find the unique combination of physical dimensions (e.g., W1=0.8mm, W2=0.45mm, H1=0.12mm, H2=0.5mm) that makes the difference between the odd and even modes of the first and second coupling lines converge to near zero, thereby achieving the impedance matching target of 3dB coupling under the constraint of non-independent adjustment.

[0065] Specifically, this invention, through a shared reference ground structure, ensures that the vertical distance from the first coupling line to the reference ground is necessarily greater than that of the second coupling line. To meet the odd-even mode impedance matching condition required for 3dB coupling, the first coupling line, which is farther away, must use a larger linewidth to compensate for the distributed capacitance to ground. This naturally results in a physical size difference where the width of the first coupling line is significantly greater than that of the second coupling line, thereby achieving a full-coverage structure where the projection of the second coupling line completely falls within the projection of the first coupling line. This causal relationship, derived from the distance difference to linewidth compensation and then to projection coverage, ensures that the facing area of ​​the two coupling lines remains constant even when there is an alignment deviation between PCB layers. This effectively eliminates the impact of alignment error on coupling degree, achieving low-cost simplification with only three conductor layers while greatly improving the performance consistency of mass production.

[0066] Specifically, such as Figure 4 As shown, an optimization design method for a hybrid transmission line 3dB coupler includes:

[0067] Step S1: Based on the stacked structure of the outer microstrip line and the inner stripline, extract the linewidth of the outer microstrip line, the linewidth of the inner stripline, the thickness of the outer dielectric, and the thickness of the inner dielectric as multidimensional physical size variables.

[0068] Specifically, based on the three-layer conductor stack structure of the 3dB coupler, under the structural constraints of eliminating the intermediate shielding ground layer and forcibly setting the projection of the inner layer stripline in the horizontal plane to completely fall within the projection area of ​​the outer layer microstrip line, the horizontal overlap width and vertical interlayer spacing of the coupling line are determined to be non-independent variables. Specifically, the horizontal overlap width is locked to be equal to the linewidth of the inner layer stripline, and the vertical interlayer spacing is locked to be the difference between the outer layer dielectric thickness and the inner layer dielectric thickness. The four independent variables that uniquely determine the odd-even mode impedance distribution parameters of the coupling line are extracted as the multidimensional physical size variables, namely: outer layer microstrip linewidth, inner layer stripline linewidth, outer layer dielectric thickness, and inner layer dielectric thickness. Physical boundary constraints based on conventional PCB manufacturing processes are assigned to the multidimensional physical size variables, specifically including: obtaining a preset minimum linewidth process threshold, and... The lower limits of the outer microstrip line width and the inner stripline line width are both constrained to be greater than or equal to the minimum line width process threshold; wherein, the minimum line width process threshold represents the minimum conductor width limit achievable by conventional PCB etching processes, and its value ranges from 0.07mm to 0.1mm; a preset minimum dielectric thickness process threshold is obtained, and the lower limits of the outer dielectric thickness and the inner dielectric thickness are both constrained to be greater than or equal to the minimum dielectric thickness process threshold; wherein, the minimum dielectric thickness process threshold represents the minimum physical spacing to prevent short circuits between adjacent conductor layers and to meet the requirements of the board lamination process, and its value ranges from 0.1mm to 0.12mm; based on the fact that the projection in the horizontal plane completely falls within the constraint conditions, the optimization lower limit value of the outer microstrip line width is forcibly limited to be greater than the optimization upper limit value of the inner stripline line width.

[0069] Step S2: Apply step perturbations to the linewidth of the outer microstrip line and the linewidth of the inner stripline respectively, and extract the changes in the electromagnetic characteristic response of the inner stripline; when it is determined that the absolute value of the change in the electromagnetic characteristic response caused by the perturbation of the outer microstrip linewidth is greater than the absolute value of the change in the electromagnetic characteristic response caused by the perturbation of the inner stripline linewidth, a cross-layer dominant marker is generated.

[0070] Specifically, step S2 includes the following process:

[0071] Step S21: Apply a preset first equal step perturbation to the linewidth of the outer microstrip line, extract the first change in the effective dielectric constant of the inner stripline, and then obtain the first relative change rate.

[0072] Specifically, the process of applying a preset first equal-step perturbation to the linewidth of the outer microstrip line includes:

[0073] Obtain the equivalent broadening of the edge electric field of the current outer microstrip line at the air-dielectric interface;

[0074] Specifically, the equivalent broadening of the edge electric field is a parameter used to characterize the physical width of the concentrated region of the edge electric field extending from the outer microstrip line into free space. In actual simulation extraction, those skilled in the art can directly calculate the equivalent broadening of the edge electric field based on the ratio of the current outer microstrip line width (W1) to the outer dielectric thickness (H1) using well-known microstrip line synthesis formulas (such as static approximation models).

[0075] The preset multiple of the equivalent broadening of the edge electric field is set to the first equal step perturbation amount;

[0076] The equivalent broadening of the edge electric field is a parameter that characterizes the physical width of the edge electric field concentration region extending from the outer microstrip line to the free space side, calculated based on the current linewidth of the outer microstrip line and the thickness of the outer dielectric.

[0077] Specifically, the preset multiple of the calculated equivalent broadening of the edge electric field is set as the first equal step perturbation amount. For example, in a typical Ku-band coupler design, if the calculated equivalent broadening of the edge electric field is 0.1 mm, the preset multiple can be set between 0.1 and 0.2.

[0078] Specifically, in the electromagnetic field simulation model, the inner stripline linewidth, outer dielectric thickness, and inner dielectric thickness—all multidimensional physical size variables—are first locked at their initial values. Then, a small, pre-defined first equal-step perturbation (e.g., increasing the outer microstrip linewidth by 1% from its current value) is applied to the outer microstrip linewidth. Since the inner stripline in this invention is completely projected and covered by the outer microstrip line and shares the same reference ground, the change in the outer microstrip linewidth directly reshapes the electromagnetic field boundary distribution above the inner stripline, resulting in a drastic change in the self-capacitance of the inner stripline. In the simulation software, the electromagnetic characteristic response of the inner stripline before and after the perturbation—the effective dielectric constant (which is directly determined by the capacitance distribution around the transmission line)—is extracted. The difference between the two is calculated and divided by the initial value of the effective dielectric constant before the perturbation, thus obtaining the first relative rate of change to eliminate absolute dimension interference.

[0079] Step S22: Apply a preset second equal step perturbation to the linewidth of the inner layer stripline, extract the second change amount corresponding to the effective dielectric constant of the inner layer stripline, and then obtain the second relative change rate;

[0080] Specifically, using the same control variable logic, the linewidth of the outer microstrip line is restored, and then the linewidth of the inner stripline is subjected to a second equal step perturbation of the same proportion. Similarly, the second relative rate of change of the effective dielectric constant of the inner stripline caused by its own linewidth perturbation is extracted and calculated.

[0081] Step S23: The ratio of the first relative rate of change to the second relative rate of change is used as the cross-layer intervention coefficient;

[0082] Specifically, in traditional stripline or symmetrical wide-side coupler designs, changing the linewidth of the lineline inevitably has the greatest impact on its electromagnetic environment, so this coefficient is usually less than or close to 1; however, in the hybrid stacked structure of this invention, since the outer microstrip line is not only the coupling arm but also constitutes the main electromagnetic field boundary of the inner stripline, this coefficient will be reversed.

[0083] Step S24: When the value of the cross-layer intervention coefficient is determined to be greater than 1, it is confirmed that the control of the outer layer lateral geometric disturbance over the inner layer electromagnetic phase velocity is higher than the control of the inner layer lateral geometry itself, so as to generate the cross-layer dominant marker.

[0084] Specifically, the system determines whether the value of the aforementioned cross-layer interference coefficient is greater than a preset dominant judgment threshold (i.e., a value of 1). When the cross-layer interference coefficient is detected to be greater than 1, the outer microstrip line undergoes only a 1% lateral geometric change. Its interference on the phase velocity of the inner stripline (determined by the effective dielectric constant) even exceeds the impact of a 1% lateral geometric change in the inner stripline itself. At this point, it is confirmed that the control of the outer layer's lateral geometric disturbance over the inner layer's electromagnetic phase velocity has substantially surpassed the control of the inner layer's lateral geometry itself, and the theoretical premise of "independent decoupling and separate tuning of parameters in each layer" in traditional RF circuits completely fails. The system then generates a cross-layer dominant flag accordingly.

[0085] Step S3: Replace the predefined odd-even mode impedance constraint target with a comprehensive evaluation function according to the cross-layer dominant marker. The comprehensive evaluation function is composed of the power distribution difference between the through end and the coupled end, the phase error value, and the port reflection coefficient weighted sum.

[0086] Specifically, step S3 includes the following process:

[0087] Step S31, in response to the cross-layer dominant marker confirming that the independent decoupling space of the even and odd mode impedances has been distorted;

[0088] Specifically, in conventional symmetrical stripline coupler designs, designers typically use predefined odd-mode impedances (e.g., 20.7 ohms and 120.7 ohms for a 3dB coupler) as hard constraints, working backward from formulas to derive physical dimensions. However, after generating the cross-layer dominant marker, the system confirms that the independent decoupling space of odd-mode impedances in this hybrid structure has been severely distorted. That is, because any fine-tuning of the outer layer parameters triggers a cascading collapse of the inner layer's distributed capacitance and inductance, there is simply no independent geometric solution within the design space that can simultaneously and precisely lock the two unequal-width transmission lines to the theoretical odd-mode impedance values.

[0089] Step S32: Construct the comprehensive evaluation function including the first weighting coefficient, the second weighting coefficient, and the third weighting coefficient; wherein, the difference between the absolute values ​​of the transmission coefficients of the through end and the coupled end is used as the power allocation difference, the absolute value of the phase difference between the transmission of the through end and the coupled end deviating from the 90-degree orthogonal state is used as the phase error value, and the absolute value of the reflection coefficient of the input port is used as the port reflection coefficient;

[0090] Specifically, the power distribution difference is used to reflect the amplitude imbalance of the 3dB bridge, the phase error value determines the degree of axial ratio degradation of the circularly polarized antenna, and the port reflection coefficient characterizes the signal reflection loss (i.e., standing wave) at the input end.

[0091] Step S33: Based on the degree of interference of the outer edge electric field to the inner phase propagation constant represented by the cross-layer dominant marker, the first weighting coefficient corresponding to the phase error value is set to be greater than the second weighting coefficient corresponding to the power distribution difference, and greater than the third weighting coefficient corresponding to the port reflection coefficient.

[0092] Specifically, step S33 includes the following processes:

[0093] Under the current parameter combination for generating the cross-layer dominant marker, the first relative sensitivity of the outer microstrip line width to the inner stripline phase propagation constant and the second relative sensitivity of the inner stripline line width to the inner stripline phase propagation constant are calculated respectively, and the ratio of the two is used as the phase interference index.

[0094] Specifically, under the current combination of multidimensional physical size variables that generate the cross-layer dominant marker (e.g., when the current outer layer linewidth, inner layer linewidth, and dielectric thickness are determined), the phase propagation constant of the inner layer stripline is extracted using electromagnetic field simulation software. Then, the two relative sensitivities were calculated separately:

[0095] First relative sensitivity: Applying only a small perturbation (e.g., an increase of 0.5%) to the linewidth of the outer microstrip line, calculate the ratio of the rate of change of the inner stripline's phase propagation constant to the rate of change of the outer microstrip linewidth, i.e. ,in, ∆W1 represents the change in the phase propagation constant of the inner microstrip line, W1 represents the current initial value of the outer microstrip linewidth, and ∆W1 represents the change in the outer microstrip linewidth. This value characterizes the cross-boundary influence of changes in the outer layer geometry on the inner layer phase.

[0096] Second relative sensitivity: Applying only a small perturbation of the same proportion to the linewidth of the inner stripline, calculate the ratio of the rate of change of the phase propagation constant of the inner stripline to the rate of change of the linewidth of the inner stripline, i.e. Where W2 represents the current initial value of the inner stripline width, and ∆W2 represents the change in the inner stripline width. This value characterizes the normal control exerted by changes in the inner stripline geometry on its own phase.

[0097] The phase interference index is multiplied by a preset phase base weight to obtain the first relative weight requirement value; at the same time, preset base power weight values ​​and base reflection weight values ​​are respectively configured for the second weight coefficient and the third weight coefficient;

[0098] Specifically, a preset phase base weight (e.g., set to 1.0 as a baseline) is obtained, and the phase interference index calculated above is multiplied by this phase base weight to obtain the first relative weight requirement value. If the interference of the outer layer on the inner layer's phase is three times the inner layer's own control force (with an exponent of 3), then during the optimization process, the algorithm's "attention" or "penalty" for phase errors must also be amplified by three times to counteract this physical distortion. Simultaneously, preset base power weight values ​​and base reflection weight values ​​are configured for the second weight coefficient corresponding to the power allocation difference and the third weight coefficient corresponding to the port reflection coefficient (e.g., both set to 1.0 to maintain a normal level of attention).

[0099] The first relative weight requirement value, the basic power weight value, and the basic reflection weight value are normalized to obtain the first weight coefficient, the second weight coefficient, and the third weight coefficient.

[0100] Specifically, since the first relative weight requirement value is amplified by the interference index, directly substituting it will lead to an imbalance in the numerical scale of the comprehensive evaluation function. Therefore, the first relative weight requirement value, the basic power weight value, and the basic reflection weight value need to be normalized (i.e., divided by the sum of the three respectively) so that the sum of the first weight coefficient, the second weight coefficient, and the third weight coefficient in the final output is always equal to 1 (or 100%).

[0101] In a specific embodiment, assuming that in a certain optimization iteration, the first relative sensitivity is measured to be 0.6 and the second relative sensitivity is 0.2, then the phase interference index = 0.6 / 0.2 = 3.

[0102] Set the phase base weight, base power weight, and base reflection weight to 1.0.

[0103] Therefore, the first relative weight requirement value = 3 × 1.0 = 3.

[0104] Summarizing the demand values ​​(3+1+1=5) and normalizing them, we finally obtain:

[0105] First weighting coefficient (phase error) = 3 / 5 = 60%;

[0106] Second weighting factor (power allocation) = 1 / 5 = 20%;

[0107] The third weighting factor (port reflection) = 1 / 5 = 20%.

[0108] Step S4: Perform parameter traversal and calculate the comprehensive evaluation function value within the manufacturing tolerance boundary of the multidimensional physical dimension variable; when it is detected that the absolute value of the rate of change of the comprehensive evaluation function value of two adjacent traversals is less than the preset gradient threshold, calculate the first partial derivative of the outer layer medium thickness with respect to the phase error value and the second partial derivative of the inner layer medium thickness with respect to the phase error value respectively.

[0109] Specifically, step S4 includes the following process:

[0110] Step S41: When it is detected that the gradient of the change of the comprehensive evaluation function value between adjacent traversal steps is less than the preset gradient threshold, independent micro-perturbations are applied to the outer layer medium thickness and the inner layer medium thickness respectively using the control variable method within the manufacturing tolerance boundary.

[0111] The preset gradient threshold is a critical criterion used to determine whether the optimization algorithm has fallen into a local minimum or an invalid oscillation. It is used to stop losses in a timely manner and avoid wasting computing power on invalid traversals that do not produce substantial performance improvements. In the RF parameter simulation context of this invention, because the objective function is extremely sensitive to size, this threshold is usually set to a very small order of magnitude, such as 1×10⁻⁶. −3 Up to 5×10 −4 .

[0112] Specifically, step S41 includes the following process:

[0113] The first tolerance zone width corresponding to the outer layer medium thickness within the manufacturing tolerance boundary and the second tolerance zone width corresponding to the inner layer medium thickness within the manufacturing tolerance boundary are extracted respectively.

[0114] Specifically, in the variable configuration table of the simulation software, the first tolerance band width corresponding to the thickness of the outer layer medium (e.g., H1 = 0.12 mm as optimized in this invention) within the manufacturing tolerance boundary is extracted. Since the outer layer medium is usually thin, based on the lamination process of high-frequency boards, its allowable thickness fluctuation range is narrow (e.g., usually ±0.02 mm), so the first tolerance band width is the difference between the upper and lower limits, 0.04 mm. The second tolerance band width corresponding to the thickness of the inner layer medium (e.g., H2 = 0.5 mm as optimized in this invention) within the manufacturing tolerance boundary is extracted. Since the inner layer medium usually uses a thicker standard core board, in the multilayer board lamination process, the thickness dispersion (absolute fluctuation value) of the thicker core board is usually significantly greater than that of the outer layer prepreg or thin medium, and its allowable thickness fluctuation range is wider (e.g., usually ±0.04 mm), so the second tolerance band width is 0.08 mm.

[0115] Multiply the width of the first tolerance band by a preset perturbation scaling factor to obtain the first perturbation level of the independent perturbation applied to the thickness of the outer medium.

[0116] Multiplying the second tolerance band width by the perturbation scaling factor gives the second perturbation magnitude of the independent perturbation applied to the thickness of the inner layer medium.

[0117] Specifically, the function of the perturbation scaling factor is to cut the coarse-grained "tolerance band" into "fine-grained perturbation slices" suitable for linear approximation. In the RF microwave simulation scenario of this invention, in order to ensure differential accuracy without triggering nonlinear transitions in the high-frequency field, the value of the perturbation scaling factor is usually set between 0.05 and 0.2 (for example, preferably set to 0.1, i.e., extracting 10% of the tolerance band width).

[0118] Step S42: Extract the first phase change caused by the independent micro-perturbation of the outer layer medium thickness and the second phase change caused by the independent micro-perturbation of the inner layer medium thickness, respectively.

[0119] Specifically, while locking the linewidths of the outer microstrip line and the inner stripline in the multidimensional physical size variables, the thicknesses of both the outer and inner dielectric layers are maintained at the initial values ​​corresponding to the current traversal stagnation point. The electromagnetic field simulation solver is invoked to calculate the absolute value of the phase difference between the through-end and coupled-end transmissions deviating from the 90-degree orthogonal state under the current model state (i.e., the aforementioned phase error value), and this value is marked as the phase reference value.

[0120] With all other physical variables locked, the thickness of the outer dielectric layer is superimposed on the initial value with the first perturbation level obtained in the preceding steps to form a first perturbation state. From a physical mechanism perspective, increasing the thickness of the outer dielectric layer alters the energy distribution ratio of the outer microstrip line at the air-dielectric interface, causing a sudden change in the microstrip line phase velocity. Since the inner stripline in the structure of this invention is completely projected and covered by the outer microstrip line, this sudden change in phase velocity is directly transmitted to the inner layer through the mutual capacitance between the two coupling lines, triggering a cross-layer secondary distortion of the inner layer phase propagation constant. On the simulation side, the system extracts the current phase error value obtained under the first perturbation state and calculates the absolute value of the first difference between the current phase error value and the phase reference value. This absolute value of the first difference is defined as the first phase change, quantifying the destructive force of the outer transverse electromagnetic boundary reshaping on the orthogonal state of the system.

[0121] The outer dielectric thickness is restored to its initial value. Similarly, while locking all other physical variables, the inner dielectric thickness is superimposed with the second perturbation level obtained in the preceding steps to form a second perturbation state. Since the inner stripline is in a fully enclosed dielectric environment, the change in the inner dielectric thickness only uniformly alters the local dielectric constant of the space where the stripline is located. This is a convergent and linear internal field perturbation mechanism that does not trigger a redistribution of field strength across different dielectric interfaces. On the simulation side, the system extracts the current phase error value obtained under the second perturbation state and calculates the absolute value of the second difference between the current phase error value and the phase reference value. This absolute value of the second difference is defined as the second phase change, characterizing the phase drift caused purely by changes in the internal dielectric environment.

[0122] Step S43: The ratio of the first phase change to the corresponding micro-perturbation is taken as the first partial derivative, and the ratio of the second phase change to the corresponding micro-perturbation is taken as the second partial derivative.

[0123] Specifically, the first partial derivative characterizes the severity of cross-layer phase distortion caused by a unit change in the thickness of the outer dielectric layer at the "deadlock point" of the current multi-dimensional size combination, within the physical scale conforming to PCB manufacturing tolerances. Since the outer microstrip line is at the interface between air and dielectric, the value of this first partial derivative typically exhibits a high absolute value, directly revealing the extreme sensitivity of the outer dielectric thickness to the 90-degree orthogonal phase index under the current conditions. The second partial derivative, because the inner stripline is completely surrounded by dielectric, has a relatively gentler field disturbance; therefore, its absolute value is normally smaller than that of the first partial derivative. If the absolute value of the first partial derivative is significantly greater than that of the second partial derivative, it indicates that, under the current linewidth configuration, the destructive force of small fluctuations in the outer dielectric thickness on the orthogonal phase state far exceeds that of the inner dielectric.

[0124] Step S5: When it is determined that the signs of the first partial derivative and the second partial derivative are opposite, the traversal step size of the outer layer medium thickness and the inner layer medium thickness is differentially reduced according to the proportional relationship between the absolute values ​​of the first partial derivative and the absolute values ​​of the second partial derivative. The parameter traversal after the reduction step size is executed until the value of the comprehensive evaluation function is lower than the preset target threshold, and the current combination of multidimensional physical size variables is output.

[0125] Specifically, step S5 includes the following process:

[0126] Step S51: Extract the first absolute value of the first partial derivative and the second absolute value of the second partial derivative, and determine the ratio of the first absolute value to the second absolute value as the adversarial sensitivity ratio;

[0127] Specifically, the first absolute value of the first partial derivative and the second absolute value of the second partial derivative are extracted to characterize the sensitivity of the corresponding medium thickness dimension to phase error under the current parameter combination, regardless of the direction of change (sign). When the resistance sensitivity ratio is significantly greater than 1, it indicates that the outer medium thickness is significantly more sensitive to phase error than the inner medium thickness; when the resistance sensitivity ratio is significantly less than 1, it indicates that the inner medium thickness is more sensitive to phase error.

[0128] Step S52: Mark the medium thickness corresponding to the larger of the first absolute value and the second absolute value as the strong adversarial dimension, and mark the medium thickness corresponding to the smaller of the two absolute values ​​as the weak adversarial dimension.

[0129] Specifically, a strongly antagonistic dimension refers to a dimension that is highly sensitive to phase errors, making it more prone to phase performance degradation during manufacturing or perturbation. A weakly antagonistic dimension refers to a dimension that is relatively less sensitive to phase errors. If the signs are opposite, it indicates that under the current parameter combination, changes in the two medium thickness dimensions have an antagonistic effect on phase errors (e.g., increasing the first perpendicular distance increases the phase error, while increasing the second perpendicular distance decreases the phase error). In this case, adjusting the step size differentially can better offset the adverse effects on phase errors. If the signs are the same, the two dimensions affect phase errors in the same direction, and the antagonistic effect is weaker, but the step size can still be reduced according to the difference in sensitivity to improve convergence efficiency and robustness.

[0130] Step S53: Obtain the current initial traversal step size, apply a first reduction factor to the initial traversal step size of the strong adversarial dimension, and apply a second reduction factor to the initial traversal step size of the weak adversarial dimension.

[0131] Specifically, the initial traversal step size can be preset by the designer based on the PCB process precision and simulation resolution. It is typically set to the same order of magnitude as or slightly larger than the manufacturing precision to balance manufacturability and optimization efficiency. For example, for geometric parameters such as line width and dielectric thickness, the initial traversal step size can be set to the order of 0.005mm to 0.020mm, a range comparable to the tolerances for conventional PCB line width and dielectric thickness. The initial traversal step size can also be gradually decreased as the optimization process progresses (e.g., decreased every few rounds), a common practice in the field and will not be elaborated further. A smaller reduction factor (more forcefully compressing the step size) is applied to strongly antagonistic dimensions to suppress the impact of large fluctuations in that dimension on phase error; a relatively larger reduction factor (less step size reduction) is applied to weakly antagonistic dimensions, retaining some degree of adjustment freedom. Example configuration:

[0132] The first reduction factor ∈ [0.1, 0.5] (e.g., 0.2 to 0.3), and the second reduction factor ∈ [0.5, 0.9] (e.g., 0.6 to 0.8); or a mapping related to the "adversarial sensitivity ratio" can be used, for example: if the strong adversarial dimension is H1, let the first reduction factor = 1 / (1 + adversarial sensitivity ratio), and limit it to [0.1, 0.5]; the second reduction factor can be 1 – the first reduction factor or an independent preset value.

[0133] Step S54: Using the new step size obtained after applying the first reduction coefficient and the second reduction coefficient as the independent traversal step size for the corresponding dimension, continue to perform parameter traversal until the value of the comprehensive evaluation function is lower than the preset target threshold, and output the current combination of multidimensional physical size variables.

[0134] Specifically, using the new step size after differential reduction as the independent traversal step size for each dimension, parameter traversal continues within the manufacturing tolerance boundaries of the multidimensional physical dimension variables, and the comprehensive evaluation function value is calculated under each new set of parameters. The comprehensive evaluation function value is monitored in real time. When it is lower than the "preset target threshold", it is determined that the current combination of multidimensional physical dimension variables meets the design requirements, optimization stops, and the set of physical dimension variables is output as the design result. If the comprehensive evaluation function value is still not lower than the preset target threshold after several rounds (e.g., N rounds, where N can be preset), the parameter boundaries can be further reduced or the overall step size can be reduced according to conventional practices in the field until convergence or the maximum number of iterations is reached.

[0135] The technical solution of the present invention has been described above with reference to the preferred embodiments shown in the accompanying drawings. However, it will be readily understood by those skilled in the art that the scope of protection of the present invention is obviously not limited to these specific embodiments. Without departing from the principles of the present invention, those skilled in the art can make equivalent changes or substitutions to the relevant technical features, and the technical solutions after these changes or substitutions will all fall within the scope of protection of the present invention.

[0136] The above description is merely a preferred embodiment of the present invention and is not intended to limit the invention. Various modifications and variations can be made to the present invention by those skilled in the art. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the scope of protection of the present invention.

Claims

1. A hybrid transmission line 3dB coupler, characterized in that, include: A first conductor layer, a first dielectric layer, a second conductor layer, a second dielectric layer, and a third conductor layer are stacked sequentially. A first coupling line is provided within the first conductor layer; A second coupling line is provided within the second conductor layer; A reference ground plane is provided within the third conductor layer; The projection areas of the first coupling line and the second coupling line overlap in the direction perpendicular to the stacking arrangement, and the orthogonal projection area of ​​the second coupling line falls completely into the orthogonal projection area of ​​the first coupling line. The first coupling line and the second coupling line both use the reference ground plane as the reference ground.

2. The hybrid transmission line 3dB coupler according to claim 1, characterized in that, The first vertical distance from the plane containing the first coupling line to the reference ground plane is greater than the second vertical distance from the plane containing the second coupling line to the reference ground plane; and under the constraint that the 3dB directional coupler satisfies the preset odd-even mode impedance matching condition, the line width of the first coupling line is greater than the line width of the second coupling line.

3. The hybrid transmission line 3dB coupler according to claim 2, characterized in that, The first coupling line is a microstrip line, whose upper surface is exposed to air or covered with a solder resist layer, and whose lower surface is in close contact with the first dielectric layer; the second coupling line is a stripline, whose upper and lower surfaces are physically wrapped by the first dielectric layer and the second dielectric layer, respectively.

4. The hybrid transmission line 3dB coupler according to claim 3, characterized in that, The physical dimension of the first vertical distance is equal to the sum of the thickness of the first dielectric layer and the thickness of the second dielectric layer; the physical dimension of the second vertical distance is equal to the thickness of the second dielectric layer.

5. The hybrid transmission line 3dB coupler according to claim 4, characterized in that, The distributed parameters of the first coupling line and the distributed parameters of the second coupling line, under non-independent adjustment conditions, satisfy the target value of 3dB coupling odd-mode impedance; wherein, the odd-mode impedance of the first coupling line is equal to the odd-mode impedance of the second coupling line, and the even-mode impedance of the first coupling line is equal to the even-mode impedance of the second coupling line.

6. A design method for a hybrid transmission line 3dB coupler, characterized in that, include: Step S1: Based on the stacked structure of the outer microstrip line and the inner stripline, extract the linewidth of the outer microstrip line, the linewidth of the inner stripline, the thickness of the outer dielectric, and the thickness of the inner dielectric as multidimensional physical size variables. Step S2: Apply step perturbations to the linewidth of the outer microstrip line and the linewidth of the inner stripline respectively, and extract the changes in the electromagnetic characteristic response of the inner stripline; when it is determined that the absolute value of the change in the electromagnetic characteristic response caused by the perturbation of the outer microstrip linewidth is greater than the absolute value of the change in the electromagnetic characteristic response caused by the perturbation of the inner stripline linewidth, a cross-layer dominant marker is generated. Step S3: Replace the predefined odd-even mode impedance constraint target with a comprehensive evaluation function according to the cross-layer dominant marker. The comprehensive evaluation function is composed of the power distribution difference between the through end and the coupled end, the phase error value, and the port reflection coefficient weighted sum. Step S4: Perform parameter traversal and calculate the comprehensive evaluation function value within the manufacturing tolerance boundary of the multidimensional physical dimension variable; when it is detected that the absolute value of the rate of change of the comprehensive evaluation function value of two adjacent traversals is less than the preset gradient threshold, calculate the first partial derivative of the outer layer medium thickness with respect to the phase error value and the second partial derivative of the inner layer medium thickness with respect to the phase error value respectively. Step S5: When it is determined that the signs of the first partial derivative and the second partial derivative are opposite, the traversal step size of the outer layer medium thickness and the inner layer medium thickness is differentially reduced according to the proportional relationship between the absolute values ​​of the first partial derivative and the absolute values ​​of the second partial derivative. The parameter traversal after the reduction step size is executed until the value of the comprehensive evaluation function is lower than the preset target threshold, and the current combination of multidimensional physical size variables is output.

7. The design method of the hybrid transmission line 3dB coupler according to claim 6, characterized in that, The process of step S3 includes: Step S31, in response to the cross-layer dominant marker confirming that the independent decoupling space of the even and odd mode impedances has been distorted; Step S32: Construct the comprehensive evaluation function including the first weighting coefficient, the second weighting coefficient, and the third weighting coefficient; wherein, the difference between the absolute values ​​of the transmission coefficients of the through end and the coupled end is used as the power allocation difference, the absolute value of the phase difference between the transmission of the through end and the coupled end deviating from the 90-degree orthogonal state is used as the phase error value, and the absolute value of the reflection coefficient of the input port is used as the port reflection coefficient; Step S33: Based on the degree of interference of the outer edge electric field to the inner phase propagation constant represented by the cross-layer dominant marker, the first weighting coefficient corresponding to the phase error value is set to be greater than the second weighting coefficient corresponding to the power distribution difference, and greater than the third weighting coefficient corresponding to the port reflection coefficient.

8. The design method of the hybrid transmission line 3dB coupler according to claim 7, characterized in that, The process of step S33 includes: Under the current parameter combination for generating the cross-layer dominant marker, the first relative sensitivity of the outer microstrip line width to the inner stripline phase propagation constant and the second relative sensitivity of the inner stripline line width to the inner stripline phase propagation constant are calculated respectively, and the ratio of the two is used as the phase interference index. The phase interference index is multiplied by a preset phase base weight to obtain the first relative weight requirement value; at the same time, preset base power weight values ​​and base reflection weight values ​​are respectively configured for the second weight coefficient and the third weight coefficient; The first relative weight requirement value, the basic power weight value, and the basic reflection weight value are normalized to obtain the first weight coefficient, the second weight coefficient, and the third weight coefficient.

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