Power conversion device

Abstract

The utility model discloses a power conversion device contains a gate driver, to provide a drive signal to a gate node, a power transistor, its gate electric connection is in the gate node, to provide a load current to a load according to the drive signal with input voltage conversion output voltage, a sensing transistor, its gate electric connection is in the gate node, to sense the load current, to produce relevant sensing current, a current to voltage conversion unit, electric connection is in the sensing transistor, to press an impedance value with sensing current conversion voltage signal, and according to a control signal adjustment impedance value, and a control circuit, to produce the control signal, and according to the voltage signal, adjustment control signal, to control impedance value.

Description

Technical Field

[0001] This utility model refers to a power conversion device, and more particularly to a power conversion device that can effectively shorten the overcurrent protection start-up time. Background Technology

[0002] Power conversion devices are widely used in electronic equipment to convert input voltage into output voltage, providing a stable current or voltage to the load. With the increasing demand for high efficiency and reliability in electronic equipment, overcurrent protection technology has become a critical element in the design of power conversion devices. Overcurrent protection aims to monitor the load current and, when the current exceeds a safe threshold, quickly limit or interrupt the current to prevent circuit components from overheating, being damaged, or the system from failing.

[0003] In existing technologies, one type of overcurrent protection mechanism is implemented using a sense transistor (Sense MOS) connected in parallel with a power transistor (Power MOS). Specifically, the sense transistor shares the same drive signal as the power transistor, generating a sense current proportional to the load current. This sense current then passes through a sense resistor to generate a voltage across the circuit, which the control circuit uses to determine whether to activate the protection mechanism. The trigger current value (i.e., the current at which the overcurrent protection mechanism begins to operate) of this architecture is typically high, influenced by several factors, including the response time of the control circuit, the discharge delay of the drive signal, and the switching of the power transistor's operating region. An excessively high trigger current value can lead to protection activation delays, increasing the risk of damage to circuit components.

[0004] To reduce trigger current, existing technologies have proposed several solutions, but all have limitations. The first common method is to improve layout design and reduce metal resistance in the sensing path. However, this method offers limited improvement and requires increasing the metal wiring area, leading to larger chip size. The second method is to reduce the resistance value of the sensing resistor to decrease the voltage across the circuit and thus lower the trigger current. However, a smaller sensing resistor results in excessively low voltage across the circuit, requiring subsequent circuits to significantly improve signal processing accuracy, increasing design complexity and cost. The third method is to reduce the size ratio of the sensing transistor to the power transistor to change the sensing current ratio. However, this requires redesigning the power transistor size, increasing layout difficulty, and may cause electromigration problems due to increased sensing current, necessitating wider metal lines to address this.

[0005] In general, existing technologies face challenges in implementing overcurrent protection, including excessively high trigger current values, delayed protection response times, increased layout area, and increased circuit design complexity. Furthermore, existing technologies lack the ability to dynamically adjust load current, making it difficult to achieve fast and accurate protection under varying load conditions. Therefore, the industry is committed to providing an overcurrent protection technology that can reduce trigger current values, shorten protection startup time, and maintain the simplicity and reliability of circuit design. Utility Model Content

[0006] Therefore, the main purpose of this utility model is to provide a power conversion device that can effectively shorten the overcurrent protection start-up time.

[0007] This utility model provides a power conversion device, comprising: a gate driver for providing a drive signal to a gate node; a power transistor, one gate of which is electrically connected to the gate node, for converting an input voltage into an output voltage according to the drive signal to provide a load current to a load; a sensing transistor, one gate of which is electrically connected to the gate node, for sensing the load current to generate a sensing current related to the load current; a current-to-voltage conversion unit electrically connected to the sensing transistor, for converting the sensing current into a voltage signal according to an impedance value, and adjusting the impedance value according to a control signal; and a control circuit electrically connected to the gate node and the current-to-voltage conversion unit, for generating the control signal and adjusting the control signal according to the voltage signal to control the impedance value.

[0008] In one embodiment of this utility model, the control circuit gradually adjusts a level of the control signal after determining that the voltage signal is higher than a reference signal, so as to change the impedance value.

[0009] In one embodiment of the present invention, at a point in time after the control circuit determines that the voltage signal is higher than the reference signal, the impedance value of the current to voltage conversion unit is gradually switched from a first impedance value to a second impedance value.

[0010] In one embodiment of the present invention, the second impedance value is less than the first impedance value, and the time it takes for the first impedance value to gradually switch to the second impedance value is related to at least one characteristic of the load.

[0011] In one embodiment of the present invention, the control circuit is further used to adjust the gate voltage according to the voltage signal generated by the current-to-voltage conversion unit.

[0012] In one embodiment of the present invention, the current-to-voltage conversion unit includes at least one resistor and at least one transistor. The control circuit controls the conduction level of the at least one transistor through the control signal to change the impedance value of the current-to-voltage conversion unit.

[0013] In one embodiment of this utility model, the control circuit includes: a voltage signal sensing unit electrically connected to both ends of the current-to-voltage conversion unit, used to detect the voltage signal and output a detection result signal; a comparator used to compare the detection result signal with a reference signal and generate a comparison result signal; a first transistor including a drain electrically connected to the gate node, a gate receiving the input voltage, and a source; a second transistor including a drain generating and outputting the control signal, a gate receiving the comparison result signal, and a source; a first resistor electrically connected between the source of the first transistor and the drain of the second transistor; and a second resistor electrically connected between the source of the second transistor and a ground terminal.

[0014] In one embodiment of the present invention, the control circuit includes: a voltage signal sensing unit electrically connected to both ends of the current-to-voltage conversion unit, used to detect the voltage signal and output a detection result signal; a comparator used to compare the detection result signal with a reference signal and generate a comparison result signal; and a gate control module coupled to the gate node, the comparator and the output voltage, used to generate and output the control signal based on the comparison result signal, the gate voltage and the output voltage.

[0015] In one embodiment of the present invention, the current-to-voltage conversion unit is electrically connected between a drain and an input terminal of the sensing transistor, and the input terminal receives the input voltage.

[0016] In one embodiment of the present invention, the current-to-voltage conversion unit is electrically connected between a source and an output terminal of the sensing transistor, and the output terminal outputs the output voltage. Attached Figure Description

[0017] Figure 1 This is a schematic diagram of a power conversion device according to an embodiment of the present invention.

[0018] Figure 2 This is a schematic diagram of a current-to-voltage conversion unit and a control circuit according to an embodiment of the present invention.

[0019] Figure 3 for Figure 1 The simulation results of the overcurrent protection process of the power conversion device.

[0020] Figure 4 This is a schematic diagram comparing the load current reaching the trigger current value.

[0021] Figure 5 for Figure 1 The simulation results of the overcurrent protection process of the power conversion device.

[0022] Figure 6 This is a schematic diagram comparing the load current reaching the trigger current value.

[0023] Figure 7 This is a schematic diagram of a power conversion device according to an embodiment of the present invention.

[0024] Figure 8 This is a schematic diagram of a power conversion device according to an embodiment of the present invention.

[0025] Figure 9A and Figure 9B This is a schematic diagram of a current-to-voltage conversion unit according to an embodiment of the present invention.

[0026] Figure 9C This is a schematic diagram of a control signal output circuit according to an embodiment of the present invention.

[0027] Figure 9D This is a schematic diagram of the current-to-voltage conversion unit and control circuit of an embodiment of the present invention.

[0028] Figure 10 This is a schematic diagram of an overcurrent protection process according to an embodiment of the present invention.

[0029] Figure reference numerals: 10 - Power conversion device; T_in - Input terminal; Vin - Input voltage; Vout - Output voltage; T_out - Output terminal; 150 - Load; 100 - Gate driver; 110 - Power transistor; 120 - Sensing transistor; 130 - Current-to-voltage conversion unit; 140 - Control circuit; G - Gate; NG - Gate node; GD - Drive signal; Iout - Load current; Isen - Sensing current; Vsen - Voltage signal; Vopb - Control signal; VG - Gate voltage; D - Drain; S - Source; Rsen - Resistor; 200 - P-type metal-oxide-semiconductor (PMOS) transistor; 202 - Voltage signal sensing unit; 204 - Comparator; 206 - First transistor; 208 - Second transistor; R1 - First resistor; R2 - Second resistor; Vdet - Detection result signal; Vref - Reference signal; Vcomp - Comparison result signal; Gnd - Ground; 300~308 - Curve; ta0~ta3 - Time points; Rx1, Rx2 - Resistance values; 400, 402 - Curves ; 500~508 - Curve; tb0~tb3 - Time points; 600, 602 - Curves; 70 - Power conversion device; 730 - Current to voltage conversion unit; 740 - Control circuit; 80 - Power conversion device; 820_1~820_n - Sensing transistor; 830_1~830_n - Current to voltage conversion unit; 840 - Control circuit; Isen_1~Isen_n - Sensing current; Vsen_1~Vsen_n - Voltage signal; Vopb_1~Vopb_n - Control signal; 90 - Current to voltage conversion unit; 92 - Current to voltage conversion unit; M1~Mn - PMOS transistor; Z1~Zn - Impedance unit; 94 - Control signal output circuit; R1_1~R1_n - Resistor; 1000 - Overcurrent protection process; 1002~1012 - Steps; Itrip30, Itrip31, Itrip32, Itrip50, Itrip51, Itrip52 - Trigger current values; Ilock - Protection current value; 900 - Logic module; Rvar - Variable resistor. Detailed Implementation

[0030] Please refer to Figure 1 , Figure 1This is a schematic diagram of a power conversion device 10 according to an embodiment of the present invention. The power conversion device 10 receives an input voltage Vin at an input terminal T_in, converts it into an output voltage Vout, and provides it to a load 150 via the output terminal T_out, achieving efficient over-current protection. For clarity, in the following description, "G" represents the gate of a transistor, "D" represents the drain of a transistor, and "S" represents the source of a transistor. Specifically, the power conversion device 10 includes a gate driver 100, a power transistor 110, a sensing transistor 120, a current-to-voltage conversion unit 130, and a control circuit 140. The gate G of the power transistor 110 is electrically connected to the gate G of the sensing transistor 120, forming a gate node NG. The gate driver 100 generates and provides a drive signal GD to the gate node NG, so the power transistor 110 can convert the input voltage Vin into an output voltage Vout according to the drive signal GD, and provide a load current Iout to the load 150. Meanwhile, since the sensing transistor 120 and the power transistor 110 share the same drive signal GD, by selecting a transistor of an appropriate size (i.e., channel length), the sensing transistor 120 can generate a sensing current Isen that is in a specific proportion to the load current Iout. In other words, the sensing transistor 120 can sense the load current Iout and generate a sensing current Isen related to the load current Iout. Furthermore, the current-to-voltage conversion unit 130 is electrically connected to the sensing transistor 120 to convert the sensing current Isen into a voltage signal Vsen according to an impedance value, and adjusts the impedance value according to a control signal Vopb. The control circuit 140 is electrically connected to the gate node NG and the current-to-voltage conversion unit 130 to generate the control signal Vopb and adjust the control signal Vopb according to the voltage signal Vsen to control the impedance value of the current-to-voltage conversion unit 130.

[0031] The overcurrent protection mechanism of the power conversion device 10 is described below. When the power conversion device 10 starts operating, the gate driver 100 drives the power transistor 110 to output a load current Iout to the load 150. The sensing transistor 120, sharing the same drive signal GD as the power transistor 110, continuously senses the load current Iout and generates a sensing current Isen related to the load current Iout. The sensing current Isen is converted into a voltage signal Vsen by the current-to-voltage conversion unit 130 based on its impedance value. In other words, the current-to-voltage conversion unit 130 senses the sensing current Isen to generate a corresponding voltage signal Vsen. The control circuit 140 can determine the sensing current Isen based on the voltage signal Vsen, and then determine whether the load current Iout exceeds a current threshold. This current threshold is the target current value that the overcurrent protection mechanism needs to limit or lock. For simplicity, this current threshold is referred to as the protection current value (i.e., the target current value that the overcurrent protection mechanism needs to limit or lock in the following text is called the protection current value). In other words, when the load current Iout changes, the sensing current Isen, which has a specific proportional relationship with it, will also change accordingly. If it continues to increase and causes the voltage signal Vsen to exceed a voltage threshold corresponding to the protection current value, the control circuit 140 can determine that an overcurrent situation has occurred and activate the protection mechanism, such as adjusting the gate voltage VG of the gate node NG (e.g., reducing the gate voltage VG by conducting the path to ground), controlling the gate driver 100 to adjust the drive signal GD, etc., so that the load current Iout is locked or limited to the protection current value.

[0032] However, in actual operation, due to the response time of the control circuit 140, the discharge delay of the drive signal GD, and the switching of the operating region of the power transistor 110, the overcurrent protection will start later than the time when the load current Iout equals the protection current value. This means that the overcurrent protection only activates when the load current Iout has actually exceeded the set protection current value. In other words, the ideal design of the power conversion device 10 is to activate the overcurrent protection mechanism when the load current Iout just exceeds the protection current value. However, in actual operation, the load current Iout at the start of the overcurrent protection mechanism will be higher than the preset protection current value. For simplicity, the load current Iout at the actual start of the overcurrent protection mechanism will be referred to as the trigger current value. In other words, although the overcurrent protection mechanism of the power conversion device 10 limits or locks the load current Iout to the protection current value, the load current Iout when the overcurrent protection mechanism is actually activated will be higher than the protection current value and equal to the trigger current value. After the overcurrent protection mechanism is activated, the control circuit 140 will lock or limit the load current Iout to the protection current value by adjusting the gate voltage VG and other mechanisms.

[0033] The difference between the protection current value and the trigger current value may cause a delay in protection activation, increasing the risk of damage to circuit components. Therefore, this embodiment of the invention incorporates a dynamic impedance adjustment mechanism. Through dynamic impedance adjustment and precise control signal regulation, the difference between the trigger current value and the protection current value after the overcurrent protection mechanism is activated is reduced, effectively shortening the protection activation time and ensuring rapid protection and precise current control.

[0034] Specifically, after the overcurrent protection mechanism is activated, in addition to adjusting the gate voltage VG of the gate node NG and controlling the gate driver 100 to adjust the drive signal GD, so that the load current Iout is locked or limited to the protection current value, the control circuit 140 can also output and adjust the control signal Vopb to control the impedance value of the current-to-voltage conversion unit 130. That is to say, in addition to determining whether to activate overcurrent protection based on the voltage signal Vsen generated by the current-to-voltage conversion unit 130, the control circuit 140 can also adjust the level of the control signal Vopb based on the voltage signal Vsen to change the impedance value of the current-to-voltage conversion unit 130 when overcurrent protection is activated, so as to ensure that the load current Iout is quickly limited after exceeding the protection current value (i.e., after the overcurrent protection is activated), and to avoid damage to circuit components due to overheating or overload. In other words, this utility model embodiment not only determines whether to activate overcurrent protection by sensing the load current Iout, but also adjusts the impedance value used to sense the sensing current Isen (i.e., the impedance value of the current-to-voltage conversion unit 130) after activating overcurrent protection, thereby dynamically changing the difference between the trigger current value and the protection current value, thereby improving the immediacy of protection.

[0035] Furthermore, the power conversion device 10 of this invention can dynamically adjust the difference between the trigger current value and the protection current value before and after the overcurrent protection mechanism is activated by adjusting the impedance value (sensing resistance value) of the current-to-voltage conversion unit 130, thereby improving protection efficiency and reducing the risk of high current. As mentioned above, by selecting a transistor of appropriate size, the sensing transistor 120 can generate a sensing current Isen in a specific proportion to the load current Iout. When the power conversion device 10 is working normally and the load current Iout gradually increases and is about to trigger the overcurrent protection, the power transistor 110 is usually fully turned on and operates in the linear region, at which time its equivalent resistance is small. Then, when the overcurrent protection is triggered, the control circuit 140 reduces the gate voltage VG of the main power transistor 110 and locks or limits the load current Iout to the protection current value. At this time, the power transistor 110 operates in the saturation region, and its equivalent resistance increases. In other words, as the power conversion device 10 transitions from normal operation to approaching overcurrent protection and then triggering overcurrent protection, the equivalent resistance of the power transistor 110 changes due to the change in the operating range. This change means that the ratio between the current sensed by the current-to-voltage conversion unit 130 and the actual current flowing through the power transistor 110 is not always precisely maintained at the originally set ratio, especially near the trigger point of overcurrent protection (the expected protection current value, but the actual trigger current value). Therefore, when the control circuit 140 detects that the voltage signal Vsen reaches the voltage that triggers overcurrent protection, this embodiment of the invention can reduce the impedance value of the current-to-voltage conversion unit 130, so that the current-to-voltage conversion unit 130 can more accurately reflect the current ratio in the saturation range, thereby tracking the protection current value more quickly under overcurrent protection triggering, and making it easier for the load current Iout to approach the protection current value.

[0036] It should be noted that the protection current value is the load current Iout limit set by the power conversion device 10 during the circuit design stage, while the trigger current value is the load current Iout when the power conversion device 10 actually activates the overcurrent protection, which is affected by many circuit conditions or operating environments. This embodiment of the invention can reduce the impedance value of the current-to-voltage conversion unit 130 when the overcurrent protection is activated, making the current-to-voltage conversion unit 130 more "sensitive" to changes in the load current Iout. This is equivalent to reducing the difference between the trigger current value and the protection current value after the overcurrent protection is activated under the same circuit conditions, thus making the load current Iout after the overcurrent protection is activated more easily approach the protection current value.

[0037] Furthermore, the current-to-voltage conversion unit 130 converts the sensed current into a voltage signal according to its impedance value and adjusts the impedance value according to the control signal Vopb. The control circuit 140, in addition to controlling the activation of the overcurrent protection, can also adjust the control signal Vopb according to the gate voltage VG and the voltage signal Vsen, thereby changing the impedance value of the current-to-voltage conversion unit 130 in a timely manner, and thus changing the difference between the trigger current value and the protection current value. The implementation of the current-to-voltage conversion unit 130 and the control circuit 140 can be determined according to application requirements and is not limited to a specific architecture. Those skilled in the art can implement the same function using different types of electronic components or modules according to system requirements. For example, the current-to-voltage conversion unit 130 may include different numbers of resistors and transistor combinations, or use other variable impedance elements (such as electrically controlled resistors or capacitors) to adjust the impedance value. Similarly, the control circuit 140 can employ various control architectures, such as a voltage sensing and comparator structure based on analog circuits, or a gate control module combined with digital signal processing, to achieve precise adjustment of the control signal Vopb. Furthermore, the protection mechanism of the control circuit 140 can be further integrated with other functions, such as over-temperature protection or short-circuit protection, to enhance system reliability. These changes do not depart from the spirit of this invention and can be adjusted according to the characteristics of the load 150 or the design objectives of the power conversion device 10.

[0038] For example, please refer to Figure 2 , Figure 2This is a schematic diagram of one embodiment of the current-to-voltage conversion unit 130 and the control circuit 140. In this embodiment, the current-to-voltage conversion unit 130 includes a resistor Rsen and a P-type metal-oxide-semiconductor (PMOS) transistor 200, which are electrically connected in parallel between the drain D of the sensing transistor 120 and the input terminal T_in. The resistor Rsen provides an initial impedance value. When the sensed current Isen passes through the resistor Rsen, a voltage signal Vsen is generated. The gate G of the PMOS transistor 200 receives a control signal Vopb, and its conduction level is adjusted by the control circuit 140 (through the control signal Vopb) to dynamically change the impedance value of the current-to-voltage conversion unit 130. The control circuit 140 includes a voltage signal sensing unit 202, a comparator 204, a first transistor 206, a second transistor 208, a first resistor R1, and a second resistor R2. A voltage signal sensing unit 202 is electrically connected to both ends of a current-to-voltage conversion unit 130 (i.e., the drain D and input T_in of sensing transistor 120) to detect a voltage signal Vsen and output a detection result signal Vdet. A comparator 204 receives the detection result signal Vdet and a reference signal Vref, compares the detection result signal Vdet with the reference signal Vref, and generates a comparison result signal Vcomp; wherein the reference signal Vref corresponds to the protection current value, i.e., the comparison result signal Vcomp can be used to indicate whether the load current Iout has reached the protection current value. The drain D of a first transistor 206 (e.g., an N-channel MOSFET) is electrically connected to the gate node NG, the gate G is electrically connected to the input T_in to receive the input voltage Vin, and the source S is electrically connected to one end of a first resistor R1. The drain D of a second transistor 208 (e.g., an N-channel MOSFET) is electrically connected to the other end of the first resistor R1, the gate G receives the comparison result signal Vcomp, and the source S is electrically connected to a second resistor R2. The second resistor R2 is further electrically connected to a ground terminal Gnd. Additionally, the drain D of the second transistor 208 generates and outputs a control signal Vopb.

[0039] pass Figure 2In an embodiment of the current-to-voltage conversion unit 130 and control circuit 140, when the power conversion device 10 is operating, the sensing transistor 120 generates a sensing current Isen proportional to the load current Iout. The sensing current Isen flows through the parallel structure of the current-to-voltage conversion unit 130 (resistor Rsen and PMOS transistor 200) to generate a voltage signal Vsen. The voltage signal sensing unit 202 continuously monitors the voltage signal Vsen and converts it into a detection result signal Vdet, which is input to the comparator 204. The comparator 204 compares the detection result signal Vdet with a reference signal Vref to generate and adjust the control signal Vopb. In one embodiment, when the detection result signal Vdet is lower than or equal to the reference signal Vref (i.e., the load current Iout does not exceed the protection current value), the comparator 204 outputs a low-level comparison result signal Vcomp, keeping the second transistor 208 off. At this time, the control signal Vopb remains at a high potential, the PMOS transistor 200 is turned off, and the impedance value of the current-to-voltage conversion unit 130 is determined by the resistor Rsen (a first impedance value, also known as the trigger impedance value). When the load current Iout increases, causing the voltage signal Vsen to rise, and the detection result signal Vdet exceeds the reference signal Vref (i.e., the load current Iout exceeds the protection current value), the comparator 204 outputs a high-level comparison result signal Vcomp, driving the second transistor 208 to turn on. The turning on of the second transistor 208 reduces the voltage of the control signal Vopb, thereby causing the PMOS transistor 200 to partially or fully turn on, reducing the equivalent impedance value of the current-to-voltage conversion unit 130 to a second impedance value (also known as the latch-up impedance value). At the same time, the turning on of the second transistor 208 opens the current path from the gate node NG to the ground terminal Gnd, correspondingly reducing the gate voltage VG, so as to control the power transistor 110 to latch or limit the load current Iout to the protection current value. In other words, when the detection result signal Vdet exceeds the reference signal Vref and the overcurrent protection is activated, in addition to adjusting the gate voltage VG to control the power transistor 110 to reduce the load current Iout to the protection current value, the control circuit 140 will also reduce the impedance value of the current to voltage conversion unit 130, so that the difference between the trigger current value and the protection current value is reduced, making it easier for the load current Iout after the overcurrent protection is activated to approach the protection current value.

[0040] In short, after the power conversion device 10 operates but before overcurrent protection is activated, the current-to-voltage conversion unit 130 uses a larger resistor Rsen (the first impedance value, i.e., the trigger impedance value) to convert the sensed current Isen to ensure detection sensitivity. When the voltage signal Vsen exceeds the reference signal Vref (corresponding to the protection current value), the control circuit 140 adjusts the control signal Vopb, driving the PMOS transistor 200 to turn on, reducing the equivalent impedance of the current-to-voltage conversion unit 130 to a smaller second impedance value (lock-in impedance value). This impedance switching reduces the influence of the resistor Rsen on the sensed current Isen, making the sensed current Isen more accurately reflect the load current Iout, while simultaneously controlling the power transistor 110 to stabilize in the saturation range, precisely locking the load current Iout to the protection current value. In this way, the difference between the trigger current value and the protection current value after overcurrent protection is activated is significantly reduced, shortening the overcurrent protection activation time and greatly improving immediacy. Furthermore, dynamic impedance adjustment eliminates the need for widened metal lines or complex post-processing, effectively avoiding electron migration problems, simplifying layout, and reducing chip area.

[0041] It should be noted that when the power conversion device 10 transitions from normal operation to overcurrent protection, the equivalent impedance of the current-to-voltage conversion unit 130 is dominated by the resistor Rsen due to the conduction of the PMOS transistor 200. The impedance changes from a larger first impedance value (trigger impedance value) to a smaller second impedance value (lock-in impedance value), making the sensed current Isen more accurately reflect the load current Iout, effectively reducing the trigger current value significantly. Although the equivalent resistance of the power transistor 110 changes due to the change in its operating range before and after entering overcurrent protection, by changing the equivalent impedance of the current-to-voltage conversion unit 130, this embodiment of the invention effectively reduces the trigger current value, thus "narrowing" the gap between the trigger current value and the protection current value. This makes the actual trigger current value closer to the target value, thereby shortening the overcurrent protection start-up time and significantly improving immediacy.

[0042] Regarding the power conversion device 10 Figure 2 For an example of how it works, please refer to [the relevant documentation / example]. Figure 3 , Figure 3 The simulation results show the overcurrent protection process for the power conversion device 10. In this example, it is assumed that the load 150 is a 220μF capacitor. Figure 3 Curves 300 to 308 are plotted from top to bottom, corresponding sequentially to the time variations of the load current Iout, the comparison result signal Vcomp, the gate voltage VG, the control signal Vopb, and the equivalent impedance value (resistance value form) of the current-to-voltage conversion unit 130. Figure 3As shown, during normal operation, before time point ta0, the load current Iout is lower than a protection current value Ilock. The comparison result signal Vcomp remains at a low level, the gate voltage VG remains high to drive the power transistor 110 in the linear range, and the control signal Vopb is high to turn off the PMOS transistor 200. The equivalent impedance value is determined by the resistor Rsen (the first impedance value, which is the resistance value Rx1 in this example). When the load current Iout gradually increases and reaches the protection current value Ilock, at time point ta0, the control circuit 140 should activate overcurrent protection. However, due to various circuit factors, the comparison result signal Vcomp has not yet reflected the situation where the load current Iout has reached the protection current value Ilock, and therefore remains at a low level. Until time point ta1, the voltage signal Vsen exceeds the reference signal Vref, causing the comparison result signal Vcomp to quickly switch to a high level. Then, the control circuit 140 activates overcurrent protection, triggering the second transistor 208 to turn on. Therefore, when the overcurrent protection is fully activated, the load current Iout has actually reached (above the protection current value Ilock) a trigger current value Itrip30. After the overcurrent protection is activated at time ta1, the gate voltage VG drops rapidly, the power transistor 110 enters the saturation range, and the control signal Vopb gradually decreases, driving the PMOS transistor 200 to turn on, causing the equivalent impedance value of the current-to-voltage conversion unit 130 to gradually (from time ta2 to time ta3) switch to a smaller second impedance value (resistance value Rx2 in this example); at the same time, the load current Iout drops rapidly and stabilizes at the protection current value Ilock. In addition, it can be seen that the difference between the trigger current value Itrip30 and the protection current value Ilock is reduced, and the overcurrent protection activation time (the time from the comparison result signal Vcomp switching to the load current Iout stabilizing at the protection current value Ilock) is extremely short.

[0043] about Figure 3 For examples of improved overcurrent protection start-up time, please refer to [the relevant documentation / reference]. Figure 4 , Figure 4 This is a schematic diagram comparing the load current Iout reaching the trigger current value when the load is a 220μF capacitor (150Ω). Figure 4 In the diagram, curve 400 represents the variation of the load current Iout under a fixed resistor architecture, which can be considered as... Figure 2 In the embodiment, the control signal Vopb output by the control circuit 140 to the PMOS transistor 200 is disconnected, or the PMOS transistor 200 is removed, resulting in the load current Iout. The trigger current value under this operating condition is Itrip31; while curve 402 represents the complete application of... Figure 2The load current Iout obtained by the control mechanism of the embodiment, i.e., when overcurrent protection is activated, the control signal Vopb can drive the PMOS transistor 200 to conduct, causing the equivalent impedance value of the current-to-voltage conversion unit 130 to switch to a smaller second impedance value. The trigger current value under this operating condition is Itrip32. Figure 4 As can be seen from curve 400, in the fixed resistor architecture, due to the influence of the sensing resistor Rsen and response delay, after the load current Iout exceeds the protection current value Ilock, it takes a relatively long time for it to start decreasing to the protection current value Ilock until the trigger current value Itrip31, increasing the risk of overcurrent. In contrast, from Figure 4 As can be seen from curve 402, this invention uses dynamic impedance adjustment. Initially, a larger resistance Rsen (Rx1 in this example) is used to ensure detection sensitivity. After overcurrent protection is triggered, the resistance is quickly reduced to Rx2, allowing the sensed current Isen to more accurately reflect the load current Iout. The load current Iout is then quickly stabilized at the protection current value Ilock by controlling the gate voltage VG. Figure 4 Simulation results show that the load current Iout of this invention rapidly decreases to the protection current Ilock after reaching the trigger current value Itrip32 (an improvement of approximately 300mA). Furthermore, the difference between the trigger current value Itrip32 and the protection current value Ilock is significantly smaller than the difference between the trigger current value Itrip31 and the protection current value Ilock, representing a significant improvement over a fixed resistor architecture. Simultaneously, the protection startup time is shortened to 70-80μs. This improvement stems from the dynamic impedance adjustment mechanism of the current-to-voltage conversion unit 130 and the control signal Vopb adjustment of the control circuit 140, eliminating the need for complex layouts or subsequent signal processing, further reducing the risk of electron migration and chip area.

[0044] It should be noted that, Figure 3 and Figure 4 The simulation results are for a load of 150Ω and a 220μF capacitor. However, this invention has similar efficacy for different loads. Please refer to... Figure 5 , Figure 5 The simulation results show the overcurrent protection process for the power conversion device 10. In this example, it is assumed that the load 150 is a 0.1μF capacitor. Figure 5 Curves 500 to 508 are plotted from top to bottom, corresponding sequentially to the time variations of the load current Iout, the comparison result signal Vcomp, the gate voltage VG, the control signal Vopb, and the equivalent impedance value (resistance value form) of the current-to-voltage conversion unit 130. Figure 5As shown, during normal operation, before time point tb0, the load current Iout is lower than the protection current value Ilock. The comparison result signal Vcomp remains at a low level, the gate voltage VG remains high to drive the power transistor 110 in the linear range, and the control signal Vopb is high to turn off the PMOS transistor 200. The equivalent impedance value is determined by the resistor Rsen (the first impedance value, which is the resistance value Rx1 in this example). When the load current Iout gradually increases and reaches the protection current value Ilock, at time point tb0, the control circuit 140 should activate overcurrent protection. However, due to various circuit factors, the comparison result signal Vcomp has not yet reflected the situation where the load current Iout has reached the protection current value Ilock, and therefore remains at a low level. Until time point tb1, the voltage signal Vsen exceeds the reference signal Vref, causing the comparison result signal Vcomp to quickly switch to a high level. Then, the control circuit 140 activates overcurrent protection, triggering the second transistor 208 to turn on. Therefore, when the overcurrent protection is fully activated, the load current Iout has actually reached (higher than the protection current value Ilock) a trigger current value Itrip50. After the overcurrent protection is activated at time tb1, the gate voltage VG drops rapidly, the power transistor 110 enters the saturation range, and the control signal Vopb gradually decreases, driving the PMOS transistor 200 to turn on, causing the equivalent impedance value of the current-to-voltage conversion unit 130 to gradually (from time tb2 to time tb3) switch to a smaller second impedance value (resistance value Rx2 in this example); simultaneously, the load current Iout drops rapidly and stabilizes at the protection current value Ilock. Compared to Figure 3 The example is under light load, such as Figure 5 As shown, when the load current Iout reaches the trigger current value Itrip50 after time point tb1, the comparison result signal Vcomp can also quickly switch to a high level, triggering the second transistor 208 to turn on. The difference is that, under light load conditions, the time for the equivalent impedance value of the current-to-voltage conversion unit 130 to switch from the resistance value Rx1 to the smaller resistance value Rx2 is shorter, i.e. Figure 5 The time from midpoint tb2 to midpoint tb3 is less than Figure 3 The time from time point ta2 to time point ta3. In addition, the load current Iout can also drop rapidly and stabilize at the protection current value Ilock, and the difference between the trigger current value Itrip50 and the protection current value Ilock can be significantly reduced, while the overcurrent protection start time (the time from the comparison result signal Vcomp to the load current Iout stabilizing at the protection current value Ilock) is extremely short.

[0045] about Figure 5 For an example of improved overcurrent protection start-up time, please refer to [the relevant documentation / reference]. Figure 6 , Figure 6This is a schematic diagram comparing the load current Iout reaching the trigger current value when the load is a 0.1μF capacitor (150Ω). Figure 6 In the diagram, curve 600 represents the variation of the load current Iout under a fixed resistor architecture, which can be considered as... Figure 2 In the embodiment, the control signal Vopb output by the control circuit 140 to the PMOS transistor 200 is disconnected, or the PMOS transistor 200 is removed, resulting in the load current Iout. The trigger current value under this operating condition is Itrip51; while curve 602 represents the complete application of... Figure 2 The control mechanism of the embodiment obtains the load current Iout, i.e., when overcurrent protection is activated, the control signal Vopb can drive the PMOS transistor 200 to turn on, causing the equivalent impedance value of the current-to-voltage conversion unit 130 to switch to a smaller second impedance value. The trigger current value under this operating condition is Itrip52. Similarly... Figure 4 The simulation results are from Figure 6 The simulation results also show that even under light load conditions, the load current Iout drops rapidly to the protection current Ilock after reaching the trigger current value Itrip52, which is about 275mA better than the fixed resistor architecture, and the protection start time is also shortened to 70-80μs.

[0046] Figures 3 to 6 Simulation results show that regardless of the size of the load 150 (e.g., a 220μF or 0.1μF capacitor), the power conversion device 10 of this invention can effectively reduce the gap between the trigger current value and the protection current value, significantly improving the immediacy and reliability of overcurrent protection. Under heavy load (220μF) and light load (0.1μF) conditions, the current-to-voltage conversion unit 130 ensures that the sensed current Isen accurately reflects the load current Iout through dynamic impedance adjustment (in this example, switching from resistance value Rx1 to resistance value Rx2), and quickly stabilizes the load current Iout to the protection current value (improvement of approximately 275-300mA). Furthermore, the impedance switching experience time (e.g.) Figure 3 Midpoint ta2 to ta3, or Figure 5 The impedance switching time (from tb2 to tb3) is related to the load characteristics. Specifically, a larger load capacitance (e.g., 220μF) results in a longer impedance switching time due to the slower current change, while a smaller load capacitance (e.g., 0.1μF) results in a shorter impedance switching time due to the faster current change. This characteristic allows this invention to be applied to different load conditions, achieving fast and accurate current control while maintaining the simplicity and reliability of the circuit design.

[0047] It should be noted that the above embodiments are merely preferred implementations of this utility model. Those skilled in the art can employ various derivative variations to achieve similar functions according to application requirements without departing from the spirit of this utility model. For example, the current-to-voltage conversion unit 130 can use other variable impedance elements (such as electrically controlled resistors or adjustable capacitors) or combinations of different numbers of resistors and transistors to achieve dynamic impedance adjustment. The control circuit 140 can also integrate a digital signal processing module or an advanced comparator structure to further precisely adjust the control signal Vopb. In addition, the protection mechanism can be extended to over-temperature protection, short-circuit protection, or other circuit safety functions to meet the needs of specific application scenarios. These variations all maintain the core advantage of reducing the gap between the trigger current value and the protection current value and are applicable to different load conditions and circuit layouts.

[0048] For example, please refer to Figure 7 , Figure 7 This is a schematic diagram of a power conversion device 70 according to another embodiment of the present invention. The power conversion device 70 is composed of... Figure 1 The power conversion device 10 is derived from it, therefore the same components are represented by the same symbols. Figure 1 The main difference is that the current-to-voltage conversion unit 730 of the power conversion device 70 is electrically connected between the source S of the sensing transistor 120 and the output terminal T_out, while the current-to-voltage conversion unit in the power conversion device 10 is electrically connected between the drain D of the sensing transistor 120 and the input terminal T_in. Otherwise, the operation of the power conversion device 70 is largely the same as that of the power conversion device 10. That is, the sensing transistor 120 generates a sensing current Isen proportional to the load current Iout, which is converted into a voltage signal Vsen by the current-to-voltage conversion unit 730. When Vsen exceeds a preset value, the control circuit 140 adjusts Vopb to reduce the equivalent impedance and simultaneously reduces the gate voltage VG, stabilizing the load current Iout at the protection current value. The power conversion device 70 also achieves dynamic impedance adjustment, narrowing the gap between the trigger current value and the protection current value, and shortening the protection start-up time. It is suitable for applications requiring current sensing at the output terminal without complex layouts or widened metal wires.

[0049] In addition, please refer to Figure 8 , Figure 8 This is a schematic diagram of a power conversion device 80 according to another embodiment of the present invention. The power conversion device 80 also comprises... Figure 1 The power conversion device 10 is derived from it, therefore the same components are represented by the same symbols. Figure 1The main difference lies in that the power conversion device 80 includes multiple parallel sensing transistors 820_1, 820_2, ..., 820_n and corresponding current-to-voltage conversion units 830_1, 830_2, ..., 830_n, allowing current sensing at different locations on the power transistor 110 and improving monitoring accuracy. Correspondingly, the control circuit 840 of the power conversion device 80 should be able to comprehensively judge and control the impedance values ​​of the current-to-voltage conversion units 830_1, 830_2, ..., 830_n. Specifically, the gate G of each sensing transistor 820_1 to 820_n is electrically connected to the gate node NG, shares the drive signal GD, and generates sensing currents Isen_1, Isen_2, ..., Isen_n proportional to Iout, respectively. Each current-to-voltage conversion unit 830_1 to 830_n converts the corresponding sensed currents Isen_1, Isen_2, ..., Isen_n into voltage signals Vsen_1, Vsen_2, ..., Vsen_n for comprehensive judgment by the control circuit 840. When any voltage signal Vsen_i exceeds a preset value (corresponding to the protection current value), the control circuit 840 adjusts the control signal Vopb, reduces the equivalent impedance of each current-to-voltage conversion unit, and controls the gate voltage VG to stabilize the load current Iout at the protection current value. This multi-point sensing design enhances the monitoring capability of the current distribution of the power transistor 110, and is particularly suitable for large-area or high-current applications, maintaining the effect of minimizing the gap between the trigger current value and the protection current value and providing rapid protection, while avoiding electron migration problems and layout complexity.

[0050] Furthermore, the current-to-voltage conversion unit 130 can employ other variable impedance elements or combinations of different numbers of resistors and transistors, and the control circuit 140 can also be appropriately varied to generate multi-stage control signals Vopb. For example, Figure 9A and Figure 9B This is a schematic diagram of the current-to-voltage conversion units 90 and 92 according to an embodiment of the present invention. Figure 9A As shown, the current-to-voltage conversion unit 90 has n variable impedance elements connected in parallel. Each variable impedance element includes a PMOS transistor (denoted as M1 to Mn) and an impedance unit (denoted as Z1 to Zn) connected in series. The impedance units Z1 to Zn can be the same or different impedance elements, and can include one or more of resistors, capacitors, and inductors. Furthermore, as... Figure 9B As shown, the current-to-voltage conversion unit 92 is connected in series with n variable impedance elements. Each variable impedance element includes a PMOS transistor (denoted as M1 to Mn) and an impedance unit (denoted as Z1 to Zn) connected in parallel. The impedance units Z1 to Zn can be the same or different impedance elements, and can include one or more of resistors, capacitors, and inductors. Further details can be found in [reference needed]. Figure 9C , Figure 9CThis is a schematic diagram of a control signal output circuit 94 according to an embodiment of the present invention. The control signal output circuit 94 consists of... Figure 2 Derived from the embodiments described above, the same symbols are used to represent the same components, and other components are omitted. The control signal output circuit 94 is... Figure 2 The first resistor R1 in the control circuit 140 is replaced by a series resistor R1_1 to R1_n to generate control signals Vopb_1 to Vopb_n. The control signals Vopb_1 to Vopb_n can be provided to the current-to-voltage conversion units 90 and 92 to realize multi-level control.

[0051] Figures 9A to 9C The current-to-voltage conversion unit and control signal generation method applicable to this invention are highly flexible, but not limited to these. Those skilled in the art can further vary the composition of impedance units Z1-Zn (e.g., combinations of resistors, capacitors, and inductors) or the dimensions of PMOS transistors M1-Mn, or adjust the order and range of control signals Vopb_1-Vopb_n, according to application requirements, to meet the requirements of precise current control. On the other hand, the control circuit 140 of this invention can also be partially implemented digitally. For example, please refer to... Figure 9D , Figure 9D This is a schematic diagram of another embodiment of the current-to-voltage conversion unit 130 and the control circuit 140. Figure 9D The embodiments are provided by Figure 2 Derived from the embodiments described above, therefore the same elements are represented by the same symbols. Figure 2 The difference in the embodiments is that, Figure 9DIn the embodiment, the current-to-voltage conversion unit 130 is implemented by a variable resistor Rvar, and the control circuit 140 further includes a logic module 900. The logic module 900 is coupled to the comparator 204, the output terminal T_out, the gate node NG, and the variable resistor Rvar, and is used to generate a control signal Vopb according to the comparison result signal Vcomp and the voltage difference between the gate node NG and the output terminal T_out (i.e., the gate-to-source voltage VGS of the power transistor 110 or the sense transistor 120, hereinafter referred to as the gate-source voltage VGS) to control the resistance value of the variable resistor Rvar. Specifically, when the comparison result signal Vcomp indicates that an overcurrent occurs, the logic module 900 compares whether VGS < lock_V + VT holds, and when it holds, triggers the variable resistor Rvar to change its resistance value (switch from a larger resistance value to a smaller resistance value) through the control signal Vopb. Here, lock_V represents the gate-source voltage value corresponding to the protection current value, that is, when the power transistor 110 is controlled to lock the load current Iout at the protection current value, its gate-source voltage VGS will drop to a specific voltage value vicinity, which is lock_V; and VT represents an additional margin value, and adding VT is to ensure that the switching action will not occur prematurely and can be appropriately adjusted according to the system requirements. Figure 9D For the remaining detailed operations of the embodiment, reference can be made to the foregoing, and details are not described herein. Through Figure 9D the embodiment, it can be seen that in the present invention, the switching of the impedance can be realized not only through the circuit but also by adding a digital judgment mechanism to achieve more accurate and stable protection startup and current locking.

[0052] In addition, in another embodiment, Figure 2 the control signal output circuit composed of the first transistor 206, the second transistor 208, the first resistor R1, and the second resistor R2 in the control circuit 140 of can be replaced with a gate control module; the gate control module can be implemented by a digital signal processing (DSP) unit, and according to the comparison result signal Vcomp, the gate voltage VG, and the output voltage Vout, generates the control signal Vopb through a digital algorithm. This digital implementation allows for a more flexible impedance adjustment strategy, such as dynamically setting multiple levels of Vopb according to the load characteristics, or integrating feedback control to optimize the protection response. In addition, the gate control module can be integrated with the gate driver 100 into a single module, sharing the digital control logic, further simplifying the circuit design and reducing the chip area.

[0053] On the other hand, the feasible implementation of each component in the power conversion device 10 (and its derivative power conversion devices 70, 80) is not limited to a specific architecture. For example, the gate driver 100 can employ an analog drive circuit, a digital pulse width modulation (PWM) controller, or a mixed-signal driver to provide a stable drive signal GD according to application requirements. The power transistor 110 and the sensing transistor 120 can be N-type or P-type MOSFETs, or other power components (such as BJTs, IGBTs), and their size ratio is designed according to the preset ratio of the sensing current Isen to the load current Iout. The current-to-voltage conversion unit 130 (or 730, 830_1 to 830_n) may include an electrically controlled resistor, an adjustable capacitor, or other impedance components in addition to a resistor connected in parallel with the PMOS transistor, and supports single-stage or multi-stage impedance switching (e.g., Figure 9A and Figure 9B The control circuit 140 (or 740, 840) can be an analog circuit (e.g. Figure 2 These implementations utilize comparator and transistor structures, digital controllers (such as DSP modules), or hybrid architectures to integrate overcurrent, overtemperature, or short-circuit protection. All of these implementations enable dynamic impedance adjustment, ensuring rapid protection and precise current control, making them suitable for various power conversion applications.

[0054] The operation of the power conversion device 10 (and its derived power conversion devices 70 and 80) can be summarized as an overcurrent protection process 1000, such as... Figure 10 As shown, but not limited to, the overcurrent protection procedure 1000 includes the following steps:

[0055] Step 1002: Begin.

[0056] Step 1004: Sensing the load current Iout through sensing transistor 120 to generate a sensing current Isen related to the load current Iout.

[0057] Step 1006: Convert the sensed current Isen into a voltage signal Vsen according to the impedance value of the current-to-voltage conversion unit 130.

[0058] Step 1008: Generate control signal Vopb based on voltage signal Vsen.

[0059] Step 1010: Adjust the current to the impedance value of the voltage conversion unit 130 according to the control signal Vopb.

[0060] Step 1012: End.

[0061] For detailed operation of the overcurrent protection process 1000, please refer to the aforementioned description, which will not be repeated here.

[0062] In existing technologies, the sensing path used to sense load current employs a fixed resistor architecture, meaning the sensing resistance value cannot be dynamically adjusted. This results in a significant difference between the trigger current and the protection current, leading to a longer protection startup time. Furthermore, it necessitates widening the metal wire to address electron migration issues, increasing layout area and design complexity. In contrast, the power conversion device 10 (and its derived power conversion devices 70 and 80) of this invention significantly reduces the difference between the trigger current and the protection current through dynamic impedance adjustment, shortening the protection startup time and ensuring rapid protection and precise current control.

[0063] In summary, this invention provides a highly efficient and reliable power conversion device. Through dynamic impedance adjustment and precise control signal regulation, it significantly reduces the gap between the trigger current and protection current values, effectively shortening the protection start-up time and ensuring rapid protection and precise current control. Furthermore, the flexible design of this invention is applicable to different load conditions and application scenarios, eliminating the need for widened metal wires or complex layouts, effectively reducing the risk of electron migration and chip area. Compared to existing technologies, this invention offers significant advantages in the immediacy, reliability, and design simplicity of overcurrent protection, providing an innovative and practical solution for the field of power conversion devices.

Claims

1. A power conversion device, characterized in that, Includes: A gate driver is used to provide a drive signal to a gate node; A power transistor, with one gate electrically connected to the gate node, is used to convert an input voltage into an output voltage according to the drive signal, so as to provide a load current to a load. A sensing transistor, with one gate electrically connected to the gate node, is used to sense the load current to generate a sensing current related to the load current. A current-to-voltage conversion unit is electrically connected to the sensing transistor to convert the sensed current into a voltage signal according to an impedance value, and to adjust the impedance value according to a control signal. as well as A control circuit, electrically connected to the gate node and the current-to-voltage conversion unit, is used to generate the control signal and adjust the control signal according to the voltage signal to control the impedance value.

2. The power conversion device as described in claim 1, characterized in that, The control circuit determines that the voltage signal is higher than a reference signal, and then gradually adjusts a level of the control signal to change the impedance value.

3. The power conversion device as described in claim 2, characterized in that, At a point in time after the control circuit determines that the voltage signal is higher than the reference signal, the impedance value of the current-to-voltage conversion unit gradually switches from a first impedance value to a second impedance value.

4. The power conversion device as described in claim 3, characterized in that, The second impedance value is less than the first impedance value, and the time it takes for the first impedance value to gradually switch to the second impedance value is related to at least one characteristic of the load.

5. The power conversion device as described in claim 1, characterized in that, The control circuit is further used to adjust the gate voltage of the gate node based on the voltage signal generated by the current-to-voltage conversion unit.

6. The power conversion device as described in claim 1, characterized in that, The current-to-voltage conversion unit includes at least one resistor and at least one transistor. The control circuit controls the conduction level of the at least one transistor through the control signal to change the impedance value of the current-to-voltage conversion unit.

7. The power conversion device as claimed in claim 1, characterized in that, The control circuit includes: A voltage signal sensing unit is electrically connected to both ends of the current-to-voltage conversion unit to detect the voltage signal and output a detection result signal. A comparator is used to compare the detection result signal with a reference signal and generate a comparison result signal; A first transistor includes a drain electrically connected to the gate node, a gate receiving the input voltage, and a source; A second transistor includes a drain that generates and outputs the control signal, a gate that receives the comparison result signal, and a source; A first resistor is electrically connected between the source of the first transistor and the drain of the second transistor; as well as A second resistor is electrically connected between the source of the second transistor and a ground terminal.

8. The power conversion device as claimed in claim 1, characterized in that, The control circuit includes: A voltage signal sensing unit is electrically connected to both ends of the current-to-voltage conversion unit to detect the voltage signal and output a detection result signal. A comparator is used to compare the detection result signal with a reference signal and generate a comparison result signal; as well as A gate control module, coupled to the gate node, the comparator and the output voltage, is used to generate and output the control signal based on the comparison result signal, the gate voltage of the gate node and the output voltage.

9. The power conversion device as claimed in claim 1, characterized in that, The current-to-voltage conversion unit is electrically connected between a drain and an input terminal of the sensing transistor, the input terminal receiving the input voltage.

10. The power conversion device as claimed in claim 1, characterized in that, The current-to-voltage conversion unit is electrically connected between a source and an output terminal of the sensing transistor, and the output terminal outputs the output voltage.

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