Glucose Monitoring Systems

Abstract

To provide a method for calibration of a glucose sensor.SOLUTION: The method according to the present invention comprises: periodically measuring, by physical sensor electronics, electrode current (Isig) signals for a working electrode; performing, by a microcontroller, an Electrochemical Impedance Spectroscopy (EIS) procedure to generate ElS-related data for the working electrode; based on the Isig signals and ElS-related data and a plurality of calibration-free SG- predictive models, calculating a sensor glucose (SG) value for each of the SG-predictive models; calculating a SG variance estimate for the SG value; when an external blood glucose (BG) value is available, incorporating the BG value into the calculation of the SG value; fusing respective SG values from the plurality of SG- predictive models to obtain a single fused SG value; applying an unscented Kalman filter to the fused SG value; and calculating a calibrated SG value for a user.SELECTED DRAWING: Figure 126

Description

[Technical Field]

[0001] Embodiments of the present invention relate generally to sensor technology, including sensors and sensor devices used to sense various physiological parameters, such as glucose concentration. More specifically, embodiments of the present invention relate to the use of complex redundancy in glucose sensors, devices, and sensor systems, including optional calibration in calibration-free systems, devices, and methods, and closed-loop insulin infusion systems, as well as fusion algorithms, electrochemical impedance spectroscopy (EIS), and application-specific integrated circuits (ASICs) for implementing the use of such redundant glucose sensors, devices, and sensor systems. [Background technology]

[0002] Subjects (e.g., patients) and medical personnel desire to monitor indications of physiological conditions within the subject's body. Illustratively, a subject desires to continuously monitor blood glucose levels within the subject's body. Currently, patients can measure their blood glucose (BG) using a BG measuring device (i.e., glucose meter) such as a test strip meter, a continuous glucose measurement system (or continuous glucose monitor), or a hospital Hema-Q. BG measuring devices measure the patient's BG level using a variety of methods, such as a sample of the patient's blood, a sensor in contact with bodily fluids, an optical sensor, an enzymatic sensor, or a fluorescent / fluorescent quenching sensor. When the BG measuring device generates a BG reading, the reading is displayed on the BG measuring device.

[0003] Infusion pump devices and systems are relatively well known in the medical field for use in delivering or dispensing prescribed medications, such as insulin, to patients. In one form, such devices include a relatively compact pump housing adapted to accept a syringe or reservoir supporting the prescribed medication for administration to the patient via infusion tubing and an associated catheter or infusion set. A programmable controller allows the infusion pump to be operated continuously or periodically to obtain tightly controlled and accurate delivery of the medication over extended periods of time. Such infusion pumps are used to administer insulin and other medications, and exemplary pump structures are shown and described in U.S. Patent Nos. 5,629,995; 5,729,995; 5,729,995; and 5,729,995, which are incorporated herein by reference.

[0004] In diabetic individuals, each body has a baseline insulin need that can generally be maintained by administering a basal amount of insulin to the patient intermittently or continuously using an infusion pump. However, when additional glucose (i.e., above the basal level) appears in the diabetic individual's body, such as when the individual eats a meal, the amount and timing of insulin administered must be determined to adequately meet the additional glucose while avoiding excessive insulin infusion. Typically, a bolus amount of insulin is administered to compensate for the meal (i.e., meal bolus). It is common to determine the amount of insulin a diabetic patient may need to compensate for an expected meal based on the carbohydrate content of the meal.

[0005] Over the years, various electrochemical glucose sensors have been developed for use in providing an indication of blood glucose levels in diabetic patients. Such measurements are useful for monitoring and / or adjusting treatment regimens, which typically involve the patient receiving periodic insulin administration. Generally, small, flexible electrochemical sensors can be used to provide periodic readings over extended periods of time. In one form, flexible subcutaneous sensors are constructed according to thin-film mask technology. Representative thin-film sensors are described in U.S. Pat. Nos. 6,111,159; ... and 6,111,159, which are incorporated herein by reference.

[0006] These electrochemical sensors have been applied in telemetry characteristic monitor systems. For example, as described in U.S. Patent Application Publication No. 2009 / 0129990, the entire contents of which are incorporated herein by reference, the telemetry system includes a remotely located data receiving device, a sensor for generating a signal indicative of a characteristic of a user, and a transmitter device that processes the signal received from the sensor and wirelessly transmits the processed signal to the remotely located data receiving device. The data receiving device can be a characteristic monitor, a data receiver that provides data to another device, an RF programmer, a medication delivery device (such as an infusion pump), or the like.

[0007] Current continuous glucose measurement systems include subcutaneous (or short-term) sensors and implantable (or long-term) sensors. For each of the short-term and long-term sensors, the patient must wait a period of time for the continuous glucose sensor to stabilize and provide an accurate reading. With many continuous glucose sensors, the subject must wait three hours for the continuous glucose sensor to stabilize before a glucose measurement can be utilized. This is inconvenient for the patient and, in some cases, may cause the patient to not utilize the continuous glucose measurement system.

[0008] Furthermore, when a glucose sensor is first inserted into a patient's skin or subcutaneous layer, it does not operate in a stable state. The electrical readings from the sensor, which represent the patient's glucose level, vary over a wide range of readings. Previously, sensor stabilization took several hours. Techniques for sensor stabilization are detailed, for example, in U.S. Pat. No. 6,279,493, in which the initialization process for sensor stabilization can be shortened to approximately one hour. A high voltage (e.g., 1.0 to 1.2 volts) can be applied for one to two minutes to stabilize the sensor, and then a low voltage (e.g., 0.5 to 0.6 volts) can be applied for the remainder of the initialization process (e.g., approximately 58 minutes).

[0009] It is also desirable to thoroughly "wet" or hydrate the sensor electrodes before utilizing them. If the sensor electrodes are not sufficiently hydrated, the results may be an inaccurate indication of the patient's physiological state. Users of current blood glucose sensors may be instructed not to power on the sensor immediately. If they are utilized too quickly, such blood glucose sensors may not operate in an optimal or efficient manner.

[0010] Much of the current state of the art in continuous glucose monitoring (CGM) is primarily incidental, meaning that readings provided by CGM devices (e.g., including implantable or subcutaneous sensors) cannot be used without a reference value to make clinical decisions. Such reference values ​​must be obtained, for example, from a fingerstick using a BG meter. Reference values ​​are necessary because the amount of information available from the sensor / sensing components is limited. Specifically, only the raw sensor value (i.e., sensor current or Isig) and counter voltage may be provided by the sensing components for processing. Therefore, during analysis, if the raw sensor signal appears abnormal (e.g., the signal is decreasing), the only way to distinguish between a sensor failure and a physiological change in the user / patient (i.e., a change in glucose levels within the body) may be by obtaining a reference glucose value via a fingerstick. As is known, a reference fingerstick is also used to calibrate the sensor.

[0011] The art has sought ways to eliminate or at least minimize the number of fingersticks required for calibration and assessment of sensor health. However, given the number and complexity of numerous sensor failure modes, no satisfactory solution has been found. Diagnostics based either on direct assessment of the Isig or on the comparison of two Isigs have been developed. In either case, the Isig, by definition, is not analyte-independent because it tracks the level of glucose in the body. Therefore, the Isig itself is not a reliable source of sensor diagnostics or a predictor of continuous sensor performance.

[0012] Another limitation of the technology to date is the lack of sensor electronics that can not only run the sensor, but also manage sensor power, perform real-time diagnostics on the sensors and electrodes, and do so for redundant electrodes, redundant sensors, complementary sensors, and redundant and complementary sensors. Indeed, the concept of electrode redundancy has been around for some time. However, until now, there has been little success in using electrode redundancy (and / or complementary and redundant electrodes) to not only obtain more than one reading at a time, but also to assess the relative health of the redundant electrodes, the overall reliability of the sensor, and how often calibration reference values ​​are needed.

[0013] Additionally, even when redundant sensing electrodes are used, their number is typically limited to two. Again, this is due in part to the lack of sophisticated electronics to implement, evaluate, and manage a large number of independent working electrodes (e.g., up to five or more) in real time. Another reason, however, is the limited view that redundant electrodes are used to obtain "independent" sensor signals, for which two redundant electrodes are sufficient. As noted, this is one function of utilizing redundant electrodes, but it is not the only one. [Prior art documents] [Patent documents]

[0014] [Patent Document 1] U.S. Patent No. 4,562,751 [Patent Document 2] U.S. Patent No. 4,678,408 [Patent Document 3] U.S. Patent No. 4,685,903 [Patent Document 4] U.S. Patent No. 5,080,653 [Patent Document 5] U.S. Patent No. 5,097,122 [Patent Document 6] U.S. Patent No. 5,390,671 [Patent Document 7] U.S. Patent No. 5,391,250 [Patent Document 8] U.S. Patent No. 5,482,473 [Patent Document 9] U.S. Patent No. 5,586,553 [Patent Document 10] U.S. Patent No. 6,809,653 [Patent Document 11] U.S. Patent No. 4,573,994 Summary of the Invention

[0015] According to an embodiment of the present invention, an optional external calibration method of a calibration-free glucose sensor for measuring a level of glucose in a user's body, the glucose sensor including physical sensor electronics, a microcontroller, and a working electrode, the method including: periodically measuring, by the physical sensor electronics, an electrode current (Isig) signal of the working electrode; performing, by the microcontroller, an electrochemical impedance spectroscopy (EIS) procedure to generate EIS-related data for the working electrode; calculating, by the microcontroller, a respective sensor glucose (SG) value for each of the SG prediction models based on the Isig signals and the EIS-related data and a plurality of calibration-free SG prediction models; calculating, by the microcontroller, an SG variance estimate for each SG value; determining, by the microcontroller, whether external blood glucose (BG) values ​​are available, and if available, incorporating the BG values ​​into the calculation of the SG value; fusing, by the microcontroller, the respective SG values ​​from the plurality of SG prediction models to obtain a single fused SG value; applying, by the microcontroller, an unscented Kalman filter to the fused SG value; and calculating, by the microcontroller, a calibrated SG value that is displayed to the user.

[0016] According to another embodiment of the present invention, a glucose monitoring system includes a glucose sensor device for determining a glucose concentration in a user's body during a total sensor device wear time, the total sensor device wear time including a first time window, a subsequent second time window, and a transition period between the first time window and the second time window, the glucose sensor device including a first glucose sensor and a second glucose sensor, the first glucose sensor and the second glucose sensor having different characteristics in at least one of hydration, stabilization, and durability. The glucose sensor device further includes sensor electronics including at least one physical microprocessor configured to: (a) periodically receive, from the first glucose sensor, respective first output signals indicative of a glucose concentration level within the user's body; (b) calculate a glucose concentration level within the user's body based entirely on the first output signals during a first time window; (c) periodically receive, from the second glucose sensor, respective second output signals indicative of a glucose concentration level within the user's body; (d) calculate a glucose concentration level within the user's body based entirely on both the first output signals and the second output signals during a transition period; and (e) calculate a glucose concentration level within the user's body based entirely on the second output signals during a second time window. [Brief explanation of the drawings]

[0017] [Figure 1] 1A and 1B are perspective views of a subcutaneous sensor insertion set and a block diagram of a sensor electronics device according to an embodiment of the present invention. [Figure 2A] 1 shows a substrate having two sides, a first side containing an electrode arrangement and a second side containing electronic circuitry. [Figure 2B] 1 shows a general block diagram of an electronic circuit for sensing the output of a sensor. [Figure 3] 1 shows a block diagram of a sensor electronics device and a sensor including multiple electrodes, according to an embodiment of the present invention. [Figure 4] 1 illustrates an alternative embodiment of the present invention including a sensor and a sensor electronics device according to an embodiment of the present invention. [Figure 5] 1 shows an electronic block diagram of a sensor electrode and voltages applied to the sensor electrode, according to an embodiment of the present invention. [Figure 6A] 10 illustrates a method of applying pulses during a stabilization time frame to shorten the stabilization time frame, according to an embodiment of the present invention. [Figure 6B] 1 illustrates a method for stabilizing a sensor according to an embodiment of the present invention. [Figure 6C] 10 illustrates the use of feedback in stabilizing a sensor, according to an embodiment of the present invention. [Figure 7] 10 illustrates the effect of stabilizing a sensor according to an embodiment of the present invention. [Figure 8A] 1 shows a block diagram of a sensor electronics device and a sensor including a voltage generating device according to an embodiment of the present invention. [Figure 8B] 1 shows a voltage generation device for implementing this embodiment of the present invention. [Figure 8C] 1 illustrates a voltage generating device for generating two voltage values ​​according to an embodiment of the present invention. [Figure 8D] 1 illustrates a voltage generation device having three voltage generation systems according to an embodiment of the present invention. [Figure 9A] 1 illustrates a sensor electronics device including a microcontroller for generating voltage pulses according to an embodiment of the present invention. [Figure 9B] 1 illustrates a sensor electronics device including an analysis module according to an embodiment of the present invention. [Figure 10] FIG. 1 shows a block diagram of a sensor system including hydration electronics, according to an embodiment of the present invention. [Figure 11] 10 illustrates an embodiment of the present invention that includes a mechanical switch to assist in determining hydration time. [Figure 12] 1 illustrates a method for detecting hydration, according to an embodiment of the present invention. [Figure 13A] 1 illustrates a method for hydrating a sensor according to an embodiment of the present invention. [Figure 13B]10 illustrates an additional method for verifying sensor hydration, according to an embodiment of the present invention. [Figure 14A] 1 illustrates a method for combining sensor hydration and sensor stabilization according to an embodiment of the present invention. [Figure 14B] 1 illustrates a method for combining sensor hydration and sensor stabilization according to an embodiment of the present invention. [Figure 14C] 1 illustrates a method for combining sensor hydration and sensor stabilization according to an embodiment of the present invention. [Figure 15A] 1 illustrates an EIS-based analysis of a system response to the application of a periodic AC signal, according to an embodiment of the present invention. [Figure 15B] 1 shows a known circuit model for electrochemical impedance spectroscopy. [Figure 16A] 1 shows an example of a Nyquist plot for a selected frequency spectrum from 0.1 Hz to 1000 MHz where a superposition of an AC voltage and a DC voltage (DC bias) is applied to a working electrode according to an embodiment of the present invention. [Figure 16B] 1 shows another example of a Nyquist plot with a linear fit at relatively low frequencies and an intercept that approximates the value of the real impedance at relatively high frequencies. [Figure 16C] 1 shows the infinite and finite glucose sensor responses to a sinusoidal operating potential, respectively. [Figure 16D] 1 shows the infinite and finite glucose sensor responses to a sinusoidal operating potential, respectively. [Figure 16E] 1 illustrates a magnitude Bode plot in accordance with an embodiment of the present invention. [Figure 16F] 1 illustrates a Bode plot of phase in accordance with an embodiment of the present invention. [Figure 17] 1 shows a Nyquist plot of sensor impedance as the sensor ages, in accordance with an embodiment of the present invention. [Figure 18] 1 illustrates the application of EIS techniques in stabilizing and detecting sensor aging, according to an embodiment of the present invention. [Figure 19]1 illustrates a timeline for performing an EIS procedure according to an embodiment of the present invention. [Figure 20] 10 illustrates a method for detecting and repairing sensors using EIS procedures in conjunction with repair actions, according to an embodiment of the present invention. [Figure 21A] 10 illustrates an example of a sensor remedial action according to an embodiment of the present invention. [Figure 21B] 10 illustrates an example of a sensor remedial action according to an embodiment of the present invention. [Figure 22] 1 shows a Nyquist plot of a properly functioning sensor, with a gradually increasing Nyquist slope and a gradually decreasing intercept as the sensor wear time progresses. [Figure 23A] 1 shows the raw current signals (Isig) from two redundant working electrodes and the real impedance of each of the electrodes at 1 kHz, according to an embodiment of the present invention. [Figure 23B] 23B shows the Nyquist plot of the first working electrode (WE1) of FIG. 23A. [Figure 23C] FIG. 23B shows the Nyquist plot of the second working electrode (WE2) of FIG. 23A. [Figure 24] 1 shows an example of a signal dip for two redundant working electrodes and the real impedance of each of the electrodes at 1 kHz, according to an embodiment of the present invention. [Figure 25A] 10 illustrates substantial glucose independence of real impedance, imaginary impedance, and phase at relatively high frequencies for a well-functioning glucose sensor in accordance with an embodiment of the present invention. [Figure 25B] 10 shows illustrative examples of various levels of glucose dependence of real impedance at relatively low frequencies, according to embodiments of the present invention. [Figure 25C] 10 shows illustrative examples of various levels of glucose dependence of phase at relatively low frequencies, according to embodiments of the present invention. [Figure 26] 10 illustrates trends in 1 kHz real impedance, 1 kHz imaginary impedance, and higher frequency phase as a glucose sensor loses sensitivity as a result of oxygen depletion at the sensor insertion site, according to an embodiment of the present invention. [Figure 27] 10 shows the Isig and phase of an in vitro simulation of oxygen starvation at different glucose concentrations, according to an embodiment of the present invention. [Figure 28A] 10 shows an example of sensitivity loss caused by oxygen depletion with redundant working electrodes WE1 and WE2 and EIS-based parameters of the electrodes, according to an embodiment of the present invention. [Figure 28B] 10 shows an example of sensitivity loss caused by oxygen depletion with redundant working electrodes WE1 and WE2 and EIS-based parameters of the electrodes, according to an embodiment of the present invention. [Figure 28C] 10 shows an example of sensitivity loss caused by oxygen depletion with redundant working electrodes WE1 and WE2 and EIS-based parameters of the electrodes, according to an embodiment of the present invention. [Figure 28D] Figures 28A-28C show EIS-induced spikes in the raw Isig for the example. [Figure 29] 10 shows an example of sensitivity loss due to oxygen depletion caused by an occlusion, according to an embodiment of the present invention. [Figure 30A] 10 shows an example of sensitivity loss due to biofouling with redundant working electrodes WE1 and WE2 and EIS-based parameters of the electrodes, according to an embodiment of the present invention. [Figure 30B] 10 shows an example of sensitivity loss due to biofouling with redundant working electrodes WE1 and WE2 and EIS-based parameters of the electrodes, according to an embodiment of the present invention. [Figure 30C] 10 shows an example of sensitivity loss due to biofouling with redundant working electrodes WE1 and WE2 and EIS-based parameters of the electrodes, according to an embodiment of the present invention. [Figure 30D] Figures 30A-30C show EIS-induced spikes in the example raw Isig. [Figure 31] 1 illustrates a diagnostic procedure for sensor fault detection according to an embodiment of the present invention. [Figure 32A] 10 illustrates another diagnostic procedure for detecting sensor faults, according to an embodiment of the present invention. [Figure 32B]10 illustrates another diagnostic procedure for detecting sensor faults, according to an embodiment of the present invention. [Figure 33A] 1 shows a top-level flowchart with the current (Isig)-based fusion algorithm, according to an embodiment of the present invention. [Figure 33B] 1 shows a top-level flowchart with a sensor glucose (SG) based fusion algorithm, according to an embodiment of the present invention. [Figure 34] FIG. 33C shows details of the sensor glucose (SG)-based fusion algorithm of FIG. 33B, according to an embodiment of the present invention. [Figure 35] 33B shows details of the current (Isig)-based fusion algorithm of FIG. 33A, according to an embodiment of the present invention. [Figure 36] FIG. 10 is an illustration of steady-state sensor calibration according to an embodiment of the present invention. [Figure 37] FIG. 10 is an illustration of calibration of a sensor during a transition according to an embodiment of the present invention. [Figure 38A] FIG. 10 is an illustration of an EIS-based dynamic gradient (with gradient adjustment) according to an embodiment of the present invention for sensor calibration. [Figure 38B] 10 illustrates an EIS-assisted sensor calibration flowchart with low actuation detection in accordance with an embodiment of the present invention. [Figure 39] 10 shows the sensor current (Isig) and impedance magnitude at 1 kHz for an in vitro simulation of an interferent in proximity to a sensor, according to an embodiment of the present invention. [Figure 40A] The Bode diagrams of the phase and impedance of the simulation shown in FIG. 39 are shown. [Figure 40B] The Bode diagrams of the phase and impedance of the simulation shown in FIG. 39 are shown. [Figure 40C] 39 shows the Nyquist plot of the simulation shown in FIG. [Figure 41] 10 shows another in vitro simulation using an interferent, according to an embodiment of the present invention. [Figure 42A]1 shows an ASIC block diagram according to an embodiment of the present invention. [Figure 42B] 1 shows an ASIC block diagram according to an embodiment of the present invention. [Figure 43] 1 illustrates a potentiostat configuration for a sensor with redundant working electrodes, according to an embodiment of the present invention. [Figure 44] 44 shows the equivalent AC interelectrode circuit for the potentiostat-configured sensor shown in FIG. 43. [Figure 45] 1 illustrates some of the major blocks of EIS circuitry in an analog front-end IC of a glucose sensor, in accordance with an embodiment of the present invention. [Figure 46A] 46 shows a simulation of the signal of the EIS circuit shown in FIG. 45 for a current of 0 degrees phase multiplied by 0 degrees phase. [Figure 46B] 46 shows a simulation of the signal of the EIS circuit shown in FIG. 45 for a current of 0 degrees phase multiplied by 0 degrees phase. [Figure 46C] 46 shows a simulation of the signal of the EIS circuit shown in FIG. 45 for a current of 0 degrees phase multiplied by 0 degrees phase. [Figure 46D] 46 shows a simulation of the signal of the EIS circuit shown in FIG. 45 for a current of 0 degrees phase multiplied by 0 degrees phase. [Figure 46E] 46 shows a simulation of the signal of the EIS circuit shown in FIG. 45 for a current of 0 degrees phase multiplied by 0 degrees phase. [Figure 46F] 46 shows a simulation of the signal of the EIS circuit shown in FIG. 45 for a current of 0 degrees phase multiplied by 0 degrees phase. [Figure 47A] 46 shows a simulation of the signal of the EIS circuit shown in FIG. 45 for a current of 0 degrees phase multiplied by 90 degrees phase. [Figure 47B] 46 shows a simulation of the signal of the EIS circuit shown in FIG. 45 for a current of 0 degrees phase multiplied by 90 degrees phase. [Figure 47C] 46 shows a simulation of the signal of the EIS circuit shown in FIG. 45 for a current of 0 degrees phase multiplied by 90 degrees phase. [Figure 47D]46 shows a simulation of the signal of the EIS circuit shown in FIG. 45 for a current of 0 degrees phase multiplied by 90 degrees phase. [Figure 47E] 46 shows a simulation of the signal of the EIS circuit shown in FIG. 45 for a current of 0 degrees phase multiplied by 90 degrees phase. [Figure 47F] 46 shows a simulation of the signal of the EIS circuit shown in FIG. 45 for a current of 0 degrees phase multiplied by 90 degrees phase. [Figure 48] 1 illustrates a circuit model according to an embodiment of the present invention. [Figure 49A] 1 shows an illustration of a circuit model according to another embodiment of the present invention; [Figure 49B] 1 shows an illustration of a circuit model according to another embodiment of the present invention; [Figure 49C] 1 shows an illustration of a circuit model according to another embodiment of the present invention; [Figure 50A] 1 is a Nyquist plot overlaid with an equivalent circuit simulation, in accordance with an embodiment of the present invention. [Figure 50B] FIG. 50B is an enlarged view of the high frequency portion of FIG. 50A. [Figure 51] 1 shows a Nyquist plot of increasing Cdl in the direction of arrow A in accordance with an embodiment of the present invention. [Figure 52] 1 shows a Nyquist plot with increasing α in the direction of arrow A, in accordance with an embodiment of the present invention. [Figure 53] 1 shows a Nyquist plot of increasing Rp in the direction of arrow A in accordance with an embodiment of the present invention. [Figure 54] 1 shows a Nyquist plot of increasing Warburg admittance in the direction of arrow A in accordance with an embodiment of the present invention. [Figure 55] 1 shows a Nyquist plot with increasing λ in the direction of arrow A, in accordance with an embodiment of the present invention. [Figure 56] 10 illustrates the effect of membrane capacitance on a Nyquist plot, in accordance with an embodiment of the present invention. [Figure 57] 1 shows a Nyquist plot of increasing membrane resistance in the direction of arrow A, according to an embodiment of the present invention. [Figure 58] 1 shows a Nyquist plot of increasing Rsol in the direction of arrow A in accordance with an embodiment of the present invention. [Figure 59A] 10 illustrates the changes in EIS parameters for circuit elements during start-up and calibration in accordance with an embodiment of the present invention. [Figure 59B] 10 illustrates the changes in EIS parameters for circuit elements during start-up and calibration in accordance with an embodiment of the present invention. [Figure 59C] 10 illustrates the changes in EIS parameters for circuit elements during start-up and calibration in accordance with an embodiment of the present invention. [Figure 60A] 10A-10C illustrate the changes in different sets of EIS parameters associated with circuit elements during start-up and calibration in accordance with an embodiment of the present invention. [Figure 60B] 10A-10C illustrate the changes in different sets of EIS parameters associated with circuit elements during start-up and calibration in accordance with an embodiment of the present invention. [Figure 60C] 10A-10C illustrate the changes in different sets of EIS parameters associated with circuit elements during start-up and calibration in accordance with an embodiment of the present invention. [Figure 61A] 10 illustrates the change in yet another set of EIS parameters associated with circuit elements during start-up and calibration in accordance with an embodiment of the present invention. [Figure 61B] 10 illustrates the change in yet another set of EIS parameters associated with circuit elements during start-up and calibration in accordance with an embodiment of the present invention. [Figure 61C] 10 illustrates the change in yet another set of EIS parameters associated with circuit elements during start-up and calibration in accordance with an embodiment of the present invention. [Figure 62] 10 shows EIS responses of multiple electrodes according to an embodiment of the present invention. [Figure 63] 10 is a Nyquist plot illustrating the effect of Isig calibration with increasing glucose, in accordance with an embodiment of the present invention. [Figure 64] 10 illustrates the effect of oxygen (Vcntr) response on a Nyquist plot in accordance with an embodiment of the present invention. [Figure 65] 1 illustrates a shift in a Nyquist plot due to temperature changes, in accordance with an embodiment of the present invention. [Figure 66] 1 illustrates the relationship between Isig and blood glucose according to an embodiment of the present invention. [Figure 67A] 1 illustrates sensor drift according to an embodiment of the present invention. [Figure 67B] 1 illustrates sensor drift according to an embodiment of the present invention. [Figure 68] 10 illustrates an increase in membrane resistance during sensitivity loss according to an embodiment of the present invention. [Figure 69] 10 illustrates the degradation of Warburg admittance during sensitivity loss according to an embodiment of the present invention. [Figure 70] 1 shows a calibration curve according to an embodiment of the present invention. [Figure 71] 1 illustrates a high frequency semicircle appearing on a Nyquist plot in accordance with an embodiment of the present invention. [Figure 72A] 10 illustrates Vcntr rail and Cdl reduction according to an embodiment of the present invention. [Figure 72B] 10 illustrates Vcntr rail and Cdl reduction according to an embodiment of the present invention. [Figure 73] 10 illustrates the varying slope of a calibration curve according to an embodiment of the present invention. [Figure 74] 1 illustrates varying lengths of Nyquist plots in accordance with an embodiment of the present invention. [Figure 75] 75 shows an enlarged view of the low and high frequency regions of the Nyquist plot of FIG. 74. [Figure 76A] 10 illustrates the combined effect of increasing membrane resistance, decreasing Cdl, and Vcntr rail according to an embodiment of the present invention. [Figure 76B] 10 illustrates the combined effect of increasing membrane resistance, decreasing Cdl, and Vcntr rail according to an embodiment of the present invention. [Figure 77] 1 shows relative Cdl values ​​for two working electrodes according to an embodiment of the present invention. [Figure 78] 1 shows relative Rp values ​​for two working electrodes according to an embodiment of the present invention. [Figure 79] 10 illustrates the combined effect of varying EIS parameters on a calibration curve, in accordance with an embodiment of the present invention. [Figure 80] 1 shows that the length of the Nyquist plot in the low frequency region becomes longer when there is a sensitivity loss, in accordance with an embodiment of the present invention. [Figure 81] FIG. 10 is a flow diagram for self-calibration of a sensor based on detection of sensitivity changes, according to an embodiment of the present invention. [Figure 82] 10 illustrates a horizontal shift in a Nyquist plot as a result of sensitivity loss, in accordance with an embodiment of the present invention. [Figure 83] 1 illustrates a method for developing a heuristic EIS metric based on a Nyquist plot, according to an embodiment of the present invention. [Figure 84] 10 illustrates the relationship between Rm and a calibration factor according to an embodiment of the present invention. [Figure 85] 10 illustrates the relationship between Rm and normalized Isig in accordance with an embodiment of the present invention. [Figure 86] 10 shows Isig plots of various glucose levels as a function of time according to an embodiment of the present invention. [Figure 87] 10 shows Cdl plots of various glucose levels as a function of time according to an embodiment of the present invention. [Figure 88] 87 illustrates a second inflection point of the plot of FIG. 86 in accordance with an embodiment of the present invention. [Figure 89] FIG. 89 illustrates the second inflection point in Rm corresponding to the peak in FIG. 88, in accordance with an embodiment of the present invention. [Figure 90] FIG. 1 shows an illustration of the relationship between calibration factor (CF) and Rmem+Rsol, in accordance with an embodiment of the present invention. [Figure 91A] 10 is a chart showing in vivo results of MARD across all valid BGs in the first approximately 8 hours of sensor life, according to an embodiment of the present invention. [Figure 91B] 10 is a chart showing the median ARD number across all valid BGs in approximately the first 8 hours of sensor life, according to an embodiment of the present invention. [Figure 92A] 10 illustrates a calibration factor adjustment according to an embodiment of the present invention. [Figure 92B]10 illustrates a calibration factor adjustment according to an embodiment of the present invention. [Figure 92C] 10 illustrates a calibration factor adjustment according to an embodiment of the present invention. [Figure 93A] 10 illustrates a calibration factor adjustment according to an embodiment of the present invention. [Figure 93B] 10 illustrates a calibration factor adjustment according to an embodiment of the present invention. [Figure 93C] 10 illustrates a calibration factor adjustment according to an embodiment of the present invention. [Figure 94A] 10 illustrates a calibration factor adjustment according to an embodiment of the present invention. [Figure 94B] 10 illustrates a calibration factor adjustment according to an embodiment of the present invention. [Figure 94C] 10 illustrates a calibration factor adjustment according to an embodiment of the present invention. [Figure 95] 10 shows an illustrative example of an initial decay in Cdl, according to an embodiment of the present invention. [Figure 96] 10 illustrates the effect on Isig of removing non-faradaic currents in accordance with an embodiment of the present invention. [Figure 97A] 10 shows the calibration factors before removing non-faradaic currents for the two working electrodes according to an embodiment of the present invention. [Figure 97B] 10 shows the calibration factor after removing the non-faradaic currents of the two working electrodes according to an embodiment of the present invention. [Figure 98A] 10 illustrates the effect of removing non-faradaic currents on a MARD, according to an embodiment of the present invention. [Figure 98B] 10 illustrates the effect of removing non-faradaic currents on a MARD, according to an embodiment of the present invention. [Figure 99] FIG. 1 is an illustration of double layer capacitance over time according to an embodiment of the present invention. [Figure 100] 10 illustrates the shift of Rmem+Rsol during sensitivity loss and the appearance of a high frequency semicircle according to an embodiment of the present invention. [Figure 101A] 1 illustrates a flow diagram for detecting sensitivity loss using combinational logic, according to one embodiment of the present invention. [Figure 101B]FIG. 10 shows a flow diagram for detecting sensitivity loss using combinatorial logic according to another embodiment of the present invention. [Figure 102] 1 illustrates an exemplary method for distinguishing between new and used sensors using the Nyquist slope as a marker, according to an embodiment of the present invention. [Figure 103A] 10 illustrates examples of Nyquist plots having different lengths for different sensor configurations, in accordance with embodiments of the present invention. [Figure 103B] 10 illustrates examples of Nyquist plots having different lengths for different sensor configurations, in accordance with embodiments of the present invention. [Figure 103C] 10 illustrates examples of Nyquist plots having different lengths for different sensor configurations, in accordance with embodiments of the present invention. [Figure 104] The Nyquist plot length is shown as a function of time for the sensors of Figures 103A-103C. [Figure 105] FIG. 10 shows a flow diagram for blanking sensor data or terminating a sensor according to an embodiment of the present invention. [Figure 106] 10 shows a flow diagram for sensor termination in accordance with an embodiment of the present invention. [Figure 107] 1 shows a flow diagram of signal dip detection according to an embodiment of the present invention; [Figure 108A] Isig and Vcntr are shown as functions of time. [Figure 108B] According to an embodiment of the present invention, glucose is shown as a function of time. [Figure 109A] Calibration ratio as a function of time. [Figure 109B] According to an embodiment of the present invention, glucose is shown as a function of time. [Figure 110A] In accordance with an embodiment of the present invention, the trend of the calibration factor is shown as a function of time. [Figure 110B] In accordance with an embodiment of the present invention, the trend of the calibration factor is shown as a function of time. [Figure 111] 1 illustrates a flow diagram for first day calibration (FDC) in accordance with an embodiment of the present invention. [Figure 112] FIG. 1 shows a flow diagram of an EIS-based calibration in accordance with an embodiment of the present invention. [Figure 113] 1 shows a flow diagram of an existing calibration method. [Figure 114] 1 shows a calibration flow diagram according to an embodiment of the present invention. [Figure 115] 10 shows a calibration flow diagram according to another embodiment of the present invention. [Figure 116] 10 shows a calibration flow diagram according to yet another embodiment of the present invention. [Figure 117] 10 shows a calibration flow diagram according to another embodiment of the present invention. [Figure 118] 1 shows a table of comparative MARD values ​​calculated according to an embodiment of the present invention. [Figure 119] 1 shows a flow diagram for calculating raw fusion weights according to an embodiment of the present invention; [Figure 120] 1 illustrates a sensor glucose (SG) fusion logic diagram according to an embodiment of the present invention. [Figure 121] 1 shows a flow diagram of a calibration-free retrospective algorithm according to an embodiment of the present invention. [Figure 122] 1 illustrates a decision tree model according to an embodiment of the present invention. [Figure 123] 1 illustrates a decision tree model for blanking data according to an embodiment of the present invention. [Figure 124] 1 is a table illustrating example parameters of a blanking algorithm, according to an embodiment of the present invention. [Figure 125] 10 illustrates fusion, filtering, and blanking results according to an embodiment of the present invention. [Figure 126] 10 shows a flow diagram of optional calibration logic according to an embodiment of the present invention. [Figure 127] 1 shows a comparison table of two different glucose sensor designs. [Figure 128] 1 illustrates an example of complex redundancy according to an embodiment of the present invention. [Figure 129] FIG. 1 shows a block diagram including a calibrated model and an uncalibrated model according to an embodiment of the present invention. [Figure 130] 1 shows a diagram of fusion logic according to an embodiment of the present invention; [Figure 131] 1 shows a diagram of one calibrated model and one uncalibrated model fusion logic according to an embodiment of the present invention. [Figure 132] 1 shows a diagram of fusion logic using two uncalibrated models according to an embodiment of the present invention. [Figure 133] 1 shows a diagram of fusion logic using two calibrated models according to an embodiment of the present invention; [Figure 134] 1 shows a diagram of fusion logic using multiple calibrated models and / or multiple uncalibrated models according to an embodiment of the present invention. DETAILED DESCRIPTION OF THE INVENTION

[0018] In the following description, reference is made to the accompanying drawings, which form a part hereof and which illustrate several embodiments of the present invention. It is understood that other embodiments may be utilized and structural and operational changes may be made without departing from the scope of the present invention. A detailed description of embodiments of the present invention will be made with reference to the accompanying drawings, wherein like numerals indicate corresponding parts therein.

[0019] The inventions herein are described below with reference to flowchart illustrations of methods, systems, devices, apparatus, and programming and computer program products. It will be understood that each block of the flowchart illustrations, and combinations of blocks in the flowchart illustrations (as well as any menu screens depicted in the illustrations), can be implemented by programming instructions comprising computer program instructions. These computer program instructions may be loaded into a computer or other programmable data processing apparatus (such as a controller, microcontroller, or processor within a sensor electronics device) to generate a machine such that the instructions, executed on the computer or other programmable data processing apparatus, create instructions for implementing the functions specified in the flowchart block or blocks. These computer program instructions may also be stored in a computer-readable memory that can instruct the computer or other programmable data processing apparatus to function in a particular manner, such that the instructions stored in the computer-readable memory generate a product including instructions that implement the functions specified in the flowchart block or blocks. Computer program instructions may also be loaded into a computer or other programmable data processing apparatus to cause the computer or other programmable apparatus to perform a series of operational steps, such that the instructions, executed on the computer or other programmable apparatus, provide steps for implementing the functions specified in the flowchart block or blocks and / or menus presented herein. The programming instructions may also be stored in and / or implemented in electronic circuits, including integrated circuits (ICs) and application specific integrated circuits (ASICs) used with sensor devices, apparatus, and systems.

[0020] FIG. 1 is a perspective view of a subcutaneous sensor insertion set and a block diagram of a sensor electronics device according to an embodiment of the present invention. As illustrated in FIG. 1, a subcutaneous sensor set 10 is provided for subcutaneous placement of an active portion, such as a flexible sensor 12 (see, e.g., FIG. 2), at a selected site on a user's body. The subcutaneous or percutaneous portion of the sensor set 10 includes a hollow, slotted insertion needle 14 and a cannula 16. The needle 14 is used to facilitate quick and easy subcutaneous placement of the cannula 16 at the subcutaneous insertion site. Inside the cannula 16 is the sensing portion 18 of the sensor 12, which exposes one or more sensor electrodes 20 to the user's bodily fluids through a window 22 formed in the cannula 16. In an embodiment of the present invention, the one or more sensor electrodes 20 may include a counter electrode, a reference electrode, and one or more working electrodes. After insertion, the insertion needle 14 is withdrawn, leaving the cannula 16 with the sensing portion 18 and sensor electrodes 20 in place at the selected insertion site.

[0021] In certain embodiments, the subcutaneous sensor set 10 facilitates precise placement of a flexible thin-film electrochemical sensor 12 of the type used to monitor specific blood parameters indicative of a user's condition. The sensor 12 monitors glucose levels within the body and may be used in conjunction with an external or implantable automatic or semi-automatic drug infusion pump to control the delivery of insulin to a diabetic patient, as described, for example, in U.S. Pat. Nos. 5,629,592; 5,729,630; 5,729,592; or 5,729,592.

[0022] Certain embodiments of flexible electrochemical sensor 12 are constructed according to thin-film mask technology and include an elongated thin-film conductor embedded or encased between layers and membranes of a selected insulating material, such as a polyimide film or sheet. A sensor electrode 20 at the tip of sensing portion 18 is exposed through one of the insulating layers for direct contact with the patient's blood or other bodily fluids when sensing portion 18 (or active portion) of sensor 12 is subcutaneously positioned at an insertion site. Sensing portion 18 is bonded to a connecting portion 24, terminating in a conductive contact pad or the like, also exposed through one of the insulating layers. In alternative embodiments, other types of implantable sensors, such as chemical-based, optical-based, etc., may be used.

[0023] As is known in the art, the connecting portion 24 and contact pads are generally adapted for direct wired electrical connection to a suitable monitor or sensor electronics device 100 that monitors a user's condition in response to signals derived from the sensor electrodes 20. Further description of this general type of flexible thin film sensor can be found, for example, in U.S. Patent No. 6,277,999, which is incorporated herein by reference. The connecting portion 24 may be conveniently connected to the monitor or sensor electronics device 100 electrically or by a connector block 28 (or the like), for example, as shown and described in U.S. Patent No. 6,277,999, which is incorporated herein by reference. Thus, according to embodiments of the present invention, the subcutaneous sensor set 10 may be configured or formed to cooperate with either a wired or a wireless characteristic monitoring system.

[0024] The sensor electrode 20 can be used in a variety of sensing applications and can be configured in a variety of ways. For example, the sensor electrode 20 can be used in physiological parameter sensing applications where some type of biomolecule is used as a catalytic agent. For example, the sensor electrode 20 may be used in a glucose and oxygen sensor having a glucose oxidase (GOx) enzyme catalyzing a reaction with the sensor electrode 20. This reaction produces gluconic acid (CH) in proportion to the amount of glucose present. 12 O7) and hydrogen peroxide (H2O2) are produced.

[0025] The sensor electrode 20, along with the biomolecule or some other catalytic agent, may be placed in the human body in a vascular or non-vascular environment. For example, the sensor electrode 20 and biomolecule may be placed intravenously and exposed to the bloodstream, or may be placed subcutaneously or in the peritoneal region of the human body.

[0026] Monitor 100 may also be referred to as sensor electronics device 100. Monitor 100 may include a power supply 110, a sensor interface 122, processing electronics 124, and data formatting electronics 128. Monitor 100 may be coupled to sensor set 10 by cable 102 via a connector electrically coupled to connector block 28 of connection portion 24. In another embodiment, the cable may be omitted. In this embodiment of the invention, monitor 100 may include a suitable connector for direct connection to connection portion 104 of sensor set 10. Sensor set 10 may be modified to allow connector portion 104 to be in a different location, e.g., to facilitate placement of monitor 100 over the sensor set.

[0027] In an embodiment of the present invention, the sensor interface 122, the processing electronics 124, and the data formatting electronics 128 are formed as separate semiconductor chips, although alternative embodiments may combine various semiconductor chips into a single or multiple customized semiconductor chips. The sensor interface 122 connects to a cable 102 that is connected to the sensor set 10.

[0028] The power supply 110 may be a battery. The battery may include three series-connected silver oxide 357 battery cells. Alternative embodiments may utilize different battery chemistries, such as lithium-based chemistries, alkaline batteries, or nickel-metal hydrides, and may use different numbers of batteries. The monitor 100 provides power to the sensor set 10 via the power supply 110 via the cable 102 and cable connector 104. In embodiments of the present invention, the power supply is a voltage provided to the sensor set 10. In embodiments of the present invention, the power supply is a current provided to the sensor set 10. In embodiments of the present invention, the power supply is a voltage provided to the sensor set 10 at a particular voltage.

[0029] 2A and 2B illustrate an implantable sensor and electronics for driving the implantable sensor, according to an embodiment of the present invention. FIG. 2A shows a substrate 220 having two sides, a first side 222 of which contains an electrode configuration and a second side 224 of which contains electronic circuitry. As seen in FIG. 2A, the first side 222 of the substrate includes two counter-working electrode pairs 240, 242, 244, and 246 on either side of a reference electrode 248. The second side 224 of the substrate includes the electronic circuitry. As shown, the electronic circuitry may be enclosed in a hermetically sealed casing 226, providing a protective housing for the electronic circuitry. This allows the sensor substrate 220 to be inserted into a vascular environment or other environment that may expose the electronic circuitry to fluids. Encapsulating the electronic circuitry within the hermetically sealed casing 226 allows the electronic circuitry to operate without risk of shorting due to surrounding fluids. FIG. 2A also shows pads 228 to which input and output lines of the electronic circuitry may be connected. The electronic circuitry itself can be fabricated in a variety of ways. According to embodiments of the present invention, the electronic circuitry can be fabricated as an integrated circuit using techniques common in the industry.

[0030] 2B illustrates a general block diagram of an electronic circuit for sensing the output of a sensor, according to an embodiment of the present invention. At least one pair of sensor electrodes 310 can be interfaced to a data converter 312, the output of which can be interfaced to a counter 314. The counter 314 can be controlled by control logic 316. The output of the counter 314 can be connected to a line interface 318. The line interface 318 can be connected to input and output lines 320, which can also be connected to the control logic 316. The input and output lines 320 can also be connected to a power rectifier 322.

[0031] The sensor electrode 310 can be used in a variety of sensing applications and can be configured in a variety of ways. For example, the sensor electrode 310 can be used in physiological parameter sensing applications where some type of biomolecule is used as a catalytic agent. For example, the sensor electrode 310 may be used in a glucose and oxygen sensor having a glucose oxidase (GOx) enzyme catalyzing a reaction with the sensor electrode 310. The sensor electrode 310, along with the biomolecule or some other catalytic agent, may be placed in the human body in a vascular or non-vascular environment. For example, the sensor electrode 310 and the biomolecule may be placed intravenously and exposed to the bloodstream.

[0032] FIG. 3 illustrates a block diagram of a sensor electronics device and sensor including multiple electrodes according to an embodiment of the present invention. Sensor set or system 350 includes sensor 355 and sensor electronics device 360. Sensor 355 includes counter electrode 365, reference electrode 370, and working electrode 375. Sensor electronics device 360 ​​includes power supply 380, regulator 385, signal processor 390, measurement processor 395, and display / transmission module 397. Power supply 380 provides power (either in the form of voltage, current, or voltage including current) to regulator 385. Regulator 385 transmits the regulated voltage to sensor 355. In an embodiment of the present invention, regulator 385 transmits a voltage to counter electrode 365 of sensor 355.

[0033] The sensor 355 generates a sensor signal indicative of the concentration of the physiological characteristic being measured. For example, the sensor signal may indicate a blood glucose reading. In embodiments of the invention utilizing a subcutaneous sensor, the sensor signal may represent the level of hydrogen peroxide in the subject. In embodiments of the invention utilizing a blood or cranial sensor, the amount of oxygen is measured by the sensor and represented by the sensor signal. In embodiments of the invention utilizing an implantable or long-term sensor, the sensor signal may represent the oxygen level of the subject. The sensor signal may be measured at the working electrode 375. In embodiments of the invention, the sensor signal may be a current measured at the working electrode. In embodiments of the invention, the sensor signal may be a voltage measured at the working electrode.

[0034] Signal processor 390 receives the sensor signal (e.g., a measured current or voltage) after it is measured at sensor 355 (e.g., a working electrode). Signal processor 390 processes the sensor signal and generates a processed sensor signal. Measurement processor 395 receives the processed sensor signal and utilizes reference values ​​to calibrate the processed sensor signal. In an embodiment of the invention, the reference values ​​are stored in a reference memory and provided to measurement processor 395. Measurement processor 395 generates sensor measurements. The sensor measurements may be stored in a measurement memory (not shown). The sensor measurements may be displayed on a display within the housing with the sensor electronics or sent to a display / transmission device for transmission to an external device.

[0035] The sensor electronics device 360 ​​may be a monitor including a display for displaying an indication of a physiological characteristic. The sensor electronics device 360 ​​may also be installed in a desktop computer, a pager, a television including communication capabilities, a laptop computer, a server, a network computer, a personal digital assistant (PDA), a cell phone including computing capabilities, an infusion pump including a display, a glucose sensor including a display, and / or an infusion pump / glucose sensor combination. The sensor electronics device 360 ​​may also be housed in a cell phone, a smartphone, a network device, a home network device, and / or other device connected to a home network.

[0036] 4 illustrates an alternative embodiment including a sensor and a sensor electronics device. Sensor set or sensor system 400 includes sensor electronics device 360 ​​and sensor 355. The sensor includes a counter electrode 365, a reference electrode 370, and a working electrode 375. Sensor electronics device 360 ​​includes a microcontroller 410 and a digital-to-analog converter (DAC) 420. Sensor electronics device 360 ​​may also include a current-to-frequency converter (I / F converter) 430.

[0037] The microcontroller 410 includes software program code or programmable logic that, when executed, causes the microcontroller 410 to send a signal to the DAC 420, the signal representing a voltage level or value to be applied to the sensor 355. The DAC 420 receives the signal and generates a voltage value at a level commanded by the microcontroller 410. In embodiments of the present invention, the microcontroller 410 may change the representation of the voltage level in the signal frequently or infrequently. Illustratively, a signal from the microcontroller 410 may command the DAC 420 to apply a first voltage value for one second and a second voltage value for two seconds.

[0038] The sensor 355 may receive a voltage level or voltage value. In an embodiment of the present invention, the counter electrode 365 may receive the output of an operational amplifier having a reference voltage and a voltage value from the DAC 420 as inputs. Application of the voltage level causes the sensor 355 to generate a sensor signal indicative of the concentration of the physiological characteristic being measured. In an embodiment of the present invention, the microcontroller 410 may measure the sensor signal (e.g., a current value) from the working electrode. Illustratively, the sensor signal measurement circuit 431 may measure the sensor signal. In an embodiment of the present invention, the sensor signal measurement circuit 431 may include a resistor, and a current may be passed through the resistor to measure the value of the sensor signal. In an embodiment of the present invention, the sensor signal may be a current level signal, and the sensor signal measurement circuit 431 may be a current-to-frequency (I / F) converter 430. The current-to-frequency converter 430 may measure the sensor signal in terms of a current reading, convert it to a frequency-based sensor signal, and transmit the frequency-based sensor signal to the microcontroller 410. In an embodiment of the present invention, the microcontroller 410 may be able to receive the frequency-based sensor signal more easily than a non-frequency-based sensor signal. The microcontroller 410 receives the sensor signal, whether frequency-based or non-frequency-based, and determines the value of the subject's physiological characteristic, such as blood glucose level. The microcontroller 410 may include program code that, when executed or operational, can receive the sensor signal and convert the sensor signal into a physiological characteristic value. In one embodiment, the microcontroller 410 can convert the sensor signal into a blood glucose level. In some embodiments, the microcontroller 410 can utilize measurements stored in internal memory to determine the subject's blood glucose level. In some embodiments, the microcontroller 410 can utilize measurements stored in memory external to the microcontroller 410 to assist in determining the subject's blood glucose level.

[0039] After the physiological characteristic value is determined by the microcontroller 410, the microcontroller 410 can store the physiological characteristic value measurements for multiple time periods. For example, blood glucose levels may be sent from a sensor to the microcontroller 410 every second or every five seconds, and the microcontroller may store sensor measurements of BG readings for five or ten minutes. The microcontroller 410 can forward the physiological characteristic value measurements to a display on the sensor electronics device 360. For example, the sensor electronics device 360 ​​can be a monitor including a display that provides blood glucose readings to the subject. In one embodiment, the microcontroller 410 can forward the physiological characteristic value measurements to an output interface of the microcontroller 410. The output interface of the microcontroller 410 can forward the physiological characteristic value, e.g., blood glucose level measurements, to an external device, e.g., an infusion pump, an integrated infusion pump / glucose meter, a computer, a personal digital assistant, a pager, a network appliance, a server, a mobile phone, or any computing device.

[0040] FIG. 5 illustrates an electronic block diagram of a sensor electrode and the voltage applied to the sensor electrode, according to one embodiment. In the embodiment illustrated in FIG. 5, an operational amplifier 530 or other servo-controlled device can be connected to the sensor electrode 510 via a circuit / electrode interface 538. The operational amplifier 530 attempts to maintain a predetermined voltage (which the DAC may desire as the applied voltage) between the reference electrode 532 and the working electrode 534 by adjusting the voltage of the counter electrode 536 using feedback via the sensor electrode. A current can then flow from the counter electrode 536 to the working electrode 534. Such a current can be measured to confirm an electrochemical reaction between the sensor electrode 510 and a sensor biomolecule placed near the sensor electrode 510 and used as a catalytic agent. The circuit disclosed in FIG. 5 can be utilized in long-term or implantable sensors, or in short-term or subcutaneous sensors.

[0041] In long-term sensor embodiments in which a glucose oxidase (GOx) enzyme is used as the sensor's catalytic agent, current can flow from the counter electrode 536 to the working electrode 534 only if oxygen is present near the enzyme and the sensor electrode 510. Illustratively, if the voltage set at the reference electrode 532 is maintained at approximately 0.5 volts, the amount of current flowing from the counter electrode 536 to the working electrode 534 has a nearly linear relationship, with a unity slope, to the amount of oxygen present in the area surrounding the enzyme and electrode. Therefore, by maintaining the reference electrode 532 at approximately 0.5 volts and utilizing this region of the current-voltage curve to vary blood oxygen levels, improved accuracy in determining the amount of oxygen in the blood can be achieved. Different embodiments can utilize different sensors having biomolecules other than the glucose oxidase enzyme and, therefore, can have voltages other than 0.5 volts set at the reference electrode.

[0042] As discussed above, during initial implantation or insertion of the sensor 510, the sensor 510 may provide inaccurate readings due to the subject's adjustment to the sensor and even electrochemical by-products caused by the catalyst utilized in the sensor. Many sensors require a stabilization period in order for the sensor 510 to provide an accurate reading of the subject's physiological parameter. During the stabilization period, the sensor 510 does not provide accurate blood glucose measurements. Sensor users and manufacturers may desire to improve the stabilization time frame of the sensor to enable rapid use of the sensor after insertion into the subject's body or the subject's subcutaneous layer.

[0043] In previous sensor-electrode systems, the stabilization period or time frame may have been in the range of one to three hours. To shorten the stabilization period or time frame and improve the timeliness of the sensor's accuracy, the sensor (or the sensor's electrodes) may undergo multiple pulses rather than applying one pulse followed by another voltage. FIG. 6A illustrates one method of applying pulses during the stabilization time frame to shorten the stabilization time frame. In this embodiment, a voltage application device applies a first voltage to the electrodes for a first time or first period, at 600. In one embodiment, the first voltage may be a constant DC voltage. This generates an anodic current. In alternative embodiments, a digital-to-analog converter or another voltage source may supply a voltage to the electrodes for the first period. An anodic current means that electrons are driven toward the electrode to which the voltage is applied. In certain embodiments, the application device may apply a current instead of a voltage. In embodiments in which a voltage is applied to the sensor, after applying a first voltage to the electrode, the voltage regulator can wait (i.e., apply no voltage) for a second time, time frame, or period at 605. That is, the voltage application device waits until the second time has elapsed. The absence of voltage application generates a cathodic current, resulting in electron capture by the electrode to which no voltage is applied. At 610, the application of a first voltage to the electrode for a first period, followed by no voltage application for a second period, is repeated multiple times. This may be referred to as an anodic and cathodic cycle. In one embodiment, the total number of iterations of the stabilization method is three, i.e., three applications of voltage for the first period, each without application of voltage for the second period. In one embodiment, the first voltage may be 1.07 volts. In additional embodiments, the first voltage may be 0.535 volts or approximately 0.7 volts.

[0044] The repeated application and removal of voltage exposes the sensor (and therefore the electrodes) to anodic-cathode cycling. The anodic-cathode cycling results in a reduction of electrochemical by-products produced by the patient's body in response to the insertion or implantation of the sensor. The electrochemical by-products cause the generation of background current, which results in inaccurate measurements of the subject's physiological parameters. Under certain operating conditions, the electrochemical by-products may be eliminated. Under other operating conditions, the electrochemical by-products may be reduced or significantly reduced. A successful stabilization process allows the anodic-cathode cycling to reach equilibrium, significantly reducing the electrochemical by-products and minimizing the background current.

[0045] In one embodiment, the first voltage applied to the electrode of the sensor may be a positive voltage. In an alternative embodiment, the applied first voltage may be a negative voltage. Additionally, the first voltage may be applied to the working electrode. In some embodiments, the first voltage may be applied to the counter electrode or reference electrode.

[0046] In some embodiments, the durations of the voltage pulse and no voltage application may be equal, e.g., 3 minutes each. In other embodiments, the durations of the voltage application or voltage pulse may be different, e.g., the first time and the second time may be different. In one embodiment, the first time period may be 5 minutes and the wait time period may be 2 minutes. In a variation, the first time period may be 2 minutes and the wait time period (or second time frame) may be 5 minutes. That is, the duration of the first voltage application may be 2 minutes and no voltage may be applied for 5 minutes. These time frames are intended for illustrative purposes only and are not limiting. For example, the first time frame may be 2, 3, 5, or 10 minutes, and the second time frame may be 5, 10, 20 minutes, etc. The time frames (e.g., the first time and the second time) may depend on the unique characteristics of the different electrodes, the sensor, and / or the physiological characteristics of the patient.

[0047] In this regard, more or less than three pulses may be utilized to stabilize the glucose sensor, i.e., the number of repetitions may be greater or less than three. For example, four voltage pulses (e.g., a high voltage followed by no voltage) may be applied to one of the electrodes, or six voltage pulses may be applied to one of the electrodes.

[0048] Illustratively, three consecutive pulses of 1.07 volts (each followed by a wait period) may be sufficient for a subcutaneously implanted sensor. In one embodiment, three consecutive voltage pulses of 0.7 volts may be utilized. The three consecutive pulses may have higher or lower voltage values, either negative or positive, for sensors implanted in blood or cranial fluid, e.g., long-term or permanent sensors. Additionally, more than three pulses (e.g., 5, 8, 12) may be utilized to generate anodic-cathodic cycling between anodic and cathodic currents in any of the subcutaneous, blood, or cranial fluid sensors.

[0049] FIG. 6B illustrates a method of stabilizing a sensor according to embodiments of the invention herein. In the embodiment illustrated in FIG. 6B, a voltage application device can apply a first voltage to the sensor for a first time period at 630 to initiate an anodic cycle at the sensor's electrodes. The voltage application device may be a DC power supply, a digital-to-analog converter, or a voltage regulator. After the first time period has elapsed, a second voltage is applied to the sensor for a second time period at 635 to initiate a cathodic cycle at the sensor's electrodes. Illustratively, rather than no voltage being applied as illustrated in the method of FIG. 6A, a different voltage (from the first voltage) is applied to the sensor during the second time frame. In one embodiment of the invention, the application of the first voltage for a first time period and the application of the second voltage for a second time period at 640 are repeated multiple times. In certain embodiments, the application of the first voltage for a first time period and the application of the second voltage for a second time period may each be applied for a stabilization time frame, e.g., 10 minutes, 15 minutes, or 20 minutes, rather than multiple repetitions. This stabilization time frame is the entire time frame for the stabilization sequence, e.g., until the sensor (and electrodes) stabilize. Advantages of this stabilization methodology include faster sensor performance, less background current (i.e., some background current is suppressed), and improved glucose response.

[0050] In one embodiment, the first voltage may be 0.535 volts applied for 5 minutes, and the second voltage may be 1.070 volts applied for 2 minutes; the first voltage of 0.535 volts may be applied for 5 minutes, and the second voltage of 1.070 volts may be applied for 2 minutes; the first voltage of 0.535 volts may be applied for 5 minutes, and the second voltage of 1.070 volts may be applied for 2 minutes. That is, in this embodiment, there are three iterations of the voltage pulsing scheme. The pulsing methodology may be modified in that the second time frame, for example, the time frame for application of the second voltage, may be extended from 2 minutes to 5 minutes, 10 minutes, 15 minutes, or 20 minutes. Additionally, after the three iterations have been applied in this embodiment of the present invention, the nominal operating voltage of 0.535 volts may be applied.

[0051] 1.070 volts and 0.535 volts are exemplary values. Other voltage values ​​may be selected based on a variety of factors. These factors may include the type of enzyme utilized in the sensor, the membrane utilized in the sensor, the sensor's operating period, the pulse length, and / or the pulse magnitude. Under certain operating conditions, the first voltage may be in the range of 1.00 to 1.09 volts, and the second voltage may be in the range of 0.510 to 0.565 volts. In other operating embodiments, the range encompassing the first and second voltages may have a higher range, e.g., 0.3 volts, 0.6 volts, or 0.9 volts, depending on the voltage sensitivity of the sensor's electrodes. Under other operating conditions, the voltage may be in the range of 0.8 volts to 1.34 volts, and other voltages may be in the range of 0.335 to 0.735 volts. Under other operating conditions, the high voltage range may be smaller than the low voltage range. Illustratively, the high voltage may be in the range of 0.9 to 1.09 volts, and the low voltage may be in the range of 0.235 to 0.835 volts.

[0052] In one embodiment, the first voltage and the second voltage may be positive voltages, or in another embodiment, they may be negative voltages. In another embodiment, the first voltage may be positive and the second voltage may be negative, or the first voltage may be negative and the second voltage may be positive. The first voltage may have a different voltage level for each iteration. In addition, the first voltage may be a constant DC voltage. Moreover, the first voltage may be a ramp voltage, a sinusoidal voltage, a stepped voltage, or other commonly used voltage waveform. In an embodiment, the second voltage may be a constant DC voltage, a ramp voltage, a sinusoidal voltage, a stepped voltage, or other commonly used voltage waveform. In an alternative embodiment, the first voltage or the second voltage may be an AC signal superimposed on a DC waveform. In general, the first voltage may be one type of voltage, e.g., a ramp voltage, and the second voltage may be a second type of voltage, e.g., a sinusoidal voltage, and the first voltage (or the second voltage) may have a different waveform shape for each iteration. For example, if the stabilization method has three cycles, in the first cycle the first voltage may be a ramp voltage, in the second cycle the first voltage may be a constant voltage, and in the third cycle the first voltage may be a sinusoidal voltage.

[0053] In embodiments, the duration of the first time frame and the duration of the second time frame may have the same value, or the duration of the first time frame and the duration of the second time frame may have different values. For example, the duration of the first time frame may be 2 minutes, the duration of the second time frame may be 5 minutes, and the number of iterations may be 3. As described above, the stabilization method may include multiple iterations. In various embodiments, the duration of each of the first time frames may vary and the duration of each of the second time frames may vary during different iterations of the stabilization method. Illustratively, during the first iteration of the anode-cathode cycle, the first time frame may be 2 minutes and the second time frame may be 5 minutes. During the second iteration, the first time frame may be 1 minute and the second time frame may be 3 minutes. During the third iteration, the first time frame may be 3 minutes and the second time frame may be 10 minutes.

[0054] In one embodiment, a first voltage of 0.535 volts is applied to the electrodes of the sensor for 2 minutes to start an anodic cycle, and then a second voltage of 1.07 volts is applied to the electrodes for 5 minutes to start a cathodic cycle.Then, a first voltage of 0.535 volts is again applied for 2 minutes to start an anodic cycle, and a second voltage of 1.07 volts is applied to the sensor for 5 minutes.In a third iteration, 0.535 volts is applied for 2 minutes to start an anodic cycle, and then 1.07 volts is applied for 5 minutes.In this case, the voltage applied to the sensor is 0.535 during the actual operating time frame of the sensor, for example, when the sensor provides an indication of the physiological characteristic of the subject.

[0055] In the embodiments of Figures 6A and 6B, shorter duration voltage pulses can be used. Short duration voltage pulses can be used to apply the first voltage, the second voltage, or both. In one embodiment, the magnitude of the shorter duration voltage pulse of the first voltage is -1.07 volts, and the magnitude of the shorter duration voltage pulse of the second voltage is about half the higher magnitude, e.g., -0.535 volts. Alternatively, the magnitude of the shorter duration pulse of the first voltage can be 0.535 volts, and the magnitude of the shorter duration pulse of the second voltage can be 1.07 volts.

[0056] In embodiments utilizing short-duration pulses, the voltage need not be applied continuously throughout the entire first period. Instead, the voltage application device may transmit multiple short-duration pulses during the first period. That is, multiple short-duration voltage pulses may be applied to the sensor electrodes over the first period. Each short-duration pulse may have a width of several milliseconds. Illustratively, the pulse width may be 30 milliseconds, 50 milliseconds, 70 milliseconds, or 200 milliseconds. These values ​​are intended to be illustrative and not limiting. In one embodiment, such as the embodiment illustrated in FIG. 6A, these short-duration pulses are applied to the sensor (electrodes) for a first period, and then no voltage is applied for a second period.

[0057] Each short-duration pulse may have the same duration within the first period. For example, each short-duration voltage pulse may have a time width of 50 milliseconds, and the pulse delay between pulses may be 950 milliseconds. In this example, if the measurement time of the first time frame is 2 minutes, 120 short-duration voltage pulses may be applied to the sensor. Alternatively, each short-duration voltage pulse may have a different time duration. In various embodiments, each short-duration voltage pulse may have the same amplitude value or different amplitude values. By utilizing short-duration voltage pulses rather than continuous application of voltage to the sensor, the same anode and cathode cycling can occur, and the sensor (e.g., electrodes) is less susceptible to total energy or charge over time. Using short-duration voltage pulses applies less energy to the sensor (and therefore the electrodes), thereby utilizing less power compared to continuous application of voltage to the electrodes.

[0058] FIG. 6C illustrates the use of feedback in stabilizing a sensor, according to one embodiment. The sensor system may include a feedback mechanism for determining whether additional pulses are needed to stabilize the sensor. In one embodiment, a sensor signal generated by an electrode (e.g., a working electrode) may be analyzed to determine whether the sensor signal is stable. At 630, a first voltage is applied to the electrode for a first time frame to begin an anodic cycle. At 635, a second voltage is applied to the electrode for a second time frame to begin a cathodic cycle. In embodiments of the invention herein, at 637, an analysis module may analyze the sensor signal (e.g., current emitted by the sensor signal, resistance at a particular point on the sensor, impedance at a particular node on the sensor) and determine whether a threshold measurement has been reached (e.g., by comparing to the threshold measurement to determine whether the sensor is providing an accurate reading). If the sensor reading is determined to be accurate, this indicates that the electrode (and therefore the sensor) is stable, and additional application of the first voltage and / or second voltage may not occur at 642. If stability is not achieved, an additional anodic / cathodic cycle can be initiated by applying a first voltage to the electrodes for a first period at 630 and a second voltage to the electrodes for a second period at 635.

[0059] In some embodiments, the analysis module may be used after three anodic / cathodic cycles of application of the first and second voltages to the electrodes of the sensor, however, as illustrated in Figure 6C, the analysis module may be used after one application of the first and second voltages.

[0060] The analysis module may be utilized to measure the voltage emitted after a current is introduced across an electrode or across two electrodes. The analysis module may monitor the voltage level at the electrodes or at the receiving level. In one embodiment, if the voltage level exceeds a certain threshold, this may mean that the sensor is stable. In one embodiment, if the voltage level is below a threshold level, this may indicate that the sensor is stable and ready to provide an indication. In one embodiment, a current can be introduced into an electrode or across several electrodes. The analysis module may monitor the current level emitted from the electrodes. In this embodiment, the analysis module may be capable of monitoring the current when the current differs by an order of magnitude from the sensor signal current. If the current is above or below a current threshold, this may indicate that the sensor is stable.

[0061] In embodiments of the invention herein, the analysis module may measure the impedance between two electrodes of the sensor. The analysis module may compare the impedance to a threshold impedance or target impedance value, and if the measured impedance is lower than the target impedance or threshold impedance, the sensor (and therefore the sensor signal) may be stable. In one embodiment, the analysis module may measure the resistance between two electrodes of the sensor. In this embodiment of the invention, the analysis module may compare the resistance to a threshold resistance value or target resistance value, and if the measured resistance value is lower than the threshold resistance value or target resistance value, the analysis module may determine that the sensor is stable and that the sensor signal is usable.

[0062] FIG. 7 illustrates the effect of stabilizing a sensor according to an embodiment of the present invention. Line 705 represents a blood glucose sensor reading of a glucose sensor utilizing a previous single-pulse stabilization method. Line 710 represents a blood glucose reading of a glucose sensor to which three voltage pulses are applied (e.g., three voltage pulses each having a duration of two minutes followed by five minutes of no voltage). The x-axis 715 represents the amount of time. Dots 720, 725, 730, and 735 represent measured glucose readings obtained using a fingerstick and then entered into a glucose meter. As illustrated by the graph, the previous single-pulse stabilization method took approximately 1 hour and 30 minutes to stabilize to a desired glucose reading, e.g., 100 units. In contrast, with the three-pulse stabilization method, the glucose sensor stabilized in only about 15 minutes, resulting in a dramatic improvement in the stabilization time frame.

[0063] FIG. 8A illustrates a block diagram of a sensor electronics device and a sensor including a voltage generating device. The voltage generating or applying device 810 includes electronics, logic, or circuitry that generates voltage pulses. The sensor electronics device 360 ​​also includes an input device 820 for receiving reference values ​​and other useful data. In one embodiment, the sensor electronics device includes a measurement memory 830 for storing sensor measurements. In this embodiment, a power supply 380 can provide power to the sensor electronics device. The power supply 380 can provide power to a regulator 385, which provides a regulated voltage to the voltage generating or applying device 810. A connection terminal 811 represents a connection terminal coupling or connecting the sensor 355 to the sensor electronics device 360 ​​in the illustrated embodiment.

[0064] 8A , a voltage generating or applying device 810 supplies a voltage, e.g., a first voltage or a second voltage, to an input terminal of an operational amplifier 840. The voltage generating or applying device 810 may also supply a voltage to the working electrode 375 of the sensor 355. Another input terminal of the operational amplifier 840 is coupled to the reference electrode 370 of the sensor. The application of a voltage from the voltage generating or applying device 810 to the operational amplifier 840 drives the voltage measured at the counter electrode 365 to be close to or equal to the voltage applied at the working electrode 375. In an embodiment, the voltage generating or applying device 810 may be utilized to apply a desired voltage between the counter electrode and the working electrode. This may be done by applying a fixed voltage directly to the counter electrode.

[0065] In one embodiment illustrated in FIGS. 6A and 6B, the voltage generating device 810 generates a first voltage to be applied to the sensor during a first time frame. The voltage generating device 810 transmits this first voltage to the operational amplifier 840, which drives the voltage at the counter electrode 365 of the sensor 355 to the first voltage. In some embodiments, the voltage generating device 810 may also transmit the first voltage directly to the counter electrode 365 of the sensor 355. In the embodiment illustrated in FIG. 6A, the voltage generating device 810 does not transmit the first voltage to the sensor 355 during a second time frame. That is, the voltage generating device 810 is turned off or switched off. The voltage generating device 810 can be programmed to continue cycling between applying the first voltage and not applying a voltage multiple times or for a stabilization time frame, e.g., 20 minutes. FIG. 8B illustrates a voltage generating device for implementing this embodiment of the present invention. Voltage regulator 385 transfers the regulated voltage to voltage generating device 810. Control circuit 860 controls the opening and closing of switch 850. When switch 850 is closed, voltage is applied. When switch 850 is open, no voltage is applied. Timer 865 provides signals to control circuit 860 that instruct control circuit 860 to turn switch 850 on and off. Control circuit 860 includes logic that can instruct the circuit to open and close switch 850 multiple times (to match the required repetition). In one embodiment, timer 865 may also send a stabilization signal to identify that the stabilization sequence is complete, i.e., the stabilization time frame has elapsed.

[0066] In one embodiment, the voltage generating device generates a first voltage for a first time frame and a second voltage for a second time frame. FIG. 8C illustrates a voltage generating device that generates two voltage values ​​to implement this embodiment. In this embodiment, a two-position switch 870 is utilized. Illustratively, when the first switch position 871 is turned on or closed by the timer 865 commanding the control circuit 860, the voltage generating device 810 generates the first voltage for the first time frame. After the first voltage is applied for the first time frame, the timer sends a signal to the control circuit 860 indicating that the first time frame has elapsed, and the control circuit 860 instructs the switch 870 to move to the second position 872. When the switch 870 is in the second position 872, the regulated voltage is directed to a voltage step-down or buck converter 880 to reduce the regulated voltage to a smaller value. The smaller value is then sent to the operational amplifier 840 for the second time. After the timer 865 signals the control circuit 860 that the second time frame has elapsed, the control circuit 860 returns the switch 870 to the first position. This continues until the desired number of iterations is completed or the stabilization time frame has elapsed. In embodiments of the invention herein, after the sensor stabilization time frame has elapsed, the sensor transmits the sensor signal 350 to the signal processor 390.

[0067] FIG. 8D illustrates a voltage application device 810 utilized to implement more complex application of voltage to the sensor. The voltage application device 810 can include a control device 860, a switch 890, a sine wave voltage generating device 891, a ramp voltage generating device 892, and a constant voltage generating device 893. In other embodiments, the voltage application may generate an AC wave on top of a DC signal or various other voltage pulse waveforms. In the embodiment illustrated in FIG. 8D, the control device 860 can move a switch to one of three voltage generation systems: 891 (sine wave), 892 (ramp), or 893 (constant DC). This results in each voltage generation system generating a distinct voltage waveform. Under certain operating conditions, for example, when a sine wave pulse is applied to three pulses, the control device 860 can cause the switch 890 to connect the voltage from the voltage regulator 385 to the sine wave voltage generator 891, causing the voltage application device 810 to generate a sine wave voltage. In other operating conditions, for example, where a ramp voltage is applied to the sensor as the first voltage of a first pulse of three pulses, a sinusoidal voltage is applied to the sensor as the first voltage of a second pulse of three pulses, and a constant DC voltage is applied to the sensor as the first voltage of a third pulse of three pulses, control device 860 may cause switch 890 to make switching movements during a first time frame of an anode / cathode cycle to connect the voltage from voltage generating or applying device 810 to ramp voltage generating system 892, then to sinusoidal voltage generating system 891, and then to constant DC voltage generating system 893. In this embodiment, control device 860 may also direct or control the switches to connect certain of the voltage generation subsystems to the voltage from regulator 385 during a second time frame, e.g., during the application of a second voltage.

[0068] FIG. 9A illustrates a sensor electronics device including a microcontroller for generating voltage pulses. An advanced sensor electronics device can include a microcontroller 410 (see FIG. 4), a digital-to-analog converter (DAC) 420, an operational amplifier 840, and a sensor signal measurement circuit 431. In one embodiment, the sensor signal measurement circuit can be a current-to-frequency (I / F) converter 430. In the embodiment illustrated in FIG. 9A, software or programmable logic within the microcontroller 410 provides instructions to send a signal to the DAC 420, which in turn instructs the DAC 420 to output a particular voltage to the operational amplifier 840. As illustrated by line 911 in FIG. 9A, the microcontroller 410 can also be instructed to output a particular voltage to the working electrode 375. As described above, application of a particular voltage to the operational amplifier 840 and the working electrode 375 can drive the voltage measured at the counter electrode to a particular voltage magnitude. That is, the microcontroller 410 outputs a signal indicating a voltage or voltage waveform to be applied to the sensor 355 (e.g., an operational amplifier 840 coupled to the sensor 355). In an alternative embodiment, a fixed voltage may be set by applying a voltage directly from the DAC 420 between the reference electrode and the working electrode 375. Similar results may also be achieved by applying a voltage to each of the electrodes with a difference equal to the fixed voltage applied between the reference electrode and the working electrode. Additionally, a fixed voltage may be set by applying a voltage between the reference electrode and the counter electrode. Under certain operating conditions, the microcontroller 410 may generate a pulse of a particular magnitude that the DAC 420 understands to represent that a voltage of a particular magnitude should be applied to the sensor. After a first time frame, the microcontroller 410 (via program or programmable logic) outputs a second signal to the DAC 420 instructing it to either not output a voltage (in the case of a sensor electronics device 360 ​​operating according to the method described in FIG. 6A) or to output a second voltage (in the case of a sensor electronics device 360 ​​operating according to the method described in FIG. 6B).After the second time frame has elapsed, the microcontroller 410 repeats the cycle of sending a signal indicating that the first voltage is to be applied (during the first time frame) and sending a signal instructing that either no voltage is to be applied or that the second voltage is to be applied (during the second time frame).

[0069] Under other operating conditions, the microcontroller 410 can generate signals to the DAC 420 instructing the DAC to output a ramp voltage. Under other operating conditions, the microcontroller 410 may generate signals to the DAC 420 instructing the DAC 420 to output a voltage that simulates a sinusoidal voltage. These signals may be incorporated into any of the pulsing methodologies described in the previous paragraph of this application or earlier. In one embodiment, the microcontroller 410 may generate a sequence of instructions and / or pulses that the DAC 420 receives and understands to mean that a particular pulse sequence should be applied. For example, the microcontroller 410 may send a sequence of instructions (via signals and / or pulses) to the DAC 420 instructing it to generate a constant voltage for the first iteration of the first time frame, a ramp voltage for the first iteration of the second time frame, a sinusoidal voltage for the second iteration of the first time frame, and a square wave having two values ​​for the second iteration of the second time frame.

[0070] The microcontroller 410 may include programmable logic or a program to continue this cycling for a stabilization time frame or multiple iterations. Illustratively, the microcontroller 410 may include counting logic to identify when the first time frame or the second time frame has elapsed. Additionally, the microcontroller 410 may include counting logic to identify when the stabilization time frame has elapsed. After any of the preceding time frames has elapsed, the counting logic may instruct the microcontroller to either send a new signal or stop sending a signal to the DAC 420.

[0071] The microcontroller 410 allows for the application of various voltage magnitudes in multiple sequences over multiple durations. In an embodiment of the present invention, the microcontroller 410 may include control logic or a program that instructs the digital-to-analog converter 420 to transmit a voltage pulse having a magnitude of approximately 1.0 volt for a first period of 1 minute, then transmit a voltage pulse of approximately 0.5 volts for a second period of 4 minutes, and repeat this cycle four times. In one embodiment, the microcontroller 420 may be programmed to transmit a signal that causes the DAC 420 to apply a voltage pulse of the same magnitude for each first voltage in each iteration. The microcontroller 410 may also be programmed to transmit a signal that causes the DAC 420 to apply a voltage pulse of a different magnitude for each first voltage in each iteration. In this embodiment, the microcontroller 410 may also be programmed to transmit a signal that causes the DAC 420 to apply a voltage pulse of a different magnitude for each second voltage in each iteration. Illustratively, the microcontroller 410 may be programmed to send signals that cause the DAC 420 to apply a first voltage pulse of approximately 1.0 volts in a first iteration, apply a second voltage pulse of approximately 0.5 volts in a first iteration, apply a first voltage of 0.7 volts and a second voltage of 0.4 volts in a second iteration, and apply a first voltage of 1.2 volts and a second voltage of 0.8 volts in a third iteration.

[0072] The microcontroller 410 may also be programmed to instruct the DAC 420 to provide multiple short duration voltage pulses during a first time frame. In this embodiment of the invention, rather than applying one voltage for the entire first time frame (e.g., 2 minutes), multiple shorter duration pulses may be applied to the sensor. In this embodiment, the microcontroller 410 may also be programmed to instruct the DAC 420 to provide multiple short duration voltage pulses to the sensor during a second time frame. Illustratively, the microcontroller 410 may send a signal that causes the DAC to apply multiple short duration voltage pulses, the short duration being 50 milliseconds or 100 milliseconds. Between these short duration pulses, the DAC may apply no voltage, or the DAC may apply a minimum voltage. The microcontroller may cause the DAC 420 to apply short duration voltage pulses for the first time frame, e.g., 2 minutes. The microcontroller 410 may then send a signal to the DAC 420 causing it to apply no voltage or short duration voltage pulses at a second voltage magnitude to the sensor for a second time frame; for example, the second voltage may be 0.75 volts and the second time frame may be 5 minutes. In one embodiment, the microcontroller 410 may send a signal to the DAC 420 causing it to apply a different voltage magnitude for each of the short duration pulses in the first time frame and / or the second time frame. In an embodiment, the microcontroller 410 may send a signal to the DAC 420 causing it to apply a pattern of voltage magnitudes to the short duration voltage pulses during the first time frame or the second time frame. For example, the microcontroller may send a signal or pulse instructing the DAC 420 to apply 30 20 millisecond pulses to the sensor during the first time frame. Each of the 30 20 millisecond pulses may have the same magnitude or different magnitudes.In this embodiment, the microcontroller 410 may instruct the DAC 420 to apply a short duration pulse during the second time frame, or may instruct the DAC 420 to apply a different voltage waveform during the second time frame.

[0073] While FIGS. 6-8 disclose the application of a voltage, a current may be applied to the sensor to initiate the stabilization process. Exemplarily, in the embodiment shown in FIG. 6B, a first current may be applied during a first time frame to initiate an anodic or cathodic response, and a second current may be applied during a second time frame to initiate an opposite anodic or cathodic response. The application of the first and second currents may continue for multiple iterations or may continue for a stabilization time frame. In one embodiment, a first current may be applied during a first time frame, and a first voltage may be applied during a second time frame. That is, one of the anodic or cathodic cycles may be triggered by applying a current to the sensor, and the other of the anodic or cathodic cycles may be triggered by applying a voltage to the sensor. As discussed above, the applied current may be a constant current, a ramp current, a stepped pulse current, or a sinusoidal current. Under certain operating conditions, the current may be applied as a sequence of short duration pulses during a first time frame.

[0074] FIG. 9B illustrates a sensor and sensor electronics utilizing an analysis module for feedback during a stabilization period, according to embodiments of the invention herein. FIG. 9B introduces an analysis module 950 in a sensor electronics device 360. The analysis module 950 utilizes feedback from the sensor to determine whether the sensor is stable. In one embodiment, the microcontroller 410 may include instructions or commands to control the DAC 420 such that the DAC 420 applies a voltage or current to a portion of the sensor 355. FIG. 9B illustrates that a voltage or current may be applied between the reference electrode 370 and the working electrode 375. However, the voltage or current may be applied between the electrodes or directly to one of the electrodes, and the present invention should not be limited by the embodiment illustrated in FIG. 9B. The application of the voltage or current is illustrated by the dotted line 955. The analysis module 950 may measure the voltage, current, resistance, or impedance of the sensor 355. While FIG. 9B illustrates measurements being taken at the working electrode 375, this is not a limitation of the present invention, as other embodiments may measure voltage, current, resistance, or impedance directly between the electrodes of the sensor or at either the reference electrode 370 or counter electrode 365. The analysis module 950 may receive the measured voltage, current, resistance, or impedance and compare the measurement to a stored value (e.g., a threshold value). The dotted line 956 represents the analysis module 950 reading or making a voltage, current, resistance, or impedance measurement. Under certain operating conditions, if the measured voltage, current, resistance, or impedance is above the threshold value, the sensor is stable and the sensor signal provides an accurate indication of the patient's physiological state. Under other operating conditions, if the measured voltage, current, resistance, or impedance is below the threshold value, the sensor is stable. Under other operating conditions, the analysis module 950 may verify that the measured voltage, current, resistance, or impedance is stable for a certain time frame, for example, one or two minutes.This may represent that the sensor 355 is stable and that the sensor signal is providing an accurate measurement of the subject's physiological parameter, e.g., blood glucose level. After the analysis module 950 determines that the sensor is stable and the sensor signal is providing an accurate measurement, the analysis module 950 may send a signal to the microcontroller 410 indicating that the sensor is stable and that the microcontroller 410 can begin using or receiving the sensor signal from the sensor 355 (e.g., a sensor stabilization signal). This is represented by the dotted line 957.

[0075] 10 illustrates a block diagram of a sensor system including hydration electronics. The sensor system includes a connector 1010, a sensor 1012, and a monitor or sensor electronics device 1025. The sensor 1012 includes electrodes 1020 and a connecting portion 1024. In one embodiment, the sensor 1012 may be connected to the sensor electronics device 1025 via the connector 1010 and a cable. In other embodiments, the sensor 1012 may be connected directly to the sensor electronics device 1025. In some embodiments, the sensor 1012 may be incorporated into the same physical device as the sensor electronics device 1025. The monitor or sensor electronics device 1025 may include a power supply 1030, a regulator 1035, a signal processor 1040, a measurement processor 1045, and a processor 1050. The monitor or sensor electronics device 1025 may also include a hydration detection circuit 1060. The hydration detection circuit 1060 interfaces with the sensor 1012 to determine whether the electrodes 1020 of the sensor 1012 are sufficiently hydrated. It is important to know when the electrodes 1020 are sufficiently hydrated because if the electrodes 1020 are not sufficiently hydrated, the electrodes 1020 will not provide accurate glucose readings. When the electrodes 1020 are sufficiently hydrated, accurate glucose readings can be obtained.

[0076] In the embodiment illustrated in FIG. 10 , the hydration detection circuit 1060 may include a delay or timer module 1065 and a connection detection module 1070. In embodiments utilizing a short-term or subcutaneous sensor, the sensor 1012 is inserted into the subcutaneous tissue, and then the sensor electronics device or monitor 1025 is connected to the sensor 1012. The connection detection module 1070 identifies that the sensor electronics device 1025 has connected to the sensor 1012 and sends a signal to the timer module 1065. This is illustrated in FIG. 10 by arrow 1084, which represents the detector 1083 detecting the connection and sending a signal to the connection detection module 1070 indicating that the sensor 1012 has connected to the sensor electronics device 1025. In embodiments utilizing an implantable or long-term sensor, the connection detection module 1070 identifies that the implantable sensor has been inserted into the body. The timer module 1065 receives the connection signal and waits for a set or established hydration time. Illustratively, the hydration time may be 2 minutes, 5 minutes, 10 minutes, or 20 minutes. These examples are intended to be illustrative and not limiting. The time frame need not be a set number of minutes, but can include any number of seconds. In one embodiment, after the timer module 1065 has waited the set hydration time, the timer module 1065 may notify the processor 1050 that the sensor 1012 is hydrated by sending a hydration signal, illustrated by line 1086.

[0077] In this embodiment, the processor 1050 receives the hydration signal and can begin utilizing the sensor signal (e.g., sensor measurement) only after the hydration signal is received. In another embodiment, the hydration detection circuit 1060 may be coupled between the sensor (sensor electrodes 1020) and the signal processor 1040. In this embodiment, the hydration detection circuit 1060 can prevent the sensor signal from being sent to the signal processor 1040 until the timer module 1065 notifies the hydration detection circuit 1060 that a set hydration time has elapsed. This is illustrated by the dotted lines labeled with reference numbers 1080 and 1081. Illustratively, the timer module 1065 can send a connect signal to a switch (or transistor) to turn the switch on and forward the sensor signal to the signal processor 1040. In an alternative embodiment, the timer module 1065 sends a connection signal to turn on a switch 1088 (or close the switch 1088) in the hydration detection circuit 1060 so that the voltage from the regulator 1035 can be applied to the sensor 1012 after the hydration time has elapsed. That is, in this embodiment, the voltage from the regulator 1035 is not applied to the sensor 1012 until the hydration time has elapsed.

[0078] FIG. 11 illustrates an embodiment including a mechanical switch to assist in determining hydration time. In one embodiment, a single housing may contain the sensor assembly 1120 and the sensor electronics device 1125. In another embodiment, the sensor assembly 1120 may be in one housing and the sensor electronics device 1125 may be in a separate housing, but the sensor assembly 1120 and the sensor electronics device 1125 may be connected to each other. In this embodiment, the connection detection mechanism 1160 may be a mechanical switch. The mechanical switch may detect that the sensor 1120 is physically connected to the sensor electronics device 1125. The timer circuit 1135 may also be activated when the mechanical switch 1160 detects that the sensor 1120 is connected to the sensor electronics device 1125. That is, the mechanical switch may close and a signal may be transmitted to the timer circuit 1135. When the hydration time has elapsed, the timer circuit 1135 sends a signal to the switch 1140 to enable the regulator 1035 to apply voltage to the sensor 1120. That is, no voltage is applied until the hydration time has elapsed. In one embodiment, once the hydration time has elapsed, a current may replace the voltage as being applied to the sensor. In an alternative embodiment, power may be first applied to the sensor 1120 when the mechanical switch 1160 identifies that the sensor 1120 is physically connected to the sensor electronics device 1125. The power sent to the sensor 1120 outputs a sensor signal from the working electrode of the sensor 1120. The sensor signal is measured and sent to the processor 1175. The processor 1175 may include a counter input. Under certain operating conditions, after a set hydration time has elapsed from the time the sensor signal is input to the processor 1175, the processor 1175 may begin processing the sensor signal as an accurate measurement of glucose in the subject's body. That is, the processor 1170 has received a sensor signal from the potentiostat circuit 1170 for a certain period of time, but does not process the signal until it receives a command from the processor's counter input identifying that the hydration time has elapsed. In one embodiment, the potentiostat circuit 1170 may include a current-to-frequency converter 1180 .In this embodiment, the current to frequency converter 1180 may receive the sensor signal as a current value and convert the current value to a frequency value, which is easier for the processor 1175 to process.

[0079] The mechanical switch 1160 may also notify the processor 1175 when the sensor 1120 is disconnected from the sensor electronics device 1125. This is represented in FIG. 11 by dotted line 1176. This may cause the processor 1170 to shut off or reduce power to multiple components, chips, and / or circuits of the sensor electronics device 1125. If the sensor 1120 is not connected, components or circuits of the sensor electronics device 1125 may drain their batteries or power sources if they are powered on. Thus, if the mechanical switch 1160 detects that the sensor 1120 has been disconnected from the sensor electronics device 1125, the mechanical switch may indicate this to the processor 1175, which may then shut off or reduce power to one or more of the circuits, chips, or components of the sensor electronics device 1125.

[0080] FIG. 12 illustrates a method for electrical detection of hydration according to embodiments of the invention herein. In one embodiment, an electrical detection mechanism can be utilized to detect sensor connection. In this embodiment, hydration detection electronics 1250 can include an AC power source 1255 and detection circuitry 1260. The hydration detection electronics 1250 can be located within a sensor electronics device 1225. The sensor 1220 can include a counter electrode 1221, a reference electrode 1222, and a working electrode 1223. As illustrated in FIG. 12, the AC power source 1255 is coupled to a voltage setting device 1275, the reference electrode 1222, and the detection circuitry 1260. In this embodiment, an AC signal from the AC power source is applied to the reference electrode connection, as illustrated by dotted line 1291 in FIG. 12. The AC signal can be coupled to the sensor 1220 through an impedance, and the coupled signal is significantly attenuated when the sensor 1220 is connected to the sensor electronics device 1225. Therefore, a low-level AC signal is present at the input to the detection circuitry 1260. This may also be referred to as a highly attenuated signal, or a signal with a high level of attenuation. Under certain operating conditions, the voltage level of the AC signal may be Vapplied*(Ccoupling) / (Ccoupling+Csensor). If the detection circuit 1260 detects that a high level AC signal (low attenuation signal) is present at the input terminal of the detection circuit 1260, no interrupt is sent to the microcontroller 410 because the sensor 1220 is not sufficiently hydrated or activated. For example, the input of the detection circuit 1260 may be a comparator. When the sensor 1220 is sufficiently hydrated (or wet), an effective capacitance is formed between the counter electrode and the reference electrode (e.g., capacitance C in FIG. 12). r-c ), an effective capacitance is formed between the reference electrode and the working electrode (e.g., capacitance C in Figure 12). w-r ), that is, effective capacitance relates to the capacitance formed between two nodes and does not represent an actual capacitor placed in circuit between the two electrodes. In one embodiment, the AC signal from AC power source 1255 is applied to the capacitance C r-c and C w-r11 , the detection circuit 1260 detects the presence of a low-level or highly attenuated AC signal from the AC power source 1255 at the input terminals of the detection circuit 1260. This embodiment is important because utilizing existing connections between the sensor 1120 and the sensor electronics device 1125 reduces the number of connections to the sensor. That is, the mechanical switch disclosed in FIG. 11 requires a switch and associated connections between the sensor 1120 and the sensor electronics device 1125. Eliminating the mechanical switch is advantageous because the sensor 1120 is continually shrinking in size, and component removal helps achieve this size reduction. In alternative embodiments, the AC signal may be applied to a different electrode (e.g., the counter electrode or working electrode), and the invention may operate in a similar manner.

[0081] As described above, after the detection circuit 1260 detects that a low-level AC signal is present at its input terminal, the detection circuit 1260 may later detect that a high-level AC signal with low attenuation is present at the input terminal. This indicates that the sensor 1220 is disconnected from the sensor electronics device 1225 or that the sensor is not operating properly. If the sensor is disconnected from the sensor electronics device 1225, the AC power source may be coupled to the input of the detection circuit 1260 with little or low attenuation. As described above, the detection circuit 1260 may generate an interrupt to the microcontroller. This interrupt may be received by the microcontroller, which may then reduce or remove power to one or more components or circuits within the sensor electronics device 1225. This may be referred to as a second interrupt. Again, this helps reduce power consumption of the sensor electronics device 1225, especially when the sensor 1220 is not connected to the sensor electronics device 1225.

[0082] In an alternative embodiment, an AC signal may be applied to the reference electrode 1222, as illustrated by reference numeral 1291, and the impedance measuring device 1277 may measure the impedance of a region within the sensor 1220. Illustratively, this region may be the region between the reference electrode and the working electrode, as illustrated by dotted line 1292 in FIG. 12 . Under certain operating conditions, the impedance measuring device 1277 may send a signal to the detection circuit 1260 if the measured impedance decreases below an impedance threshold or other set criterion, indicating that the sensor is sufficiently hydrated. Under other operating conditions, the impedance measuring device 1277 may send a signal to the detection circuit 1260 when the impedance increases above the impedance threshold. The detection circuit 1260 then sends an interrupt to the microcontroller 410. In another embodiment, the impedance measuring device 1277 may send an interrupt or signal directly to the microcontroller.

[0083] In alternative embodiments, AC power supply 1255 may be replaced by a DC power supply. If a DC source is utilized, a resistance measuring element may be utilized in place of impedance measuring element 1277. In embodiments utilizing a resistance measuring element, when the resistance drops below a resistance threshold or set criterion, the resistance measuring element may send a signal directly to detection circuit 1260 (represented by dotted line 1293) or a microcontroller indicating that the sensor is sufficiently hydrated and that power may be applied to the sensor.

[0084] In the embodiment illustrated in FIG. 12 , if the detection circuit 1260 detects a low level or highly attenuated AC signal from the AC power source, an interrupt is generated to the microcontroller 410. This interrupt indicates that the sensor is sufficiently hydrated. In this embodiment, in response to the interrupt, the microcontroller 410 generates a signal that is forwarded to the digital-to-analog converter 420, causing the digital-to-analog converter 420 to apply a voltage or current to the sensor 1220. Any of the different sequences of pulses or short duration pulses described above in FIG. 6A , FIG. 6B , or FIG. 6C , or the associated text describing the application of pulses, may be applied to the sensor 1220. Illustratively, the voltage from the DAC 420 may be applied to an operational amplifier 1275, the output of which is applied to the counter electrode 1221 of the sensor 1220. This results in a sensor signal being generated by the sensor, e.g., the working electrode 1223 of the sensor. Because the sensor is sufficiently hydrated, the sensor signal produced by the working electrode 1223 is accurately measuring glucose, as identified by the interrupt. The sensor signal is measured by the sensor signal measuring device 431, which transmits the sensor signal to the microcontroller 410, where a parameter of the subject's physiological state is measured. The generation of an interrupt indicates that the sensor is sufficiently hydrated and that the sensor 1220 is now providing an accurate glucose measurement. In this embodiment, the hydration period may depend on the type and / or manufacturer of the sensor and the sensor's response to insertion or implantation into the subject. Illustratively, one sensor 1220 may have a hydration time of 5 minutes, and one sensor 1220 may have a hydration time of 1 minute, 2 minutes, 3 minutes, 6 minutes, or 20 minutes. Again, any amount of time may be an acceptable amount of hydration time for the sensor, although shorter amounts of time are preferred.

[0085] If the sensor 1220 is connected but not sufficiently hydrated or wetted, the effective capacitance C r-c and C w-rThe AC power source 1255 may not attenuate the AC signal from the AC power supply 1255. The electrodes of the sensor 1120 are dry before insertion, and because the electrodes are dry, there is no good electrical (or conductive) path between the two electrodes. Therefore, high-level or weakly attenuated AC signals can still be detected by the detection circuit 1260 and no interrupts can be generated. When the sensor is inserted, the electrodes are immersed in a conductive body fluid. This creates a leakage path with lower DC resistance. Also, a boundary layer capacitor forms at the metal / fluid interface. That is, a significant capacitance forms between the metal / fluid interface, and this large capacitance appears as two capacitors in series between the sensor electrodes. This is sometimes referred to as effective capacitance. In effect, the conductivity of the electrolyte on the electrodes is being measured. In some embodiments of the present invention, the glucose limiting membrane (GLM) also exhibits an impedance that blocks electrical efficiency. An unhydrated GLM exhibits high impedance, while a hydrated GLM exhibits low impedance. Low impedance is desirable for accurate sensor measurements.

[0086] FIG. 13A illustrates a method of hydrating a sensor according to embodiments of the invention herein. In one embodiment, at 1310, the sensor may be physically connected to the sensor electronics device. After connection, in one embodiment, a timer or counter may be started at 1320 to count the hydration time. After the hydration time has elapsed, at 1330, a signal may be sent to a subsystem of the sensor electronics device to begin applying voltage to the sensor. As mentioned above, in one embodiment, a microcontroller may receive this signal and instruct a DAC to apply voltage to the sensor, or in another embodiment of the invention, a switch may receive a signal that enables a regulator to apply voltage to the sensor. The hydration time may be 5 minutes, 2 minutes, 10 minutes, and may vary depending on the subject and even the type of sensor.

[0087] In an alternative embodiment, after connecting the sensor to the sensor electronics device, an AC signal (e.g., a low-voltage AC signal) may be applied to the sensor, e.g., to the sensor's reference electrode, at 1340. The AC signal may be applied because the connection of the sensor to the sensor electronics device allows the AC signal to be applied to the sensor. After the AC signal is applied, an effective capacitance forms between the electrode of the sensor to which the voltage is applied and the other two electrodes, at 1350. The detection circuit determines, at 1360, what level of AC signal is present at the detection circuit's input. If a low-level AC signal (or a highly attenuated AC signal) is present at the detection circuit's input, an interrupt is generated by the detection circuit and sent to the microcontroller, at 1370, due to the effective capacitance forming a good electrical conduit between the electrodes and the resulting attenuation of the AC signal.

[0088] The microcontroller receives the interrupt generated by the detection circuit and sends a signal to the digital-to-analog converter at 1380 instructing or causing the digital-to-analog converter to apply a voltage to an electrode of the sensor, e.g., a counter electrode. At 1390, upon applying a voltage to the electrode of the sensor, the sensor creates or generates a sensor signal. The sensor signal measuring device 431 measures the generated sensor signal and sends the sensor signal to the microcontroller. At 1395, the microcontroller receives the sensor signal from the sensor signal measuring device coupled to the working electrode and processes the sensor signal to extract a measure of a physiological characteristic of the subject or patient.

[0089] FIG. 13B illustrates an additional method for verifying sensor hydration according to embodiments of the invention herein. In the embodiment illustrated in FIG. 13B, a sensor is physically connected to a sensor electronics device at 1310. An AC signal is applied to an electrode in the sensor, e.g., a reference electrode, at 1341. Alternatively, in another embodiment, a DC signal is applied to the electrode of the sensor at 1341. If the AC signal is applied, an impedance measuring element measures impedance at a point in the sensor at 1351. Alternatively, if a DC signal is applied, a resistance measuring element measures resistance at a point in the sensor at 1351. If the resistance or impedance is lower than a resistance or impedance threshold (or other set criterion), respectively, at 1361, the impedance (or resistance) measuring element signals (or enables a signal to be sent) a detection circuit, which sends an interrupt to the microcontroller identifying that the sensor is hydrated. Reference numbers 1380, 1390, and 1395 are the same in FIGS. 13A and 13B because they represent the same actions.

[0090] The microcontroller receives the interrupt and sends a signal to the digital-to-analog converter at 1380 to apply a voltage to the sensor. In an alternative embodiment, the digital-to-analog converter can apply a current to the sensor, as described above. At 1390, the sensor, e.g., the working electrode, creates a sensor signal representative of a physiological parameter of the patient. At 1395, the microcontroller receives a sensor signal from a sensor signal measuring device that measures the sensor signal at an electrode in the sensor, e.g., the working electrode. The microcontroller processes the sensor signal to extract a measurement of a physiological characteristic of the subject or patient, e.g., the patient's blood glucose level.

[0091] 14A and 14B illustrate a method for combining sensor hydration and sensor stabilization according to embodiments of the invention herein. In the embodiment of the invention illustrated in FIG. 14A, a sensor is connected to a sensor electronics device at 1405. An AC signal is applied to the sensor's electrodes at 1410. A detection circuit determines at 1420 what level of AC signal is present at the detection circuit's input. If the detection circuit determines that a low level of AC signal is present at the input (representing attenuation of a high level of AC signal), an interrupt is sent to the microcontroller at 1430. Once the interrupt is sent to the microcontroller, the microcontroller knows to initiate or begin a stabilization sequence at 1440, i.e., the application of multiple voltage pulses to the sensor's electrodes, as described above. For example, the microcontroller may cause the digital-to-analog converter to apply three voltage pulses (magnitude +0.535 volts) to the sensor, each of the three voltage pulses followed by a period of three voltage pulses (applied with a magnitude of 1.07 volts). This may be referred to as sending a voltage stabilization sequence. The microcontroller may cause this by executing a software program in read-only memory (ROM) or random access memory. After the stabilization sequence has finished executing, at 1450, the sensor may generate a sensor signal that is measured and sent to the microcontroller.

[0092] The detection circuit may determine at 1432 that a high-level AC signal continues to be present at the detection circuit's input (e.g., the comparator's input) after the hydration time threshold has elapsed. For example, the hydration time threshold may be 10 minutes. After 10 minutes have elapsed, the detection circuit may still detect that a high-level AC signal is present. At this point, the detection circuit may send a hydration assistance signal to the microcontroller at 1434. If the microcontroller receives the hydration assistance signal, the microcontroller may send a signal at 1436 that causes the DAC to apply a voltage pulse or series of voltage pulses to assist in hydration of the sensor. In one embodiment, the microcontroller may send a signal that causes the DAC to apply part of a stabilization sequence or other voltage pulses to assist in hydration of the sensor. In this embodiment, the application of the voltage pulse results in a low-level AC signal (or a highly attenuated signal) being detected by the detection circuit at 1438. At this point, the detection circuit may send an interrupt and the microcontroller may begin the stabilization sequence, as disclosed in step 1430.

[0093] FIG. 14B illustrates a second embodiment of a combined hydration and stabilization method in which feedback is utilized in the stabilization process. At 1405, the sensor is connected to a sensor electronics device. At 1411, an AC signal (or DC signal) is applied to the sensor. In an embodiment, the AC signal (or DC signal) is applied to an electrode of the sensor, e.g., a reference electrode. At 1416, an impedance measuring device (or resistance measuring device) measures the impedance (or resistance) within a designated region of the sensor, e.g., between the reference electrode and the working electrode. At 1421, the measured impedance (or resistance) is compared with an impedance or resistance value to determine whether the impedance (or resistance) within the sensor is low enough to indicate that the sensor is hydrated. If the impedance (or resistance) is below the impedance (or resistance) value or other set criterion (which may be a threshold), an interrupt is sent to the microcontroller at 1431. After receiving the interrupt, the microcontroller sends a signal to the DAC instructing the DAC to apply a stabilization sequence of voltage (or current) to the sensor at 1440. After the stabilization sequence is applied to the sensor, a sensor signal is generated at the sensor (e.g., at the working electrode), measured by the sensor signal measuring device, transmitted by the sensor signal measuring device, and received by the microcontroller at 1450. Because the sensor is hydrated and a stabilization sequence of voltages has been applied to the sensor, the sensor signal is an accurate measurement of the physiological parameter (i.e., blood glucose).

[0094] FIG. 14C illustrates a third embodiment in which the stabilization and hydration methods are combined. In this embodiment, at 1500, a sensor is connected to a sensor electronics device. After physically connecting the sensor to the sensor electronics device, at 1510, an AC signal (or DC signal) is applied to an electrode (e.g., a reference electrode) of the sensor. Simultaneously, or nearly simultaneously, at 1520, the microcontroller sends a signal to the DAC to apply a stabilization voltage sequence to the sensor. In an alternative embodiment, a stabilization current sequence may be applied to the sensor instead of a stabilization voltage sequence. At 1530, the detection circuit determines what level of AC signal (or DC signal) is present at the detection circuit's input terminal. At 1540, if there is a low-level AC signal (or DC signal) present at the detection circuit's input terminal, representing a highly attenuated AC signal (or DC signal), an interrupt is sent to the microcontroller. Because the microcontroller has already initiated the stabilization sequence, at 1550, the microcontroller receives the interrupt and sets a first indicator that the sensor is sufficiently hydrated. At 1555, once the stabilization sequence is complete, the microcontroller sets a second indicator indicating the completion of the stabilization sequence. Application of the stabilization sequence voltage causes the sensor, e.g., the working electrode, to generate a sensor signal at 1560, which is measured by the sensor signal measurement circuit and sent to the microcontroller. If, at 1570, the second indicator that the stabilization sequence is complete is set and the first indicator that hydration is complete is set, the microcontroller may utilize the sensor signal. If one or both of the indicators are not set, the microcontroller may not utilize the sensor signal because the sensor signal may not represent an accurate measurement of the subject's physiological measurement.

[0095] The hydration and stabilization process described above may generally be used as part of a larger continuous glucose monitoring (CGM) methodology. The current state of the art for continuous glucose monitoring is primarily incidental, meaning that readings provided by CGM devices (e.g., including implantable or subcutaneous sensors) cannot be used without a reference value to make clinical decisions. Such reference values ​​must generally be obtained from a fingerstick, for example, using a BG meter. Reference values ​​are necessary because the amount of information available from the sensor / sensing components is limited. Specifically, only the raw sensor value (i.e., sensor current or Isig) and the counter voltage, which is the voltage between the counter and reference electrodes (see, e.g., Figure 5), can be provided for processing by the sensing components. Therefore, during analysis, if the raw sensor signal appears abnormal (e.g., a decreasing signal), the only way to distinguish between sensor failure and a physiological change in the user / patient (i.e., a change in glucose levels within the body) may be by obtaining a reference glucose value via a fingerstick. As is known, a reference fingerstick is also used to calibrate the sensor.

[0096] Embodiments of the invention described herein are directed to advances and improvements in continuous glucose monitoring that result in more autonomous systems and associated devices and methodologies that may minimize or eliminate the requirement for a reference fingerstick, thereby enabling clinical decisions to be made with high confidence based on information obtained from the sensor signal alone. From a sensor design perspective, according to embodiments of the invention, such autonomy may be achieved through electrode redundancy, sensor redundancy (e.g., including complex redundancy between two or more sensors), sensor diagnostics, and Isig and / or sensor glucose (SG) fusion.

[0097] For purposes of the present discussion, and the present invention, "redundancy" refers to the presence / use of two or more electrodes, whether contained on / in a single probe (or "flex") or on / in two or more flexes, and "complex redundancy" refers to the presence / use of two (or more) sensors, where (at least) two of the sensors are identical. Thus, "redundant" electrodes may be contained on / in a single flex, on / in two or more flexes that are identical, or on / in two or more flexes that are not identical. As discussed further below, redundancy may be achieved, for example, by using multiple working electrodes to generate multiple signals indicative of a patient's blood glucose (BG) level. The multiple signals may be used to generate a fused glucose value as well as to assess the relative health of the (working) electrodes, the overall reliability of the sensor, and the required frequency of calibration reference values, if desired.

[0098] For example, it is known that acquiring signals from multiple electrochemical sensors can provide improved performance as a form of simple redundancy, either through multiple electrodes on the same probe (or flex) or by utilizing spatial separation and two separate probes. For example, Medtronic sells a hospital glucose sensor that includes two probes, with two working electrodes on each probe, generating four independent glucose signals.

[0099] In contrast to simple redundancy, orthogonal redundancy can be defined as two devices using two different technologies to reach the same goal, where the failure modes of the two devices are completely unique and do not intersect. Therefore, orthogonal redundancy can be created, for example, by combining optical and electrochemical sensing technologies. Clearly, the advantage of orthogonal redundancy is that two types of sensors, e.g., optical and electrochemical (or "echem") sensors, are subject to different types of interference, failure modes, and bodily responses. On the other hand, the use of two completely different technologies introduces additional design layers and computational complexity into the measurement and analysis of glucose levels in a patient's body.

[0100] Pseudo-orthogonal redundancy, on the other hand, can be achieved by utilizing the same techniques but with variations to generate complementary glucose measurements with minimal additional design and / or computational complexity. For example, two or more electrochemical sensors may be used, where one (or more) sensors are conventional peroxide-based sensors and one (or more) sensors measure glucose by the calculated difference in oxygen between two working electrodes (usually on the same sensor).

[0101] In yet another particular type of redundancy, discussed in more detail below, in embodiments of the invention herein, a sensor system using complex redundancy includes two (or more) sensors, (at least) two of which are dissimilar in design (and may also use different chemistries and / or sizes). Here, one (or more) of the sensors may be designed to have, for example, fairly good hydration and / or stabilization properties, but may not last two or three days. Meanwhile, the other sensor(s) may have long-term durability, but slow initial hydration and / or stabilization. In such cases, an algorithm may be designed to generate glucose data during early application using a first sensor(s), subsequently use the first sensor(s) to calibrate a second sensor(s), and then switch over to the second sensor(s) to generate glucose data for the remainder of the glucose sensor system's lifetime.

[0102] Sensor diagnostics involve the use of additional (diagnostic) information that can provide real-time insight into the health of a sensor. In this regard, electrochemical impedance spectroscopy (EIS) has been found to provide such additional information in the form of sensor impedance and impedance-related parameters at different frequencies. Advantageously, it has further been discovered that, for certain ranges of frequencies, the impedance and / or impedance-related data is substantially glucose-independent. This glucose independence enables the use of various EIS-based markers or indicators not only to generate robust and reliable sensor glucose values ​​(via fusion methodologies described in more detail below), but also to assess the condition, health, aging, and efficiency of individual electrode(s) and the sensor(s) as a whole, substantially independently of glucose-dependent Isig.

[0103] For example, analysis of glucose-independent impedance data provides information about the efficiency of the sensor, e.g., how quickly the sensor hydrates and becomes ready for data acquisition using values ​​for real impedance at 1 kHz, imaginary impedance at 1 kHz, and Nyquist slope (discussed in more detail below). Furthermore, glucose-independent impedance data provides information about potential blockage(s) that may exist on the sensor membrane surface, which may temporarily block the passage of glucose into the sensor and thus cause a dip in the signal (e.g., using values ​​for real impedance at 1 kHz). Additionally, glucose-independent impedance data provides information about loss of sensor sensitivity during prolonged wear, which may be due, for example, to local oxygen deficiency at the wear site (e.g., using values ​​for phase angle and / or imaginary impedance at frequencies above 1 kHz).

[0104] Within the context of electrode redundancy and EIS, a fusion algorithm may be used to obtain the diagnostic information provided by EIS for each redundant electrode and assess the reliability of each electrode individually. A weight, a measure of reliability, may then be added to each independent signal, and a single fusion signal may be calculated that can be used to generate a patient / subject-perceived sensor glucose value. As can be seen from the above, the combined use of redundancy, EIS-based sensor diagnostics, and an EIS-based fusion algorithm can improve the reliability of the overall CGM system. Additionally, EIS diagnostics autonomously probe the health of each electrode without requiring a reference glucose value (fingerstick), thereby reducing the number of required reference values.

[0105] EIS, or AC impedance, studies a system's response to the application of a periodic, small-amplitude AC signal. This is shown illustratively in FIG. 15A, where E is the applied potential, I is the current, and impedance (Z) is defined as ΔE / ΔI. However, although impedance itself can be mathematically defined simply as ΔE / ΔI, to date, there has been no commercial success in applying EIS technology to continuous glucose monitoring. This is in part because glucose sensors are highly complex systems, and to date, no mathematical model has been developed that can fully explain the complexity of a glucose sensor's EIS output.

[0106] One simplified electrical circuit model that has been used to describe electrochemical impedance spectroscopy is shown in Figure 15B. In this illustration, IHP represents the inner Helmholtz plane, OHP represents the outer Helmholtz plane, CE represents the counter electrode, WE represents the working electrode, and C d is the double layer capacitance, R p is the polarization resistance, Z w is the Warburg impedance, R s is the solution resistance. Each of the latter four components, double layer capacitance (C d ), Warburg impedance (Z w ), polarization resistance (R p ), and solution resistance (Rs ) can play an important role in sensor performance and can be measured individually by applying low- or high-frequency interaction potentials. For example, the Warburg impedance is closely related to the diffusion impedance of the electrochemical system, which is primarily a low-frequency impedance, and is therefore present in all diffusion-limited electrochemical sensors. Therefore, EIS techniques may be used as a sensor diagnostic tool by correlating one or more of these components with one or more components and / or layers of a glucose sensor.

[0107] As is known, impedance is defined in terms of its magnitude and phase, where magnitude (|Z|) is the ratio of the voltage difference amplitude to the current amplitude, and phase (θ) is the phase shift by which the current leads the voltage. When a circuit is driven solely by direct current (DC), impedance is the same as resistance, i.e., resistance is the special case of impedance with a zero phase angle. However, as a complex quantity, impedance may also be expressed in terms of its real and imaginary parts. In this regard, real and imaginary impedances can be derived from the magnitude and phase of the impedance using the following formula: Real impedance (ω) = magnitude (ω) × cos(phase (ω) / 180 × π) Imaginary impedance (ω) = magnitude (ω) × sin(phase (ω) / 180 × π) where ω represents the input frequency at which the magnitude (ohms) and phase (degrees) are measured. The relationship between impedance on the one hand and current and voltage on the other hand, including how the former may be calculated based on measurements of the latter, is discussed more fully below in connection with the sensor electronics, including application specific integrated circuits (ASICs), developed for use in embodiments of the invention described herein.

[0108] Continuing with the circuit model shown in FIG. 15B, the total system impedance can be simplified to:

number

number

[0109] A Nyquist plot is a graphical representation in which the real part of the impedance (Real Z) is plotted against its imaginary part (Img Z) across a spectrum of frequencies. Figure 16A shows a generalized example of a Nyquist plot, where the X value is the real part of the impedance and the Y value is the imaginary part of the impedance. The phase angle is the angle between the impedance point (X,Y) and the X axis, which defines a vector with magnitude |Z|.

[0110] The Nyquist plot in FIG. 16A is generated by applying an AC voltage plus a DC voltage (DC bias) between the working and counter electrodes at selected frequencies from 0.1 Hz to 1000 MHz (i.e., a frequency sweep). Starting from the right, the frequency increases from 0.1 Hz. Using each frequency, real and imaginary impedances can be calculated and plotted. As shown, a typical Nyquist plot for an electrochemical system may appear as a semicircle joined by a straight line at an inflection point, with the semicircle and line representing the plotted impedance. In certain embodiments, the impedance at the inflection point is of particular interest because it is the easiest to identify on the Nyquist plot and may define the intercept. Typically, the inflection point is close to the X-axis, and the X-value at the inflection point is the sum of the polarization resistance and the solution resistance (R p +R s ) is approximated.

[0111] Referring to FIG. 16B , a Nyquist plot can generally be described in terms of a low-frequency region 1610 and a high-frequency region 1620, where the terms “high frequency” and “low frequency” are used in a relative sense and are not intended to be limiting. Thus, for example, the low-frequency region 1610 may illustratively include data points obtained for a frequency range of about 0.1 Hz to about 100 Hz (or higher), and the high-frequency region 1620 may illustratively include data points obtained for a frequency range of about 1 kHz (or lower) to about 8 kHz (and higher). In the low-frequency region 1610, the Nyquist slope represents the slope of a linear fit 1630 of the low-frequency data points in the Nyquist plot. As shown, in the high-frequency region 1620, the value of the imaginary impedance is minimal and may become negligible. Thus, the intercept 1600 is essentially the value of the real impedance at high frequencies (e.g., in this case, in the range of about 1 kHz to 8 kHz). In FIG. 16B, the intercept 1600 is about 25 kilohms.

[0112] Figures 16C and 16D illustrate how the glucose sensor responds to a sinusoidal (i.e., alternating) working potential. In these figures, GLM is the glucose-limiting membrane of the sensor, AP is the adhesion promoter, HSA is human serum albumin, GOX is the glucose oxidase enzyme (layer), and E dc is the DC potential, Eac is the AC potential, and C´ peroxide 過酸化物 is the peroxide concentration during AC application. As shown in Figure 16C, when the sensor diffusion length, which is a function of AC potential frequency, molecular diffusion coefficient, and membrane thickness, is small compared to the membrane (GO) length, the system gives a relatively linear response (i.e., infinite) with a constant phase angle. In contrast, when the diffusion length is equal to the membrane (GO) length, the system response becomes finite, resulting in a semicircular Nyquist plot, as shown in Figure 16D. The latter is typically true for low-frequency EIS where non-Faraday processes are negligible.

[0113] When performing EIS analysis, AC voltages and DC biases of various frequencies may be applied, for example, between a working electrode and a reference electrode. In this respect, EIS offers an improvement over previous methodologies that may have been limited to simple DC currents or AC voltages of a single frequency. Generally, EIS may be performed at frequencies ranging from μHz to MHz, although a narrower range of frequencies (e.g., about 0.1 Hz to about 8 kHz) may be sufficient for the embodiments of the invention described herein. Thus, in some embodiments, an AC potential may be applied within a frequency range of about 0.1 Hz to about 8 kHz, with a programmable amplitude of at least 100 mV, preferably about 50 mV.

[0114] Within the above frequency range, relatively high frequencies, i.e., frequencies generally in the range of about 1 kHz to about 8 kHz, are used to probe the capacitive nature of the sensor. Depending on the film thickness and transmittance, a typical range of impedance at relatively high frequencies may be, for example, about 500 ohms to 25 kilohms, and a typical range of phase may be, for example, 0 degrees to -40 degrees. On the other hand, relatively low frequencies, i.e., frequencies generally in the range of about 0.1 Hz to about 100 Hz, are used to probe the resistive nature of the sensor. Here, depending on the electrode design and degree of metallization, a typical functional range of output real impedance may be, for example, about 50 kilohms to 300 kilohms, and a typical range of phase may be between about -50 degrees and about -90 degrees. The above exemplary ranges are shown, for example, in the Bode plots of Figures 16E and 16F.

[0115] As mentioned above, the terms "high frequency" and "low frequency" are meant to be used relative to each other, not in an absolute sense, and are meant to be illustrative and not limiting, as are the typical impedance and phase ranges described above. However, the basic principle remains the same: the capacitive and resistive behavior of a sensor can be probed by analyzing impedance data across the frequency spectrum, with lower frequencies typically providing more information about resistive components (e.g., electrodes) and higher frequencies providing more information about capacitive components (e.g., membranes). However, the actual frequency range in each case depends on the overall design, including, for example, the type(s) of electrode(s), the surface area of ​​the electrode(s), the membrane thickness, the membrane permeability, etc. See also FIG. 15B for the general correspondence between high-frequency circuit components and the sensor membrane, as well as between low-frequency circuit components and Faradaic processes, including, for example, the electrode(s).

[0116] EIS may be used in sensor systems that include a single working electrode, as well as in sensor systems where the sensor includes multiple (redundant) working electrodes. In one embodiment, EIS provides valuable information about the aging (or aging) of a sensor. Specifically, the magnitude and phase angle of the impedance change at various frequencies. As can be seen in FIG. 17, the sensor impedance, particularly the sum of Rp and Rs, reflects the age of the sensor as well as its operating conditions. Therefore, as can be seen from the various plots in FIG. 17, new sensors typically have higher impedance than used sensors. Thus, by considering the X value of the sum of Rp and Rs, a threshold can be used to determine when the sensor has aged beyond its specified operating life. For the illustrative examples shown in FIGS. 17-21 and described below, the real impedance value at the inflection point (i.e., Rp+Rs) is used to determine the aging, state, stabilization, and hydration of the sensor; however, it should be noted that alternative embodiments of the sensor may use other EIS-based parameters in addition to or instead of the real impedance, such as, for example, imaginary impedance, phase angle, Nyquist slope, etc.

[0117] FIG. 17 illustrates an example Nyquist plot over the lifetime of a sensor. The points indicated by the arrows are the respective inflection points for each sweep across the frequency spectrum. For example, before initialization (time t = 0), Rs + Rp was higher than 8.5 kOhms, and after initialization (time t = 0.5 hours), the Rs + Rp value dropped to less than 8 kOhms. Over the next six days, Rs + Rp continued to decrease, causing Rs + Rp to drop below 6.5 kOhms at the end of the specified sensor life. Based on this example, a threshold can be set to specify when the Rs + Rp value will indicate the end of the sensor's specified operating life. Thus, the EIS approach can plug the eyelet that allows the sensor to be reused beyond the specified operating time. That is, if the patient attempts to reuse the sensor by disconnecting and reconnecting it after the sensor has reached its specified operating time, the EIS will measure an abnormally low impedance, which can cause the system to reject the sensor and request a new one from the patient.

[0118] Additionally, EIS may enable detection of sensor failure by detecting when the sensor's impedance drops below a low impedance threshold level. In that case, the system may terminate the sensor before its specified operational life. As discussed in more detail below, sensor impedance may also be used to detect other sensor failure modes. For example, if the sensor enters a low current state (i.e., sensor failure) due to a variety of reasons, the sensor impedance may also increase above a certain high impedance threshold. If the impedance becomes abnormally high during sensor operation due to protein or polypeptide fouling, macrophage adhesion, or any other factor, the system may also terminate the sensor before its specified operational life.

[0119] FIG. 18 illustrates how EIS techniques can be applied during sensor stabilization and in detecting sensor aging, according to certain embodiments. The logic of FIG. 18 begins at 1800 after the previously described hydration and sensor initialization procedures are completed. That is, the sensor is deemed sufficiently hydrated, and an initial initialization procedure is applied to initialize the sensor. The initialization procedure may preferably take the form of a voltage pulse, as previously described in the detailed description. However, in alternative embodiments, a different waveform may be used for the initialization procedure. For example, a sine wave may be used instead of a pulse to accelerate wetting or conditioning of the sensor. Additionally, a portion of the waveform may be required to be greater than the sensor's normal operating voltage, i.e., 0.535 volts.

[0120] At block 1810, an EIS procedure is applied, comparing the impedance to both a first high threshold and a first low threshold. Example first high and first low thresholds are 7 kilohms and 8.5 kilohms, respectively, although the values ​​can be set higher or lower as needed. If the impedance, e.g., Rp + Rs, is higher than the first high threshold, the sensor undergoes an additional initialization procedure (e.g., application of one or more additional pulses) at block 1820. Ideally, the total number of initialization procedures applied to initialize the sensor would be optimized to limit the impact on both the sensor's battery life and the overall amount of time required to stabilize the sensor. Therefore, by applying EIS, fewer initializations can be performed initially, and the number of initializations can be added incrementally to provide an appropriate amount of initialization to prepare the sensor for use. Similarly, in an alternative embodiment, EIS can be applied to the hydration procedure to minimize the number of initializations required to assist the hydration process, as described in FIGS. 13-14 .

[0121] On the other hand, if the impedance, e.g., Rp+Rs, falls below a first low threshold, the sensor will be determined to be faulty, in which case the process will immediately terminate at block 1860. A message will be given to the user to replace the sensor and restart the hydration process. If the impedance is within the high and low thresholds, the sensor will begin operating normally at block 1830. The logic proceeds to block 1840, where an additional EIS is performed to check the aging of the sensor. The first time the logic reaches block 1840, the microcontroller will perform EIS to age the sensor and close an eyelet that allows the user to plug in and out the same sensor. When the logic returns to block 1840, in future iterations of the EIS procedure, the microprocessor will perform EIS at fixed intervals for the specified life of the sensor. In a preferred embodiment, the fixed intervals are set to two hours, although longer or shorter periods can easily be used.

[0122] At block 1850, the impedance is compared to a second set of high and low thresholds. Examples of such second high and low thresholds may be 5.5 kilohms and 8.5 kilohms, respectively, although the values ​​can be set higher or lower as needed. As long as the impedance value is within the second high and low thresholds, the logic proceeds to block 1830, and the sensor operates normally until the specified sensor life is reached, e.g., 5 days. Of course, as described with respect to block 1840, EIS would be performed at regularly scheduled intervals throughout the specified sensor life. However, if, after EIS is performed, it is determined at block 1850 that the impedance has dropped below the second low threshold or risen above the second high threshold, the sensor is terminated at block 1860. In a further alternative embodiment, a secondary check for a faulty sensor reading can be implemented. For example, if the EIS indicates that the impedance is outside of a second high and low threshold, the logic may perform a second EIS to confirm that the second set of thresholds is in fact not met (to confirm that the first EIS was performed correctly) before determining the end of the sensor at block 1860.

[0123] FIG. 19 builds on the above discussion and details a possible schedule for performing diagnostic EIS procedures. Each diagnostic EIS procedure is optional, and it is possible to schedule no diagnostic EIS procedure or combine one or more diagnostic EIS procedures, if desired. The schedule in FIG. 19 begins with sensor insertion at point 1900. Following sensor insertion, the sensor undergoes a hydration period 1910. As previously discussed, this hydration period is important because an insufficiently hydrated sensor can provide an inaccurate reading to the user. A first optional diagnostic EIS procedure at point 1920 is scheduled during this hydration period 1910 to ensure the sensor is sufficiently hydrated. The first diagnostic EIS procedure 1920 measures sensor impedance values ​​to determine whether the sensor is sufficiently hydrated. If the first diagnostic EIS procedure 1920 determines that the impedance is within established high and low thresholds, indicating sufficient hydration, the sensor controller will allow sensor activation at point 1930. Conversely, if the first diagnostic EIS procedure 1920 determines that the impedance is outside the set high and low thresholds, indicating insufficient hydration, the sensor hydration period 1910 may be extended. After an extended period of hydration, when a certain capacitance is reached between the sensor's electrodes, the sensor is sufficiently hydrated and activation may occur at point 1930.

[0124] A second optional diagnostic EIS procedure 1940 is scheduled after sensor power-up at point 1930 but before sensor initialization begins at 1950. This scheduled second diagnostic EIS procedure 1940 can detect whether a sensor is being reused before initialization begins at 1950. The test to determine whether a sensor is being reused was detailed in the description of FIG. 18 . However, unlike the immediately preceding description of FIG. 18 , in which the aging test occurs after initialization is complete, the aging test is shown in FIG. 19 as being performed before initialization. It is important to understand that the timeline of the EIS procedure described in FIG. 19 can be rearranged and the order of some of the steps can be swapped without affecting the overall teachings of the present application. As previously described, the second diagnostic EIS procedure 1940 detects reused sensors by determining the sensor's impedance value and then comparing it to set high and low thresholds. If the impedance is outside the set thresholds, indicating the sensor is being reused, the sensor is rejected and the user is prompted to replace it with a new sensor. This prevents problems that may arise from reusing an old sensor. Conversely, if the impedance is within the set threshold, sensor initialization 1950 can begin with confidence that a new sensor is being used.

[0125] A third optional diagnostic EIS procedure 1960 is scheduled after initialization begins at point 1950. The third diagnostic EIS procedure 1960 tests the sensor's impedance value to determine whether the sensor is fully initialized. The third diagnostic EIS procedure 1960 should be run for the minimum amount of time necessary to fully initialize the sensor. Running it at this point maximizes sensor life by limiting the amount of time a fully initialized sensor is unused, and over-initialization is avoided by ensuring the sensor is fully initialized before over-initialization occurs. Preventing over-initialization is important because it can suppress current and cause inaccurate readings. However, under-initialization is also a problem, and if the third diagnostic EIS procedure 1960 indicates the sensor is under-initialized, an optional initialization may be run at point 1970 to fully initialize the sensor. Under-initialization is disadvantageous because it generates excessive current that is unrelated to the actual glucose concentration. Because of the risk of under-initialization and over-initialization, the third diagnostic EIS procedure plays an important role in ensuring the sensor functions properly when in use.

[0126] Additionally, an optional periodic diagnostic EIS procedure 1980 can be scheduled for a time after the sensor is fully initialized. The EIS procedure 1980 can be scheduled at set intervals. As described in more detail below, the EIS procedure 1980 may also be triggered by other sensor signals, such as an abnormal current or an abnormal counter electrode voltage. Additionally, fewer or more EIS procedures 1980 can be scheduled, as desired. In a preferred embodiment, the EIS procedure used during the hydration process, sensor life check, initialization process, or periodic diagnostic test is the same procedure. In alternative embodiments, for various EIS procedures, the EIS procedure can be shortened or lengthened (i.e., a fewer or greater range of frequencies is tested) depending on the need to focus on specific impedance ranges. The periodic diagnostic EIS procedure 1980 monitors impedance values ​​to ensure the sensor continues to operate at an optimal level.

[0127] If the sensor current drops due to contaminant species, sensor aging, or a combination of contaminant species and sensor aging, the sensor may not be operating at an optimal level. Sensors that have aged beyond a certain period of time are no longer usable, whereas sensors hindered by contaminant species may be repaired. Contaminant species can reduce the electrode surface area or diffusion paths for analytes and reaction byproducts, thereby causing the sensor current to drop. These contaminant species become charged and gradually collect on the electrode or membrane surface under a certain voltage. Previously, contaminant species impaired the usefulness of the sensor. Now, if the periodic diagnostic EIS procedure 1980 detects impedance values ​​indicative of the presence of contaminant species, remedial action can be taken. Instances where remedial action is taken are described with reference to FIG. 20. Periodic diagnostic EIS procedure 1980 can therefore be extremely useful, as it can trigger sensor remedial action that can return the sensor current to normal levels and potentially extend the sensor's lifespan. Two possible embodiments of sensor remedial action are described below in the discussion of FIGS. 21A and 21B.

[0128] Additionally, a scheduled diagnostic EIS procedure 1980 may be interrupted or rescheduled if a particular event is determined to be imminent. Such events include any situation in which the patient needs to confirm a sensor reading, including, for example, when a patient measures their BG reading using a test strip meter to calibrate the sensor, when a patient is alerted to a calibration error and the need to measure their BG reading a second time using a test strip meter, or when a high or low blood glucose warning has been issued but not confirmed.

[0129] FIG. 20 illustrates a method for combining a diagnostic EIS procedure with sensor remediation measures. The diagnostic procedure of block 2000 can be any of the periodic diagnostic EIS procedures 1980, as detailed in FIG. 19. The logic of this method begins when a diagnostic EIS procedure is performed in block 2000 to detect a sensor impedance value. As previously described, in certain embodiments, the EIS procedure applies a combination of DC bias and AC voltage at various frequencies, and the impedance detected by performing the EIS procedure is mapped to a Nyquist plot, with the inflection points of the Nyquist plot approximating the sum of the polarization resistance and the solution resistance (i.e., the real impedance value). After the diagnostic EIS procedure of block 2000 detects the sensor impedance value, the logic moves to block 2010.

[0130] In block 2010, the impedance value is compared to set high and low thresholds to determine whether the impedance value is normal. If the impedance is within the set boundaries of the high and low thresholds in block 2010, normal sensor operation resumes in block 2020, and the logic of FIG. 20 ends until such time as another diagnostic EIS procedure is scheduled. Conversely, if the impedance is determined to be abnormal in block 2010 (i.e., outside the set boundaries of the high and low thresholds), remedial action is triggered in block 2030. Example high and low thresholds that would be acceptable over the life of the sensor would be 5.5 kilohms and 8.5 kilohms, respectively, although the values ​​can be set higher or lower as needed.

[0131] Remedial action is taken in block 2030 to remove any contaminating species that may have caused the abnormal impedance value. In a preferred embodiment, the remedial action is taken by applying a reverse current or voltage between the working and reference electrodes. The details of the remedial action are described in more detail with respect to FIG. 21 . After the remedial action is taken in block 2030, the impedance value is again tested by a diagnostic EIS procedure in block 2040. The impedance value from the diagnostic EIS procedure in block 2040 is then compared to a set high or low threshold, and the success of the remedial action is then determined in block 2050. Similar to block 2010, if the impedance is within the set threshold, the impedance is considered normal; if the impedance is outside the set threshold, the impedance is considered abnormal.

[0132] If the sensor impedance value is determined to have returned to normal at block 2050, normal sensor operation at block 2020 will occur. If the impedance is still abnormal, indicating that sensor aging is the cause of the abnormal impedance or that remedial actions to remove contaminant species were unsuccessful, the sensor will terminate at block 2060. In an alternative embodiment, instead of immediately terminating the sensor, the sensor may first generate a sensor message requesting the user to wait and then perform further remedial actions after a set time has elapsed. This alternative step may be combined with another logic that determines whether the impedance value is approaching the boundaries of high and low thresholds after the first remedial action is performed. For example, if no change is observed in the sensor impedance value, the sensor may decide to terminate. However, if the sensor impedance value is approaching the preset boundaries but is still outside the boundaries after the first remedial action, additional remedial actions may be performed. In yet another alternative embodiment, the sensor may generate a message requesting the user to calibrate the sensor by performing a finger stick meter measurement to further verify whether the sensor is indeed faulty. All of the above embodiments act to prevent a user from using a defective sensor that produces an inaccurate reading.

[0133] FIG. 21A illustrates one embodiment of the aforementioned sensor remediation procedure. In this embodiment, blockages caused by contaminating species are removed by reversing the voltage applied to the sensor between the working and reference electrodes. The reverse DC voltage lifts the charged contaminating species from the electrode or membrane surface, clearing the diffusion pathway. Once the pathway is cleared, the sensor current returns to normal levels and the sensor can provide accurate readings. Therefore, the remediation procedure saves the user the time and expense associated with replacing an otherwise valid sensor.

[0134] FIG. 21B illustrates an alternative embodiment of the sensor remediation procedure described above. In this embodiment, a reverse DC voltage applied between the working and reference electrodes is combined with an AC voltage. By adding the AC voltage, the AC voltage extends the force of the AC voltage farther from the electrodes and penetrates all layers of the sensor, thereby removing tightly adsorbed or surface species. The AC voltage can be generated in any number of different waveforms. Some examples of waveforms that can be used include square waves, triangular waves, sine waves, or pulses. As with the previous embodiment, once the contaminating species are cleared, the sensor returns to normal operation, improving both sensor life and accuracy.

[0135] While the above examples primarily illustrate the use of real impedance data in sensor diagnostics, embodiments of the invention described herein also contemplate the use of other EIS-based substantially analyte-independent parameters (in addition to real impedance) in sensor diagnostic procedures. For example, as discussed above, analysis of (substantially) glucose-independent impedance data, such as the 1 kHz real impedance and 1 kHz imaginary impedance and Nyquist slope, provides information about the efficiency of the sensor, relating to how quickly the sensor hydrates and becomes ready for data acquisition. Furthermore, glucose-independent impedance data, such as the 1 kHz real impedance value, provides information about potential blockage(s) that may be present on the sensor membrane surface, which may temporarily block the passage of glucose into the sensor and thus cause the signal to dip.

[0136] In addition, (substantially) glucose-independent impedance data, such as high-frequency phase angle and / or imaginary impedance values ​​at frequencies above 1 kHz, can provide information about loss of sensor sensitivity due to prolonged wear, which may be attributed to local oxygen deficiency at the insertion site. In this regard, the underlying mechanism of sensitivity loss caused by oxygen deficiency can be explained as follows: When local oxygen is deficient, the sensor output (i.e., Isig and SG) becomes dependent on oxygen rather than glucose, and thus the sensor loses sensitivity to glucose. Other markers, including 0.1 Hz real impedance, counter electrode voltage (Vcntr), and EIS-induced spikes in Isig, may also be used to detect sensitivity loss caused by oxygen deficiency. Furthermore, in sensor systems with redundant electrodes, the relative differences in 1 kHz real impedance, 1 kHz imaginary impedance, and 0.1 Hz real impedance between two or more working electrodes may be used to detect sensitivity loss due to biofouling.

[0137] According to embodiments of the invention described herein, EIS-based sensor diagnostics involve the consideration and analysis of EIS data related to one or more of at least three primary potential sensor failure modes: (1) signal activation, (2) signal dip, and (3) sensitivity loss. Importantly, the discovery that most impedance-related parameters used in such diagnostic analyses and procedures are examined at frequencies or within frequency ranges where the parameters are substantially analyte-independent allows for the implementation of sensor diagnostic procedures independent of the level of analyte in the patient's body. Thus, while EIS-based sensor diagnostics may be driven by, for example, large variations in Isig that are analyte-dependent, the impedance-related parameters used in such sensor diagnostic procedures are themselves substantially independent of analyte levels. As discussed in more detail below, in most situations where glucose may be found to affect the magnitude (or other characteristic) of an EIS-based parameter, such effects are typically sufficiently small—e.g., at least an order of magnitude difference between the EIS-based measurement and the effect of glucose thereon—that the glucose effect can be filtered out of the measurement, for example, via software in the IC.

[0138] By definition, "start-up" refers to the integrity of the sensor signal during the first few hours after insertion (e.g., t = 0–6 hours). For example, in many current devices, the signal during the first two hours after insertion is considered unreliable, and thus the sensor glucose value is hidden from the patient / user. In situations where the sensor takes a long time to hydrate, the sensor signal is low for several hours after insertion. With EIS, additional impedance information is available immediately after inserting the sensor (by performing an EIS procedure). In this regard, it is possible to use the equation for total impedance to explain the principle behind low start-up detection using real impedance at 1 kHz. At relatively high frequencies, in this case above 1 kHz, the imaginary impedance is very small (as confirmed by in vivo data), and the total impedance simplifies to:

number

[0139] As the sensor gradually becomes fully wet, the double layer capacitance (C d ) increases. As a result, the total impedance increases as shown in the above equation. d , the total impedance will decrease. This is illustrated, for example, in the form of intercept 1600 on the real impedance axis shown in Figure 16B. Importantly, the imaginary impedance at 1 kHz can also be used for the same purpose, as it contains a capacitive component and is inversely proportional to the capacitive component.

[0140] Another marker of low activation detection is the Nyquist slope, which depends only on the impedance at relatively low frequencies, which corresponds to the Warburg impedance component of the total impedance (see, for example, FIG. 15B). FIG. 22 shows a Nyquist plot of a properly functioning sensor, with arrow A indicating the progression of time starting at t=0, i.e., the time of sensor attachment. Thus, EIS at relatively low frequencies is performed immediately after sensor insertion (time t=0), producing real and imaginary impedance data that are plotted with a first linear fit 2200 having a first (Nyquist) slope. At a time interval after t=0, a second (lower) frequency sweep is performed, which produces a second linear fit 2210 having a second (Nyquist) slope that is greater than the first Nyquist slope, and so on. As the sensor becomes more hydrated, the Nyquist slope increases and the intercept decreases, becomes steeper, and moves closer to the y-axis, as reflected by lines 2200, 2210, etc. Related to low-initialization detection, clinical data indicates that there is generally a sharp increase in the Nyquist slope after sensor insertion and initialization, which then stabilizes to a certain level. One explanation for this is that as the sensor becomes increasingly wet, the diffusion coefficients and concentrations of species undergo dramatic changes, which are reflected in the Warburg impedance.

[0141] In FIG. 23A, the Isig 2230 of the first working electrode, WE1, starts lower than expected (at approximately 10 nA) and takes a while to catch up with the Isig 2240 of the second working electrode, WE2. Therefore, in this particular example, WE1 is designated as having low activation. The EIS data reflects this low activation in two ways. First, as shown in FIG. 23A, the real impedance at 1 kHz (2235) of WE1 is much higher than the real impedance at 1 kHz (2245) of WE2. Second, compared to the Nyquist slope of WE2 (FIG. 23C), the Nyquist slope of WE1 (FIG. 23B) starts lower, has a larger intercept 2237, and takes more time to stabilize. As described below, these two signatures, the 1 kHz real impedance and the Nyquist slope, can be used as diagnostic inputs to the fusion algorithm to determine which of the two electrodes can hold a higher weight when calculating the fused signal. Additionally, one or both of these markers may be used in a diagnostic procedure to determine whether the sensor is generally acceptable or whether it should be retired and replaced.

[0142] By definition, a signal (or Isig) dip refers to an instance of low sensor signal, which is in fact almost always temporary, lasting, for example, a few hours. Such a low signal can be caused, for example, by some form of biological blockage on the sensor surface or by pressure applied to the insertion site (e.g., while sleeping in a lateral position). During this period, the sensor data is considered unreliable, although the signal eventually recovers. In EIS data, as shown in Figure 24, this type of signal dip is reflected in the 1 kHz real impedance data, as opposed to one caused by blood glucose changes in the patient's body.

[0143] Specifically, in Figure 24, both Isig 2250 for the first working electrode, WE1, and Isig 2260 for the second working electrode, WE2, start at approximately 25 nA (i.e., at 6:00 PM) at the far left. As time progresses, both Isigs fluctuate, reflecting glucose fluctuations near the sensors. For the first 12 hours or so (i.e., until 6:00 AM), both Isigs are fairly stable, as are their respective 1 kHz real impedances 2255 and 2265. However, from approximately 12 to 18 hours, i.e., from 6:00 AM to noon, Isig 2260 for WE2 begins to dip and continues its downward trend for the next few hours until approximately 9:00 PM. During this period, Isig 2250 for WE1 also shows some dipping, but Isig 2250 is much more stable than Isig 2260 for WE2, with significantly fewer dips. The behavior of the Isigs for WE1 and WE2 is also reflected in their respective 1 kHz real impedance data. Thus, as shown in Figure 24, the 1 kHz real impedance of WE1 (2255) remains fairly stable over the above period, while there is a significant increase in the 1 kHz real impedance of WE2 (2265).

[0144] By definition, sensitivity loss refers to the sensor signal (Isig) becoming low and unresponsive for an extended period of time and is usually irreversible. Sensitivity loss can occur for a variety of reasons. For example, electrode poisoning rapidly reduces the active surface area of ​​the working electrode, thereby severely limiting current amplitude. Sensitivity loss can also occur due to hypoxia or lack of oxygen at the insertion site. In addition, sensitivity loss can occur due to certain forms of extreme surface obscuration (i.e., more permanent forms of signal dips caused by biological or other factors) that limit the passage of both glucose and oxygen through the sensor membrane, thereby reducing the number / frequency of chemical reactions that generate current at the electrode and ultimately the sensor signal (Isig). Note that the various causes of sensitivity loss described above apply to both short-term (7–10 days of wear) and long-term (6 months of wear) sensors.

[0145] In EIS data, a loss of sensitivity is often preceded by an increase in the absolute value of the phase (|phase|) and the absolute value of the imaginary impedance (|imaginary impedance|) in the relatively high frequency range (e.g., above 128 Hz and above 1 kHz, respectively). Figure 25A shows an example of a properly functioning glucose sensor where the sensor current 2500 responds to glucose, i.e., Isig 2500 tracks glucose fluctuations, but the real impedance at 1 kHz 2510, the imaginary impedance at 1 kHz 2530, and the phase (2520) for frequencies above about 128 Hz remain stable, for example, because they are substantially glucose independent.

[0146] Specifically, the top graph of FIG. 25A shows that after the first few hours, the real impedance 2510 at 1 kHz remains fairly stable at about 5 kilohms (and the imaginary impedance 2530 at 1 kHz remains fairly stable at about −400 ohms). That is, at 1 kHz, the real impedance data 2510 and imaginary impedance data 2530 are substantially glucose-independent, and as a result, they can be used as signatures or independent indicators of the health, condition, and ultimately the reliability of the particular sensor being analyzed. However, as noted above, different impedance-related parameters may exhibit glucose independence at different frequency ranges, which in any case may depend on the overall sensor design, e.g., electrode type, electrode surface area, membrane thickness, membrane permeability, etc.

[0147] Thus, in exemplary FIG. 25B, for the 90% short tubeless electrode design, the top graph again shows that the sensor current 2501 responds to glucose and that the real impedance 2511 at 1 kHz remains fairly stable at about 7.5 kilohms after the first few hours. The bottom graph in FIG. 25B shows the real impedance data for frequencies from 0.1 Hz (2518) to 1 kHz (2511). As can be seen, the real impedance data at 0.1 Hz (2518) is quite glucose dependent. However, as indicated by reference numerals 2516, 2514, and 2512, the real impedance becomes increasingly glucose independent as the frequency increases from 0.1 Hz to 1 kHz, i.e., for impedance data measured at frequencies approaching 1 kHz.

[0148] Returning to FIG. 25A, the middle graph shows that phase 2520 at relatively high frequencies is substantially glucose-independent. Note, however, that "relatively high frequencies" in relation to this parameter (phase) of the sensor under analysis refers to frequencies above 128 Hz. In this regard, the graph shows that phase for all frequencies between 128 Hz and 8 kHz is stable throughout the time period shown. Meanwhile, as can be seen in the bottom graph of FIG. 25C, phase 2522 at 128 Hz (and above) is stable, but with increasing frequencies below 128 Hz, phase 2524 fluctuates, i.e., phase 2524 becomes increasingly glucose-dependent to varying degrees. Note that the electrode design for the example in FIG. 25C is the same as that used in FIG. 25B, and the top graph in the former is identical to the top graph in the latter.

[0149] FIG. 26 shows an example of sensitivity loss due to oxygen deprivation at the insertion site. In this case, the insertion site becomes oxygen-deprived shortly after day 4 (indicated by the dark vertical line in FIG. 26), causing the sensor current 2600 to be low and unresponsive. The 1 kHz real impedance 2610 remains stable, indicating that there is no physical blockage in the sensor. However, changes in the relatively high frequency phase 2622 and the 1 kHz imaginary impedance 2632, as indicated by the respective downward arrows, are consistent with a loss of sensitivity, indicating that this type of loss is due to oxygen deprivation at the insertion site. Specifically, FIG. 26 shows that before the sensor loses sensitivity, the phase at high frequencies (2620) and the 1 kHz imaginary impedance (2630) become more negative, as indicated by the dark vertical line, and continue their downward trend as sensor sensitivity loss continues. Therefore, as noted above, an increase in the absolute value of the phase (|phase|) and the absolute value of the imaginary impedance (|imaginary impedance|) in the relatively high frequency range (e.g., above 128 Hz and above 1 kHz, respectively) precedes or predicts this sensitivity loss.

[0150] The above-described signatures can be verified by in vitro testing, an example of which is shown in FIG. 27. FIG. 27 shows the results of in vitro testing of the sensor, simulating oxygen deprivation at various glucose concentrations. In the top graph, Isig varies with glucose concentration as the glucose concentration increases from 100 mg / dL (2710) to 200 mg / dL (2720), 300 mg / dL (2730), and 400 mg / dL (2740), and then decreases to 200 mg / dL (2750). In the bottom graph, the phase at relatively high frequencies is generally stable and glucose-independent. However, at very low oxygen concentrations, such as 0.1% O, the phase at relatively high frequencies fluctuates, as indicated by the circled region and arrows 2760 and 2770. Note that the magnitude and / or direction (i.e., positive or negative) of the fluctuations depend on various factors. For example, the higher the ratio of glucose concentration to oxygen concentration, the higher the magnitude of phase fluctuations. In addition, the specific sensor design as well as the age of the sensor (i.e., as measured in time since implantation) will affect such fluctuations. Thus, for example, the older the sensor, the more susceptible it will be to fluctuations.

[0151] 28A-28D show another example of sensitivity loss caused by oxygen depletion using redundant working electrodes WE1 and WE2. As shown in FIG. 28A, the real impedance 2810 at 1 kHz remains stable, even as the sensor current 2800 fluctuates and eventually becomes unresponsive. And, as before, the change in the imaginary impedance 2820 at 1 kHz is consistent with a loss of sensor sensitivity. However, in addition, FIG. 28B shows real and imaginary impedance data at 0.105 Hz (2830 and 2840, respectively). The latter is sometimes more commonly referred to as the "0.1 Hz data," but while the imaginary impedance at 0.1 Hz appears fairly stable, the real impedance 2830 at 0.1 Hz increases significantly as the sensor loses sensitivity. Furthermore, as shown in FIG. 28C, the loss of sensitivity due to oxygen depletion results in a significant increase in V cntrThe 2850 rails to 1.2 volts.

[0152] In summary, this figure shows that the loss of sensitivity caused by oxygen depletion is due to a lowered 1 kHz imaginary impedance (i.e., the latter becomes more negative) and an increased 0.105 Hz real impedance (i.e., the latter becomes more positive), and V cntr Furthermore, the oxygen depletion process and V cntr -rail is often associated with an increased capacitive component in the electrochemical circuit. Note that in some of the diagnostic procedures described below, the 0.105 Hz real impedance may not be used, as it appears that this relatively low frequency real impedance data may be analyte dependent.

[0153] Finally, with respect to the example of Figures 28A-28D, it should be noted that impedance measurements at frequencies above 1 kHz typically cause EIS-induced spikes in Isig. This is shown in Figure 28D, where the raw Isig for WE2 is plotted against time. The sudden increase in Isig at the onset of the spike, due to the charge on the double layer capacitance, is a non-Faradaic process. Therefore, as discussed above, the loss of sensitivity caused by oxygen depletion is reflected in the lowered 1 kHz imaginary impedance, the elevated 0.105 Hz real impedance, and the increased V cntr In addition to rails, they may also be associated with heightened EIS-induced spikes.

[0154] FIG. 29 illustrates another example of sensitivity loss, in this case, which may be considered an extreme version of the Isig dip described above in connection with FIG. 24. Here, sensor current 2910 is observed to be low from the time of insertion, indicating a problematic insertion procedure resulting in electrode blockage. While 1 kHz real impedance 2920 is significantly elevated, both the relatively high frequency phase 2930 and the 1 kHz imaginary impedance 2940 are significantly shifted to more negative values ​​compared to the same parameter values ​​of a properly functioning sensor shown in FIG. 25A. The shifts in the relatively high frequency phase 2930 and the 1 kHz imaginary impedance 2940 indicate that the sensitivity loss may be due to oxygen depletion, which may be caused by blockage of the sensor surface.

[0155] 30A-30D show data from another redundant sensor, where the relative differences in real impedance at 1 kHz, imaginary impedance at 1 kHz, and real impedance at 0.1 Hz between two or more working electrodes can be used to detect sensitivity loss due to biofouling. In this example, WE1 exhibits a greater loss of sensitivity than WE2, as evidenced by the high real impedance at 1 kHz 3010, low imaginary impedance at 1 kHz 3020, and much higher real impedance at 0.105 Hz (3030) for WE2. However, in addition, in this example, V cntr Moreover, as shown in Figure 30D, the spike heights of the raw Isig data do not change significantly over time. This suggests that the sensitivity loss due to biofouling is due to the V cntr -rail is correlated with the increase in spike height. Additionally, the fact that the spike height of the raw Isig data does not change significantly with time indicates that the capacitive component of the circuit does not change significantly with time, and consequently, the sensitivity loss due to biofouling is related to the resistive component of the circuit (i.e., diffusion).

[0156] The various impedance-related parameters described above may be used individually or in combination as inputs to (1) an EIS-based sensor diagnostic procedure and / or (2) a fusion algorithm to generate more reliable sensor glucose values. Regarding the former, FIG. 31 illustrates how EIS-based data, i.e., impedance-related parameters or characteristics, can be used in a diagnostic procedure to determine in real time whether a sensor is operating normally or whether the sensor should be replaced.

[0157] The diagnostic procedure illustrated in the flow diagram of FIG. 31 is based on periodic collection of EIS data, such as, for example, hourly, 30 minutes, 10 minutes, or any other interval, including continuously, as may be appropriate for the particular sensor under analysis. At each such interval, EIS may be performed across the entire frequency spectrum (i.e., a "full sweep"), or across a selected frequency range, or at a single frequency. Thus, for example, for an hourly data collection scheme, EIS may be performed at frequencies ranging from μHz to MHz, or EIS may be performed across a narrower range of frequencies, such as about 0.1 Hz to about 8 kHz, as previously discussed. In various embodiments, EIS data acquisition may be implemented alternating between full sweeps and narrower ranges of the spectrum, or according to other schemes.

[0158] The temporal frequency of EIS implementation and data collection can be dictated by various factors. For example, each EIS implementation consumes a certain amount of power, typically provided by the sensor's battery—i.e., the battery that operates the sensor electronics, including the ASIC, described below. Therefore, battery capacity and remaining sensor life can help determine the number of times EIS is performed and the frequency range sampled for each such execution. Additionally, certain situations may require monitoring an EIS parameter at a particular frequency (e.g., real impedance at 1 kHz) on a first schedule (e.g., once every few seconds or minutes), while other parameters, and / or the same parameters at other frequencies, can be monitored on a second schedule (e.g., less frequently). In these situations, diagnostic procedures can be adapted to specific sensors and requirements, thereby conserving battery power and avoiding unnecessary and / or redundant EIS data acquisition.

[0159] Note that in certain embodiments, a diagnostic procedure such as that shown in FIG. 31 involves a series of separate “tests” performed to perform real-time monitoring of the sensor. Multiple tests or markers, also referred to as “multi-markers,” are implemented because each time EIS is performed (i.e., each time an EIS procedure is performed), data can be gathered regarding multiple impedance-based parameters or characteristics that can be used to detect the status or quality of the sensor, including, for example, whether the sensor has failed or is about to fail. In performing sensor diagnostics, there may be diagnostic tests that may indicate a fault, while there may be other diagnostic(s) that may indicate the absence of a fault. Thus, being able to use multiple impedance-related parameters and implementing multiple test procedures is advantageous because some of the multiple tests can serve as validity checks for some of the other tests. Therefore, real-time monitoring using multi-marker procedures may include a degree of built-in redundancy.

[0160] With the above in mind, the logic of the diagnostic procedure shown in FIG. 31 begins at 3100 by providing EIS data as input after the sensor is inserted / implanted and an EIS run is performed. At 3100, the EIS data is used as input to first determine whether the sensor is still in place. Therefore, if the |Z| slope is found to be constant across the tested frequency band (or range) and / or the phase angle is approximately −90°, the sensor is determined to be out of place, and a warning is sent to the patient / user, for example, indicating that sensor withdrawal has occurred. The specific parameters (and their respective values) described herein for detecting sensor withdrawal are based on the discovery that once the sensor is outside the body and the membrane is no longer hydrated, the impedance spectral response looks just like capacitance.

[0161] If it is determined that the sensor is still in place, the logic proceeds to step 3110 to determine if the sensor has been properly initialized. As shown, the "initial check" is (i) a 1 kHz |(Z n -Z1) / Z1|>30% (Z1 is the real impedance measured at the first time, Z n is the measured impedance at the next interval mentioned above), and (2) whether the change in phase angle is greater than 10° at 0.1 Hz. If the answer to any one of the questions is "yes," the test is satisfactory, i.e., Test 1 is not failed. Otherwise, Test 1 is marked as failed.

[0162] At step 3120, Test 2 asks whether the difference in frequency (f2 - f1) between two consecutive EIS runs at a phase angle of -45° is greater than 10 Hz. Again, a "no" answer is marked as a fail, otherwise Test 2 is satisfactorily met.

[0163] Test 3 in step 3130 is a hydration test, where the query is the current impedance Z nis the impedance Z after initialization at 1kHz pi If so, this test is satisfied, otherwise test 3 is marked as failed. Test 4 in step 3140 is also a hydration test, but this time at a lower frequency. Therefore, this test is performed during sensor operation after initialization, using Z at 0.1 Hz. n is less than 300 kilohms. Again, a "no" answer indicates the sensor failed test 4.

[0164] In step 3150, test 5 asks whether the low frequency Nyquist slope is increasing overall from 0.1 Hz to 1 Hz. As previously mentioned, for a properly operating sensor, the relatively low frequency Nyquist slope should be increasing over time. Therefore, if the answer to the query is "yes," this test is satisfied; if not, the test is marked as failed.

[0165] Step 3160 is the final test in this embodiment of the diagnostic procedure. Here, the question is whether there is an overall decrease in real impedance. Again, as previously discussed, it is assumed that a properly operating sensor should have a decrease in real impedance over time. Therefore, a "yes" answer here means the sensor is operating properly; if not, the sensor has failed test 6.

[0166] Once all six tests have been implemented, a determination is made at 3170 as to whether the sensor is operating normally or whether the sensor has failed. In this embodiment, the sensor is determined to be functioning normally if it passes at least three of the six tests (3172). Stated differently, the sensor must fail at least four of the six tests to be deemed to have failed (3174). In alternative embodiments, different rules may be used to evaluate normal operation versus sensor failure. Additionally, in some embodiments, each test may be weighted, with the assigned weight reflecting, for example, the importance of that test in determining overall sensor operation (normal vs. failed) or the particular parameter(s) questioned for that test. For example, one test may be weighted twice as much as another test, while a third test may be weighted only half as much, etc.

[0167] Other alternative embodiments may use a different number of tests and / or a different set of EIS-based parameters for each test. FIGS. 32A and 32B illustrate an example of a real-time monitoring diagnostic procedure involving seven tests. Referring to FIG. 32A, the logic begins at 3200 by providing EIS data as input after a sensor is inserted / implanted and an EIS procedure is performed. At 3200, the EIS data is used as input to first determine whether the sensor is still in place. Therefore, if the |Z| slope is found to be constant across the tested frequency band (or range) and / or the phase angle is approximately −90°, it is determined that the sensor is no longer in place, and a warning is sent to the patient / user, for example, indicating that sensor withdrawal has occurred. On the other hand, if the sensor is determined to be in place, the logic transitions to start (3202) the diagnostic test.

[0168] In 3205, Test 1 is similar to Test 1 in the diagnostic procedure described above in connection with FIG. 31, except that this Test 1 specifies that the subsequent measurement, Zn, be performed two hours after the initial measurement. Thus, in this example, Zn = Z2hr. More specifically, Test 1 compares the real impedance two hours after initialization (sensor implantation and initialization) to the value before initialization. Similarly, the second part of Test 1 asks whether the difference between the phase two hours after initialization and the phase before initialization is greater than 10° at 0.1 Hz. As before, if the answer to any one of the queries is affirmative, the sensor is determined to be successfully hydrated and initialized, and Test 1 is satisfied; otherwise, the sensor fails this test. It should be noted that even though this test queries regarding changes in impedance and phase two hours after initialization, the time interval between two consecutive EIS runs may be shorter or longer depending on a variety of factors, including sensor design, level of electrode redundancy, the degree to which redundant testing is included in the diagnostic procedure, battery power source, etc.

[0169] Moving to 3210, the logic then performs a sensitivity loss check by querying whether the rate of change of the impedance magnitude and the rate of change of Isig at 1 kHz are greater than 30% after a two time interval (n+2). If the answer to both queries is yes, then the sensor is determined to have lost sensitivity and fails Test 2. Note that although Test 2 is illustrated herein based on a preferred rate of difference of 30%, in other embodiments the rate of difference of the impedance magnitude and the rate of difference of Isig at 1 kHz can be in the range of 10% to 50% for purposes of performing this test.

[0170] Test 3 (at 3220) is similar to test 5 of the algorithm illustrated in FIG. 31. Here, as before, the question is whether the low-frequency Nyquist slope is increasing overall from 0.1 Hz to 1 Hz. If so, the test passes; if not, the test fails. As shown in 3220, this test is also subject to the setting of a threshold or tolerance for the rate of change of the low-frequency Nyquist slope, beyond which the sensor may be deemed to have failed, or at least may trigger further diagnostic testing. In embodiments of the present invention, such a threshold / tolerance for the rate of change of the low-frequency Nyquist slope may fall within a range of about 2% to about 20%. In some preferred embodiments, the threshold may be about 5%.

[0171] The logic then transitions to 3230, which is another low-frequency test, this time involving phase and impedance magnitude. More specifically, the phase test queries whether the phase at 0.1 Hz is continuously increasing over time. If so, the test fails. Like other tests where a parameter trend is monitored, the low-frequency phase test of test 4 also follows a threshold or tolerance setting for the rate of change of low-frequency phase, beyond which the sensor may be deemed to have failed or at least raised concern. In some preferred embodiments, such a threshold / tolerance for the rate of change of low-frequency phase may fall within a range of about 5% to about 30%. In some preferred embodiments, the threshold may be about 10%.

[0172] As mentioned above, Test 4 also includes a low-frequency impedance magnitude test, in which the inquiry is whether the impedance magnitude at 0.1 Hz is continuously increasing over time. If so, the test fails. Note that if either the phase test or the impedance magnitude test fails, Test 4 is considered a "fail." The low-frequency impedance magnitude test of Test 4 also sets a threshold or tolerance for the rate of change of the low-frequency impedance magnitude, beyond which the sensor may be considered to have failed or at least raise concerns. In some preferred embodiments, such a threshold / tolerance for the rate of change of the low-frequency impedance magnitude may fall within a range of about 5% to about 30%. In some preferred embodiments, the threshold may be about 10%, where the impedance magnitude range for typical sensors is generally about 100 kilohms to about 200 kilohms.

[0173] Test 5 (at 3240) is another sensitivity loss check that can be considered a supplement to Test 2. Here, if both the rate of change of Isig and the rate of change of impedance magnitude at 1 kHz are greater than 30%, the sensor is determined to have recovered from sensitivity loss. That is, the sensor is determined to have previously suffered a sensitivity loss, even if for some reason the sensitivity loss was not detected by Test 2. As with Test 2, Test 5 is illustrated based on a preferred difference rate of 30%, although in other embodiments the difference rate between Isig and impedance magnitude at 1 kHz may fall within the range of 10% to 50% for purposes of performing this test.

[0174] Transitioning to 3250, Test 6 provides a sensor functionality test using specific failure criteria determined based on observed data and the particular sensor design. Specifically, in one embodiment, if at least two of the following three criteria are met, the sensor may be determined to be faulty and therefore not responsive to glucose. Therefore, Test 6 is determined to pass if any two of (1) through (3) are not met. Note that in other embodiments, an Isig log of this test may fail if Isig is less than about 5 nA to about 20 nA. Similarly, a second log may fail if imaginary impedance at 1 kHz is less than about -1000 ohms to about -2000 ohms. Finally, the phase projection may be rejected if the phase at 1 kHz is less than about -10° to about -20°.

[0175] Finally, step 3260 provides another sensitivity test, where the parameters are evaluated at low frequencies. Thus, test 7, at 0.1 Hz, asks whether the magnitude of the difference between the ratio of imaginary impedance to Isig(n+2), on the one hand, and the previous value of this ratio, on the other hand, is greater than 30% of the magnitude of the previous value of this ratio. If so, the test fails; otherwise, the test passes. Here, test 7 is illustrated based on a preferred difference ratio of 30%, but in other embodiments, the difference ratio may fall within the range of 10% to 50% for purposes of performing this test.

[0176] Once all seven tests have been implemented, a determination is made at 3270 as to whether the sensor is operating normally or whether to issue a warning indicating that the sensor has failed (or may be about to fail). As shown, in this embodiment, if at least four of the seven tests pass, the sensor is determined to be functioning normally (3272). Stated differently, a sensor must fail at least four of the seven tests to be determined to have failed, or at least raise a concern (3274). If the sensor is determined to be "bad" (3274), a warning to that effect may be issued, for example, to the patient / user. As noted above, alternative embodiments may use different rules to evaluate normal operation versus sensor failure / concern. Additionally, in some embodiments, each test may be weighted, with the assigned weight reflecting, for example, the importance of that test or the particular parameter(s) questioned for that test in determining overall sensor operation (normal vs. failing).

[0177] As previously mentioned, in the embodiments of the invention described herein, the various impedance-related parameters discussed above may be used individually or in combination as inputs to one or more fusion algorithms that generate more reliable sensor glucose values. Specifically, unlike single-sensor (i.e., single working electrode) systems, multiple sensing electrodes are known to provide more reliable glucose readings because multiple signals obtained from two or more working electrodes can be fused to provide a single sensor glucose value. Such signal fusion utilizes quantitative inputs provided by EIS to calculate the most reliable output sensor glucose value from redundant working electrodes. While the following discussion may describe various fusion algorithms with respect to a first working electrode (WE1) and a second working electrode (WE2) as redundant electrodes, it should be noted that this is by way of example and not limitation, as the algorithms described herein and their underlying principles are applicable to and may be used in redundant sensor systems having three or more working electrodes. Additionally, redundant electrodes can be included on (identical) sensors on / in a single flex or multiple flexes, or redundant electrodes can be included on non-identical sensors on / in a single flex or multiple flexes (e.g., a complex redundant sensor system having two or more sensors, where at least two of the sensors have different designs from each other).

[0178] FIG. 33A and FIG. 33B show top-level flowcharts of two alternative methodologies, each including a fusion algorithm. Specifically, FIG. 33A is a flowchart involving the current (Isig)-based fusion algorithm, while FIG. 33B is a flowchart directed to sensor glucose (SG) fusion. As can be seen, the primary difference between the two methodologies is the time of calibration. Thus, FIG. 33A shows that for Isig fusion, calibration 3590 is performed after fusion 3540 is completed. That is, redundant Isigs from WEI through WEn are fused into a single Isig 3589, which is then calibrated to generate a single sensor glucose value 3598. On the other hand, for SG fusion, calibration 3435 is completed for each individual Isig from WEI through WEn, generating calibrated SG values ​​(e.g., 3436, 3438) for each of the working electrodes. Therefore, the SG fusion algorithm provides independent calibration of each of multiple Isigs, which may be preferred in some embodiments of the invention described herein. Once calibrated, the multiple calibrated SG values ​​are merged into a single SG value 3498.

[0179] It is important to note that each of the flowcharts shown in Figures 33A and 33B includes a spike filtering process (3520, 3420). As previously discussed in the discussion regarding sensitivity loss, frequency impedance measurements above 1 kHz typically cause EIS-induced spikes in the Isig signal. Therefore, for both SG fusion and Isig fusion, once the EIS procedure is performed for each of electrodes WEI through WEn, it is desirable to first filter Isigs 3410, 3412, etc., and 3510, 3512, etc., to obtain respective filtered Isigs 3422, 3424, etc., and 3522, 3524, etc. The filtered Isigs are then used in Isig fusion, as described below, or are first calibrated and then used in SG fusion. As will become apparent in the following discussion, both fusion algorithms involve the calculation and assignment of weights based on various factors.

[0180] FIG. 34 shows details of the fusion algorithm 3440 for SG fusion. Essentially, there are four elements that need to be tested before determining the fusion weights. First, an integrity check 3450 involves determining whether each of the following parameters is within a specified range (e.g., predetermined lower and upper thresholds) for normal sensor operation: (i) Isig, (ii) 1 kHz real and imaginary impedance, (iii) 0.105 Hz real and imaginary impedance, and (iv) Nyquist slope. As shown, the integrity check 3450 includes a boundary check 3452 and a noise check 3456, and for each of the tests, the above-mentioned parameters are used as input parameters. Note that for simplicity, the real and / or imaginary impedance at one or more frequencies is simply represented as "Imp" in FIGS. 33A-35. Additionally, the magnitude and phase of the impedance may be used (this is shown as an input in Figures 33A and 33B) to calculate both real and imaginary impedances.

[0181] The output from each of the boundary check 3452 and noise check 3458 is a respective reliability index (RI) for each of the redundant working electrodes. Thus, outputs from the boundary check include, for example, RI_bound_We1 (3453) and RI_bound_We2 (3454). Similarly, for the noise check, outputs include, for example, RI_noise_We1 (3457) and RI_noise_We2 (3458). The boundary and noise reliability indexes for each working electrode are calculated based on compliance with the above-mentioned ranges for normal sensor operation. Thus, if any of the parameters are outside the specified range for a particular electrode, the reliability index for that particular electrode is reduced.

[0182] It should be noted that the thresholds or ranges for the above-mentioned parameters may depend on various factors, including the particular sensor and / or electrode design. Meanwhile, in a preferred embodiment, typical ranges for some of the above-mentioned parameters may be, for example, the following: boundary threshold for real impedance at 1 kHz = [0.3e+4, 2e+4]; boundary threshold for imaginary impedance at 1 kHz = [-2e+3, 0]; boundary threshold for real impedance at 0.105 Hz = [2e+4, 7e+4]; boundary threshold for imaginary impedance at 0.105 Hz = [-2e+5, -0.25e+5]; and boundary threshold for Nyquist slope = [2, 5]. The noise may be calculated, for example, using a second-order central difference method, where noise is considered to be outside the noise boundary if it exceeds a certain percentage (e.g., 30%) of the median of each variable buffer.

[0183] Second, sensor dips may be detected using the sensor current (Isig) and 1 kHz real impedance. Therefore, as shown in FIG. 34, Isig and “Imp” are used as inputs for dip detection 3460. Here, the first step is to determine whether there is divergence between Isigs and whether such divergence is reflected in the 1 kHz real impedance data. This may be accomplished by using a mapping 3465 between an Isig similarity index (RI_sim_Isig12) 3463 and a 1 kHz real impedance similarity index (RI_sim_imp12) 3464. This mapping helps avoid false positives when the dip is not real, so that if the significant Isig divergence is real, the algorithm will select the sensor with the higher Isig.

[0184] According to one embodiment, the divergence / convergence of two signals (eg, two Isig, or two 1 kHz real impedance data points) can be calculated as follows: diff_va1=abs(va1-(va1+va2) / 2) diff_va2=abs(va2-(va1+va2) / 2) RI_sim=1-(diff_va1+diff_va2) / (mean(abs(va1+va2)) / 4) where va1 and va2 are two variables, and RI_sim (similarity index) is an index for measuring the convergence or divergence of signals. In this embodiment, RI_sim must be bounded between 0 and 1. Therefore, if RI_sim calculated above is less than 0, RI_sim will be set to 0, and if RI_sim is greater than 1, RI_sim will be set to 1.

[0185] The mapping 3465 is performed using ordinary linear regression (OLR). However, if OLR does not perform well, robust median slope linear regression (RMSLR) can be used. For example, for an Isig similarity index and a 1 kHz real impedance similarity, two mapping procedures are required: (i) mapping the Isig similarity to a 1 kHz real impedance similarity index, and (ii) mapping the 1 kHz real impedance similarity index to an Isig similarity index. Both mapping procedures produce two residues, res12 and res21. Then, the dip reliability indices 3467, 3468 are each calculated as: It can be calculated as RI_dip=1-(res12+res21) / (RI_sim_Isig+RI_sim_1K_real_impedance).

[0186] The third factor is sensitivity loss 3470, which can be detected using the 1 kHz imaginary impedance trend over, for example, the past 8 hours. If one sensor's trend becomes negative, the algorithm will rely on the other sensor. If both sensors lose sensitivity, a simple average is taken. By using a strong low-pass filter to smooth the 1 kHz imaginary impedance, which tends to be noisy, and by using a correlation coefficient or linear regression over time over, for example, the past 8 hours, a trend can be calculated to determine whether the correlation coefficient is negative or whether the slope is negative. Each of the sensitivity loss reliability indices 3473, 3474 is then assigned a binary value of 1 or 0.

[0187] The overall reliability index (RI) for each of we1, we2, ... wen is calculated as follows: RI_we1=RI_dip_we1×RI_sensitivity_loss_we1×RI_bound_we1×RI_noise_we1 RI_we2=RI_dip_we2×RI_sensitivity_loss_we2×RI_bound_we2×RI_noise_we2 RI_we3=RI_dip_we3×RI_sensitivity_loss_we3×RI_bound_we3×RI_noise_we3 RI_we4=RI_dip_we4×RI_sensitivity_loss_we4×RI_bound_we4×RI_noise_we4 · · · RI_we n =RI_dip_we n ×RI_sensitivity_loss_we n ×RI_bound_we n ×RI_noise_we n

[0188] To calculate the reliability index for each individual working electrode, the weight of each of the electrodes is: weight_we1=RI_we1 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n ) weight_we2=RI_we2 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n ) weight_we3=RI_we3 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n ) weight_we4=RI_we4 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n ) · · · weight_we n =RI_we n / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n ) can be calculated as follows:

[0189] Based on the above, Fusion SG3498: SG=weight_we1×SG_we1+weight_we2×SG_we2+weight_we3×SG_we3+ weight_we4×SG_we4+...+weight_we n ×SG_we n It is calculated as follows:

[0190] The last factor relates to artifacts in the final sensor readings, such as those caused by momentary weight changes in the sensor fusion. This can be avoided by applying a low-pass filter 3480 to smooth the RI of each electrode, or by applying a low-pass filter to the final SG. If the former is used, the filtered reliability indices, e.g., RI_We1* and RI_We2*(3482, 3484), are used in calculating the weights for each electrode and thus the fused SG 3498.

[0191] Figure 35 shows details of the fusion algorithm 3540 for Isig fusion. As can be seen, this algorithm is substantially similar to that shown in Figure 34 for SG fusion, with two exceptions. First, as noted above, for Isig fusion, calibration constitutes the final step in the process, where a single fused Isig 3589 is calibrated to produce a single sensor glucose value 3598. See also Figure 33B. Second, while SG fusion uses SG values ​​from multiple electrodes to calculate the final SG value 3498, the fused Isig value 3589 is calculated using filtered Isigs (3522, 3524, etc.) from multiple electrodes.

[0192] One closed-loop study with a nondiabetic population found that the fusion algorithm described above significantly improved both mean absolute relative differences (MARDs) on day 1, when the low start-up issue is most pronounced, potentially impacting sensor accuracy and reliability overall (i.e., over the sensor's 7-day lifetime). The study evaluated data from an 88% variance layout design with high current density (nominal) plating using three different methodologies: (1) calculating a single sensor glucose value (SG) by fusion using Medtronic Minimed's Ferrari algorithm 1.0 (the SG fusion algorithm discussed above); (2) calculating a single SG by identifying a better ISIG value using 1 kHz EIS data (the ISIG fusion algorithm discussed above); and (3) calculating a single SG by using a higher ISIG value (i.e., without EIS). Details of the study's data are presented below.

[0193] (1) SG based on Ferrari 1.0 algorithm with 88% dispersion layout with high current density (nominal) plating [Table 1] [Table 2] [Table 3] [Table 4]

[0194] (2) Better ISIG-based SG by using 1kHz EIS in an 88% dispersion layout with high current density (nominal) plating [Table 5] [Table 6] [Table 7] [Table 8]

[0195] (3) SG based on higher ISIG for 88% dispersion layout with high current density (nominal) plating [Table 9] [Table 10] [Table 11] [Table 12]

[0196] Using the above data, the first approach yielded a Day 1 MARD (%) of 19.52% and an overall MARD of 12.28%. The second approach yielded a Day 1 MARD of 15.96% and an overall MARD of 11.83%. Finally, the third approach yielded a Day 1 MARD of 17.44% and an overall MARD of 12.26%. Therefore, for this design with redundant electrodes, the better IS-based SG calculation using 1 kHz EIS (i.e., the second methodology) appears to offer the greatest benefit. Specifically, the lower Day 1 MARD may be due to better low activation detection using EIS, for example. Additionally, the overall MARD percentage is over 1% lower than the overall average MARD of 13.5% for WE1 and WE2 in this study. It should be noted that in the above approach, data transitions may be processed by a filtering method that minimizes the severity of the transitions, such as by using the low-pass filter 3480 discussed above in connection with Figures 33A-35.

[0197] It is worth reiterating that sensor diagnostics, including evaluation of low actuation, sensitivity loss, and signal dip events, depend on a variety of factors, including sensor design, the number of electrodes (i.e., redundancy), and electrode distribution / configuration. Therefore, the actual frequency or frequency range at which EIS-based parameters may be substantially glucose-independent, and thus an independent marker or predictor of one or more of the aforementioned failure modes, may also depend on the particular sensor design. For example, as described above, it has been found that sensitivity loss can be predicted using imaginary impedance at relatively high frequencies where the imaginary impedance is substantially glucose-independent, but the level of glucose dependence, and thus the particular frequency range for using imaginary impedance as a marker of sensitivity loss, may shift (higher or lower) depending on the actual sensor design.

[0198] More specifically, as sensor designs increasingly move toward using redundant working electrodes, the latter must be of increasingly smaller size to maintain the overall size of the sensor. The size of the electrode impacts the frequency at which it can be queried for a particular diagnosis. In this regard, it is important to note that the fusion algorithms described herein and shown in Figures 33A-35 are intended to be exemplary rather than limiting, as each algorithm can be modified as needed to use EIS-based parameters at frequencies that exhibit the least amount of glucose dependence based on the type of sensor being analyzed.

[0199] In addition, experimental data indicate that human tissue structure can also affect glucose dependence at various frequencies. For example, in children, real impedance at 0.105 Hz has been found to be a substantially glucose-independent indicator of low-start detection. This is believed to result from changes in children's tissue structure, e.g., Warburg impedance, which is associated with a mostly resistive component. See also the following discussion on the detection of interferents.

[0200] The embodiments of the invention described herein are also directed to the use of EIS in optimizing sensor calibration. By way of background, current methodologies require that the slope of a BG-body Isig plot, which may be used to calibrate subsequent Isig values, be determined by:

number

[0201] However, there are situations where the linear relationship described above does not hold, such as during periods when the sensor is experiencing a transition. As shown in Figure 37, it is clear that Isig-BG pairs 1 and 2 are significantly different from pairs 3 and 4 in terms of their Isig and BG relationships. In these types of conditions, using a constant offset tends to produce inaccurate results.

[0202] To address this issue, one embodiment is directed to the use of EIS-based dynamic offset, where the EIS measurement is: V={real_imp_1K, img_imp_1K, Nyquist_slope, Nyquist_R_square} The dynamic offset method is used to define a sensor status vector such as (V2-V3|), where all elements of the vector are substantially BG-independent. Nyquist_R_square is the R-squared of the linear regression used to calculate the Nyquist slope, i.e., the squared correlation coefficient between real and imaginary impedances at relatively low frequencies. Note that a low R-squared indicates abnormal sensor performance. A status vector is assigned to each Isig-BG pair. If a significant difference in the status vectors is detected, e.g., |V2-V3| in the example shown in Figure 37, different offset values ​​are assigned to 3 and 4 compared to 1 and 2. Therefore, by using this dynamic offset method, it is possible to maintain a linear relationship between Isig and BG.

[0203] In a second embodiment, an EIS-based segmentation approach may be used for calibration. Using the example of FIG. 37 and vector V, it can be determined that the sensor state between 1 and 2 is significantly different from the sensor state between 3 and 4. Therefore, the calibration buffer can be divided into two segments as follows: Isig_buffer1=[Isig1, Isig2], BG_buffer1=[BG1, BG2] Isig_buffer2=[Isig3, Isig4], BG_buffer2=[BG3, BG4] Therefore, if the sensor operates between 1 and 2, Isig_buffer1 and BG_buffer1 will be used for calibration. However, if the sensor operates between 3 and 4, i.e., during the transition period, Isig_buffer2 and BG_buffer2 will be used for calibration.

[0204] In yet another embodiment, an EIS-based dynamic gradient technique, in which EIS is used to adjust the gradient, may be used for calibration purposes. FIG. 38A shows an example of how this method can be used to improve sensor accuracy. In this illustration, data points 1-4 are discrete blood glucose levels. As can be seen in FIG. 38A, there is a sensor dip 3810 between data points 1 and 3, which can be detected using the sensor state vector V described above. During the dip, the gradient can be adjusted upward to reduce the under-indication, as indicated by reference numeral 3820 in FIG. 38A.

[0205] In further embodiments, EIS diagnostics may be used to determine the timing of sensor calibration, which can be very useful, for example, for low start-up events, loss-of-sensitivity events, and other similar situations. As is known, most current methodologies require periodic calibration based on a pre-set schedule, such as four times a day. However, with EIS diagnostics, calibration can be event-driven, so that it can be performed as frequently as needed and at the most productive times. Again, the status vector V may be used to determine when the sensor state has changed, and if so, to request calibration.

[0206] More specifically, in an illustrative example, FIG. 38B shows a flowchart for EIS-assisted sensor calibration with low start-up detection. Using Nyquist slope, 1 kHz real impedance, and bounds check 3850 (see, e.g., the aforementioned bounds checks and associated thresholds for EIS-based parameters related to the fusion algorithms of FIGS. 33A-35), a confidence index 3853 can be developed for start-up such that if 1 kHz real impedance 3851 and Nyquist slope 3852 are lower than their corresponding upper limits, then RI_startup=1 and the sensor is ready for calibration. That is, confidence index 3853 is “high” (3854), and the logic can proceed to calibration at 3860.

[0207] On the other hand, if the 1 kHz real impedance and Nyquist slope are higher than the corresponding upper limits (or thresholds), then RI_startup=0 (i.e., “low”) and the sensor is not ready for calibration (3856), i.e., there may be a low startup issue. Here, the trends in the 1 kHz real impedance and Nyquist slope can be used to predict when both parameters will be within range (3870). If this is estimated to take only a very short amount of time (e.g., less than an hour), the algorithm waits until the sensor is ready, i.e., until the above-mentioned EIS-based parameters are within bounds (3874), at which point the algorithm proceeds to calibration. However, if the wait time is relatively long (3876), the sensor can be calibrated now, and the slope or offset can be gradually adjusted according to the trends in the 1 kHz real impedance and Nyquist slope (3880). Note that performing the adjustments can avoid significant over- or under-indications caused by low startup. As noted above, the EIS-based parameters and associated information used in the instant calibration algorithm are substantially glucose-independent.

[0208] Note that while the above description in connection with FIG. 38B illustrates a single working electrode and the calculation of a reliability index for that working electrode's activation, this is by way of example and not limitation. Therefore, in a redundant sensor including two or more working electrodes, a boundary test can be performed and an activation reliability index can be calculated for each of the multiple (redundant) working electrodes. Based on the respective reliability indices, at least one working electrode that can proceed to acquire a glucose measurement can then be identified. That is, in a sensor with a single working electrode, if the latter exhibits low activation, actual use of the sensor (to measure glucose) may need to be delayed until the low activation period has ended. This period can typically be on the order of an hour or more, which is clearly disadvantageous. In contrast, in a redundant sensor, utilizing the methods described herein enables adaptive, or "smart," activation, allowing the identification of an electrode that can proceed to data collection in fairly short order, for example, on the order of minutes. This, in turn, reduces MARD, since low activation typically increases MARD by about 1 / 2%.

[0209] In yet another embodiment, EIS can assist in adjusting the calibration buffer. For existing calibration algorithms, the buffer size is always 4, i.e., 4 Isig-BG pairs, and the weights are based on α, which is an exponential function of the time constant, and β, which is a function of blood glucose fluctuations, as described above. Here, EIS can help determine when to flush the buffer, how to adjust the buffer weights, and what the appropriate buffer size is.

[0210] In some embodiments, EIS may be used for interferent detection. Specifically, it may be desirable to provide a drug infusion set that includes a combination of a sensor and a drug infusion catheter, with the sensor being disposed within the infusion catheter. In such systems, the physical positioning of the infusion catheter relative to the sensor may be somewhat important, primarily due to potential effects (i.e., interference) on the sensor signal that may be caused by the drug being infused and / or its inactive components.

[0211] For example, diluents used with insulin contain m-cresol as a preservative. In vitro studies have shown that m-cresol adversely affects glucose sensors when insulin (and therefore m-cresol) is injected in close proximity to the sensor. Therefore, a system combining a sensor and an infusion catheter in a single needle must be able to detect and adjust for the effect of m-cresol on the sensor signal. Because m-cresol affects the sensor signal, it would be desirable to have a means of detecting this interferent independently of the sensor signal itself.

[0212] Experiments have shown that the effect of m-cresol on the sensor signal is temporary and therefore reversible. On the other hand, if an insulin injection is administered too close to the sensor, m-cresol tends to "poison" the electrode(s), resulting in the latter being unable to detect glucose until insulin (and m-cresol) is absorbed into the patient's tissues. In this regard, we have found that there is typically a period of approximately 40 minutes between the start of insulin injection and the time when the sensor regains the ability to detect glucose again. However, advantageously, we have also found that during this same period, the magnitude of the 1 kHz impedance increases significantly, completely independent of the glucose concentration.

[0213] Specifically, Figure 39 shows Isig and impedance data from an in vivo experiment in which the sensor was placed in a 100 mg / dL glucose solution and impedance was measured at 1 kHz every 10 minutes, as indicated by the circled data points 3920. Cresol was then added to bring the solution to 0.35% m-cresol (3930). As can be seen, when m-cresol was added, Isig 3940 first increased dramatically and then began to drift. The glucose concentration in the solution was then doubled by adding 100 mg / dL glucose, however, this had no effect on Isig 3940 because the electrode was unable to detect glucose.

[0214] On the other hand, m-cresol dramatically affected both the magnitude and phase of the impedance. Figure 40A shows the Bode plot of the phase, and Figure 40B shows the Bode plot of the impedance magnitude, both before and after the addition of m-cresol. As can be seen, after m-cresol was added, the impedance magnitude 4010 increased by at least one order of magnitude across the frequency spectrum from its post-initialization value 4020. At the same time, the phase 4030 completely changed compared to its post-initialization value 4040. Regarding the Nyquist plot in Figure 40C, here, the pre-initialization curve 4050 and post-initialization curve 4060 appear as expected for a properly functioning sensor. However, after the addition of m-cresol, curve 4070 becomes dramatically different.

[0215] The above experiment identifies an important practical pitfall of continuing to rely on the Isig signal after m-cresol has been added. Referring again to Figure 39, a patient / user monitoring the sensor signal may be under the mistaken impression that their glucose level has just spiked and that a bolus should be administered. The user then administers the bolus, and at the bolus, the Isig signal has already begun to drift. To the patient / user, everything may appear normal. However, what has actually happened is that, depending on the patient's glucose level before the bolus, the patient has simply received an unnecessary dose of insulin, putting the patient at risk of experiencing a hypoglycemic event. This scenario reinforces the desirability of a means of detecting interferents that is as glucose-independent as possible.

[0216] Figure 41 shows another experiment in which the sensor was initialized with a 100 mg / dL glucose solution, then the glucose was increased to 400 mg / dL for 1 hour, then returned to 100 mg / dL. m-cresol was then added to increase the concentration to 0.35%, and the sensor was held in this solution for 20 minutes. Finally, the sensor was placed in a 100 mg / dL glucose solution to allow the Isig to recover after exposure to m-cresol. As can be seen, after initialization, the impedance magnitude 4110 at 1 kHz was approximately 2 kilohms. Addition of m-cresol caused a sharp increase in the Isig 4120, as well as the impedance magnitude 4110. Furthermore, when the sensor was placed back in the 100 mg / dL glucose solution, the impedance magnitude 4110 also returned to near normal levels.

[0217] As can be seen from the above experiment, EIS can be used to detect the presence of an interferent, in this case m-cresol. Specifically, because interferents affect the sensor in a way that increases the magnitude of impedance across the entire frequency spectrum, the magnitude of impedance can be used to detect interference. When interference is detected, the sensor's operating voltage can be changed to a point where the interferent is not measured, or the sensor can temporarily suspend data reporting, indicating to the patient / user that the sensor is unable to report data due to the administration of medication (until the measured impedance returns to pre-infusion levels). Note that because the effect of the interferent is due to preservatives contained in the insulin, the magnitude of impedance will exhibit the same behavior described above regardless of whether the insulin being injected is fast-acting or slow-acting.

[0218] Importantly, as noted above, the impedance magnitude, and indeed the magnitude at 1 kHz, is substantially glucose-independent. Referring to Figure 41, it can be seen that a four-fold increase in glucose concentration from 100 mg / dL to 400 mg / dL results in an increase in the 1 kHz impedance magnitude from approximately 2000 ohms to approximately 2200 ohms, or an increase of approximately 10%. In other words, the effect of glucose on the impedance magnitude measurement appears to be approximately one order of magnitude smaller than the measured impedance. Because the "signal-to-noise" ratio at this level is typically small enough to eliminate noise (i.e., the glucose effect), the resulting impedance magnitude is substantially glucose-independent. Additionally, it should be emphasized that the impedance magnitude exhibits an even greater degree of glucose independence in actual human tissue compared to the buffer solution used in the in vivo experiments described above.

[0219] The embodiments of the invention described herein are also directed to analog front-end integrated circuits (AFE ICs), which are custom application-specific integrated circuits (ASICs) that provide the analog electronics necessary to, among other things, (i) support interfacing with multiple potentiostats and multi-terminal glucose sensors based on either oxygen or peroxide, (ii) interface with a microcontroller to form a micropowered sensor system, and (iii) implement EIS diagnostics, fusion algorithms, and other EIS-based processes based on measurements of EIS-based parameters. More specifically, the ASIC incorporates diagnostic capabilities to measure real and imaginary impedance parameters of the sensor(s) over a wide range of frequencies, as well as digital interface circuitry that enables bidirectional communication with a microprocessor chip. Additionally, the ASIC includes power control circuitry, a real-time clock, and a crystal oscillator that enable operation at very low standby and operating power, allowing the external microprocessor to be powered off.

[0220] Figures 42A and 42B show a block diagram of the ASIC, and Table 1 below provides a description of the pad signals (shown on the left side of Figures 42A and 42B), with some signals multiplexed onto a single pad. [Table 13]

[0221] The ASIC will now be described with reference to Figures 42A and 42B and Table 1.

[0222] Power Plane The ASIC has one power plane powered by a power supply pad, VBAT (4210), which has an operating input range of 2.0 volts to 4.5 volts. This power plane has a regulator that drops the voltage of some of the circuits within this plane. The power supply is called VDDBU (4212) and has an output pad for test and bypass. The VBAT power supply circuitry includes an RC oscillator, a real-time clock (RC osc) 4214, battery protection circuitry, regulator control, a power-on reset circuit (POR), and various inputs and outputs. The pad on the VBAT power plane is configured to draw less than 75 nA at 40°C and VBAT=3.50V.

[0223] The ASIC also has a VDD power supply to power the logic. The VDD power supply voltage range is programmable from at least 1.6 volts to 2.4 volts. The circuitry on the VDD power plane includes most of the digital logic, timers (32 kHz), and a real-time clock (32 kHz). The VDD power plane includes level shifters to interface with other voltage planes as needed. The level shifters have regulated interfaces so that the current on a powered power plane does not increase by more than 10 nA when another power plane is unpowered.

[0224] The ASIC includes an on-board regulator (with shutdown control) and an option for an external VDD source. The regulator input is a separate pad, REG_VDD_IN (4216), which has electrostatic discharge (ESD) protection in common with the other I / O on VBAT. The on-board regulator has an output pad, REG_VDD_OUT (4217). The ASIC also has an input pad for VDD that is separate from the REG_VDD_OUT pad.

[0225] The ASIC is powered by either the VDD on-board regulator or an external source, and includes an analog power plane called VDDA (4218), which is typically powered by a filtered VDD. Circuits powered by VDDA are configured to operate within 0.1 volts of VDD, thus eliminating the need for level shifting between the VDDA and VDD power planes. The VDDA-powered source powers the sensor analog circuitry, analog measurement circuitry, as well as any other noise-sensitive circuitry.

[0226] The ASIC includes a pad-powered VPAD for dedicated digital interface signals. The pad power has an operating voltage range of at least 1.8V to 3.3V. These pads have separate power pad(s) and are powered from an external source. The pads also incorporate level shifters to other on-board circuitry to allow a flexible pad power supply range independent of the VDD logic power supply voltage. The ASIC can adjust the VPAD pad ring signal so that other power supply currents do not increase by more than 10nA when the VPAD power supply is not enabled.

[0227] Bias Generator The ASIC has a bias generator circuit, BIAS_GEN (4220), which is powered from the VBAT supply and generates a stable bias current at the system supply voltage. The output current has the following specifications: (i) power supply sensitivity of less than ±2.5% from a supply voltage of 1.6V to 4.5V, and (ii) current accuracy of less than ±3% after trimming.

[0228] The BIAS_GEN circuit generates switched and non-switched output currents to power circuits that require bias current to operate. The operating current drain of the BIAS_GEN circuit is less than 0.3uA at 25°C with VBAT between 2.5V and 4.5V (excluding bias output current). Finally, the temperature coefficient of the bias current is typically between 4,000ppm / °C and 6,000ppm / °C.

[0229] Voltage Reference The ASIC described herein is configured with a low power supply reference voltage that is powered from the VBAT power supply. The voltage reference has an enable input that can accept a signal from logic powered by VBAT or VDDBU. The ASIC is designed so that when powered by VBAT, the enable signal does not cause a current increase from this signal interface of more than 10 nA from any power supply.

[0230] The reference voltage has the following specifications: (i) output voltage: 1.220 ±3 mV after trimming; (ii) power supply sensitivity: less than ±6 mV for inputs from 1.6 V to 4.5 V; (iii) temperature sensitivity: less than ±5 mV from 0°C to 60°C; and (iv) default accuracy of output voltage (untrimmed): 1.220 V ±50 mV. Additionally, the supply current is less than 800 nA at 4.5 V and 40°C. In this embodiment, the reference output is forced to VSSA when the reference is disabled to prevent the VDD voltage regulator from overshooting to levels above the logic breakdown voltage.

[0231] 32kHz oscillator The ASIC includes a low-power 32.768 kHz crystal oscillator 4222, powered by power drawn from the VDDA supply and with software-trimmable capacitance on the crystal oscillator pads (XTALI, XTALO). Specifically, the frequency trimming range is at least -50 ppm to +100 ppm, with a maximum step size of 2 ppm across the entire trimming range. The crystal may have a load capacitance of 7 pF, Ls = 6.9512 kH, Cs = 3.3952 fF, Rs = 70 kH, shunt capacitance = 1 pF, and 2 pF of PC board parasitic capacitance on each crystal terminal.

[0232] The ASIC has a VPAD level output CLK_32kHz available on a pad, in which case the output can be disabled under software and logic control. By default, the output drives a 32kHz oscillator. An input pin OSC32K_BYPASS (4224) allows the 32kHz oscillator to be disabled (without power consumption) and allows digital input to the XTALI pad. The circuitry associated with this function is configured so that no state of the OSC32K_BYPASS signal other than the oscillator current adds more than 10nA to the ASIC current when OSC32K_BYPASS is low.

[0233] The 32kHz oscillator is required to be running at all times when the VDDA plane is powered, except in a bypass condition. When OSC32K_BYPASS is true, the 32kHz oscillator analog circuitry is in a low-power state and the XTALI pad is configured to accept digital inputs whose levels are between 0 and VDDA. Note that the 32kHz oscillator output has a duty cycle of 40% to 60%.

[0234] Timer The ASIC includes a timer 4226 that runs on a divide-by-two clock from the 32 kHz oscillator. The timer 4226 is preconfigurable and has two programmable timeouts. The timer 4226 has 24 programmable bits, giving a total time count of up to 17 minutes and 4 seconds. The timer also has a programmable delay to disable the clock to the CLK_32kHz pad and set the microprocessor (uP) interface signals on the VPAD plane to a predetermined state (see the Microprocessor Wakeup Control Signals section below). This allows the microprocessor to enter suspend mode without an external clock. However, this feature may be disabled by software using a programmable bit.

[0235] The timer also includes a programmable delay to wake up the microprocessor by enabling the CLK_32kHz clock output and setting UP_WAKEUP high. When POR2 (VDD POR) transitions from the powered low state to the powered OK state, the 32kHz oscillator, the CLK_32kHz clock output, and UP_WAKEUP are enabled and set high. Power down and power up are configured to be controlled by programmable control bits.

[0236] Real-time clock (RTC) The ASIC also contains a 48-bit readable / writeable binary counter that runs off an ungated, free-running 32 kHz oscillator. Writes to the real-time clock 4228 require a write to the keyed address before the clock can be written. Write access to the clock is configured to complete between 1 ms and 20 ms after a write to the keyed address.

[0237] The real-time clock 4228 is configured to be reset to half count (MSB=1, all other bits 0) by a power-on reset from either POR1_IN (VBAT POR) or POR2_IN (VDD_POR). In embodiments of the present invention, the real-time clock has programmable interrupt capability and is designed to be robust against single event upsets (SEUs), which may be achieved by layout techniques or by adding capacitance to appropriate nodes as needed.

[0238] RC oscillator The ASIC also includes an RC clock powered by VBAT or a VBAT-derived source. The RC oscillator is always running, except that it can be bypassed by writing a register bit in analog test mode (see the Digital Test section) and applying a signal to GPIO_VBAT with a level between 0 and VBAT. The RC oscillator is not trimmable and has the following specifications: (i) a frequency between 750 Hz and 1500 Hz; (ii) a duty cycle of 50% ±10%; (iii) a current consumption of less than 200 nA at 25°C; (iv) a frequency variation of less than ±2% from 1 V to 4.5 V VBAT and greater than 1% from 1.8 V to 4.5 V VBAT; and (v) a frequency variation of less than +2, -2% from 15°C to 40°C at VBAT = 3.5 V. The RC frequency can be measured using a 32 kHz crystal oscillator or with an external frequency source (see Oscillator Calibration Circuit).

[0239] Real-time RC clock (RC oscillator based) The ASIC contains a 48-bit readable / writeable binary ripple counter based on an RC oscillator. Writes to the RC real-time clock require a write to a keyed address before the clock can be written. Write access to the clock completes 1 ms to 20 ms after a write to the keyed address, and a protection window time is configured to be generated by the RC clock.

[0240] The Real-Time RC Clock allows for relative time stamps if the crystal oscillator is configured to shut down and reset to half count (MSB=1, all others 0) on POR1_IN (BAT POR). The Real-Time RC Clock is designed to be robust against Single Event Upsets (SEUs) through layout techniques or by adding capacitance to appropriate nodes where necessary. On the falling edge of POR2_IN or when the ASIC goes to battery low, the RT Real-Time Clock value may be captured in a register that can be read through the SPI port. This register and associated logic reside on the VBAT or VDDBU power plane.

[0241] battery protection circuit The ASIC includes a battery protection circuit 4230 that uses a comparator to monitor the battery voltage and is powered by power drawn from the VBAT power plane. The battery protection circuit is configured to operate constantly with power applied to the VBAT supply. The battery protection circuit uses an RC oscillator for its clock signal and can have an average current drain of less than 30 nA, including a 3 megohm total resistance external voltage divider.

[0242] The battery protection circuit uses an external switched voltage divider with a ratio of 0.421 for a 2.90V battery threshold. The ASIC also has an internal voltage divider with a ratio of 0.421 ±0.5%. This divider is connected between BATT_DIV_EN (4232) and VSSA (4234), and the divider output is a pin called BATT_DIV_INT (4236). To save pins on packaged parts, BATT_DIV_INT in this embodiment is connected to BATT_DIV internally in the package. This configuration also saves two package pins because BATT_DIV_EN does not need to be brought out of the package.

[0243] The battery protection circuit is configured to sample the voltage on input pin BATT_DIV (4238) approximately twice per second, with the sampling time being generated from an RC oscillator. The ASIC adjusts the RC oscillator divider to adjust the sampling time interval to 0.500 seconds ±5 milliseconds, allowing the RC oscillator to operate within its operating tolerances. In a preferred embodiment, the ASIC has a test mode that allows for more frequent sampling intervals during testing.

[0244] The comparator input is configured to accept an input between 0 and VBAT volts. The input current to the comparator input, BATT_DIV, is less than 10 nA with an input between 0 and VBAT volts. The comparator sampling circuit outputs a positive pulse on pad BATT_DIV_EN that can be used by external circuitry to enable an off-chip resistor divider only during the sampling time to save power. The high logic level of the voltage is the VBAT voltage, and the low level is the VSS level.

[0245] The output resistance of the BATT_DIV_EN pad should be less than 2 kOhms at VBAT=3.0V. This allows a voltage divider to be driven directly from this output. After a programmable number of consecutive samples indicating a low battery condition, the comparator control circuit triggers an interrupt UP_INT to the interrupt output pad. The default number of samples is 4, but the number of consecutive samples is programmable from 4 to 120.

[0246] The comparator control circuit is configured to generate a signal that will put the ASIC into low-power mode after a programmable number of consecutive samples indicating low battery after the generation of UP_INT mentioned above. The VDD regulator will be disabled and a low signal will be asserted on pad VPAD_EN. This can be called a low battery state. Again, the number of consecutive samples is programmable from 4 to 120 samples, with a default of 4 samples.

[0247] The comparator has separate programmable thresholds for the falling and rising voltage of the BATT_DIV line. This is implemented in digital logic to multiplex the two values ​​into the circuit depending on the state of the battery low condition. Therefore, the falling threshold applies when the battery low condition is low, and the rising threshold applies when the battery low condition is high. Specifically, the comparator has 16 programmable thresholds ranging from 1.22 to 1.645 ±3%, with the DNL of the programmable thresholds set to less than 0.2 LSB.

[0248] The comparator thresholds vary less than + / -1% from 20°C to 40°C. The default threshold for falling voltage is 1.44V (3.41V VBAT threshold using the nominal voltage divider), and the default threshold for rising voltage is 1.53V (3.63V VBAT threshold using the nominal voltage divider). After the ASIC enters a low battery state, if the comparators detect four consecutive indicators of battery OK, the ASIC will initiate the microprocessor start-up sequence.

[0249] Battery power plane power on reset A power-on reset (POR) output is generated at pad nPOR1_OUT (4240) when the input VBAT slews above 1.2 volts for a 50 microsecond period, or when the VBAT voltage falls below 1.6 ±0.3 volts. This POR is stretched to a minimum pulse width of 5 milliseconds. The output of the POR circuit is configured to be active low and goes to pad nPOR1_OUT on the VBAT power plane.

[0250] The IC has an input pad for the battery power plane POR, nPOR1_IN (4242). This input pad provides RC filtering so that pulses shorter than 50 nsec do not cause a logic reset. In this embodiment, nPOR1_OUT is externally connected to nPOR1_IN during normal operation, thereby isolating the analog circuitry from the digital circuitry for testing. nPOR1_IN causes a reset of all logic on any of the power planes and initializes all registers to their default values. Therefore, the POR bit in the reset status register is set and all other reset status register bits are cleared. The POR reset circuit is configured to consume no more than 0.1 uA from the VBAT power supply for more than 5 seconds after power-on.

[0251] VDD Power-On Reset (POR) The ASIC also has a voltage comparator circuit that generates a VDD voltage plane reset signal at power-up or if VDD drops below a programmable threshold. The range is programmable with several voltage thresholds. The default value is 1.8V to 15% (1.53V). POR2 has a programmable threshold for the rising voltage that implements hysteresis. The rising threshold is also programmable, with a default value of 1.60V ±3%.

[0252] The POR signal is active low and has an output pad nPOR2_OUT (4244) on the VDD power plane. The ASIC also has an active low POR open-drain output nPOR2_OUT_OD (4246) on the VBAT power plane, which can be used to apply the POR to other system components.

[0253] The VDD powered logic has a POR derived from the input pad nPOR2_IN (4248). The nPOR2_IN pad is on the VDD power plane and provides RC filtering to prevent pulses shorter than 50 nsec from resetting the logic. nPOR2_OUT is configured to be externally connected to the nPOR2_IN input pad in normal use, which separates the analog circuitry from the digital circuitry.

[0254] The generated reset is extended to an active period of at least 700ms after VDD exceeds a programmable threshold to ensure the crystal oscillator is stable. The POR reset circuitry is designed to consume less than 0.1uA from the VDD supply for more than 5 seconds after power-on, and less than 0.1uA from the VBAT supply for more than 5 seconds after power-on. The register that stores the POR threshold is powered from the VDD power plane.

[0255] Sensor Interface Electronics In the embodiment of the invention described herein, the sensor circuitry supports up to five sensor WORK electrodes (4310) for any combination of peroxide or oxygen sensors, although additional embodiments may accommodate more such electrodes. The oxygen sensor WORK electrode sinks current while the peroxide sensor WORK electrode sources current. For this embodiment, the sensor can be configured in a potentiostat configuration, as shown in FIG. 43.

[0256] The sensor electronics has a programmable power controller for each electrode interface circuit, minimizing current drain by turning off current to the electronics of unused sensors. The sensor electronics also includes electronics that drive the COUNTER electrode 4320 using feedback from the RE (reference) electrode 4330. Current to this circuit can be programmed to be turned off when not in use to conserve power. The interface electronics includes a multiplexer 4250 so that the COUNTER and RE electrodes can be connected to either of the (redundant) WORK electrodes.

[0257] The ASIC is configured to provide the following sensor interfaces: (i) RE: a reference electrode that establishes a reference potential for the electronics solution to set the WORK voltage; (ii) WORK1-WORK5: sensor electrodes where the desired reduction / oxidation (redox) reactions occur; and (iii) COUNTER: the output from this pad maintains a known voltage on the RE electrode relative to the system's VSS. In this embodiment, the ASIC is configured to allow the WORK voltage to be set independently for up to five WORK electrodes with a resolution and accuracy of 5 mV or better.

[0258] The WORK voltage(s) are programmable from at least 0 to 1.22 V relative to VSSA in oxygen mode. In peroxide mode, the WORK voltage(s) are programmable from at least 0.6 V to 2.054 V relative to VSSA. When VDDA is less than 2.15 V, the WORK voltage operates at VDDA-0.1 V. The ASIC includes a current measurement circuit that measures the WORK electrode current in peroxide sensor mode. This may be implemented with a current-to-voltage or current-to-frequency converter that may have the following specifications, for example: (i) current range: 0 to 300 nA; (ii) voltage output range: same as the WORK electrode in peroxide / oxygen mode; (iii) output offset voltage: maximum ±5 mV; and (iv) uncalibrated resolution: ±25 nA.

[0259] After applying a calibration factor to the gain and assuming an acquisition time of 10 seconds or less, the current measurement accuracy is: 5pA~1nA:±3%±20pA 1nA-10nA: ±3% ±20pA 10nA~300nA: ±3% ±0.2nA.

[0260] For the current-to-frequency converter (ItoF) only, the frequency range can be 0 Hz to 50 kHz. The current converter must operate within the specified voltage range relative to the VSS of the WORK electrode in peroxide mode, where the current drain is less than 2 uA from a 2.5 V supply and the WORK electrode current is less than 10 nA per converter, including the digital-to-analog (DAC) current.

[0261] The current converter can be enabled or disabled by software control. When disabled, the WORK electrode presents a very high impedance value, i.e., greater than 100 megaohms. Again, for ItoF only, the output of the IF converter goes to a 32-bit counter that can be read, written, and cleared by the microprocessor and test logic. During counter readings, counter clocking is suspended to ensure accurate readings.

[0262] In the embodiment of the invention described herein, the ASIC also includes a current measurement circuit for measuring the WORK electrode current in oxygen sensor mode. This circuit may be implemented as a current-to-voltage converter or a current-to-frequency converter, and a programmable bit may be used to configure the current converter for operation in oxygen mode. As before, the current converter must operate within the specified voltage range of the WORK electrode relative to VSS in oxygen mode. Here, the current range is 3.7 pA to 300 nA, the voltage output range is the same as the WORK electrode in oxygen mode, the output offset voltage is up to ±5 mV, and the uncalibrated resolution is 3.7 pA ±2 pA.

[0263] After applying a calibration factor to the gain and assuming an acquisition time of 10 seconds or less, the current measurement accuracy is: 5pA-1nA: ±3% ±20pA 1nA-10nA: ±3% ±20pA 10nA-300nA: ±3% ±0.2nA.

[0264] For the current-to-frequency converter (ItoF) only, the frequency range may be between 0 Hz and 50 kHz, and the current drain is less than 2 μA from a 2.5 V supply with WORK electrode current, including DAC current, being less than 10 nA per converter. The current converters can be enabled or disabled via software control. When disabled, the WORK electrodes present very high impedance values, i.e., greater than 100 megaohms. Also, for ItoF only, the output of the IF converter goes to a 32-bit counter that can be read, written, and cleared by the microprocessor and test logic. During counter reads, counter clocking is suspended to ensure accurate readings.

[0265] In the embodiment of the invention described herein, the reference electrode (RE) 4330 has an input bias current of less than 0.05 nA at 40° C. The COUNTER electrode adjusts the output of the COUNTER electrode to maintain the desired voltage on the RE electrode. This is achieved using an amplifier 4340 whose output to the COUNTER electrode 4320 attempts to minimize the difference between the actual RE electrode voltage and the target RE voltage set by the DAC.

[0266] The RE set voltage is programmable from at least 0 to 1.80 V, and the COUNTER amplifier common-mode input range includes at least 0.20 to (VDD-0.20) V. If desired, register bits may be used to select the common-mode input range and program the COUNTER operating mode. The WORK voltage is set with a resolution and accuracy of 5 mV or better. Note that in normal mode, the COUNTER voltage seeks a level that maintains the RE voltage at the programmed RE target value. However, in force-opposed mode, the COUNTER electrode voltage is forced to the programmed RE target voltage.

[0267] All electrode drive circuits are configured to drive the electrodes into a load and to avoid oscillation in any use scenario. Figure 44 shows an equivalent AC inter-electrode circuit for the embodiment having the potentiostat configuration shown in Figure 43. The equivalent circuit shown in Figure 44 may be between any of the electrodes, i.e., WORK1-WORK5, COUNTER, and RE, and has the following ranges of values ​​for each circuit component: Ru = [200~5k] ohms Cc=[10~2000]pF Rpo = [1 to 20] kiloohms Rf = [200~2000] kiloohms Cf=[2~30]uF

[0268] During initialization, the drive currents for the WORK and COUNTER electrodes must provide higher currents than in normal potentiostat operation as described above. Therefore, if additional drive is required, a programmable register bit may be used to program the electrode drive circuitry to a higher power state. It is important to achieve low-power operation in normal potentiostat mode, where electrode currents are typically less than 300 nA.

[0269] In a preferred embodiment, during initialization, the WORK1 through WORK5 electrodes are programmable from 0 to VDD volts in steps of 5 mV or less, and their drive or sink current output capability is a minimum of 20 μA from 0.20 V to (VDD - 0.20 V). Also, during initialization, the ASIC is typically configured to be able to measure the current of one WORK electrode up to 20 μA with an accuracy of ±2% of the measured value, ±40 nA. Furthermore, during initialization, the RE set voltage is programmable as described above, the COUNTER DRIVE CIRCUIT output can source or sink a minimum of 50 μA using the COUNTER electrodes from 20 V to (VDD - 0.20 V), and the power supply currents to the initialization circuit (VDD and VDDA) must exceed any output current sourced by less than 50 μA.

[0270] Current Calibrator In an embodiment of the present invention, the ASIC has a current reference that can be directed to any WORK electrode for calibration purposes. In this regard, the calibrator includes a programmable bit to sink or source the current output. Programmable currents include at least 10 nA, 100 nA, and 300 nA, with an accuracy of better than ±1% ±1 nA, assuming an external precision resistor with zero tolerance. The calibrator uses a 1-megohm precision resistor, TP_RES (4260), connected to a pad for the reference resistance. Additionally, a current reference can be directed to the COUNTER or RE electrode for initialization and sensor status purposes. A constant current may be applied to the COUNTER or RE electrode, and the electrode voltage may be measured using an ADC.

[0271] High-Speed ​​RC Oscillator Referring again to FIG. 42, the ASIC further includes a fast RC oscillator 4262 that supplies an analog-to-digital converter (ADC) 4264, an ADC sequencer 4266, and other digital functions that require a clock faster than 32 kHz. The fast RC oscillator is phase-locked to the 32 kHz clock (32.768 kHz) and provides a programmable output frequency from 524.3 kHz to 1048 kHz. In addition, the fast RC oscillator has a duty cycle of 50% ±10%, a phase jitter of less than 0.5% rms, a current of less than 10 μA, and a frequency that is stable throughout the VDD operating range (voltages ranging from 1.6 V to 2.5 V). The fast RC oscillator defaults to "off" (i.e., disabled), in which case the current draw is less than 10 nA. However, the ASIC has a programmable bit to enable the fast RC oscillator.

[0272] Analog-to-Digital Converter The ASIC has the following: (i) the ability to perform conversions in less than 1.5 milliseconds when running from a 32 kHz clock; (ii) the ability to perform faster conversions when clocked from a fast RC oscillator; (iii) at least 10-bit accuracy (12-bit ±4 counts); (iv) a reference voltage input of 1.220 V and a temperature sensitivity of less than 0.2 mV / °C from 20°C to 40°C; (v) full-scale input ranges of 0 to 1.22 V, 0 to 1.774 V, 0 to 2.44 V, and 0 to VDDA, with the 1.774 and 2.44 V ranges having a programmable bit to reduce the conversion range to lower values ​​to accommodate lower VDDA voltages; and (vi) a current draw from the ASIC's power supply of 50 (vi) have a converter that can operate with a 32 kHz clock or a fast RC clock; (vii) have a DNL of less than 1 LSB; and (viii) issue an interrupt at the end of conversion.

[0273] As shown in Figures 42A and 42B, the ASIC has an analog multiplexer 4268 at the input of the ADC 4264, both of which are software controllable. In a preferred embodiment, at least the following signals are connected to the multiplexer: (i) VDD- Core voltage and regulator output (ii) VBAT-Battery power supply (iii) VDDA - Analog power supply (iv) Reference electrode of RE-sensor (v) COUNTER - Counter electrode of the sensor (vi) WORK1 to WORK5 - Working electrodes of the sensor (vii) Temperature sensor (viii) At least two external pins for analog signal input (ix)EIS integrator output (x) ItoV current converter output

[0274] The ASIC is configured so that the loading of the ADC does not exceed ±0.01 nA for inputs COUNTER, RE, WORK1 through WORK5, the temperature sensor, and other inputs adversely affected by the loading. The multiplexer includes a divider for inputs having a voltage higher than the input voltage range of the ADC, and a buffer amplifier adapted to reduce the input resistance of the divided input for the load-sensing input to less than 1 nA. The buffer amplifier has a common-mode input range of at least 0.8 V to VDDA voltage and an offset of less than 3 mV from the input range of 0.8 V to VDDA-0.1 V.

[0275] In a preferred embodiment, the ASIC has a mode in which ADC measurements are taken in a programmed sequence. Therefore, the ASIC includes a programmable sequencer 4266 that monitors the measurement of up to eight input sources for ADC measurements, using the following programmable parameters: (i) ADC MUX input (ii) ADC Range (iii) Delay time before measurement. The delay is programmable from 0 to 62 ms in 0.488 ms steps. (iv) Number of measurements for each input, 0 to 255 (v) Number of measurement cycles: 0 to 255, where a measurement cycle refers to repeating a measurement sequence of up to 8 inputs multiple times (for example, as an outer loop of a program). (vi) Delay between measurement cycles. The delay is programmable from 0 to 62 ms in steps of 0.488 ms.

[0276] The sequencer 4266 is configured to start upon receiving an automatic measurement start command, and the measurement can be stored in the ASIC for retrieval via the SPI interface. Note that the sequencer's time base is programmable between a 32 kHz clock and the fast RC oscillator 4262.

[0277] Sensor Diagnostics As discussed in detail above, embodiments of the invention described herein are directed to the use of impedance and impedance-related parameters, for example, in sensor diagnostic procedures and Isig / SG fusion algorithms. To that end, in preferred embodiments, the ASIC described herein has the capability to measure the impedance magnitude and phase angle of any WORK sensor electrode relative to the RE and COUNTER electrodes when in a potentiostat configuration. This is done, for example, by measuring the amplitude and phase of the current waveform in response to a sinusoidal waveform superimposed on the WORK electrode voltage. See, for example, diagnostic circuit 4255 in FIG. 42B.

[0278] The ASIC has the capability to measure the resistive and capacitive components of any electrode, for example, via electrode multiplexer 4250. Note that such measurements may interfere with the balance of the sensor and may require settling time or sensor initialization to record stable electrode currents. As previously discussed, the ASIC may be used for impedance measurements over a wide spectrum of frequencies, but for purposes of embodiments of the present invention, a relatively narrow frequency range may be used. Specifically, the ASIC's sinusoidal wave measurement capability may include test frequencies from about 0.10 Hz to about 8192 Hz. When making such measurements, the minimum frequency resolution according to embodiments of the present invention may be limited as shown in Table 2 below. [Table 14]

[0279] The amplitude of the sine wave is programmable from at least 10 mV p-p to 50 mV p-p in 5 mV steps, and from 60 mV p-p to 100 mV p-p in 10 mV steps. In preferred embodiments, amplitude accuracy is better than ±5% or ±5 mV, whichever is greater. Additionally, the ASIC may measure electrode impedance to the accuracy specified in Table 3 below. [Table 15]

[0280] In embodiments of the present invention, the ASIC can measure the input waveform phase relative to a time base, which can be used in the impedance calculations to improve accuracy. The ASIC may also have an on-chip resistor for calibrating the electrode impedance circuitry. The on-chip resistor may be calibrated by comparison with a known 1 megaohm off-chip precision resistor.

[0281] Data sampling of the waveform may be used to determine impedance. The data may be sent to an external microprocessor with a serial peripheral interface (SPI) for calculations and processing. The converted current data is sufficiently buffered that 2000 ADC conversions of data can be transferred to an external device via the SPI interface without data loss. This assumes a maximum latency of 8 milliseconds for servicing the data transfer request interrupt.

[0282] In embodiments of the present invention, rather than or in addition to measuring electrode impedance using a sine wave, the ASIC may measure electrode current using a step input. Here, the ASIC may apply programmable amplitude steps of 10 to 200 mV with a resolution better than 5 mV to the electrode and sample (measure) the resulting current waveform. The duration of the sampling may be programmable to at least 2 seconds in 0.25 second steps, and the sampling interval for measuring the current may include at least five programmable binary-weighted steps of approximately 0.5 milliseconds to 8 milliseconds.

[0283] The resolution of the electrode voltage sample is less than 1 mV with a range of up to ±0.25 volts. This measurement can be referenced to a suitable stable voltage to reduce the dynamic range required for data conversion. Similarly, the resolution of the electrode current sampling is less than 0.04 uA with a range of up to 20 uA. The current measurement can be unipolar, where the measurement polarity is programmable.

[0284] In embodiments of the present invention, current measurements may use an IV converter. Additionally, the ASIC may have an on-chip resistor for calibrating the current measurements. The on-chip resistor may then be calibrated by comparison with a known 1 megohm off-chip precision resistor. The accuracy of the current measurement sampling is better than ±3% or ±10 nA, whichever is greater. As before, the converted current data is sufficiently buffered that 2000 ADC conversions of data can be transferred to an external device via the SPI interface without data loss. This assumes a maximum latency of 8 milliseconds for servicing the data transfer request interrupt.

[0285] Calibration Voltage The ASIC includes a precision voltage reference for calibrating the ADC. The output voltage is 1.000V ±3% with less than ±1.5% manufacturing variation and better than ±3mV stability over a temperature range of 20°C to 40°C. This precision calibration voltage may be calibrated during manufacturing by comparison to an external precision voltage via the on-chip ADC. During manufacturing, calibration coefficients may be stored in the system's non-volatile memory (not in the ASIC) to achieve even greater accuracy.

[0286] The current drain of the calibration voltage circuit is preferably less than 25uA. Additionally, the calibration voltage circuit can power down to less than 10nA to conserve battery power when not in use.

[0287] Temperature Sensor The ASIC has a temperature transducer with a sensitivity of 9-11 mV per degree Celsius over a range of -10°C to 60°C. The output voltage of the temperature sensor is such that the ADC can measure temperature-related voltages over an ADC input range of 0 to 1.22 V. The current drain of the temperature sensor is preferably less than 25 uA, and the temperature sensor can power down to less than 10 nA to conserve battery power when not in use.

[0288] VDD voltage regulator The ASIC has a VDD voltage regulator with the following characteristics: (i) Minimum input voltage range: 2.0V~4.5V. (ii) Minimum output voltage: 1.6~2.5V±5%, default is 2.0V. (iii) Dropout voltage: Vin-Vout<0.15V when Iload=100uA and Vin=2.0V. (iv) The output voltage is programmable and accurate to within 2% of the values ​​shown in Table 4 below. [Table 16] (v) The regulator can provide an output of 2.5V at 1mA with an input voltage of 2.8V. (vi) The regulator also has input and output pads that can be open circuited when using an external regulator. The current draw of the regulator circuit is preferably less than 100 nA in this non-operating mode. (vii) The change in output voltage from a load of 10 uA to 1 mA is preferably less than 25 mV. (viii) With a 1mA load, the current drain, excluding the output current, is less than 100uA from the source. (ix) With a 0.1mA load, the current drain, excluding the output current, is less than 10uA from the source. (x) With a 0.1mA load, the current drain, excluding output current, is less than 10uA from the source.

[0289] General-purpose comparator The ASIC contains at least two comparators 4270, 4271 powered by VDDA. The comparators use 1.22V as a reference to generate thresholds. The outputs of the comparators can be read by the processor and are adapted to create maskable interrupts on either a rising or falling edge, as determined by a configuration register.

[0290] The comparators have power controls to reduce power when not in use, and draw less than 50 nA of current per comparator. The comparator response time is preferably less than 50 microseconds to a 20 mV overdrive signal, and the offset voltage is less than ±8 mV.

[0291] The comparators also have programmable hysteresis; hysteresis options include threshold = 1.22V + Vhyst on rising inputs, threshold = 1.22 - Vhyst on falling inputs, or no hysteresis (Vhyst = 25 ± 10mV). The output from either comparator is available to any GPIO on any power plane (see GPIO section).

[0292] Sensor connection detection circuit for RE An analog switched-capacitor circuit monitors the impedance of the RE connection to determine whether the sensor is connected. Specifically, a roughly 20 pF capacitor is switched at a frequency of 16 Hz, driven by an inverter whose output swings from VSS to VDD. A comparator detects the voltage swing on the RE pad; if the swing is below a threshold, the comparator output indicates a connection. This comparison is made on both transitions of the pulse. A swing below the threshold on both transitions is required to indicate a connection; a comparison showing a high swing on either phase indicates a disconnect. The connect / disconnect signal is debounced so that its state transition requires a stable indication to the new state for at least 1 / 2 second.

[0293] This circuit has six thresholds defined by the following resistors in parallel with a 20pF capacitor: 500k, 1M, 2M, 4M, 8M, and 16M. This parallel equivalent circuit is between the RE pad and a virtual ground, which can be at any voltage between the power rails. The threshold accuracy is better than ±30%.

[0294] The output of the sensor connection sensor circuit can programmably generate an interrupt or wake up a processor when a sensor is connected or disconnected. This circuit is active whenever nPOR2_IN is high and VDD and VDDA are present. The current drain of this circuit is less than 100nA on average.

[0295] WAKEUP Pad The WAKEUP circuit is powered by the VDD supply and has an input range of 0V to VBAT. The WAKEUP pad 4272 has a weak pull-down of 80±40nA. This current is drawn from the output of BIAS_GEN 4220. The average current consumed by the circuit is less than 50nA with a 0V input.

[0296] The WAKEUP input has a rising input voltage threshold, Vih, of 1.22 ± 0.1 V, and a falling input threshold of -25 mV ± 12 mV of the rising threshold. In a preferred embodiment, the circuitry associated with the WAKEUP input draws less than 100 nA for any input with a value between -2 V and the VBAT voltage (this current excludes input pull-down current). The WAKEUP pad is debounced for at least 1 / 2 second.

[0297] The output of the WAKEUP circuit can programmably generate an interrupt or wake up a processor if the WAKEUP pad changes state (see the "Event Handler" section). It is important to note that the WAKEUP pad circuit is configured to assume a low current of less than 1 nA when the battery protection circuit indicates a low battery condition.

[0298] UART WAKEUP The ASIC is configured to monitor the nRX_EXT pad 4274. If the nRX_EXT level is continuously high (UART BREAK) for longer than 1 / 2 second, a UART WAKEUP event will be generated. The UART WAKEUP event can be generated with a continuously high value for as short as 1 / 4 second due to sampling. The UART WAKEUP event can programmably generate an interrupt, WAKEUP, or a microprocessor reset (nRESET_OD). (See the "Event Handler" section.)

[0299] In a preferred embodiment, the circuitry associated with the UART WAKEUP input draws 100 nA or less, and the UART WAKEUP pad circuit is configured to assume a low current of less than 1 nA when the battery protection circuit indicates a low battery condition. The UART Wakeup input has a rising input voltage threshold, Vih, of 1.22 ± 0.1 V, and a falling input threshold that is −25 mV ± 12 mV of the rising threshold.

[0300] Microprocessor WAKEUP control signal The ASIC may generate signals useful for controlling the power management of the microprocessor. Specifically, the ASIC may generate the following signals: (i) nSHUTDN - nSHUTDN may control the power enable of the off-chip VDD regulator. The nSHUTDN pad is on the VBAT power rail. If the battery protection circuit indicates a low battery condition, nSHUTDN shall be low; otherwise, nSHUTDN shall be high. (ii) VPAD_EN—VPAD_EN may control the power enable of the external regulator that supplies VPAD power. An internal signal corresponding to this external signal ensures that input from the VPAD pad does not cause excess current due to a floating input when the VPAD power supply is disabled. The VPAD_EN pad is an output on the VBAT power rail. If the battery protection signal indicates a low battery, the VPAD_EN signal is low. The VPAD_EN signal may be set low by a software command that starts a timer, whose terminal count forces VPAD_EN low. If the battery protection signal indicates a good battery, the following events may cause the VPAD_EN signal to go high: nPOR2_IN low-to-high transition, SW / Timer (programmable), WAKEUP transition from low to high and / or high to low (programmable), sensor connection transition from low to high and / or high to low (programmable), UART break, and RTC time event (programmable) (see Event Handler for details). (iii) UP_WAKEUP - UP_WAKEUP may be connected to a microprocessor wake-up pad. It is intended to wake the microprocessor from sleep mode or a similar power-down mode. The UP_WAKEUP pad is an output on the VPAD power rail. The UP_WAKEUP signal can be programmed to be active-low, active-high, or pulsed. The UP_WAKEUP signal may be set low by a software command that starts a timer, the terminal count of which forces UP_WAKEUP low. If the battery protection signal indicates a good battery, the following events may cause the UP_WAKEUP signal to go high: nPOR2_IN low-to-high transition, SW / Timer (programmable), WAKEUP transition from low to high and / or high to low (programmable), sensor connection transition from low to high and / or high to low (programmable), UART break, and RTC time event (programmable) (see Event Handler for details). The WAKEUP signal may be delayed by a programmable amount. If WAKEUP is programmed to be pulsed, the pulse width can be programmed. (iv) CLK_32kHz - The CLK_32kHz pad can be connected to a microprocessor to provide a low-speed clock. The clock is programmable on / off and has a programmable wake-up event. The CLK_32kHz pad is an output on the VPAD power rail. The CLK_32kHz signal is low when the battery protection signal indicates a low battery. The CLK_32kHz output can be programmed off by a programmable bit. The default is on. The CLK_32kHz signal can be disabled by a software command that starts the timer. The timer's terminal count forces CLK_32kHz low. If the battery protection signal indicates a good battery, the following events may enable the CLK_32kHz signal: nPOR2_IN low to high transition, SW / Timer (programmable), WAKEUP transition from low to high and / or high to low (programmable), sensor connection transition from low to high and / or high to low (programmable), UART break, RTC time event (programmable), and detection of low battery by the battery protection circuit (see Event Handler for details). (v) nRESET_OD - nRESET_OD connects to the microprocessor and can cause it to reset. nRESET_OD is programmable for wake up events. The nRESET_OD pad is an output on the VPAD power rail. This pad is open drain (nfet output). The nRESET_OD signal is low when the battery protection signal indicates a low battery. The active time of nRESET_OD is programmable from 1 to 200 ms. The default is 200 ms. The following events can cause the nRESET_OD signal to assert low: nPOR2_IN, SW / Timer (programmable), WAKEUP transitions from low to high and / or high to low (programmable), sensor connection transitions from low to high and / or high to low (programmable), UART break, and RTC time event (programmable) (see Event Handler for details). (vi) UP_INT - UP_INT connects to the microprocessor to communicate interrupts. UP_INT is programmable to wake up events. The UP_INT pad is an output on the VPAD power rail. If the battery protection signal indicates a low battery, the UP_INT signal is low. The UP_INT signal may be set high by a software command that starts a timer, the terminal count of which forces UP_INT high. If the battery protection signal indicates a good battery, any of the following events may cause the UP_INT signal to assert high: SW / Timer (programmable), WAKEUP transition from low to high and / or high to low (programmable), sensor connection transition from low to high and / or high to low (programmable), UART break, RTC time event (programmable), low battery detection by the battery protection circuit, and ASIC interrupts when not masked (see Event Handler for details).

[0301] The ASIC has GPIO1 and GPIO0 pads that can act as boot mode controls for the microprocessor. A POR2 event will reset a 2-bit counter whose bits are mapped to GPIO1 and GPIO0 (MSB, LSB respectively). The rising edge of the UART break will increment the counter by 1; the counter counts modulo 4 and reaches zero when incremented in state 11. The boot mode counter can be pre-configured via SPI.

[0302] Event Handlers / Watchdogs The ASIC incorporates event handlers that define responses to events, including changes in system state and input signals. Events include all interrupt sources (e.g., UART_BRK, WAKE_UP, sensor connection, etc.). The event handler's response to stimuli is software programmable via the SPI interface; however, some responses may be hardwired (non-programmable).

[0303] Event handler actions include enabling / disabling VPAD_EN, enabling / disabling CLK_32kHz, asserting nRESET_OD, asserting UP_WAKEUP, and asserting UP_INT. Event watchdog timers 1 through 5 are individually programmable from 250 milliseconds to 16,384 seconds in 250 millisecond increments. The timeouts for event watchdog timers 6 through 8 are hard-coded. Timers 6 and 7 have a 1-minute timeout, and timer 8 has a 5-minute timeout.

[0304] The ASIC also has a watchdog function that monitors the microprocessor's response when triggered by an event. The event watchdog is activated when the microprocessor fails to acknowledge event-induced activity. When activated, the event watchdog executes a programmable sequence of actions, Event Watchdog Timers 1 through 5, followed by a hardwired sequence of actions, Event Watchdog Timers 6 through 8, to regain the microprocessor's response. The sequence of actions includes an interrupt, reset, wakeup, assertion of the 32 kHz clock, power down, and power up to the microprocessor.

[0305] During a sequence of actions, if the microprocessor regains its ability to acknowledge the logged activity, the event watchdog is reset. If the ASIC does not receive acknowledgement from the microprocessor, the event watchdog will power down the microprocessor, providing UART_BRK is able to restart the microprocessor, and will activate an alarm. When activated, the alarm condition generates a square wave on pad ALARM with a frequency of approximately 1 kHz in a programmable repeating pattern. The programmable pattern has two programmable sequences with programmable burst on and off times. The alarm has another programmable pattern that can be programmable via the SPI port. In the case of another programmable pattern, it has two programmable sequences with programmable burst on and off times.

[0306] Digital-to-Analog (D / A) In the preferred embodiment, the ASIC has two 8-bit D / A converters 4276, 4278 with the following characteristics: (i) The D / A settles in less than 1 millisecond with a load of less than 50 pF. (ii) The D / A has at least 8-bit precision. (iii) The output range is programmable to either 0 to 1.22V or 0 to VDDA. (iv) The temperature sensitivity of the D / A voltage reference is less than 1 mV / °C. (v) DNL is less than 1 LSB. (vi) The current consumed by the D / A is less than 2uA from the VDDA supply. (vii) Each D / A has an output I to a pad. (viii) The D / A output is high impedance. The load current must be less than 1 nA. (ix) The D / A pad can be programmed to output a digital signal from the register. The output swing is VSSA to VDDA.

[0307] Charger / Data Downloader Interface The TX_EXT_OD 4280 is an open-drain output whose input is the signal on the TX_UP input pad. This allows the TX_EXT_OD pad to be released in UART idle conditions. The TX_EXT_OD pad has a comparator that monitors its voltage. When the voltage exceeds the comparator threshold voltage for the debounce period (1 / 4 second), the output nBAT_CHRG_EN (4281) goes low. This comparator and other associated circuitry with this function reside on the VBAT and / or VDDBU planes.

[0308] The circuitry associated with this function must tolerate a low on the TX_EXT_OD pad resulting from normal communication with an external device without disabling the assertion of nBAT_CHRG_EN. nBAT_CHRG_EN is high (not asserted) when POR1 is active. The comparator threshold voltage is 0.50V to 1.2V. The comparator has hysteresis, with the falling threshold approximately 25mV lower than the rising threshold.

[0309] The nRX_EXT pad inverts the signal on this pad and outputs it to RX_UP. This way, the nRX_EXT signal idles low. nRX_EXT must accept inputs up to the VBAT voltage. The nRX_EXT threshold is 1.22V ±3%. The output of this comparator is made available over the SPI bus for reading by the microprocessor.

[0310] The nRX_EXT pad also incorporates a means to programmably source a current of 80 ±30 nA, with a maximum voltage of VBAT. The ASIC layout has a mask-programmable option to adjust this current from 30 nA to 200 nA in steps of less than 50 nA, with a minimal number of mask layers. A programmable bit is available to block UART break detection and force RX_UP high. In normal operation, this bit is set high before enabling the current source to nRX_EXT, and then set low after the current source is disabled, ensuring that no glitches are generated on RX_UP or UART break events are generated. Note that a wet connector detector is implemented; while the current source to nRX_EXT is active, an RX comparator output indicating a low input voltage indicates leakage current. The ASIC includes a pulldown resistor of approximately 100 kOhm on the nRX_EXT pad. This pulldown is disconnected when the current source is active.

[0311] Sensor connection switch The ASIC shall have a pad, SEN_CONN_SW (4282), capable of detecting low resistance to VSS (4284). SEN_CONN_SW sources a current of 5 to 25 uA at SEN_CONN_SW=0V and has a maximum open circuit voltage of 4 V. The ASIC layout shall have a mask programmable option to adjust this current from 1 uA to 20 uA in steps of less than 5 uA, with a minimum number of mask layers changed. SEN_CONN_SW shall have associated circuitry to detect the presence of resistance between SEN_CONN_SW and VSSA (4234) with a threshold of 2 kOhm to 15 kOhm. The average current drain of this circuit shall be a maximum of 50 nA. Sampling shall be used to achieve this low current.

[0312] Oscillator Calibration Circuit The ASIC has counters whose inputs can be driven from internal or external clock sources. One counter generates a programmable gating interval for the other counter. Gating intervals range from 1 to 15 seconds from a 32 kHz oscillator. Clocks that can be driven to either counter include 32 kHz, RC oscillator, fast RC oscillator, and input from any GPIO pad.

[0313] Oscillator Bypass The ASIC can replace the output of each oscillator with an external clock. The ASIC has a register that can only be written to when a specific TEST_MODE is asserted. This register has a bit that enables the external input of the RC oscillator and can be shared with other analog test control signals. However, this register does not allow the oscillator bypass bit to be activated unless TEST_MODE is active.

[0314] The ASIC also has an input pad for an external clock that bypasses the RC oscillator. Pad GPIO_VBAT is on the VBAT power plane. The ASIC also includes a bypass enable pad OSC32K_BYPASS for the 32kHz oscillator. When high, the 32kHz oscillator output is provided by driving the OSC32kHz_IN pad. Note that the OSC32kHz_IN pad is typically connected to a crystal.

[0315] The ASIC has an input for an external clock that bypasses HS_RC_OSC. The bypass is enabled by a programmable register bit. HS_RC_OSC can be programmably supplied by either a GPIO on the VDD plane or a GPIO on the VPAD plane.

[0316] SPI Slave Port The SPI slave port contains an interface consisting of a chip select input (SPI_nCS) 4289, a clock input (SPI_CK) 4286, a serial data input (SPI_MOSI) 4287, and a serial data output (SPI_MISO) 4288. The chip select input (SPI_nCS) is an active-low input that is asserted by an off-chip SPI master to initiate and qualify SPI transactions. When SPI_nCS is asserted low, the SPI slave port configures itself as an SPI slave and performs data transactions based on the clock input (SPI_CK). When SPI_nCS is inactive, the SPI slave port resets itself and remains in reset mode. This SPI interface supports block transfers, so the master must hold SPI_nCS low until the transfer is complete.

[0317] The SPI clock input (SPI_CK) is always intended to be asserted by the SPI master. The SPI slave port uses the rising edge of SPI_CK to latch incoming data on the SPI_MOSI input and the falling edge of SPI_CK to drive outgoing data on the SPI_MISO output. The serial data input (SPI_MOSI) is used to transfer data from the SPI master to the SPI slave. All data bits are asserted after the falling edge of SPI_CK. The serial data output (SPI_MISO) is used to transfer data from the SPI slave to the SPI master. All data bits are asserted after the falling edge of SPI_CK.

[0318] SPI_nCS, SPI_CK, and SPI_MOSI are always driven by the SPI master unless the SPI master is powered down. When VPAD_EN is low, these inputs are regulated so that the current drain associated with these inputs is less than 10 nA, and the SPI circuitry is held in reset or inactive. SPI_MISO is driven by the SPI slave port only when SPI_nCS is active; otherwise, SPI_MISO is tri-stated.

[0319] Chip select (SPI_nCS) defines and frames the data transfer packet for SPI data transactions. The data transfer packet consists of three parts: a 4-bit command section followed by a 12-bit address section, followed by any number of 8-bit data bytes. Command bit 3 is used as a direction bit: a "1" indicates a write operation and a "0" indicates a read operation. Combinations of command bits 2, 1, and 0 have the following definitions. Unused combinations are undefined. (i) 0000: Read data and increment address. (ii) 0001: Read data, no address change (iii) 0010: Read data and decrement address (iv) 1000: Write data and increment address (v) 1001: Write data, no change to address (vi) 1010: Write data and decrement address (vii) x011: Test port address specification

[0320] The 12-bit address section defines the starting byte address. If SPI_nCS remains active after the first data byte to indicate a multi-byte transfer, the address is incremented by 1 after each byte transferred. The address bits (in address <11:0>) <11> indicates the most significant address bits. The address wraps around after reaching a boundary.

[0321] The data is in byte format, and by extending SPI_nCS to perform block transfers, all the bytes can be transferred in one packet.

[0322] Microprocessor Interrupts The ASIC has a VPAD logic level output UP_INT for sending an interrupt to the host microprocessor. The microprocessor interrupt module consists of an interrupt status register, an interrupt mask register, and a function that logically ORs all interrupt statuses into one microprocessor interrupt. Interrupts are implemented to support both edge- and level-sensitive styles. Interrupt polarity is programmable. The default interrupt polarity is TBD.

[0323] In the preferred embodiment, all interrupt sources on the AFE ASIC are recorded in an interrupt status register. Writing a '1' to the corresponding interrupt status bit clears the corresponding pending interrupt. All interrupt sources on the AFE ASIC are maskable via the interrupt mask register. Writing a '1' to the corresponding interrupt mask bit enables masking of the corresponding pending interrupt. Writing a '0' to the corresponding interrupt mask bit disables masking of the corresponding interrupt. The default state of the interrupt mask register is TBD.

[0324] General Purpose Input / Output (GPIO) / Parallel Test Port In an embodiment, the ASIC may have eight GPIOs that operate with VPAD level signals. The ASIC has one GPIO that operates with VBAT level signals and one GPIO that operates with VDD level signals. All of the GPIOs have at least the following characteristics: (i) Register bits control the selection and direction of each GPIO. (ii) The ASIC has a means to configure the GPIO as an input that can be read via the SPI interface. (iii) The ASIC has a means to configure the GPIO as an input for generating an interrupt. (iv) The ASIC has means to configure each GPIO as an output controlled by a register bit that can be written via the SPI interface. (v) Programmably, the ASIC can output an input signal applied to GPIO_VBAT or GPIO_VDD to a GPIO (on the VPAD power plane) (level shift function). (vi) The ASIC has means to configure each GPIO as an input to an oscillator calibration circuit. (vii) The ASIC has means for configuring the output of each general-purpose comparator to at least one GPIO on each power plane, the polarity of the comparator output being programmable by a programmable bit. (viii) GPIO has the ability to generate interrupts for the microprocessor. (ix) GPIOs are programmable for open-drain outputs. (x) GPIOs on the VPAD power plane can be configured to implement boot control of the microprocessor.

[0325] The parallel test port shares an 8-bit GPIO on the VPAD voltage plane. The test port is used to monitor register contents and various internal signals. The output of this port is controlled by the port configuration registers in normal mode. Writing 8'hFF to both the GPIO_O1S_REG and GPIO_O2S_REG registers will direct the test port data on the GPIO outputs, while writing 8'h00 to the GPIO_ON_REG register will disable the test port data and enable the GPIO data on the GPIO outputs.

[0326] Registers and pre-grouped internal signals can be monitored through this test port by addressing the target register through the SPI slave port. The SPI packet has the command bits set to 4'b0011, followed by the 12-bit target register address. The parallel test port continues to display the contents of the addressed register until the next test port addressing command is received.

[0327] Analog Test Port The IC has a multiplexer feeding pad TP_ANAMUX (4290), which exposes an internal analog circuit node for testing. The IC also has a multiplexer feeding pad TP_RES (4260), which exposes an internal analog circuit node for testing. This pad also accepts a precision 1 meg resistor for general use to perform various system calibrations.

[0328] Chip ID The ASIC contains a programmable ID, a 32-bit mask. A microprocessor using an SPI interface will be able to read this ID. This ID is intended to be located in the analog electronic block, so changing the ID does not require rerouting the chip. It should be designed so that changing the ID requires only one metal or contact mask change.

[0329] Preliminary Test Output The ASIC has 16 spare digital output signals that can be multiplexed onto the 8-bit GPIOs under command sent over the SPI interface. These signals are organized as two 8-bit bytes and are tied to VSS if not used.

[0330] Digital Testing The ASIC has a test mode controller that uses two input pins, TEST_CTL0 (4291) and TEST_CTL1 (4292). The test controller generates signals from a combination of test control signals (TEST_CTL<1:0>) with the following functionality: (i) 0 is the normal operating mode. (ii) 1 is the analog test mode. (iii) 2 is the scan mode. (iv) 3 is an analog test mode in which VDD_EN is controlled by the input to GPIO_VBAT.

[0331] The test controller logic is split across the VDD and VDDBU power planes. During scan mode, LT_VBAT can be asserted high to regulate the analog outputs to the digital logic. The ASIC has scan chains implemented in as much of the digital logic as reasonably possible for high-speed digital testing.

[0332] Leak Test Pin The ASIC has a pin called LT_VBAT which, when high, puts all analog blocks into an inactive mode so that only leakage current is drawn from the power supply. LT_VBAT forces all digital outputs from the analog blocks into a stable high or low state so as not to affect the current drain of the interface logic. The LT_VBAT pad is on the VBAT plane and has a pull-down resistor of 10k-40k ohms.

[0333] power requirements In the embodiment of the invention herein, the ASIC includes a low-power mode where, at a minimum, the microprocessor clock is off, the 32 kHz real-time clock is running, and circuitry is active to detect sensor connections, a change in level on the WAKE_UP pin, or a BREAK on the nRX_EXT input. This mode has a total current drain of up to 4.0 uA from VBAT (VDDBU), VDD, and VDDA. When the battery protection circuit detects a low battery (see Battery Protection Circuit Description), the ASIC transitions to a mode where only the VBAT and VDDBU power planes are active. This is called the low battery state. The VBAT current in this mode is less than 0.3 uA.

[0334] When the ASIC is programmed for a potentiostat configuration with one WORK electrode active in H2O2 (peroxide) mode with its voltage set to 1.535V, the COUNTER amplifier turned on with VSET_RE set to 1.00V, a 20 meg load resistor connected between WORK and COUNTER, and the COUNTER and RE connected together, assuming one WORK electrode current measurement per minute, the average current drain for all supplies is less than 7uA. After calibration, the measured current should be 26.75nA ±3%. Enabling additional WORK electrodes increases the combined current drain by less than 2uA at a WORK electrode current of 25nA.

[0335] When the ASIC is programmed for a potentiostat configuration with diagnostics enabled to measure the impedance of one of the WORK electrodes relative to the COUNTER electrode, the ASIC is configured to satisfy the following conditions: (i) Test frequencies: 0.1, 0.2, 0.3, 0.5 Hz, 1.0, 2.0, 5.0, 10, 100, 1000, 4000 Hz. (ii) The measurement of the above frequencies is not to exceed 50 seconds. (iii) The total charge delivered to the ASIC is less than 8 millicoulombs.

[0336] environment In a preferred embodiment of the present invention, the ASIC: (i) Operates and meets all specifications within the commercial temperature range of 0 to 70°C. (ii) It functions well at temperatures between -20°C and 80°C, but its accuracy may decrease at these temperatures. (ii...

Claims

1. 1. A glucose monitoring system comprising:

1. A glucose sensor device for determining a glucose concentration in a body of a user during a sensor device wearing time that includes a first time window, a subsequent second time window, and a transition period between the first time window and the second time window, the glucose sensor device comprising: a first glucose sensor, the first glucose sensor being a sensor suitable for use with an uncalibrated model that includes a calibration-free algorithm; a second glucose sensor, the second glucose sensor being a sensor suitable for use in a calibration model including a calibration algorithm using a reference glucose value, the first glucose sensor and the second glucose sensor having different characteristics in at least one of hydration, stabilization, and durability; A sensor electronic device, comprising: (a1) periodically receiving, from the first glucose sensor, a respective first output signal indicative of a glucose concentration level within the body of the user; (b1) periodically receiving, from the second glucose sensor, a respective second output signal indicative of a glucose concentration level within the body of the user; (c1) calculating a glucose concentration level in the user using the uncalibrated model based entirely on the first output signal during the first time window; (d1) during the transition period, calculating a first sensor glucose output using the uncalibrated model based on the first output signal; inputting the first sensor glucose output into the calibration model as the reference glucose value and calculating a second sensor glucose output using the calibration model based on the reference glucose value and the second output signal, where the calibration model calibrates the calibration model using the reference glucose value to calculate the second sensor glucose output; calculating a glucose concentration level within the user's body based on the first sensor glucose output and the second sensor glucose output; (e1) sensor electronics including at least one physical microprocessor configured to calculate a glucose concentration level in the user's body during the second time window using the calibration model calibrated entirely based on the second output signal without using the first sensor glucose output.

2. The glucose monitoring system of claim 1 , wherein the glucose sensor device is implanted or disposed subcutaneously within the user's body.

3. 2. The glucose monitoring system of claim 1, wherein the physical microprocessor determines start and end times for each of the first time window, the transition period, and the second time window based on the hydration, stabilization, and durability characteristics of the first glucose sensor and the second glucose sensor.

4. 4. The glucose monitoring system of claim 3, wherein during the transition period, the physical microprocessor periodically fuses the first sensor glucose output and the second sensor glucose output to calculate a single fused glucose value.

5. 2. The glucose monitoring system of claim 1, wherein during the transition period, the physical microprocessor compares the first output signal and the second output signal to diagnose whether each glucose sensor is functioning properly.

6. 6. The glucose monitoring system of claim 5, wherein based on the comparison and the diagnosis, the physical microprocessor assigns respective weights to the first sensor glucose output and the second sensor glucose output to generate respective weighted first and second signals.

7. 7. The glucose monitoring system of claim 6, wherein the physical microprocessor periodically calculates a single fused glucose value based on the weighted first and second signals.

8. The glucose monitoring system of claim 1 , further comprising a transmitter, said transmitter being worn on the user's body.

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