Wireless communication device and wireless communication method

Abstract

To make it possible to perform FFT window position control taking a front ghost wave into consideration even in a signal format with a sparse pilot density.SOLUTION: A radio communication device of this example includes: an FFT section 102 that performs FFT processing by setting an FFT window, which is a cut-out interval corresponding to an OFDM effective symbol length, to an OFDM signal; an FFT window position control section 104 that provides symbol timing detected from the OFDM signal to the FFT section 102 in order to control the position of the FFT window; and an optimal FFT window position search section 106 that applies a plurality of different phase rotations to an FFT-processed common signal to perform demodulation processing, identifies the number of phase rotations that minimizes a demodulation error, and calculates the amount of timing correction corresponding to the number of phase rotations. The FFT window position control section 104 is configured to correct the symbol timing according to the amount of timing correction and provide it to the FFT section 102.SELECTED DRAWING: Figure 1

Description

[Technical field]

[0001] The present invention relates to a wireless communication device that receives and processes OFDM signals. [Background technology]

[0002] In wireless communication systems using the Orthogonal Frequency Division Multiplexing (OFDM) method, FFT (Fast Fourier Transform) processing is required for demodulation processing. In FFT processing, a predetermined number of samples (effective symbol length) is cut out from the received signal. Hereinafter, the cut-out section of the effective OFDM symbol length is referred to as the "FFT window." Depending on the position of this FFT window, inter-symbol interference increases, and the spread of each point of the constellation becomes larger.

[0003] The importance of the FFT window will be explained below with reference to Figs. As shown in Fig. 2, the symbol length is the sum of the effective symbol length and the GI (guard interval) length. The GI is a copy of the latter half of the effective symbol and added to the beginning. As shown in Fig. 3(a), when the FFT window does not include the OFDM symbol boundary, highly accurate demodulation is possible. In contrast, as shown in Fig. 3(b), when the FFT window includes the OFDM symbol boundary, inter-symbol interference increases and the constellation spreads. Therefore, the FFT window needs to be controlled so that it does not include the OFDM symbol boundary.

[0004] The FFT window position is also important when demodulating received waves with different delay times with high accuracy. As shown in Figure 4, in addition to the received signal with the highest power (hereafter referred to as the "main wave"), radio waves with long delay times (hereafter referred to as the "late ghost wave") may be received. By adding GI, which is a feature of OFDM, the FFT window can be set so as not to include the symbol boundary even when late ghost waves are present. Therefore, in order to use GI most effectively against late ghost waves, it is desirable to set the FFT window from the beginning of the effective symbol of the received wave with the shortest delay time (the main wave in Figure 4).

[0005] As an example of a method for determining the FFT window position, a method using the correlation of GI will be explained. The waveform shown in Fig. 5 is obtained by taking the absolute value of the correlation between the received signal and a signal obtained by delaying the received signal by the effective symbol length, performing a moving average with the GI length, and then performing symbol averaging. Since the GI is a copy of a part of the effective symbol, the correlation level of the GI part is high. By performing the moving average, a waveform like that shown in Fig. 5 is obtained, and by using the peak position of this waveform to detect the OFDM symbol timing, the FFT window can be controlled to a desired position.

[0006] Prior art in the technical field of the present invention includes the following: For example, Patent Document 1 discloses an invention of a data transmission system that transmits data using frames in a format in which the symbol length of a data symbol is longer than the symbol length of a pilot symbol, and the guard interval ratio of a data symbol is smaller than the guard interval ratio of a pilot symbol. [Prior art documents] [Patent documents]

[0007] [Patent Document 1] Patent Publication No. 2021-005803 Summary of the Invention [Problem to be solved by the invention]

[0008] In the conventional symbol synchronization method described above, when a pre-ghost wave exists, it is difficult to control the FFT window to a position where no inter-symbol interference occurs. Here, a pre-ghost wave is defined as a radio wave that arrives earlier than the main wave. Below, two examples shown in Figures 6 and 7 are described as situations in which a pre-ghost wave can occur.

[0009] In Fig. 6(a), radio waves propagating directly from a transmitting station 601 to a receiving station 604 are blocked by a building 602, and the power is attenuated. In addition to this, radio waves reflected by a building 603 also arrive at the receiving station 604. In this case, the power of the reflected wave from the building 603 is greater than the power of the direct wave blocked by the building 602. Therefore, at the receiving station 604, the reflected wave from the building 603 becomes the main wave, and the direct wave blocked by the building 602 becomes the pre-ghost wave.

[0010] In FIG. 6(b), there is a propagation path that propagates directly from the transmitting station 601 to the receiving station 604, and a propagation path that arrives at the receiving station 604 via the relay station 605. The relay station 605 also amplifies the radio wave, and the transmission distance from the relay station 605 to the receiving station 604 is shorter than the transmission distance from the transmitting station 601 to the receiving station 604, and attenuation is small. Therefore, the received power at the receiving station 604 is higher for the radio wave received from the relay station 605 than for the radio wave received directly from the transmitting station 601. Also, the delay time of the radio wave is longer for the propagation path via the relay station 605 than for the propagation path that propagates directly from the transmitting station 601 to the receiving station 604, because the delay time at the relay station 605 is added. Therefore, in this case as well, the direct wave is observed as a pre-ghost wave with respect to the main wave (the radio wave from the relay station 605) at the receiving station 604.

[0011] In an environment where a front ghost wave occurs as described above, it is necessary to detect the symbol timing of the front ghost wave and control the FFT window position as shown in Fig. 7(a). This ensures that the symbol boundary is not included in the FFT window. On the other hand, as shown in Fig. 7(b), if the front ghost wave cannot be detected and the FFT window position is controlled based on the main wave, the symbol boundary of the front ghost wave will be included in the FFT window, resulting in inter-symbol interference. The method using the GI correlation described above has low timing detection accuracy, making it difficult to detect the symbol timing of the front ghost wave.

[0012] One of the methods for detecting the position of the ghost wave is a conventional technique that uses a delay profile. An example of a method for calculating a delay profile will be described below. When a radio wave with a different delay time from the main wave is mixed in, the main wave and the mixed radio wave are superimposed, and the received signal strengthens or weakens each other depending on the OFDM subcarrier frequency. The interval between the strengthening and weakening is proportional to the delay time. By converting the frequency characteristic due to this superposition into the time domain, it is possible to calculate a delay profile that separates the main wave and the mixed radio wave.

[0013] A known pseudo-random pattern is inserted at equal intervals into the subcarriers of OFDM. This carrier is called a "pilot carrier." At the receiving side, the propagation path characteristics of the pilot carrier can be obtained by extracting the pilot carrier and multiplying it by the inverse pattern of the pseudo-random pattern. In addition, the propagation path characteristics of the pilot carrier can be estimated for the entire band by interpolating the propagation path characteristics of the pilot carrier with a low-pass filter (hereinafter referred to as a "propagation path estimation filter"). The delay profile can be calculated by performing IFFT processing on the result of this propagation path estimation and converting it into the time domain. The previous ghost wave can be detected by observing this delay profile and detecting the earliest signal in time. Then, inter-symbol interference can be suppressed by controlling the FFT window position with the symbol timing of the previous ghost wave.

[0014] However, as shown in Fig. 8(a), the pilot carriers need to be arranged sufficiently densely in the frequency direction. Here, assuming the effective symbol length is T sym and the pilot interval is P, the time width T obs for which no image occurs and we want to observe can be expressed by the following equation (1).

[0015]

Equation

[0016] When the pilot interval P is large, that is, when the pilot carriers are sparse, T obs becomes a small value. In this case, as shown in Fig. 8(b), an image of the post ghost wave is generated near the main wave when interpolating the pilot carriers. For this reason, in the method of detecting the earliest signal in time, there is a problem that it is impossible to distinguish between the pre ghost wave and the post ghost wave, and since the FFT window position cannot be controlled at the symbol timing of the pre ghost wave, the inter symbol interference increases.

[0017] As an example, in the OFDM standards (ARIB STD - B71, STD - B57, STD - B33) of the FPU (Field Pickup Unit), the pilot carriers are set at a density of one per eight carriers. In this case, in the delay profile waveform obtained by performing IFFT on the propagation path estimation result estimated from the pilot carriers, an image occurs at a period of approximately 12.5 μs, which is 1 / 8 of the symbol length (about 100 μs). Depending on the frequency, since a post ghost wave of 12.5 μs or more may exist, there is a problem that it is impossible to distinguish between the pre ghost wave and the post ghost wave.

[0018] Regarding the prior art required to solve the problem of the pre ghost wave, the relationship between the FFT window position and the passband of the propagation path estimation filter will be described. As described above, when interpolating the pilot carriers in the frequency direction, an image occurs at the interval of T obs shown in equation (1). Therefore, the time passband width of the propagation path estimation filter is T obsThe time passband width must be set to exclude images, so that it is desirable to design the passband so that long-delay ghost waves are included as much as possible within this constraint.

[0019] When the FFT window is controlled to the beginning of the effective symbol, the phase rotation number when observed in the frequency direction of the pilot carrier is 0 rad / Hz, so the main wave component is located in the center of the passband of the channel estimation filter, as shown in Figure 9(a). When a late ghost wave is mixed in, the component occurs in the late time direction from the center of the filter. Therefore, the late ghost wave 2 that exists at a position beyond the passband of the filter is attenuated. Therefore, as shown in Figure 9(b), by multiplying the pilot carrier by the phase rotation component in the frequency domain and shifting the main wave to the early band edge of the filter, the long-delayed late ghost wave 2 can be placed within the passband of the filter. This makes it possible to effectively utilize the passband of the filter.

[0020] When performing the above processing, if a front ghost wave is present, the front ghost wave will be outside the band of the filter, as shown in Figure 9(c). As a result, the front ghost wave will be attenuated by the filter, and the propagation path estimation error will increase. In contrast, as shown in Figure 9(d), if the phase rotation number multiplied by the pilot carrier can be changed to bring the front ghost wave into the passband of the filter, the propagation path estimation error will decrease. However, as mentioned above, when the density of pilot carriers is sparse, it is not possible to distinguish between the images of the front ghost wave and the rear ghost wave, and therefore it is not possible to determine the optimal phase rotation number.

[0021] The present invention has been made in consideration of the above-mentioned conventional circumstances, and aims to make it possible to control the FFT window position taking into account pre-ghost waves, even in a signal format with a sparse pilot carrier density. [Means for solving the problem]

[0022] In order to achieve the above object, a wireless communication device according to an aspect of the present invention is configured as follows: That is, the wireless communication device according to the present invention is a wireless communication device that receives and processes an OFDM signal, and includes an FFT unit that sets an FFT window, which is an extraction section having an effective symbol length of OFDM, for the OFDM signal and performs FFT processing, an FFT window position control unit that provides the FFT unit with symbol timing detected from the OFDM signal in order to control the position of the FFT window, and an optimal FFT window position search unit that provides a plurality of different phase rotations to a common signal that has been FFT processed, performs demodulation processing, identifies a phase rotation number that minimizes demodulation error, and calculates a timing correction amount corresponding to the phase rotation number, and is characterized in that the FFT window position control unit corrects the symbol timing according to the timing correction amount and provides the symbol timing to the FFT unit.

[0023] Here, in the wireless communication device according to the present invention, the optimal FFT window position search unit can be configured to perform demodulation processing each time a predetermined timing elapses by applying a phase rotation corresponding to the timing to the common signal.

[0024] Furthermore, in the wireless communication device according to the present invention, the optimal FFT window position search unit can be configured to have a memory for storing one symbol's worth of a signal that has been FFT processed, and to perform demodulation processing each time a predetermined timing elapses by applying a phase rotation corresponding to the timing to the signal stored in the memory, and to update the contents of the memory after a number of timings equal to the number of types of phase rotation have elapsed.

[0025] Also, in the wireless communication device according to the present invention, the optimum FFT window position search unit has a plurality of processing units that perform demodulation processing by applying different phase rotations to a common signal, and operates the plurality of processing units in parallel.

[0026] Also, a wireless communication method according to another aspect of the present invention is configured as follows: That is, the wireless communication method receives and processes an OFDM signal, and is characterized by comprising the steps of: setting an FFT window, which is an extraction section having an OFDM effective symbol length, for the OFDM signal and performing FFT processing; applying a plurality of different phase rotations to the FFT-processed signal and performing demodulation processing, identifying a phase rotation number that minimizes demodulation error, and calculating a timing correction amount corresponding to the phase rotation number; and correcting symbol timing detected from the OFDM signal in accordance with the timing correction amount, and controlling the position of the FFT window based on the corrected symbol timing. Effect of the Invention

[0027] According to the present invention, even in a signal format in which the density of pilot carriers is sparse, it is possible to control the FFT window position taking into account the pre-ghost wave. [Brief description of the drawings]

[0028] [Figure 1] 1 is a diagram illustrating an example of the configuration of a wireless communication device according to an embodiment of the present invention. [Diagram 2] FIG. 2 is a diagram illustrating the symbol length of an OFDM signal. [Diagram 3] FIG. 1A is a diagram showing an example of a constellation when the FFT window does not include an OFDM symbol boundary, and FIG. 1B is a diagram showing an example of a constellation when the FFT window includes an OFDM symbol boundary. [Figure 4] FIG. 1 is a diagram for explaining a trailing ghost wave that arrives with a delay from the main wave. [Diagram 5] FIG. 1 is a diagram for explaining GI correlation. [Figure 6] FIG. 1A is a diagram for explaining the generation of a pre-ghost wave due to an obstruction, and FIG. 1B is a diagram for explaining the generation of a pre-ghost wave due to a relay station. [Figure 7](a) is a diagram showing the FFT window when the pre-ghost wave has been detected, and (b) is a diagram showing the FFT window when the pre-ghost wave has not been detected. [Figure 8] (a) is a diagram showing an example of a delay profile when pilot carriers are sufficiently dense, and (b) is a diagram showing an example of a delay profile when pilot carriers are sparse. [Figure 9] (a) is a diagram showing an example of a delay profile before the phase rotation of pilot carriers, (b) is a diagram showing an example of a delay profile after the phase rotation of pilot carriers, (c) is a diagram showing an example of a delay profile when phase rotation is performed without detecting the pre-ghost wave, and (d) is a diagram showing an example of a delay profile when phase rotation is performed after detecting the pre-ghost wave.

Embodiments for Carrying Out the Invention

[0029] An embodiment of the present invention will be described with reference to the drawings. FIG. 1 shows a configuration example of a wireless communication device according to an embodiment of the present invention. The wireless communication device in FIG. 1 includes an ADC 101, an FFT unit 102, a symbol synchronization unit 103, an FFT window position control unit 104, a main line reception processing unit 105, and an optimal FFT window position search unit 106. The main line reception processing unit 105 includes a fixed phase rotation unit 107, an equalization unit 108, a pilot extraction unit 109, an interpolation filter 110, and a decoding unit 111. The optimal FFT window position search unit 106 includes a scan control counter 112, a one-symbol memory 113, a phase rotation scan unit 114, an equalization unit 108, a pilot extraction unit 109, an interpolation filter 110, an MER calculation unit 115, an optimal rotation number calculation unit 116, and a correction value calculation unit 117.

[0030] The ADC 101 converts the received analog signal into a digital signal and outputs it to the symbol synchronization unit 103 and the FFT unit 102. The symbol synchronization unit 103 detects the symbol timing W of the OFDM signal by using the correlation of the GI described above. The timing signal detected by the symbol synchronization unit 103 is input to the FFT window position control unit 104. The FFT window position control unit 104 calculates the symbol timing W based on the correction value W input from the correction value calculation unit 117 described later. comp In accordance with (S), the timing signal W' obtained by correcting the symbol timing W is output to the FFT unit 102. Here, the corrected timing W' is expressed by the following equation (2).

[0031]

number

[0032] In the FFT unit 102, an FFT window is set for the signal input from the ADC 101 based on a timing signal W' input from an FFT window position control unit 104, and the signal is converted to a frequency domain signal F(ω) by FFT processing, where ω indicates a subcarrier number in the frequency domain. The FFT processed signal is output to a main line reception processing unit 105 which performs demodulation and decoding, and to an optimal FFT window position search unit 106 which searches for an optimal window position.

[0033] In the following, first the main line reception processing unit 105 which is responsible for main line reception processing will be described, and then the optimum FFT window position searching unit 106 which searches for the optimum window position will be described.

[0034] In the main line reception processing unit 105, the signal that has been subjected to the FFT processing is input to the fixed phase rotation unit 107. In the fixed phase rotation unit 107, as shown in the following equation (3), a fixed phase rotation signal e j2πωRfix / NFFT Multiply by N FFT indicates the number of points for FFT processing. Also, the fixed rotation number R fix is the phase rotation signal N FFT It indicates the number of revolutions in the sample, which does not have to be an integer.

[0035]

number

[0036] 9, this process changes the number of phase rotations in the frequency direction of the pilot carriers, and shifts them to the band edge of the passband of the interpolation filter 110, which will be described later. This allows the interpolation filter to be used efficiently when a post-ghost wave is mixed in.

[0037] Fixed phase rotation signal e j2πωRfix / NFFT The signal F'(ω) multiplied by is output to equalization section 108 and pilot extraction section 109. Pilot extraction section 109 extracts pilot carriers from the input signal and outputs them to interpolation filter 110. Interpolation filter 110 performs filtering to interpolate pilot carriers in the frequency direction, thereby making it possible to estimate the frequency characteristics of subcarriers where no pilot carriers exist.

[0038] The frequency-interpolated pilot signal (hereinafter referred to as the "propagation path estimation result") is output to an equalization unit 108. The equalization unit 108 performs equalization processing using the signal F'(ω) input from the fixed phase rotation unit 107 and the propagation path estimation result input from the interpolation filter 110, and outputs the equalization result to a decoding unit 111. The decoding unit 111 performs error correction processing and the like, and outputs the decoded result as the result of wireless transmission. This completes the flow of the main line reception processing.

[0039] Next, the process of the optimum FFT window position search unit 106 for controlling the FFT window position with high precision will be described. The optimum FFT window position search unit 106 performs demodulation processing by applying a plurality of different phase rotations to the signal for one symbol, searches for the phase rotation number that results in the highest demodulation quality, and determines the correction value W of the FFT window position based on the result. compIn this example, the current FFT window position is controlled using past received signals, so it is assumed that any fluctuations in the propagation path during that time can be ignored. However, with recent advances in signal processing devices, it is now possible to control the FFT window position in real time by increasing the amount of calculations per unit time, in which case it is possible to handle even large fluctuations in the propagation path.

[0040] The received signal F(ω) that has been FFT processed by the FFT unit 102 is input to a scan control counter 112 and a one-symbol memory 113 . Scan control counter 112 is a counter that counts up every time one symbol's worth of data is input, and when it reaches its maximum value, it counts again from 0. Count value c by scan control counter 112 is output to one symbol memory 113, phase rotation scanner 114, and optimum rotation number calculator 116. The maximum count value is set to the same number as the number of types of phase rotation numbers performed by phase rotation scanner 114, which will be described later.

[0041] In 1 symbol memory 113, when count value c input from scan control counter 112 is 0, the signal input from FFT unit 102 is written into the memory for one symbol, and when count value c is other than 0, no new signal is written. Reading from 1 symbol memory 113 is performed for each count, and the signal read from 1 symbol memory 113 is input to phase rotation scan unit 114.

[0042] The phase rotation scanning unit 114 changes the phase rotation number R(c) for each count input from the scan control counter 112, and generates a phase rotation signal e corresponding to R(c) for the input signal. j2πωR(c) / NFFT to calculate the phase-rotated signal F″(ω) (see equation (4)).

[0043]

number

[0044] This process makes it possible to change the position of the main wave with respect to the interpolation filter 110. The phase-rotated signal F"(ω) is subjected to the same processing as that for the main line described above by the pilot extraction unit 109, the interpolation filter 110, and the equalization unit 108.

[0045] The signal after equalization by the equalization unit 108 is input to the MER calculation unit 115. The MER calculation unit 115 calculates the MER (Modulation Error Ratio) based on the input signal. The MER is an index often used to judge the quality of a transmission or reception signal. Here, the ratio of the error power between the reception point after equalization and the ideal reception point on the receiving side and the average power of the signal is calculated, and this ratio is averaged over all carriers to calculate. In this case, the MER is maximum when the result of the equalized signal is closest to the ideal reception point. Here, the MER is adopted as a method for calculating the signal quality after equalization, but a method for measuring the signal quality may be reused using other values ​​such as EVM (Error Vector Magnitude) and mutual information.

[0046] The MER calculated by the MER calculation unit 115 is input to the optimum rotation speed calculation unit 116. The optimum rotation speed calculation unit 116 calculates the phase rotation speed R at which the MER becomes maximum while the count value c input from the scan control counter 112 changes from 0 to the maximum value. opt and outputs it to the correction value calculation unit 117.

[0047] The correction value calculation unit 117 calculates the phase rotation number R opt Using this, the correction value W for the symbol timing W is comp (S) is calculated, where S indicates the scan number. Correction value W comp (S) and MER maximum phase rotation R opt The relationship with (S) is expressed by the following equation (5).

[0048]

number

[0049] There is a one-to-one relationship between the FFT window position in the time domain and the phase rotation number in the frequency domain. In the optimal FFT window position search unit 106, the FFT window position W′ is fixed and the optimal phase rotation number R opt Therefore, at this stage, the FFT window position is not the optimal position, so the fixed rotation number R fix From the optimal phase rotation number R opt The integer part of the value obtained by subtracting comp Add this to (S-1) to get the correction value W comp Calculate (S).

[0050] As described above, the FFT window position control unit 104 uses the correction value W comp The FFT window position is corrected using (S). The above process makes it possible to control the FFT window position taking into account the pre-ghost wave, and it is possible to suppress the occurrence of inter-symbol interference even in a reception environment where a pre-ghost wave occurs.

[0051] As described above, the wireless communication device of this example comprises an FFT unit 102 that performs FFT processing on the OFDM signal by setting an FFT window, which is an extraction section having the effective symbol length of OFDM, an FFT window position control unit 104 that provides symbol timing detected from the OFDM signal to the FFT unit 102 in order to control the position of the FFT window, and an optimal FFT window position search unit 106 that performs demodulation processing by applying a plurality of different phase rotations to the FFT-processed common signal, identifies the number of phase rotations that minimizes the demodulation error, and calculates the amount of timing correction corresponding to that number of phase rotations, and the FFT window position control unit 104 is configured to correct the symbol timing in accordance with the amount of timing correction and provide it to the FFT unit 102.

[0052] With this configuration, it is possible to search for the optimal FFT window position taking into account the front ghost wave even in a signal format with a sparse pilot carrier density. In other words, by testing multiple positional relationships of the main wave, front ghost wave, and rear ghost wave with respect to the passband of the propagation path estimation filter, the optimal FFT window position can be calculated from the phase rotation number that minimizes the demodulation error. This makes it possible to control the FFT window position taking into account the front ghost, and to perform demodulation processing that suppresses the occurrence of inter-symbol interference, thereby improving reception performance. In addition, since the optimal FFT window position can be searched for using a signal with a determined FFT window position, it is possible to search for the optimal FFT processing window position separately from the main line reception processing.

[0053] Here, optimal FFT window position search section 106 in this example has a one-symbol memory 113 that stores one symbol's worth of signal that has been FFT-processed, and each time one symbol's worth of timing has passed, demodulation processing is performed by applying a phase rotation corresponding to that timing to the signal stored in one-symbol memory 113, and after a number of timings equal to the number of types of phase rotation have passed, the contents of one-symbol memory 113 are updated. With this configuration, it is possible to search for an optimal FFT window position in a time period equal to the number of symbol timings equal to the number of types of phase rotation.

[0054] As a first modification, the components of the optimum FFT window position search unit 106 (for example, the phase rotation unit 14, the pilot extraction unit 109, the interpolation filter 110, the equalization unit 108, and the MER calculation unit 115) may be made more powerful so that the demodulation process with different phase rotations can be performed at a shorter timing. This makes it possible to complete the search for the optimum FFT window position in a shorter time.

[0055] As a second modification, a plurality of processing units (e.g., phase rotation unit 14, pilot extraction unit 109, interpolation filter 110, equalization unit 108, and MER calculation unit 115) may be provided to perform demodulation processing by applying different phase rotations to the signal stored in one-symbol memory 113, and these processing units may be operated in parallel. This configuration also makes it possible to complete the search for the optimal FFT window position in a shorter time. It is also possible to combine the first and second modifications.

[0056] Although the embodiments of the present invention have been described above, these embodiments are merely illustrative and do not limit the technical scope of the present invention. The present invention can take various other embodiments, and various modifications such as omissions and substitutions can be made without departing from the gist of the present invention. These embodiments and modifications are included in the scope and gist of the invention described in this specification, etc., and are included in the scope of the invention described in the claims and their equivalents.

[0057] Furthermore, the present invention can be provided not only as devices such as those described above or as a system composed of these devices, but also as methods executed by these devices, programs for causing a processor to realize the functions of these devices, and storage media for storing such programs in a computer-readable format. [Industrial Applicability]

[0058] The present invention can be used in a wireless communication device that receives and processes OFDM signals. [Explanation of symbols]

[0059] 101: ADC, 102: FFT section, 103: symbol synchronization section, 104: FFT window position control section, 105: main line reception processing section, 106: optimum FFT window position search section, 107: fixed phase rotation section, 108: equalization section, 109: pilot extraction section, 110: interpolation filter, 111: decoding section, 112: scan control counter, 113: 1 symbol memory, 114: phase rotation scanning section, 115: MER calculation section, 116: optimum rotation number calculation section, 117: correction value calculation section

Claims

1. 1. A wireless communication device for receiving and processing an OFDM signal, comprising: an FFT unit that performs FFT processing on the OFDM signal by setting an FFT window that is an extraction section having an effective symbol length of OFDM; an FFT window position control unit that supplies a symbol timing detected from the OFDM signal to the FFT unit in order to control a position of the FFT window; a main line reception processing unit which performs demodulation processing by applying a phase rotation of a fixed number of rotations to the FFT-processed signal; an optimal FFT window position search unit that performs demodulation processing by applying phase rotation of a plurality of different rotation numbers to a common signal for one symbol in the FFT processed signal, identifies an optimal rotation number that minimizes a demodulation error, and calculates a timing correction amount corresponding to the optimal rotation number based on the fixed rotation number, the optimal rotation number, and the previous timing correction amount, separate from the main line reception processing unit; The wireless communication device, wherein the FFT window position control section corrects the symbol timing in accordance with the timing correction amount and provides the corrected symbol timing to the FFT section.

2. 2. The wireless communication device according to claim 1, The wireless communication device, wherein the optimal FFT window position search unit performs demodulation processing by applying a phase rotation of a number of rotations corresponding to a predetermined timing to the common signal each time the predetermined timing elapses.

3. 3. The wireless communication device according to claim 2, the optimal FFT window position search unit has a memory for storing the one symbol's worth of signal that has been subjected to the FFT processing, and each time a predetermined timing elapses, performs a demodulation process on the signal stored in the memory by applying a phase rotation of a number of rotations corresponding to the timing, and updates the contents of the memory after a number of timings equal to the number of types of phase rotation rotations have elapsed.

4. 4. The wireless communication device according to claim 1, The wireless communication device, characterized in that the optimal FFT window position search unit has a plurality of processing units that perform demodulation processing by applying phase rotation of different rotation numbers to the common signal, and the plurality of processing units are operated in parallel.

5. 1. A wireless communication method for receiving and processing an OFDM signal, comprising: a step of performing an FFT process on the OFDM signal by setting an FFT window which is an extraction section having an effective symbol length of OFDM; a step of performing a demodulation process by applying a phase rotation of a fixed number of rotations to the FFT-processed signal, separately from a main line reception process in which the FFT-processed signal is subjected to a phase rotation of a plurality of different number of rotations to a common signal of one symbol in the FFT-processed signal, thereby specifying an optimal number of rotations at which demodulation error is minimized, and calculating a timing correction amount corresponding to the optimal number of rotations based on the fixed number of rotations, the optimal number of rotations, and the previous timing correction amount; correcting symbol timing detected from the OFDM signal in accordance with the timing correction amount, and controlling a position of the FFT window based on the corrected symbol timing.

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