Base station device, weight generation method, and wireless communication system

Abstract

To effectively perform an interference rejection in a low operation amount.SOLUTION: A base station device in a radio communications system that a base station device having a plurality of antenna elements and a plurality of terminal devices perform a spatial multiplex transmission at the same time on the same frequency, calculates a weight vector that is for performing a spatial multiplex transmission to the plurality of terminal devices on the basis of a channel matrix in which a channel vector generated by channel information between a virtual antenna element obtained by synthesizing the antenna element of a terminal device or the antenna element, and the antenna element provided by the base station device and an additional channel vector related to the antenna element of the terminal device that is different from the channel vector are arranged, and calculates a weight vector by selecting a second channel vector of which an alternative similarity is lower than that of a first channel vector from one or a plurality of second channel vectors, acquired before the first channel vector as the additional channel vector.SELECTED DRAWING: Figure 12

Description

[Technical Field]

[0001] The present invention relates to a base station apparatus, a weight generation method, and a wireless communication system. [Background technology]

[0002] To meet the ever-increasing demand for wireless communications, progress is being made in implementing multi-user MIMO (Multiple-Input Multiple-Output) technology, which multiplexes and transmits signals for multiple terminals over the same space, time, and frequency (see, for example, Non-Patent Document 1). In multi-user MIMO, it is assumed that the number of antenna elements at a terminal is smaller than the number of antenna elements at a base station. In this case, since it is difficult for the terminal to separate signals for multiple terminals, it is common for the base station to perform interference suppression processing (hereinafter referred to as "null formation") to prevent interference between multiple terminals before transmitting signals.

[0003] To perform this null formation, it is necessary to accurately grasp the state of the propagation path between the antenna element of the base station that transmits the signal and the antenna element of the terminal that receives the signal. For this purpose, propagation path estimation (hereinafter also referred to as "channel estimation") is required, in which a known signal is transmitted in advance between the transmitter and receiver, and the state of the propagation path is estimated based on information on how much the signal has changed. Since this propagation path estimation requires the transmission of a known signal, it results in communication overhead. Therefore, it is generally performed discretely when communication is required or at regular intervals specified by communication standards.

[0004] As the frequency bands used in wireless communications expand, studies are also underway on Massive MIMO technology, which uses multiple antenna elements for transmitting and receiving antennas, in order to compensate for the increased noise power that accompanies increased distance attenuation and wideband transmission (see, for example, Non-Patent Document 2). Massive MIMO has a high degree of antenna freedom, as it uses multiple antenna elements, enabling more flexible signal control than conventional methods.

[0005] On the other hand, wireless communication propagation paths change constantly due to terminal movement and changes in the surrounding environment. Therefore, if the propagation path acquired by propagation path estimation changes from the propagation path actually used for communication, the effectiveness of null formation weakens, resulting in inter-user interference and degrading transmission characteristics. Therefore, focusing on the high antenna flexibility of Massive MIMO, null space extension technology has been studied, which suppresses inter-user interference by forming multiple nulls per terminal antenna, instead of the conventional spatial multiplexing transmission in which one null is formed per terminal antenna (see, for example, Non-Patent Documents 3 and 4 and Patent Document 1). In the conventional null space extension technology, after the minimum necessary null formation is performed, the remaining antenna freedom is used to obtain the gain of the desired signal. However, by performing additional null formation, it is possible to realize multi-user MIMO transmission that is robust against changes in propagation paths.

[0006] [About Multi-User MIMO Technology] (Overview of Multi-User MIMO) Coherent transmission and phased array antenna technology are basically technologies for improving line gain, but in order to increase the line capacity when a wide service area is covered by a single base station device, another wireless communication technology is required. On the other hand, since frequency resources are limited, here we will explain, for example, the multi-user MIMO technology discussed in Non-Patent Document 1 as a technology for using limited resources with high frequency utilization efficiency.

[0007] 2 is a schematic diagram showing an example of the configuration of a multi-user MIMO system. As shown in the figure, the multi-user MIMO system includes a base station device 801 and terminal devices 802-1, 802-2, and 802-3 (terminal devices #1 to #3). In reality, one base station device 801 accommodates a large number of terminal devices 802, but several of these are selected (terminal devices 802-1 to 802-3 in the figure) for communication. Each terminal device 802 generally has fewer transmitting and receiving antennas than the base station device 801. Below, a case where communication (downlink) from the base station device 801 to the terminal device 802 is described.

[0008] The base station device 801 forms multiple directional beams using a large number of antenna elements. For example, consider a case where three MIMO channels are assigned to each of the terminal devices 802-1 to 802-3, and a total of nine signal sequences are transmitted. In this case, the signal transmitted to the terminal device 802-1 is adjusted so that the directional gain is extremely low (so that nulls are formed) in the directions of the terminal devices 802-2 and 802-3, thereby suppressing interference to the terminal devices 802-2 and 802-3. Similarly, the signal transmitted to the terminal device 802-2 is adjusted so that the directional gain is extremely low in the directions of the terminal devices 802-1 and 802-3. Similar processing is performed on the terminal device 802-3. The reason for performing directivity control in this manner is that, for example, terminal device 802-1 has no way of knowing information about signals received by terminal device 802-2 and terminal device 802-3, and therefore cooperative reception processing between the terminal devices 802 is not possible; in other words, it is extremely difficult to separate all nine signal sequences in the reception processing of only terminal device 802-1, which has only three antennas. Therefore, interference separation is performed in advance on the transmitting side so that each of terminal devices 802-1 to 802-3 does not receive signals from other terminal devices 802. This concludes the overview of existing multi-user MIMO systems.

[0009] Next, a method for forming a directional beam will be described below. Here, a case will be described in which the base station device 801 has nine antenna elements and each of the terminal devices 802-1 to 802-3 has three antenna elements. For example, in FIG. 2, the channel information between the jth (j=1, ..., 9) antenna element of the base station device 801 and the first antenna element of the terminal device 802-1 is expressed as h 1j Using channel information between each antenna element (j=1, . . . , 9) of the base station device 801 and the first antenna element of the terminal device 802-1, the row vector h1 is expressed as (h 11 ,h 12 ,h 13 ,…,h 18 ,h 19Similarly, the channel information between the j-th antenna element of the base station device 801 and the second and third antenna elements of the terminal device 802-1 is expressed as h 2j and h 3j and the corresponding row vectors h2 and h3 are written as (h 21 ,h 22 ,h 23 ,…,h 28 ,h 29 ) and (h 31 ,h 32 ,h 33 ,…,h 38 ,h 39 ) The antenna elements of the terminal device 802-2 and the terminal device 802-3 are assigned the same consecutive numbers, and the row vectors h4 to h9 are expressed as (h 41 ,h 42 ,h 43 ,…,h 48 ,h 49 )~(h 91 ,h 92 ,h 93 ,…,h 98 ,h 99 ) is written as

[0010] Channel information is a value that represents the state of the wireless communication propagation path, i.e., the channel, which is the actual propagation coefficient between the transmitting antenna and the receiving antenna, estimated or acquired by the transmitter or receiver (strictly speaking, it includes the effects of amplifiers and filters within the transmitter and receiver).

[0011] In addition, the nine signals transmitted by the base station device 801 are represented as t1 to t9, and a column vector having these as components is denoted as Tx [all] =(t1,t2,t3,…,t8,t9) T Here, the letter T on the right hand side represents the transposition of a vector or matrix. Similarly, the received signals at the nine antenna elements of the terminal devices 802-1 to 802-3 are represented as r1 to r9, and a column vector having these as components is represented as Rx [all] =(r1,r2,r3,…,r8,r9) T Finally, the matrix with row vectors h1 to h9 as the first to ninth row components is defined as the overall channel matrix H [all]Also, noise is written as n.

[0012] In this case, the relationship of the following equation (1) holds for the entire multi-user MIMO system.

[0013]

number

[0014] In order to control the transmission directivity, a 9-row, 9-column transmission weight matrix W is introduced, and equation (1) is rewritten as equation (2) below.

[0015]

number

[0016] Furthermore, if the transmission weight matrix W is decomposed into column vectors w1 to w9 and written as W = (w1, w2, w3, ..., w8, w9), then "H [all] ·W" can be expressed as the following equation (3).

[0017]

number

[0018] Here, for example, consider selecting values ​​of w1 to w3 so that the multiplication of six row vectors h4 to h9 by three column vectors w1 to w3 (the sum of the products of each component; in the case of complex vectors, this is different from the dot product) is all zero. At the same time, select values ​​of w4 to w9 so that the multiplication of row vectors h1 to h3 and h7 to h9 by column vectors w4 to w6, and the multiplication of row vectors h1 to h6 by column vectors w7 to w9 are all zero.

[0019] Then, the 9-row, 9-column matrix H shown in equation (3) [all] ·W can be expressed as the following equation (4) using a submatrix of 3 rows and 3 columns.

[0020]

number

[0021] In equation (4), H [1] , H [2] , and H [3] is a 3-row, 3-column matrix, and "0" is a 3-row, 3-column matrix whose elements are all zero. By selecting a transformation matrix that satisfies these conditions as the transmission weight matrix W, equation (4) can be decomposed into the three relational equations expressed by the following equations (5-1) to (5-3).

[0022]

number

[0023] where Tx [1] =(t1,t2,t3) T , Tx [2] =(t4,t5,t6) T , Tx [3] =(t7,t8,t9) T , Rx [1] =(r1,r2,r3) T , Rx [2] =(r4,r5,r6) T , Rx [3] =(r7,r8,r9) T In this way, one base station device can be considered to be performing one-to-one MIMO communication, or so-called single-user MIMO communication, simultaneously and in parallel on three systems.

[0024] Next, an example of a method for determining transmission weight vectors w1 to w9 will be described below. The procedure is to determine transmission weight vectors w1 to w3 for terminal device 802-1, then determine transmission weight vectors w4 to w6 for terminal device 802-2, and transmission weight vectors w7 to w9 for terminal device 802-3.

[0025] First, as a first step, six basis vectors e4 to e9 in a six-dimensional subspace spanned by six row vectors h4 to h9 for terminal devices 802-2 and 802-3 are obtained. There are various methods for obtaining these vectors, including Gram-Schmidt orthogonalization, but here we will explain the Gram-Schmidt orthogonalization method as an example. First, we focus on one row vector h4, and define a vector in this direction with an absolute value of 1 as a basis vector e4. The basis vector e4 is expressed as the following equation (6).

[0026]

number

[0027] (h4h4 H ) is a scalar quantity that means the square of the absolute value of the same vector, and division by the square root of this value means normalizing the row vector h4. Also, "h4 H " is a Hermitian conjugate vector for the row vector h4, and is a vector obtained by transposing the rows and columns and taking the complex conjugate of each component.

[0028] Next, we focus on row vector h5, and calculate row vector h5' by canceling the component in the direction of basis vector e4 from this row vector, and then normalize it. Row vector h5' and basis vector e5 are expressed by the following equations (7-1) and (7-2).

[0029]

number

[0030] (h5e4 H ) means the projection of row vector h5 in the direction of basis vector e4. Similar processing is performed as in the following equations (8-1) and (8-2).

[0031]

number

[0032] Here, the range of the sum of Σ in equation (8-1) is the sum for integer i in the range 4≦i≦(j-1) (j is an integer between 5 and 9). In other words, this means canceling out the component in the direction of the already determined specified vector. In this way, the six basis vectors e4 to e9 can be determined.

[0033] Next, in the second step, transmission weight vectors w1 to w3 for terminal device 802-1 are found. First, the components of the six-dimensional subspace spanned by basis vectors e4 to e9 are cancelled from row vectors h1 to h3. Specifically, this is expressed by the following equation (9).

[0034]

number

[0035] Here, j in equation (9) is an integer between 1 and 3, and the range of the sum of Σ is the sum for integer i in the range 4≦i≦9. The three-dimensional space spanned by the three row vectors h1' to h3' thus obtained is orthogonal to all of the row vectors h4 to h9 described above. By selecting three vectors (which do not necessarily have to be orthogonal vectors) within this three-dimensional space and setting the complex conjugate vectors of these vectors as transmission weight vectors w1 to w3, it is possible to suppress interference with other terminal devices 802-2 and 802-3.

[0036] Note that the three vectors can be selected in any manner, but for example, if three orthogonal vectors that form a unitary matrix obtained by performing singular value decomposition are used, eigenmode transmission limited to a subspace that does not cause interference to other terminal devices 802 becomes possible, enabling efficient transmission.

[0037] Finally, in the third step, if the same processing is performed for the terminal device 802-2 and the terminal device 802-3, the overall transmission weight vectors w1 to w9 can be finally determined.

[0038] The above is an example of how to find the transmission weight matrix W.

[0039] FIG. 3 is a flowchart showing an example of a procedure for calculating a transmission weight matrix W in a multi-user MIMO system. First, in calculating the transmission weight matrix W, a channel matrix H for all terminal devices 802 to be multiplexed is acquired (step S801). If a serial number is assigned to the destination terminal device 802 and a variable indicating the serial number is k, k is first initialized (step S802). Furthermore, k is counted up (step S803), and a partial channel matrix H for the terminal device 802 (#k) corresponding to the current value indicated by k (here, for convenience, H main ) is extracted (step S804), and the partial channel matrix (H sub ) is extracted (step S805).

[0040] Furthermore, the partial channel matrix H sub The orthogonal basis vectors of the subspace spanned by each row vector of are calculated, and these are called basis vectors {e j} (step S806). Next, as a process corresponding to equation (9), the partial channel matrix H main The basis vector {e j} and this is the matrix H main (Step S807). Here, in step S807, H with a "~ (tilde)" above it is written as "~H". Similarly, in the following, when writing a character with a symbol above it such as "^ (hat)" in mathematical expressions, the symbol is written before the character.

[0041] Furthermore, the matrix ~H main Calculate any orthogonal basis vector of the subspace spanned by the row vector of k} (step S808). Here, the arbitrary basis vector is, for example, a matrix H mainYou can also select the vectors that make up the right singular matrix when singular value decomposition is performed. Then, k}, the transmission weight vector {w k} is determined (step S809).

[0042] Here, it is determined whether the transmission weight vectors of all the destination terminal devices 802 have been determined (step S810), and if there are any remaining terminal devices 802, the processing from step S803 to step S809 is repeated. If the transmission weight vectors of all the terminal devices 802 have been determined, the transmission weight vector {w k} as each column vector (step S811), and the process ends.

[0043] In addition, since channel information generally differs for each frequency component, in the case of a wideband signal, for example, a signal using an OFDM (Orthogonal Frequency Division Multiplexing) modulation method, a similar transmission weight is calculated for each frequency component, i.e., for each subcarrier. In addition, since the case where each of the terminal devices 802-1 to 802-3 has three antenna elements has been described here, the matrix ~H main However, if the terminal device has only one antenna, step S808 simply calculates the orthogonal basis vectors of the subspace spanned by the matrix ∼H main This corresponds to normalizing the row vector corresponding to

[0044] The above is a method for calculating the transmission and reception weights of a general multi-user MIMO, assuming that the terminal device has multiple antennas, and is a method of block diagonalizing the overall channel matrix as shown in Equation (4). However, as other methods for calculating the same transmission and reception weights, there are also several other variations. These variations do not necessarily require the terminal device to have only one antenna, but for simplicity in the following explanation, it is assumed that N terminal devices each with one antenna perform spatial multiplexing simultaneously. The following is an explanation of other methods for calculating the transmission and reception weights.

[0045] First, regarding the transmission and reception weights of the base station device 801, for the overall channel matrix H shown in Equation (1) etc., [all] a ZF (Zero Forcing) type pseudo-inverse matrix represented by the following Equations (10-1) and (10-2) may be calculated and used as the transmission weight and reception weight.

[0046]

Equation

[0047] Here, assuming the number of terminal devices performing spatial multiplexing is N and the number of antenna elements of the base station device 801 is K (N < K), for example, taking the downlink as an example, the size of the overall channel matrix H [all] is N × K (N rows and K columns). If the rank of H [all] is N, the size of the matrix H [all] ·H [all]H is N × N and its inverse matrix exists, and a pseudo-inverse matrix can be obtained using Equation (10-1). Generally, if the value of K is sufficiently redundant with respect to N, the rank of this N × N matrix will be stably N and the inverse matrix will stably exist. Similarly, regarding the reception weight of the uplink corresponding to the reception of the base station device 801, the size of the overall channel matrix H [all] is K × N (K rows and N columns), and the matrix H [all]H ·H [all]The size of is also N×N, and generally an inverse matrix exists. A ZF type pseudo-inverse matrix expressed by the following equation (10-2) may be calculated and used as the receiving weight.

[0048] In addition, in the MMSE (Minimum Mean Square Error) weight, which is known as a similar transmitting and receiving weight, the noise power is σ 2 Then, the following equations (11-1) and (11-2) may be used instead of equations (10-1) and (10-2): Note that "I" in equations (11-1) and (11-2) is an NxN (N rows, N columns) unit matrix.

[0049]

number

[0050] (Example of multi-user MIMO device configuration) 4 is a schematic block diagram showing an example of the configuration of a base station device 80 in a multi-user MIMO system. As shown in the figure, the base station device 80 includes a transmitter 81, a receiver 85, an interface circuit 87, a MAC (Medium Access Control) layer processing circuit 88, and a communication control circuit 820. The MAC layer processing circuit 88 includes a scheduling processing circuit 881.

[0051] The base station device 80 inputs and outputs data to and from external devices or networks via an interface circuit 87. The interface circuit 87 detects data to be transferred over a wireless channel from the input data and outputs the detected data to a MAC layer processing circuit 88. The MAC layer processing circuit 88 performs MAC layer processing in accordance with instructions from a communication control circuit 820, which manages and controls the operation of the entire base station device 80. Here, the MAC layer processing includes converting data input and output by the interface circuit 87 and data transmitted and received over a wireless channel, and adding MAC layer header information. During this processing, a scheduling processing circuit 881 performs various scheduling processes, including combining terminal devices that simultaneously perform spatial multiplexing in multi-user MIMO transmission. The scheduling processing circuit 881 outputs the scheduling results to the communication control circuit 820. In multi-user MIMO, signals are transmitted to multiple terminal devices at once, so multiple signal sequences are output from the MAC layer processing circuit 88 to the transmitter 81.

[0052] 5 is a schematic block diagram showing an example of the configuration of a transmitter 81 in a base station apparatus 80 in a multi-user MIMO system. As shown in the figure, the transmitter 81 includes transmission signal processing circuits 811-1 to 811-L (L is an integer of 2 or more), addition and synthesis circuits 812-1 to 812-K (K is an integer of 2 or more), IFFT (Inverse Fast Fourier Transform) & GI (Guard Interval) adding circuits 813-1 to 813-K, D / A (Digital / Analog) converters 814-1 to 814-K, a local oscillator 815, mixers 816-1 to 816-K, filters 817-1 to 817-K, high power amplifiers (HPAs) 818-1 to 818-K, antenna elements 819-1 to 819-K, and a transmission weight processing unit 830. The transmission signal processing circuits 811-1 to 811-L and the transmission weight processing unit 830 are connected to the communication control circuit 820 shown in FIG.

[0053] The transmission weight processing unit 830 includes a channel information acquisition circuit 831, a channel information storage circuit 832, and a multi-user MIMO (MU-MIMO) transmission weight calculation circuit 833. Here, the subscript L of the transmission signal processing circuits 811-1 to 811-L in the figure indicates the number of simultaneous spatial multiplexing operations. Also, the subscript K of the circuits from the addition and synthesis circuits 812-1 to 812-K to the antenna elements 819-1 to 819-K indicates the number of antenna elements included in the base station device 80.

[0054] In multi-user MIMO, to transmit signals to multiple terminal devices at once, multiple signal sequences are input from the MAC layer processing circuit 88 to the transmitter 81, and the input multiple signal sequences are input to the transmission signal processing circuits 811-1 to 811-L. When data (data inputs #1 to #L) to be transmitted to each of the destination terminal devices is input from the MAC layer processing circuit 88, the transmission signal processing circuits 811-1 to 811-L generate wireless packets to be transmitted over wireless channels and perform modulation processing. Here, if, for example, an OFDM modulation scheme is used, the signals of each signal sequence are modulated for each frequency component. Furthermore, the modulated baseband signals are multiplied by transmission weights for each frequency component. The signals multiplied by the transmission weights corresponding to the antenna elements 819-1 to 819-K undergo the remaining signal processing as necessary, and are input to the addition and synthesis circuits 812-1 to 812-K as sampled data of the transmission signal in baseband.

[0055] The signals input to summing circuits 812-1 to 812-K are combined for each frequency component. The combined signals are converted from frequency-domain signals to time-domain signals by IFFT & GI adding circuits 813-1 to 813-K, and further processed to insert guard intervals and shape the waveform between OFDM symbols (between blocks of block transmission in the case of SC-FDE (Single-Carrier Frequency Domain Equalization)). Then, for each of antenna elements 819-1 to 819-K, D / A converters 814-1 to 814-K convert the digital sampling data into a baseband analog signal. Furthermore, each analog signal is multiplied by a local oscillation signal input from local oscillator 815 by mixers 816-1 to 816-K, and upconverted to a radio frequency signal. Here, since the up-converted signals contain frequency components outside the band of the channel to be transmitted, filters 817-1 to 817-K remove the frequency components outside the band to generate electrical signals to be transmitted. The generated signals are amplified by high-power amplifiers 818-1 to 818-K and transmitted from antenna elements 819-1 to 819-K.

[0056] 5, after additive synthesis of the signals of each frequency component is performed in additive synthesis circuits 812-1 to 812-K, processing such as IFFT processing, insertion of guard intervals, waveform shaping, etc. is performed, but it is also possible to configure in such a way that these processes are performed in transmission signal processing circuits 811-1 to 811-L and the IFFT & GI adding circuits 813-1 to 813-K are omitted. In this case, the remaining signal processing as necessary after transmission weight multiplication in transmission signal processing circuits 811-1 to 811-L refers to processing such as IFFT processing, insertion of guard intervals, waveform shaping, etc.

[0057] Furthermore, the transmission weights multiplied in the transmission signal processing circuits 811-1 to 811-L are obtained from a multi-user MIMO transmission weight calculation circuit 833 provided in the transmission weight processing unit 830 during signal transmission processing. In the transmission weight processing unit 830, a channel information acquisition circuit 831 separately acquires channel information acquired by the receiving unit 85 via the communication control circuit 820, and stores this information in a channel information storage circuit 832 while updating it as needed. When transmitting a signal, in accordance with an instruction from the communication control circuit 820, the multi-user MIMO transmission weight calculation circuit 833 reads channel information corresponding to the destination station from the channel information storage circuit 832 and calculates transmission weights based on the read channel information. The multi-user MIMO transmission weight calculation circuit 833 outputs the calculated transmission weights to the transmission signal processing circuits 811-1 to 811-L.

[0058] Furthermore, the communication control circuit 820 manages the overall communication control, such as the management of the destination station and overall timing control. The communication control circuit 820 outputs information indicating the destination station, etc. to the transmission weight processing unit 830, which performs signal processing related to the calculation of the transmission weight described above.

[0059] 6 is a schematic block diagram showing an example of the configuration of a receiving unit 85 in a base station device 80 in a multi-user MIMO system. As shown in the figure, the receiving unit 85 includes antenna elements 851-1 to 851-K, low-noise amplifiers (LNAs) 852-1 to 852-K, a local oscillator 853, mixers 854-1 to 854-K, filters 855-1 to 855-K, A / D (analog-to-digital) converters 856-1 to 856-K, FFT (Fast Fourier Transform) circuits 857-1 to 857-K, receiving signal processing circuits 858-1 to 858-L, and a receiving weight processing unit 860. The receiving signal processing circuits 858-1 to 858-L and the receiving weight processing unit 860 are connected to the communication control circuit 820 shown in FIG. The receiving weight processing unit 860 includes a channel information estimation circuit 861 and a multi-user MIMO (MU-MIMO) receiving weight calculation circuit 862.

[0060] Signals received by antenna elements 851-1 to 851-K are amplified by low-noise amplifiers 852-1 to 852-K. The amplified signals are multiplied by the local oscillation signal output from local oscillator 853 in mixers 854-1 to 854-K, and the amplified signals are down-converted from radio frequency signals to baseband signals. Because the down-converted signals contain frequency components outside the frequency band to be received, the out-of-band components are removed by filters 855-1 to 855-K. The signals from which the out-of-band components have been removed are converted into digital baseband signals by A / D converters 856-1 to 856-K. All digital baseband signals are input to FFT circuits 857-1 to 857-K, which convert the signals on the time axis into signals on the frequency axis at the specified symbol timing (separating them into signals of each frequency component). The signals separated into each frequency component are input to received signal processing circuits 858-1 to 858-L, and also to channel information estimation circuit 861.

[0061] The channel information estimation circuit 861 estimates channel information between the antenna elements of each terminal device and each of antenna elements 851-1 to 851-K of base station device 80 for each frequency component based on known signals for channel estimation (such as a preamble signal added to the beginning of a wireless packet) separated into each frequency component, and outputs the estimation results to a multi-user MIMO receiving weight calculation circuit 862. The multi-user MIMO receiving weight calculation circuit 862 calculates a receiving weight to be multiplied for each frequency component based on the input channel information. At this time, the receiving weights that combine the signals received by each of antenna elements 851-1 to 851-K differ for each signal sequence, and are input to the receiving signal processing circuits 858-1 to 858-L corresponding to the signal sequence to be extracted.

[0062] The reception signal processing circuits 858-1 to 858-L multiply the signals for each frequency component input from the FFT circuits 857-1 to 857-K by the reception weights input from the multi-user MIMO reception weight calculation circuit 862, and add and combine the signals received by the antenna elements 851-1 to 851-K for each frequency component. The reception signal processing circuits 858-1 to 858-L demodulate the added and combined signals, and output the recovered data to the MAC layer processing circuit 88.

[0063] Here, the different reception signal processing circuits 858-1 to 858-L perform signal processing of different signal sequences. Furthermore, the MAC layer processing circuit 88 performs processing related to the MAC layer (for example, conversion between data input / output to / from the interface circuit 87 and data transmitted / received over a wireless channel, termination of MAC layer header information, etc.). In this processing, the scheduling processing circuit 881 performs various scheduling processes including combinations of terminal devices that simultaneously perform spatial multiplexing in multi-user MIMO transmission, and outputs the scheduling results to the communication control circuit 820. The reception data processed by the MAC layer processing circuit 88 is output to an external device or a network via the interface circuit 87.

[0064] Furthermore, the communication control circuit 820 manages the overall communication related control, such as management of the terminal device of the transmission source and overall timing control. Furthermore, information indicating the terminal device of the transmission source, etc. is input from the communication control circuit 820 to the reception weight processing unit 860, which performs signal processing related to the calculation of the reception weight described above.

[0065] As with transmission, in signal reception, in a wideband system using the OFDM modulation scheme or the SC-FDE scheme, the multiplication of the above-mentioned receiving weights is performed for each frequency component. That is, the signals output from the A / D converters 856-1 to 856-K are subjected to FFT in the FFT circuits 857-1 to 857-K to separate them into frequency components, and for each separated frequency component, signal processing is performed in the channel information estimation circuit 861 and received signal processing is performed in the received signal processing circuits 858-1 to 858-L.

[0066] (Multi-user MIMO transmission processing) 7 is a flowchart showing transmission processing of the base station device 80 in multi-user MIMO. In multi-user MIMO, downlink channel information is periodically fed back separately from data transmission. When the channel information acquisition circuit 831 acquires downlink channel information (step S831), it stores the channel information of each frequency component for each terminal device in the channel information storage circuit 832 (step S832). The processing of steps S831 and S832 is performed sequentially.

[0067] When the signal transmission process from the base station device 80 is started (step S821), the multi-user MIMO transmission weight calculation circuit 833 reads out the channel information of each frequency component corresponding to the destination terminal device from the channel information storage circuit 832 (step S822).

[0068] The multi-user MIMO transmission weight calculation circuit 833 calculates a multi-user MIMO transmission weight for each frequency component based on the read channel information by the process described above (step S823). Separately from the processes of steps S822 and S823, the transmission signal processing circuits 811-1 to 811-L generate a transmission signal of each frequency component for each destination station by transmission signal processing such as various modulation processes for data to be transmitted to each destination station (step S824).

[0069] The transmission signal processing circuits 811-1 to 811-L multiply the generated transmission signals by the transmission weights calculated by the multi-user MIMO transmission weight calculation circuit 833 in step S823 (step S825). The transmission signal processing circuits 811-1 to 811-L also perform a series of signal processing, and the addition and synthesis circuits 812-1 to 812-K add and synthesize the transmission signals of each frequency component addressed to each terminal device for each antenna element 819-1 to 819-K. The signals are then converted from signals on the frequency axis to signals on the time axis by the IFFT & GI addition circuits 813-1 to 813-K, and further processed such as insertion of guard intervals and waveform shaping between OFDM symbols (between blocks of block transmission in the case of SC-FDE), before being output to the D / A converters 814-1 to 814-K (steps S826-1 to S826-K).

[0070] The signals output from the IFFT & GI adding circuits 813-1 to 813-K are subjected to signal processing in the D / A converters 814-1 to 814-K and high power amplifiers 818-1 to 818-K, and are transmitted from the antenna elements 819-1 to 819-K, respectively (steps S827-1 to S827-K), and the processing ends (steps S828-1 to S828-K).

[0071] The processing in steps S827-1 to S827-K includes up-conversion processing from a baseband signal to a radio frequency, removal of frequency components in a band using a filter, and signal amplification using a high-power amplifier.

[0072] (Multi-user MIMO receiving processing) 8 is a flowchart showing the reception processing of base station device 80 in multi-user MIMO. First, when the reception processing starts (step S840), signals are received by first to Kth antenna elements 851-1 to 851-K (steps S841-1 to S841-K). Here, reception includes processing up to and including analog-to-digital conversion of the received signal or a signal obtained by down-converting the received signal. The subsequent signal processing refers to processing of the digitized received signal.

[0073] Next, FFT circuits 857-1 to 857-K perform signal processing such as separating the received signals corresponding to antenna elements 851-1 to 851-K into frequency components (steps S842-1 to S842-K). Furthermore, channel information estimation circuit 861 performs channel estimation for each frequency component based on the reception state of a preamble signal of a known pattern that was added to the wireless packet (steps S843-1 to S843-K). Here, the attenuation of the signal on the propagation path and the rotation state of the complex phase are determined. In the channel estimation performed in steps S843-1 to S843-K, channel estimation must be performed individually for each spatially multiplexed signal sequence, as shown in steps S843-1, S843-2, ..., S843-K.

[0074] This individual channel estimation needs to be performed in a state where the signals transmitted from each of the source terminal devices can be separated. Taking the OFDM modulation method as an example, a preamble signal for channel estimation, the number of symbols of which is equal to the number of spatial multiplexing, is generally required. Each terminal device transmits a signal with the same number of symbols as the number of spatial multiplexing (or more) and with a preamble signal of a different pattern, and the base station device 80 uses the differences in the patterns to perform individual channel estimation in steps S843-1 to S843-K.

[0075] The multi-user MIMO receiving weight calculation circuit 862 calculates appropriate receiving weights for each spatially multiplexed signal sequence and each frequency component using the channel information estimated by the channel information estimation circuit 861 (step S844). Furthermore, the receiving signal processing circuits 858-1 to 858-L multiply the receiving signals of each antenna element separated for each frequency component by the receiving weights calculated for each signal sequence and each frequency component (steps S845-1 to S845-K).

[0076] Here, since a receiving weight is prepared for each spatially multiplexed signal sequence, the multiplication results in steps S845-1 to S845-K are different for each spatially multiplexed signal sequence. For each signal sequence, the signals of antenna elements 851-1 to 851-K are added and combined for each frequency component (steps S846-1 to S846-L), and the combined signal sequence is subjected to signal processing from the first signal sequence (step S847-1) to the Lth signal sequence (step S847-L), after which the processing ends (steps S848-1 to S848-L).

[0077] For simplicity, an example using linear receiving weights has been shown here. However, in general, nonlinear signal processing such as MLD (Maximum Likelihood Detection) may be performed for MIMO. In this case, the processes in steps S845-1 to S845-L, steps S846-1 to S846-L, and steps S847-1 to S847-L are integrated into a nonlinear signal detection process. Furthermore, the linear receiving weights can be calculated using a method similar to the calculation process for the transmitting weights shown in FIG. 3. Alternatively, receiving weights using a pseudo-inverse matrix or MMSE weights can also be used. Furthermore, in the above description, the number of spatially multiplexed signal sequences is L, relative to the number K of antenna elements 851-1 to 851-K used for reception. However, generally, K and L do not need to be equal. As long as the value of the spatial multiplexing number L is equal to or less than the number K of antennas, signals of a large number of signal sequences can be spatially multiplexed.

[0078] It is generally known that MIMO transmission improves in performance as the number of antenna elements on the transmitting and receiving stations increases relative to the number of spatially multiplexed signal sequences. This improvement in performance is utilized in the form of improving the SINR (Signal to Interference and Noise Ratio) of each spatially multiplexed signal sequence and further increasing the number of spatial multiplexing sequences. In recent years, the implementation of Massive MIMO technology, such as that discussed in Non-Patent Document 2, which expands the number of antenna elements on the base station side to an ultra-large number of 100 or more, has also progressed. Null space expansion technology is being studied as an interference suppression technology that utilizes this ultra-large number of antenna elements on the base station side (see, for example, Non-Patent Document 3 and Patent Document 1).

[0079] In multi-user MIMO technology, a base station device grasps channel information for all terminal devices to be multiplexed, and transmits signals by performing interference suppression processing based on the grasped channel information so as not to cause interference between the terminal devices. As mentioned above, there are multiple methods for interference suppression processing, but all of these methods have in common the point that they add pre-processing based on channel information regarding terminal devices other than the destination terminal device so that the interfering signal is not received by terminal devices other than the destination terminal device (so that a null is formed). For example, when focusing on the i-th terminal device, the partial channel matrix H i sub Based on this, the partial channel matrix H for the other j (j≠i)-th terminal device is j sub A weight that is orthogonal to each row vector of is generated and multiplied by the signal directed to the i-th terminal device to perform interference suppression.

[0080] On the other hand, in the null space expansion technique, the partial channel matrix H i sub is expanded by inserting additional channel vectors, and the expanded partial channel matrix H i ' subBy suppressing interference based on this, nulls are formed over a wider area than with conventional techniques. This means that in addition to the null in the direction that should originally be pointed, new nulls are formed in other directions, but when channels related to terminal devices other than the i-th one fluctuate over time, even if they deviate from the original null point, as long as the destination of the fluctuation is near the additional null point, the same interference suppression effect can be obtained, and as a result, a high inter-user interference reduction effect can be obtained even in a time-varying environment.

[0081] As an example, consider the case where a base station device with 100 antenna elements simultaneously transmits spatially multiplexed signals to 10 terminal devices, each with a single antenna element. In this case, the channel vector for each terminal device is a 100-dimensional vector. In conventional technology, 10 degrees of freedom are used for spatial multiplexing of 10 terminals, i.e., interference suppression for the 10 terminals, and the remaining 90 (=100-10) degrees of freedom are used to improve the line gain of each terminal device through in-phase combining. Here, processing is performed assuming that there is another virtual terminal device (actually, the original terminal device has moved) nearby that should suppress interference. In conventional technology, when the channel fluctuates due to the movement of each terminal device, the effect of interference suppression is weakened and interference increases significantly. However, if the terminal device moves to the location of the virtual terminal device due to time fluctuation, interference suppression is expected to be similarly achieved. In this case, interference suppression is also performed for the additional virtual terminal device, consuming one additional degree of freedom for each terminal device.

[0082] In other words, the total number of degrees of freedom used for interference suppression becomes 20, and the number of degrees of freedom available for improving the link gain through in-phase combining is reduced to 80. If the link gain improvement rate is proportional to the square root of the degree of freedom, the difference in the number of degrees of freedom used for link gain improvement is 10Log(80 / 90) = -0.51 dB, which is only about 0.5 dB. However, if the time variation of the channel is within the range of the additional interference suppression implemented as described above, the power of inter-user interference can be significantly reduced. In other words, although the SNR (Signal-to-Noise Ratio) will deteriorate by about 0.5 dB, a significant improvement in the SIR (Signal-to-Interference power Ratio) is expected. Although the transmission capacity of multi-user MIMO is ultimately determined by the SINR (Signal-to-Interference plus Noise power Ratio), this method is effective because the SIR characteristic is considered to be more dominant than the SNR in a time-varying environment.

[0083] Next, the key points of the null space extension technology will be explained. Fig. 1 shows a beam pattern generated by a base station (BS) included in a wireless communication system using the null space extension technology, and Fig. 9 shows a beam pattern generated by a base station (BS) included in a wireless communication system using conventional technology. In a wireless communication system, a base station and multiple terminal devices can perform spatial multiplexing transmission on the same frequency at the same time. In the wireless communication systems shown in Figs. 1 and 9, a base station communicates with terminal device T1 and terminal device T2 by spatial multiplexing transmission. The time variation prediction destination of terminal device T1 is T1', and the time variation prediction destination of terminal device T2 is T2'.

[0084] As shown in Figure 9, a base station device of the prior art transmits a signal to a destination terminal device T1 located in the direction of arrow D1 using a beam pattern B8 that directs a null toward a terminal device T2 other than the terminal device T1. However, when the terminal device T2 moves to a position of the time-varying predicted destination T2' indicated by the dotted line, the terminal device T2 moves away from the null in the direction of arrow D2, and interference power increases in the terminal device T2. On the other hand, as shown in Figure 1, a base station device using null space extension technology generates a beam pattern B1 by adding a null to a point T2' where time variation is expected in the direction of arrow D2', which is added based on the extended channel matrix. As a result, even if the actual movement of the terminal device T2 is to a point T2' where time variation is expected or if the terminal device T2 remains at its original position, it is possible to suppress interference power as long as it falls within the range of the null formed by the above procedure.

[0085] Fig. 11 shows an example of the configuration of an extended channel matrix used to calculate weights using the null space extension technique, and Fig. 10 shows an example of the configuration of a channel matrix used to calculate weights using the conventional technique. For simplicity, we consider a case where each terminal device is equipped with one antenna and L terminal devices are spatially multiplexed. h1(t) to h L (t) is the channel vector of each terminal device at time t, which has the dimension of the number of base station antenna elements. Note that the channel vector h i is (h i1 ,h i2 ,…,h ij ) and h ij is the channel information between the j-th antenna element of the base station device and the i-th terminal device.

[0086] As shown in Fig. 10, in the conventional technology, a channel matrix is ​​used in which channel vectors corresponding to the number of terminals for simultaneous multiplex transmission are arranged. When multiplex transmission is performed toward terminal device #1, channel vectors h2(t) to h3(t) for other terminal devices #2 to #L are arranged as L (t)(=H1 sub ) is multiplied by the weights to perform processing such as orthogonalization so that the wireless signals cancel each other out and become null, thereby suppressing interference between users.

[0087] On the other hand, as shown in FIG. 11, in the null space expansion technology, the partial channel matrix H1main is common to the conventional technology, but the partial channel matrix H1 sub the current channel vectors h2(t) to h L In addition to (t), past channel vectors h2(t-nT) to h2(t-nT) for other terminal devices #2 to #L are L (t-nT) is added to the past channel vector h i (t-nT) is the channel vector acquired before time nT for the i-th terminal device #i. Here, propagation path estimation is performed every time period T, and n=1... is an integer indicating how many periods ago the channel vector was estimated, but if propagation path estimation is not performed periodically, any past time may be given as an argument. Also, the number of past channel vectors to be expanded may differ depending on the terminal device #i.

[0088] The key point of the null space expansion technique is the partial channel matrix H sub This is a process to extend the partial channel matrix H sub By performing weight calculation after extending the extended partial channel matrix H'1 as described above, it is possible to suppress interference over a wider range than with existing techniques. sub By performing processing such as orthogonalization on the partial channel matrix H, the beam is directed toward the desired terminal device #1, and in addition to forming nulls for other terminal devices, weights with additional null formation can be obtained, making it possible to suppress interference between users in a time-varying environment. In other words, the null space extension technology calculates weights that align the phase of the radio signal when multiplied by the channel vector of the destination terminal device, and that cancel each other out to form a null when multiplied by the channel vector of terminal devices other than the destination and the past channel vector. Note that the partial channel matrix H subThe past channel vector to be added for generating the past channel vector does not necessarily have to be one for each terminal device, and multiple past channel vectors may be generated and added. Furthermore, a different number of past channel vectors may be generated for each terminal device.

[0089] The weights generated by the null space expansion technique are used to calculate the current channel vectors h2(t) to h L (t) and the past channel vector h2(t-nT)~h L (t-nT), the null is not only directed to both the current channel vector h j (t) and the past channel vector h j For (t-nT), for any complex coefficient γ, h j (t)+γ×{h j (t-nT)-h j Therefore, it is not necessary to predict pinpoint propagation path fluctuations expressed as highly accurate "points," but it is sufficient to appropriately extract the subspace where the null should be directed.

[0090] In other words, if the channel varies over time within the subspace, interference will be suppressed at any time δt. Conventional channel prediction techniques used to address time variations predict the channel at the desired transmission time δt and pinpoint weights that suppress interference for the predicted channel. Therefore, it was necessary to predict the channel and generate weights for each transmission request, or to predict all channels at the expected transmission times in advance and store the generated weights. On the other hand, weights based on null space expansion techniques can be obtained by forming nulls once in accordance with the update of channel information. [Prior art documents] [Patent documents]

[0091] [Patent Document 1] Japanese Patent Application Laid-Open No. 2016-136706 [Non-patent literature]

[0092] [Non-Patent Document 1] Yasushi Takatori et al., "Application of Downlink Multi-User MIMO Technology to Next-Generation High-Speed ​​Wireless Access Systems," Institute of Electronics, Information and Communication Engineers, IEICE Transactions on Wireless Access Systems, Vol. J93-B, No. 9, pp. 1127-1139, September 2010. [Non-patent document 2] Kazuki Maruta et al., "Proposal for a Large-Scale Antenna Wireless Entrance System - Performance Evaluation by Computer Simulation -", Institute of Electronics, Information and Communication Engineers, Technical Report of the Institute of Electronics, Information and Communication Engineers, RCS2013-6, vol.113, no.8, pp.31-36, April 2013 [Non-patent document 3] T.Iwakuni, et. al, "Null-Space Expansion for Multiuser Massive MIMO Inter-User Interference Suppression in Time Varying Channels," IEICE TRANSACTIONS on Communications, Vol.E100-B, No.5, pp.865-873 [Non-patent document 4] Taromaru, Makoto et al., "A Study on Beam Patterns of Multi-User MIMO Base Stations Formed by Null Space Expansion Method - Consideration of the Principle of Improving Tolerance to Channel Fluctuations by Orthogonalizing to Past Interference Channels -", ​​Institute of Electronics, Information and Communication Engineers, Technical Report of the Institute of Electronics, Information and Communication Engineers, RCS2017-226, vol.117, no.284, pp.123-127, November 2017 Summary of the Invention [Problem to be solved by the invention]

[0093] Weight generation in null space expansion technology has the following problem. When attempting to form multiple nulls when the amount of propagation path variation is small, the null formation consumes a large amount of freedom, resulting in a decrease in the desired signal gain. Furthermore, while the calculation of null formation requires a large amount of calculation, the interference suppression effect is limited.

[0094] The present invention has been made in view of the above circumstances, and aims to provide a base station apparatus, a wireless communication method, and a wireless communication system that are capable of extracting appropriate channel vectors and generating weights to efficiently suppress inter-user interference caused by channel fluctuations.

[0095] The present invention has been made in view of the above circumstances, and aims to provide a base station apparatus, a weight generation method, and a wireless communication system that can efficiently perform interference suppression with a low amount of calculation. [Means for solving the problem]

[0096] One aspect of the present invention is a base station device in a wireless communication system that includes a base station device having a plurality of antenna elements and a plurality of terminal devices, and that is capable of performing spatial multiplexing transmission on the same frequency at the same time, the base station device having a weight calculation unit that calculates weight vectors for performing spatial multiplexing transmission for the plurality of terminal devices based on a channel matrix that arranges channel vectors generated from channel information between the antenna elements of the terminal devices or a virtual antenna element obtained by combining the antenna elements and the antenna elements of the base station device, and additional channel vectors that are different from the channel vectors and are associated with the antenna elements of the terminal devices, and the weight calculation unit selects, as the additional channel vector for the terminal device, from one or more second channel vectors acquired before a first channel vector, which is the channel vector of the terminal device, the second channel vector that has a low mutual similarity with the first channel vector, and calculates the weight vector.

[0097] Another aspect of the present invention is a weight generation method performed by a base station device in a wireless communication system that includes a base station device having a plurality of antenna elements and a plurality of terminal devices, and that is capable of performing spatial multiplexing transmission on the same frequency at the same time. The weight generation method includes the steps of: calculating weight vectors for performing spatial multiplexing transmission for the plurality of terminal devices based on a channel matrix that arranges channel vectors generated using channel information between the antenna elements of the terminal devices or a virtual antenna element obtained by combining the antenna elements and the antenna elements of the base station device, and additional channel vectors that are different from the channel vectors and are associated with the antenna elements of the terminal devices; and selecting, as the additional channel vector for the terminal device, a second channel vector that has a low degree of mutual similarity with the first channel vector from one or more second channel vectors acquired before the first channel vector, which is the channel vector of the terminal device, and calculating the weight vector.

[0098] Another aspect of the present invention is a wireless communication system comprising a base station device having a plurality of antenna elements and a plurality of terminal devices, wherein the base station device and the terminal devices are capable of performing spatial multiplexing transmission on the same frequency at the same time, wherein the base station device comprises a weight calculation unit that calculates weight vectors for performing spatial multiplexing transmission for the plurality of terminal devices based on a channel matrix that arranges channel vectors generated from channel information between the antenna elements of the terminal devices or a virtual antenna element obtained by combining the antenna elements and the antenna elements of the base station device, and additional channel vectors that are different from the channel vectors and are associated with the antenna elements of the terminal devices, and the weight calculation unit selects, as the additional channel vector of the terminal device, from one or more second channel vectors acquired before a first channel vector, which is the channel vector of the terminal device, the second channel vector that has a low mutual similarity with the first channel vector, and calculates the weight vector. [Effects of the Invention]

[0099] According to the present invention, it is possible to efficiently suppress interference with a low amount of calculation. [Brief explanation of the drawings]

[0100] [Figure 1] 10A and 10B are diagrams illustrating examples of beam patterns generated by a base station device according to an embodiment of the present invention. [Figure 2] FIG. 1 is a schematic diagram illustrating an example of the configuration of a conventional multi-user MIMO system. [Figure 3] 1 is a flowchart showing a procedure for calculating a transmission weight matrix W in a multi-user MIMO system according to the prior art. [Figure 4] FIG. 1 is a schematic block diagram showing an example of the configuration of a base station device 80 in a conventional multi-user MIMO system. [Figure 5]1 is a schematic block diagram showing an example of the configuration of a transmitter 81 provided in a base station device 80 in a multi-user MIMO system according to the prior art. [Figure 6] 1 is a schematic block diagram showing an example of the configuration of a receiver 85 included in a base station device 80 in a multi-user MIMO system according to the prior art. [Figure 7] 10 is a flowchart showing a transmission process of a base station device 80 in multi-user MIMO according to the prior art. [Figure 8] 10 is a flowchart showing a receiving process of a base station device 80 in multi-user MIMO according to the prior art. [Figure 9] FIG. 1 is a diagram illustrating an example of a beam pattern generated by a base station device according to the prior art. [Figure 10] FIG. 1 is a diagram illustrating an example of the configuration of a channel matrix used for weight calculation in the prior art. [Figure 11] FIG. 10 is a diagram illustrating an example of the configuration of an extended channel matrix used for weight calculation in an embodiment of the present invention. [Figure 12] 1 is a schematic block diagram showing an example of the configuration of a transmission weight processing unit 130 according to an embodiment of the present invention. [Figure 13] 5 is a flowchart showing the operation of a transmission weight processing unit 130 according to the embodiment of the present invention. [Figure 14] 5 is a flowchart showing the operation of a transmission weight processing unit 130 according to the embodiment of the present invention. DETAILED DESCRIPTION OF THE INVENTION

[0101] Hereinafter, an embodiment of the present invention will be described with reference to the drawings.

[0102] The basic principle of an embodiment of the present invention will be explained. It is known that the shape of the null formed in the null space extension technology forms a null in the propagation path element wave that causes channel fluctuations at the interfering terminal (see, for example, Non-Patent Document 4). On the other hand, it is difficult to estimate the arrival direction of the element wave itself. Therefore, the null space extension technology can be said to be a technology that automatically suppresses interference in the element wave direction by adding past channel vectors for the interfering terminal as an extended channel matrix.

[0103] In this case, forming a null on a propagation path with small channel fluctuations results in forming a null on adjacent rays, which reduces the interference suppression effect. sub In the above, the current channel vector h i (t), when forming a null for the past channel vector hi(t-nT), h i (t) and h i If (t-nT) are similar, it can be said that there is no need to perform null formation for each of them. Therefore, in the embodiment of the present invention, an index of similarity is introduced between the current channel vector and past channel vector for these other terminal devices, and only those with low similarity are used in the weight calculation of the null space expansion technology.

[0104] This will enable multi-user MIMO transmission using only channel vectors that are effective for interference suppression.

[0105] The difference between the configuration of the base station device 80 according to the embodiment of the present invention and the configuration of the conventional base station device 80 shown in Fig. 4 is that the configuration of the transmission weight processing unit 830 of the transmitting unit 81 shown in Fig. 5 is the configuration of the transmission weight processing unit 130 described below. Hereinafter, components having the same configuration as those of the conventional base station device 80 will be assigned the same reference numerals, and descriptions thereof will be omitted.

[0106] As mentioned above, the multi-user MIMO system of the embodiment described below is a wireless communication system that includes a base station device 80 equipped with multiple antenna elements and multiple terminal devices 802, and that allows the base station device 80 and the terminal devices 802 to perform spatial multiplexing transmission on the same frequency at the same time.

[0107] [Configuration of transmission wait processing unit] The transmission weight processing unit 130 (weight calculation unit) in the embodiment calculates weight vectors for performing spatial multiplexing transmission for multiple terminal devices 802 based on a channel matrix that arranges channel vectors generated from channel information between the antenna elements of the terminal device 802 or virtual antenna elements obtained by combining the antenna elements and the antenna elements provided by the base station device 80, and additional channel vectors associated with the antenna elements of the terminal device 802 that are different from the channel vectors.

[0108] Here, the transmission weight processing unit 130 selects, as an additional channel vector for the terminal device 802, a second channel vector that has a low degree of mutual similarity with the first channel vector from one or more second channel vectors obtained before the first channel vector, which is the channel vector of the terminal device 802, and calculates a weight vector.

[0109] 12 shows a block diagram of the transmission weight processing unit 130 in the embodiment of the present invention, and the weight calculation will be described in detail below. The case where the transmission weight is calculated will be described below.

[0110] 12 is a block diagram showing the functional configuration of the transmission weight processing unit 130 in the embodiment of the present invention. As shown in FIG. 12, the transmission weight processing unit 130 includes a channel information acquisition circuit 831, a channel information storage circuit 132, a MU-MIMO transmission weight calculation circuit 133, a channel similarity calculation circuit 135, and a channel information selection circuit 136.

[0111] The channel similarity calculation circuit 135 calculates the current channel vector h i (t) and the past channel vector h i Based on the absolute value of the inner product with (t-nT), the current channel vector h i (t) and the past channel vector h i For example, the channel similarity calculation circuit 135 calculates the mutual similarity between the current channel vector h i (t) and the past channel vector h i The correlation coefficient between (t-nT) is calculated using the following equation (12).

[0112]

number

[0113] Here, H represents a conjugate transpose vector. The channel similarity calculation circuit 135 calculates the correlation coefficient γ i (n) is the absolute value of |γ i (n)| is less than a certain value, the similarity between the propagation paths is determined to be low, and h i A record is made that the similarity to (t-nT) is low.

[0114] The correlation coefficient may be normalized by the norm of each channel vector. i (t) and the past channel vector h i The absolute value of the inner product of (t-nT) and the current channel vector h i The norm of (t) and the past channel vector h i Based on the cross-correlation value divided by the product with the norm of (t-nT), the current channel vector h i (t) and the past channel vector h i (t-nT) may be calculated. In this case, equation (12) can be rewritten as equation (13) below.

[0115]

number

[0116] In addition, in equation (13), each channel vector is normalized using the norm of each channel vector, but the norm of the channel vector is expected to have a certain magnitude depending on the number of elements of the base station antenna, etc. Therefore, the channel similarity calculation circuit 135 calculates the current channel vector h i (t) and the past channel vector h i The absolute value of the inner product of (t-nT) and the current channel vector h i (t) and the past channel vector h i (t-nT) or an approximate correlation value divided by the square of either norm, and the current channel vector h i (t) and the past channel vector h i The mutual similarity with (t-nT) may be calculated.

[0117] That is, for example, instead of equation (13), the approximate norm expressed by the following equation (14) or (15) may be used.

[0118]

number

[0119]

number

[0120] Furthermore, the difference vector between the channel vectors used for null formation represents the difference in the direction of the null to be formed for that channel vector, and by obtaining the norm, it is possible to evaluate it as the similarity between the vectors. Therefore, the channel similarity calculation circuit 135 calculates the current channel vector h i (t) and the past channel vector h i(t-nT) or the norm of the difference vector is used to calculate the current channel vector h i (t) and the past channel vector h i Based on the value obtained by dividing by the norm of either one of (t-nT), the current channel vector h i (t) and the past channel vector h i The mutual similarity with (t-nT) may be calculated.

[0121] That is, for example, the difference vector d i The norm of (n) can be used as the similarity measure.

[0122]

number

[0123] After calculating the similarity between channel vectors using these methods, the past channel vectors h i The (t−nT) information is sent to the channel information selection circuit 136 .

[0124] The channel information selection circuit 136 selects from the channel information storage circuit 132 a past channel vector h i (t-nT) and transfers it to the MU-MIMO transmission weight calculation circuit 133. As a result, the MU-MIMO transmission weight calculation circuit 133 calculates the extended partial channel matrix H i ' sub and the transmission weights can be calculated based on the obtained values.

[0125] The channel information selection circuit 136 may determine the weight vectors by excluding weight vectors calculated before the time when the similarity between the channel vectors having a value higher than a predetermined value appears more than a predetermined number of times.

[0126] [Operation of the transmission wait processing unit] The operation of the transmission weight processing unit 130 is shown as a flowchart in, for example, FIGS.

[0127] First, the channel similarity calculation circuit 135 calculates the similarity of the above-mentioned channel vector for each terminal that performs spatial multiplexing transmission. That is, for terminal #k, the most recent channel vector h k (t) and the past channel vector h obtained in the nth channel estimation. k The similarity between (t-nT) is calculated using one of the formulas (12) to (16) (step S050).

[0128] If the similarity thus obtained is equal to or less than a certain value (YES in step S050), the channel similarity calculation circuit 135 calculates the past channel vector h k (t-nT) is recorded as the matrix to be used for the partial channel matrix (step S060). If n reaches a predetermined upper limit or if there are no more past channel vectors (e.g., they are not stored), the process ends. Otherwise (step S080 YES), n is incremented by 1 (step S040), and the similarity is calculated in the same way.

[0129] When the calculation of the similarity of the past channel vectors is completed, the channel similarity calculation circuit 135 performs the same similarity calculation for all terminals that simultaneously transmit by spatial multiplexing (steps S020 to S090). When the above processing is completed for all terminals (step S090: YES), the channel similarity calculation circuit 135 calculates the partial channel matrices Hmain and H'sub for each terminal device #k (steps S120 to S130).

[0130] The MU-MIMO transmission weight calculation circuit 133 calculates the extended partial channel matrix H' sub The orthogonal basis vectors of the subspace spanned by each row vector of j} (step S135). Next, the MU-MIMO transmission weight calculation circuit 133 calculates the partial channel matrix H main The basis vector {e' j} and this is the matrix ~H' main (Step S140). Furthermore, the MU-MIMO transmission weight calculation circuit 133 calculates the matrix H' main Calculate any orthogonal basis vector of the subspace spanned by the row vector of k} (step S145).

[0131] Here, an arbitrary basis vector is, for example, a matrix H main Alternatively, the MU-MIMO transmission weight calculation circuit 133 may select a vector that constitutes a right singular matrix when singular value decomposition is performed on the basis vector {e' k}, the transmission weight vector {w' k} is determined (step S150).

[0132] The MU-MIMO transmission weight calculation circuit 133 determines whether the transmission weight vectors for all the destination terminal devices 802 have been determined (step S155). If the MU-MIMO transmission weight calculation circuit 133 determines that there is an unprocessed terminal device 802 (step S155: NO), it repeats the processing from step S120 to step S150. Then, if the MU-MIMO transmission weight calculation circuit 133 determines that the transmission weight vectors for all the destination terminal devices 802 have been determined (step S155: YES), it calculates the transmission weight vector {w' k The transmission weight matrix W' is determined as a matrix with each column vector being {k, k, k} (step S160), and the process ends. k This becomes:

[0133] In this way, by performing orthogonalization processing, the base station device 80 of the embodiment can extract only channel vectors that are effective in interference suppression and perform null formation, thereby efficiently generating multi-user MIMO weights.

[0134] The transmitter 81 of the base station device 80 performs transmission processing using the transmission weight matrix W' determined in the transmission weight calculation processing shown in Fig. 13 and Fig. 14 described above. As a result, the transmission signal s from the base station device 80 becomes the transmission weight matrix W' x the transmission signal t from the base station device to each terminal. However, the transmission signal s is composed of transmission signals s1, s2, ..., s from each of the antenna elements 819-1 to 819-K. K The transmission signal t is a column vector having elements t1, t2, ..., t from the base station device 80 to the terminal devices 802-1 to 802-L, respectively. L is a column vector with elements.

[0135] As described above, when multiple channel vectors used in the null space extension technology are similar, the interference suppression effect of null formation using both of them decreases. Therefore, the base station device 80 in the embodiment of the present invention calculates the similarity between channel vectors and uses only channel vectors whose similarity is below a certain level for weight formation. This allows the base station device 80 to efficiently suppress inter-user interference. According to the base station device 80 in the embodiment of the present invention, by extracting only channel vectors that are effective for interference suppression in the null space extension technology and generating weights, it becomes possible to efficiently suppress interference with a low amount of calculation.

[0136] As described above, the present invention relates to a null-forming technology using multiple antenna elements that is applied to multi-user MIMO wireless communications and the like. Null-forming technologies that generate weights for signals from other users, other beams, and other streams and perform matrix calculations to suppress interference between users have been studied. However, null-forming consumes degrees of freedom, i.e., the number of antenna elements used. While interference can be suppressed, the gain of the desired signal decreases. Furthermore, conventional methods have the problem of requiring a huge amount of calculations for null generation. In response to this problem, the present invention devise a new formula for calculating the similarity between channel vectors of each channel, and uses those channel vectors for weight generation only when the similarity is high, thereby enabling efficient interference suppression and resolving the above-mentioned problems.

[0137] [Other supplementary points regarding the embodiment of the present invention] Below, some supplementary points regarding the embodiments of the present invention are summarized.

[0138] Any method can be used as the downlink channel estimation method in the embodiment of the present invention. For example, when acquiring downlink channel information, a method (explicit feedback method) can be used in which a base station device transmits a predetermined training signal on the downlink, a terminal device performs channel estimation based on the training signal, and the channel information is directly fed back to the base station device using a wireless packet containing predetermined control information. Similarly, a method (implicit feedback method) can be used in which a terminal device transmits a predetermined training signal on the uplink, a base station device performs uplink channel estimation based on the training signal, and the downlink channel information is estimated after a predetermined calibration process based on the obtained uplink channel information.

[0139] 12 can be realized in a variety of ways, but in the embodiment of the present invention, the processing is performed after the channel information acquisition circuit 831 has completed acquisition of channel information, so there is no effect no matter what channel information acquisition method is used in the conventional channel information acquisition circuit 831. In other words, the embodiment of the present invention is applicable to any channel information acquisition method.

[0140] In the above description, for simplicity, subscripts representing frequency components are omitted, and furthermore, descriptions of individual frequency components are omitted. However, generally, all signal processing, such as channel information, transmit / receive weights, and transmit signals and receive signals, is defined and processed individually for each frequency component on the frequency axis. Within each signal processing circuit, for example, signal processing up to the stage preceding IFFT processing on the transmitting side (bit sequence interleaving, signal point mapping, signal modulation, transmit weight multiplication, etc.) is performed for each frequency component. Similarly, signal processing from FFT processing on the receiving side (receive weight multiplication, signal detection, signal demapping, deinterleaving, etc.) is also performed for each frequency component. Therefore, acquisition of downlink channel information is also performed for each frequency component, and similarly, prediction of future channel information is also performed for each frequency component.

[0141] However, although downlink channel information is typically acquired individually for all frequency components, it is expected that the accuracy of future channel prediction will be lower than that of simple channel information acquisition. In this case, it is not necessary to perform channel prediction for all subcarriers, and it is expected that characteristics will not deteriorate significantly even if channel prediction is performed by thinning out frequency components to a certain extent and using predicted channel vectors for nearby frequency components. For example, if channel prediction is performed once for three frequency components, it is also acceptable to use predicted channel vectors for frequency components before and after the frequency component for which prediction was performed. If the correlation in the frequency direction is even stronger, it is also acceptable to perform channel prediction by thinning out even more.

[0142] In terms of circuit configuration, individual circuits may be provided for each frequency component, or since the same processing is performed, processing may be performed serially for each frequency component in order, and a circuit may be shared for all frequency components.Furthermore, as an intermediate step, multiple circuits may be provided, the frequency components may be appropriately divided, and parallel processing may be performed serially by the multiple circuits.These are common to all embodiments.

[0143] The terminal device is equipped with a plurality of antenna elements, and the partial channel matrix H main When the dimension of is two or more, the partial channel matrix H main It is not essential that the orthogonalization be performed using the transmission weights on the base station device side. For example, if signal separation between different terminal devices is possible using a transmission weight generation method such as block diagonalization, desired signal separation within the same terminal device (suppression of inter-stream interference) can be handled by signal processing on the terminal device side.

[0144] Furthermore, in the OFDM modulation method, all subcarriers are used for communication with the same terminal device, and therefore the transmit and receive weights (averaged transmit and receive weight vectors and real-time transmit and receive weight matrices) used at this time are transmit and receive weights for the terminal device with a common combination for all subcarriers. However, in OFDMA (Orthogonal Frequency-Division Multiple Access), different combinations of allocations to terminal devices are gathered together in a patchwork pattern on the time axis and frequency axis, so it is necessary to use transmit and receive weights for the assigned terminal device for each time (OFDM symbol) and frequency (subcarrier). However, apart from this difference, OFDM and OFDMA can be processed in exactly the same way, and although this specification has mainly described OFDM, the embodiments of the present invention can also be applied to OFDMA in exactly the same way.

[0145] Furthermore, there are various operational variations for SC-FDE. However, in both the received signal processing after the transmitting side multiplies the averaged transmission weight and the signals transmitted from each antenna element are spatially combined, and the received signal processing after the receiving side multiplies the averaged reception weight and the signals of each antenna element are additively combined, the above-described configuration examples are configured to directly apply the processing performed by conventional SC-FDE, and therefore can be applied to all variations of SC-FDE. In this case, instead of signal processing using an OFDM modulation scheme, signal processing using a single carrier is performed, and then, in the downlink, signal components for each frequency component are generated by performing FFT processing on the signal on the time axis of the single carrier. Then, these signal components are regarded as signals of each subcarrier generated using the OFDM modulation scheme and multiplied by the transmission weight generated according to an embodiment of the present invention.

[0146] Similarly, in the uplink, signals obtained by performing FFT processing on received signals are treated in the same manner as in the OFDM modulation method, and are separated by multiplying them by transmission weights generated by an embodiment of the present invention. Then, the separated frequency component signals are converted into single-carrier signals on the time axis by performing IFFT processing. Thus, although there are differences between the OFDM modulation method and SC-FDE in some signal processing, the generation of transmission and reception weights and multiplication processing are common, and the embodiment of the present invention can be applied to either of these signal methods.

[0147] Furthermore, although this specification is written assuming a wideband system such as OFDM or OFDMA, the embodiments of the present invention are also applicable to narrowband systems.

[0148] Furthermore, for the sake of convenience, this specification does not distinguish between "row vectors" and "column vectors." In a strict mathematical notation that unifies the direction of vector arrangement, a symbol such as "transpose" should be used. However, the information required to implement this invention is the value of each component of a vector, and whether the vector is a row vector or a column vector does not make much sense. Therefore, for ease of understanding, this specification does not distinguish between "row vectors" and "column vectors."

[0149] For example, the channel vector h in Eq. (3) i is a row vector, and the transmit weight vector w j is a column vector. As shown in FIG. 11, the extended partial channel matrix H' used for calculating the transmission weights in the embodiment of the present invention is sub is a matrix in which the channel vectors and predicted channel vectors of terminal devices not subject to transmission weight calculation are arranged in each row. That is, in the case of the downlink, the partial channel matrix H sub The predicted channel vector of the row vector is added to

[0150] On the other hand, in the case of the uplink, the channel vector of each terminal device is a column vector, and the receiving weight vector is also a column vector. Therefore, the extended partial channel matrix H' used to calculate the receiving weight is sub is a matrix in which the channel vectors and predicted channel vectors of terminal devices not subject to receiving weight calculation are arranged in each column. That is, in the case of the uplink, the partial channel matrix H sub The predicted channel vector of the column vector is added to

[0151] According to the above-described embodiment, the wireless communication system includes a base station device having a plurality of antenna elements and a plurality of terminal devices, and the base station device and the terminal devices can perform spatial multiplexing transmission on the same frequency at the same time. For example, the base station device is base station device 80 in the embodiment, and the terminal device is terminal device 802 in the embodiment. The above-described base station device includes a weight calculation unit. For example, the weight calculation unit is transmission weight processing unit 130 in the embodiment.

[0152] The weight calculation unit calculates weight vectors for spatial multiplexing transmission to multiple terminal devices based on a channel matrix that arranges channel vectors generated from channel information between the antenna elements of the terminal devices or virtual antenna elements obtained by combining the antenna elements and the antenna elements provided by the base station device, and additional channel vectors associated with antenna elements of the terminal devices that are different from the channel vectors.

[0153] The weight calculation unit selects, as an additional channel vector of the terminal device, a second channel vector that has a low degree of mutual similarity with the first channel vector from among one or more second channel vectors acquired before the first channel vector, which is the channel vector of the terminal device, and calculates a weight vector.

[0154] In the above base station device, the weight calculation unit may calculate the mutual similarity between the first channel vector and the second channel vector based on the absolute value of the inner product of the first channel vector and the second channel vector, or based on a cross-correlation value obtained by dividing the absolute value of the inner product by the product of the norm of the first channel vector and the norm of the second channel vector. For example, the first channel vector may be the current channel vector h of the terminal device #i in the embodiment. i (t), and the second channel vector is the past channel vector h i (t-nT), and the absolute value of the inner product is the correlation coefficient γ i(n), and the cross-correlation value is the correlation coefficient γ i (n).

[0155] In the above base station device, the weight calculation unit may calculate the mutual similarity between the first channel vector and the second channel vector based on an approximate correlation value obtained by dividing the absolute value of the inner product of the first channel vector and the second channel vector by the norm of either the first channel vector or the second channel vector or by the square of the norm of either the first channel vector or the second channel vector. For example, the approximate correlation value may be calculated using a correlation coefficient γ i (n).

[0156] In the above base station device, the weight calculation unit may calculate the mutual similarity between the first channel vector and the second channel vector based on the norm of the difference vector between the first channel vector and the second channel vector, or based on a value obtained by dividing the norm of the difference vector by the norm of either the first channel vector or the second channel vector. For example, the difference vector is a difference vector d i (n).

[0157] In addition, in the above-mentioned base station device, the weight calculation unit may determine the weight vectors excluding the weight vectors calculated before the point at which mutual similarities higher than a predetermined value appear more than a predetermined number of times.

[0158] The above-described processing may be performed by recording a program for implementing the functions of the base station device 80 in the embodiment on a computer-readable recording medium, and then loading and executing the program recorded on the recording medium into a computer system. The term "computer system" as used herein includes hardware such as an OS and peripheral devices. The term "computer system" also includes a WWW system equipped with a homepage provision environment (or display environment). The term "computer-readable recording medium" refers to portable media such as flexible disks, optical magnetic disks, ROMs, and CD-ROMs, as well as storage devices such as hard disks built into computer systems. The term "computer-readable recording medium" also refers to devices that retain a program for a certain period of time, such as volatile memory (RAM) within a computer system that serves as a server or client when the program is transmitted via a network such as the Internet or a communication line such as a telephone line.

[0159] The program may also be transmitted from a computer system storing the program in a storage device or the like to another computer system via a transmission medium or by transmission waves in the transmission medium. Here, the "transmission medium" that transmits the program refers to a medium that has the function of transmitting information, such as a network (communication network) such as the Internet or a communication line (communication line) such as a telephone line. The program may also be a program that realizes part of the above-mentioned functions. Furthermore, the program may be a so-called differential file (differential program) that can realize the above-mentioned functions in combination with a program already recorded in the computer system. [Industrial Applicability]

[0160] The present invention can be applied to a wireless communication system in which a base station device having a plurality of antenna elements communicates with a plurality of terminal devices using the same frequency channel. [Explanation of symbols]

[0161] 80...base station device, 81...transmitting unit, 85...receiving unit, 87...interface circuit, 88...MAC layer processing circuit, 130...transmitting weight processing unit, 132...channel information storage circuit, 133...MU-MIMO transmit weight calculation circuit, 135...channel similarity calculation circuit, 136...channel information selection circuit, 801...base station device, 802, 802-1 to 802-k, 802-L...terminal device, 811-1, 811-L...transmitting signal processing circuit, 812-1 to 812-K...addition and synthesis circuit, 813-1 to 813-K...GI providing circuit, 814-1 to 814-K...D / A converter, 815...local oscillator, 816-1 to 816-K...mixer, 817-1 to 817-K...filter, 818-1 to 818-K...high power amplifier, 819-1 to 819-K...antenna elements, 820...communication control circuit, 830...transmission weight processing unit, 831...channel information acquisition circuit, 832...channel information storage circuit, 833...multi-user MIMO transmission weight calculation circuit, 851-1 to 851-K...antenna elements, 852-1 to 852-K...low-noise amplifier, 853...local oscillator, 854-1 to 854-K...mixer, 855-1 to 855-K...filter, 856-1 to 856-K...A / D converter, 857-1 to 857-K...FFT circuit, 858-1 to 858-L...reception signal processing circuit, 860...reception weight processing unit, 861...channel information estimation circuit, 862...multi-user MIMO reception weight calculation circuit, 881...scheduling processing circuit

Claims

1. A base station apparatus in a wireless communication system including a base station apparatus having a plurality of antenna elements and a plurality of terminal apparatuses, wherein the base station apparatus and the terminal apparatuses can perform spatial multiplexing transmission on the same frequency at the same time, a weight calculation unit that calculates weight vectors for performing spatial multiplexing transmission for a plurality of the terminal devices based on a channel matrix in which first channel vectors for a plurality of the terminal devices, which are generated based on channel information between the antenna elements of the terminal devices or virtual antenna elements obtained by combining the antenna elements and the antenna elements provided in the base station device, and second channel vectors for a plurality of the terminal devices, which are associated with antenna elements of the terminal devices different from the first channel vectors, are arranged. Equipped with The weight calculation unit uses, as the second channel vector of each of the plurality of terminal devices arranged in the channel matrix, one or more of the second channel vectors that are channel vectors of a terminal device included in the plurality of terminal devices and that are acquired before the first channel vector of the terminal device, the second channel vector having a low degree of mutual similarity with the first channel vector. Base station equipment.

2. The weight calculation unit calculates the mutual similarity between the first channel vector and the second channel vector based on an absolute value of an inner product of the first channel vector and the second channel vector, or based on a cross-correlation value obtained by dividing the absolute value of the inner product by a product of a norm of the first channel vector and a norm of the second channel vector. The base station device according to claim 1 .

3. The weight calculation unit calculates the mutual similarity between the first channel vector and the second channel vector based on an approximate correlation value obtained by dividing the absolute value of an inner product of the first channel vector and the second channel vector by the norm of either the first channel vector or the second channel vector or by the square of the norm of the either one of the first channel vector and the second channel vector. The base station device according to claim 1 .

4. The weight calculation unit calculates the mutual similarity between the first channel vector and the second channel vector based on a norm of a difference vector between the first channel vector and the second channel vector, or based on a value obtained by dividing the norm of the difference vector by the norm of either the first channel vector or the second channel vector. The base station device according to claim 1 .

5. The weight calculation unit calculates the weight vector based on the channel matrix excluding the second channel vector acquired before a time point at which the mutual similarity having a value higher than a predetermined value appears more than a predetermined number of times. The base station device according to claim 1 .

6. A weight generation method executed by a base station device in a wireless communication system including a base station device having a plurality of antenna elements and a plurality of terminal devices, wherein the base station device and the terminal devices can perform spatial multiplexing transmission on the same frequency at the same time, the weight generation method comprising: a step of calculating weight vectors for performing spatial multiplexing transmission for the plurality of terminal devices based on a channel matrix in which first channel vectors for the plurality of terminal devices, which are generated based on channel information between the antenna elements of the terminal devices or virtual antenna elements obtained by combining the antenna elements, and the antenna elements of the base station device, and second channel vectors for the plurality of terminal devices, which are associated with the antenna elements of the terminal devices different from the first channel vectors, are arranged; As the second channel vector of each of the plurality of terminal devices arranged in the channel matrix, one or more of the second channel vectors, which are channel vectors of a terminal device included in the plurality of terminal devices and acquired before the first channel vector of the terminal device, are used, and the second channel vector has a low degree of similarity to the first channel vector; A weight generation method having the following.

7. A wireless communication system comprising a base station device having a plurality of antenna elements and a plurality of terminal devices, wherein the base station device and the terminal devices can perform spatial multiplexing transmission at the same time on the same frequency, The base station device a weight calculation unit that calculates weight vectors for performing spatial multiplexing transmission for a plurality of the terminal devices based on a channel matrix in which first channel vectors for a plurality of the terminal devices, which are generated based on channel information between the antenna elements of the terminal devices or virtual antenna elements obtained by combining the antenna elements and the antenna elements provided in the base station device, and second channel vectors for a plurality of the terminal devices, which are associated with antenna elements of the terminal devices different from the first channel vectors, are arranged. Equipped with The weight calculation unit uses, as the second channel vector of each of the plurality of terminal devices arranged in the channel matrix, one or more second channel vectors that are channel vectors of one terminal device included in the plurality of terminal devices and that are acquired before the first channel vector of the one terminal device, the second channel vector having a low degree of similarity with the first channel vector. Wireless communication system.

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