Quasi-resonant switching power supply

Abstract

To improve accuracy of bottom detection of a pseudo resonance switching power supply device.SOLUTION: A control section 10 of a pseudo resonance switching power supply device 1 comprises: a comparator CP1 which compares a winding voltage V14 of a secondary coil n2 of a reactor L1 with a reference voltage Vref2; and a bottom detection section 13 which detects an edge of an output signal of the comparator CP1 and outputs a bottom signal. The bottom detection section 13 changes a voltage value of the reference voltage Vref2 to a lower limit value after the first bottom detection in such a manner that a ringing voltage of the winding voltage V14 crosses the varied reference voltage.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] The present invention relates to a quasi-resonant switching power supply device whose switching frequency varies depending on the load, and more particularly to a technique for reducing switching loss in a quasi-resonant switching power supply device under a light load. [Background technology]

[0002] A known example of a conventional quasi-resonant switching power supply control circuit is the control circuit for a quasi-resonant switching power supply that enters switching-on operation in synchronization with the bottom timing detected by a bottom detection circuit under light load conditions, as described in Patent Document 1. This control circuit is characterized by having a function for counting a dummy signal in place of a bottom detection signal when the bottom detection count reaches a predetermined number of times under light load conditions, before the bottom detection circuit can no longer detect the bottom. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] Patent No. 6037207 Summary of the Invention [Problem to be solved by the invention]

[0004] However, the switching power supply device described in Patent Document 1 needs to include an oscillator that generates a dummy signal, and needs to correspond to the ringing frequency with a predetermined frequency, which means that accurate bottom timing may not be achieved even based on the dummy signal. Furthermore, if the processing time from bottom detection to output of the turn-on signal by the control circuit is long, it is necessary to anticipate this processing time and perform detection using the ringing waveform just before the bottom. An object of the present invention is to provide a switching power supply device that can be turned on accurately at the bottom of a ringing waveform. [Means for solving the problem]

[0005] In order to solve the above problem, the present invention provides a bottom detection unit that detects the bottom of the ringing voltage by comparing the ringing voltage with a reference voltage and outputs a bottom signal, and the bottom detection is performed when the load is light. Department A control circuit for a quasi-resonant switching power supply that starts a switching-on operation in accordance with the bottom timing detected by As the ringing voltage attenuates over time, the voltage value of the reference voltage is changed to a lower limit value; The ringing voltage and the varied reference voltage are caused to intersect with each other. [Effects of the Invention]

[0006] According to the present invention, by changing the voltage value of the reference voltage to the lower limit value as the ringing voltage decays over time, it is possible to reliably detect the bottom timing taking into account the processing time of the control circuit. Furthermore, bottom detection can be performed reliably without increasing the circuit scale of the control circuit of the quasi-resonant switching power supply device. [Brief explanation of the drawings]

[0007] [Figure 1] 1 is a circuit block diagram of a switching power supply device according to a first embodiment of the present invention. [Figure 2] FIG. 2 is a diagram showing an example of a characteristic diagram of a threshold variable voltage of the switching power supply device according to the first embodiment of the present invention. [Figure 3] 3 is a diagram showing the operating waveforms of each part under a light load condition in the quasi-resonant switching power supply device 1 according to the first embodiment of the present invention. FIG. [Figure 4] 1 is a circuit block diagram showing an application example of the switching power supply device according to the first embodiment of the present invention. [Figure 5] FIG. 5 is a diagram showing an example of a characteristic diagram of a threshold variable voltage of the switching power supply device shown in FIG. [Figure 6] FIG. 5 is a circuit block diagram of a switching power supply device according to a second embodiment of the present invention. [Figure 7]4 is a timing chart for explaining quasi-resonance and bottom skip control of the switching power supply device according to the first embodiment of the present invention. DETAILED DESCRIPTION OF THE INVENTION

[0008] Several embodiments of the switching power supply of the present invention will be described in detail below with reference to the drawings. I will explain in detail. [Example]

[0009] FIG. 1 is a circuit diagram of a quasi-resonant switching power supply device according to a first embodiment of the present invention. The quasi-resonant switching power supply device 1 shown in FIG. 1 is a switching power supply device that performs full-wave rectification of AC voltage from an AC power source AC to convert it into DC voltage, and controls the DC voltage via a quasi-resonant switching circuit so as to adjust the Iled current flowing through the load, LED 40, based on a signal from an external dimmer 30. 1, a full-wave rectifier circuit DB full-wave rectifies an AC voltage from an AC power source AC and outputs a full-wave rectified voltage Vin to both ends of a series circuit of resistors R1 and R2. The configuration consisting of the AC power source 1 and the full-wave rectifier circuit DB can also be regarded as a DC power source that outputs a DC input voltage, such as a battery.

[0010] A series circuit of reactor L1 and switching element Q1 is connected across the series circuit of resistors R1 and R2. Switching element Q1 is a MOSFET, and a series circuit of diode D1 and capacitor C2 is connected between the drain and source of switching element Q1. A series circuit of resistors R5 and R6 is connected across capacitor C2.

[0011] This switching power supply rectifies AC voltage from an AC power source AC and converts it into a DC input voltage. The DC input voltage is switched through a series circuit consisting of a reactor L1 and a switching element Q1, and the resulting voltage is rectified and smoothed by a rectifying and smoothing circuit consisting of a diode D1 and a capacitor C2 to obtain a predetermined output voltage Vout. The output voltage Vout is connected to an LED 40. The current Iled flowing through the LED 40 is dimmed via the control unit 10 in accordance with a command value of the external dimming device 30.

[0012] The control unit 10 is made up of a microcontroller unit (MCU) that calculates an input voltage signal obtained by dividing the DC input voltage Vin using resistors R1 and R2, an output voltage signal obtained by dividing the output voltage Vout using resistors R5 and R6, and the on-time of the switching element Q1 from the aforementioned signals, and controls the on-off of the switching element Q1 based on the on-time obtained, thereby keeping the output voltage at a predetermined value and improving the power factor.

[0013] The control unit 10 includes analog-to-digital converters ADC1, ADC2, and ADC3, a feedback control unit 11, a comparator 12, a bottom counting unit 13, a bottom skip control unit 14, a PWM waveform forming unit 15, and reference voltages Vref1 and Vref2.

[0014] The analog-to-digital converter ADC1 converts the voltage obtained by dividing the full-wave rectified voltage Vin by the resistors R1 and R2 into a digital value V11 and outputs the digital value V11 to the bottom skip control unit .

[0015] The analog-to-digital converter ADC2 converts the voltage obtained by dividing the output voltage Vout by resistors R5 and R6 into a digital value V12 and outputs it to the feedback control unit 11 and bottom skip control unit .

[0016] The analog-to-digital converter ADC3 converts the voltage of the resistor R7, which detects the LED current Iled, into a digital value V13 and outputs it to the feedback control unit 11.

[0017] The comparator 12 has a role of detecting the bottom timing of the ringing waveform of the reactor L1, and receives the winding voltage of the secondary winding n2 of the reactor L1 via the diode D2 and resistors R3 and R4.

[0018] Feedback control unit 11 calculates the on-time of switching element Q1 in the next switching cycle based on the error between a reference value Vref and either a digital value V12 based on the output voltage Vout from analog-digital converter ADC2 or a digital value V13 based on the LED current Iled from analog-digital converter ADC3, whichever is larger. (In steady-state operation, the on-time is calculated based on the error between the reference value Vref and digital value V13 based on the LED current Iled from ADC3.) The on-time is controlled by proportional-plus-integral (PI) control, and by slowing down the response time, the on-time in one cycle of DC input voltage Vin (for example, times t1 to t2 and t3 to t4) is kept constant, as shown in FIG.

[0019] As a result, the average value of the input current IIN becomes proportional to the DC input voltage Vin, and the power factor can be improved.

[0020] The bottom skip control unit 14 calculates the off time Toff of the switching element Q1 in the next switching period based on the on time Ton of the switching element Q1, the DC input voltage Vin, and the output voltage Vout (or the LED current Iled).

[0021] The details of the calculation process of the bottom skip control unit 14 for the switching element Q1 in the boost type critical mode (a mode in which the current flowing through the reactor L1 is zero) will be described. When the on-time is Ton, the peak current Ip flowing through the reactor L1 can be calculated using equation (1).

[0022] Ip = Vin × Ton / L … (1) Vin is the DC input voltage, and L is the inductance value of the reactor L1 (primary winding n1).

[0023] When the current peak is Ip, the off time Toff for transition to critical mode operation is given by equation (2): Required.

[0024] Ip-(Vout-Vin)×Toff / L=0 Ip=(Vout-Vin)×Toff / L Toff=Ip / (Vout-Vin) / L =(Vin×Ton / L) / (Vout-Vin) / L =(Vin × Ton) / (Vout-Vin) …(2) From equation (2), the timing when the reactor current becomes zero is determined by the DC input voltage Vin and the output voltage The difference between the output voltage Vout and the peak current Ip is determined by the inductance L. The peak current Ip in the critical conduction mode is determined by the DC input voltage Vin, the on-time Ton, and the It is determined by the inductance value L.

[0025] Therefore, the timing when the reactor current becomes zero is when the DC input voltage Vin and the output voltage V It can be calculated from the on-time Ton. Therefore, the bottom skip control unit 14 can calculate the off time of the switching element Q1 by determining the timing at which the reactor current becomes zero from the on time Ton of the switching element Q1, the DC input voltage Vin, and the output voltage Vout.

[0026] The bottom skip control unit 14 corresponds to the calculation unit / control unit of the present invention, and calculates the switching frequency f of the switching element Q1 based on the switching period T of the switching element Q1, which is the sum of the calculated off-time and on-time. Note that in the case of a light load, the bottom skip control unit 14 controls the switching element Q1 so that the frequency becomes the reference frequency based on the result of comparing the calculated switching frequency f with the reference frequency.

[0027] For example, if the command value of the external dimmer 30 sets the current Iled flowing through the LED 40 to a minute current, the switching power supply 1 enters a light load state, causing the switching frequency f calculated by the bottom skip control unit 14 to increase. If the calculated frequency exceeds a preset reference frequency, the bottom skip control unit 14 adjusts the frequency to match the cycle calculated based on the reference frequency. This results in a long off-time being given, even though the actual off-time is shorter than the cycle calculated based on the reference frequency. As a result, ringing occurs in the voltage (drain-source voltage) Vds across the switching element Q1. The bottom skip control unit 14 sets the number of bottom skips (one to multiple) based on the command value of the external dimmer 30 and the DC input voltage Vin, counts the bottom detection signals from the bottom counter, and turns on the switching element Q1.

[0028] The bottom skip control unit 14 obtains a signal V14 from the secondary winding n2 of the reactor L1 shown in FIG. 1 via resistors R3 and R4, which indicates the bottom timing when the voltage between both ends (drain-source voltage) Vds undergoes damped oscillation while the switching element Q1 is off. The signal V14 from the secondary winding n2 of the reactor L1 is input to the non-inverting terminal of the comparator 12. A reference voltage Vref2 is connected to the inverting terminal of the comparator 12. Here, the signal V14 is compared with the reference voltage Vref2, and the result of the comparison is output to the bottom counting unit 13 as a bottom detection signal. The bottom counting unit 13 detects the falling edge of the bottom detection signal, counts the bottom count, and outputs the result to the bottom skip control unit 14 as a bottom timing signal.

[0029] The PWM waveform forming unit 18 generates a PWM signal based on the signal from the bottom skip control unit 14. The generated PWM signal is transmitted to the switching element Q1 via the driver (Drv) 20. Output to the gate.

[0030] As shown in FIG. 2, the bottom counter 13 attenuates the reference voltage Vref2 immediately after detecting the first falling edge of the bottom detection signal, thereby changing the threshold of the comparator 12. This reference voltage Vref2 is varied from its initial value over time from immediately after the first down edge of the bottom detection signal is detected as shown in Figures 2(a) and 2(c), and is reduced to a predetermined Min voltage, which is the lower limit. Alternatively, immediately after detecting the first down edge of the bottom detection signal as shown in Figures 2(b) and (d), the voltage of the reference voltage Vref2 is changed in stages by a predetermined threshold value along with the elapsed time or the bottom count from the first time, and is lowered to a predetermined Min voltage, which is the lower limit value. 2(c) and 2(d), the bottom counting unit 13 changes the threshold value (Vref2) of the comparator 12 in accordance with the decay of the ringing voltage when the voltage across both terminals (drain-source voltage) Vds undergoes damped oscillation while the switching element Q1 is off, thereby enabling reliable bottom detection in accordance with the ringing voltage. Furthermore, because the threshold value (Vref2) of the comparator 12 is changed in accordance with the decay of the ringing voltage, the initial value of the threshold value (Vref2) can be started from a high value, which has the advantage of reducing false detection due to noise. Furthermore, the variable operation of the reference voltage Vref2 can be set without using a special circuit by using a microcontroller unit (MCU), so it can be realized without increasing the circuit scale.

[0031] 3 is a diagram showing the operating waveforms of each part under a light load condition in the quasi-resonant switching power supply 1 according to the first embodiment of the present invention. From the top of the waveforms shown in FIG. 3, the voltage waveform of the secondary winding n2 of the reactor L1, the waveform of the signal V14, and the drain current waveform of the switching element Q1 are shown. The slope of the attenuation of the ringing of the signal V14 is slightly larger than the slope of the attenuation of the ringing of the voltage of the secondary winding n2 of the reactor L1. For reference, the linearly attenuating slope of the reference voltage Vref2 is shown on the waveform of the signal V14 in FIG. 3, but in setting it, it is desirable to correct the slope of the reference voltage Vref2 to match the slope of the signal V14.

[0032] FIG. 4 is a circuit block diagram showing an application example of the switching power supply device according to the first embodiment of the present invention. FIG. 5 is an example of a characteristic diagram of the threshold variable voltage of the switching power supply device shown in FIG. The difference between the application example in Figure 4 and Figure 1 is that in Figure 4, the polarity of the secondary winding n2 of the reactor L1 in Figure 1 is reversed, and the bottom signal is obtained by positively biasing the negative voltage signal with a reference voltage (such as Vref1) via resistor R4. Note that other parts with the same configuration are denoted with the same reference symbols. Here, as shown in FIG. 5, the bottom counting section 13 increases the reference voltage Vref2 immediately after detecting the first down edge of the bottom detection signal, thereby changing the threshold value of the comparator 12. This reference voltage Vref2 is changed from its initial value over time from immediately after the first down edge of the bottom detection signal as shown in FIG. 5(a) is detected, and is increased up to a predetermined Max voltage, which is the upper limit. Alternatively, immediately after detecting the first down edge of the bottom detection signal as shown in Figure 5(b), the voltage of the reference voltage Vref2 is changed in stages by a predetermined threshold value along with the elapsed time or the bottom count from the first time, and is raised to a predetermined Max voltage, which is the upper limit value. This allows accurate bottom timing to be detected in the same way as in FIG. [Example]

[0033] FIG. 6 is a circuit block diagram of a switching power supply device according to a second embodiment of the present invention. 6 is a quasi-resonant switching power supply device 1b in which the non-insulated buck converter configuration of Example 1 is replaced with an isolated flyback converter configuration. The same reference numerals are used to denote parts with the same configuration.

[0034] In the quasi-resonant switching power supply device 1b shown in Fig. 6, the reactor L1 in Fig. 1 is replaced with a transformer T1. Accordingly, a snubber circuit (diode D3, capacitor C3, and resistor R8) for resetting the primary winding P1 of the transformer T1 is added to the primary side circuit. Furthermore, the tertiary winding P2 of the transformer T1 corresponds to the secondary winding n2 of the reactor L1. In the control unit 10b, the analog-to-digital converter ADC3 is omitted compared to the control unit 10, and the analog-to-digital converter ADC2 collectively receives the feedback signal and the error signal from the secondary side feedback control unit via the photocoupler PC1.

[0035] The rectifying and smoothing circuit, consisting of a diode D1 and a capacitor C2 connected to the secondary winding S1 of the transformer T1, is the same as in the first embodiment. The secondary-side circuit differs from the first embodiment in that a secondary-side feedback control unit 50, a reference voltage Vref3, and a photocoupler PC1 have been newly added to provide an isolated configuration. The secondary-side feedback control unit 50 compares the detection signal of either the output voltage Vout or the LED current Iled, whichever is larger, with the reference value Vref3, and outputs the result as a feedback signal. The feedback signal output from the secondary-side feedback control unit 50 is sent to the analog-to-digital converter ADC2 of the primary-side control unit 10a via the newly added photocoupler PC1.

[0036] In the flyback converter configuration of the second embodiment, in a light load state, a ringing voltage occurs in the primary winding P1 of the transformer T1 while the switching element Q1 is off, which causes a similar ringing voltage to occur in the tertiary winding P2 of the transformer T1, thereby achieving the same effect as in the first embodiment.

[0037] The present invention is not limited to the quasi-resonant switching power supply devices of the first and second embodiments. [Explanation of symbols]

[0038] 1, 1a, 1b Quasi-resonant switching power supply 2 AC power supply 10, 10a, 10b Control unit 11 Feedback control section 12 Comparators 13 Bottom counting unit 14 Bottom skip control section 15 PWM waveform forming section 20 Drivers 30 External dimmer 40 LED ADC1, ADC2, ADC3 analog-to-digital converters C1, C2, C3 capacitors D1, D2, D3 diodes DB full wave rectifier circuit L1 reactor Q1 switching element R1~R9 Resistors T1 transformer Vref1, Vref2, Vref3 reference voltages

Claims

1. a bottom detection unit that detects the bottom of the ringing voltage by comparing the ringing voltage with a reference voltage and outputs a bottom signal; A control circuit for a quasi-resonant switching power supply that starts a switching-on operation in accordance with a bottom timing detected by the bottom detection unit under a light load, A control circuit for a quasi-resonant switching power supply, characterized in that the voltage value of the reference voltage is changed to a lower limit value as the ringing voltage attenuates over time, so that the ringing voltage and the changed reference voltage intersect.

2. 2. The control circuit for a quasi-resonant switching power supply according to claim 1, wherein the voltage value of the reference voltage is changed to a second set voltage lower than the initial set voltage in proportion to the time elapsed after the ringing voltage intersects with the initial set voltage of the reference voltage.

3. 2. The control circuit for a quasi-resonant switching power supply according to claim 1, wherein the voltage value of the reference voltage is changed in a stepwise manner up to a second set voltage lower than the initial set voltage each time the ringing voltage intersects with the initial set voltage of the reference voltage and a subsequent bottom is detected.

4. 2. The control circuit for a quasi-resonant switching power supply according to claim 1, wherein the voltage value of the reference voltage is changed in proportion to the time elapsed after the ringing voltage intersects with the initial setting voltage of the reference voltage, up to a third setting voltage higher than the initial setting voltage.

5. 2. The control circuit for a quasi-resonant switching power supply according to claim 1, wherein the voltage value of the reference voltage is changed in a stepwise manner up to a third set voltage higher than the initial set voltage each time the ringing voltage intersects with the initial set voltage of the reference voltage and a subsequent bottom is detected.

6. 6. The control circuit of the quasi-resonant switching power supply according to claim 1, wherein the control circuit of the quasi-resonant switching power supply is digitally controlled.

7. A semiconductor device comprising a control circuit for the quasi-resonant switching power supply device according to any one of claims 1 to 6, which is configured as an integrated circuit.

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