Dual three-phase permanent magnet synchronous motor model prediction torque control method based on time-sharing control

Abstract

The invention discloses a dual three-phase permanent magnet synchronous motor model prediction torque control method based on time-sharing control, and the method comprises the steps: firstly constructing a virtual voltage vector according to a principle that a harmonic space current is suppressed to be zero; and secondly, stator flux linkage and torque decoupling control is realized by utilizing the stator flux linkage and torque decoupling characteristics in a stator flux linkage oriented two-phase rotating coordinate system. And then, selecting a pair of mutually vertical virtual voltage vectors to respectively control the stator flux linkage and the torque. Finally, the duty ratio of the virtual voltage vector is calculated, and stator flux linkage and torque time-sharing control is achieved. By adopting the method, the setting of the weight coefficient and the repeated iteration of the cost function are avoided, the stator flux linkage and the torque ripple are reduced, the current harmonic is inhibited, and the steady-state and dynamic performance of the system is improved.

Description

Technical Field

[0001] The invention relates to a dual three-phase permanent magnet synchronous motor model prediction torque control method based on time-sharing control, belonging to the technical field of motor drive and control. Background Art

[0002] Permanent magnet synchronous motor (PMSM) has been widely used in industrial transmission, household appliances, aerospace and other fields due to its high power density, wide speed regulation range and high efficiency. Field-oriented control (FOC) and direct torque control (DTC) are the two most commonly used control strategies for PMSM. Compared with FOC and DTC, finite control set-model predictive control (FCS-MPC) selects the optimal voltage vector through online optimization, which is more direct, accurate and effective in vector selection. It has the characteristics of simple principle and easy handling of system nonlinear constraints, so it has attracted widespread attention. The model predictive direct torque control strategy uses stator flux and torque error to construct the cost function, selects the voltage vector that minimizes the objective function, and can effectively reduce the torque ripple, but its weight coefficient adjustment is complex and the cost function needs to be iterated repeatedly.

[0003] In order to reduce the number of iterations of the cost function, the Chinese invention patent "A simplified method for predicting PMSM direct torque control based on a finite state set model" (patent number CN110266238B) discloses a finite set model predictive control method. This method uses the current torque angle, sector position, etc. as constraints to simplify the set of alternative voltage vectors. However, the patent constructs a cost function through stator flux and torque amplitude, and the weight coefficient still needs to be adjusted.

[0004] In view of the complex problem of adjusting the weight coefficient of FCS-MPC, IET Electric Power Applications, 2017, 11(5):749-760 (Finite control set predictive torque control for induction motor drive with simplified weighting factor selection using TOPSIS method) introduces the superiority and inferiority distance method (TOPSIS) of multi-objective decision theory into the cost function to eliminate the weight coefficient, but this method has a large amount of calculation and a relatively complicated implementation process. Therefore, it is necessary to further study the model predictive direct torque control strategy to further reduce the amount of calculation and improve its steady-state performance under the premise of solving the complex adjustment of the weight coefficient of the cost function. Summary of the invention

[0005] The purpose of the present invention is to solve the problems of difficulty in setting weight coefficients and large iterative calculation load in the prior art, and to propose a control method that effectively improves the driving performance of a permanent magnet synchronous motor, and controls the stator flux and torque respectively, which not only avoids the difficulty in setting weight coefficients, but also reduces the amount of calculation. To this end, a dual three-phase permanent magnet synchronous motor model prediction torque control method based on time-sharing control is proposed.

[0006] The present invention achieves the above technical objectives through the following technical means.

[0007] A dual three-phase permanent magnet synchronous motor model prediction torque control method based on time-sharing control comprises the following steps:

[0008] Step 1: Use the Clarke transformation matrix to transform the phase voltages corresponding to different switching states of the power devices in the voltage source inverter driving the dual three-phase permanent magnet synchronous motor into a two-phase stationary coordinate system to obtain the base voltage vectors of the fundamental wave plane α-β and the harmonic wave plane xy.

[0009] Step 2, based on the principle of suppressing the harmonic current in the xy plane to zero, the large and medium base voltage vectors with the same direction in the α-β plane are used to synthesize a virtual voltage vector, and the amplitude of the virtual voltage in the xy plane is made zero. The action time of the large base voltage vector in the fundamental plane is 0.732T s , the action time of the base voltage vector is 0.268T s , and then obtain the virtual voltage vector synthesized by the above base voltage vector in the fundamental wave plane; the virtual voltage vector V i 3 (i＝1-12) The amplitude is 0.5977U dc (U dcrepresents bus voltage).

[0010] Step 3: Divide the α-β plane into 12 sectors by taking the angle bisector between two adjacent virtual voltage vectors as the dividing line.

[0011] Step 4: To reduce the computational complexity and difficulty of the reference stator voltage and realize the decoupling control of stator flux and torque, define a two-phase rotating coordinate system (mt coordinate system) based on the orientation of stator flux; differentiate and discretize the torque equation with respect to time, and combine the equation of stator voltage in the mt coordinate system to calculate the reference voltage u of stator flux and torque. sm * 、u st * .

[0012] Step 5: Considering the one-step delay of digital control, the reference voltage equation obtained in step 4 is delayed and compensated. The value at time k+2 is used as the reference value to obtain the reference voltage u for controlling the stator flux and torque. sm * 、u st * .

[0013] Step 6: According to the reference voltage u sm * 、u st * and the angle between the corresponding voltage vector and the α axis, which are expressed in vector form in the α-β plane as u sm * 、u st * .

[0014] Step 7, according to u sm * 、u st * The relationship between the sector and the voltage vector on the stator flux and torque is established, and a switch table is formulated to select the virtual voltage vector (V FC ) and the virtual voltage vector (V TC ).

[0015] Step 8, construct a duty cycle calculation equation group, and control the stator flux and torque separately within a control cycle by calculating the action time of the two selected virtual voltage vectors, so that the stator flux and torque can follow the reference value at the end of the control cycle.

[0016] In step 9, a virtual voltage vector and a calculated duty cycle are selected from the established switch table and input into a PWM module to generate a PWM waveform to control the on and off of power devices in the voltage source inverter.

[0017] The present invention has the beneficial effects:

[0018] 1) Based on the decoupling characteristics of the stator flux oriented coordinate system, the present invention adopts a virtual voltage vector to independently control the torque and stator flux, which not only avoids the difficulty of setting the weight coefficient, but also realizes the decoupling control of the torque and stator flux, can quickly follow the reference torque and stator flux, improves its following performance, reduces the torque and stator flux pulsation, improves its control accuracy, improves the dynamic response speed, and improves the steady-state performance of the system.

[0019] 2) The present invention divides the space into sectors, and then directly selects the virtual voltage vector in combination with the reference voltage vector position angle. This not only saves the process of determining the minimum distance between the reference voltage vector and the adjacent virtual voltage vector, but also avoids repeated iterative calculations of the cost function. The optimal voltage vector is obtained in a more concise and intuitive manner, greatly reducing the amount of calculation.

[0020] 3) The present invention is based on the deadbeat control concept and calculates the duty cycle corresponding to the selected virtual voltage vector, making the control more precise and further improving the steady-state performance of the system. BRIEF DESCRIPTION OF THE DRAWINGS

[0021] Figure 1 The block diagram of torque control for the dual three-phase permanent magnet synchronous motor model based on time-sharing control.

[0022] Figure 2 It is the virtual voltage vector distribution diagram.

[0023] Figure 3 This is the stator flux vector distribution block diagram.

[0024] Figure 4 (a) is the stator flux trajectory diagram; (b) is the torque trajectory diagram.

[0025] Figure 5 (a) is u sm * 、u st * When the numbers are the same, the duty cycle calculation principle diagram;

[0026] Figure 5 (b) is u sm * 、u st * Duty cycle calculation principle diagram when the signs are different;

[0027] Figure 6 It is the dual three-phase PMSM current simulation waveform.

[0028] Figure 7 (a) is the torque simulation waveform of the dual three-phase PMSM in steady state; (b) is the stator flux simulation waveform.

[0029] Figure 8 (a) is the dual three-phase PMSM speed simulation waveform; (b) is the load step simulation waveform.

[0030] Fig. 9 It is the simulation waveform of the dual three-phase PMSM flux circle. DETAILED DESCRIPTION

[0031] The present invention will be further described below in conjunction with the accompanying drawings and specific embodiments.

[0032] The control block diagram designed by the present invention is as follows Figure 1 As shown in the figure, the present invention is composed of a double closed-loop control system based on a model predictive control inner loop and a PI control speed outer loop. First, the given speed and the feedback speed are subtracted to obtain the speed error, and the given torque T is obtained through PI adjustment. e * The stator flux ψ is observed by the flux and torque observer s and torque T e . Given torque T e * and estimated torque T e Subtract and get ΔT e ; Given the flux linkage ψ s * and the estimated flux ψ s Subtract and get Δψ s According to ΔT e and Δψ s , combined with the stator flux angle θ s Calculate the reference voltage u related to torque and flux m * 、u t * and express it in the form of a vector in the α-β coordinate system as u m * 、u t * , determine the sector in which it is located in the α-β coordinate system, select the corresponding virtual voltage vector from the formulated switch table and calculate its duty cycle to achieve model predictive torque control of the motor.

[0033] The offset angle between the two sets of windings abc and def of the dual three-phase permanent magnet synchronous motor is 30 degrees. The Clarke transformation matrix shown in formula (1) is used to transform the phase voltages corresponding to different switching states of the power devices in the voltage source inverter driving the dual three-phase permanent magnet synchronous motor into a two-phase stationary coordinate system to obtain the base voltage vectors of the fundamental wave plane α-β and the harmonic wave plane xy.

[0034] Among them Ua , U b , U c , U d , U e , U f Indicates the phase voltage, U α , U β , U x , U y , U z1 , U z2 Represents the voltage of the two-phase stationary coordinate system.

[0035] In order to suppress harmonic current to zero, the large and medium base voltage vectors with the same direction are used in the α-β plane to synthesize the virtual voltage vector to make it zero in the xy plane. Among them, the action time of the large base voltage vector in the α-β plane is 0.732T s , the action time of the base voltage vector is 0.268T s , and then the synthesized virtual voltage vector in the α-β plane is obtained as Figure 2 As shown. The synthesized virtual voltage vector V i 3 (i＝1-12) The amplitude in the α-β plane is 0.5977U dc (U dc = represents the bus voltage). Therefore, the relationship between the virtual voltage vector and the large and medium base voltage vectors is shown in Table 1.

[0036] Table 1 Virtual voltage vector synthesis table

[0037] Vector action time <![CDATA[V1 3 ]]> <![CDATA[V2 3 ]]> <![CDATA[V3 3 ]]> <![CDATA[V4 3 ]]> <![CDATA[V5 3 ]]> <![CDATA[V6 3 ]]> <![CDATA[V7 3 ]]> <![CDATA[V8 3 ]]> <![CDATA[V9 3 ]]> <![CDATA[V 10 3 ]]> <![CDATA[V 11 3 ]]> <![CDATA[V 12 3 ]]> <![CDATA[0.732T s ]]> <![CDATA[V 44 ]]> <![CDATA[V 64 ]]> <![CDATA[V 66 ]]> <![CDATA[V 26 ]]> <![CDATA[V 22 ]]> <![CDATA[V 32 ]]> <![CDATA[V 33 ]]> <![CDATA[V 13 ]]> <![CDATA[V 11 ]]> <![CDATA[V 51 ]]> <![CDATA[V 55 ]]> <![CDATA[V 45 ]]> <![CDATA[0.268T s ]]> <![CDATA[V 65 ]]> <![CDATA[V 46 ]]> <![CDATA[V 24 ]]> <![CDATA[V 62 ]]> <![CDATA[V 36 ]]> <![CDATA[V 23 ]]> <![CDATA[V 12 ]]> <![CDATA[V 31 ]]> <![CDATA[V 53 ]]> <![CDATA[V 15 ]]> <![CDATA[V 41 ]]> <![CDATA[V 54 ]]>

[0038] In the above table, V i 3 (i＝1-12) Represents a virtual voltage vector; V i(i＝0-64) represents the base voltage vector; T s Indicates the sampling period.

[0039] Taking the straight line passing through the center point o and bisecting two adjacent virtual voltage vectors as the sector dividing line, the α-β plane can be equally divided into 12 sectors with an angle of 30°, such as Figure 2 shown.

[0040] In the traditional model predictive torque control calculation, since the magnitude of the reference torque angle is unknown, the components of the reference stator voltage on the dq axis are difficult to calculate. To this end, the torque equation is first differentiated with respect to time and discretized, and then the torque angle increment is moved to the left side of the equation to obtain

[0041]

[0042] Where, δ represents the torque angle, Ls Indicates the inductance value, R s represents stator resistance, P represents pole pair number, T e and T e * denote the torque and reference torque respectively, ψ f represents the permanent magnet flux, T s represents the sampling period, k represents the current time, and k+1 represents the current time.

[0043] In order to reduce the computational complexity and difficulty of the reference stator voltage and realize the decoupling control of the stator flux and torque, a two-phase rotating coordinate system (mt coordinate system) based on the orientation of the stator flux is defined. In the mt coordinate system, the stator flux vector is located on the m-axis, and the torque is located on the t-axis perpendicular to the m-axis, such as Figure 3 At this time, the motor voltage model can be expressed in the mt coordinate system as

[0044]

[0045] Among them, u sm 、u st 、i sm 、i st , sm , st They represent the components of stator voltage, current and stator flux in the mt coordinate system, ω s Represents the electrical angular velocity of the stator flux vector rotation.

[0046] Since the stator flux vector in the defined mt coordinate system is located on the m-axis, ψ sm Equal to the stator flux ψ s , ψ st Equal to zero. Discretize equation (3) and use the value at time k+1 as the reference value. Substitute equation (2) into equation (3) to obtain the reference voltage u for controlling the stator flux and torque: sm * 、u st *

[0047]

[0048] Among them, ω r represents the electrical angular velocity of the rotor, δ * represents the reference torque angle, ψ s * Represents the reference flux linkage.

[0049] Considering the delay of digital control, one-step prediction is used for delay compensation. Formula (4) is iterated once, and the value at time k+2 is used as the reference value to obtain the reference voltage u for controlling the stator flux and torque. sm* 、u st *

[0050]

[0051] According to the reference voltage u sm * 、u st * And the angle θ between the corresponding voltage vector and the α axis m ,θ t , which are expressed in vector form in the α-β plane as

[0052]

[0053] According to u sm * 、u st * The phase angle in the α-β coordinate system determines the sector in which it is located, and then selects the virtual voltage vector (V FC ) and virtual voltage vector (V TC ).

[0054] Table 2 Candidate voltage vector selection table

[0055] <![CDATA[u sm * 、u st * Sector]]> I II III IV V VI VII VIII IX X XI XII <![CDATA[V FC 、In TC ]]> <![CDATA[V1 3 ]]> <![CDATA[V2 3 ]]> <![CDATA[V3 3 ]]> <![CDATA[V4 3 ]]> <![CDATA[V5 3 ]]> <![CDATA[V6 3 ]]> <![CDATA[V7 3 ]]> <![CDATA[V8 3 ]]> <![CDATA[V9 3 ]]> <![CDATA[V 10 3 ]]> <![CDATA[V 11 3 ]]> <![CDATA[V 12 3 ]]>

[0056] V selected by the switch table FC 、V TC The stator flux and torque are controlled respectively, and V is applied FC The stator flux is controlled, the stator flux changes rapidly, and the change in torque is negligible; applying V TC The torque is controlled so that the torque changes rapidly while the change of stator flux is negligible. At the end of each control cycle, the stator flux and torque reach the reference values ​​respectively.

[0057] like Figure 5 As shown, first calculate the angle θ between the selected virtual voltage vector and the reference voltage vector

[0058] θ=θ vv -θ v (7)

[0059] where θ vv ,θ v They respectively represent the angles between the selected virtual voltage vector and the reference voltage vector and the α-axis.

[0060] Secondly, the duty cycle calculation equations are established to calculate the action time of the two selected virtual voltage vectors so that the stator flux and torque can follow the reference flux and torque in each control cycle. Figure 5 As shown, when u sm * 、u st * Same number

[0061]

[0062] Obtain the action time T corresponding to the virtual voltage vector FC 、T TC for

[0063]

[0064] When u sm * 、u st * Different sign time

[0065]

[0066] Obtain

[0067]

[0068] When the system is in steady state, if T TC <T s 、T FC <T s , and (T TC +T FC ) <T s When the zero vector action time T0 is T s -(T TC +T FC ); if T TC <T s 、T FC <T s , and (T TC +T FC )>T s When T TC 、T FC When the system undergoes a speed or load step, the T calculated by the duty cycle TC >T s , then select V TC Acts on the entire control cycle and makes T TC =T s , in order to shorten the dynamic response time of the system and improve the dynamic performance of the system. The vector action time optimization result is shown in formula (12):

[0069]

[0070] The selected virtual voltage vector and action time are input into the PWM module to generate a PWM waveform, control the on and off of the power devices in the inverter, and realize the control of the dual three-phase PMSM.

[0071] When the motor is in steady state, the current, torque and stator flux waveforms are as follows: Figure 6 and 7 The current waveform has good sinusoidality, and the total harmonic distortion (THD) is 4.97%; the pulsation of torque and stator flux is very small. Figure 8 The speed and load step waveforms show that when the speed jumps from 300rpm to 400rpm, there is no obvious overshoot; when the load jumps from 7N·m to 5N·m, the torque tracking is timely and accurate. Fig. 9 It is the stator flux circular waveform, and the stator flux circle is a standard circle. Therefore, it is proved that the algorithm of the present invention has excellent steady-state and dynamic performance.

[0072] In summary, the present invention performs decoupling control on the stator flux and torque in the mt coordinate system based on the idea of ​​stator flux and torque time-sharing control. The stator flux is not affected when the torque is controlled, and the torque is not affected when the stator flux is controlled, thereby realizing the decoupling control of the stator flux and torque. The u sm * 、u st * The position angle is used to determine the sector it is in, and the virtual voltage vector is directly selected from the switch table, which avoids repeated iterations of the cost function and reduces the calculation load of the CPU. The simulation verifies that the invented algorithm can effectively reduce the stator flux and torque pulsation and suppress current harmonics, and has good steady-state and dynamic performance.

[0073] It should be understood that the above embodiments are only used to illustrate the present invention and are not used to limit the scope of the present invention. After reading the present invention, various equivalent modifications of the present invention by those skilled in the art all fall within the scope defined by the claims attached to this application.

Claims

1. A dual three-phase permanent magnet synchronous motor model predictive torque control method based on time-sharing control, characterized in that: The following steps are involved: Step 1, using the Clarke transformation matrix to transform the phase voltages corresponding to different switching states of the power devices in the voltage source inverter driving the dual three-phase permanent magnet synchronous motor into a two-phase stationary coordinate system, and obtain the base voltage vectors of the fundamental wave plane α-β and the harmonic wave plane xy; Step 2, based on the principle of suppressing the harmonic current in the xy plane to zero, the large and medium base voltage vectors with the same direction in the α-β plane are used to synthesize a virtual voltage vector, and the amplitude of the virtual voltage in the xy plane is made zero; the action time of the large base voltage vector in the fundamental plane is 0.732T s , the action time of the base voltage vector is 0.268T s , and then obtain the virtual voltage vector synthesized by the above base voltage vector in the fundamental wave plane; the virtual voltage vector V i 3 (i＝1-12) The amplitude is 0.5977U dc (U dc represents bus voltage); therefore, the relationship between each virtual voltage vector and the large and medium base voltage vectors is shown in the following table In the above table, V i 3 (i＝1-12) Represents a virtual voltage vector; V i(i＝0-64) represents the base voltage vector; T s Indicates the sampling period. Step 3, using the angle bisector between two adjacent virtual voltage vectors as a dividing line, dividing the fundamental wave plane α-β into 12 sectors; Step 4: To reduce the computational complexity and difficulty of the reference stator voltage and realize the decoupling control of stator flux and torque, define a two-phase rotating coordinate system (mt coordinate system) based on the orientation of stator flux; differentiate and discretize the torque equation with respect to time, and combine the equation of stator voltage in the mt coordinate system to calculate the reference voltage u of stator flux and torque. sm * 、u st * ; Step 5: Considering the one-step delay of digital control, the reference voltage equation obtained in step 4 is delayed and compensated. The value at time k+2 is used as the reference value to obtain the reference voltage u for controlling the stator flux and torque. sm * 、u st * Step 6: According to the reference voltage u sm * 、u st * And the angle θ between the corresponding voltage vector and the α axis m ,θ t , which are expressed in vector form in the α-β plane as u sm * 、u st * Step 7, according to u sm * 、u st * The relationship between the sector and the voltage vector on the stator flux and torque is formulated as follows. The virtual voltage vector (V FC ) and virtual voltage vector (V TC ); V selected by the switch table FC 、V TC The stator flux and torque are controlled respectively, and V is applied FC The stator flux is controlled, the stator flux changes rapidly, and the change in torque is negligible; applying V TC The torque is controlled, the torque changes rapidly, and the change of stator flux is negligible. At the end of each control cycle, the stator flux and torque reach the reference value respectively; Step 8, construct a duty cycle calculation equation group, and control the stator flux and torque separately within a control cycle by calculating the action time of the two selected virtual voltage vectors, so that the stator flux and torque can follow the reference value at the end of the control cycle; In step 9, a virtual voltage vector and a calculated duty cycle are selected from the established switch table and input into a PWM module to generate a PWM waveform to control the on and off of power devices in the voltage source inverter.

2. A dual three-phase permanent magnet synchronous motor model predictive torque control method based on time-sharing control according to claim 1, characterized in that: The specific process in step 4 is as follows: Step 4.1, differentiate the torque equation with respect to time and discretize it, and move the torque angle increment to the left side of the equation to obtain Where, δ represents the torque angle, L s Indicates the inductance value, R s represents stator resistance, P represents pole pair number, T e and T e * denote the torque and reference torque respectively, ψ s represents the stator flux, ψ f represents the permanent magnet flux, T s represents the sampling period, k represents the current moment, and k+1 represents the next moment; Step 4.2, define the two-phase rotating coordinate system mt based on the stator flux orientation, so that the m-axis coincides with the stator flux vector, and the t-axis coincides with the torque. The stator voltage in the mt coordinate system can be expressed as Among them, u sm 、u st 、i sm 、i st , sm , st Represent the stator voltage, current and voltage components of the stator flux on the m and t axes respectively, ω s Represents the electrical angular velocity of the stator flux vector rotation; the above formula is discretized, and the value at time k+1 is used as the reference value to obtain the reference voltage u for controlling the stator flux and torque sm * 、u st * Among them, ω r represents the electrical angular velocity of the rotor, δ * represents the reference torque angle, ψ s * Represents the reference flux linkage.

3. The method for predicting torque control of a dual three-phase permanent magnet synchronous motor based on time-sharing control according to claim 1 is characterized in that: The specific process in step 8 is as follows: Step 8.1, calculate the angle θ between the selected virtual voltage vector and the reference voltage vector θ=θ vv -θ v where θ vv ,θ v Respectively represent the angles between the selected virtual voltage vector and the reference voltage vector and the α-axis; Step 8.2, establish the duty cycle calculation equation group, and calculate the action time of the selected virtual voltage vector so that the stator flux and torque can follow the reference value at the end of each control cycle. sm * 、u st * When the signs are the same, the action time T corresponding to the virtual voltage vector is obtained. FC 、T TC When u sm * 、u st * Different sign time Step 8.3, when the system is in steady state, if T TC <T s 、T FC <T s , and (T TC +T FC ) <T s , then the zero vector action time T0 is T s -(T TC +T FC ); if T TC <T s 、T FC <T s , and (T TC +T FC )>T s When T TC 、T FC When the system undergoes a speed or load step, the T calculated by the duty cycle TC >T s , then select V TC Acts on the entire control cycle and makes T TC =T s , in order to shorten the dynamic response time of the system and improve the dynamic performance of the system. The vector action time optimization result is shown in the following formula

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