Motor and control method and device thereof, storage medium and computer program product

Abstract

The invention discloses a motor and a control method and device thereof, a storage medium and a computer program product, and the method comprises the steps: injecting a pulsating voltage with a preset frequency into a d-axis part outputted by a PI controller in a current loop at a zero-speed starting stage and a low-speed operation stage of the motor; in the whole stage of the motor, three-phase current of the motor is obtained; in the whole stage of the motor, in a current loop and a speed loop, according to the three-phase current of the motor, a full-order observation module, a phase locking module and a switching function module are utilized to obtain a rotor position estimation value and a rotating speed estimation value of the motor; and in the whole stage of the motor, starting and running of the motor are controlled according to the rotor position estimation value and the rotating speed estimation value of the motor. According to the scheme, the improved high-frequency injection algorithm is combined with the full-order observer based on the back electromotive force, and the exponential weighting function is adopted to smoothly switch the position and the speed of the motor rotor estimated by the high-frequency injection algorithm and the full-order observer, so that the complexity and the cost are reduced, and the reliability is improved.

Description

TECHNICAL FIELD

[0001] The present application belongs to the technical field of electric machines, and particularly relates to a control method and device of an electric machine, the electric machine, a storage medium and a computer program product, in particular to a control method and device of a full-speed threshold position sensorless permanent magnet synchronous motor, the electric machine, a storage medium and a computer program product. BACKGROUND

[0002] Electric machines (such as permanent magnet synchronous motors) are widely used in industrial control, new energy vehicles, robotics, aerospace and household appliances due to their high efficiency, high power density, low rotational inertia and excellent output characteristics. When building a high-performance control system for a permanent magnet synchronous motor, it is crucial to accurately obtain the position and speed information of the rotor. In related schemes, the control system of the permanent magnet synchronous motor relies on sensors for direct measurement. Common sensors include Hall sensors, optical encoders and rotary transformers. However, the use of sensors inevitably increases the complexity of the control system of the permanent magnet synchronous motor, resulting in reduced reliability and increased cost.

[0003] The above content is only used to assist in understanding the technical solutions of the present application and does not represent the acknowledgement of the above content as prior art. SUMMARY

[0004] The present application aims to provide a control method and device of an electric machine, the electric machine, a storage medium and a computer program product, to solve the problem of the electric machine (such as a permanent magnet synchronous motor) relying on sensors for direct measurement to obtain the position and speed information of the rotor, increasing the complexity and cost of the control system of the electric machine, and reducing the reliability of the control system of the electric machine. The improved high-frequency injection algorithm is combined with the full-order observer based on back electromotive force, and an exponential weighting function is used to smoothly switch the rotor position and speed estimated by the high-frequency injection algorithm and the full-order observer, reducing the complexity and cost of the control system of the electric machine, and improving the reliability of the control system of the electric machine.

[0005] The application provides a motor control method and a motor control system, and the motor control system has a current loop and a speed loop; a full-order observation module, a phase-locked module and a switching function module are arranged in the speed loop; the motor control method comprises the following steps: dividing a full stage from power-on to running of the motor into a zero-speed starting stage, a low-speed running stage and a medium-high speed running stage; injecting a preset frequency pulse voltage into a d-axis part of a PI controller output in the current loop in the zero-speed starting stage of the motor and the low-speed running stage of the motor; obtaining three-phase currents of the motor in the full stage from power-on to running of the motor; obtaining rotor position estimation values of the motor and rotational speed estimation values of the motor by using the full-order observation module, the phase-locked module and the switching function module according to the three-phase currents of the motor in the full stage from power-on to running of the motor in the current loop and the speed loop; and controlling starting and running of the motor according to the rotor position estimation values of the motor and the rotational speed estimation values of the motor in the full stage from power-on to running of the motor.

[0006] In some embodiments, the preset frequency pulse voltage comprises a high frequency pulse voltage; and injecting the preset frequency pulse voltage into the d-axis part of the PI controller output in the current loop comprises injecting the high frequency pulse voltage into the d-axis part of the PI controller output in the current loop according to the following formula:

[0007]

[0008] wherein, is a d-axis component of a stator high frequency pulse voltage in an estimated rotating coordinate system is a d-axis component of a stator high frequency pulse voltage in an estimated rotating coordinate system h is an amplitude of the high frequency pulse voltage, ω h is an electrical angular frequency of the high frequency pulse voltage, and t is time.

[0009] In some embodiments, in the current loop and the speed loop, the rotor position estimation value and the rotor speed estimation value of the motor are obtained according to the three-phase current of the motor by using the full-order observation module, the phase-locked module and the switching function module, including: according to the three-phase current of the motor, the alpha-axis current of the motor, the beta-axis current of the motor, the d-axis current of the motor and the q-axis current of the motor are obtained through coordinate transformation; in the current loop, the d-axis voltage of the motor output by the PI controller is obtained; in the speed loop, the q-axis voltage of the motor output by the PI controller is obtained; based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor and the q-axis current of the motor, the first estimation signal of the rotor position and the rotor speed of the motor is obtained by using the full-order observation module; based on the alpha-axis current of the motor, the beta-axis current of the motor and the preset modulation signal, the second estimation signal of the rotor position and the rotor speed of the motor is obtained by using the phase-locked module; according to the first estimation signal of the rotor position and the rotor speed of the motor and the second estimation signal of the rotor position and the rotor speed of the motor, the switching function module is used for smooth switching according to the current stage of the motor in the zero-speed starting stage, the low-speed running stage and the medium-high speed running stage, so as to obtain the rotor position estimation value and the rotor speed estimation value of the motor in the current stage of the motor.

[0010] In some embodiments, the full-order observation module includes: a full-order observer; based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor and the q-axis current of the motor, the first estimation signal of the rotor position and the rotor speed of the motor is obtained by using the full-order observer, including: based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor and the q-axis current of the motor, the first estimation signal of the rotor position and the rotor speed of the motor is obtained by using the full-order observer according to the following formula:

[0011]

[0012] wherein Δθ is the rotor angle estimation error, is the estimated value of the d-axis electromotive force observed by the full-order observer, is the estimated value of the q-axis electromotive force observed by the full-order observer; is the estimation signal of the rotor position of the motor, k p is the proportional coefficient of the PI controller, k i is the integral coefficient of the PI controller, an estimated signal of the rotor position of the motor; and an estimated signal of the speed of the motor; and the estimated signal of the rotor position of the motor and the estimated signal of the speed of the motor are the first estimated signals of the rotor position and the speed of the motor.

[0013] In some embodiments, the phase-locked module comprises a multiplier, an LPF filter and a phase-locked loop arranged in sequence; and the second estimated signals of the rotor position and the speed of the motor are obtained by using the phase-locked module based on the current of the motor, the current of the motor and a preset modulation signal, comprising: obtaining a product value by passing the current of the motor, the current of the motor and the preset modulation signal through the multiplier; obtaining a filtered value by passing the product value through the LPF filter; and obtaining the estimated signal of the rotor position of the motor and the estimated signal of the speed of the motor by passing the filtered value, the electromagnetic torque of the motor and the load torque of the motor through the phase-locked loop, as the second estimated signals of the rotor position and the speed of the motor.

[0014] In some embodiments, the switching function module is configured to perform smooth switching according to the first estimated signals of the rotor position and the speed of the motor and the second estimated signals of the rotor position and the speed of the motor based on the current stage of the motor being in the zero-speed starting stage, the low-speed running stage or the medium-high speed running stage, and the rotor position estimated value and the speed estimated value of the motor in the current stage are obtained by using the switching function module, comprising: determining whether the current stage of the motor is in the zero-speed starting stage, the low-speed running stage or the medium-high speed running stage; and performing smooth switching by using the switching function module according to the first estimated signals of the rotor position and the speed of the motor and the second estimated signals of the rotor position and the speed of the motor, so as to: if it is determined that the current stage of the motor is in the zero-speed starting stage or the low-speed running stage, taking the second estimated signals of the rotor position and the speed of the motor as the rotor position estimated value and the speed estimated value of the motor; and if it is determined that the current stage of the motor is in the medium-high speed running stage, taking the first estimated signals of the rotor position and the speed of the motor as the rotor position estimated value and the speed estimated value of the motor.

[0015] In some embodiments, the switching function module comprises an exponential weighting function module; and the smooth switching is performed by using the exponential weighting function module according to the first estimated signals of the rotor position and the speed of the motor and the second estimated signals of the rotor position and the speed of the motor, comprising:

[0016]

[0017] wherein, is an estimation signal of the rotor position of the motor, and δ is an exponential weighting function, is an estimation signal of the rotor position and the speed under the condition of high-frequency pulsating voltage injection, is an estimation signal of the rotor position and the speed obtained by the full-order observer; is an estimation signal of the speed of the motor; e is a natural number, is also an estimation signal of the rotor speed of the motor, and n1 and n2 are respectively the starting rotor speed and the ending rotor speed of the switching process using the exponential weighting function module for smooth switching.

[0018] To match the above method, another aspect of the present application provides a motor control device, a motor control system having a current loop and a speed loop; a full-order observation module, a phase-locked module and a switching function module are arranged in the speed loop; the motor control device comprises: a control unit configured to divide the full stage from starting to running after the motor is powered on into a zero-speed starting stage, a low-speed running stage and a medium-high-speed running stage; in the zero-speed starting stage of the motor and the low-speed running stage of the motor, a preset frequency pulsating voltage is injected into the d-axis part of the PI controller output in the current loop; a unit configured to obtain the three-phase current of the motor from the full stage from starting to running after the motor is powered on; the control unit is also configured to obtain the rotor position estimation value of the motor and the speed estimation value of the motor by using the full-order observation module, the phase-locked module and the switching function module according to the three-phase current of the motor in the current loop and the speed loop from the full stage from starting to running after the motor is powered on; the control unit is also configured to control the starting and running of the motor according to the rotor position estimation value of the motor and the speed estimation value of the motor from the full stage from starting to running after the motor is powered on.

[0019] In some embodiments, the preset frequency pulsating voltage includes a high-frequency pulsating voltage; the control unit injects the high-frequency pulsating voltage into the d-axis part of the PI controller output in the current loop according to the following formula:

[0020]

[0021] wherein, is the d-axis component of the stator high-frequency pulsating voltage in the estimated rotating coordinate system U h is the amplitude of the high-frequency pulsating voltage, and ω h is the electrical angular frequency of the high-frequency pulsating voltage, and t is time.

[0022] In some embodiments, the control unit, in the current loop and the speed loop, obtains the rotor position estimation value of the motor and the speed estimation value of the motor according to the three-phase current of the motor, by using the full-order observer module, the phase-locked loop module and the switching function module, comprises: according to the three-phase current of the motor, the α-axis current of the motor, the β-axis current of the motor, the d-axis current of the motor and the q-axis current of the motor are obtained through coordinate transformation; in the current loop, the d-axis voltage of the motor output by the PI controller is obtained; in the speed loop, the q-axis voltage of the motor output by the PI controller is obtained; based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor and the q-axis current of the motor, the first estimation signal of the rotor position and the speed of the motor is obtained by using the full-order observer module; based on the α-axis current of the motor, the β-axis current of the motor and the preset modulation signal, the second estimation signal of the rotor position and the speed of the motor is obtained by using the phase-locked loop module; according to the first estimation signal of the rotor position and the speed of the motor and the second estimation signal of the rotor position and the speed of the motor, the switching function module is used for smooth switching according to the current stage of the motor in the zero-speed starting stage, the low-speed running stage and the medium-high speed running stage, to obtain the rotor position estimation value of the motor and the speed estimation value of the motor in the current stage of the motor.

[0023] In some embodiments, the full-order observer module comprises: a full-order observer; the control unit, based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor and the q-axis current of the motor, uses the full-order observer to obtain the first estimation signal of the rotor position and the speed of the motor, comprises: based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor and the q-axis current of the motor, the full-order observer is used to obtain the first estimation signal of the rotor position and the speed of the motor according to the following formula:

[0024]

[0025] Wherein, Δθ is the rotor angle estimation error, is the estimated value of the d-axis electromotive force observed by the full-order observer, is the estimated value of the q-axis electromotive force observed by the full-order observer; is the estimation signal of the rotor position of the motor, k p is the proportional coefficient of the PI controller, k i is the integral coefficient of the PI controller, an estimated signal of a rotor position of the motor; and an estimated signal of a speed of the motor; and the estimated signal of the rotor position of the motor and the estimated signal of the speed of the motor are the first estimated signals of the rotor position and the speed of the motor.

[0026] In some embodiments, the phase-locked module comprises a multiplier, an LPF filter and a phase-locked loop arranged in sequence; and the control unit obtains the second estimated signals of the rotor position and the speed of the motor by using the phase-locked module based on the current of the motor, the current of the motor and the preset modulation signal, including: obtaining a product value by passing the current of the motor, the current of the motor and the preset modulation signal through the multiplier; obtaining a filtered value by passing the product value through the LPF filter; and obtaining the estimated signal of the rotor position of the motor and the estimated signal of the speed of the motor by passing the filtered value, the electromagnetic torque of the motor and the load torque of the motor through the phase-locked loop, as the second estimated signals of the rotor position and the speed of the motor.

[0027] In some embodiments, the control unit performs smooth switching by using the switching function module according to the first estimated signals of the rotor position and the speed of the motor and the second estimated signals of the rotor position and the speed of the motor based on the current stage of the motor being in the zero-speed starting stage, the low-speed running stage or the medium-high speed running stage, to obtain the rotor position estimated value of the motor and the speed estimated value of the motor in the current stage of the motor, including: determining whether the current stage of the motor is in the zero-speed starting stage, the low-speed running stage or the medium-high speed running stage; and performing smooth switching by using the switching function module according to the first estimated signals of the rotor position and the speed of the motor and the second estimated signals of the rotor position and the speed of the motor, to: if it is determined that the current stage of the motor is in the zero-speed starting stage or the low-speed running stage, take the second estimated signals of the rotor position and the speed of the motor as the rotor position estimated value of the motor and the speed estimated value of the motor; and if it is determined that the current stage of the motor is in the medium-high speed running stage, take the first estimated signals of the rotor position and the speed of the motor as the rotor position estimated value of the motor and the speed estimated value of the motor.

[0028] In some embodiments, the switching function module comprises an exponential weighting function module; and the control unit performs smooth switching by using the exponential weighting function module according to the first estimated signals of the rotor position and the speed of the motor and the second estimated signals of the rotor position and the speed of the motor, including:

[0029]

[0030]

[0031] wherein, is an estimated signal of the rotor position of the motor, and δ is an exponential weighting function, is an estimated signal of the rotor position and speed under high-frequency pulsating voltage injection, is an estimated signal of the rotor position and speed obtained by a full-order observer; is an estimated signal of the speed of the motor; e is a natural number, is also an estimated signal of the rotor speed of the motor, and n1 and n2 are respectively an estimated signal of the starting rotor speed and an estimated signal of the ending rotor speed of the switching process using the exponential weighting function module for smooth switching.

[0032] In order to achieve the above object, the motor control device according to the present application comprises: a current loop PI controller, a high-frequency pulsating voltage injection module, a full-order observer, an exponential weighting function module, and a motor control unit.

[0033] In order to achieve the above object, the motor control device according to the present application comprises: a current loop PI controller, a high-frequency pulsating voltage injection module, a full-order observer, an exponential weighting function module, and a motor control unit.

[0034] In order to achieve the above object, the motor control device according to the present application comprises: a current loop PI controller, a high-frequency pulsating voltage injection module, a full-order observer, an exponential weighting function module, and a motor control unit.

[0035] Therefore, the scheme of the present application, aiming at the zero-speed starting stage, low-speed running stage and medium-high speed running stage of the motor, in the zero-speed starting stage and low-speed running stage, a high-frequency pulsating signal is injected into the d-axis voltage output by the current loop PI controller, so as to convert the analysis of the stator current component in the dq coordinate system into the analysis of the stator current component in the αβ coordinate system, and obtain the position information and speed information of the rotor; in the high-speed running stage, a full-order observer is used to obtain the position information and speed information of the rotor; and a smooth switching strategy of the exponential weighting function is set to switch the estimated values of the motor rotor position and speed between the high-frequency injection algorithm and the full-order observer; according to the position information and speed information of the rotor, the motor is controlled; thus, by combining the improved high-frequency injection algorithm with the full-order observer based on the back electromotive force, and using the exponential weighting function to smoothly switch the estimated motor rotor position and speed of the high-frequency injection algorithm and the full-order observer, the complexity and cost of the motor control system are reduced, and the reliability of the motor control system is improved.

[0036] Other features and advantages of the present application will be set forth in the following description, and in part will become apparent to those skilled in the art upon examination of the following or can be learned by practice of the application. The objects and advantages of the application can be realized and attained by means of the instrumentalities and combinations particularly pointed out in the appended claims. BRIEF DESCRIPTION OF DRAWINGS

[0037] Fig. 1 Flow chart of an embodiment of the method for controlling the motor of the present application;

[0038] Fig. 2 Flow chart of an embodiment of the method of the present application for obtaining the rotor position estimate of the motor and the rotor speed estimate of the motor using the full-order observer module, the phase-locked loop module and the switching function module;

[0039] Fig. 3 Flow chart of an embodiment of the method of the present application for obtaining the second estimate signal of the rotor position and the rotor speed of the motor using the phase-locked loop module;

[0040] Fig. 4 Flow chart of an embodiment of the method of the present application for performing smooth switching using the switching function module according to the first estimate signal of the rotor position and the rotor speed of the motor and the second estimate signal of the rotor position and the rotor speed of the motor;

[0041] Fig. 5 Structure diagram of an embodiment of the control device for the motor of the present application;

[0042] Fig. 6 Structure diagram of the control system for the motor;

[0043] Fig. 7 Structure diagram of the phase current sampling circuit;

[0044] Fig. 8 Diagram showing the relationship between the estimated coordinate system and the actual coordinate system;

[0045] Fig. 9 Control flow diagram of the control system for the permanent magnet synchronous motor, wherein (a) is a diagram showing the control flow of the Fig. 1 , and (b) is a diagram showing the control flow of the Fig. 2 ;

[0046] Fig. 10 Control flow diagram of the rotor electrical angle position phase-locked loop tracking observer, specifically the control flow diagram of the PLL phase-locked loop part in Fig. 9 ;

[0047] Fig. 11 Implementation flow diagram of the rotor electrical angle position estimation by high-frequency pulsating voltage injection method, specifically the part shown by the blue line in Fig. 9 ;Fig. 11 Observer_PLL is shown in part of FIG. 1; Fig. 10

[0048] Fig. 12 a curve diagram of an exponential weighting function δ;

[0049] Fig. 13 a curve diagram of an estimated rotor angle;

[0050] Fig. 14 a speed waveform diagram for switching the control algorithm in the 200rpm-400rpm stage.

[0051] In combination with the drawings, the following are the meanings of the reference signs in the embodiments of the present application:

[0052] 102 - acquisition unit; 104 - control unit. DETAILED DESCRIPTION

[0053] In order to make the purposes, technical solutions and advantages of the present application clearer, the technical solutions of the present application will be described clearly and completely below in combination with specific embodiments of the present application and corresponding drawings. Obviously, the described embodiments are only some of the embodiments of the present application, but not all the embodiments. Based on the embodiments in the present application, all other embodiments obtained by those skilled in the art without creative labor fall within the scope of protection of the present application.

[0054] It is considered that the motor (such as a permanent magnet synchronous motor) relies on a sensor for direct measurement to obtain rotor position information and speed information, which increases the complexity and cost of the control system of the motor, and causes the reliability of the control system of the motor to be reduced. In view of this, the control method without position sensor has gradually become a hotspot of research and application. In the control system of the permanent magnet synchronous motor without position sensor, the means for obtaining the position information of the rotor is based on the back electromotive force to solve the motor electrical angle. However, this method has great difficulty in signal acquisition and calculation when the motor is started at zero speed and runs at low speed, and it is difficult to accurately obtain the position information of the rotor, thereby affecting the performance of the control system of the permanent magnet synchronous motor. In order to overcome this problem, the method used in the related scheme is to use a high-frequency injection algorithm to drive the motor when the motor is started at zero speed and runs at low speed. However, this method has high dependence on the salient structure of the motor, and the signal processing process is relatively complex, especially the excessive use of filters, which will have a great impact on the amplitude and phase of the control system of the permanent magnet synchronous motor, and further affect the stability of the control system of the permanent magnet synchronous motor. In addition, when the motor switches from the medium-low speed stage to the medium-high speed running stage, the linear weighting switching mode of angle and speed is usually used. This switching strategy may cause the motor to vibrate in actual application, and even cause switching failure in severe cases, affecting the normal operation of the motor.​

[0055] Therefore, the scheme of the present application proposes a control method of a motor, in particular, a control method of a full-speed threshold position sensorless permanent magnet synchronous motor, which combines an improved high-frequency injection algorithm with a full-order observer based on back electromotive force, realizes the use of high-frequency injection in the zero-speed starting and low-speed running stages of the motor, uses the full-order observer in the medium and high-speed running stages, and uses an exponential weighting function to smoothly switch the motor rotor position and speed estimated by the two algorithms (i.e., the high-frequency injection and full-order observer two sensorless algorithms), thereby reducing the complexity and cost of the control system of the motor and improving the reliability of the control system of the motor.

[0056] According to an embodiment of the present application, a control method of a motor is provided, as shown in the flowchart of an embodiment of the method of the present application. Fig. 1 The control system of the motor has a current loop and a speed loop; in the speed loop, a full-order observation module, a phase-locked module and a switching function module are arranged, the full-order observation module is a full-order observer as shown in Fig. 9 , the phase-locked module is a multiplier, an LPF module and a PLL phase-locked loop module as shown in Fig. 9 , and the switching function module is an exponential weighting function δ as shown in Fig. 9 and Fig. 12 In the scheme of the present application, as shown in Fig. 1 , the control method of the motor comprises steps S110 to S140.

[0057] At step S110, the full stage from starting to running of the motor after power-on is divided into a zero-speed starting stage, a low-speed running stage and a medium and high-speed running stage; in the zero-speed starting stage of the motor and the low-speed running stage of the motor, a preset frequency pulse voltage is injected to the d-axis part of the output of the PI controller in the current loop; wherein the preset frequency pulse voltage is a high-frequency pulse voltage injected by using an improved high-frequency pulse injection algorithm.

[0058] At step S120, the three-phase current of the motor is obtained in the full stage from starting to running after power-on of the motor; the full stage from starting to running after power-on of the motor includes the zero-speed starting stage of the motor, the low-speed running stage of the motor and the medium and high-speed running stage of the motor; that is, the three-phase current of the motor, such as the A-phase current i a , the B-phase current i b and the C-phase current i c , is obtained in the zero-speed starting stage, the low-speed running stage and the medium and high-speed running stage.

[0059] At step S130, during the full stage from starting to running after the motor is powered on, according to the three-phase current of the motor, the rotor position estimation value of the motor and the rotating speed estimation value of the motor are obtained by using the full-order observation module, the phase-locked module and the switching function module in the current loop and the speed loop; wherein the estimated rotor position estimation value of the motor is like a rotor angle estimation value the rotating speed estimation value of the motor is like an estimated rotating speed

[0060] At step S140, during the full stage from starting to running after the motor is powered on, according to the rotor position estimation value of the motor and the rotating speed estimation value of the motor, the starting and running of the motor are controlled.

[0061] The scheme of the application combines the improved high-frequency injection algorithm with the full-order observer based on back electromotive force, designs a control system of the permanent magnet synchronous motor with full-speed threshold, realizes the use of high-frequency injection in the zero-speed starting and low-speed running stage of the motor, uses the full-order observer in the medium and high-speed running stage, and uses the exponential weighting function to smoothly switch the motor rotor position and speed estimated by the two algorithms (i.e. the high-frequency injection and the full-order observer two non-inductive algorithms), reduces the complexity and cost of the control system of the motor, improves the reliability of the control system of the motor, and solves the problem of switching the position sensorless control algorithm of the motor in different running stages in the related scheme. The full-speed threshold is the whole process from zero-speed starting to high-speed running, including the process of zero-speed starting, low-speed running and medium and high-speed running.

[0062] In some embodiments, the preset frequency pulse voltage includes a high-frequency pulse voltage.

[0063] The step of injecting the preset frequency pulse voltage into the d-axis part of the PI controller in the current loop in step S110 includes: injecting a high-frequency pulse voltage into the d-axis part of the PI controller in the current loop according to the following formula:

[0064]

[0065] wherein, is the d-axis component of the stator high-frequency pulse voltage in the rotating coordinate system is the d-axis component of the stator high-frequency pulse voltage in the rotating coordinate system h is the amplitude of the high-frequency pulse voltage, ω h is the electrical angular frequency of the high-frequency pulse voltage, and t is time.

[0066] The improved high frequency injection in the scheme of the present application will be described in detail below.

[0067] Fig. 6 is a structural schematic diagram of the control system of the motor. As Fig. 6As shown, the AC power provided by the power grid is fed through a three-phase rectifier bridge, bus capacitors, and a three-phase inverter bridge to power the motor. The motor control device includes: a bus voltage sampling circuit, a main controller, a current sampling circuit, a DC / DC (i.e., DC to DC conversion) voltage converter, and a protection circuit. The current sampling circuit samples the three-phase current of the motor from the three-phase windings and feeds it back to the main controller. The target voltage sampling circuit samples the bus voltage from the output end of the bus capacitor and feeds it back to the main controller and the DC / DC voltage converter respectively. The DC / DC voltage converter respectively powers the protection circuit, the main controller, and the current sampling circuit.

[0068] Fig. 6 The schematic diagram of the entire control system is shown in Figure 1. During the zero-speed start and low-speed operation of the motor, the high-frequency pulse injection method is used to estimate the rotor position information. All the following algorithms, including the high-frequency pulse injection algorithm, are run on Fig. 6 in the main controller.

[0069] Fig. 7 It is the structural diagram of the phase current sampling circuit, that is, Fig. 6 The structural diagram of the current sampling circuit is shown in FIG. Fig. 7 As shown, the U phase current I U After passing through resistor R1, it is input to the non-inverting input terminal of the comparator. Resistor R2 and capacitor C2 are connected in parallel between the non-inverting input terminal of the comparator and the analog ground AGND. Capacitor C1 and resistor R3 are connected in parallel between the inverting input terminal of the comparator and the output terminal of the comparator. The output terminal of the comparator is connected to the analog ground AGND after passing through resistor R4 and capacitor C3. The common terminal of resistor R4 and capacitor C3 outputs the sampling current I of the U phase U _DSP to host controller.

[0070] Fig. 7 The diagram shows the sampling circuit diagram of the U-phase current. The sampling of the V-phase current is consistent with that of the U-phase current. Fig. 8 The diagram below shows the relationship between the estimated coordinate system and the actual coordinate system. After Clarke transformation and Park transformation of the U-phase current and the V-phase current, the dq-axis current can be obtained, as shown in Figure 2. Fig. 8 Shown is the estimated rotation coordinate system The relationship between the actual rotating coordinate system dq and the stationary coordinate system αβ, the estimated rotating coordinate system The rotor angle error between the actual rotating coordinate system dq is:

[0071]

[0072] Where, is the estimated value of the rotor angle, θ is the actual value of the rotor angle, and Δθ is the error in the rotor angle estimation.

[0073] Assume that the injection is into the estimated rotation coordinate system The high frequency pulse voltage is:

[0074]

[0075] Where, Estimate the rotation coordinate system The stator high-frequency pulse voltage dq axis component, where To estimate the rotation coordinate system The d-axis component of the stator high-frequency pulse voltage, To estimate the rotation coordinate system The q-axis component of the stator high-frequency pulse voltage; U h is the high-frequency pulse voltage amplitude; ω h is the electrical angular frequency of the high-frequency pulse voltage, and t is the time.

[0076] When analyzing the electrical characteristics of a permanent magnet synchronous motor under high-frequency excitation, if the angular frequency in the motor fundamental wave equation is significantly lower than the angular frequency of the injected high-frequency pulse voltage signal, the inductive reactance in the motor stator winding impedance will dominate the total impedance, and the influence of resistance will be relatively small and can be approximately ignored. Based on this premise, the permanent magnet synchronous motor can be simplified and equivalent to a pure inductance model. This simplification makes it easier to analyze the electromagnetic relationship of the motor under high-frequency excitation, and thus obtain the stator voltage equation of the permanent magnet synchronous motor under high-frequency excitation:

[0077]

[0078] Where u dh 、u qh They are the dq axis components of the stator high frequency pulse voltage, i dh 、i dh are the dq axis components of the stator high-frequency pulsating current respectively, Ld and Lq are the direct-axis and quadrature-axis inductances of the motor, and p is the differential operator.

[0079] Depend on Fig. 8 It can be seen that the estimated rotation coordinate system The relationship between the dq high-frequency pulse voltage and the actual rotating coordinate system is:

[0080]

[0081] according to Fig. 8 The estimated rotated coordinate system shown The relationship between the actual rotating coordinate system dq, the coordinate transformation principle, and the mathematical model (3) under high-frequency excitation of the permanent magnet synchronous motor can be obtained as follows:

[0082]

[0083] where i αh and i βh are stator high-frequency pulsating current αβ-axis components. Transforming the above equation, we have:

[0084]

[0085] Substituting equation (2) and equation (4) into equation (6), we have:

[0086]

[0087] The actual coordinate system stator high-frequency pulsating current αβ-axis components, i.e. actual coordinate system αβ-axis current response under high-frequency excitation, are as follows:

[0088]

[0089] When the rotor electric angle position estimation converges to the rotor electric angle position actual value, and the rotor electric angle position estimation error is small enough, we have:

[0090]

[0091] In the permanent magnet synchronous motor field-oriented vector control system, the sampled stator current αβ-axis components i αβ include: fundamental current i αβf , high-frequency pulsating response current i αβh , and high-order harmonic current i αβx generated by PWM switching signal, i.e.:

[0092]

[0093] where I f and θ f are fundamental current amplitude and phase angle error, respectively. Multiply the αβ-axis current with the modulation signal 2sinω h t, and use a low-pass filter to process the product signal. If the motor works in zero speed and low speed range, the rotor electric angle speed ω is small ω h , i.e. the rotor angle actual value θ = ωt is small ω h t; and the high-order harmonic current i αβx multiplied by the modulation signal is still a high-frequency current. The above amplitude modulation and low-pass filter processing can be represented by the following equation:

[0094]

[0095] After sampling the three-phase current and then performing coordinate transformation, we have i α , i β , and after LPF (low-pass filter), we have iα1 、i β1 .i αβ1 for i α1 、i β1 The unified representation of LPF is the parameter of the low-pass filter. According to the above formula, the stator current αβ axis component i αβ After modulation and low-pass filtering, the current signal i αβ The rotor electrical angle position information is included in the formula (14). The rotor angle θ is included at the end. The specific waveform is the sine-cosine waveform under a certain amplitude, so i αβ The data contains the rotor electrical angle position information. To extract the angle, we need to use the phase locked loop (PLL) described below.

[0096] After obtaining the i containing the rotor position information αβ1 After that, a phase-locked loop can be used to estimate the rotor position and speed. This method is not only stable, but also can effectively suppress and filter the high-frequency components in the current signal, ensuring the accuracy of position and speed estimation.

[0097] In the solution of the present invention, a high-frequency pulse signal injection method is adopted, and by analyzing the current components of the αβ axes, the dependence on the salient polarity structure of the motor body is effectively reduced, making the algorithm more universal. As shown at the end of formula (14), in the formula containing the motor angle, only the d-axis inductance Ld is included, rather than the difference between the q-axis inductance Lq and the d-axis inductance Ld (the larger the difference Lq-Ld, the greater the salient polarity of the motor), so the dependence on the salient polarity of the motor body is reduced. In response to the problem of the high-frequency injection algorithm's dependence on the salient polarity structure of the permanent magnet synchronous motor in the related solution, the solution of the present invention is improved so that even motors with an unclear salient polarity structure of the permanent magnet synchronous motor can be applied to the high-frequency injection algorithm, thereby broadening the scope of application of the algorithm.

[0098] In the solution of the present invention, the medium- and high-speed operation stage observer uses a full-order observer to estimate the motor angle and speed information during the medium- and high-speed operation stage of the motor. The rotor position and speed information extraction method extracts components from the αβ axes, eliminating the need for bandpass filters, reducing the use of filters, and reducing reliance on the salient polarity structure of the motor body. The sensorless algorithm switching method uses an exponential weighting function. In this way, an improved high-frequency pulse injection algorithm is used during the motor startup and low-speed operation stages to analyze the shaft current components, reducing the use of filters, achieving zero-speed startup of the motor, and reducing reliance on the salient polarity structure of the motor body. During the medium- and high-speed operation stage, an exponential weighting function is used to switch between the two sensorless algorithms, high-frequency injection and full-order observer, to determine the motor rotor position and speed, gradually switching to the full-order observer to smooth the switching process.

[0099] In some embodiments, in step S130, in the current loop and the speed loop, according to the three-phase current of the motor, the full-order observation module, the phase-locked module and the switching function module are used to obtain the rotor position estimation value of the motor and the speed estimation value of the motor. See the following exemplary description for the specific process.

[0100] The following combination Fig. 2 The flowchart of an embodiment of the method of the present invention using the full-order observation module, the phase-locked module and the switching function module to obtain the estimated value of the rotor position of the motor and the estimated value of the speed of the motor is shown, further illustrating the specific process of using the full-order observation module, the phase-locked module and the switching function module in step S130 to obtain the estimated value of the rotor position of the motor and the estimated value of the speed of the motor, including: steps S210 to S250.

[0101] Step S210, according to the three-phase current of the motor, through coordinate transformation, obtain the α-axis current of the motor, the β-axis current of the motor, the d-axis current of the motor and the q-axis current of the motor; wherein the α-axis current is the current i α , β axis current such as current i β , d-axis current such as current i d , q-axis current such as current i q , please refer to Fig. 9 Example shown.

[0102] Step S220, in the current loop, obtain the d-axis voltage of the motor output by the PI controller; in the speed loop, obtain the q-axis voltage of the motor output by the PI controller; wherein the d-axis voltage is such as the voltage u d , q-axis voltage is like voltage u q , please refer to Fig. 9 Example shown.

[0103] Step S230 , obtaining a first estimation signal of the rotor position and speed of the motor using the full-order observation module based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor, and the q-axis current of the motor.

[0104] Step S240 : Based on the α-axis current of the motor, the β-axis current of the motor, and a preset modulation signal, the phase-locked module is used to obtain a second estimation signal of the rotor position and speed of the motor.

[0105] Step S250, based on the current stage of the motor in the zero speed starting stage, low speed running stage and medium-high speed running stage of the motor, according to the first estimation signal of the rotor position and speed of the motor and the second estimation signal of the rotor position and speed of the motor, the switching function module is used for smooth switching to obtain the rotor position estimation value of the motor and the rotor speed estimation value of the motor in the current stage of the motor.

[0106] In the scheme of the application, by injecting a high-frequency pulsating signal into the d-axis of the two-phase rotating coordinate system output by the current loop PI of the control system of the motor, the analysis of the stator current components in the dq coordinate system in the related scheme is converted into the analysis of the stator current components in the αβ coordinate system, so as to obtain the position information and speed information of the rotor. This method not only saves the band-pass filter and simplifies the system structure, but also significantly improves the dynamic control performance of the system. Among them, the analysis of the stator current components of the αβ axis simplifies the system structure because the band-pass filter is no longer needed. Because of the lack of the band-pass filter, the system bandwidth is increased and the hysteresis is reduced, so the dynamic control performance of the system is improved. In view of the problem that the high-frequency injection algorithm in the related scheme uses a large number of filters in the signal processing process. The scheme of the application saves the band-pass filter, simplifies the control system structure of the permanent magnet synchronous motor, and improves the efficiency and reliability of the control system of the permanent magnet synchronous motor.

[0107] In some embodiments, the full-order observer module comprises a full-order observer.

[0108] In step S230, based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor and the q-axis current of the motor, the full-order observer module is used to obtain the first estimation signal of the rotor position and speed of the motor, comprising:

[0109] Based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor and the q-axis current of the motor, the full-order observer is used to obtain the first estimation signal of the rotor position and speed of the motor according to the following formula:

[0110]

[0111] Where Δθ is the rotor angle estimation error, is the estimated value of the d-axis electromotive force observed by the full-order observer, is the estimated value of the q-axis electromotive force observed by the full-order observer. is the estimation signal of the rotor position of the motor, k p is the proportional coefficient of the PI controller, k i is the integral coefficient of the PI controller, an estimated signal of the rotor position of the motor and an estimated signal of the speed of the motor as the first estimated signals of the rotor position and the speed of the motor.

[0112] The full-order observer will be described in detail below.

[0113] In the high-speed running stage of the motor, the full-order observer is used to estimate the position information of the rotor. The dynamic equation of the state observer is: where H is the feedback matrix. According to the controllability and observability of the control system, the four poles of the observer are configured at -ω k , and the feedback matrix H is:

[0114]

[0115] where R s is the stator resistance of the motor, ω k is the system bandwidth, and the dynamic equation of the observer is:

[0116]

[0117] First, the observation variable ω is calculated by formula (17).

[0118]

[0119] where Rs is the winding resistance of the motor, Ld and Lq are the direct-axis and quadrature-axis inductances of the motor, ω k is the system bandwidth, and according to the different adjustment values of the system, is the estimated speed, E rr I d , E rr I q is the estimated error of the current I d , I q , E is the derivative of the estimated I d , I q current, V d , V q is the output of the current loop PI.

[0120] The PI regulator (i.e., PI controller) is designed according to the position error, and the speed ω

[0121]

[0122] where k p is the proportional coefficient of the PI controller, and k iis the integral coefficient of the PI controller. The motor rotor position angle is obtained by integrating the speed:

[0123]

[0124] The estimated rotor angle is Fig. 13 shown. Fig. 13 Schematic diagram of the estimated rotor angle curve. Fig. 13 The horizontal axis is time t (unit s), and the vertical axis is angle (unit rad). Fig. 13 It is shown that the high-frequency injection + full-order observer algorithm can accurately estimate the motor rotor angle.

[0125] In the solution of the present invention, the use of the full-order observer in the medium and high-speed operation stage fully utilizes its high adaptability and high-precision estimation ability to the changes in motor parameters during high-speed operation. Under the premise of ensuring the stability of the motor control system, the dynamic control performance of the motor is further improved, so that the motor can maintain an efficient and accurate operating state throughout the entire speed range, meeting the requirements of modern industrial control for high performance, high reliability and a wide speed regulation range of permanent magnet synchronous motor systems.

[0126] In some implementations, the phase-locked module includes: a multiplier, an LPF filter, and a phase-locked loop arranged in sequence.

[0127] For the specific process of obtaining the second estimated signal of the rotor position and speed of the motor based on the α-axis current of the motor, the β-axis current of the motor, and the preset modulation signal using the phase-locked module in step S240, please refer to the following exemplary description.

[0128] The following combination Fig. 3 The flowchart of an embodiment of the method of the present invention for obtaining the second estimated signal of the rotor position and speed of the motor by using the phase-locked module is shown, which further illustrates the specific process of obtaining the second estimated signal of the rotor position and speed of the motor by using the phase-locked module in step S240, including: steps S310 to S330.

[0129] In step S310 , the α-axis current of the motor, the β-axis current of the motor, and a preset modulation signal are multiplied by the multiplier to obtain a product value.

[0130] Step S320: Pass the product value through the LPF filter to obtain a filtered value.

[0131] In step S330 , the filtered value, the electromagnetic torque of the motor, and the load torque of the motor are passed through the phase-locked loop to obtain estimated signals of the rotor position and speed of the motor as second estimated signals of the rotor position and speed of the motor.

[0132] Fig. 9 The control flow diagram of the permanent magnet synchronous motor control system is shown in Figure 2, where (a) is a schematic diagram of the control flow of the permanent magnet synchronous motor control system. Fig. 1 , (b) is a schematic diagram Fig. 2 .like Fig. 9 As shown, the d-axis given current i dref = 0, and the actual d-axis current i output by Park transformation d , is input to the PI controller after passing through the comparator; the PI controller outputs the d-axis voltage u d , and the high-frequency pulse voltage u injected by the high-frequency pulse h cosω h After passing through the comparator, t is input to the Park inverse transformation module. Given speed ω ref , and the estimated speed output by the exponential weighting function After passing through the comparator, it is input to the PI controller, and the PI controller outputs the q-axis given current i qref , and the actual q-axis current i output by the Park converter q , which is output to the PI controller after passing through the comparator. The PI controller outputs the q-axis voltage u q To the Park inverse transform module. The Park inverse transform module outputs the rotor angle estimate based on the exponential weighting function Output α-axis voltage u α , β-axis voltage u β To the space vector pulse width modulation (SVPWM) module. The SVPWM module outputs the control signal to the three-phase inverter, and the three-phase inverter outputs the three-phase current to the motor. The three-phase current i a 、i b and i c After Clarke transformation, the output α-axis current i α , β-axis current i β To the Park transform module. The Park transform module outputs the rotor angle estimate based on the exponential weighting function Output d-axis actual current i d , q-axis actual current i q d-axis voltage u d , q-axis voltage u q , d-axis actual current i d , q-axis actual current i q , input to the full-order observer. The stator current αβ axis component i output by Clarke transformation αβ With the modulation signal 2sinω h t multiplied, and the product signal is processed using a low-pass filter (LPF) to obtain signal i αβ1, and then input to the PLL phase-locked loop. The output signal of the full-order observer and the output signal of the PLL phase-locked loop are output after the exponential weighting function to estimate the speed. and the rotor angle estimate

[0133] Fig. 10 This is a schematic diagram of the control flow of the rotor electrical angle position phase-locked loop tracking observer. Fig. 9 The control flow diagram of the PLL phase-locked loop part in FIG. Fig. 10 As shown, the signal i β1 and The product value of, and signal i α1 and The difference ε of the product value is obtained by K p +k i / s module and then output to the comparator. The electromagnetic torque T of the motor e , the motor's load torque T l The output signal of the comparator is sent to the 1 / J module, 1 / s module, and n p Module output estimated speed Estimated speed Output rotor angle estimation after passing through 1 / s module J is the moment of inertia; 1 / s is the integral; n p is the number of motor pole pairs. Fig. 10 The figure shows the implementation block diagram of the rotor position phase-locked loop tracking observer. Fig. 10 where ε is obtained by signal processing of the phase-locked loop feedback loop and linearization of the error signal, as shown in the following formula:

[0134]

[0135] Among them, signal i α1 , signal i β1 is signal i αβ1 The α and β axis components of the motor are shown in the above formula as the motor rotor position angle estimation error signal value obtained by using the improved high frequency injection method.

[0136] Fig. 11 This is a schematic diagram of the implementation process of the rotor electrical angle position estimation using the high-frequency pulse voltage injection method. Fig. 9 The part indicated by the blue line in Figure (b) is Fig. 11 Observer_PLL is Fig. 10 The part shown. Fig. 11 The block diagram of the improved high frequency injection method for estimating rotor angular speed is shown in Figure 2. Fig. 11 As shown, the signal i A and signal i BAfter the coordinate transformation of ABC / αβ, the stator current αβ axis component i is obtained αβ , stator current αβ axis component i αβ With the modulation signal 2sinω h t multiplied, and the product signal is processed using a low-pass filter (LPF) to obtain signal i αβ1 , and then input to Observer_PLL (flux observer + phase-locked loop). Observer_PLL (flux observer + phase-locked loop) is based on the electromagnetic torque T of the motor. e , the motor's load torque T l , output estimated speed and the rotor angle estimate Among them, i A 、i B i.e. i a 、i b , see The exponential weighting function will be described in detail below The purple part shown in (b).

[0137] Compared with the high-frequency injection method in related schemes, this improved high-frequency pulse voltage injection method achieves significant optimization in the generation mechanism of the rotor electrical angle position estimation error signal. Specifically, the improved error signal value no longer depends on the salient polarity structure or saturation salient polarity of the motor, but is only associated with the parameter of the stator winding d-axis inductance Ld. This feature makes this method more widely applicable and more adaptable when facing motors with different salient polarity characteristics. Moreover, the improved algorithm successfully eliminates the use of bandpass filters. This change not only simplifies the system structure and reduces the complexity of the system, but also effectively avoids the problems caused by the use of bandpass filters in the methods in related schemes, such as unnecessary attenuation of the system signal amplitude and phase angle delay. These problems often have a negative impact on the dynamic performance of the system in the control system of related schemes. The improved method in the scheme of the present invention significantly improves the dynamic control performance of the system by optimizing the system architecture, and improves the control accuracy and response speed.

[0138] In some embodiments, in step S250, based on the current stage of the motor among the zero-speed starting stage, low-speed operation stage and medium-high-speed operation stage of the motor, according to the first estimated signal of the rotor position and speed of the motor and the second estimated signal of the rotor position and speed of the motor, the switching function module is used for smooth switching to obtain the specific process of the rotor position estimated value of the motor in the current stage of the motor and the speed estimated value of the motor. See the following exemplary description.

[0139] The following combination Fig. 12An embodiment flowchart of the method of the application shown in the figure for smoothly switching according to the first estimated signal of the rotor position and speed of the motor and the second estimated signal of the rotor position and speed of the motor using the switching function module, further illustrates the specific process of smoothly switching according to the first estimated signal of the rotor position and speed of the motor and the second estimated signal of the rotor position and speed of the motor using the switching function module in step S250, comprising steps S410 to S440.

[0140] Step S410, determine whether the current stage of the motor is in the zero-speed starting stage, low-speed running stage, or in the medium-high speed stage of the motor.

[0141] Step S420, according to the first estimated signal of the rotor position and speed of the motor and the second estimated signal of the rotor position and speed of the motor, use the switching function module to perform step S430 or step S440.

[0142] Step S430, if it is determined that the current stage of the motor is in the zero-speed starting stage, low-speed running stage, the second estimated signal of the rotor position and speed of the motor is taken as the rotor position estimation value of the motor and the motor speed estimation value.

[0143] Step S440, if it is determined that the current stage of the motor is in the medium-high speed running stage, the first estimated signal of the rotor position and speed of the motor is taken as the rotor position estimation value of the motor and the motor speed estimation value.

[0144] The related scheme is to replace the band-pass filter with a two-stage series pure delay filter to extract the high-frequency current response, while the scheme of the application is to extract the high-frequency current response by analyzing the stator current components of the αβ axis. The scheme of the application uses an improved high-frequency pulse injection algorithm in the zero-speed starting and low-speed running stage of the motor, and innovatively analyzes the current components of the αβ axis, thereby effectively saving the use of the band-pass filter. This improvement not only realizes the zero-speed starting of the motor, but also significantly reduces the dependence on the salient structure of the motor body, making the algorithm more universal and practical. By reducing the use of the band-pass filter, the structure of the motor control system is simplified, the complexity and potential failure points of the motor control system are reduced, and the overall reliability and stability are improved. At the same time, the simplified motor control system is also more advantageous in cost and maintenance, providing a more economical and efficient solution for industrial applications.

[0145] When the motor enters the medium-high speed running stage, the scheme of the application introduces a smooth switching strategy based on an exponential weighting function, which is used for switching the estimated values of the motor rotor position and speed between the high-frequency injection algorithm and the full-order observer. The related scheme uses a new variable gain sliding mode observer method with S-curve function and resistance feedback when the motor runs at medium-high speed, and the switching algorithm is linear weighting switching. Alternatively, the related scheme uses a sliding mode observer when the motor runs at medium-high speed, and the switching algorithm is linear weighting switching. However, the scheme of the application uses a full-order observer when the motor runs at medium-high speed, and the switching algorithm is exponential weighting switching. Compared with the switching method of the related scheme, the exponential weighting function can dynamically adjust the weights of the two algorithms (i.e. the high-frequency injection and the full-order observer two non-inductive algorithms) according to the running state of the motor, ensuring the smoothness and stability of the switching process. This smooth transition effectively avoids the problems of motor jitter and torque fluctuation that may be caused by sudden switching of the control algorithm, and improves the smoothness and comfort of the motor operation. In addition, the use of the full-order observer in the medium-high speed running stage fully utilizes its high adaptability to parameter changes of the motor and high-precision estimation ability at high speed, which further improves the dynamic control performance of the motor under the premise of ensuring the stability of the control system of the motor, so that the motor can maintain high efficiency and accurate operation state in the entire speed range, meeting the requirements of modern industrial control for high performance, high reliability and wide speed range of the permanent magnet synchronous motor system.

[0146] In some embodiments, the switching function module comprises an exponential weighting function module.

[0147] In step S420, the smooth switching is performed on the first estimated signal of the rotor position and speed of the motor and the second estimated signal of the rotor position and speed of the motor by using the exponential weighting function module, comprising:

[0148]

[0149] wherein, is the estimated signal of the rotor position of the motor, and δ is an exponential weighting function, is the estimated signal of the rotor position and speed under the condition of high-frequency pulsating voltage injection, is the estimated signal of the rotor position and speed obtained by the full-order observer; is the estimated signal of the speed of the motor; e is a natural number, is also the estimated signal of the rotor speed of the motor, and n1 and n2 are respectively the estimated signals of the starting rotor speed and the ending rotor speed of the switching process of the smooth switching by using the exponential weighting function module.

[0150] Fig. 12In order to enable the motor to switch smoothly from the high-frequency injection algorithm control stage to the full-order observer control stage, the scheme of the present application proposes to use the switching strategy of the exponential weighting function δ, that is, in the switching stage of the sensorless control algorithm, the rotor electric angle position and the speed estimation value changes according to the exponential weighting function δ. The formula of the exponential weighting function δ is as follows:

[0151]

[0152] is the rotor position and speed estimated by the high-frequency injection, is the rotor position and speed estimated by the full-order observer, and δ is the exponential weighting function; Fig. 12 is a curve diagram of the exponential weighting function δ, and the exponential weighting function δ is as shown in Fig. 14 .

[0153]

[0154] Fig. 14 , wherein 0 to n1, n1 to n2, and n2 correspond to the zero-speed starting stage, the low-speed running stage, and the medium-high-speed running stage, respectively. e is a natural number, is the rotor speed estimation value, n1 and n2 are the starting rotor speed and the ending rotor speed of the control algorithm switching process, respectively, and the effect is shown in Fig. 14 . The unit of the estimated speed is rad / s, and the unit of the rotor speed estimation value is RPM (r / min), that is, the number of revolutions per minute. Usually, the estimated speed of the non-inductive algorithm is in rad / s, but in debugging and daily use, the unit of the speed is often RPM. Since n1 and n2 should be obtained by debugging, for the convenience of debugging, this letter is used to represent it. n1 and n2 should be obtained by debugging, and the two switching points most suitable for debugging are obtained. Fig. 14 is a speed waveform diagram for controlling the algorithm switching in the 200 rpm-400 rpm stage. Fig. 12 The abscissa is time (unit: s), and the ordinate is speed (r / min), Fig. 5 indicates that when n1 and n2 are 200 RPM and 400 RPM obtained by debugging, the speed estimation value follows the actual value well when the two non-inductive algorithms are switched, and there is no large speed fluctuation.

[0155] ​​In the scheme of the application, an exponential weighting function is introduced for switching between the high-frequency injection and full-order observer two kinds of non-inductive algorithms in the estimation of motor rotor position and speed, which ensures the smooth transition of the motor in different operating stages and avoids problems such as jitter and switching failure. Since the high-frequency injection has good low-speed performance and general high-speed performance, switching of the non-inductive algorithm is needed to make the motor control system meet the full-speed threshold control. At the same time, in the scheme of the application, the full-order observer is used in the medium and high-speed operating stage, which further improves the dynamic control performance of the motor under the premise of ensuring the stability of the system, so that the entire control system can maintain efficient, stable and accurate operation in the full-speed range.

[0156] In related schemes, the system is generally directly switched or linearly weighted between the high-frequency injection and full-order observer two kinds of non-inductive algorithms, which may have the problem of large error in the estimation of angle by the high-frequency injection and full-order observer two kinds of non-inductive algorithms, resulting in jitter and switching failure. The scheme of the application uses an exponential weighting function for switching, as shown in Fig. 9 , which exponentially weights the angles observed by the high-frequency injection and full-order observer two kinds of non-inductive algorithms, and can avoid the case of large angle error. The use of the exponential weighting function makes the switching of the motor from low-speed operation to medium and high-speed operation smooth, thereby improving the dynamic control performance.

[0157] By using the technical scheme of the embodiment, for the zero-speed starting stage, low-speed operating stage and medium and high-speed operating stage of the motor, in the zero-speed starting stage and low-speed operating stage, a high-frequency pulsating signal is injected into the d-axis voltage output by the current loop PI controller, so as to convert the analysis of the stator current component in the dq coordinate system into the analysis of the stator current component in the αβ coordinate system, and obtain the position information and speed information of the rotor; in the high-speed operating stage, a full-order observer is used to obtain the position information and speed information of the rotor; and a smooth switching strategy of an exponential weighting function is set to switch the estimated values of the motor rotor position and speed between the high-frequency injection algorithm and the full-order observer; according to the position information and speed information of the rotor, the motor is controlled; thereby, by combining the improved high-frequency injection algorithm with the full-order observer based on the back electromotive force, and using the exponential weighting function to smoothly switch the motor rotor position and speed estimated by the high-frequency injection algorithm and the full-order observer, the complexity and cost of the motor control system are reduced, and the reliability of the motor control system is improved.

[0158] According to the embodiment of the application, a motor control device corresponding to the control method of the motor is also provided. Referring to Fig. 9 , a structural schematic diagram of an embodiment of the device of the application is shown. The motor control system has a current loop and a speed loop; a full-order observation module, a phase-locked loop module and a switching function module are arranged in the speed loop, the full-order observation module is as shown inFig. 9 The full-order observer, the phase-locked module as Fig. 12 The multiplier, the LPF module and the PLL phase-locked loop module, the switching function module as Fig. 5 And The improved high frequency injection in the scheme of the present application will be described in detail below. The exponential weighting function δ as shown; in the scheme of the application, as Fig. 6 The control device of the motor is shown, comprising: an acquisition unit 102 and a control unit 104.

[0159] The control unit 104 is configured to divide the full stage from starting to running after the motor is powered on into a zero-speed starting stage, a low-speed running stage and a medium-high speed running stage; in the zero-speed starting stage of the motor and the low-speed running stage of the motor, a preset frequency pulse voltage is injected into the d-axis part of the PI controller output in the current loop; wherein the preset frequency pulse voltage is a high-frequency pulse voltage injected by using an improved high-frequency pulse injection algorithm. For specific functions and processing of the control unit 104, see step S110.

[0160] The acquisition unit 102 is configured to acquire the three-phase current of the motor in the full stage from starting to running after the motor is powered on; the full stage from starting to running after the motor is powered on includes the zero-speed starting stage of the motor, the low-speed running stage of the motor and the medium-high speed running stage of the motor; that is, the three-phase current of the motor, such as the A-phase current i a , the B-phase current i b and the C-phase current i c , is acquired in the zero-speed starting stage, the low-speed running stage and the medium-high speed running stage. For specific functions and processing of the acquisition unit 102, see step S120.

[0161] The control unit 104 is further configured to, in the full stage from starting to running after the motor is powered on, obtain the rotor position estimation value of the motor and the speed estimation value of the motor by using the full-order observation module, the phase-locked module and the switching function module according to the three-phase current of the motor in the current loop and the speed loop; wherein the estimated rotor position estimation value of the motor, such as the rotor angle estimation value The speed estimation value of the motor, such as the estimated speed For specific functions and processing of the control unit 104, see step S130.

[0162] The control unit 104 is further configured to control the starting and operation of the motor according to the estimated rotor position and the estimated speed of the motor during the entire process from starting to running after the motor is powered on. The specific functions and processing of the control unit 104 are further described in step S140.

[0163] The solution of the present invention combines an improved high-frequency injection algorithm with a full-order observer based on back electromotive force to design a control system for a permanent magnet synchronous motor with a full-speed threshold. This allows the motor to use high-frequency injection during zero-speed startup and low-speed operation, and use a full-order observer during medium- and high-speed operation. An exponential weighting function is used to smoothly switch the motor rotor position and speed estimated by the two algorithms (i.e., the two sensorless algorithms, high-frequency injection and full-order observer). This reduces the complexity and cost of the motor control system, improves the reliability of the motor control system, and solves the problem of switching between position sensorless control algorithms for motors in different operating stages in related solutions. The full-speed threshold refers to the entire process from zero-speed startup to high-speed operation, including zero-speed startup, low-speed operation, and medium- and high-speed operation.

[0164] In some embodiments, the pulse voltage of the preset frequency includes a high-frequency pulse voltage.

[0165] The control unit 104 injects a pulse voltage of a preset frequency into the d-axis portion output by the PI controller in the current loop, including: the control unit 104 is further configured to inject a high-frequency pulse voltage into the d-axis portion output by the PI controller in the current loop according to the following formula:

[0166]

[0167] in, To estimate the rotation coordinate system The d-axis component of the stator high-frequency pulse voltage, U h is the amplitude of the high-frequency pulse voltage, ω h is the electrical angular frequency of the high-frequency pulse voltage, and t is the time.

[0168] Fig. 6

[0169] Fig. 6 This is a schematic diagram of the motor control system. Fig. 6As shown, the AC power provided by the power grid is fed through a three-phase rectifier bridge, bus capacitors, and a three-phase inverter bridge to power the motor. The motor control device includes: a bus voltage sampling circuit, a main controller, a current sampling circuit, a DC / DC (i.e., DC to DC conversion) voltage converter, and a protection circuit. The current sampling circuit samples the three-phase current of the motor from the three-phase windings and feeds it back to the main controller. The target voltage sampling circuit samples the bus voltage from the output end of the bus capacitor and feeds it back to the main controller and the DC / DC voltage converter respectively. The DC / DC voltage converter respectively powers the protection circuit, the main controller, and the current sampling circuit.

[0170] Fig. 7 The schematic diagram of the entire control system is shown in Figure 1. During the zero-speed start and low-speed operation of the motor, the high-frequency pulse injection method is used to estimate the rotor position information. All the following algorithms, including the high-frequency pulse injection algorithm, are run on Fig. 6 in the main controller.

[0171] Fig. 7 It is the structural diagram of the phase current sampling circuit, that is, Fig. 7 The structural diagram of the current sampling circuit is shown in FIG. Fig. 8 As shown, the U phase current I U After passing through resistor R1, it is input to the non-inverting input terminal of the comparator. Resistor R2 and capacitor C2 are connected in parallel between the non-inverting input terminal of the comparator and the analog ground AGND. Capacitor C1 and resistor R3 are connected in parallel between the inverting input terminal of the comparator and the output terminal of the comparator. The output terminal of the comparator is connected to the analog ground AGND after passing through resistor R4 and capacitor C3. The common terminal of resistor R4 and capacitor C3 outputs the sampling current I of the U phase U _DSP to host controller.

[0172] Fig. 8 The diagram shows the sampling circuit diagram of the U-phase current. The sampling of the V-phase current is consistent with that of the U-phase current. Fig. 8 The diagram below shows the relationship between the estimated coordinate system and the actual coordinate system. After Clarke transformation and Park transformation of the U-phase current and the V-phase current, the dq-axis current can be obtained, as shown in Figure 2. Fig. 8 Shown is the estimated rotation coordinate system The relationship between the actual rotating coordinate system dq and the stationary coordinate system αβ, the estimated rotating coordinate system The rotor angle error between the actual rotating coordinate system dq is:

[0173]

[0174] Where, is the estimated value of the rotor angle, θ is the actual value of the rotor angle, and Δθ is the error in the rotor angle estimation.

[0175] Assume that the injection is into the estimated rotation coordinate system The high frequency pulse voltage is:

[0176]

[0177] Where, Estimate the rotation coordinate system The stator high-frequency pulse voltage dq axis component, where To estimate the rotation coordinate system The d-axis component of the stator high-frequency pulse voltage, To estimate the rotation coordinate system The q-axis component of the stator high-frequency pulse voltage; U h is the high-frequency pulse voltage amplitude; ω h is the electrical angular frequency of the high-frequency pulse voltage, and t is the time.

[0178] When analyzing the electrical characteristics of a permanent magnet synchronous motor under high-frequency excitation, if the angular frequency in the motor fundamental wave equation is significantly lower than the angular frequency of the injected high-frequency pulse voltage signal, the inductive reactance in the motor stator winding impedance will dominate the total impedance, and the influence of resistance will be relatively small and can be approximately ignored. Based on this premise, the permanent magnet synchronous motor can be simplified and equivalent to a pure inductance model. This simplification makes it easier to analyze the electromagnetic relationship of the motor under high-frequency excitation, and thus obtain the stator voltage equation of the permanent magnet synchronous motor under high-frequency excitation:

[0179]

[0180] Where u dh 、u qh They are the dq axis components of the stator high frequency pulse voltage, i dh 、i dh are the dq axis components of the stator high-frequency pulsating current respectively, Ld and Lq are the direct-axis and quadrature-axis inductances of the motor, and p is the differential operator.

[0181] Depend on Fig. 9 It can be seen that the estimated rotation coordinate system The relationship between the dq high-frequency pulse voltage and the actual rotating coordinate system is:

[0182]

[0183] according to Fig. 9 The estimated rotated coordinate system shown The relationship between the actual rotating coordinate system dq, the coordinate transformation principle, and the mathematical model (3) under high-frequency excitation of the permanent magnet synchronous motor can be obtained as follows:

[0184]

[0185] where i αh and i βh are stator high-frequency pulsating current αβ-axis components. Transforming the above equation, we have:

[0186]

[0187] Substituting equation (2) and equation (4) into equation (6), we have:

[0188]

[0189] The actual coordinate system stator high-frequency pulsating current αβ-axis components, i.e. the actual coordinate system αβ-axis current response under high-frequency excitation, are as follows:

[0190]

[0191] When the rotor electric angle position estimation value converges to the rotor electric angle position actual value, and the rotor electric angle position estimation error is small enough, we have:

[0192]

[0193] In the field-oriented vector control system of permanent magnet synchronous motor, the sampled stator current αβ-axis components i αβ include: fundamental current i αβf , high-frequency pulsating response current i αβh , and high-order harmonic current i αβx generated by PWM switching signal, i.e.:

[0194] i αβ = i αβf + i αβh + i αβx (11);

[0195]

[0196] where I f and θ f are fundamental current amplitude and phase angle error, respectively. Multiply the αβ-axis current with the modulation signal 2sinω h t, and use a low-pass filter to process the product signal. If the motor works in zero speed and low speed range, the rotor electric angle speed ω is small ω h , i.e. the rotor angle actual value θ = ωt is small ω h t; and the high-order harmonic current i αβx multiplied by the modulation signal is still high-frequency current. The above amplitude modulation and low-pass filtering process can be represented by the following equation:

[0197]

[0198] LPF is the parameter of low pass filter. According to the analysis of the above formula, the stator current αβ axis component i αβ After modulation and low pass filter processing, the current signal i αβ contains the rotor electric angle position information. The last formula (14) has included the rotor angle θ, and the specific waveform is the sine and cosine waveform with certain amplitude, so that the i αβ contains the rotor electric angle position information, and how to extract the angle still needs to use the phase locked loop (PLL) to be introduced below.

[0199] After obtaining the i αβ1 containing the rotor position information, the phase locked loop can be used to estimate the rotor position and speed, which not only has strong stability, but also can effectively suppress and filter the high frequency components in the current signal, and guarantees the accuracy of position and speed estimation.

[0200] In the scheme of the application, the high-frequency pulse signal injection method is adopted, and the current components of the αβ axis are analyzed, which effectively reduces the dependence on the saliency structure of the motor body, and makes the algorithm more universal. In the formula at the end of formula (14), only the d-axis inductance Ld is included in the formula containing the motor angle, and not the difference between the q-axis inductance Lq and the d-axis inductance Ld (the larger the difference Lq-Ld, the more obvious the saliency of the motor), so the dependence on the saliency of the motor body is reduced. In view of the dependence of the high-frequency injection algorithm in the related scheme on the saliency structure of the permanent magnet synchronous motor, the scheme of the application is improved, so that even the motor with not obvious saliency structure of the permanent magnet synchronous motor can also apply the high-frequency injection algorithm, which widens the application range of the algorithm. In the scheme of the application, the observer in the medium and high speed running stage: the full-order observer is used to estimate the motor angle and speed information in the medium and high speed running stage of the motor; the rotor position and speed information extraction method: the components are extracted in the αβ axis, which saves the band pass filter and reduces the use of the filter and the dependence on the saliency structure of the motor body; the inductance-free algorithm switching mode: an exponential weighting function is used. In this way, the improved high-frequency pulse injection algorithm is used in the starting and low speed running stage of the motor to analyze the current components of the shaft, reduce the use of the filter, realize the zero speed starting of the motor, and reduce the dependence on the saliency structure of the motor body; in the medium and high speed running stage, the exponential weighting function is used to switch the rotor position and speed of the motor between the high-frequency injection and the full-order observer two inductance-free algorithms, and gradually switch to the full-order observer, so that the switching process is smooth.

[0201] In some embodiments, the control unit 104, in the current loop and the speed loop, obtains the rotor position estimation value of the motor and the speed estimation value of the motor according to the three-phase current of the motor, by using the full-order observer module, the phase-locked loop module and the switching function module, including:

[0202] The control unit 104 is further configured to obtain the α-axis current of the motor, the β-axis current of the motor, the d-axis current of the motor and the q-axis current of the motor through coordinate transformation according to the three-phase current of the motor; wherein the α-axis current is current i α , the β-axis current is current i β , the d-axis current is current i d , and the q-axis current is current i q . For details, please refer to the example shown in FIG. 10. The specific functions and processes of the control unit 104 also refer to step S210. The full-order observer will be described in detail below.

[0203] The control unit 104 is further configured to obtain the d-axis voltage of the motor output by the PI controller in the current loop and the q-axis voltage of the motor output by the PI controller in the speed loop; wherein the d-axis voltage is voltage u d , and the q-axis voltage is voltage u q . For details, please refer to the example shown in FIG. 11. The specific functions and processes of the control unit 104 also refer to step S220. Fig. 13

[0204] The control unit 104 is further configured to obtain the first estimation signal of the rotor position and speed of the motor by using the full-order observer module based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor and the q-axis current of the motor. The specific functions and processes of the control unit 104 also refer to step S230.

[0205] The control unit 104 is further configured to obtain the second estimation signal of the rotor position and speed of the motor by using the phase-locked loop module based on the α-axis current of the motor, the β-axis current of the motor and the preset modulation signal. The specific functions and processes of the control unit 104 also refer to step S240.

[0206] ​​The control unit 104 is further configured to, based on the current stage of the motor being in a zero-speed starting stage, a low-speed running stage, and a medium-to-high-speed running stage, according to the first estimation signal of the rotor position and speed of the motor and the second estimation signal of the rotor position and speed of the motor, utilize the switching function module to perform smooth switching to obtain the rotor position estimation value of the motor in the current stage of the motor and the rotor speed estimation value of the motor. The specific functions and processes of the control unit 104 are also described in step S250.

[0207] In the scheme of the application, by injecting a high-frequency pulsating signal into the d-axis of the two-phase rotating coordinate system output by the current loop PI of the control system of the motor, the analysis of the stator current components in the dq coordinate system in the related scheme is converted into the analysis of the stator current components in the αβ coordinate system, so as to obtain the position information and speed information of the rotor. This method not only saves the band-pass filter and simplifies the system structure, but also significantly improves the dynamic control performance of the system. In the analysis of the stator current components in the αβ axis, the system structure is simplified because the band-pass filter is no longer needed. The system bandwidth is increased and the hysteresis is reduced because the band-pass filter is less used, so the dynamic control performance of the system is improved. In view of the problem that the high-frequency injection algorithm in the related scheme uses a large number of filters in the signal processing process. The scheme of the application saves the band-pass filter, simplifies the control system structure of the permanent magnet synchronous motor, and improves the efficiency and reliability of the control system of the permanent magnet synchronous motor.

[0208] In some embodiments, the full-order observation module includes a full-order observer, and the control unit 104 utilizes the full-order observation module to obtain the first estimation signal of the rotor position and speed of the motor based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor, and the q-axis current of the motor, including:

[0209] The control unit 104 is further configured to utilize the full-order observer to obtain the first estimation signal of the rotor position and speed of the motor based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor, and the q-axis current of the motor according to the following formula:

[0210]

[0211] wherein Δθ is the rotor angle estimation error, is the estimated value of the d-axis electromotive force observed by the full-order observer, is the estimated value of the q-axis electromotive force observed by the full-order observer. is the estimation signal of the rotor position of the motor, k p is the proportional coefficient of the PI controller, ki kint is an integral coefficient of the PI controller, is an estimated signal of the speed of the motor; the estimated signal of the rotor position of the motor and the estimated signal of the speed of the motor are taken as the first estimated signals of the rotor position and the speed of the motor.

[0212] Fig. 13

[0213] In the high-speed running stage of the motor, the full-order observer is used to estimate the position information of the rotor. The dynamic equation of the state observer is: where H is a feedback matrix. According to the controllability and observability of the control system, the four poles of the observer are configured at -ω k , and the feedback matrix H is:

[0214]

[0215] where R s is the stator resistance of the motor, ω k is the system bandwidth, and the dynamic equation of the observer is:

[0216]

[0217] First, the observation variable ω is calculated by formula (17). Then, the estimated angle error is calculated:

[0218]

[0219] According to the design of the PI regulator (i.e., the PI controller) based on the position error, the speed ω is observed.

[0220]

[0221] where k p is the proportional coefficient of the PI controller, and k i is the integral coefficient of the PI controller. The rotor position angle θ is obtained by integrating the speed ω

[0222] .

[0223] The estimated rotor angle is shown in FIG. 4. Fig. 9 FIG. 4 is a curve diagram of the estimated rotor angle. Fig. 1

[0224] In the solution of the present invention, the use of the full-order observer in the medium and high-speed operation stage fully utilizes its high adaptability and high-precision estimation ability to the changes in motor parameters during high-speed operation. Under the premise of ensuring the stability of the motor control system, the dynamic control performance of the motor is further improved, so that the motor can maintain an efficient and accurate operating state throughout the entire speed range, meeting the requirements of modern industrial control for high performance, high reliability and a wide speed regulation range of permanent magnet synchronous motor systems.

[0225] In some implementations, the phase-locked module includes: a multiplier, an LPF filter, and a phase-locked loop arranged in sequence.

[0226] The control unit 104 obtains a second estimated signal of the rotor position and speed of the motor using the phase-locked module based on the α-axis current of the motor, the β-axis current of the motor, and a preset modulation signal, including:

[0227] The control unit 104 is further configured to obtain a product value by passing the α-axis current of the motor, the β-axis current of the motor, and the preset modulation signal through the multiplier. The specific functions and processing of the control unit 104 are also shown in step S310.

[0228] The control unit 104 is further configured to obtain a filtered value by passing the product value through the LPF filter. The specific functions and processing of the control unit 104 are also shown in step S320.

[0229] The control unit 104 is further configured to pass the filtered value, the electromagnetic torque of the motor, and the load torque of the motor through the phase-locked loop to obtain an estimated signal of the rotor position and speed of the motor as a second estimated signal of the rotor position and speed of the motor. The specific functions and processing of the control unit 104 are further described in step S330.

[0230] Fig. 2 The control flow diagram of the permanent magnet synchronous motor control system is shown in Figure 2, where (a) is a schematic diagram of the control flow of the permanent magnet synchronous motor control system. Fig. 9 , (b) is a schematic diagram Fig. 10 .like Fig. 9 As shown, the d-axis given current i dref = 0, and the actual d-axis current i output by Park transformation d , is input to the PI controller after passing through the comparator; the PI controller outputs the d-axis voltage u d , and the high-frequency pulse voltage u injected by the high-frequency pulse h cosω h After passing through the comparator, t is input to the Park inverse transformation module. Given speed ω ref , and the estimated speed output by the exponential weighting function After passing through the comparator, it is input to the PI controller, and the PI controller outputs the q-axis given current i qref , and the actual q-axis current i output by the Park converter q , which is output to the PI controller after passing through the comparator. The PI controller outputs the q-axis voltage u q To the Park inverse transform module. The Park inverse transform module outputs the rotor angle estimate based on the exponential weighting function Output α-axis voltage u α , β-axis voltage u β To the space vector pulse width modulation (SVPWM) module. The SVPWM module outputs the control signal to the three-phase inverter, and the three-phase inverter outputs the three-phase current to the motor. The three-phase current i a 、i b and i c After Clarke transformation, the output α-axis current i α , β-axis current i β To the Park transform module. The Park transform module outputs the rotor angle estimate based on the exponential weighting function Output d-axis actual current i d , q-axis actual current i q d-axis voltage u d , q-axis voltage u q , d-axis actual current i d , q-axis actual current i q , input to the full-order observer. The stator current αβ axis component i output by Clarke transformation αβ With the modulation signal 2sinω h t multiplied, and the product signal is processed using a low-pass filter (LPF) to obtain signal i αβ1 , and then input to the PLL phase-locked loop. The output signal of the full-order observer and the output signal of the PLL phase-locked loop are output after the exponential weighting function to estimate the speed. and the rotor angle estimate

[0231] Fig. 10 This is a schematic diagram of the control flow of the rotor electrical angle position phase-locked loop tracking observer. Fig. 10 The control flow diagram of the PLL phase-locked loop part in FIG. Fig. 10 As shown, the signal i β1 and The product value of, and signal i α1 and The difference ε of the product value is obtained by K p +k i / s module and then output to the comparator. The electromagnetic torque T of the motor e , the motor's load torque Tl The output signal of the comparator is sent to the 1 / J module, 1 / s module, and n p Module output estimated speed Estimated speed Output rotor angle estimation after passing through 1 / s module like Fig. 11 The figure shows the implementation block diagram of the rotor position phase-locked loop tracking observer. Fig. 9 where ε is obtained by signal processing of the phase-locked loop feedback loop and linearization of the error signal, as shown in the following formula:

[0232]

[0233] Among them, signal i α1 , signal i β1 is signal i αβ1 The α and β axis components of the motor are shown in the above formula as the motor rotor position angle estimation error signal value obtained by using the improved high frequency injection method.

[0234] Fig. 11 This is a schematic diagram of the implementation process of the rotor electrical angle position estimation using the high-frequency pulse voltage injection method. Fig. 10 The part indicated by the blue line in Figure (b) is Fig. 11 Observer_PLL is Fig. 11 The part shown. The exponential weighting function will be described in detail below The block diagram of the improved high frequency injection method for estimating rotor angular speed is shown in Figure 2. Fig. 12 As shown, the signal i A and signal i B After the coordinate transformation of ABC / αβ, the stator current αβ axis component i is obtained αβ , stator current αβ axis component i αβ With the modulation signal 2sinω h t multiplied, and the product signal is processed using a low-pass filter (LPF) to obtain signal i αβ1 , and then input to Observer_PLL (flux observer + phase-locked loop). Observer_PLL (flux observer + phase-locked loop) is based on the electromagnetic torque T of the motor. e , the motor's load torque T l , output estimated speed and the rotor angle estimate

[0235] Compared with the high-frequency injection method in the related scheme, the improved high-frequency pulse voltage injection method realizes significant optimization in the generation mechanism of the rotor electric angle position estimation error signal. Specifically, the improved error signal value is no longer dependent on the salient pole structure or saturated salient pole of the motor, but is only associated with the parameter of the stator winding d-axis inductance Ld. This characteristic makes the method have more extensive applicability and higher adaptability when facing motors with different salient pole characteristics. Moreover, the improved algorithm successfully eliminates the use of a band-pass filter. This change not only simplifies the system structure and reduces the complexity of the system, but also effectively avoids the problems caused by the use of a band-pass filter in the related scheme, such as unnecessary attenuation of the system signal amplitude and phase angle delay. These problems often have a negative impact on the dynamic performance of the system in the control system of the related scheme. The improved method in the scheme of the present application significantly improves the dynamic control performance of the system, and improves the control accuracy and response speed by optimizing the system architecture.

[0236] In some embodiments, the control unit 104, based on the current stage of the motor in the zero-speed starting stage, low-speed running stage and medium-high speed running stage of the motor, uses the switching function module to perform smooth switching according to the first estimated signal of the rotor position and speed of the motor and the second estimated signal of the rotor position and speed of the motor, to obtain the rotor position estimation value of the motor in the current stage of the motor and the speed estimation value of the motor, including:

[0237] The control unit 104 is specifically further configured to determine whether the current stage of the motor is in the zero-speed starting stage, low-speed running stage, or medium-high speed stage of the motor. The specific functions and processes of the control unit 104 are also referred to as step S410.

[0238] The control unit 104 is specifically further configured to use the switching function module to perform smooth switching according to the first estimated signal of the rotor position and speed of the motor and the second estimated signal of the rotor position and speed of the motor, to execute step S430 or step S440. The specific functions and processes of the control unit 104 are also referred to as step S420.

[0239] The control unit 104 is specifically further configured to, if it is determined that the current stage of the motor is in the zero-speed starting stage, low-speed running stage of the motor, use the second estimated signal of the rotor position and speed of the motor as the rotor position estimation value of the motor and the speed estimation value of the motor. The specific functions and processes of the control unit 104 are also referred to as step S430.

[0240] The control unit 104 is further configured to, if it is determined that the current stage of the motor is in the medium-high speed running stage of the motor, take the first estimated signal of the rotor position and the speed of the motor as the rotor position estimation value of the motor and the speed estimation value of the motor. The specific functions and processes of the control unit 104 are also described in step S440.

[0241] The related scheme is to replace the band-pass filter with a two-stage series pure delay filter to extract the high-frequency current response, while the scheme of the present application is to extract the high-frequency current response by analyzing the stator current components of the αβ axis. The scheme of the present application uses an improved high-frequency pulse injection algorithm during the motor zero-speed starting and low-speed running stage, and innovatively analyzes the current components of the αβ axis, thereby effectively saving the use of the band-pass filter. This improvement not only realizes the zero-speed starting of the motor, but also significantly reduces the dependence on the salient structure of the motor body, making the algorithm more universal and practical. By reducing the use of the band-pass filter, the structure of the motor control system is simplified, the complexity and potential failure points of the motor control system are reduced, and the overall reliability and stability are improved. At the same time, the simplified motor control system is also more advantageous in terms of cost and maintenance, providing a more economical and efficient solution for industrial applications.

[0242] When the motor enters the medium-high speed running stage, the scheme of the present application introduces a smooth switching strategy based on an exponential weighting function for switching the estimated values of the motor rotor position and speed between the high-frequency injection algorithm and the full-order observer. The related scheme uses a new variable gain sliding mode observer method with an S-curve function and resistance feedback when the motor is running at medium-high speed, and the switching algorithm is linear weighting switching. Alternatively, the related scheme uses a sliding mode observer when the motor is running at medium-high speed, and the switching algorithm is linear weighting switching. However, the scheme of the present application uses a full-order observer when the motor is running at medium-high speed, and the switching algorithm is exponential weighting switching. Compared with the switching method of the related scheme, the exponential weighting function can dynamically adjust the weights of the two algorithms (i.e. high-frequency injection and full-order observer) according to the running state of the motor, ensuring the smoothness and stability of the switching process. This smooth transition effectively avoids problems such as motor jitter and torque fluctuation that may be caused by sudden switching of the control algorithm, improving the smoothness and comfort of the motor operation. In addition, the use of the full-order observer during the medium-high speed running stage fully utilizes its high adaptability to parameter changes of the motor and high-precision estimation capability during high-speed running, further improving the dynamic control performance of the motor under the premise of ensuring the stability of the motor control system, so that the motor can maintain efficient and accurate operation in the entire speed range, meeting the requirements of modern industrial control for high performance, high reliability and wide speed regulation range of permanent magnet synchronous motor systems.

[0243] In some embodiments, the switching function module comprises an exponential weighting function module.

[0244] The control unit 104 uses the exponential weighting function module to perform smooth switching according to the first estimated signal of the rotor position and speed of the motor and the second estimated signal of the rotor position and speed of the motor, comprising:

[0245]

[0246] wherein, is the estimated signal of the rotor position of the motor, and δ is an exponential weighting function, is the estimated signal of the rotor position and speed under high-frequency pulsating voltage injection, is the estimated signal of the rotor position and speed obtained by the full-order observer; is the estimated signal of the speed of the motor; e is a natural number, is also the estimated signal of the rotor speed of the motor, and n1 and n2 are respectively the estimated signals of the starting rotor speed and the ending rotor speed of the switching process using the exponential weighting function module to perform smooth switching.

[0247] Fig. 12 In order to enable the motor to be smoothly switched from the high-frequency injection algorithm control stage to the full-order observer control stage, the scheme of the present application proposes a switching strategy using an exponential weighting function δ, that is, during the switching stage of the sensorless control algorithm, the rotor electrical angle position and the speed estimation value vary according to the exponential weighting function δ. The formula of the exponential weighting function δ is as follows:

[0248]

[0249] wherein, is the rotor position and speed estimated by high-frequency injection, is the rotor position and speed estimated by the full-order observer, and δ is an exponential weighting function; Fig. 12 is a schematic diagram of the curve of the exponential weighting function δ, and the exponential weighting function δ is as shown in Fig. 14 .

[0250]

[0251] wherein, Fig. 14 wherein, 0 to n1, n1 to n2, and n2 thereafter correspond to the zero-speed starting stage, the low-speed running stage, and the medium-to-high-speed running stage respectively. e is a natural number, is the rotor speed estimation value, n1 and n2 are respectively the starting rotor speed and the ending rotor speed of the control algorithm switching process, and the effect is shown in Fig. 12 .​ A schematic diagram of a speed waveform for switching control algorithms in the 200rpm-400rpm phase.

[0252] In the scheme of the application, an exponential weighting function is introduced for switching between the high-frequency injection and full-order observer two sensorless algorithms in the estimation of the motor rotor position and speed. This switching method ensures smooth transition of the motor in different operating phases, avoiding problems such as jitter and switching failure. Since the high-frequency injection has good low-speed performance and general high-speed performance, switching of the sensorless algorithm is needed to make the motor control system meet the full-speed threshold control. At the same time, in the scheme of the application, a full-order observer is used in the medium and high-speed operating phase, which further improves the dynamic control performance of the motor under the premise of ensuring system stability, so that the entire control system can maintain efficient, stable and accurate operation in the full-speed range.

[0253] In related schemes, the system is generally a direct switching or linear weighting switching between the high-frequency injection and full-order observer two sensorless algorithms, which may have the problem of large error in the estimation of the angle by the high-frequency injection and full-order observer two sensorless algorithms, leading to jitter and switching failure. The scheme of the application uses an exponential weighting function for switching, as shown in ​ which exponentially weights the angles observed by the high-frequency injection and full-order observer two sensorless algorithms, avoiding the case of large angle error. The use of the exponential weighting function makes the switching of the motor from low-speed operation to medium and high-speed operation smooth, thus improving the dynamic control performance.

[0254] Since the processing and functions realized by the device of the embodiment basically correspond to the embodiments, principles and examples of the foregoing method, details not described in the description of the embodiment can be referred to the related description in the foregoing embodiments, which will not be repeated here.

[0255] According to the embodiments of the application, a motor corresponding to the motor control device is also provided. The motor can include the motor control device described above.

[0256] Since the processing and functions realized by the motor of the embodiment basically correspond to the embodiments, principles and examples of the foregoing device, details not described in the description of the embodiment can be referred to the related description in the foregoing embodiments, which will not be repeated here.

[0257] According to the embodiments of the application, a computer program product corresponding to the motor control method is also provided, including a computer program which, when executed by a processor, realizes the steps of the motor control method described above.

[0258] Since the processing and functions realized by the product of the embodiment are basically corresponding to the embodiments, principles and examples of the foregoing method, the unexplained parts in the description of the embodiment can be seen in the related description in the foregoing embodiments, which will not be repeated here.

[0259] According to the embodiment of the present application, a storage medium corresponding to the control method of the motor is also provided, which comprises a stored program, wherein when the program is running, the device where the storage medium is located executes the steps of the control method of the motor as described above.

[0260] Since the processing and functions realized by the storage medium of the embodiment are basically corresponding to the embodiments, principles and examples of the foregoing method, the unexplained parts in the description of the embodiment can be seen in the related description in the foregoing embodiments, which will not be repeated here.

[0261] In summary, the skilled in the art can easily understand that the above-mentioned advantageous modes can be freely combined and superimposed without conflict.

[0262] The above only describes the embodiments of the present application and is not used to limit the present application. The present application can have various changes and variations for the skilled in the art. Any modification, equivalent replacement, improvement, etc. made within the spirit and principles of the present application shall be included in the scope of claims of the present application.

Claims

1. A control method of an electric motor, characterized by, The control system of the motor has a current loop and a speed loop; a full-order observation module, a phase-locked module and a switching function module are arranged in the speed loop; The control method of the motor comprises: The full stage from starting to running after the motor is powered on is divided into a zero-speed starting stage, a low-speed running stage and a medium-high speed running stage; in the zero-speed starting stage of the motor and the low-speed running stage of the motor, a preset frequency pulse voltage is injected into the d-axis part of the PI controller output in the current loop; In the full stage from starting to running after the motor is powered on, the three-phase current of the motor is obtained; In the full stage from starting to running after the motor is powered on, the rotor position estimation value of the motor and the speed estimation value of the motor are obtained by using the full-order observation module, the phase-locked module and the switching function module according to the three-phase current of the motor in the current loop and the speed loop; In the full stage from starting to running after the motor is powered on, the starting and running of the motor are controlled according to the rotor position estimation value of the motor and the speed estimation value of the motor.

2. The control method of an electric motor according to claim 1, characterized by, The preset frequency pulse voltage comprises a high-frequency pulse voltage; Injecting the preset frequency pulse voltage into the d-axis part of the PI controller output in the current loop comprises: The high-frequency pulse voltage is injected into the d-axis part of the PI controller output in the current loop according to the following formula: wherein for estimating the d-axis component of the stator high-frequency ripple voltage of the rotating coordinate system U h is the amplitude of the high-frequency ripple voltage, ω h is the electrical angular frequency of the high-frequency ripple voltage, t is time.

3. The control method of an electric motor according to claim 1 or 2, characterized by, In the current loop and the speed loop, the rotor position estimation value of the motor and the speed estimation value of the motor are obtained by using the full-order observation module, the phase-locked module and the switching function module according to the three-phase current of the motor, comprising: According to the three-phase current of the motor, the alpha-axis current of the motor, the beta-axis current of the motor, the d-axis current of the motor and the q-axis current of the motor are obtained through coordinate transformation; In the current loop, the d-axis voltage of the motor output by the PI controller is obtained; in the speed loop, the q-axis voltage of the motor output by the PI controller is obtained; Based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor and the q-axis current of the motor, the first estimation signal of the rotor position and speed of the motor is obtained by using the full-order observation module; Based on the alpha-axis current of the motor, the beta-axis current of the motor and the preset modulation signal, the second estimation signal of the rotor position and speed of the motor is obtained by using the phase-locked module; Based on the current stage of the motor in the zero-speed starting stage, the low-speed running stage and the medium-high speed running stage, the rotor position estimation value of the motor and the speed estimation value of the motor in the current stage of the motor are obtained by using the switching function module to perform smooth switching according to the first estimation signal of the rotor position and speed of the motor and the second estimation signal of the rotor position and speed of the motor.

4. The control method of an electric motor according to claim 3, characterized by The full-order observation module comprises a full-order observer; Based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor and the q-axis current of the motor, the full-order observer is used to obtain the first estimation signal of the rotor position and speed of the motor, including: Based on the d-axis voltage of the motor, the q-axis voltage of the motor, the d-axis current of the motor and the q-axis current of the motor, the full-order observer is used to obtain the first estimation signal of the rotor position and speed of the motor according to the following formula: wherein Δθ is a rotor angle estimation error, Ed is an estimated value of a d-axis electromotive force observed by the full-order observer, Eq is an estimated value of a q-axis electromotive force observed by the full-order observer; θ is an estimated signal of a rotor position of the motor, k p kP is a proportional coefficient of a PI controller, k i kI is an integral coefficient of the PI controller, ω is an estimated signal of a rotational speed of the motor; the estimated signal of the rotor position of the motor and the estimated signal of the rotational speed of the motor are used as first estimated signals of the rotor position and the rotational speed of the motor.

5. The control method of an electric motor according to claim 3, characterized by, The phase-locked loop module includes a multiplier, an LPF filter and a phase-locked loop arranged in sequence; Based on the α-axis current of the motor, the β-axis current of the motor and the preset modulation signal, the phase-locked loop module is used to obtain the second estimation signal of the rotor position and speed of the motor, including: The α-axis current of the motor, the β-axis current of the motor and the preset modulation signal are multiplied by the multiplier to obtain a product value; The product value is filtered by the LPF filter to obtain a filtered value; The filtered value, the electromagnetic torque of the motor and the load torque of the motor are input into the phase-locked loop to obtain the estimation signal of the rotor position and speed of the motor as the second estimation signal of the rotor position and speed of the motor.

6. The control method of an electric motor according to any one of claims 3 to 5, characterized by, Based on the current stage of the motor in the zero-speed starting stage, the low-speed running stage and the medium-high speed running stage, the first estimation signal of the rotor position and speed of the motor and the second estimation signal of the rotor position and speed of the motor are input into the switching function module for smooth switching to obtain the rotor position estimation value and the speed estimation value of the motor in the current stage of the motor, including: Determine whether the current stage of the motor is in the zero-speed starting stage, the low-speed running stage or the medium-high speed stage of the motor; According to the first estimation signal of the rotor position and speed of the motor and the second estimation signal of the rotor position and speed of the motor, the switching function module is used for smooth switching, including: If it is determined that the current stage of the motor is in the zero-speed starting stage or the low-speed running stage of the motor, the second estimation signal of the rotor position and speed of the motor is taken as the rotor position estimation value and the speed estimation value of the motor; If it is determined that the current stage of the motor is in the medium-high speed running stage of the motor, the first estimation signal of the rotor position and speed of the motor is taken as the rotor position estimation value and the speed estimation value of the motor.

7. The control method of an electric motor according to claim 3 or 6, characterized by, The switching function module includes an exponential weighting function module; According to the first estimation signal of the rotor position and speed of the motor and the second estimation signal of the rotor position and speed of the motor, the exponential weighting function module is used for smooth switching, including: wherein, is an estimation signal of the rotor position of the motor, and δ is an exponential weighting function, is an estimation signal of the rotor position and speed under high-frequency pulsating voltage injection, is an estimation signal of the rotor position and speed obtained by a full-order observer; is an estimation signal of the rotor speed of the motor; e is a natural number, is also an estimation signal of the rotor speed of the motor, and n1 and n2 are respectively an estimation signal of the starting rotor speed and an estimation signal of the ending rotor speed of the switching process using the exponential weighting function module for smooth switching.

8. A control device of an electric motor characterized by comprising: The control system of the motor has a current loop and a speed loop; the full-order observer, the phase-locked loop module and the switching function module are arranged in the speed loop; The control device of the motor includes: The control unit is configured to divide the full stage of starting to running after the motor is powered on into a zero-speed starting stage, a low-speed running stage and a medium-high speed running stage; in the zero-speed starting stage of the motor and the low-speed running stage of the motor, a d-axis part of the output of the PI controller in the current loop is injected with a preset frequency of pulse voltage; The acquisition unit is configured to acquire three-phase currents of the motor in the full stage of starting to running after the motor is powered on; The control unit is further configured to, in the full stage of starting to running after the motor is powered on, obtain a rotor position estimation value of the motor and a rotating speed estimation value of the motor according to the three-phase currents of the motor in the current loop and the speed loop by using the full-order observation module, the phase-locked module and the switching function module; The control unit is further configured to control starting and running of the motor according to the rotor position estimation value of the motor and the rotating speed estimation value of the motor in the full stage of starting to running after the motor is powered on.

9. An electric machine characterized by The motor control device comprises: The motor control device according to claim 8.

10. A storage medium, characterized by The storage medium comprises a stored program, wherein the program controls a device where the storage medium is located to execute the motor control method according to any one of claims 1 to 7 when the program is executed.

11. A computer program product comprising a computer program, characterized in that, The computer program is executed by a processor to implement the steps of the motor control method according to any one of claims 1 to 7.

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