Drive circuit for clocking the control input of the power transistor of a converter and use thereof

Abstract

The invention relates to a drive circuit for the control input of a power transistor in a clocked converter, in particular in a bridge leg, which comprises two power transistors to be connected. The drive circuit is coupled in parallel to the control input of each power transistor. The drive circuit comprises two first auxiliary transistors, which are connected together at their emitter at a first connection point. One PNP and one NPN auxiliary transistor are operated in a base circuit, respectively. Each drive circuit is activated by applying a current to its first connection point. The activation current is a charging or discharging current of at least one capacitor, which is also connected to the first connection point. The at least one capacitor is connected in parallel to the output of the bridge leg. The combination of the bridge capacitor and / or the activation capacitor with the drive circuit causes the control input of the power transistor connected thereto to be short-circuited for as long as the operating voltage of the same power transistor changes, in order to place it in a standby position for this length of time.

Description

TECHNICAL FIELD

[0001] The present invention relates to a drive circuit for a bridge leg of an electrical inverter, which is designed self-oscillating or for external control or for a high-side driver. The invention especially relates to a drive circuit for a control input between a control electrode and a reference electrode of a power transistor. BACKGROUND

[0002] Even in the age of digitization, which is accompanied by an increasing integration of functions and now extends to the control of personal devices or end devices, self-oscillating circuits retain their importance wherever the disadvantages of digitization prevail. This is given, among other things, in the case of high demands on at least one of the following parameters:

[0003] - reliability

[0004] - robustness

[0005] - independence

[0006] - efficiency

[0007] - electromagnetic compatibility

[0008] - resolution of amplitude or frequency

[0009] - avoidance of aliasing effects

[0010] - spontaneous synchronization.

[0011] It can be seen from this that self-oscillators are, for example, destined in many cases to supply electrical energy, which essentially equals a conversion of this energy. One of the greatest disadvantages of self-oscillators is their difficulty to be controlled from the outside, which is why they often occur in systems which are highly independent and thus require little or no information from the outside, for example in converters for a known input voltage and a known load, which have strict tolerances, respectively. As soon as the input voltage is applied, the converter starts and performs the conversion. In this case, for example, advanced lighting technology exhibits a large field of application of such self-oscillators in the case of light sources which run at a frequency which is significantly higher than the supply frequency.

[0012] The first widely known self-oscillator excited by magnetic positive feedback is a half-bridge inverter consisting of NPN bipolar transistors, whose control inputs, i.e. the lines between the base and the emitter, respectively, are connected to the oppositely oriented drive windings of a three-winding drive transformer. Its measurement winding is connected in series with the midpoint of the half-bridge, so that the current output to the load is measured. Here, negative feedback occurs by means of the periodic saturation of the drive transformer shortly before the desired turn-off point of the just conducting transistor. Therefore, this type of drive is referred to in the following as "saturation drive".

[0013] In order to make this negative feedback current-dependent, a relatively high-impedance resistor is connected in series to each base, and each transistor is controlled at a level far above its saturation current gain. In this way, each drive winding "sees" a notable voltage-time area, thus driving the transformer into saturation, which depends on the base current and thus on the half-bridge output current. A further advantage of this circuit is the time needed for the drive transformer to recover from its saturation again. The half-bridge can utilize this so-called dead-time or duty gap for polarity inversion of its midpoint, so that a relatively large ladder capacitor can be implemented, which connects the midpoint of the bridge leg to which it is connected with at least one pole of the supply DC voltage of the same bridge leg. A third advantage is that the hotter the transformer, the earlier it saturates. This counteracts or even overcompensates the inverse behavior of bipolar transistors, which slows down upon heating. Thus, a bipolar self-oscillator with saturation drive is inherently temperature-stable. All this means that this variant has remained the dominant inverter circuit for the electronic operation of fluorescent lamps, especially in the form of energy-saving lamps, for more than ten years.

[0014] The many disadvantages described above forced further development. Since the circuit described above depends not only on a number of parameters of the transistors, which are difficult to limit, but also on parameters of the ferrite core of the drive transformer, which are likewise difficult to limit. The biggest disadvantage, however, is the limitation to a maximum achievable inverter efficiency of about 90%. The overdrive and the main cause of this by the base series resistance are at least partially eliminated by the teaching of EP 982 862 B2 ("Low resistance bipolar bridge circuit"), in which the base current is reduced and the base series resistance is replaced by a base series capacitor, whose voltage spans the time area of the transformer saturation.

[0015] The duty gap, however, is thus largely lost, so that the ladder capacitor is used in order to recover the duty gap and at the same time to control. In EP 992 113 B2 ("Half-bridge drive without collector bias current peak"), this half-bridge is connected directly between the base terminals of both transistors transversely to the entire half-bridge, and thus the time in which the transformer remains saturated is as long as the time needed by the half-bridge for polarity inversion. The bipolar transistors, which are already slow, however, become even slower upon switching off, since this position of the ladder capacitor makes the Miller capacitances of both transistors increase their values, respectively. This, i.e. the current overload of the still underlying saturation drive and the basic series capacitor, does not ultimately lead to a significant efficiency increase, which is why this variant has not yet found a foothold on the market.

[0016] In another method, saturation is replaced by the inherent magnetization of the drive transformer for the negative feedback. This method is therefore referred to in the following as "flux drive". For this purpose, the drive transformer must be changed in its construction, either by a correspondingly small construction form or by an air gap, or an external inductance is connected in parallel. Since its magnetizing inductance essentially determines the converter frequency and therefore must be relatively small. This drive transformer almost completely loses its sensitivity to the core tolerances, since it is electrically or mechanically regulated by the parallel circuit. Compared to all saturation drives, the base current can be further reduced, the transistor can be fully utilized up to its saturation current gain. The drive circuit together shrinks onto the individual unique drive winding, which first feeds back the excess basic energy into the half-bridge, since the drive transformer always intervenes as a linear transformer. Now the important transistor parameters are its current gain and its base-emitter saturation voltage, both of which influence the converter frequency and are precisely specified.

[0017] Due to the uninterrupted linearity of the drive transformer, both half-bridge transistors can in most cases even turn off with a negative base current, the value of which corresponds approximately to half the collector current, which is still positive at the time. This is the fourth frequency-determining parameter: the transistors independently and autonomously determine and optimize their turn-off process for each cycle. The less active power is transmitted, the more negative the base current is relative to the collector current. The self-oscillator with flux drive achieves inverter efficiencies of over 98%. But this inverter has no space for a polarity-changing gap in principle, so there the ladder capacitor is to be avoided.

[0018] The teaching of EP 1 071 210 B1 ("turn-off acceleration") offers a remedy, which not only allows the ladder capacitor again, but even requires it. However, unlike EP 982 862 B2 or EP 992 113 B2, it is not directly connected between the base terminals here, but rather two base terminals are connected here essentially in parallel with the operating path of the associated power transistor and drive an NPN auxiliary transistor, which short-circuits the drive of the associated power transistor. An additional diode network is responsible for the necessary charge balance in the ladder capacitor, where a single such diode network can also be connected directly between the control electrodes of the auxiliary transistor and then be used double. The almost immeasurable positive influence of this development is not proportional to the enormous space and cost overhead, which is why it has not been used.

[0019] In EP 1 071 210 B1 auxiliary transistors are applied in the emitter circuit, whose emitters are each at the reference potential of the associated power transistor. For the already mentioned charge balancing, diodes are responsible, which are each connected anti-parallel to the base-emitter path of the auxiliary transistor.

[0020] A variant is also known, in which the same auxiliary transistors are each operated in the basic circuit. At least one ladder capacitor is there connected to the emitter of the auxiliary transistor. The diodes for charge balancing remain in place, but have the opposite polarity.

[0021] Neither of the two variants with a single NPN auxiliary transistor provides the necessary variability or scalability between the two independent variables "load current" and "magnetization current in the magnetic core of the drive transformer". The "turn-off acceleration" is therefore even more suitable for a half-bridge with saturated drive than it is actually designed for, and has been used in this way by at least one major lighting manufacturer.

[0022] In general, not only too small a negative base current is detrimental to the turn-off speed of a bipolar power transistor, but also too large, in particular pulse-shaped, base currents. If this current is even larger than the collector current at the turn-off point in time, the base is actively removed, instead of allowing its residual charge to be consumed by conducting a part of the collector current. The base-collector diode is then also actively removed, where not only the voltage reference of the entire power transistor with respect to its emitter is lost, but the removal itself is similar to a hard turn-off of a freewheeling diode. The power transistor generates the excess current almost from itself by establishing a collector voltage with corresponding losses.

[0023] The trigger device is therefore not suitable for external control of a bipolar self-oscillator. Rather, the shape of the collector current and the base current should be similar: the proportion of the direct current component corresponds to the current gain currently present, the curvature of the base current corresponds to the turns ratio between the measurement winding and the drive winding of the drive transformer, which is smaller than the curvature of the collector current, and the negative ramp of the drive transformer magnetization current is superimposed on the base current. The invention builds on this most complex of all driving situations, because the highest efficiency can be expected there, and because a solution for this complex case can more easily be scaled down to simpler cases, but not vice versa.

[0024] All variants described so far require a powerful trigger device as start-up circuit, and thus always also a so-called "quenching circuit" which "holds" the trigger as long as the inverter is oscillating, thus preventing the trigger from pulsing into the running self-oscillator in an asynchronous and sporadic manner. As an improvement over the known discharge diode (sometimes also called "dead-time diode"), whose anode is at the positive pole of the trigger buffer capacitor and whose cathode is at the half-bridge midpoint, such a quenching circuit is disclosed in EP 0 682 464 B1 as emitter circuit and in EP 2298 036 B1 as basic circuit, which here is applied in that the R6 there is particularly advantageously not connected in series with the base of the quenching transistor T3, but is connected in series with its emitter with significantly lower impedance. The advantage of the basic circuit itself is its phase-correct intervention after the actual trigger pulse, the here shifting of the resistance is independent of the current gain of the quenching transistor, and this transistor is better protected.

[0025] In parallel to all of the above there is a variant which first no longer requires a trigger to start up, but only a slow and high-impedance start-up aid. If the level is reached, the inverter starts up and the start-up aid is bridged by a low-impedance path at high frequency. In addition, the drive transformer is dispensed with, which is replaced by at least two control windings which are directly coupled to the always present converter inductances. Finally, MOS field-effect transistors are used as power switches. Since the voltage over each inductance (= case here) leads the current through the same inductance (= case above), the gate controller here consists of a phase shifter or phase regulator in order to be able to drive the bridge legs, for example, phase-correctly. These relatively high-impedance drive circuits additionally open up significantly more possibilities for controlling the inverter from the outside. One of the basic patents of this technology (even without the high-impedance start-up aid) is the "phase regulator" EP 0 781 077 B1, which is a feature of the second generation of energy-saving lamps. The complete control and regulation of the half-bridge is taken over by the "AS" block, which consists of two control windings of opposite polarity, a series resistance for each gate controller and a parallel resonance circuit.

[0026] In a further embodiment, the "LED phase regulator" DE 10 2017 105 560 Al, the same scheme is extended to LED operation by inserting the HF rectifier in series with the converter inductance. Due to the now nonlinear load, the above-mentioned parallel oscillation circuit driver no longer works correctly, so that the time or voltage control is taken over by a monoflop of each gate controller, which always switches off the half-bridge transistor controlled by it again after a given time and is called "monoflop control", which ensures a fixed or in particular maximum on-time of each half-bridge. A high-impedance start-up aid is also shown here, which consists of a pull-up resistor R14 and a support capacitor C9 and is again muted by D5 and D7.

[0027] In a further implementation for the drive of a bridge leg consisting of two MOS field effect transistors, i.e. the "ultra-sensitive auxiliary thyristor" DE 10 2009 042 433 B4, the regulation and drive of the half-bridge is no longer completely entrusted to the design and its dimensions. There a current measurement resistor is connected in series with each power transistor, the voltage of which triggers a small thyristor when the respective necessary level is reached at each converter clock control, which short-circuits the drive of the currently active power transistor and thus switches off this power transistor. This design is self-oscillating, but current-regulated, which is why it is called "current-regulated self-oscillator" in the following.

[0028] Here it is not specified that the value of the measurement resistor on the ground side can be easily changed from the outside. Here, in addition to the general change of the half-bridge frequency as the most important regulation intervention for the power quantity converted, the duty cycle of the bridge leg considered here is also changed here away from 50:50, which leads to a tilt control that acts in the same direction as the frequency increase and particularly well on the power quantity transmitted in the half-bridge topology with a capacitive passive branch.

[0029] The intentionally unequal duty cycle in the drive of a half-bridge will therefore be referred to as "tilt control" in the following in order to influence the power quantity transmitted by this half-bridge.

[0030] "Drive pause", EP 1 379 109 B1 describes an on-time gap control by integrating a bridge capacitor into a so-called bootstrap circuit, which can drive a bridge leg equipped with MOS field-effect transistors without using a drive transformer. However, it is limited there that the control signal for the high side is only obtained as a complement to the control signal for the low side, and there only the polarity change of the half-bridge center point from positive to ground is under control, in such a way that the actual switching point of the power transistor on the ground side is delayed compared to the above-mentioned complement until the point in time at which the operating voltage of this power transistor becomes zero or minimal.

[0031] This corresponds to the repeatedly occurring "zero voltage switching" or ZVS (Zero Voltage Switching) later on. Here, the voltage-free switching of the switching transistors or switching elements of clocked converters is known to the person skilled in the art. In the opposite polarity change direction, the required on-time is always so short that the dead time of the bootstrap itself is sufficient as on-time. The circuit there is explicitly designed for tilt control, so that the low side is switched on for at most half the cycle duration or shorter. It can therefore also be directly driven by a simple PWM-IC, which has a switching time limited to a maximum of 50% of the instantaneous cycle duration and is usually used to control simpler electronic power converters with only one power transistor.

[0032] The application EP 0 409 328 A2 describes the same with on-time gap control, which in the basic circuit has only one NPN auxiliary transistor per bridge leg for the potential side. In its basic version, only the low side gate is actively clamped to negative during the polarity change of the center point from positive to negative, while in the opposite oscillation direction only the high side gate is clamped to the center point. The incoming power transistor is therefore switched on delayed in order to achieve ZVS (= zero voltage switching). In its improved version, the high side gate is also clamped to the center point during the polarity change of the center point from positive to negative. For this, however, an NPN current mirror is used, which is coupled to the bridge capacitor at its input and whose output transistor additionally clamps the bootstrap line to the center point during the same oscillation process.

[0033] In the case of higher requirements for the flexibility of the bridge legs, the switching regulator or the PWM-IC and all its passive environment circuitry also disappear, and its power transistors are almost directly driven by an ASIC or microprocessor. There is then purely external control, which almost always only takes place at the gate of a MOS field effect transistor that can be applied in the power section. Here, the main focus is on the significantly more freely selectable turn-off point in time of the transistor that is just turned on, respectively. In addition, the faster this turn-off process takes place, the smaller the turn-off losses are. The slowing effect caused by the Miller capacitance additionally burdens the driver and power transistors. The first task of the present invention is therefore the assistance for the actual driving, which deactivates the Miller capacitance by means of a time-correct and low-ohmic short circuit of the control input of the power transistor that is just to be turned off. This control input here and in the following always consists of the control electrode, i.e. the base or gate, and the reference potential of the associated power transistor, i.e. its emitter or source.

[0034] This is sought in the solution according to US 4 481 434 A. However, the two auxiliary transistors of each drive circuit shown there always work simultaneously, not time staggered as will be shown below for the first auxiliary transistor proposed here. In addition, the thyristor formed there always shorts the control input of the associated power transistor by its anode (= emitter of the PNP auxiliary transistor) and its cathode (= emitter of the NPN auxiliary transistor), i.e. always with both auxiliary transistors, whereas the thyristor or similar thyristor formed here further below shorts the control input by means of its gate, i.e. for this always only one of its base transistors is used. In comparison with this prior art, a smaller residual voltage for the shorted control input and a shorter recovery time result therefrom.

[0035] In particular in zero voltage switching (ZVS), which is optimal for modern electronic power converters, the passive topology of the power section determines directly after such a turn-off point of time, respectively, when the next power transistor or the same power transistor should be switched on again: This dead time or occupation gap lies within the time range defined by the passive topology and is roughly multiplied by the size of the operating window in which the power converter should work. Here, the most important passive element of the power section is the so-called total trapezoidal capacitance, which is essentially parallel to the operating path of at least one power transistor. As the name suggests, such a capacitance can be formed by a plurality of individual capacitors, which do not even need to be connected directly in parallel or in series within the entire circuit.

[0036] In transversal direction to this, the current level of the load current, including the inductive part of the load, its instantaneous mass and the turn-off point of the power transistor. So far, the operating window of the inverter considered is very small (for example, only one type of lamp can be connected). Therefore, the length of the dead time gap can be designed as an Ansteuermimik, because the dead time gap is always the same length. As soon as the operating window increases, as it almost always does in LED operation, the variability and the absolute length range of the dead time gap increases. In the case of ramp control, different lengths are added within the same bridge leg depending on the direction of the transistor commutation.

[0037] A fixed dead time gap soon becomes a stranglehold. A variable dead time gap, pre-calculated by an ASIC or microprocessor, is necessary. For this, the dimensions and the instantaneous operating point of the passive power topology must be known. The tolerances of both strongly consume the freedom of the length of the dead time gap, which is already very narrow in ZVS, so that their calculation usually requires more code than all other regulation tasks of the ASIC or microprocessor together.

[0038] The above-mentioned half-bridge circuit is an inverter, which contains only one bridge leg as the active inverting circuit part or power part. Therefore, so far, the terms "inverter", "half-bridge" or "bridge leg" are synonyms. However, all of the following also apply to more complex inverters, for example, a full-bridge, a six-pulse bridge or a multi-level converter can thus be used, which contain several bridge legs, which can be said to be side by side or on top of each other or both. Such more complex inverters are indispensable components, especially for photovoltaic systems, wind turbines, large charging devices, on-board power supplies, independent network power supplies, uninterruptible power supplies and entire electric cars on rails and roads. Therefore, the focus is on the individual bridge leg together with its drive as the smallest recurring component of such more complex inverters. If only one bridge leg is used, the known half-bridge inverter arises again. SUMMARY

[0039] The technical problem addressed by the present invention is therefore to give a drive circuit for two power transistors of a bridge leg, which directly after the independently identified turn-off process of the power transistors, turns off or shorts its control input with low ohmic resistance and reliably shorts an additional positive control voltage, which is output to its control input by its assigned driver for the turn-on of the power transistor, in order to put the power transistor to be turned on next respectively in a waiting position for the length of time of the change of its operating voltage.

[0040] If the operating voltage under consideration is in a standstill, usually by switching the reverse diode of the power transistor into the waiting position, then the same drive circuit should directly initiate or execute the switching on delayed to this point itself and should also terminate the short circuit of the control input on the other bridge half. Thus, each drive circuit proposed here is connected essentially in parallel to the control input between the control input of the associated power transistor and its actual driver.

[0041] Because, in contrast to the prior art, a greater operating window is required for the possible load, which, viewed from the bridge midpoint, unfolds in both axes "range for the required active power output" and "reactive power associated with the required active power" and leads to a significantly variable on-space gap or drive pause or waiting time between the individual switching-off processes and the corresponding next switching-on processes. In addition to this, the smallest possible switching losses must always be observed, thus a priori enlarging the operating window.

[0042] The above technical problem is solved by a drive circuit for a control input between a control electrode and a reference electrode of a power transistor of a clock-controlled converter, which is coupled in parallel to the control input, wherein, for energy transmission, the power transistor is clock-controlled switched on or off by a control voltage, and wherein a waiting time is provided between the switching processes. The drive circuit has a first semiconductor chain and a second semiconductor chain, wherein on the control electrode of the control input an N region of a first PN junction of the first semiconductor chain is connected, which assists the power transistor switching off by short-circuiting the control input, and on the reference electrode a P region of a first PN junction of the second semiconductor chain is connected, which in the same way generates a waiting time for the switching on of the power transistor.

[0043] In the following, a series circuit with at least two PN junctions of opposite polarity is considered a semiconductor chain, i.e. for example P-N and subsequent N-P. Here, the PN junctions can be part of one or more components, for example transistors, diodes or thyristors, which are also integrated into one another. In the following, always reference is made to PN junctions, even if reference is made to junctions of opposite polarity, i.e. N-P junctions. Thus, the polarity is indicated respectively. For the realization of two control voltage polarities, a first and a second PN junction are formed between the control electrode and the reference electrode of the control input of each semiconductor chain, wherein the first and the second PN junction are polarized inversely. This measure realizes a highly efficient and low-cost drive circuit, which can advantageously switch the power transistor quickly and efficiently and avoids a bridge short circuit.

[0044] In a particularly preferred embodiment, the respective second PN junction is used to hold the control input at high impedance in order to achieve the turn-on of the power transistor. This advantageously ensures a fast and low-loss turn-on of the power transistor. In order to be able to simultaneously short-circuit the control input in order to comply with the latency, the N region of the second PN junction of the first semiconductor chain is connected to the reference potential and the P region of the second PN junction of the second semiconductor chain is connected to the control electrode. As a general feature, both semiconductor chains are thus connected to the control input with the same doping, respectively.

[0045] In a further particularly advantageous embodiment, a particularly heavily doped P region of the third PN junction of the first semiconductor chain or a particularly heavily doped N region of the third PN junction of the second semiconductor chain is provided in order to trigger the drive circuit. Here, the third PN junction of the first semiconductor chain shares the N region of the control electrode with the first PN junction or the third PN junction of the second semiconductor chain shares the P region of the reference electrode with the first PN junction. The P region of the third PN junction of the first semiconductor chain and the N region of the third PN junction of the second semiconductor chain are coupled to one another in the first connection point and a current for triggering is applied at this point.

[0046] By this measure, it is advantageously possible to trigger using all known measures for applying a current and thus to be particularly advantageously very simply matched to existing converter topologies. Any measure for applying a current using inductances, capacitances, etc. is suitable here.

[0047] In a further embodiment, the second PN junction of the first semiconductor chain shares a P region coupled to the P region of the first PN junction with a fourth PN junction of the first semiconductor chain, wherein the fourth PN junction actively supports the turn-off of the power transistor. The fourth PN junction has the advantage that the turn-off of the power transistor becomes faster and thus less lossy.

[0048] In a further embodiment, the second PN junction of the second semiconductor chain shares an N region with a fourth PN junction of the second semiconductor chain, said N region being coupled to the N region of the first PN junction, wherein the fourth PN junction ensures the latency even in the case of a minimum current for triggering the drive circuit. This measure offers the advantage that the latency is still correctly maintained even in the case of very small loads. A particularly long latency can thus be achieved and a converter having a drive circuit according to the application can cover a significantly larger output power range.

[0049] In a further embodiment, in order to trigger the drive circuit, a bridge capacitor and / or an activation capacitor is connected to the first connection point in order to provide the required current. This capacitive variant of the current application is provided in the case of every voltage jump occurring in clocked converters in order to generate a limited current while advantageously preventing a DC component in the trigger signal and thus inherently ensuring the correct mode of operation of the drive circuit.

[0050] In one embodiment, the first semiconductor chain and the second semiconductor chain are each formed by at least one first auxiliary transistor and a flux diode, and each semiconductor chain jointly forms at least three PN junctions of opposite polarity. Here, the first PN junction and the third PN junction each form a first auxiliary transistor, which is arranged in each semiconductor chain. Here, the first PN junction forms the collector-base junction of the first auxiliary transistor, and the third PN junction forms the emitter-base junction of the first auxiliary transistor. Here, the emitter of the first auxiliary transistor of the first semiconductor chain is a P emitter, and the emitter of the first auxiliary transistor of the second semiconductor chain is an N emitter.

[0051] In one embodiment, each second PN junction can be formed by a fast diode. In a further embodiment, each second PN junction is formed by a Schottky diode.

[0052] In a further embodiment, the second PN junction of the first semiconductor chain is formed by a base-emitter diode of a second auxiliary transistor, which has an NPN polarity, and whose collector is coupled to the base of the first auxiliary transistor or is particularly advantageously directly connected to the control electrode of the power transistor.

[0053] In a further embodiment, the second PN junction of the second semiconductor chain is formed by an emitter-base diode of a second auxiliary transistor, which has a PNP polarity, and whose collector is connected to the base of the first auxiliary transistor, i.e. to the reference potential.

[0054] Furthermore, the above-mentioned technical problem is solved by a bridge branch having a first and a second power transistor to be connected, which are connected in series, wherein the first power transistor is coupled to a reference potential and the second power transistor is coupled to a supply potential, wherein the above-mentioned first drive circuit is coupled in parallel to the control input of the first power transistor and the above-mentioned second drive circuit is coupled in parallel to the control input of the second power transistor.

[0055] By using the drive circuit according to the application in the bridge leg of a clocked converter, the advantages of the application can be fully exploited and the clocked converter likewise offers the above-mentioned advantages, such as an increased output power window, intrinsic immunity to interference and protection against bridging short circuits.

[0056] In a particularly preferred embodiment, a bridge capacitor is connected between the two first connection points of the first and second drive circuit. This ensures that the current required for driving the power transistors can advantageously flow between the drive circuits during a polarity change in order to achieve ZVS (zero voltage switching).

[0057] In a further preferred embodiment, a ladder capacitor is connected in parallel to the operating path, i.e. the collector-emitter path or the drain-source path of the second power transistor. This advantageously achieves a de-loading of the two drive circuits and likewise ensures ZVS if dimensioned correctly. The capacitance of the ladder capacitor is preferably less than three times the capacitance of the bridge capacitor here.

[0058] In a further embodiment, a lower activation capacitor is connected between the first connection point of the first drive circuit and the operating electrode of the first power transistor and an upper activation capacitor is connected between the first connection point of the second drive circuit and the operating electrode of the second power transistor, wherein the recharge currents of the two activation capacitors simultaneously trigger the drive circuits coupled thereto, respectively. Preferably, the capacitances of the two activation capacitors differ by a maximum of 10%.

[0059] In a further embodiment, the two activation capacitors have substantially the same capacitance. In a further embodiment, a further ladder capacitor is connected in parallel to one or to the other power transistor. Depending on the converter topology and design, these measures can positively influence the behavior of the converter in terms of efficiency and output power window. Here, preferably, the total capacitance of all other ladder capacitors is less than five times the capacitance of the two activation capacitors.

[0060] In a particularly preferred embodiment, the two drive circuits each contain a bead choke, advantageously either for high-frequency decoupling of the first semiconductor chain from the second semiconductor chain or for high-frequency decoupling of the first PN junction from the fourth PN junction of the first semiconductor chain. Thereby, the turn-off process of each power transistor is further accelerated.

[0061] Here, the two bead chokes of the two drive circuits are preferably coupled to one another such that two terminals, which project from a common core or from a common bead on the same end side, are coupled to the control electrodes of the power transistors, respectively. This advantageously leads to a particularly effective decoupling in order to reliably avoid bridging short circuits.

[0062] The invention likewise relates to a high-side driver having two of the above-mentioned drive circuits, which high-side driver is used to drive the second power transistor and additionally the first power transistor of the above-mentioned bridge branch, wherein terminals are provided for connecting the drive circuits in parallel with the control paths of the two power transistors, in particular two first connection points of the drive circuits are provided as terminals. This integrated assembly has all the advantages of the invention and can be realized in particular space-saving and at low cost for use in many different converter topologies with all the advantages.

[0063] The invention also relates to a clocked converter having the above-mentioned bridge branch, wherein the clocked converter has an inductance having three windings, wherein the energy converted by the converter flows through one of the windings and two further windings are each coupled with the above-mentioned drive circuits and are reverse-polarized, wherein the converter is arranged in such a way that the current flowing through the respective one of the further windings provides the energy required for switching on the power transistors and defines the switching-off point in time. This converter is characterized advantageously by high efficiency, intrinsic immunity to interference, in particular against bridging short circuits, and a large output power window at minimum outlay.

[0064] The invention is described precisely in the following and in particular the precise mode of operation within the drive circuit according to the invention and the current flow occurring there is set out. Here, the terms "reference potential", "control electrode" and "control line" include in principle the paragraph "power transistor supported by the drive circuit just considered at the time of switching on", so that this paragraph is mostly omitted. Since many regions and PN junctions are to be described, the "viewing direction" from above the control line in the direction of the connection line of the relevant reference potential applies to all other considerations.

[0065] A drive circuit for larger and stronger transistors or power transistors is proposed, which drive circuit comprises at least two smaller bipolar transistors as actively switchable components, which two bipolar transistors are connected together at a first connection point at their respective emitters. In order to make the drive circuit as universally usable as possible, this also requires, in particular, a non-short circuiting of the two control voltage polarities, each of the smaller auxiliary transistors, which are referred to as "first auxiliary transistors", must be expanded to the above-mentioned semiconductor chain. In the first semiconductor chain, the smaller transistor or first auxiliary transistor is a PNP transistor, while in the second semiconductor chain it is an NPN transistor. The group consisting of these two first auxiliary transistors is also referred to as "complementary auxiliary transistors" in the following.

[0066] Due to the high demands on the speed and the delay-free nature of the drive circuit, two first auxiliary transistors are operated in the basic circuit. The first auxiliary transistor in the first semiconductor chain completes the short circuit of the control input of the power transistor after the turn-off process of the power transistor, and the first auxiliary transistor in the second semiconductor chain completes the short circuit of the control input time staggered before the next turn-on process of the power transistor, in order to generate a latency time for the power transistor during which the power transistor remains in the standby position. For this purpose, the base of the first auxiliary transistor, i.e. the smaller PNP transistor, of the first semiconductor chain is coupled to the control electrode. In contrast, the base of the first auxiliary transistor, i.e. the smaller NPN transistor, of the second semiconductor chain is at a reference potential.

[0067] The actual drivers of the two power transistors of the bridge branch, which provide the auxiliary energy required for the clock control of these transistors as well as the switching-on point in time and in particular the switching-off point in time, remain unaffected by the drive circuit, with the sole exception that the drivers must be short-circuit-proof, i.e. must have a series impedance. As soon as the above-mentioned drive transformer is used as often as in the bridge branch, a pure alternating voltage, i.e. also a negative drive signal, occurs in principle on the winding of the drive transformer which is provided for the control input of the power transistor. The drive circuit is not allowed to interfere with this negative drive signal and does not do so, because the drive on the opposite arranged potential side or half of the bridge branch would be distorted or even made impossible, respectively, by this.

[0068] The collector of the first auxiliary transistor with N emitter is thus completed to the second semiconductor chain by a second PN junction, the P region of which is coupled to the control electrode, and the N region of which is connected to the likewise doped collector of the first auxiliary transistor. This second PN junction can be a flux diode, i.e. for example a fast diode or a Schottky diode, the cathode of which is connected to the collector of the first auxiliary transistor, and the anode of which is connected to the control line.

[0069] It is therefore necessary that a semiconductor chain in parallel to the control input of the connected power transistor, which consists at least of the base-collector diode of the first auxiliary transistor and the above-mentioned flux diode. Here, it is characteristic that the two regions have a p-type doping, respectively, and that the semiconductor components of the second semiconductor chain, which are observed in more detail above, are connected to the control input of the connected power transistor with said two regions.

[0070] To avoid a direct short circuit of the control input, or to achieve a blocking capability for the drive voltage in both directions, at least two PN junctions are required, furthermore, at least two of which must be oriented in reverse. In the observation direction, this is the second PN junction between the anode and the cathode of the flux diode of the second semiconductor chain, and the "NP junction" of the collector to the base of the first auxiliary transistor, which is in series with it, i.e. its first PN junction.

[0071] A third feature of this semiconductor chain is that it is triggered or activated by a further semiconductor region, which is not located between the two regions with which the semiconductor chain is coupled to the control input, but rather can be said to be located outside of the two regions. The collector of the first auxiliary transistor of the second semiconductor chain is located between the control line and the reference potential, and the cathode of its flux diode is connected to this collector. Its base is located at the reference potential. Its emitter is opposite the collector, and thus located "outside" of the control input. In this way, continuing the above observation direction, a third PN junction is created between the base and the emitter of the first auxiliary transistor as a third doping boundary below the reference potential. As will also be shown later, the second semiconductor chain ensures the flux balance in the drive transformer and the maintenance of the latency time.

[0072] A second PN junction also exists between the collector of the first auxiliary transistor of the first semiconductor chain and the reference potential for the same reason, which can be composed of a fast diode or a Schottky diode, the cathode of which is at the respective reference potential and the anode of which is connected to the collector of the first auxiliary transistor: following the "NP junction" on the n-doped base of the first auxiliary transistor, which is connected to the control line, is the first PN junction to the collector of the first auxiliary transistor, which is connected to the anode of its flux diode. This is followed by the second PN junction to the cathode of this flux diode, which is located at the reference potential. The two connection regions of this first semiconductor chain to the control input are n-doped, and the collector and the anode are p-doped between them, respectively.

[0073] Because the emitter is also arranged opposite the collector of the first auxiliary transistor here, the emitter, which is also responsible for activating the first semiconductor chain, can also be found outside of the control input, here especially above the control line. The third PN junction separates this emitter from the base of the first auxiliary transistor, which is connected to the control line. As will also be shown later, the first semiconductor chain is responsible for conducting a negative drive current when switching off the associated power transistor, especially in the case of flux driving, and for compensating for the charging current difference between the activation capacitance and the trapezoidal capacitance.

[0074] The second PN junction of the first semiconductor chain can also form a base-emitter path of a further NPN auxiliary transistor, the collector of which (delimited from its base by a fourth PN junction) is connected to the control line, and which is also referred to hereinafter as the "second auxiliary transistor of the first semiconductor chain". In this way, the first semiconductor chain obtains a branch in its P region between the control line and the reference potential, which P region consists of the collector of the first auxiliary transistor and the base of the second auxiliary transistor, and the collector of the second auxiliary transistor is there docked via a further NP junction (again viewed from the control line in the direction of the reference potential), the fourth PN junction. The former flux diode anode becomes its emitter.

[0075] The orientation of the region layers and the doping boundaries between them, which have been determined above for the first semiconductor chain, is thereby not changed. Since the new collector is again n-doped, only the N region of the first semiconductor chain is in contact with the control input, now in total three such regions, of which two are connected to the control line. This, together with the otherwise unconnected collector-base connection between the two auxiliary transistors of the same semiconductor chain, indicates a thyristor-like structure, which is produced by the first and second auxiliary transistors. As will also be shown later, only the first semiconductor chain branched in this way can actively assist the power transistor at its turn-off, for example by the second auxiliary transistor conducting a Miller current to the reference potential.

[0076] Finally, but equally importantly, the second PN junction or flux diode of the second semiconductor chain can be implemented as an emitter-base path of a further PNP auxiliary transistor, the collector of which (again delimited from its base by a fourth PN junction) is connected to the reference potential, and which is referred to hereinafter as the "second auxiliary transistor of the second semiconductor chain".

[0077] The second semiconductor chain thus also obtains a branch, which now lies in an N region between the control line and the reference potential, which N region again consists of the collector of the first auxiliary transistor and the base of the second auxiliary transistor. The newly added collector of the second auxiliary transistor is docked to the N region via the fourth PN junction, which collector, with its p-doping, as a second contact, is connected here to the reference potential. The anode of the flux diode just replaced becomes the emitter of this second auxiliary transistor of the second semiconductor chain. As will also be shown later, only this branch of the second semiconductor chain can guarantee the retention of the latency time up to very small bridging output powers.

[0078] The reference potential double contact through the two p-doped regions "base of the first auxiliary transistor" and "collector of the second auxiliary transistor" of the second semiconductor chain and the otherwise unconnected n-doped collector-base connection between the two auxiliary transistors indicates that the two auxiliary transistors can be merged into a small thyristor, which is referred to as "second auxiliary thyristor", which is connected in parallel between its anode and its gate to the control input of the power transistor and whose cathode forms together with the emitter of the first auxiliary transistor of the first semiconductor chain the first connection point between two differently doped emitters.

[0079] This second auxiliary thyristor corresponds to a monolithically continuous second semiconductor chain from the control line to the activation. Seen from the usual viewing direction, its p-doped anode as the uppermost region is the emitter of the second auxiliary transistor, the subsequent second PN junction delimits its base, which is at the same time the collector of the first auxiliary transistor. This n-doped region, the drift region of the second auxiliary thyristor, is located between the control line and the reference potential, is defined below by the first PN junction to the p-doped base of the first auxiliary transistor and by the fourth PN junction to the likewise p-doped collector of the second auxiliary transistor, wherein the first and fourth PN junctions merge in the thyristor. Both are at the reference potential and likewise merge to form the gate of the second auxiliary thyristor. The third PN junction as the third doped boundary then delimits the lowermost of the semiconductor chain, the n-doped emitter of the first auxiliary transistor upwards. On this emitter, which is at the same time the cathode of the second auxiliary thyristor, the activation of this second semiconductor chain, although classically by a positive gate voltage, is actually a negative cathode voltage. The control input is short-circuited by the anode-gate path of the second auxiliary thyristor, whereby a lower residual voltage on the control input and a shorter extinction time of the second auxiliary thyristor are possible, since only its PNP transistor is used.

[0080] The double basic circuit is formed by two first auxiliary transistors, whose complementary transistors are connected together to their emitters. The circuit diagram of the complete drive circuit according to the transistor resolution has three or four rectifying diode symbols, including all occurring basic symbols, which always end in the flow direction in the reference potential of the power transistor to which they are connected and in the case of four such symbols begin on their control line. Characteristic is also that each connected control input according to the invention is commutated by the two basic diodes of the respective first auxiliary transistor, which are connected in series via the first connection point. The node between their emitters is connected with a current source, which applies its current to the node, thereby activating the drive circuit. The current source can consist of at least one capacitor, whereby a pulsed current flows exactly when the drive circuit is activated and should be activated and with its length of time.

[0081] The larger and stronger transistors or power transistors are particularly advantageously N-channel MOS field-effect transistors or NPN bipolar transistors based on a base, for example silicon, silicon carbide or gallium nitride, respectively. In the bipolar transistor, a flyback diode can be integrated, or an external flyback diode can be connected in antiparallel to the operating path between the collector, which is the operating electrode, and the emitter, which is the reference potential, of the bridge branch. In the MOS field-effect transistor, there is already a flyback diode, the anode of which is at the reference potential and the cathode of which is at the operating electrode thereof, i.e. in antiparallel to the operating path between the drain and the source thereof.

[0082] This component or component group is preferably grouped in a bridge circuit in series with a so-called inverter, whereby in the following a single bridge leg is considered exemplarily in more detail. It consists of two power transistors connected in series. This series circuit is supplied by a DC voltage source. Here, the reference potential of the first power transistor is at ground, and the operating electrode of the second power transistor is connected with positive. The reference potential of this second power transistor is connected with the operating electrode of the first power transistor, and this connection point between the operating paths of the two power transistors forms the midpoint of the bridge leg. The first power transistor is also referred to as "Low-Side-Schalter" or "Low-Side-Transistor" in the following, the second power transistor is also referred to as "High-Side-Schalter" or "High-Side-Transistor", and each power transistor is also referred to as "bridge transistor". Each of the drive circuits disclosed here is coupled in parallel with the control inputs of the two bridge transistors of the bridge leg considered, i.e. either between base and emitter or between gate and source.

[0083] A load is connected at the midpoint at which the high-frequency alternating voltage is to be tapped, which load is advantageously tuned inductance-ohmic and is connected with at least one pole of the same DC voltage via at least one DC decoupling capacitor, which also energizes the bridge leg. The load and the at least one DC decoupling capacitor are not shown in principle. If the load should exhibit a resonant behavior, the at least one DC decoupling capacitor can be intentionally reduced in order to form an intrinsic frequency together with the inductive part of the load.

[0084] In particular, but also already in the case of an inductance-ohmic load, the above-mentioned bridge leg can particularly advantageously be constructed self-oscillating. For this purpose, the primary winding or the measuring winding of a drive transformer is connected with at least one turn, for example, to the bridge midpoint in series with the load. The drive transformer also comprises at least two secondary windings or drive windings for forming a control voltage or a control current for the control inputs of the two power transistors participating in the same bridge leg.

[0085] In the case of a flux driver and two NPN bipolar transistors as bridge transistors, the drive transformer is magnetically fixed coupled. At least one primary winding or measurement winding of the drive transformer, which has at least one turn, is generally located in series in a branch of the circuit, at least a portion of the load current flowing in this branch at least per switching phase. For the purpose of forming an inverter for push-pull operation, the orientation of the secondary windings of the drive transformer is oppositely oriented. Each drive winding is oriented with respect to the measurement winding such that there is positive feedback at the beginning of the on phase of the power transistor it drives, respectively. The portion of the load current or coil voltage corresponding to the turns ratio is applied in the on direction to the control input of the power transistor that should be activated next. The detailed structure of the drive transformer is not an element of the invention.

[0086] In the case of bipolar transistors as bridge transistors, the turns ratio between the measurement winding and the drive winding can be from about half to the full saturation current gain of the bridge transistors. In the two drive circuits of the bridge branch under consideration, there are in particular no further series components, such as for example resistors. Instead, a magnetization current is used for the respective switching off, which is integrated in the drive transformer in each on phase of one of the power transistors and is subtracted from the load current as base current before being transmitted. Thus, in addition to other parameters, the inherent magnetization of the drive transformer determines the operating frequency of the flux-driven bridge branch.

[0087] At the end of each on phase, there are two inductively imposed currents that are independent of one another. More predominantly is the current of the load, but almost less predominantly is the current of the current inherent magnetization of the drive transformer, which is connected in series with the load or a further current sink, which represents the load current at least partially or to some extent. This arrangement is theoretically forbidden, it works because the drive transformer has a reserve for the difference current its drive winding. Thus, in each switching process, the control input of the two bipolar power transistors receives the difference of these two currents, which is deliberately "wrong" even during the on phase: If the collector current is constant or even slightly increasing, the base current has already decreased again or has become negative due to the inherent magnetization. Because this serves to switch off the currently activated power transistor. But after that, where does this current difference go?

[0088] The winding orientation on the drive transformer allows only one possibility: into the base of the power transistor to be switched on at the moment, without any shoot-through gap. Each bridge leg with a flux driver is in principle hard switched, although from the phase of the load current, a very advantageous resonant ZVS operation can usually be achieved. This requires a waiting position for the next power transistor to be switched on, respectively, after the corresponding previous power transistor has been switched off, which lasts as long as the time required for the natural polarity change of the bridge midpoint, i.e. the so-called total trapezoidal capacitance of the inductive part of the load current must be recharged until its voltage is either zero or the instantaneous DC supply voltage "positive". The duration of this waiting position corresponds to the same waiting time as the known shoot-through gap or dead time.

[0089] At very high midpoint frequencies of approx. 1 MHz, this total trapezoidal capacitance is formed only by the parasitic output capacitance parallel to the working path of the bridge transistors, the required load inductance is produced by the wiring, i.e. by the circuit layout. At the desired 20 kHz to 50 kHz frequencies at the bridge output, both must be structurally increased: a discrete trapezoidal capacitor connected in parallel to the working path of at least one of the bridge transistors, at least one discrete inductance connected in series with the load.

[0090] Osram for 12 V halogen lamps designed exclusively with a flux driver The device has a very narrow load window. Therefore, there, a simple buffer is tuned in parallel or in series with at least one driver to the tiny trapezoidal capacitance. The required inductance is produced by the leakage inductance of the safety isolation transformer, which is mandatory for SELV approved devices. This simple solution no longer meets the requirements of the Osram device for LED operation. Due to the now mandatory HF rectifier and especially due to the LED load itself (both not shown), the load is coupled to the bridge leg nonlinearly and more strongly reactive in order to mitigate this effect. For this a large allowed operating range is required, which is limited by the six lines "maximum output voltage, maximum output power, maximum output current, minimum output voltage, minimum output power and minimum output current" in the IU diagram and results in an operating window that recently even reaches the Y axis. More and more the "minimum output current = 0" is required, the so-called deep dimming.

[0091] Since the base-emitter flux voltage as the maximum voltage at the control input of the bipolar transistor as power transistor is low, it must be reliably avoided that the control voltage of the actually switched-off bipolar transistor ever reaches this flux voltage level. Otherwise a short-time forming bridge short-circuit would immediately lead to a complete failure of the associated bridge leg by a strongly increased radio interference and losses. Usually, this short-time level overshoot is the result of a leakage inductance in the drive transformer combined with the partial dynamic capacitance of the currently switched-off power transistor, for example its Miller capacitance. Therefore, it is not only important that both mutually independent applied currents find a path for continued flow after each of the switched-off bridge transistors, but also where exactly these paths are located: Before and after the considered switching-off process, both currents should use the same winding in the drive transformer with the same amplitude and direction as at the end of the preceding on-phase, since each current jump in one of the windings excites high-frequency oscillations in the control voltage due to said leakage inductance, which in the worst case leads to such a short-time bridge short-circuit.

[0092] Due to the ladder capacitances parallel to the operating paths of the at least one bridge transistor, during such a polarity change process the load current continues to flow in the measurement winding as described before. And the negative base current should be able to continue to flow in the exact same drive winding by which the just switched-off bridge transistor is driven. For this purpose it is provided that an additional ladder capacitor, for example a bridge capacitor, is connected between the control inputs of the bridge transistors. If not only the control voltage of the currently switched-off bridge transistor, but also the control voltage of the not yet switched-on bridge transistor is controlled, an additional advantage arises: As will also be shown later, in the control winding for the power transistor to be switched on a current is prepared which, although it is usually not matched in its amplitude, is at least matched in its direction to the current which continues to flow in the same control winding after the upcoming switching-on process.

[0093] For this purpose the drive circuit disclosed here enables, almost independently of the type of bridge transistor and the general driving method, to short-circuit the two control inputs of the power transistor of the same bridge leg just during the waiting time in which no bridge transistor should be on, or in which the potential of the midpoint changes. Finally, the low-loss steering of the two control voltages during the polarity change phase, by which said transient bridge short-circuit is avoided, significantly increases the oscillatory behavior, in particular when said steering ends immediately at the beginning of the new on-phase, in order to free the two applied currents again.

[0094] The above-mentioned more detailed task, i.e. the two applied currents occurring in each bridge leg with flux driver first independently of each other and secondly even after the turn-off process of the bridge transistor in as unchanged a path as possible through the drive transformer, is solved by inserting a PN junction at the two ends of the bridge capacitor, which is the starting point of the drive circuit of the two power transistors of the same bridge leg. Due to its "transverse to the load current" position, the "bridge capacitor" thus formed always forms part of the total ladder capacitor connected in parallel to the output of the bridge leg considered.

[0095] As will also be shown below, the bridge capacitor can also be replaced by two active capacitors of essentially the same size, in particular when the power of the bridge leg considered is relatively large, for example more than 300 W. In addition to the capacitances of these two active capacitors, which together are called "active capacitance", the capacitance acting at the bridge midpoint, which although it is formed at the midpoint and in particular slows down the voltage curve of these polarity change processes, but here does not activate the drive circuit, i.e. the capacitance of the further ladder capacitor, can add up to five times the value of the active capacitance. In the case of a smaller total ladder capacitance allowed, i.e. in the case of a smaller total sum of active and ladder capacitance allowed, the former has priority, so the ladder capacitance can also be zero.

[0096] Even in the case of a bridge capacitor with its bridge capacitor, which is functionally equivalent to the above-mentioned active capacitor, there can be a further ladder capacitor in the total circuit containing the bridge leg considered, the sum of which can amount to three times the capacitance of the bridge capacitor. It is particularly advantageous that a single such ladder capacitor is connected in parallel to the operating path of the positive bridge transistor, the capacitance of which is essentially equal to the bridge capacitor. Here, too, the bridge capacitor is preferred, i.e. the further ladder capacitor can also be zero.

[0097] The drive circuit consisting of the PN junction only interfaces with each participating drive circuit and thus does not change the drive circuit itself. The following considers the turn-off process of an NPN bipolar power transistor as a high-side switch. Immediately after, i.e. in the polarity change process of the voltage at the bridge midpoint from positive to ground, the bridge capacitor must be discharged due to its ladder position. As a result, a current flows out of its high-side end, which of course has to flow downstream at its low-side end.

[0098] If the bridge capacitor is connected directly between the two bridge transistor base terminals and their associated drive windings, it is possible to accomplish this task on the high-voltage side: The discharge current of the bridge capacitor after the turn-off process can correspond exactly to the negative base current before the same turn-off process. But on the low side, the same current would suddenly be added from the drive winding there in the positive direction there. Not only is the flux balance in the drive transformer impaired, but the current jump on the low side also produces uncontrollable oscillations.

[0099] If the bridge capacitor is halved, although the flux balance can be restored again, at the same time two current jumps in opposite directions occur, which can again destroy this restoration due to the already mentioned leakage inductance in the drive transformer.

[0100] The PN junction directly from the ground to the low-side terminal of the bridge capacitor, which makes the lower drive winding free of current flow during the just described polarity reversal, and the PN junction already described above between the high-side terminal of the bridge capacitor and the control electrode of the high-side transistor, which initially activates the drive circuit, provide a remedy.

[0101] In order for the turn-off of the low-side switch to function equally well, the bridge capacitor must be charged again thereafter, which requires a PN junction from the low-side terminal of the bridge capacitor to the control electrode of the low-side switch, which further directs a negative drive current in the lower drive winding, and as a "current source", a PN junction from the half-bridge midpoint to the high-side terminal of the bridge capacitor, which prevents the effects of the impaired flux balance in the drive transformer even when the polarity reversal is to the reverse direction.

[0102] This part of the drive circuit described here forms the basic structure of the invention, namely the series circuit of two rectifying PN junctions, which starts at the reference potential of the power transistor to be connected and ends at its control electrode in the flow direction, wherein an activation capacitor and / or a bridge capacitor, for example the capacitance of an activation capacitor or especially the capacitance of a bridge capacitor, respectively, acts between the two PN junctions as part of the total ladder capacitance. Thus, the control voltage of all bipolar power transistors connected according to the invention, although it can and often does become negative, is here limited to the flux voltage of the 2 PN junctions.

[0103] The current, for example present in the bridge capacitor, does not necessarily correspond to the negative base current directly before the considered turn-off and polarity reversal process, but is in fact always different from it due to the voltage shape of the polarity reversal process occurring at the half-bridge midpoint in ZVS, in particular in its course. Since the voltage shape, which reflects the load current, is always curved, the negative drive current is approximately fixed at its starting value at the time of turn-off during the polarity reversal process.

[0104] If the current flowing through the bridge capacitor is greater than the current required by the flux in the drive transformer visible on the negative drive current, which is usually the case after the high-side transistor is switched off at the beginning of the polarity change phase, additional paths are required for these excess currents on the high side. This is achieved by shifting the PN junction from the high side of the bridge capacitor to the control electrode of the high-side transistor in the direction of flow, which has already been known as the third PN junction, into the emitter-base diode of a PNP auxiliary transistor, the collector of which is associated with the reference potential of the high-side transistor and thus with the half-bridge midpoint, from which the "too much" current is just tapped.

[0105] However, since its parasitic collector-base diode, which has already been introduced above as the first PN junction, shorts the positive driver of the low-side transistor by coupling it back down through the drive transformer, the second PN junction must be inserted into the new link in the opposite polarity to the first (as seen in the direction of flow of the positive collector current for the auxiliary transistor, i.e. for example the flux diode), the anode of which is connected to the collector of the PNP auxiliary transistor or first auxiliary transistor, the cathode of which is connected to the reference potential of the high-side transistor. In this way, the first semiconductor chain is formed. The PNP auxiliary transistor is its first auxiliary transistor, the PN junction between its base and its emitter being its third PN junction.

[0106] Because the load current decreases significantly faster during each polarity change process than the approximately constant inherent magnetization current of the drive transformer due to its significantly higher ohmic component, the voltage slope of each polarity change process also decreases during its run, which produces the curvature described above, and thus the current through the bridge capacitor also decreases. A point is soon reached in which the recharge current through the bridge capacitor and the current magnitude applied in its high-side drive winding by the drive transformer are identical. Then, the first auxiliary transistor switches off.

[0107] Immediately after that, the recharge current becomes too small, i.e. smaller than the current actually applied in the high-side drive winding. For this case, the effect of the reverse coupling already described above is fully utilized, i.e. the "becoming too small negative" current in the high-side drive winding establishes a corresponding positive current in the low-side drive winding. For this purpose, a new path must be opened next to the control input of the low-side transistor in order to put it in standby position.

[0108] To this end, the previous low-side PN junction is extended in the flow direction between the ground and the lower end of the bridge capacitor as a base-emitter diode of an NPN auxiliary transistor, which takes over such an excess or premature positive current in the low-side drive winding. Its collector is connected to the control electrode of the low-side transistor, its base to ground. Here, the parasitic base-collector diode also shorts the positive driver of the high voltage side via its first PN junction, so here the second PN junction must be connected in series with the just newly added concatenation in the opposite orientation to the first (in the flow direction for the positive collector current of the NPN auxiliary transistor, i.e. for example the flux diode). Its anode is now connected to the low-side control electrode, however, and its cathode to the collector of the auxiliary NPN transistor. In this way, a second semiconductor chain is formed. The NPN auxiliary transistor is its first auxiliary transistor, the PN junction between its base and emitter is its third PN junction.

[0109] The advantage of this arrangement is that the current flow in the low-side drive winding, which is already prepared at the end of the polarity change phase, also occurs in the subsequent switching phase (turn-on of the low-side transistor with base-collector current in parallel to the current through its freewheeling diode or reverse diode), albeit with a higher amplitude, but in the same direction. This reduces the upcoming current jump and thus the amplitude of the subsequently excited oscillation. The task of the continuity of all currents described above, which flow through the drive transformer, even across the switching phase boundaries, is at least partially fulfilled in the positions where a current change for driving the bridge leg is actually unavoidable: in the case of a changeover of the drive current from the high-side drive winding to the low-side drive winding and vice versa.

[0110] Directly following the turn-off process of the high-side transistor or the second power transistor, the first semiconductor chain of the second drive circuit (because this first semiconductor chain is also located above in the bridge leg) becomes active, which at least low-ohmically switches off the control input of the second power transistor. Before the low-side transistor or the first power transistor is turned on, the second semiconductor chain of the first (because located below) drive circuit is usually active in order to put the first power transistor into a standby position by shorting its control input if necessary.

[0111] It goes without saying that, in order to design the drive circuit for the turn-off process of the first power transistor symmetrically, identical means must exist on the ends of the bridge capacitor and between the reference potential and the control electrode of the two bridge transistors respectively, i.e. all means described so far for the high side additionally exist on the low side and vice versa. This applies in particular to the extension of the PN junction to the auxiliary transistor and to the flux diode connected to its collector. Directly after turning off the first power transistor, the first semiconductor chain of the first drive circuit becomes active and, before turning on the second power transistor, usually the second semiconductor chain of the second drive circuit becomes active.

[0112] Thereby the same circuit topology of the first connection point between the two emitters is obtained on all potentials of the possible drive, the belonging complementary auxiliary transistors of the two emitters each operating in a basic circuit. Furthermore new is the connection of the first connection point with a capacitor which just leads the current in the size in which it exists when the drive circuit according to the invention has to be active and also approximately. The basic circuit is sufficient therefore and the form of the capacitor is just called "activation capacitor" therefore. The combination of two such capacitors results in a "bridge capacitor". The advantage of these basic circuits is their enormous speed which exists even when the auxiliary transistors are saturated, because the emitter current is limited by the recharge of the bridge capacitor or activation capacitor respectively.

[0113] The mandatory flux diode or other PN junction on all collectors of the respective first auxiliary transistor of each semiconductor chain and drive circuit allows slightly negative drive voltages and thus saves the drive of the respective oppositely arranged bridge half in a bridge branch with flux driver or usually with fixedly coupled drive transformer.

[0114] The two mentioned flux diodes can also be implemented as base-emitter paths of the respective second auxiliary transistor, the collector of the second auxiliary transistor being connected at the potential of the considered drive circuit respectively, at which the original flux diode cannot be connected respectively. Thereby, as a new characteristic circuit diagram a chain consisting of four rectifying diode symbols is obtained, including all existing base-emitter symbols, which, seen in the flow direction, starts at the control electrode of a power transistor and ends at the reference potential of the same power transistor. In the middle of this chain, i.e. at the first connection point, a bridge capacitor is exemplarily connected. The PN junction with its anode starting at the control electrode can be absent.

[0115] In the case of a corresponding switching on of the bipolar power transistor, in particular when the reverse diode is connected in parallel to the bipolar power transistor or is integrated in the bipolar power transistor, the base-emitter voltage of the bipolar power transistor is clamped to a value close to zero. The reason for this is that, when the total current is greater than the magnetization current stored in the drive transformer, the overall negative total current is conducted in parallel through the base-collector diode of the power transistor and the reverse diode. Here, the distribution between them is precisely defined by the load inductance and the flux in the drive transformer, as long as no protection diodes are installed (see below). Due to its reverse diode, the emitter of the power transistor is one flux voltage higher than its collector, and due to its base-collector current, its base is also one flux voltage higher than its collector.

[0116] Thus, in the case of identical flux voltages on both paths, the base-emitter voltage of the power transistor that has just become switched on is still precisely zero until the point at which the negative collector current is smaller in terms of absolute value than the positive base current, i.e. when the reverse diode becomes currentless and its regular base-emitter diode is switched on. Then, the base-emitter voltage reaches its regular positive value, and here the collector-emitter voltage is correspondingly increased to a value slightly below zero. Only with the change in the sign of the collector current does the power transistor finally switch on, wherein, as it were, the power transistor falls from below into its saturation voltage, which corresponds to the ideal state.

[0117] In reality, however, a negative switching-on voltage peak occurs, because the path through the reverse diode, in particular, is not impedanceless. However, because the impedance of each drive circuit is still much higher, said voltage peak occurs almost unattenuated in the control voltage, which is even irrelevant for the currently switched-on bridge transistor. However, when this control voltage is transmitted to the other potential side of the bridge leg (in the following also referred to as bridge half), with the opposite polarity, in particular in the case of flux driving with a drive transformer without saturation, a positive control voltage peak occurs at the control input of the bridge transistor that was switched off shortly before, exactly when the other bridge transistor is switched on. This effect, like the already mentioned effect of the leakage inductance of the drive transformer, can lead to a dreaded short-time bridge short circuit, especially because at exactly this time all control voltages and control currents are released.

[0118] In the bridge branch with flux driver, it has proven suitable to string a bead choke in series in at least one of the drive circuits. The bead choke, i.e. a small inductance that has been brought into saturation at low direct current, is either located between the base of the power transistor and its drive circuit or in a section of the control line of the drive circuit according to the invention, for example between the first semiconductor chain and the second semiconductor chain, respectively. There, for example, a bead ferrite ring core with a diameter of 3.5 mm, a length of 5 mm, a bore of 1.5 mm is passed through the control line

[0119] It is also advantageous here that the drive circuit is mirrored in such a way that the second semiconductor chain is connected directly to the base of the bridge transistor and the first semiconductor chain is decoupled from it by means of a bead choke. If a second auxiliary transistor with NPN polarity is used in the first semiconductor chain (whose function is described more precisely below than above), its collector should most advantageously be connected directly between the control electrode of the associated bridge transistor and the bead choke.

[0120] It is particularly advantageous if the two bead chokes are coupled by means of a double-hole magnetic core or by means of the two control lines of the same bridge branch that pass through the same ferrite bead, so that a common-mode choke is obtained from the perspective of the drive transformer towards the two control electrodes of the bridge transistor. For this purpose, the ends of the control lines that are to be connected to the control electrodes of the bridge transistor must protrude from the same end face of the double-hole magnetic core or the common ferrite bead. This perfectly avoids the bridging short circuit: at all four jumps, the base voltage has a clear negative peak without any subsequent oscillations.

[0121] If the PN junction in series with the collector of the first auxiliary transistor in the first semiconductor chain is implemented as a base-emitter path of a second auxiliary transistor with NPN polarity as already mentioned above, the collector of the second auxiliary transistor is connected to the control electrode of the power transistor to which it is connected, a small thyristor is generated by the two auxiliary transistors. Especially in the case of small currents stored in the drive transformer at the turn-off point, the first auxiliary transistor can become high-impedance despite the large recharge current of the bridge capacitor on the emitter of the first auxiliary transistor, because its base current (exactly the current stored above) is too small. However, if the collector current of the first auxiliary transistor is amplified by the second auxiliary transistor of the same first semiconductor chain, the base of the first auxiliary transistor is unloaded and the first auxiliary transistor can then be switched on completely as required.

[0122] In the above case without a second auxiliary transistor, the Miller current from the base of the just switched-off power transistor can even cause, due to its base current inversion, the first auxiliary transistor to switch off completely again. Conversely, if this Miller current has a very low impedance path through the collector of the second auxiliary transistor, this Miller current is guided as required to the reference electrode of the power transistor to support and accelerate its switching-off process, thereby not disturbing the drive circuit in its actual on-space gap control operation any more.

[0123] Another possible case, in which the recharge current through the bridge capacitor is significantly smaller than the stored drive current directly after the switching-off process, leads without a second auxiliary transistor to the fact that no voltage guidance of the control input of the just switched-off power transistor takes place, because the first auxiliary transistor is not switched on at all due to the lack of collector current. This switching-off process is thereby very sensitive to the current through the Miller capacitor of the just switched-off power transistor.

[0124] With the second auxiliary transistor, the first auxiliary transistor is unloaded, as described above, and therefore, through its possibly small positive collector current, the second auxiliary transistor is switched on inversely through its base-collector diode, which again leads to a low-impedance path for the Miller current. In this case, the difference made up of the stored drive current and the too small recharge current for this immediately establishes itself in the drive winding on the further (i.e. oppositely arranged) bridge half.

[0125] If there is a series bead choke in the drive circuit under consideration, there is no or a coupling for control on the oppositely arranged potential side of the bridge branch under consideration, in short, for driving the further bridge half, the bead choke can be passed through the control line at any position. Particularly advantageously, the collector of the second auxiliary transistor of the first semiconductor chain is connected directly with the base of the power transistor connected thereto, whereas the base of the first auxiliary transistor is shielded by the bead choke. It can thus become an integral part of the small thyristor, which increases its trigger and extinction speed.

[0126] However, it is not possible to use a real thyristor as a monolithic first semiconductor chain instead of the first and second auxiliary transistors, because the connections required by the above-mentioned bead choke are not exposed in the n-doped region between its "upper base" and "lower collector". Moreover, it is precisely there, i.e. by current consumption at the "upper gate", that this small thyristor is triggered each time, which is not allowed for the overall assembly. At the same time, this unexposed "upper gate" is always at least part of the control line, the voltage of which should be guided by the driving circuit under consideration. Therefore, in the following, there is never any mention of a "first auxiliary thyristor", but always only of a "small thyristor". Thus, as already mentioned above, if a second auxiliary thyristor is mentioned, it can actually be constructed as such a small thyristor.

[0127] It has already been mentioned at the beginning of this general description that the fact is important, especially for flux driving, that the driving circuit presented here is ready for driving the current path after the turn-off process and before the power transistor whose work the control input is waiting for is switched on on the bridge half arranged opposite to the turn-off process, the time the driving current passes in the waiting position at the control input of the transistor is exactly as long as the time the transistor has to wait for. This requires the duty gap control in the case of ZVS as always the basis. The current path is based on the basic circuit, here the basic circuit of the first auxiliary transistor of the second semiconductor chain, which can only conduct the current flowing in the belonging emitter, here in the bridge capacitor or activation capacitor. For this purpose, the base current in the auxiliary transistor must additionally be positive in order to always keep it switched on. When this base current is exhausted, the point equation "current in the lower drive winding" = "current through the bridge capacitor" = "current in the upper drive winding" = "instantaneous load current" applies. Thereafter, the current path is high-impedance and the waiting transistor is switched on.

[0128] However, the same point equation also shows that a load current less than half the magnetization current currently stored in the driving transformer invalidates the duty gap control during a polarity change process. The waiting power transistor is still switched on, even if current is still flowing through the bridge capacitor or activation capacitor, i.e. its operating voltage has not yet become zero, although this is possible. This is particularly disturbing in the case of small power of the bridge branch, for example important strong dimming operation or for "minimum load current = 0". This problem can be solved by amplifying the current path considered here.

[0129] To this end, in the second semiconductor chain, the second PN junction between the control electrode of the power transistor that is waiting and the collector of the associated first auxiliary transistor can be expanded to the emitter-base path of a second auxiliary transistor, which now has the PNP polarity and has its collector connected to the respective reference potential. Its base is connected to the collector of the first auxiliary transistor, whereby the amplification is obtained. The base of the first auxiliary transistor and the collector of this new second auxiliary transistor, both P layers, are at the same potential, namely at the respective reference potential, and form the gate of a further small thyristor, which can thus be implemented here and contrary to above actually as a concentrated small-signal thyristor and is therefore also called "second auxiliary thyristor". Because it triggers each time by means of a bridge capacitor or activation capacitor, which leads its cathode potential below its gate potential, i.e. ultimately classically by means of a positive gate voltage.

[0130] It is characterized in that its path between anode and gate is coupled in parallel to the control input to which the power transistor is connected. It thus uses only one of its two transistors to short-circuit the control input, whereby a minimum residual voltage at this control input and a minimum recovery time after such a short-circuit is produced.

[0131] For the sake of clarity, only some possible variants are shown as examples of NPN bipolar transistors or N-channel MOS field effect transistors as drive circuits for the bridge transistors and are explained in more detail. Any combination of the details of these figures and obvious modifications are likewise included in the present disclosure and its description. In particular, the person skilled in the art can thus derive by mirroring, polarity conversion, inversion or forming a complement variants for the same effect of PNP bipolar transistors or P-channel MOS field effect transistors as drive circuits for the power transistors.

[0132] Furthermore, it makes sense to use two different drive circuits for the same bridge leg at the same time, which, in addition to, for reasons of clarity, only two drive circuits of the same type are always shown separately in the entire circuit. This applies in particular to the so-called "complementary bridge leg" space-occupying gap-optimized circuit, in which PNP and NPN bipolar transistors or P-channel and N-channel MOS field effect transistors are used as power transistors, respectively.

[0133] In the self-oscillator described at the outset with two types of bridge transistors and MOS field effect transistor bridge branches with external controllers, the bipolar self-oscillator with flux driver is the most demanding, which is why the starting point of the entire invention disclosed here is it and the above derivation, which is always only generally formulated in the possible passages. In the case of the bipolar self-oscillator, the reason is pure current control with the smallest control voltage rise and with two control voltage polarities. At least three independent variables must be kept under control at the same time and without interruption, even in the case of exceeding the switching phase boundaries.

[0134] For all considerations, resonant zero voltage switching (ZVS) is assumed, which minimizes losses and radio interference. The solution found in this way fulfills its unchanged tasks without or only with slight modifications, also in the case of MOS field effect transistor self-oscillations, i.e. for example in monostable drives or in current-regulated self-oscillators, or in external control, where only those examples of N-channel MOS field effect transistor bridge branches are elaborated in detail depending on the frequency of the actual implementation. For bridge branches equipped with field effect transistors, it is particularly easy to trigger their two drivers directly from an integrated control IC (alias μProcessor or ASIC), which at the same time monitors and regulates the entire remaining circuit containing the bridge branch considered.

[0135] Since the gate voltage, which is significantly higher than the common base-emitter saturation voltage (max. 1 V), for example 15 V, exceeds the reverse voltage of the emitter-base diodes of most auxiliary transistors and thus destroys these transistors, a so-called base series diode must be connected between the base of the first auxiliary transistor in the first semiconductor chain and the gate line of the MOS field effect power transistor, with its anode on the base of each first auxiliary transistor and its cathode on each gate line. In order to reliably cut off the high gate voltage by this diode, a so-called base protection diode is additionally required from its anode, which is connected with its cathode to the first connection point. Finally, the base protection diode is anti-parallel to the emitter-base diode of the first auxiliary transistor to be protected. Thus, the majority of the positive gate voltage, in inverse proportion to the capacitances acting in parallel with it respectively, is distributed to this then cut-off series diode and to the likewise cut-off base-emitter diode of the first auxiliary transistor of the second semiconductor chain.

[0136] Since the operating voltage of the subsequently switched-on bridge transistor remains very small and is mainly quiescent, the activation capacitance of the bridge capacitor or activation capacitor is in parallel with the hitherto unprotected base-emitter diode of the first auxiliary transistor of the second semiconductor chain and keeps its cut-off voltage very small, since the capacitance of the above-mentioned base series diode is very small compared to the activation capacitance. However, the turn-on voltage peak, the flux voltage change of the currently switched-on power transistor and possible changes in the supply voltage are almost completely added to this actually very small reverse voltage on the same path. Therefore, in addition (not shown) each drive circuit can connect a capacitor with a capacitance of approximately the activation capacitance in parallel with the base-emitter diode of the first auxiliary transistor of the second semiconductor chain, or (not shown) the same protective measures can also be applied to the second semiconductor chain, which are described in the form of the base series diode and the base protection diode for the first semiconductor chain. In contrast thereto, in the case of all bridge legs equipped with MOS field effect transistors, the second PN junction of the second semiconductor chain is omitted and replaced by a direct connection from the gate to the collector of the switched-on first auxiliary transistor of the shortened second semiconductor chain.

[0137] However, when the drive circuit is matched to the efficiency of the actual drive or at least partially merged therewith, the externally controlled bipolar bridge leg can likewise be improved by the same drive circuit and in the same direction. The self-oscillator with saturation drive has different disadvantages and is not considered further below.

[0138] However, if in the case of external control, in particular the real-time space- gap determination or the subtask of space-gap optimization (the latter is necessary in the case of boundary ZVS or non-ZVS) is again transferred into analog hardware as suggested herein, for the remaining control and regulation tasks, smaller and thus less costly ICs can be installed in the same overall circuit. The drive circuit according to the invention takes over the required real-time space-gap optimization. In contrast, the same properties are easily adjustable from the outside for the most complex bipolar self-oscillator bridge legs to be driven in a hitherto unknown large operating range.

[0139] A further advantage is the intrinsic non-ZVS turn-off independent of the class of the bridge transistor, since in the case that the recharge current in the bridge capacitor is exhausted before its final charging or discharging, the said derivation path becomes high-impedance, whereby the bridge leg wants to enter a Valley-Detect-Modus (switching on shortly before or exactly at the voltage minimum or maximum of the polarity change phase after one turn-off in the bridge transistor). However, the further beyond the so-called ZVS border in the operating area of the bridge leg, the more strongly this has switching-on losses.

[0140] If these switching-on losses exceed the power reserve required for self-oscillation, the self-oscillator turns off. In the case of external control, this is an additional positive feature, since the ZVS border can thereby be identified and the corresponding optimum switching-on point in time can subsequently be identified, and since it is also possible to work slightly beyond this border due to the minimization of real-time losses connected with the switching-on point in time, without additional code being required for this, provided that the switching-on losses produced here do not exceed the flux losses at the rated operating point.

[0141] If, in the case of external control for a MOS field effect transistor bridge leg, the so-called high-side driver (which essentially only bridges the potential jumps for the high side of the bridge leg and otherwise only transmits the control signal applied on its input for the respective high-side transistor to its output) obtains a channel for the low side, then the high-side driver can additionally fulfill the task of the space occupation control independently of the actual external control when the invention disclosed here is integrated into said high-side driver. Due to the series impedance required for this and especially due to the intervention possibilities for the drive circuit, the thus expanded high-side driver requires an additional channel for its low side. In the case of a merger of the output transistor, especially the always present high-side amplifier, with the drive circuit, a variant with a second auxiliary transistor in the second semiconductor chain is formed, but this variant does not impose a second auxiliary transistor in the first semiconductor chain.

[0142] Preferred embodiments can be found throughout the disclosure, wherein in the illustrations not always a detailed distinction is made between the device aspect and the use aspect; in any case, the disclosure is to be read implicitly with respect to all categories of the invention.

[0143] Further advantageous refinements and design options of the drive circuit for the base-emitter path of a bipolar transistor or for the gate-source path of a MOS field effect transistor according to the invention result from the following description. BRIEF DESCRIPTION OF DRAWINGS

[0144] Further advantages, features and details of the application result from the following description of exemplary embodiments and from the figures, in which identical or identically acting elements are provided with the same reference signs. In the figures:

[0145] Figure 1a The basic structure of a drive circuit for the low side of a bridge leg is shown in a circuit diagram and a semiconductor chain diagram,

[0146] Figure 1b The basic structure of a drive circuit for the high side of a bridge leg is shown in a circuit diagram and a semiconductor chain diagram,

[0147] Figure 2a The basic structure of a known self-oscillating bridge leg with bipolar transistors as power transistors, a flux driver, a ladder capacitor and a bridge capacitor is shown,

[0148] Figure 2b The turn-on phase of a bipolar transistor according to Figure 2a is shown,

[0149] Figure 2c The polarity change phase of the bridge center point of Figure 2a in the case of zero voltage switching (ZVS) is shown,

[0150] Figure 2d The polarity change phase of the bridge center point of Figure 2a in the case of boundary ZVS is shown,

[0151] Figure 2e The polarity change phase of the bridge center point of Figure 2a in the case of the so-called valley-detect modus in the case of non-ZVS is shown,

[0152] Figures 3a to 3h The current path in the case of a polarity change of the center point of a half bridge from positive to ground in the case of ZVS is shown, the current path having bipolar transistors as power transistors and having a flux driver,

[0153] Figures 4a to 4h The current path in the same circuit is shown, only for the polarity change of the center point from ground to positive in the case of ZVS,

[0154] Figure 5a A first embodiment of a fully self-oscillating bipolar bridge leg with a bridge capacitor is shown,

[0155] Figure 5b A corresponding embodiment of a fully self-oscillating bipolar bridge leg with an activation capacitor is shown, by means of which a symmetrical total ladder capacitance can be achieved,

[0156] Figure 6 switching processes measured in a circuit according to the prior art are shown,

[0157] Figure 7 those switching processes measured in a circuit according to the prior art are shown, Figure 6 Figure 5a

[0158] Figure 8 Figure 7

[0159] Figure 9a

[0160] Figure 9b

[0161] Figure 9c

[0162] Figure 9d

[0163] Figure 9e

[0164] Figure 9f

[0165] Figure 9g

[0166] Figure 9h Figure 9c ​​​​​​​​​​​​​

[0167] Figure 9j a further embodiment of the structure of the drive circuit according to Figure 9e is shown, but wherein the PN junction in series with the collector of the first auxiliary transistor (NPN) of the second semiconductor chain is extended to a second auxiliary transistor in PNP polarity for the same semiconductor chain and with a collector on the reference potential, respectively, whereby a second auxiliary thyristor is functionally formed by the two auxiliary transistors of the second semiconductor chain,

[0168] Figure 9k a further embodiment of the structure of the drive circuit according to Figure 9j is shown, wherein the first and second auxiliary transistors of the second semiconductor chain are configured as an integral second auxiliary thyristor,

[0169] Figure 9l a further embodiment of the structure of the drive circuit according to Figure 9g is shown, which has a second auxiliary transistor (PNP) of the second semiconductor chain according to Figure 9j ,

[0170] Figure 9m a further embodiment of the structure of the drive circuit according to Figure 9k is shown, which has a second auxiliary thyristor according to Figure 9g ,

[0171] Figure 10a a third embodiment of a full self-oscillating bipolar bridge leg of a bridge capacitor with a drive circuit according to Figure 9l is shown,

[0172] Figure 10b a corresponding fourth embodiment of a full self-oscillating bipolar bridge leg with an activated capacitor is shown, whereby a symmetrical total ladder capacitance can be achieved,

[0173] Figure 10c a coupled bead choke is shown,

[0174] Figure 10d a lower bead choke and an upper bead choke in a common double-hole magnetic core are shown,

[0175] Figure 10e voltages on the drive transformer and the coupled bead choke are shown when the low-side reverse diode just turns on, and

[0176] Figure 10f those voltages are shown when the high-side reverse diode turns on,

[0177] Figure 11aa self-oscillating bridge branch with MOS field effect transistors according to the prior art with monostable flip-flop-time controller is shown,

[0178] Figure 11b a bridge branch according to Figure 11a the bridge branch according to the invention is improved in such a way that the bridge branch is driven with the optimum on-space gap respectively,

[0179] Figure 12a a current-regulated self-oscillator according to the prior art is shown,

[0180] Figure 12b an improved version of a self-oscillator according to Figure 12a the invention is shown,

[0181] Figure 13a a simple circuit structure of a bridge branch with external controller and drive circuit according to Figure 1a and Figure 1b with MOS field effect transistors is shown,

[0182] Figure 13b a circuit according to Figure 9c is shown, which is improved by a drive circuit according to Figure 13a especially adapted to tilt control and placed with a series resistor in such a way that small thyristors are triggered thereby at the same time,

[0183] Figure 14a the possibility to improve a high-side driver according to the invention is shown,

[0184] Figure 14b an improved version of a bridge branch according to Figure 14a is shown, in which the PNP transistor of the emitter follower becomes at least on the high side a second auxiliary transistor of the second semiconductor chain. DETAILED DESCRIPTION

[0185] Figure 1a The left-hand part shows the basic structure of a drive circuit 600, 610 according to the invention with a control line 103 on top and a connection line 400 for the reference potential on the bottom, which together on the low side of the bridge branch form the control input of the first power transistor. On one side the actual driver (not shown) is connected, which provides the necessary drive auxiliary energy and especially the switch-off point in time, on the other side the control electrode is connected to the control line and the reference potential of the power transistor to be switched on (not shown) is connected at least ohmically to the connection line, i.e. in summary to its control input. Here it is not important to which side of the drive circuit 600 or 610 the connection is made.

[0186] The essential feature is the PNP auxiliary transistor 1, which is also referred to as "first auxiliary transistor of the first semiconductor chain" or simply as "transistor 1" in the following, and whose base is connected to the control line 103, and the NPN auxiliary transistor 2, which is also referred to as "first auxiliary transistor of the second semiconductor chain" or simply as "transistor 2" in the following, and whose base is connected to the connection line 400 for the reference potential. Both transistors are thus operated in a basic circuit, because by the actual driving connection, or the connection from the control electrode to the reference potential, especially in the case of bipolar transistors as the power transistor to be connected, or both can be very low-impedance, i.e. in figurative terms, between the left-hand terminal on the left-hand side and the right-hand terminal on the right-hand side of the not-shown vertical connection of Figure 1a the maximum possible negative control voltage.

[0187] Also essential is the connection of both emitters of the transistors 1 and 2. On the first connection point 6 formed thereunder, at least one capacitor is connected, through which the current flows for exactly as long as the driving circuit is to be activated, and also approximately in the amount to be processed by the driving circuit. The basic circuit is thus sufficient, in which no amplification takes place, but which is virtually arbitrarily fast for this purpose.

[0188] The entire driving circuit is thus activated by the first connection point 6. The at least one capacitor connected there can be connected on its other end to virtually all potentials of, for example, a bridge leg equipped with the driving circuit 600 or 610, without being connected to the respective reference potential 400. The capacitor connected to the first connection point 6 thus results in a higher potential in principle. It can be the activation capacitor 8 or the bridge capacitor 5. The activation capacitor 8 is advantageously connected to the operating potential of the power transistor to be connected and completes the driving circuit 610. The bridge capacitor 5 is advantageously connected to the first connection point between the two emitters of the second driving circuit of the same bridge leg and completes the driving circuit 600.

[0189] If the negative control voltage for switching off the power transistor is not required, or is not generated by the actual driver, or is not disturbed when such a negative control voltage is clamped in a low-ohmic manner by the driving circuit, for example when a coupling capacitor connected in series with the actual driver must be recharged, then the driving circuit is now complete. In Figure 1aThe diodes 3 and 4 drawn on the left side of the figure can be replaced conceptually by a direct connection. Since both transistors 1 and 2 are high-impedant for a positive control voltage when not activated: the collector-base diode of transistor 1 is cut off, and also the base-collector diode of transistor 2. Both PN junctions defined by these parasitic diodes are therefore also referred to as "first PN junctions" in the following. A negative control voltage is clamped by a negative collector current flowing through both first PN junctions.

[0190] However, in order to be able to achieve a negative control voltage, this is absolutely necessary for a driver of pure inductive coupling for each other potential side or bridge half in the bridge branch without a coupling capacitor in series with the actual driver, it is necessary to deactivate the parasitic collector-base diodes of both transistors 1 and 2. This is achieved in that in the direction of flow of the positive collector current, a further PN junction in the form of a flux diode 3, which can be a fast diode or a Schottky diode, is inserted between the collector of transistor 1 and the reference potential 400, and in the direction of flow of the positive collector current of the same, a further PN junction in the form of a flux diode 3, which can likewise be a fast diode or a Schottky diode, is likewise inserted between the control line 103 and the collector of transistor 2.

[0191] Seen from the known viewing direction, the cathodes of both flux diodes 3 and 4 point downwards. In the case of a positive control voltage, both flux diodes are theoretically conductible and in fact also conductible when the auxiliary transistors connected in series are activated, whereas in the case of a negative control voltage, both flux diodes 3 and 4 generally enter a cut-off operation and thereby ensure that the drive circuit does not further interfere with the actual drive. Both PN junctions defined by these flux diodes are also referred to as "second PN junctions" in the following.

[0192] It should be mentioned here Figure 1a the right-hand part of the figure. The P or N regions of the spherical expansion of the semiconductor chain show the respective doping, their delimitation from the adjacent regions by means of the dashed PN junctions, and at the same time the control line 103 or the reference line 400 running horizontally through them. Due to their higher doping, each emitter is marked with a double letter, i.e. "nn" or "pp". The same nomenclature applies to the right-hand side of the following Figure 1b the right-hand side of the figure series 9 as well as to all right-hand sides of the figure series 9.

[0193] By inserting the flux diodes 3 and 4 between the collector electrodes of the transistors 1 and 2 and the line, which would actually be connected to the line if these collector electrodes only fulfilled their primary function, each drive circuit 600, 610 forms a first semiconductor chain consisting of the transistor 1 and the flux diode 3 and a second semiconductor chain consisting of the transistor 2 and the flux diode 4. Each semiconductor chain respectively connects two identically doped regions of its constituent parts to each other: the first semiconductor chain comprises the mutual connection of the two P regions, the connection of the collector electrode of the transistor 1 to the anode of the flux diode 3, and the second semiconductor chain accordingly comprises the connection of the two N regions, the now connection of the collector electrode of the transistor 2 to the cathode of the flux diode 4.

[0194] Thus, between the control line 103 and the reference potential 400, i.e. in parallel with the control input of the power transistor to be connected, two connections are formed consisting of the two semiconductor chains described above, which always comprise at least two PN junctions of opposite polarity when viewed in the known viewing direction. Here, the base-collector junction la or 2a belonging to the transistor 1 or 2 is always the first PN junction of the semiconductor chain in question, and the PN junction 3a or 4a of the diode 3 or 4 is the second PN junction thereof. By this construction, the two connections or semiconductor chains in parallel with the control input have an inherent high impedance for both directions of the control voltage when the two connections or semiconductor chains are not activated.

[0195] Thus, each semiconductor chain is in principle connected to the control line 103 and the reference line 400 with identically doped regions, the first semiconductor chain with two N regions and the second semiconductor chain with two P regions. In contrast thereto, the two connections of the semiconductor chains to the control line 103 and to the reference line 400 are respectively doped differently.

[0196] The base-emitter junction lb or 2b of the transistor 1 or 2 is respectively the third PN junction of the semiconductor chain in question. Again, when viewed in the known viewing direction, the polarity of this third PN junction of each semiconductor chain is respectively opposite to the polarity of the adjacent first PN junction. In Figure 1a In the right-hand part of Fig. 1, the non-amplification can be seen by the illustration: the activation of each semiconductor chain respectively takes place "outside the control input", i.e. from above the control line 103 or from below the reference potential 400.

[0197] In addition to two exceptions, it is further given below that, when again viewing Figure 1athe left-hand part of the figure, as a continuous feature circuit diagram, a chain consisting of four diode symbols of the same orientation (both diode symbols or all occurring base-emitter symbols below are taken into account) is obtained, viewed in the flow direction, starting at the control line 103 and ending at the associated reference potential 400. In the middle of this chain, i.e. in the first connection point 6, one pole of the bridge capacitor 5 or the activation capacitor 8, respectively, is connected.

[0198] The same applies to functionally identical components in all circuits or measurements described below, so that in those cases the respective description is no longer repeated. The same applies to all other designations that have not yet been introduced. They are only described at the introduction and these descriptions subsequently apply likewise or at least by analogy. Thus, instead of components, only their position or their electrical effective value can be described, and instead of nodes or connection points, the entire line or subgrid connected thereto can be described, and vice versa.

[0199] Figure 1b The high-side variant 700 or 710 of the drive circuit according to the invention is shown according to Figure 1a the upper control line 203 and the lower connection line 206 for the now jumping or oscillating reference potential, which in most cases on the high side simultaneously shows the midpoint of the bridge branch considered or the AC voltage output. Otherwise, the working principle and the surroundings are identical to Figure 1a the working principle and the surroundings of the drive circuit according to the invention, so that they are not explained again here. The left-hand part of the figure shows the circuit diagram, the right-hand part the semiconductor chain diagram.

[0200] The functionally identical elements of the drive circuit 700 or 710 for the high side, with one exception, have the respective designations increased by ten with respect to the designations of the low side of the same bridge branch considered:

[0201] The emitter of the first auxiliary transistor 11 (PNP) or 12 (NPN) is connected to the first connection point 16

[0202] above, the flux diode 13 is connected in series with the collector of the transistor 11 in the flow direction of the positive collector current, and the further flux diode 14 is connected in the same way with the collector of the transistor 12.

[0203] The same applies to the functionally identical components in all circuits or measurements described below, so that in those cases the respective description is no longer repeated. The same applies to all other designations that have not yet been introduced. They are only described at the introduction and these descriptions subsequently apply likewise or at least by analogy. Thus, instead of components, only their position or their electrical effective value can be described, and instead of nodes or connection points, the entire line or subgrid connected thereto can be described, and vice versa. Figure 1aThe opposite is true for the low-side circuit or first drive circuit 600 or 610 shown in the middle: for the second pole of the activation capacitor on the high side of the bridge leg, two potential directions are possible: in the drive circuit 710, as on the low side, the so-called activation capacitor 18 advantageously leads up to the operating potential of the second power transistor to be connected, whereas in the drive circuit 700 the bridge capacitor 5 leads down, since on the high side only its second end is involved. Therefore, each bridge capacitor 5 always belongs to two drive circuits 600 and 700 connected thereto, which together form a bridge circuit 800. In addition to the bridge leg output or internal bridge midpoint 206 as a variable reference potential, which marks the only technical difference between the drive circuit 710 and the drive circuit 610, the different direction of the bridge capacitor 5 is the second point that distinguishes the high-side circuit or second drive circuit 700 from the low-side circuit or first drive circuit 600.

[0204] The combination of the drive circuit 700 with the drive circuit 600 forms the entire so-called bridge circuit 800, the combination of the circuit 710 with the circuit 610 forms a bridge circuit 810. As long as only such drive circuits of the same type are always combined (which is not mandatory, however, is obvious and therefore always taken as a basis in the following for the sake of simplicity), the individual units and tens of the individual drive circuits are transferred to the numbering 8** of the entire bridge circuit.

[0205] Figure 2a The initial position of the invention is shown. The main feature of the prior art EP 992 113 B2, the bridge capacitor 5 (there C6) in combination with a bridge leg equipped with bipolar transistors as power transistors, works here with flux drive, unlike the prior art. The decisive difference between the ladder capacitor 202 (there C1) and the bridge capacitor 5 has become apparent here: the latter is located on the drive side of the bridge leg in principle, the former on its output side or load side.

[0206] All power elements of the bridge leg, such as its power transistors 100, 200 or its reverse diodes 101, 201 and its nodes, lines, voltages or currents, have a figure reference on the high side that is one hundred greater than on the respective low side, except for the respective reference potentials 206 or 400. In contrast, the figure references of the components of the driver for the same bridge leg, as already outlined above in Figure 1a and 1b differ by the value ten between the low side and the high side. This numbering principle continues in all subsequent figures with circuit diagrams and is therefore no longer re-stated there.

[0207] A load (not shown) is connected at the node 300 to which the high-frequency alternating voltage is applied, which is advantageously tuned inductively-ohmically at all occurring frequencies of the alternating voltage and is connected via at least one DC decoupling capacitor (not shown) to at least one pole of the same DC voltage from ground 400 and positive 500, which also energizes the bridge branch. If this load should have a resonant characteristic, the at least one DC decoupling capacitor can be intentionally reduced in order to form an inherent frequency together with the inductive part of the load, which should be lower than the lowest occurring frequency of the alternating voltage at the node 300, wherein an attenuation through the ohmic load can be taken into account.

[0208] The inductance of the measuring winding 27 of the drive transformer for the flux driver is not sufficient for this inductive part of the load, which is why at least one further inductance (not shown), for example an isolation transformer, must always be connected in series. The formation of a resonant load characteristic is meaningful, in particular in the energy supply of LEDs, because such a load becomes nonlinear not only due to itself but also due to a rectifier (not shown) which is forced and suitable for the above-mentioned high-frequency alternating voltage. This effect is attenuated by the resonant coupling of such a load to the bridge branch, which is always given in the following text once the node 300 or 206 is shown in the drawing.

[0209] The load current 351 should therefore lag behind the output voltage 206. If the voltage is inverted from ground 400 to positive 500, the load current 351 is still positive according to the depicted arrow direction, and if the voltage on the node 206 oscillates from positive to ground again, it is correspondingly still negative. This current 351 is permanently applied by the inductive part of the load into the path of the bridge branch, which is different depending on the conduction phase, which supplies the load. In contrast, the bridge branch applies the voltage at its node 206 into the load. In this way, an alternating current 351 is generated from the DC voltage 500.

[0210] As long as the lower bridge transistor or first power transistor 100 is conducting, its collector current 311 corresponds to the load current 351, which is essentially positive in this conduction phase as depicted. For this, a base current 321 is required, which flows into its base at the control input, i.e. between the lower control line 103 and ground 400, with a positive voltage. The further reference signs 310 and 312 describe the starting value and the end value of the collector current at the beginning and at the end of the conduction phase of the first power transistor 100. The reference signs 320 and 322 designate the respective starting value and end value of the associated base current.

[0211] If the load current 351 is just negative, the upper bridge transistor or the second power transistor 200 is thus mostly on. In order to be able to further clearly show this, the marking arrow of the upper base current 421 is drawn here negatively, especially in the series of figures 2c to 2e.

[0212] Figure 2b It is helpful to explain why, in the flux drive, two mutually independent fluxes or two independent currents resulting therefrom have to be continued to be guided during each polarity change process of the midpoint 206 of the bridge branch according to Figure 2a The typical on phase with a resonantly coupled load, for which just the above has been set out, at a given inverter frequency, at least slightly inductive tuning can be identified by the fact that the transistor 100 is switched on with a negative collector current 310 and has already been switched off again with a positive collector current 312. This is a prerequisite for the bridge branch considered to be able to work in a zero voltage on operation or "zero voltage switching (ZVS)" with particularly low losses and low interference. This mode of operation is, in contrast, the basis for the entire explanation below.

[0213] The on time or on duration 551 of the signal curves shown begins at the on time point 550 and ends at the off time point 552. The number "3" in the hundreds place indicates all that is related to the load directly or indirectly. Then, in the tens place, "1" describes the collector current of the power transistor 100, "2" the real base current of the same transistor, "3" the theoretical base current on the same path, and "4" the difference of the last two, i.e. indirectly the magnetic flux in the drive transformer. The units place indicates the time points: "0" the beginning of the on phase of the bridge transistor 100, "1" its course, "2" the off time point of the same on phase, and "3" the end of the subsequent polarity change phase or the beginning of the on phase of the opposite bridge half. The same applies to the time points and durations whose reference signs usually begin with 55*. In addition, there is also a duration 554 corresponding to the already introduced waiting time. This waiting time is the same as the on space gap or dead time known from the literature.

[0214] The base current 320 is positive when turned on and therefore negative when turned off, as shown in 322. This is due to the inherent magnetization of the drive transformer (stored during polarity reversal) and the inherent magnetization of the inductor that may be connected in parallel with it (if the effect of this magnetization must be amplified). Therefore, this method is called "flux-driven" because the resulting positive turn-on base current 320 and negative turn-off base current 322 not only fully realize the operation of the bridge branch under consideration but also benefit the operation of the bridge branch under consideration.

[0215] Without the inductor and with an ideal drive transformer (L M →infinity) The theoretical base current 331 corresponds to the turns ratio n between the drive windings 7, 17 and the measuring winding 27. Figure 2b The collector current 311 decreases by 4, and the curvature of the base current 321 is defined. The difference between the theoretical and actual base current at turn-off, the “tilt position” 342 mentioned above and imposed by the main inductance of the drive transformer and possible parallel inductance, enables the power transistor to be turned off again and remains substantially constant during the subsequent polarity reversal process, as can be clearly seen at a similar size 343. Conversely, the load current 351 (which is imposed through the load circuit and corresponds to the collector current 311 during the conduction phase observed previously and shown in solid line) rapidly decreases to a value 313. All the curves of this polarity reversal process (which can be seen, for example, by the trapezoidal capacitor already mentioned) Figure 2a The position 202 in the middle is shown as a dashed line. The independence of these two fluxes can be clearly seen by the fact that 313 is significantly smaller than 312 compared to 343 relative to 342, or at the bend at point 322 in the direction of 321.

[0216] For the following explanation, it is assumed that the symmetrical duty cycle in the output AC voltage of this bridge branch is such that the positive voltage block at node 206 is as long as the zero pause between them, or that durations 551 and 554 always have the same length, regardless of which bridge half is currently conducting. The end of this polarity reversal process then transitions to the beginning of the considered conduction phase in such a way that the absolute value 313 of the "positive dashed line" and the initial negative collector current 310 is the same. This, i.e., 330 = -333, also applies to the theoretical base current. At the beginning of the conduction phase, the actual base current 320 is generated by adding the difference 343 to the theoretical base current 330 at the beginning of the considered conduction phase, the difference being lower than the theoretical base current 333 at the end of the polarity reversal phase.

[0217] Between these two, separate conduction phases occur in the other bridge half, symmetrical to the phase shown, and separate polarity reversal phases (both not shown), which in practice achieve the aforementioned symmetrical transmission. The base current value at the start of this conduction phase in the other bridge half corresponds to point 323, where the theoretical base current 333 decreases by the total difference 343 up to that point. Because it remains substantially constant not only during a polarity reversal process but also across multiple successive polarity reversal processes with alternating signs, 323 = -320 also applies here.

[0218] During the polarity reversal phase, the current generated by the flux must find two at least partially independent pathways through the circuit, which is effectively high impedance during each polarity reversal phase due to the two off power transistors. If in Figure 2a If there is no capacitor at position 202 or 5, this results in an infinite voltage slope during the polarity reversal phase or during an infinitely short polarity reversal phase. Therefore, no space gap is needed. This continued approach of avoiding capacitors, driven by increasingly faster power components, has practically led to inverters with ultrafast or ultrahard switching. Two problems remain: their significant radio interference and their switching losses, which are formed by energy stored in the capacitors of the fast power components, and this energy is lost as heat during each switching process.

[0219] The aforementioned radio interference problem requires flattening the voltage slope during these polarity switching processes to achieve a finite and measurable duration. Switching losses are eliminated if this duration is achieved by waiting in a manner that keeps both bridge transistors off throughout the period, and especially the bridge transistor to be turned on next, in the waiting position.

[0220] The second method outlined here results in a resonant inverter, achieved by intentionally using capacitors connected in parallel with at least the operating path of the power transistors, i.e., for example in... Figure 2a At position 202. The objective is the aforementioned zero-voltage turn-on or zero-voltage switch (ZVS). The present invention, as a further step in this second method, involves directly using at least a portion of the capacitance of the aforementioned "intentionally used capacitor" for time control of the waiting position.

[0221] Therefore, in Figure 2a A bridge capacitor C is provided at position 5. BIt is connected in parallel with the bridge branch between the two drive circuits used to guide the current, which acts essentially on the drive side of the bridge branch because the current is generated by the difference 342 or 343. Conversely, the load current 351, having an initial value 312 for the polarity reversal process, must be able to continue flowing on the output side of the bridge branch, and therefore uses according to Figure 2a A known trapezoidal capacitor C is located between the midpoint 206 of the bridge and the ground or positive position 202. T Consequently, the entire bridge branch also becomes lower impedance during its polarity reversal phase, as shown by a flat voltage slope. The output voltage at node 206 is trapezoidal, which is why the "intentionally used capacitor" that causes this is usually referred to accordingly.

[0222] During the transition from the solid line to the dashed line, the ratio C between the bridge capacitor and the trapezoidal capacitor... B / C T According to Figure 2b Estimate at the corresponding height ratio of 342 / 312. The unremarkable result is that the bridge capacitor and the trapezoidal capacitor should be approximately the same size. If a smaller total trapezoidal capacitance (corresponding to C) is permissible... B +C T If a bridge capacitor is used, it is preferred because it is essential for the function of the circuit.

[0223] Using the turns ratio n (the number of turns in each drive winding 7, 17 is n times that of the measuring winding 27 of the drive transformer) Figure 2b In the case of n=4), the theoretical final value of the base current is calculated as follows:

[0224] 332 = 312 / n.

[0225] The actual base current at turn-off time 322 is formed by the reduction of the theoretical base current 332 at the same time point by the magnetization 342 of the flux driver's drive circuit at the end of the considered conduction phase, i.e.

[0226] 322 = 332 - 342

[0227] Therefore, combining the previous equations, we get

[0228] 322 = 312 / n - 342

[0229] Assuming that the negative base current 322 has half the amount of the positive collector current 312 when the associated bridge transistor is turned off, i.e., if 322 = -312 / 2 applies, C can be calculated more accurately. B / C T :

[0230] 312 / n - 342 = -312 / 2.

[0231] For the above capacity ratio, this can be converted to:

[0232] C B / C T = 342 / 312 = 1 / 2 + 1 / n.

[0233] Thus, the higher the number of turns ratio n, the smaller the bridge capacitance. In the same direction, in the case of higher power in the bridge branch, the intrinsic magnetization of the drive transformer plus the current in its optional parallel inductance is reduced relative to the collector current to be switched off, so that the ladder capacitance can be up to three times as large as the bridge capacitance. In other words, the higher the power, the smaller the value of the term "1 / 2" in the above equation, which only represents the ratio of the switched-off collector current to the switched-off base current and thus in fact depends on the respective operating point.

[0234] If the base current at the switching-off point of time is 1 / x of the positive collector current, the following equation results for the sought-after capacity ratio:

[0235] C B / C T = 342 / 312 = 1 / x + 1 / n

[0236] Figure 2b It is also shown that the capacity ratio is almost never exactly matched, so that the drive circuit in the form presented here is absolutely necessary. In fact, experiments for the present invention started with a circuit according to Figure 2a , i.e. with a bridge capacitor 5 fixedly connected between the two base lines in addition to the ladder capacitor 202: the measurement quickly showed that this can work, but always only at one operating point and also always only at the only point in time within the polarity change process. Because at the switching-on point of time, i.e. at the end of each polarity change process, C T ' must be much smaller than C B '. This gives a second possible estimate of the ratio of the bridge capacitance to the ladder capacitance C B ' / C T ' from the same figure is 343 / 313.

[0237] In the same opposite direction, it is also shown that the fact that at low active power taken from the bridge branch, the above x < 2, i.e. relatively more negative base current is required to switch off. Thus, low power means relatively large bridge capacitance. Therefore, instead of reducing its bridge capacitance on the path from small to large power through the required large operating window of the same bridge branch, both power transistors are equipped with a drive circuit.

[0238] However, with the relatively large bridge capacitor C B or without a further ladder capacitor C T "too much" current must flow through the bridge capacitor directly after each turn-off process. Thus, the load applies more current to the driver than the drive winding itself, for which excess an additional path must be provided in the driver circuit. "Too little" current can also flow in the bridge capacitor before each turn-on process, which means that the drive winding applies more current than the load, for which an additional path is to be provided in the driver circuit. Figure 8 The measurement at this operating point is shown. Thus, during the entire operating cycle of a self-oscillating bridge branch with a flux driver and equipped with bipolar power transistors, the operating point passes through at least 10, usually 12 phases according to the circuit diagram Figures 3a to 4h However, first, in the series of figures 2c to 2e, zero voltage switching (ZVS) will be described in more detail:

[0239] Figure 2c The conventional ZVS is shown. It arises from the right of Figure 2b with the subsequent turn-on phase, the course of which is produced by the mirror image of its left side on the time axis. Here, the theoretical base current is omitted and for this reason the bridge branch output voltage 206, which is significantly more important here, is plotted. The curve of the load current 351 right of the time point 553 thus corresponds to the beginning of the curve of 311 mirrored on the time axis from the time point 550 of Figure 2b The newly recorded voltage curve 206 at the bridge midpoint clearly shows the turn-off time point 552 at which the curve leaves the zero line, the turn-on time point 553 at which it reaches the "positive" line of the opposite bridge half, and the waiting time or on-space gap 554 during which.

[0240] The characteristic of the ZVS or "conventional ZVS" is that the target voltage "positive" is reached by the bridge midpoint 206 before the load current 351 reaches the zero line. The shaded area 350 under the load current curve 351 after the time point 553 at which the voltage "positive" is reached is likewise characteristic of this operating mode. This area 350 describes the energy that is fed back into the voltage supply each inverter half-cycle. The conventional ZVS requires a certain amount of reactive power in the load circuit, which justifies the name "resonant converter". The energy is fed back at the beginning of the half-cycle, respectively, is caused by the lagging load current 351, and shows a slightly inductive tuning of the load.

[0241] The base current 321 is shown as in Figure 2bcontinues beyond the end point 322 as indicated by the dashed line and thereafter although continuously transitioning to the base current 421, shown as a negative value matching the curve of 351 on the other bridge half, is also bent at this second seam point 323, as already at point 322, protruding around this second seam point by two dashed diamonds. These two bends illustrate that if the bridge leg according to Figure 2a is to be properly applied to this ZVS mode, the involved base currents have to be handled by additional drive circuitry.

[0242] In Figure 2d the reactive power is minimized and the hatched area 350 disappears. For this, the time interval 554' between points 552' and 553' is here significantly longer than the time interval between 552 and 553 above. Thereby, a second important task for the drive circuitry becomes apparent, as detailed below with respect to Figure 3d or 4d, to always first put the bridge transistor to be switched on next into the standby position just as long as required by the waiting time for the current operating point. Here, in Figure 2d the "borderline ZVS" is reached, which is characterized in that the point in time 553' at which the midpoint voltage 206' reaches the input voltage "positive" coincides exactly with the point in time at which the load current 351' changes its sign. Due to the residual current 313' which then disappears, the midpoint voltage 206' here asymptotically reaches its target level. This is caused by the significantly smaller turn-off current 312' at the beginning of such a borderline polarity change process compared to 312 in Figure 2c .

[0243] In the case of a slightly further reduced turn-off current 312" according to Figure 2e ZVS can no longer be achieved, so this operating range is often referred to as "non-ZVS". The energy stored in the inductive part of the load at the turn-off point in time 552" is no longer sufficient to cause the polarity change of the bridge midpoint 206" to the voltage "positive". This can be seen at the bend at point 313" in the curve of the load current 351", which is highlighted by the dashed square around it. This switching process is lossy, as can also be seen from the fact that the curve of the bridge output voltage 206" simultaneously performs a jump. The resulting loss corresponds to the square of the height of this jump multiplied by half the total trapezoidal capacitance, to which the output capacitances of both bridge transistors add.

[0244] In Figure 2eThe so-called "Valley-Detect-Modus" mode is illustrated below. This valley-detection mode, a special characteristic of non-ZVS, turns on the bridge transistor precisely when its operating voltage is at its minimum, i.e., when the midpoint voltage 206" is closest to the target value "positive", or then when the load current crosses the neutral line. Therefore, the switching losses unavoidable in non-ZVS are at least minimized. Since this resonant current crossing the neutral line flows at least proportionally, the notoriously difficult maximum or minimum value identification is perfectly accomplished by the trapezoidal capacitor and / or bridge capacitor. Therefore, the external controller further illustrated below (…) Figures 13a to 14b In the case of self-oscillation, the "valley detection mode" is generally ensured, and conversely, in the case of self-oscillation, it only continues until the point where the aforementioned unavoidable switching losses prevent self-oscillation capability. Therefore, in the case of self-oscillation, the present invention acts as a "non-ZVS shutdown," thereby always ensuring optimal efficiency and optimal radio protection operation.

[0245] Figure 3a Through bridge capacitor 5 Figure 1a and 1b The left portion, combined with the two drive windings 7 and 17 that act as the actual drivers on its left, and one of the bridge branches connected to its right, consists of two bipolar power transistors 100 and 200 without additional trapezoidal capacitors. The state of the second bipolar power transistor 200 directly on the high side before turn-off is shown, with current flowing from its base terminal or control line 203. (Since some of the components numbered above are activated later, their reference numerals are only from...) Figure 3b (Start drawing.) The maximum collector current has typically passed at this point in time, but the still high collector current of power transistor 200 is split into a positive emitter current and approximately the same amount of negative base current. The positive emitter current flows directly out of node 206, while the negative base current flows into the upper drive winding 17, thus the base current must detour through here. In this way, the stored charge in the base-emitter diode of power transistor 200 is optimally cleared, allowing it to be turned off with low loss.

[0246] For simplicity, this figure and all fifteen figures below (up to and including) Figure 4h The reference numerals in the accompanying drawings also serve as reference numerals for the switching stages or stage-switching diagrams shown, or for stages based on conventional ZVS and permissible boundary conditions where no other trapezoidal capacitors exist in the surrounding circuitry, for example in Figure 2a At position 202. Therefore, only the capacitance of bridge capacitor 5 causes the flattening of voltage jumps in the output voltage of the bridge branch.

[0247] Figure 3b The current path directly after the turn-off of the high-side switch or second power transistor 200, i.e. at the beginning of the polarity change process when the voltage at the half-bridge center point 206 is transformed from the current supply voltage to ground, is shown. Here, the bridge capacitor 5 has to discharge. The discharge current, which corresponds exactly to the current from the node 206, flows in full size through the PN junction of the emitter-base diode of the upper PNP auxiliary transistor 11, the base of which is tied to the base 203 of the transistor 200, and through the further PN junction of the base-emitter diode of the lower NPN auxiliary transistor 2, which, as required, at the beginning of the polarity change process, leaves the lower drive winding 7 free of any current.

[0248] Since the current from the node 206 directly after the turn-off, i.e. at the beginning of the polarity change process described here, is mostly greater than the current applied in the upper drive winding 17 at this point in time, this larger current is divided in the upper PNP auxiliary transistor 11 into a part which flows through its base into the upper drive winding 17 and a further part which has to be able to flow out of its collector. Therefore, an additional path on the high side is required for this excess current, which is created in that the upper PNP auxiliary transistor 11 is already switched on by the discharge current. The collector of the PNP auxiliary transistor 11 is tied to the emitter 206 of the high-side switch 200 via the flux diode 13, the anode of which is connected to the collector of the PNP auxiliary transistor 11 and its cathode is connected to the reference potential 206 of the emitter or high-side switch 200.

[0249] In the case of a very high current output from the node 206 to the load, the polarity change process can end directly from the switching phase 3b. Then, as shown directly below in Figure 3e , the current flowing in the upper high-side winding 17 is suddenly commutated in the opposite direction onto the low-side drive winding 7 and further out of the node 206 through the base-collector diode of the low-side switch 100. The part of the current from the node 206, which exceeds the current in 7, flows through the simultaneously switched lower reverse diode 101. Due to the parallel and often even monolithically integrated power PN junctions for both current components, the base 103 of the low-side switch 100 is essentially clamped to its reference potential or to ground during this phase, and therefore, the base 203 of the high-side switch 200 is also clamped to the bridge center point 206.

[0250] According to Figure 3fThe switching phase of the low-side switch 100 only occurs point-wise when the absolute value of the current flowing from 206 drops to the value of the current in the lower drive winding 7 and thus switches off the reverse diode 101. The base-emitter voltage of the low-side switch 100 now depends on the current potential of the bridge output 206, but starts to rise due to the just dropped current amount and accordingly rises with the bridge output voltage 206. If this happens simultaneously with the state "bridge capacitor 5 completely discharged", a direct phase transition from 3b to 3f occurs.

[0251] When the base-emitter voltage of the low-side switch 100 directly reaches its turn-on threshold, its base-emitter diode turns on with a current that is just greater than the current from the node 206 in the low-side drive winding 7. Soon, the low-side base-emitter voltage is approximately at its saturation value and the base-emitter voltage of the high-side switch 200 becomes negative. The resulting phase circuit diagram is according to Figure 3g However, the process described further below does not apply to the just described phase transition.

[0252] If the turn-off current is smaller than the current required for a direct transition from the switching phase 3b to 3e or 3f, the polarity change process becomes flatter and longer. Since the load current decreases significantly faster during each polarity change process than the current in the drive winding 17 due to its significantly higher ohmic component, the voltage slope of each polarity change process of the node 206 also decreases during its run and the current through the bridge capacitor accordingly decreases.

[0253] The point shown in Figure 3c is reached soon, in which the discharge current through the bridge capacitor 5 or the current flowing from the node 206 just corresponds to the current imposed in the drive winding 17. The collector current in the upper PNP auxiliary transistor 11 dries up.

[0254] The phase transition from 3c to 3f is not possible, because both switching phases describe a point-like situation, respectively, which explicitly fixes the differentiating factor "the voltage on the bridge capacitor 5 is either greater than zero or equal to zero".

[0255] Figure 3d The state directly after the balancing of Figure 3c is shown. The discharge current through the bridge capacitor 5 becomes too small, i.e. smaller than the current imposed in the high-side drive winding 17 until the end. Instead of now painstakingly providing a path anti-parallel to the control input of the just switched-off second power transistor for the current difference formed, it is explained that with the above already described effect of the reverse coupling, i.e. the "becoming too small negative" current in the high-side drive winding 17 establishes a corresponding positive current in the low-side drive winding 7.

[0256] To this end, a new path is opened next to the base 103 of the low-side switch 100, which places the low-side switch or its driver into the already mentioned waiting position. Only then can the current in 17 be further reduced according to the value predefined by the bridge capacitor 5, and the new positive current in the low-side drive winding 7 can be correspondingly increased, the sum of both corresponding to the current in 17 in 3c or 3c. Figure 3b To this end, the hitherto PN junction on the low side of the bridge capacitor 5 is expanded to the base-emitter diode of the lower NPN auxiliary transistor 2, which has thus been on since the switch phase 3b (thus in there and in 3c marked in dashed lines) and the collector of the lower NPN auxiliary transistor is coupled with the base 103 of the low-side switch 100 to derive the excess or advanced positive current in the low-side drive winding 7. To this end, the flux diode 4 is also connected in series here. Its anode is connected to the low-side base 103, and its cathode is connected to the collector of the NPN auxiliary transistor 2. Figure 3c

[0257] Figure 3g The current path is shown, starting from the switch phase 3d, which results after the low-side switch 100 has been switched on after the bridge capacitor 5 has been completely discharged. Since the high-side drive winding 17 now becomes currentless, the current in the low-side drive winding 7 jumps to the value applied by the high-side drive winding 17 during phases 3b and 3c with its previously directly discharged current and this value is greater than the current that is currently also output from the node 206 to the load, which flows through the base-collector diode of the low-side switch 100. The rest of the current now flows through the base-emitter diode of the low-side switch 100 and switches this low-side switch on without losses.

[0258] In other words, according to Figure 3d The current that also flows through the bridge capacitor 5, i.e. the load current, now flows through the base-collector diode of the low-side switch 100, and the current that has flowed through the lower drive winding 7 there is now flowing through the base-emitter diode of the low-side switch 100, the base voltage of which thus rises approximately to its saturation value and causes the base voltage of the high-side switch 200 to become correspondingly negative.

[0259] The replacement transition from phase 3f onwards has already been described above.

[0260] Shortly thereafter, the load current changes its sign at the node 206 and the state with the just switched-on low-side switch 100 shown in 3g occurs. Figure 3h

[0261] Since the respective or complementary Figures 3a to 3h ​​some of the symmetries and also to illustrate the return path, the opposite load current direction and the polarity change from ground to positive compared to above, the following eight Figures 4a to 4h arranged in the opposite order, i.e. from the lower right to the upper left on its page.

[0262] Figure 4a The state directly before the first bipolar power transistor 100 on the low side is switched off is shown. The normally no longer maximum collector current of the first bipolar power transistor 100 in the load node 206 is divided in the power transistor into a positive emitter current and an absolute value substantially identical negative base current, which is received by the lower drive winding 7. In this way, the memory charge of the base-emitter diode of the power transistor 100 is optimally cleared and the power transistor 100 can be switched off with low losses.

[0263] The left-hand Figure 4b The current path directly after the lower bridge transistor or the first power transistor 100 is switched off, i.e. at the start of the polarity change from ground to positive of the voltage at the half-bridge midpoint 206, is shown. Here, the bridge capacitor 5 has to be charged. The charging current required for this and essentially applied by the load flows through the PN junction of the base-emitter diode of the upper NPN auxiliary transistor 12 directly from the load node 206 into the emitter of the lower PNP auxiliary transistor 1. Directly after the switch-off, i.e. at the start of the polarity change described here, the current flowing out of the load into the node 206 is mostly greater than the current applied to the lower drive winding 7 at this point in time. This greater current in the lower PNP auxiliary transistor 1 is therefore divided into a part, which is subtracted from the current in the lower drive winding 7 by its base, and another part, which has to be able to flow out by its collector.

[0264] An additional path for this excess current, which has been generated by the discharge current switching on of the lower PNP auxiliary transistor 1, is therefore required on the low side. The collector of the lower PNP auxiliary transistor 1 is connected to ground via the flux diode 3, the anode of which is connected to the collector of the PNP auxiliary transistor 1 and the cathode of which is connected to the emitter of the low-side switch 100 or to ground.

[0265] Directly above it Figure 4eThe switching phase shown in the middle is produced by phase 4b, where the collector current that is switched off there is so high that when the bridge capacitor 5 is fully charged, this current is still higher than the negative current that is essentially constant during the preceding switching phase in the low-side drive winding 7. Because the high-side reverse diode 201 is just switched on with this current excess, because the current that flowed essentially constantly in the negative direction through the low-side drive winding 7 before is now commutated onto the high-side drive winding 17 with the same magnitude but in the positive direction, and now flows through the base-collector diode of the high-side switch 200. As in switching phase 3e, during this phase the two base-emitter voltages of the power transistors are also clamped to zero, the only difference being that here the two power PN junctions on the high side cause the clamping.

[0266] As soon as the absolute value of the current in 206 falls to the value of the current in 17, the switching phase according to Figure 4f is formed on its left side. The reverse diode 201 is then switched off. If this occurs simultaneously with the state "bridge capacitor 5 fully charged", a phase transition from 4b directly to 4f is produced.

[0267] This pure point-like state is immediately replaced by a loop current through the base-collector diode of the high-side switch 200, which supplements the decreasing current in node 206 to the magnitude of the current applied in 17. The base-emitter voltage of the high-side switch 200 approximately reaches its saturation value, and the base-emitter voltage of the low-side switch 100 becomes negative. The phase circuit diagram according to Figure 4g is obtained, which, however, is not applicable to the phase transition just described as a process as will be described further below.

[0268] Otherwise, i.e. when the load current becomes smaller than the current applied into the drive winding 7 during the charging of the bridge capacitor 5, Figure 4b the switching phase transitions into the switching phase shown in the left-hand adjacent Figure 4c . Because the current through the bridge capacitor 5 corresponds to the decreasing load current. The point shown in Figure 4c is reached soon, in which the charging current through the bridge capacitor 5 just corresponds to the current applied in the drive winding 7. The collector current in the lower PNP auxiliary transistor 1 then dries up.

[0269] Because there cannot be a direct transition between the two point-like states, which are distinguished by the criterion "bridge capacitor 5 fully charged, yes or no" that is just fixed as a result, there is also no direct transition from the switching phase 4c to the switching phase 4f here.

[0270] Figure 4d is shown directly in Figure 4cthe state after the balancing of the bridge capacitor 5. The charging current through the bridge capacitor 5, as it has been applied by the load at the node 206 before, becomes too small, i.e. smaller than the current that is applied in the low-side drive winding 7 until the end. Instead of bothering to provide a path in parallel to the control input of the just turned off first power transistor for the current difference that is formed in the low side, the effect of the reverse coupling that has been described above is exploited, i.e. the "becoming too small negative" current in the low-side drive winding 7 builds up a corresponding positive current in the high-side drive winding 17. For this purpose, a new path is opened next to the base 203 of the high-side switch 200, which places the low-side switch or its driver into the already mentioned waiting position.

[0271] For this purpose, the hitherto PN junction on the high side of the bridge capacitor 5 is extended to the base-emitter diode of the upper NPN auxiliary transistor 12, which is thus turned on (thus marked in dashed lines there and in Figure 4b from the state shown in Figure 4c and the collector of the upper NPN auxiliary transistor is coupled to the base 203 of the high-side switch 200 to lead out the excess or advanced positive current in the high-side drive winding 17. For this purpose, the flux diode 14 is also connected in series here. Its anode is now connected to the high-side base 203, and its cathode is connected to the collector of the NPN auxiliary transistor 12.

[0272] directly above it Figure 4g depicts the current path, which results from the case according to Figure 4d after the bridge capacitor 5 has been fully charged to the current supply voltage, after the high-side switch 200 is turned on. Since the low-side drive winding 7 now becomes currentless, the current in the high-side drive winding 17 jumps to the value that the low-side winding 7 applied during the phases 4b and 4c, which is greater than the load current that is still entering through the node 206 at the moment. This load current has to bypass the upper drive winding 17 in order to flow through the base-collector diode of the high-side switch 200. Because the rest of the current now flows through the base-emitter diode of the high-side switch 200 in the circuit and turns this high-side switch on without losses.

[0273] In other words, the current that has flowed through the bridge capacitor 5 before now flows through the base-collector diode of the high-side switch 200, and the current that has flowed through the upper drive winding 17 before now flows through the base-emitter diode of the high-side switch 200 in the loop.

[0274] The further transition from the phase 4f to the switching phase has been described above.

[0275] Shortly thereafter, the load current changes its sign at the node 206 and appears Figure 4h with the just switched-on high-side switch 200 as shown in the state Figure 2b and 2c The possible time curves of some of the just explained quantities of the phase sequence 3g-3h-4a-4b-4c-4d-4g are shown. Furthermore, it can be seen from Figure 3b , 3c 3d or 4b, 4c and 4d that the recharging of the bridge capacitor 5, which causes the voltage rise (Spanungshub) at the bridge midpoint 206, is always defined only by the load current at the node 206. The current stored in the control winding 7 or 17 or in the possibly parallel connected inductance at the switching-off time point or at the beginning of the polarity change process therefore does not help in the polarity change at the bridge midpoint after the switching-off of one of the power transistors.

[0276] The above-mentioned phases 3d or 4d, however, can also end in a different way than by switching on the further bridge half as described above. For this purpose, the critical point is the base current for the respective auxiliary transistor 2 or 12, which must exist and be positive for these phases and which is obtained from the instantaneous load current by subtracting the just newly formed current in the one control winding, which is in the waiting position, respectively. If the load current happens to correspond to half of the current stored directly in one of the drive windings at the last switching-off time point of the bridge transistors, a new special case "load current = current in upper drive winding = current through bridge capacitor = current in lower drive winding" arises. For the said critical base current, there is no surplus, and a further reduced load current, i.e. which is smaller than half of the current previously stored in the drive winding, switches off the previously activated auxiliary transistor 2 or 12, respectively.

[0277] This happens independently of the charging state of the bridge capacitor 5, i.e. when its voltage is not yet positive or zero, even if the load current surplus is even still sufficient for its final charging or discharging. A non-ZVS due to the too short duration of the waiting position is therefore superfluous. Conversely, this problem is also solved if the flux diode 4 or 14 is replaced by a small amplifier. Since the auxiliary transistor 2 or 12 also provides a relatively small control current for this amplifier, the above-mentioned critical base current can be almost arbitrarily small. The auxiliary transistor 2 or 12 then actually only has a high impedance as required for a perfect ZVS when the charging current through the bridge capacitor 5 is really zero.

[0278] In addition to the first principle "auxiliary transistor in the basic circuit" as can be seen from the present disclosure, a second principle "merging of diode and matching transistor" is applied for this purpose. In other words, the diode 4 or 14 constitutes the emitter-base path of the respective second auxiliary transistor in PNP polarity, the newly added collector of which is connected to the respective reference potential, for example in Figure 9j and 9l is shown. The initial flux diode cannot be connected to the respective reference potential. The collector-emitter path of the second auxiliary transistor forms the sought amplifier for the drive current in the waiting position, which is driven by the collector current of the respective first auxiliary transistor 2 or 12 of the second semiconductor chain only, so that its base current is almost completely unloaded from the above-mentioned subtraction as required. The bridge branch improved as proposed here can therefore also utilize its theoretically existing operating area significantly better in practice, i.e. oscillation and inversion take place entirely within a significantly larger operating window, or even particularly low-loss and low-disturbance ZVS is achieved here Figure 2b , 2c or 2d.

[0279] Figure 5a As a first embodiment, a functional self-oscillating bridge branch 900 is shown. The same circuit topology 600 or 700 of the first connection point 6 or 16 occurs on both reference potentials 400 and 206 of the driver, the emitters of the complementary auxiliary transistors 1 + 2 or 11 + 12 are connected to the first connection point 6 or 16, the complementary auxiliary transistors 1 + 2 or 11 + 12 operate in the double basic circuit. The connection points 6 and 16 are directly coupled by the capacitor 5, which is why the components located at this location are called "bridge capacitor". Thus, the two separate drive circuits 600 and 700 form the entire bridge circuit 800, with which the considered bridge branch is equipped.

[0280] The bridge capacitor 5 only precisely guides the current in the case where both drive circuits have to be activated at the same time, respectively, and also approximately in the size of the current to be processed. The basic circuit is therefore sufficient, since an amplification is usually not required. As a particular advantage of this arrangement, the current for activating both drive circuits 600 and 700 is always the same size, respectively. For all triggers, respectively, the turn-off process of one of the power transistors 100 or 200 forming the bridge branch, which is based on the respective signal actually driven, is here represented by the drive winding 7 and 17. Subsequently, the polarity change process of the midpoint 206 of the considered bridge branch, during which both of its drive circuits 600 and 700 should be activated in the ideal case in order to ensure the required duty gap for this and, in particular, to place the respective power transistor to be switched on in the waiting position.

[0281] The mandatory PN junctions or flux diodes 3, 4, 13 and 14 in the flow direction to all the collectors of the total of at least four auxiliary transistors 1, 2, 11 and 12 of each bridge leg remove the control of the respective further or opposite bridge half. In other words, all four PN junctions of the flux diodes are oriented opposite to the PN junctions of the four collector diodes, which are actually "NP junctions" according to the same observation direction, and are the first PN junctions of all four semiconductor chains participating in the entire bridge circuit 800, respectively, when observed in the known observation direction. The PN junctions of the four flux diodes form the respective second PN junctions.

[0282] The chain consisting of four identically oriented diode symbols, which takes both base-emitter symbols into account, is thus obtained as a new characteristic circuit diagram, the observation starting from the base 103 or 203 of the respective power transistor 100 or 200 in the flow direction and ending at the emitter or reference potential 400 or 206 of the same transistor. The above-mentioned oppositely oriented collector diodes are not shown. In the middle of the chain, the first connection point 6 or 16 is connected to one pole of the bridge capacitor 5, respectively. More generally, each of the drive circuits 600 and 700 is activated by applying a current to its first connection point 6 or 16.

[0283] The known protection diodes 104 and 204 are likewise shown at the node 300 between the measurement winding 27 of the drive transformer and the load. As long as the collector current in the respective power transistor 100 or 200 is still negative, they significantly contribute to the control of the two half-bridge control voltages 103 to 400 and 203 to 206 after the respective switching on of the subsequent bridge half. In Figure 6 and Figure 7 these protection diodes are active, whereas in Figure 8 these protection diodes are inactive. If these protection diodes are present, the integrated or additionally parallel connected freewheel diodes 101 and 201 can be dispensed with directly on the half-bridge. An alternative solution is proposed in the figure series 9 (part) and 10, which makes the protection diodes 104 and 204 largely superfluous and the reverse diodes 101 and 201 mandatory again.

[0284] Furthermore, four possible positions 102, 105, 202 and 205 of ladder capacitors are depicted, whose capacitances, if connected, are all added to the capacitance of the mandatory bridge capacitor 5 for the edge flattening on each polarity change process at the bridge midpoint 206, but do not activate the drive circuits 600 and 700, respectively. If there are no active capacitors, at least one of the above-mentioned positions 102, 105, 202 and 205 is occupied by a corresponding capacitor. The so-called ladder capacitances and bridge capacitances together form the total ladder capacitance. Conversely, in the case of a small total ladder capacitance, the bridge capacitance dominates, because otherwise the bridge circuit cannot work, and the additional ladder capacitance can be zero.

[0285] The bridge capacitor 5 functions globally parallel to the low side of the half-bridge, i.e. between its midpoint 206 and ground 400. Therefore, in the case of a limited total ladder capacitance, it is advantageous to provide an additional ladder capacitor only on the high side position 202 or 205, because then the total ladder capacitance can be distributed between positive 300 and ground 400, which is advantageous for radio interference filtering. In order to further optimize the switching behavior of the half-bridge transistors, it is also possible to provide both ladder capacitors 202 and 205 at the same time, thereby having a correspondingly smaller individual capacitance. The most important ladder capacitor is located in position 202. Its capacitance can be up to three times the capacitance of the bridge capacitor 5, especially in the case of high bridge power. In the case of a total ladder capacitance greater than twice the bridge capacitance, in the ideal case two additional ladder capacitors are provided, and the ladder capacitor on position 202 is respectively larger than the ladder capacitor on position 102 by the bridge capacitance, in order to distribute its effect evenly over the bridge branch for the sake of simplifying radio interference filtering. In the case of a total ladder capacitance equal to twice the bridge capacitance, it is particularly advantageous to provide a ladder capacitor on position 202, which has the same capacitance as the bridge capacitor.

[0286] Finally, an advantageous implementation of the start-up circuit and the mute circuit is shown, which consists of a pull-up resistor 81, a buffer capacitor 82, a DIAC or similar pulse-providing element or other trigger element 83, a mute transistor 84 and an emitter resistor or protection resistor 85: its function is described in EP 2 298 036 B1 as already mentioned above. There, R6 (here 85) is particularly advantageously not connected in series with the base of the mute transistor T3 (here 84), but in series with its emitter with significantly lower impedance. By this displacement of the resistor 85, the circuit becomes independent of the current amplification of the transistor 84, at least the output voltage of the low-side driver obtains a small attenuation step by step, and the transistor 84 is better protected.

[0287] Figure 5bA functional self-oscillating half-bridge inverter 910 is shown as a second embodiment. The only difference to Figure 5a the first embodiment is the activation of the drive circuit, which here is done separately by two activation capacitors 8 and 18. In this additional way, the two separate drive circuits 610 and 710 here complete the entire bridge circuit 810 for the bridge leg. Because the connection points 6 and 16 are no longer directly capacitively bridged, but are only also coupled to the collector potential or operating potential 206 and 500, respectively, located thereon, this different designation is required, which indicates that the two capacitors 8 and 18 equally activate the drive circuits 610 and 710 and thus stand out with respect to the remaining ladder capacitors.

[0288] The two activation capacitors 8 and 18 must have essentially the same capacitance, which in turn is similar to half the capacitance of the bridge capacitor 5 of the similar structure 900, and should not differ by more than 10% from the respective further activation capacitor. Thus, the bridge capacitor itself is already symmetrical on the bridge leg in the form of the two activation capacitors, since one half (18) acts in parallel to the high side and the other half (8) acts in parallel to the low side. This variant is meaningful in the case of a forced symmetry and a relatively small allowed total ladder capacitance, i.e. in the case that no further ladder capacitor should be installed at all on the positions 102, 105, 202 or 205, or in the case of high power and thus required total symmetry, where additional ladder capacitors can even be installed at all positions 102, 105, 202 and 205 at the same time. Similar to Figure 5a the first embodiment, the positions 102 and 202 are the more important positions here. In contrast to the low-side positions 102 and 105 in the first embodiment, Figure 5a the preference of the high-side positions 202 and 205 is cancelled here.

[0289] At very high powers, the intrinsic magnetization of the toroidal magnetic core plus the current in its parallel inductance is reduced with respect to the collector current to be switched off, so that the ladder capacitance can be up to five times as large as the total capacitance of the two activation capacitors.

[0290] Of course, it is also possible for two activation capacitors 8, 18 of the same size to occur in the same circuit under consideration and at the same time for a bridge capacitor 5 to occur (not shown). This is meaningful in the case of a relatively small allowed total ladder capacitance and a high necessary activation capacitance, i.e. the capacitance of the bridge capacitor 5 plus the capacitance of the two activation capacitors 8 and 18, while there is a high requirement for symmetry of the capacitances on the bridge leg under consideration. In order to perfectly complete this symmetry, it is also necessary for the ladder capacitor on the position 202 or 205 to have the capacitance of the bridge capacitor 5.

[0291] Figure 6The voltage and current curves on the lower bipolar power transistor 100 of the OSRAM company's OT FIT 30 / 220-240 / 700 CS, which has a two-armed voltage-guided passive power factor correction circuit consisting only of diodes and capacitors, and which, moreover, works according to the principles of EP 1705 961 A2 or EP 2 111 730 B1, which, like the power factor correction and drive transformer structure, is not the subject of the present invention, and which, therefore, is only briefly described as follows: The half-bridge with flux driver regulates its frequency by periodically short-circuiting the winding with 18 turns in such a way that the drive transformer, with a turns ratio of 6:1 :6:18, is only partially and controllably saturated for switching off the just conducting power transistor, whereby the bridge frequency can be increased and, thus, the transmitted power can be reduced.

[0292] The only change compared to the original device is that two 1.5 ohm base series resistors are bridged before the control input of the power transistor, and the ladder capacitance is increased by almost half and is connected directly to the half-bridge midpoint 206. The uppermost curve shows the half-bridge midpoint voltage or collector voltage 206 with respect to the reference potential 400, the slightly lower and, so to speak, in between curve shows the collector current 311, the third curve in the same position shows the base current 321, and the lowermost fourth curve shows the base voltage 103 with respect to the reference potential 400. The slow turn-off of the bipolar transistor can be clearly seen on the edges of the completely non-perpendicular drop of its collector current 311, which lasts almost as long as the polarity change process formed by the tilting of the ladder capacitance of the half-bridge midpoint 206.

[0293] Figure 7 The same device at the same operating point in Figure 6 is shown after the removal of the periodic controller and its replacement by a 120 μΗ inductance in parallel to the 18-turn winding of the drive transformer (both not shown) and after the installation of the two drive circuits 600 and 700 according to the invention in Figure 8 . The illustration of the base current 321 is omitted here and in Figure 5a because the space for the current measuring clamp for measuring the base current is no longer sufficient. The 470 pF ladder capacitance on the position 102 and the 820 pF bridge capacitance on the position 5 complete the circuit. The magnetic core of the drive transformer is a ferrite high-permeability soft magnetic ring core with an outer diameter = 8 mm and a width = 4 mm.

[0294] The fast turn-off of the transistor 100 during the change at 311, and the steady guidance of the base voltage 103 in the non-dangerous voltage band with respect to the reference potential 400 during the polarity change, i.e. during the ramp of the collector voltage 206 with respect to the reference potential 400, can be well seen. The slopes of these ramps are hardly reduced in comparison with the slopes of Figure 6 the ramp 206, despite the increased total trapezoidal capacitance. This shows that, in addition to the current flowing in parallel to the trapezoidal capacitance and the bridge capacitance, a considerable further current must also flow during the falling time of the half-bridge transistor: the collector current. Conversely, this means that, with the same slope of the ramp, the sharper the turn-off process, the higher the total trapezoidal capacitance, which is advantageous.

[0295] Finally, Figure 8 the effectiveness of the space-occupying gap control obtained by the present application is shown, in particular the effectiveness of the obtained switching phase shown in Figure 3d and Figure 4d . At the operating point with the minimum active power, the collector current 311 is even turned off before the maximum of the almost purely reactive resonant load current, so that the above-mentioned voltage ramp 206 with respect to the reference potential 400 has the shape of a sine cut. The effect of the space-occupying gap control is evident in the relatively short section with a base voltage 103 that is not equal to zero compared to the reference potential 400, and the small amount of active power transmitted at this operating point is evident during the same section of the very small collector current 311, which in addition has hardly any DC component.

[0296] In the following figures of series 9, as a possible improvement of the drive circuit according to Figure 1a , a bead choke 9 or a further NPN auxiliary transistor 23 or a further PNP auxiliary transistor 24 or some combination thereof is inserted. The auxiliary transistor 24 (PNP) can be merged with the auxiliary transistor 2 (NPN) to a so-called "second auxiliary thyristor" 22, both the auxiliary transistor 24 (PNP) and the auxiliary transistor 2 (NPN) belonging to the second semiconductor chain. Since in one application the collector of the first auxiliary transistor 2 of the second semiconductor chain can be used differently from the coupling thereof to the control line of the belonging power transistor, there the flux diode 4 or the auxiliary transistor 24 is missing.

[0297] Each such drive circuit connects an actual driver with the control input of the power transistor, which actual driver has to provide the necessary energy and the necessary time information, in particular the time information for the switch-off point in time. For this purpose, a respectively lower connection line 400 connects the driver at least low-ohmically to the reference potential of the power transistor to be driven, and a respectively upper control line 103 connects the second pole of the driver with the control electrode of the associated power transistor.

[0298] As soon as at least one bead choke is inserted into the control line 103, the control line is at least divided into a first part 113 and a second part 123. Here, it depends on the actual situation of the bridge branch to be connected at the moment, which one of the two parts 113 or 123 is better connected with the second pole of the actual driver, and which one of the two parts 113 or 123 is better connected with the control electrode 103 of the power transistor to be driven. In principle, both are possible.

[0299] In the following Figures 9a to 9m , for the sake of clarity, only the respective low-side variant 60* or 61* or 620 or 630 for the drive circuit is shown. The transition to the respective high-side variant 70* or 71* or 720 or 730 occurs according to the logic as from Figures 1a to 1b the transition from the circuit 600 or 610 to the circuit 700 or 710. Here, for the variants with the bridge capacitor 5, respectively, the digit "0" or "2" is on its ten's place, and for the variants with the activation capacitor 8 and 18, the respective digit "1" is on its ten's place.

[0300] Figure 9a The drive circuit 601 or 611 is shown with an additional bead choke 9 at the edge, so that the second part 123 of the control line comprises only one connection node, and the first part 113 additionally comprises a connection with the base of the auxiliary transistor 1 and with the anode of the flux diode 4.

[0301] In Figure 9b , the bead choke according to Figure 9a is moved to the middle of the control line and now between the anode of the flux diode 4 and the base of the auxiliary transistor 1, the anode of the flux diode 4 together with one connection node forming the first part 113 of the control line, and the base of the auxiliary transistor 1 together with another connection node forming the second part 123 of the control line. Feasible in this way is that the current introduced into the control line from the drive circuit 602 or 612 is separated for a short time from the current drawn from the control line by the same drive circuit.

[0302] Figure 9cBy means of the second PN junction of the first semiconductor chain (according to Figure 1a , 9a or the flux diode 3 of the flux diode 13 according to Figure 1b ) at the collector of the first auxiliary transistor 1 of the same semiconductor chain to the further NPN auxiliary transistor 23 and by means of its connection with the control line 103, a path for the Miller current is provided, which is actively switched on by the drive circuit directly during the turn-off process. The collector of the further NPN auxiliary transistor 23 can be well blocked against the gate voltage. The right-hand part of the drive circuit 603 or 613 is triggered by the "pulling of the gate line", i.e. by the actual driving turn-off command of the control line 103, which is almost always electrically negative. The emitter-base diode and the base series diode 21 of the first auxiliary transistor 1 become conductive, triggering the second auxiliary transistor 23 of the first semiconductor chain via the collector of the first auxiliary transistor. This in turn reliably shorts the control input of the power transistor connected thereto and this for as long as the holding current flows through the bridge capacitor and / or the activation capacitor 5 and / or 8. The term holding current is therefore justified, as the wiring of the components 1, 21 and 23 is similar to that of a small thyristor.

[0303] This structure 603 or 613 is particularly suitable for the improved version according to the application of a bridge leg equipped with MOS field effect transistors as power transistors. Thus, the base series diode 21 and the base protection diode 46 are drawn here for the first time. Without the former, a positive gate voltage, which can be much larger than the base-emitter breakdown voltage, would destroy the first auxiliary transistor 1 of the first semiconductor chain when its emitter-base diode has to absorb this much larger voltage at least partially as off voltage.

[0304] Since, as already explained above, the base-emitter diode of the first auxiliary transistor 2 of the second semiconductor chain is partially protected by the functional parallel circuit of the activation capacitor and / or the bridge capacitor, its off voltage remains at a value which is tolerable for the transistor 2, which is much smaller, for example, than the above-mentioned large off voltage, as the bridge capacitor and / or the activation capacitor is larger than the input capacitance of the transistor 1. If the emitter-base diode of the transistor 1 can receive the gate voltage as off voltage, the base series diode 21 can also be dispensed with. Otherwise, the gate voltage is divided between the emitter-base diode and the base series diode 21 of the transistor 1 in the off direction. This halving of the gate voltage to be switched off can also mean too much stress for the emitter-base diode of the transistor 1. In order to have the gate voltage approximately completely switched off by the base series diode 21, the base protection diode 46 is connected antiparallel to the emitter-base diode of the transistor 1 to be protected.

[0305] Lower down appears the lower drive circuit 621, which is not described separately here, since it is derived from Figure 1a the drive circuit 600, if there two diodes 21 and 46 are additionally provided as just described.

[0306] In Figure 9d , the drive circuit 604 or 614 is composed according to Figure 9a and Figure 9c the circuit combination without the diodes 21 and 46. Thus, at the edge, i.e. at the connection node 123, a drive circuit is formed with a small thyristor composed of the auxiliary transistors 1 and 23 and the bead choke 9. The bead choke 9 separates this node from the first part 113 of the control line, to which, in addition to the second connection node, the anode of the flux diode 4 of the first semiconductor chain, the base of the transistor 1 and the collector of the nearest second auxiliary transistor 23 are connected.

[0307] Figure 9e It is shown in the drive circuit 605 or 615 how the bead choke 9 can be enclosed in the small thyristor composed of the auxiliary transistors 1 and 23. Since, by this choke, only a voltage in pulse form occurs, this significantly changes the triggering and especially the extinguishing properties of the thyristor. Starting from Figure 9d for this, the bead choke 9 only has to be moved between the collector of the second auxiliary transistor 23 of the first semiconductor chain and the base of the first auxiliary transistor 1, respectively. Thus, the first part 113 of the control line only has the connections to the anode of the flux diode 4 and to the base of the first auxiliary transistor 1, while the second part 123 of the control line has the connection to the collector of the auxiliary transistor 23.

[0308] In Figure 9f , the small thyristor of the drive circuit 606 or 616 is again chokeless, and the bead choke is located in the control line between the anode of the flux diode 4 and the base of the first auxiliary transistor 1, as in Figure 9b . Thus, the first part 113 of the control line only has the connection to the anode of the flux diode 4, while the second part 123 of the control line has the connections to the base of the first auxiliary transistor 1 and to the collector of the second auxiliary transistor 23 of the same first semiconductor chain.

[0309] Figure 9g is a combination of the previous two figures. The advantage of this configuration 607 or 617 is that from one side of the bead choke 9, i.e. from the first part 113 of the control line, auxiliary currents can only be drawn off, and on the other side, i.e. on the second part 123 of the control line, only fed in correspondingly. And in Figure 9eThe advantages of the bead choke enclosed in the small thyristor described in the middle are utilized at the same time. Thus, the second part 123 of the control line only has the connection to the base of the first auxiliary transistor 1 of the first semiconductor chain additionally, while the first part 113 of the control line is connected to the collector of the second auxiliary transistor 23 of the first semiconductor chain and to the anode of the flux diode 4, respectively, in contrast.

[0310] Figure 9h The open-collector variant of the circuit 603 or 613 is shown in the drive circuit 620 or 630. Figure 9c The flux diode 4 is missing here, which has an anode on the control line 103 and a cathode on the collector of the auxiliary transistor 2. This variant can only be meaningfully used in the case of a bridge branch equipped with MOS field-effect transistors, in particular in the case of such a bridge branch with monostable trigger drive. Thus, the base series diode 21 and the base protection diode 46 are again depicted here.

[0311] In Figure 9j , the second auxiliary transistor 24 in the second semiconductor chain is extended to the drive circuit 608 or 618 according to the drive circuit 605 or 615. Figure 9e If the flux diode 4 there is completed as a PNP transistor, whose emitter is connected to the anode there, whose base is connected to the cathode there, and whose collector is connected where the diode 4 cannot be connected there, i.e. at the respective reference potential, here 400, the second auxiliary transistor 24 of the second semiconductor chain is formed.

[0312] In the description of the previous figures, a first or lower drive circuit 628 occurs, which is formed from the upper drive circuit 608 by a direct connection between the control line parts 113 and 123 instead of the bead choke 9, and the transistor 23 is replaced back into the original flux diode 3 from the collector of the transistor 1 to the reference potential 400, again. An upper or second drive circuit 728 is similarly configured in the same position.

[0313] A further small thyristor is generated from the second auxiliary transistor 24 and the first auxiliary transistor 2 of the same semiconductor chain, which amplifies as desired and in combination with Figure 3d or Figure 4dDescribed occurs between the collector of the auxiliary transistor 2 and the base of the auxiliary transistor 24, the connection associated therewith is moreover not necessarily accessible, and its gate consists as "lower P-layer" of the base of the auxiliary transistor 2 and the collector of the auxiliary transistor 24. Although this gate is connected with the corresponding reference potential (here 400), the further thyristor is triggered classically by a positive gate voltage when the drive circuit 608 or 618 holds the bridge transistor or power transistor to be switched on in the waiting position, i.e. establishes a current path parallel to the control path to the associated power transistor. As soon as this is exactly required, at least a part of the instantaneous load current is taken from the first connection point 6 via the activation capacitor 8 and / or the bridge capacitor 5, so that the emitter potential of the auxiliary transistor 2 is below its base potential, which equals the above positive gate voltage.

[0314] The further small thyristor triggered thereby takes its holding current from the activation capacitor 8 and / or the bridge capacitor 5, and (not typical for thyristors, but typical for all applications shown here) shorts the actual driver between the node 113, the anode of the thyristor or the emitter of the auxiliary transistor 24, and the node 400, its gate. Due to all this, the two auxiliary transistors 2 and 24 here can actually be replaced by a real second auxiliary thyristor 22, resulting in a drive circuit 608' or 618' in which, as shown in Figure 9k , the second auxiliary thyristor 22 is coupled between its anode and its gate to the associated control input.

[0315] In Figure 9l , the drive circuit 609 or 619 is derived from the flux diode 4 of Figure 9g by extending it to a second auxiliary transistor 24 of PNP polarity according to Figure 9j .

[0316] Finally, Figure 9m , the drive circuit 609' or 619' is shown, which is formed from the corresponding "un-dashed" variant of Figure 9l by the fact that there the auxiliary transistors 2 and 24 are also replaced by a second auxiliary thyristor 22 connected according to Figure 9k .

[0317] Figure 10a Corresponding to Figure 5a , instead of the first drive circuit in Figure 9l and Figure 1a , the bridge branch 909 is here improved to a third embodiment with the first drive circuit 609 and the corresponding high-side variant or the second drive circuit 709 according to Figure 1b . Like above, the entire bridge circuit is designated 809.

[0318] Figure 10b corresponding to Figure 10a and 5b . The fourth embodiment 919 thus formed differs from the third embodiment described above only in that instead of the bridge capacitor 5 two activation capacitors 8 and 18 of the same size are used, and thus also the two drive circuits and the entire bridge circuit formed thereby are designated 619, 719 and 819. In Figure 5a and Figure 5b between, for the further ladder capacitors at the possible positions 102, 105, 202 or 205, respectively different preferences also apply with the same differences and associations as Figure 10a and Figure 10b .

[0319] Thus, the drive circuit 609, 619 according to Figure 9l and its corresponding high-side variant 709, 719 is selected here and connected as shown, because then between the respective base 103 or 203 of the power transistor and the bead choke 9 or 19, i.e. by the first part 113 or 213 of the respective control line, through the emitter of the auxiliary transistor 24 or 34 or through the collector of the auxiliary transistor 23 or 33, a current can always only be drawn, i.e. from the respective second auxiliary transistor of the two semiconductor chains, and because then only after the bead choke 9 or 19, i.e. in the second part of the respective control line 123 or 223, a current can be added through the base of the first auxiliary transistor 1 or 11. The optional coupler 29 does not belong to 609, 619 nor to 709, 719, but to the entire bridge circuit 809 in Figure 10a or 819 in Figure 10b , since it only makes sense for a complete bridge leg.

[0320] It should be repeated for the sake of completeness that in Figure 10a and Figure 10bIn the node 300, a load (not shown) is connected, to which a high-frequency alternating voltage is applied, which is advantageously inductively ohmically tuned at all occurring frequencies of the alternating voltage and is connected to at least one pole via at least one DC decoupling capacitor (not shown), the poles coming from the same DC voltage ground 400 and positive 500, which also supplies the bridge branch. If this load should have a resonant behavior, the at least one DC decoupling capacitor can be deliberately reduced in order to form an inherent frequency together with the inductive part of the load. Here, the frequency-reducing effect of the attenuation of this inherent frequency by the real component of the load can be jointly exploited in order to fall below the bridge frequency together with all components connected to the node 300. This is particularly meaningful in the case of the energy supply of LEDs, since such a load becomes nonlinear not only by itself but also due to the rectifier that is then necessarily applied to the above-mentioned alternating voltage. This effect is weakened by the resonant coupling of such a load on the bridge branch. All this applies analogously to the figure series 11, 12 and 13.

[0321] In these third and fourth embodiments 909 and 919 according to Figure 10a and Figure 10b the protection diodes 104 and 204, although shown as built-in, including the possible locations 105 and 205 for further ladder capacitors, are no longer as important in their function as for the bridge branches 900 and 910 according to Figure 5a and Figure 5b Since the bead chokes 9 and 19 recently inserted into the control circuit solve the same problem, only in a different place and in a different way.

[0322] Furthermore, it has been shown in practice that a small inductance in series with the base of a bipolar transistor as power transistor significantly increases its turn-off speed, thereby correspondingly reducing its turn-off losses.

[0323] As can be seen from Figure 3e or Figure 4e , in the case of an ideal reverse diode or freewheel diode 101 or 201, the control voltage will be clamped to zero due to the parallel current flow through the base-collector diode of the power transistor 100 or 200. This is even more true for integrated or intrinsic reverse diodes, since the two power PN junctions used simultaneously and in parallel in these switching phases have the same doping profile and thus form essentially the same flux voltage.

[0324] The problem, however, is the parasitic impedances, in particular the parasitic impedances for the high-frequency processes in series with each reverse diode or freewheel diode 101 and 201, such as the inductance of the bond wires. By virtue of the above-mentioned Figure 3e or Figure 4e These individual or integrated diodes take over parts of the comparatively high current abruptly, resulting in voltage drops over these parasitic impedances. Similar to driving through a pothole, the "working potential is far below zero". The above-mentioned protection diodes 104 and 204 mitigate this effect by conducting a part of the reverse diode conduction current through them.

[0325] The same Figure 3e or Figure 4e The effect of these voltage drops is also shown: these voltage drops also occur almost unattenuated on the drive windings 7 or 17, as they are many times higher in impedance. In the case of the anti-coupled drive windings, the negative voltage peaks, which are generated on the low side at the beginning of the switching phase shown in Figure 3e , are positively transferred to the high side. Thereby, when the first power transistor on the low side just becomes conductive, the high-side switch or the second power transistor can just be conductive again, which can lead to a very dangerous short-time bridge short circuit. Exactly and only at this point in time, the opposite coupling of both drive windings 7 and 17, which otherwise always works without fault for the same bridge leg, acts as an error control. This can also be seen very well in Figure 8 , in the measurement of Figure 8 , where the bead chokes 9+19+29 have not been installed: the base potential 103 of the lower bridge transistor or the first bridge transistor performs a very high-frequency oscillation at the border between the seventh and the eighth time period, the peak of which just falls below the turn-on level of the potential 103, which lies to its left. Just at this point in time, the working potential 206 of the same bridge transistor reaches a positive level, which can be seen at the last upper corner of its time curve, which corresponds to the point in time at which the upper bridge transistor or the second bridge transistor or its reverse diode turns on, thus causing this oscillation. If this peak is about 0.2 V higher, there is a short-time bridge short circuit.

[0326] If now the sub-actual drive, which is represented by the drive windings 7 and 17 in Figure 10a and Figure 10b , is also in series with the control electrodes 103 or 203 of the power transistors, there is an additional impedance, in particular for high frequencies or short times, which is able to filter out this error control, as it only occurs for a very short time. The above-mentioned ferrite beads or bead chokes 9 and 19, i.e. small inductances that are already saturated at low DC currents, are very suitable for this task. The possible coupling 29 of the two ferrite beads 9 and 19 results in a common-mode current choke (Gleich-taktdrossel) in the viewing direction of the two control electrodes of the half-bridge transistors as shown in Figure 10a and Figure 10b , further accelerates the turn-off and perfectly avoids the bridge short circuit.

[0327] To this end, as shown in Figure 10c Fig. 2, two control lines for the same bridge leg pass through the same bead choke, wherein the terminals thereof for the control electrodes 103 and 203 of the two power transistors protrude at the same end side 29 of the bead. The lower control line passes through the ferrite bead to produce a lower bead choke 9. The upper control line passes through the ferrite bead to produce an upper bead choke 19. Since both control lines pass through the same ferrite bead and since both ends to be connected to the control electrodes 103 and 203 of the bridge transistors protrude from the same end side 29 of the common ferrite bead, the desired coupling is obtained.

[0328] Figure 10d A further possibility is shown: a dual-hole magnetic core provides the same orientation but a slightly reduced coupling, wherein one hole 9' is filled by the low-side control line and the other hole 19' is filled by the high-side control line, the ends of both holes protruding from the same end side 29' of the core being connected to the control electrodes 103 and 203 of the power transistors. The ferrite material between the two control lines causes the reduction of the coupling.

[0329] Here, the two control lines must be insulated from each other and from the common bead choke or the common dual-hole magnetic core.

[0330] Figure 10e The effect of the described common-mode coupling 29 between the bead chokes 9 and 19 is shown, which for this purpose can even be threaded through the respective control line at arbitrary locations. Since the voltage on the coupled bead chokes 9 + 19 + 29 as described above is rectified and thus exactly opposite to the voltage on the drive windings 7 and 17, which are exactly oppositely, i.e. inversely, coupled. This can compensate for the short-time false control. The voltage positions are shown directly at the switching-on of the low-side inverse diode 101. Even if the high-voltage side is temporarily "grounded", its potential 206 and 203 and the control inductances 17 and 19 are here shown raised, especially due to the comparability with the lower figure.

[0331] Due to the above-mentioned, albeit small, but particularly inductive impedance in series with the inverse diode 101, the bridge output voltage 206 is exactly slightly below zero, i.e. below ground 400 (not shown for clarity, see above). Thereby, the control potential 103 of the low-side transistor 100 is brought to negative (this is illustrated by the lower position of the point 103 relative to the point 400 in Figure 10e Fig. 3), since the base-collector diode of the transistor 100 is simultaneously conducting, in combination with Figure 3eThis can be seen best. Since the negative voltage peaks of 103 in relation to ground 400 formed here are distributed approximately equally over the impedances 7 and 9, they are compensated quite precisely on the high side by the corresponding impedances 17 and 19, which can be seen at the same height of the points 203 and 206: the upper bridge transistor can no longer be switched on incorrectly, since its control voltage remains steadily at zero between 203 and 206.

[0332] Figure 10f Now in connection with Figure 4e the corresponding Figure 10e , only here the negative voltage peaks of the high-side control potential are shown as the height of the point 203, and the bridge midpoint is shown in relation as the height of the point 206, which, when the high-side reverse diode 201 is switched on, exceeds the pulse potential, and due to the base-collector diode of the same transistor, the upper power transistor base is clamped to this pulse potential at the height of the point 203. The complete symmetry of the proposed arrangement is shown, since now (at the same height of the points 400 and 103 can be seen) the control voltage on the low side remains calmly at zero, as Figure 10e the control voltage on the high side. In this way, a transient bridge short circuit is effectively prevented.

[0333] Figure 11a As part of the disclosure of DE 10 2017 105 560 A1 ("LED phase regulator"), a practical bridge branch with its complete self-oscillating driver is shown, which forms the basis of the fifth embodiment elaborated below in detail. With respect to all that has been said before, the bridge branch is here for the first time equipped with N-channel MOS field-effect transistors as power transistors 99 and 199 (with their belonging gates 103 and 203).

[0334] Due to the non-linearity of the load (not shown) always present in the LED power supply and due to its rectifier (not shown), in each drive instead of the parallel oscillation circuit possible for linear loads and disclosed in EP 781 077 B1 a monostable flip-flop is used as robust timer, whose timer capacitor 55 or 65 advantageously has the same capacitance on both potential sides of the bridge branch in question. As soon as the auxiliary transistor 26 or 36 remote from the power transistor or simply "remote" becomes low-ohmic and thus switches off the auxiliary transistor 23 or 33 closest to the power transistor or simply "proximal", each such driver switches on the power transistor by charging its gate 103 or 203. Immediately the gate is charged by the gate series resistance 54 or 64 closest to the gate of the power transistor. From this point on, the timer capacitor 55 or 65 is discharged by the discharge resistance 53 or 63 directly next to it, until the proximal auxiliary transistor 23 or 33 switches on again and the associated power transistor 99 or 199 switches off again by zeroing of the voltage of its gate 103 or 203.

[0335] In order to enable the monostable flip-flop to work as described, the remote auxiliary transistor 26 or 36 associated with the just switched-on power transistor 99 or 199 must likewise remain switched on during the entire on phase. If the auxiliary transistor switches off earlier, the associated proximal auxiliary transistor 23 or 33 is immediately switched on by the timer capacitor 55 or 65, irrespective of its instantaneous charge, the associated power transistor 99 or 199 is immediately switched off, and the timer capacitor 55 or 65 begins to charge again by the two further pull-up resistors 51 and 52 or 61 and 62. Thus, superimposed on the two timers, the monostable flip-flop limiting the maximum on time, and the switching-off process of the remote auxiliary transistor 26 or 36, which is controlled by the voltage on the control winding 57 or 67 and which can shorten the on time, the control winding 57 or 67 is coupled with the converter inductance, for example in the form of a concentrated implementation of the load inductance 97 or the primary winding 97 of a load transformer.

[0336] As already in the drive transformer with its secondary windings 7 and 17, here the control windings 57 and 67 are also coupled to the converter inductance in reverse orientation and also each so that at the beginning of each on phase there is a positive feedback in one of the bridge transistors.

[0337] Due to the voltage on the control winding 57 or 67, delayed by the low pass filter 58+59 or 68+69, in normal operation, shortly after the turn-off of the power transistor 99 or 199 triggered by the monostable flip-flop, the remote auxiliary transistor 26 or 36 is also turned off again and the timer capacitor 55 or 65 is charged again through the pull-up resistors 51 and 52 or 61 and 62, since their further terminal is clamped to the respective flux voltage with respect to the respective reference potential by the base-emitter diode of the near auxiliary transistor 23 or 33, respectively. In series with the control winding, the feed diode 37 or 47 is directed into the buffer capacitor 25 or 35, which provides the auxiliary supply voltage for each such driver.

[0338] For starting the whole, a flip-flop is no longer required at the first time, but only a slow and high-impedance start-up auxiliary, which here is formed by the support capacitor 88 and its discharge diode 89, among others. Its pull-up resistors here consist of the resistors 61, 62, 63, 51, 52 and 53 and the transistors 33 and 23, which are thus switched on during each initial start-up process, switch the resistors 64 and 54 into the pull-up path and reliably turn off both power transistors. In this way, the low-side driver is raised as a whole, supported by the support capacitor 88 and realizing a galvanic isolation between the control winding 57 and the load inductance 97, until a sufficient level is reached at the gate 103 of the low-side switch 99, which initially switches it on. Thus, the inverter is started, and the start-up auxiliary is bridged by a low-impedance path at high frequency, here by the discharge diode 89 and the support capacitor 88. The periodically conducting low-side switch 99 thus keeps its driver at the correct potential itself.

[0339] The remote auxiliary transistor 26 or 36 is turned on and off by the voltage on the control winding 57 or 67 by a low pass filter consisting of the capacitor 58 or 68 and the resistor 59 or 69. The emitter flux diode 39 or 49 is responsible for the fact that this low pass filter can work more precisely by doubling the reference voltage. It has already been explained above that this low pass filter first realizes the work of the monostable flip-flop, since the switching edges of the remote auxiliary transistor 26 or 36 force the time and direction-synchronous switching edges of the associated power transistor 99 or 199, respectively. The low pass filter is thus also used to ensure an occupation dead time in order to be able to realize an orderly operation of a self-oscillating bridge branch with a monostable flip-flop driver. Due to the double dependency on the low pass filter on the one hand and on the control winding voltage on the other hand, this occupation dead time is difficult to scale.

[0340] Figure 11b It is shown that from the application according to the invention disclosed here Figure 11aPossible extensions to the fifth embodiment 920 in order to make the occupation of the space gap flexible and to match the load situation in real time, respectively, as now claimed. For this purpose the drive circuits 620 and 720 according to Figure 9h are used.

[0341] Since the control of the flux balance in the winding 57 or 67 here does not work due to its coupling with the inductive part 97 of the load, and since the gate 103 or 203 can be actively connected to the respective source 400 or 206 by the near-end auxiliary transistor 23 or 33, and since this is even the normal state due to the discharge resistor 53 or 63, the provision of a current path for the driver of the short-circuited power transistor before the actual on-time of the power transistor can be cancelled. The diode 4 or 14 from the control electrode, for example from the gate 103 or 203 to the collector of the auxiliary transistor 2 or 12, whose base is at the respective reference potential, becomes superfluous. This gives Figure 9h the circuit 620 or 720 as its high-side variant. Two such open-collector circuits are installed into the circuit according to Figure 11a their connection points 6 and 16 are connected to each other by a bridge capacitor 5 and generate the circuit 920 according to Figure 11b Since the waiting time that is mandatory for ZVS is ensured by other means:

[0342] In any case, as mentioned above, the power transistor 99 or 199 is switched off as soon as the associated far-end auxiliary transistor 26 or 36 is also high-impedance. The now free collector of the auxiliary transistor 2 or 12 is therefore connected to the base of the far-end auxiliary transistor 26 or 36, which is decoupled from the above-mentioned low-pass filter by an additional resistor 60 or 70. In this way, the far-end auxiliary transistor, and thus the associated power transistor, remains switched off as soon as the bridge midpoint 206 polarity is changed. The additional resistor 60 or 70 allows the low-pass filter to continue to charge in the meantime, so that the far-end auxiliary transistor 26 or 36 can be switched on immediately after deactivation of the drive circuit 620 and 720, if necessary, and the associated power transistor 99 or 199 can also be switched on.

[0343] The other side of the drive circuit starting from the first connection point 6 or 16 is connected in the above-described manner, i.e. the base of the first auxiliary transistor 1 or 11 is connected via the base series diode 21 or 31 to the control electrode 103 or 203 of the power transistor 99 or 199, while the collector of the same first auxiliary transistor (PNP transistor, respectively) is connected via a PN junction to the reference potential 400 or 206. It is provided here to incorporate the PN junction with the base-emitter diode of the proximal auxiliary transistor 23 or 33, whereby the auxiliary transistor adds the function of a second auxiliary transistor already introduced above into the first semiconductor chain with NPN polarity.

[0344] Thus, a small thyristor is formed by the first auxiliary transistor 1 or 11 and the proximal or second auxiliary transistor 23 or 33, respectively, which is prevented from saturation by the base series diode 21 or 31, which is triggered classically by a positive gate voltage between the lower P layer and its cathode (= base-emitter path of the proximal auxiliary transistor 23 or 33) when switching off the associated power transistor, and which maintains the current flow from the bridge capacitor 5, the two ends 6 and 16 of which are connected to the two emitters of the respective first auxiliary transistor of the drive circuit according to the application. In this way, the thyristor is immediately extinguished after the polarity reversal process just observed has ended, since then no more holding current can flow through the bridge capacitor 5. This small thyristor neutralizes the Miller effect of the decelerated switching off in an ideal manner: it is sufficiently low-impedant itself, and due to its special circuit it is also sufficiently fast.

[0345] The base protection diode 46 or 76 is the only component that does not apply the "difference = 10-rule" between the reference numerals of the lower drive circuit and the upper drive circuit.

[0346] However, the overall only makes sense in the case of a load. From the description of Figure 2a known, the same applies to Figure 5a , Figure 5b , Figure 10a or Figure 10b effective, respectively not shown inductive-ohmic loads with at least one DC decoupling capacitor can here even be coordinated in ohmic-capacitive manner, since the inductive-ohmic load is connected to the node 301 differently than above. Since on its other side there is already the necessary load inductance 97, since it is necessary for obtaining the drive voltage. The same applies to the following Figure 12a and Figure 12b .

[0347] In Figure 12aTwo such small thyristors are used in the middle, which, as above, are each composed of an interconnection of two transistors, in order to constitute a current-regulated self-oscillator from the bridge leg equipped with MOS field-effect transistors, as shown in DE 10 2009 042433 B4 ("ultrafast auxiliary thyristor"). Here, as there, the current measurement resistors 10 or 20 in series with the internal reference potential 109 or 209 of the two power transistors replace the hitherto known timers. The small thyristors composed of the transistors 1 and 23 or 11 and 33 are triggered by the voltage on these measurement resistors 10 or 20 in order to switch off the respective power transistor. Now, as an additional task, these small thyristors are to ensure the respective optimum on-space gap.

[0348] For this purpose, the circuit just explained is retrofitted with the drive circuits 603 and 703 according to Figure 9c and thus forms the sixth embodiment 903 according to Figure 12b For each driver in Figure 12a , although a holding current is provided for the small thyristor, but the resistor 53 or 63, which ultimately discharges the gate, is replaced by a direct connection of the P-emitter of the thyristor, i.e. the emitter of the actually first auxiliary transistor 1 or 11 of the first semiconductor chain, to the bridge capacitor 5, according to the invention the bridge capacitor must provide the holding current. The upper N-layer of each such small thyristor corresponds to the collector of the auxiliary transistor 23 or 33 and must therefore be connected to the control electrode, i.e. the gate 103 or 203 of the power transistor 99 or 199.

[0349] For this purpose, the gate lines 103 and 203 are not connected between the components 54 and 37 or 64 and 66, as in Figure 12a , but directly to the collector of the auxiliary transistor 23 or 33, which is already present in the small thyristor. The base of the associated first auxiliary transistor 1 or 11 is also connected to the gate lines 103 and 203, respectively, via the base series diodes 21 and 31, as shown in Figure 12b . The pull-up resistors 54 or 64 of each such small thyristor, initially according to Figure 12a , become more ohmic from the control potential to the upper N-layer and assume the entire feed from the gate driver and drive circuit of the control winding 57 or 67 in the form of a gate series resistor 54 or 64 without a corresponding timer capacitor in Figure 12b , the voltage of which is unidirectionally rectified by the feed diode 37 or 47, the current is limited by the feed resistor 56 or 66 and is slightly buffered in the capacitor 25 or 35.

[0350] Finally, to ensure the required occupation of the space gap, the drive circuit is completed as described above. The emitter of the complementary first auxiliary transistor 2 or 12 of the second semiconductor chain is connected to the P layer of the upper part of the small thyristor, i.e. to the emitter of the first auxiliary transistor 1 or 11, which forms the mandatory first connection point. The base of this auxiliary transistor 2 or 12, which is an NPN transistor, respectively, is at the reference potential 400 or 206, and the anode of the flux diode 4 or 14, which has the cathode at the gate 103 or 203 of the corresponding power transistor 99 or 199, connects this gate with the collector of the first auxiliary transistor 2 or 12 of the second semiconductor chain.

[0351] The feed diode 37 or 47 blocks the negative voltage of the control winding 57 or 67, so the voltage over the entire gate driver (indicated by the voltage over the buffer capacitor 25 or 35) never becomes negative. Therefore, the flux diode 4 or 14 can be dispensed with here and replaced by a direct connection between the gate 103 or 203 and the collector of the auxiliary transistor 2 or 12 (not shown).

[0352] The original form of the strong trigger circuit is shown in Figure 12a . It consists of the pull-up resistor 81, the buffer capacitor 82, the DIAC or similar trigger element 83, the protection resistor 85 and the discharge diode 89 already mentioned at the beginning. The crux of this circuit is that, just when the buffer capacitor 82 should actually output its energy to the trigger element 83, it is additionally discharged through the circuit in order to prevent a new trigger pulse. Therefore, the capacitance of the buffer capacitor 82 must be selected unreasonably large.

[0353] Without the diode 89, the buffer capacitor 82 can be reduced and therefore the resistor 85 can be bridged. The discharge diode 89 is replaced by the advantageous mute circuit already introduced in Figure 5a and employed in Figure 12b , which consists of the mute transistor 84 and the emitter resistor 85, which is previously connected in series with the DIAC 83. Due to the high positive voltage on the control winding 57, the base-emitter diode of the mute transistor 84 must additionally be protected here against breakdown by means of the diode 86, which is connected in parallel to it and in the reverse conduction direction measured at the aforementioned base-emitter diode. The pull-up resistor 81, the buffer capacitor 82 and the trigger element 83 remain in place and function identically.

[0354] Therefore, there is a space duty optimization in real time (periodically), however, the bridge branch is simultaneously current-regulated due to the current measuring resistors 10 or 20 present for each power transistor 99 or 199. Because the triggering of the two small thyristors remains unchanged, these two small thyristors are already present, but are now additionally integrated into the space duty controller.

[0355] As in the widely used prior art, the reference potentials 400 and 206 for the drive circuit remain unchanged, however, the internal reference potentials 109, 209 of the power transistors 99, 100, 199, 200 are not directly connected to the reference potentials 400 and 206, but only via the associated low-impedance current measuring resistors 10 and 20. Otherwise, i.e. when the drive circuit is connected to the internal reference potentials 109 and 209, they would have to process the negative current measuring signals, respectively. The disadvantage of this common approach is that the control voltage is distorted by the current measuring voltage, which can be neglected in the case of relatively high gate voltages in bridge branches equipped with MOS field-effect transistors, but leads to additional current feedback in bridge branches equipped with bipolar transistors.

[0356] The two small thyristors (which consist of the auxiliary transistors 1 and 23 or 11 and 33, which typically have a positive gate voltage) are alternately triggered by the slightly smoothed current measuring voltage connected to the internal reference potentials 109 and 209, i.e. by raising their lower P layer or by switching on the auxiliary transistors 23 or 33. As already in the fifth embodiment of Figure 11b In the sixth embodiment according to Figure 12b In the sixth embodiment according to

[0357] The value of the measuring resistor 10, which is connected to the ground of the overall circuit, can be changed externally in a simple manner (not shown), because its reference potential 400, i.e. the ground, is calm. Therefore, not only can the most important parameter for controlling the power of the bridge branch of the resonant load, i.e. its switching frequency, be changed, but at the same time the duty cycle between the low side and the high side of the same bridge branch can be brought into a skew control (Schiefkontrollung), which increases with increasing frequency. This acts on the transmitted power equally and in the same direction as the switching frequency, in particular when the inverter bridge branch considered is operated on a common supply DC voltage between positive 500 and ground 400 in a half-bridge arrangement with a capacitive passive branch.

[0358] to Figure 13aThe path, which shows the principle circuit diagram 921 of the seventh embodiment, is no longer remote. There it is shown that the bridge leg is completely externally controlled, which is equipped with MOS field effect transistors 99 and 199 as power transistors and at least proportionally stressed by the burden of the resonance (not shown). If the requirements of flexibility or versatility of this part in the overall circuit indicate the need for a fully digital solution, and if at the same time the actual driving of the two power transistors 99 and 199 is successfully implemented directly from the then mandatory IC, alias microcontroller, microprocessor, FPGA or ASIC, then this represents a complete external control. This is possible mainly in the case that the main or bridge transistors 99 and 199 are designed, for example, as N-channel MOS field effect transistors.

[0359] The high requirements, however, also bring with them a wide range of possible duty cycle gaps and, in particular, their particularly poor predictability. A large part of the code required by the IC must be used for duty cycle calculation, and the IC must additionally have several measurement analogues in order to be able to complete this task. However, especially if the subtask of real-time duty cycle determination or optimization is transferred back into analog hardware, a smaller and thus less expensive IC can be installed in the same overall circuit, as this is sufficient for the remaining control and regulation tasks.

[0360] Here, the IC output for driving the bridge leg can even be reduced to a simple alternating voltage source 107, whose voltage rise must be slightly larger than the size of the gate voltage expected in the case of the power transistors 99 or 199 being on, and which outputs a voltage-symmetrical or DC voltage-free square wave signal at the desired frequency and the necessary duty cycle, usually 50:50. The possible zero pauses in between can be cancelled or fixedly set to the minimum necessary or possible value, as the bridge circuit 821 consisting of two drive circuits 621 and 721 inserted as described above ensures compliance with the necessary and respectively ideal duty cycle. If the respective Figure 13a , the base series diode 21 or 31 and the base protection diode 46 or 76 required for the operation of the MOS field effect transistor are installed there, then the drive circuit 621 or 721 used here is formed by the drive circuit from Figure 1a and Figure 1b , i.e. from the drive circuit 600 or 700.

[0361] As already described above, the same bridge circuit 821 also works correctly in the case of the boundary ZVS according to Figure 2d , and even according to Figure 2eThe on-space gap is optimized in the case of non-ZVS, thus maintaining the corresponding suboptimal valley detection mode. Therefore, the bridge branch connected in this way can even be operated at a distance beyond its ZVS boundary without additional code in the IC, as long as the switching losses formed here do not exceed the flux losses formed at the rated load.

[0362] For this purpose, the respective positive voltage on the secondary side 7 or 17 of the 1 : 1 : 1 drive transformer is always short-circuited by the drive circuit 621 or 721 for the standby position of the associated power transistor at the beginning of the respective upcoming conduction phase, the core of the drive transformer can be designed as in the first four embodiments with bipolar power transistors, and the two secondary windings 7 and 17 of the drive transformer are oppositely polarized as drive windings for generating push-pull. In order to prevent the output 107 of the IC from being overloaded or even damaged as a result, at least one impedance, preferably a series resistor 44, must be connected in series with the primary side 27 of the drive transformer.

[0363] The additional feature resulting from the double base circuit of the respective first auxiliary transistor of the two semiconductor chains and the mutual connection of the emitter and base series diode, which clamps each control voltage connected according to the invention to the maximum negative value of the three PN threshold voltages, leads to a feature that, although largely known, proven and used, is also mandatory here. Since the range of the control voltage is between plus and minus one PN threshold voltage in all four previous embodiments with bipolar transistors as power transistors of the bridge branch, such a clamping simply does not occur. Embodiments five and six present a circuit in which, for each driver, the control signal is also evaluated directly, but is rectified unidirectionally for the generation of a positive auxiliary voltage for each drive circuit, respectively. A negative gate voltage is therefore a priori impossible.

[0364] However, here the two drive circuits are coupled to the 1 : 1 : 1 drive transformer with relatively low impedance, which must also provide a likewise low-impedance negative voltage. In order to cut off the negative voltage against the above-mentioned clamping, each drive circuit of the seventh embodiment 921 requires a series or cut-off capacitor 28 or 38, respectively, which reduces the maximum negative voltage occurring on the drive transformer with respect to the three PN threshold voltages. For the same reason, a protection circuit against overvoltage on the gate 103 or 203 is also depicted here for the first time, which consists of a Zener diode or similar 96 or 196, respectively, actually protecting, and a decoupled flux diode 98 or 198, respectively.

[0365] Again, the whole only makes sense in the case of a load. From the description of Figure 2a which is already known from the description of Figure 5a , Figure 5b, Figure 10a or Figure 10b the inductive-ohmic load, not shown separately, with at least one DC decoupling capacitor can even be connected directly to the node 206 here, since the current measurement is mostly implemented differently in the bridge leg equipped with MOS field effect transistors. The same applies to the following Figure 13b .

[0366] However, as shown in Figure 13b , the normal case for the output of the IC 107 is that the control pin 77 of the low side and the similar pin 87 of the high side have a positive logic, i.e. a positive voltage relative to the IC reference potential, when the respective power transistor 99 or 199 is to be switched on and as long as it is to be conducting. This arrangement also allows the output of the dead time gap to be output by the IC itself (two control pins = zero), which can be fixedly set here to a constant and minimally necessary or possible value. Thus a modified implementation 923 of the seventh embodiment of the principle circuit diagram 921 is shown. Figure 13a

[0367] The nominal switching-on time point output by the IC is usually located before the actual switching-on time point of the respective power transistor 99 or 199 which is to be subsequently conducting according to ZVS permission. Because a time must be waited in between until the trapezoidal capacitor 202 and / or the bridge capacitor 5 is charged or discharged therefrom to the voltage positive. In this time, the bridge circuit 803 consisting of the drive circuits 603 and 703 according to Figure 9c short-circuits the two control inputs of the power transistors of the same bridge leg and thus always holds the next power transistor to be switched on in the waiting position. For this purpose, the output consisting of the two pins 77 and 87 can be made more robust by the amplifier 117 and at least one series impedance is provided in order for the output of the amplifier 117 or the output of the IC 107 not to be damaged. Here this is achieved by the gate series resistors 54 and 64.

[0368] The primary winding 27 of the above-mentioned 1 : 1 : 1 drive transformer is connected directly between these pins 77 and 87 or between the corresponding outputs of the amplifier 117. Because a pure alternating voltage is always applied by its drive winding 7 or 17 and because the negative gate voltage is clamped to an absolute value close to zero according to the invention, a bulk capacitor 28 or 38 in series with each drive winding is required. This and the said clamping make the entire device intrinsically suitable for a tilt control of the bridge leg contained therein, wherein for actually conscious tilt control a third bulk capacitor 48 is to be connected in series between the IC outputs 77 and 87 or the corresponding outputs of the amplifier 117 and the primary winding 27 of the 1 : 1 : 1 drive transformer.

[0369] ​If the necessary longitudinal impedance is additionally cleverly transferred to the sides of the two drivers, in particular as gate series resistance 54 or 64 into the connection line of each drive circuit, then the voltage falling thereon can additionally actively trigger according to Figure 9c The small thyristor formed in the drive circuit 603 or 703, which should be conductive at the turn-off point of the associated power transistor 99 or 199. The small thyristor is formed by the first auxiliary transistor 1 or 11, whose PN junction on the collector side extends in the flow direction to the auxiliary transistor 23 or 33, whose collector in turn is on the control potential 103 or 203, respectively. The "gate" of this small thyristor is the otherwise unconnected node on the collector of the first auxiliary transistor 1 or 11 or also on the base of the second auxiliary transistor 23 or 33. By means of the current into it, the transistor 23 or 33 is switched on and thus the small thyristor is triggered classically. By this, it can compensate the Miller effect in real time and in particular supports the power transistor connected in this way at turn-off in the respective overall circuit 923. Since the small thyristor is prevented from saturation by the base series diode 21 and 31, the bridge capacitor 5 is also immediately extinguished when a small holding current flows through it.

[0370] For the respective turn-off, controlled from the outside, the voltage on the associated drive winding 7 or 17 jumps from positive to negative, therefore the series resistance 54 or 64 must be placed into the return line, respectively away from the gate, in order to thereby generate the positive voltage jump. At the nodes between the drive winding 7 or 17 and the series resistance 54 or 64, respectively, the already known decoupling resistance 60 or 70 is connected in order to obtain the positive jump, which, due to its second connection to the base of the auxiliary transistor 23 or 33, respectively, thereby generates the required positive trigger current for the small thyristor.

[0371] The 1 : 1 : 1 drive transformer can also deviate from this uniform turns ratio, i.e. have x: 1 : 1, in order to adjust the control voltage present in the IC or at the amplifier output to the required gate voltage.

[0372] If, as Figure 14aAs shown, the so-called high-side driver 941, which essentially only bridges the potential jumps of the high side of the bridge branch and thus only transfers the control signal applied on its input 87 to its output 203, obtains for the low side a channel which consists of the connection between 77 and 103 and contains the series resistor 54, then in the eighth embodiment the task of the general or direction-independent duty gap control can even be realized decoupled from the actual drivers 77 and 87 by the assembly, for example if one of the disclosed driver circuits 621 and 721 or one of its improvements is integrated there. Here, the lower first connection point 6 and the upper first connection point 16 are formed as new terminals for such an improved high-side driver 941, which are brought out as pins, since there between it should be possible to connect the bridge capacitor 5, in particular preferably. The terminals 77 and 103 for the low-side driver are likewise new, they transform the high-side driver, so to speak, into a bridge branch driver.

[0373] The rest of the circuit required for this is only indicated in a form that is suitable in principle by the American resistance symbols and the unnumbered transistor symbols, and since it is not part of the invention, it does not need to be further explained within the scope of this disclosure. The same applies to the components shown only in dashed lines externally, which are indispensable for the normal function of the high-side driver 941 itself and its duty gap controller. Since the described high-side driver is unipolar powered only between VCC and GND, it cannot generate negative gate voltages. Therefore, here, as already in the Figure 12b above, the flux diodes 4 and 14 can be dispensed with and replaced by a direct connection between the gate 103 or 203 and the collector of the auxiliary transistor 2 or 12 (not shown).

[0374] As shown in Figure 14b , the PNP transistor of the high-side amplifier, which can be designed, for example, as an emitter follower around the series resistor 64, merges with the flux diode 14 into the second auxiliary transistor 34 of the second semiconductor chain, resulting in at least one transistor pair similar to Figure 9jImproved high-side driver 948 of the high-side drive circuit 728 shown there as a low-side variant 608. However, in contrast to all further described variants with an auxiliary transistor 24 or 34, here the auxiliary transistor 23 or 33 can be dispensed with instead of the PN junction or the flux diode 3 or 13, and furthermore here also the flux diode 3 or 13 can be dispensed with by a direct connection (not shown) between the collector of the transistor 1 or 11 and the reference potential 400 or 206, since the gate voltage to be output here also never becomes negative. However, the base series diode 21 or 31 and the base protection diode 46 or 76 are mandatory. If the low-side channel is also to be amplified, then as shown, for this the same low-side drive circuit 628 is provided, wherein here too the auxiliary transistor 24 can form part of the amplifier output stage.

[0375] The amplification of the low-side channel, in particular when it is implemented by a double inversion (not shown) similar to the amplification on the high-side channel, also brings with it the possibility of obtaining a gate voltage of approximately VCC size at the points 103 and 203, which is different from the voltage rise at the inputs 77 and 87.

[0376] List of reference signs

[0377] Low-side drive circuit

[0378] 1: lower PNP auxiliary transistor of the first semiconductor chain or first auxiliary transistor

[0379] 1a: base-collector junction of the transistor 1 or first PN junction of the first semiconductor chain

[0380] 1b: base-emitter junction of the transistor 1 or third PN junction of the first semiconductor chain

[0381] 21: lower base series diode of the transistor 1

[0382] 2: lower NPN auxiliary transistor of the second semiconductor chain or first auxiliary transistor

[0383] 2a: base-collector junction of the transistor 2 or gate-drift region junction in the thyristor 22 or first PN junction of the second semiconductor chain

[0384] 2b: base-emitter junction of the transistor 2 or gate-cathode junction in the thyristor 22 or third PN junction of the second semiconductor chain

[0385] 22: lower second auxiliary thyristor instead of the transistors 2 and 24

[0386] 3: lower flux diode in series with the collector of the auxiliary transistor 1

[0387] 3a: PN junction in flux diode 3 or second PN junction of first semiconductor chain

[0388] 23: Lower second auxiliary transistor (NPN) of first semiconductor chain in place of diode 3

[0389] 23a: Base-emitter junction of transistor 23 or second PN junction of first semiconductor chain

[0390] 23b: Base-collector junction of transistor 23 or fourth PN junction of first semiconductor chain

[0391] 4: Lower flux diode in series with collector of transistor 2

[0392] 4a: PN junction in flux diode 4 or second PN junction of second semiconductor chain

[0393] 24: Lower second auxiliary transistor (PNP) of second semiconductor chain in place of diode 4

[0394] 24a: Emitter-base junction of transistor 24 or anode-drift region junction in thyristor 22 or second PN junction of second semiconductor chain

[0395] 24b: Collector-base junction of transistor 24 or fourth PN junction of second semiconductor chain

[0396] 5: Bridge capacitor

[0397] 6: Lower end or lower first connection point of bridge capacitor

[0398] 46: Lower base protection diode of transistor 1 with cathode at node 6

[0399] 7: Lower secondary winding or drive winding of drive transformer

[0400] 8: Lower activation capacitor

[0401] 9: Lower bead choke (low side)

[0402] 9': Hole for lower control circuit in double-hole magnetic core

[0403] High-side drive circuit

[0404] 11: Upper PNP auxiliary transistor or first auxiliary transistor of first semiconductor chain

[0405] 11a: Base-collector junction of transistor 11 or first PN junction of first semiconductor chain

[0406] 11b: Base-emitter junction of transistor 11 or third PN junction of first semiconductor chain

[0407] 31: upper base series diode of transistor 11

[0408] 12: upper NPN auxiliary transistor of second semiconductor chain or first auxiliary transistor

[0409] 12a: base-collector junction of transistor 12 or first PN junction of second semiconductor chain

[0410] 12b: base-emitter junction of transistor 12 or third PN junction of second semiconductor chain

[0411] 13: upper flux diode in series with collector of auxiliary transistor 11

[0412] 13a: PN junction in flux diode 13 or second PN junction of first semiconductor chain

[0413] 33: upper second auxiliary transistor (NPN) of first semiconductor chain in place of diode 13 or upper proximal auxiliary transistor

[0414] 14: upper flux diode in series with collector of auxiliary transistor 12

[0415] 14a: PN junction in flux diode 14 or second PN junction of second semiconductor chain

[0416] 34: upper second auxiliary transistor (PNP) of second semiconductor chain in place of diode 14

[0417] 16: upper end of bridge capacitor or upper first connection point

[0418] 76: upper base protection diode of transistor 11 with cathode at node 16

[0419] 17: upper secondary winding of drive transformer or drive winding

[0420] 18: upper activation capacitor

[0421] 19: upper bead choke

[0422] 19': upper control circuit hole for dual hole magnetic core

[0423] Other drive components

[0424] 10, 20: emitter resistor or source resistor in series with power transistor

[0425] 25, 35: buffer capacitor for drive auxiliary voltage

[0426] 26, 36: distal auxiliary transistor

[0427] 27: primary winding of the drive transformer or measuring winding

[0428] 28, 38: series capacitor or block capacitor

[0429] 29: coupling between the bead chokes 9 and 19 or end side of the common bead choke facing the control electrodes 103 and 203

[0430] 29': end side of the double-hole magnetic core facing the control electrodes 103 and 203

[0431] 37, 47: feed diode

[0432] 39, 49: emitter flux diode

[0433] 44: series resistor of the external control

[0434] 48: series capacitor or block capacitor in series between the outputs of the external control

[0435] 51, 61: pull-up resistor for the start-up circuit for high impedance

[0436] 52, 62: charging resistor

[0437] 53, 63: discharging resistor

[0438] 54, 64: gate series resistor

[0439] 55, 65: timer capacitor or its capacitance

[0440] 56, 66: feed resistor

[0441] 57, 67: control winding coupled to the discrete load inductance

[0442] 58, 68: delay capacitor

[0443] 59, 69: control resistor

[0444] 60, 70: decoupling resistor

[0445] Start-up circuit and muting circuit or external control

[0446] 81: pull-up resistor

[0447] 82: buffer capacitor

[0448] 83: trigger element, for example DIAC

[0449] 84: muting transistor

[0450] 85: emitter resistor or protection resistor

[0451] 86: Mute transistor protection diode

[0452] 77, 87: Output channel of the external driver or input terminal of the high-side driver provided for this purpose

[0453] 88: Support capacitor

[0454] 89: Discharge diode

[0455] 107: Control IC for external control

[0456] 117: Output amplifier for the control IC 107

[0457] Power components and nodes of the bridge leg

[0458] 96: Low-side gate voltage protection diode

[0459] 97: Discrete load inductance for coupling the control winding

[0460] 98: Low-side gate voltage decoupling diode

[0461] 99: Low-side switch or lower bridge transistor or first power transistor as N-channel MOS field effect transistor

[0462] 100: Low-side switch or lower bridge transistor or first power transistor as NPN power transistor

[0463] 101: Reverse diode or freewheeling diode integrated directly in the transistor 100 or directly in parallel with the collector-emitter path thereof

[0464] 102: Low-side position of the ladder capacitor in parallel with the transistor 99 or 100

[0465] 103: Base terminal or gate terminal of the low-side switch, associated control line, or associated control potential

[0466] 104: Low-side protection diode

[0467] 105: Low-side position of the ladder capacitor in parallel with the protection diode 104

[0468] 109: Internal reference potential, node between the low-side or lower bridge transistor and the current measurement resistor

[0469] 113: First part of the low-side control line

[0470] 123: Second part of the low-side control line

[0471] 196: High-side gate voltage protection diode

[0472] 198: High-side gate voltage decoupling diode

[0473] 199: High-side switch or upper bridge transistor or second power transistor as N-channel MOS field effect transistor

[0474] 200: High-side switch or upper bridge transistor or second power transistor as NPN power transistor

[0475] 201: Reverse diode or freewheeling diode directly integrated in transistor 200 or directly connected in parallel with its collector-emitter path

[0476] 202: High-side position of ladder capacitor in parallel with transistor 199 or 200

[0477] 203: Base terminal or gate terminal of high-side switch, associated control line, or associated control potential

[0478] 204: High-side protection diode

[0479] 205: High-side position of ladder capacitor in parallel with protection diode 204

[0480] 206: Emitter potential or source potential or emitter terminal or source terminal of high-side switch = collector potential or drain potential of low-side switch = internal bridge midpoint or its connection line or its potential

[0481] 209: Internal reference potential, node between high-side or upper bridge transistor and current measurement resistor

[0482] 213: First part of high-side control line

[0483] 223: Second part of high-side control line

[0484] Load node or load current and currents associated therewith

[0485] 300: Inverter output to load or node between protection diodes 104 and 204

[0486] 301: Inverter output to load, where load inductance belongs to inverter

[0487] 310: Collector current value or drain current value directly at bridge transistor turn-on

[0488] 311: Collector current curve or drain current curve

[0489] 312: Collector current value or drain current value directly before bridge transistor turn-off

[0490] 313: value of the load current at the end of the polarity change phase of the bridge leg

[0491] 320: initial value of the base current after switching on the bridge transistor with the flux driver

[0492] 321: curve of the same base current

[0493] 322: final value of the base current before switching off the bridge transistor with the flux driver

[0494] 323: negative initial value of the base current on the oppositely arranged bridge half

[0495] 330: initial value of the theoretical base current in case of flux driving

[0496] 331: curve of the same theoretical base current

[0497] 332: final value of the theoretical base current

[0498] 333: value of the same theoretical base current at the end of the polarity change phase of the bridge leg

[0499] 342: difference between the theoretical and the actual base current at the bridge transistor switching off point in case of flux driving

[0500] 343: difference between the theoretical and the actual base current at the end of the polarity change phase of the bridge leg or at the bridge transistor switching on point

[0501] 350: typical negative current time area of ZVS after switching on the bridge transistor

[0502] 351: load current

[0503] 421: curve of the base current on the oppositely arranged bridge half

[0504] 400: emitter terminal or (external) source terminal of the low-side switch or lower bridge transistor, reference potential of the low voltage side, relevant connection line of the entire circuit or ground

[0505] 500: direct current positive or positive (DC supply voltage)

[0506] time and time point

[0507] 550: start of the on phase of the power transistor on one half of the bridge leg,

[0508] 551: on time or on duration of the same power transistor

[0509] 552: switching off point of the same power transistor

[0510] 553: start of the subsequent conduction phase of the power transistor on the other half of the bridge leg

[0511] 554: wait time or space gap or dead time

[0512] component

[0513] 600, 610,

[0514] 601, 611,

[0515] 602, 612,

[0516] 603, 613,

[0517] 604, 614,

[0518] 605, 615,

[0519] 606, 616,

[0520] 607, 617,

[0521] 608, 618, 608', 618',

[0522] 609, 619, 609', 619',

[0523] 620, 630,

[0524] 621, 628: drive circuit, low side

[0525] 700, 710,

[0526] 709, 719,

[0527] 703, 720,

[0528] 721, 728: drive circuit, high side

[0529] 800, 810,

[0530] 809, 819,

[0531] 803, 821: entire drive circuit for the bridge leg it is matched with

[0532] entire circuit

[0533] 900, 910,

[0534] 909, 919,

[0535] 920, 903,

[0536] 921, 923: entire half-bridge inverter in chronological order

[0537] 941, 948: high-side driver with integrated footprint gap control

Claims

1. A drive circuit for a control input between a control electrode (103, 203) and a reference electrode (400, 206) of a power transistor of a clocked converter, the drive circuit being coupled in parallel to the control input, wherein, For the purpose of energy transmission, the power transistor is clocked on or off by means of a control voltage, and wherein between switching processes a waiting time is provided, characterized in that the drive circuit comprises a first semiconductor chain (1; 11) and a second semiconductor chain (2; 12), wherein a semiconductor chain is a series circuit with at least two PN junctions of opposite polarity, wherein the N region of the first PN junction (1a) of the first semiconductor chain (1; 11) is connected to a control electrode (103, 203) of the control input, which first semiconductor chain assists in switching off the power transistor by short-circuiting its control input, while the P region of the first PN junction (2a) of the second semiconductor chain (2; 12) is connected to a reference electrode (400, 206), which second semiconductor chain in the same way generates a waiting time before the power transistor is switched on.

2. The drive circuit according to claim 1, characterized in that, For triggering the drive circuit, the P region of the third PN junction (1b) of the first semiconductor chain or the N region of the third PN junction (2b) of the second semiconductor chain is provided, wherein the third PN junction (1b) of the first semiconductor chain and the third PN junction (2b) of the second semiconductor chain share the N region of the control electrode (103, 203) or the P region of the reference electrode (400, 206), and the P region of the third PN junction (1b) of the first semiconductor chain and the N region of the third PN junction (2b) of the second semiconductor chain are coupled to each other in a first connection point (6, 16), and in which first connection point a current for triggering is applied.

3. The drive circuit according to claim 2, characterized in that, For triggering the drive circuit, a bridge capacitor (5) and / or an activation capacitor (8, 18) is connected to the first connection point (6, 16) in order to provide the required current.

4. The drive circuit according to any one of claims 1 to 3, characterized by, Between the control electrode and the reference electrode of the control input, each semiconductor chain forms a first PN junction and a second PN junction, respectively, in order to achieve both control voltage polarities, wherein the first PN junction and the second PN junction are oppositely polarized.

5. The drive circuit according to claim 4, characterized in that, The second PN junction is used to keep the control input high-impedance, respectively, in order to enable switching on of the power transistor.

6. The drive circuit according to claim 4, characterized by The second PN junction (3a, 23a) of the first semiconductor chain shares a P region with the fourth PN junction (23b) of the first semiconductor chain, which is coupled to the P region of the first PN junction, wherein the fourth PN junction actively supports switching off of the power transistor.

7. The drive circuit according to claim 4, characterized by The second PN junction (4a, 24a) of the second semiconductor chain shares an N region with the fourth PN junction (24b) of the second semiconductor chain, which is coupled to the N region of the first PN junction of the second semiconductor chain, wherein the fourth PN junction (24b) of the second semiconductor chain ensures a waiting time even in the case of a minimum current for triggering the drive circuit.

8. A bridge leg having a first power transistor (99, 100) and a second power transistor (199, 200) to be connected, the first and second power transistors being connected in series, wherein, The first power transistor (99, 100) is coupled with a reference potential and the second power transistor (199, 200) is coupled with a supply potential, characterized in that a first drive circuit (600) is coupled in parallel to the control input of the first power transistor (99, 100) and a second drive circuit (700) is coupled in parallel to the control input of the second power transistor (199, 200), wherein the first drive circuit and the second drive circuit are configured according to any one of the preceding claims 1 to 7.

9. The bridge leg of claim 8, wherein, Between the two first connection points (6, 16) of the first drive circuit (600) and the second drive circuit (700) a bridge capacitor (5) is connected.

10. The bridge leg of claim 9, wherein, A ladder capacitor (202) is connected in parallel to the collector-emitter path or the drain-source path of the second power transistor (199, 200).

11. The bridge leg of claim 10, wherein, The capacitance of the ladder capacitor (202) is less than three times the capacitance of the bridge capacitor (5).

12. The bridge leg of claim 8, wherein, Between the first connection point of the first drive circuit (600) and the working electrode of the first power transistor (99, 100) a lower activation capacitor is connected and between the first connection point of the second drive circuit (700) and the working electrode (500) of the second power transistor (199, 200) an upper activation capacitor is connected, wherein the recharge currents of both activation capacitors simultaneously trigger the coupled drive circuits.

13. The bridge leg of claim 12, wherein, The capacitances of both activation capacitors differ by a maximum of 10%.

14. The bridge leg of claim 12, wherein, Further ladder capacitors are connected in parallel to one power transistor and / or to the other power transistor.

15. The bridge leg of claim 14, wherein, The total capacitance of all further ladder capacitors is less than five times the capacitance of both activation capacitors (8, 18).

16. The bridge leg of any one of claims 8 to 15, wherein, Both drive circuits comprise a bead choke (9, 19) for either high-frequency decoupling of the first semiconductor chain from the second semiconductor chain or for high-frequency decoupling of the first PN junction from the fourth PN junction of the first semiconductor chain.

17. The bridge leg of claim 16, wherein, The two bead chokes (9, 19) of both drive circuits are coupled to each other such that the two terminals, which protrude from a common core or from the same end side (29, 29') of a common magnetic bead, are coupled to the control electrodes (103, 203) of the power transistors, respectively.

18. A high-side driver (941, 948) for driving the second power transistor and additionally the first power transistor of a bridge leg according to any one of claims 8 to 17, characterized in that The high-side driver has a drive circuit according to any one of the preceding claims 1 to 7, respectively, wherein terminals are provided for connecting the drive circuit in parallel to the control paths of both power transistors.

19. The high-side driver (941, 948) of claim 18, wherein, The two first connection points (6, 16) of the drive circuit are provided as terminals.

20. A clocked converter having a bridge leg according to any one of claims 8 to 17, characterized in that The clock-controlled converter has an inductance with three windings, wherein the energy converted by the converter flows through one of the windings (27) and two further windings (7, 17) are coupled to a drive circuit according to any one of the preceding claims 1 to 7, respectively, wherein the converter is designed such that the current in each further winding triggers the drive circuit.

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