Motor control device, mechatronic unit, electric vehicle system, motor control method

Abstract

The present invention aims to properly perform motor control at the time of overmodulation. In the motor control device (1) of the present invention, the carrier frequency adjusting section (16) adjusts the carrier frequency (fc) in a manner that changes the voltage phase error (Δθv) that indicates the phase difference of the three-phase voltage command (Vu*, Vv*, Vw*) and the triangular wave signal (Tr). When the modulation ratio (H) corresponding to the ratio of the voltage amplitude of the direct current power supplied from the high voltage battery to the inverter and the alternating current power output from the inverter to the motor exceeds a prescribed value, for example 1.15, the current control section (14) corrects the amplitude and phase of the d-axis voltage command (Vd*) and the q-axis voltage command (Vq*) based on the carrier phase difference (Δθcarr) that indicates the phase of the triangular wave signal (Tr).

Description

Technical Field

[0001] This invention relates to motor control devices, mechatronic units, electric vehicle systems, and motor control methods. Background Technology

[0002] In inverter drive systems that use PWM (Pulse Width Modulation) control to drive motors based on voltage commands, most employ asynchronous PWM control where the carrier frequency is fixed relative to the inverter's variable output frequency. Therefore, when the inverter output frequency increases and the number of output pulses per cycle of the voltage command decreases, the inverter's output error increases. Consequently, synchronous PWM control, which changes the carrier frequency based on the inverter's variable output frequency, is adopted.

[0003] In synchronous PWM control, the following overmodulation PWM control is proposed: in the overmodulation mode where the amplitude of the voltage command becomes larger than the amplitude of the carrier wave of the triangular wave or sawtooth wave, the amplitude of the voltage command is increased nonlinearly, so that the voltage amplitude output from the inverter becomes the desired value (for example, Patent Document 1 below).

[0004] Existing technical documents

[0005] Patent documents

[0006] Patent Document 1: Japanese Patent Application Publication No. 2008-312420 Summary of the Invention

[0007] The problem the invention aims to solve

[0008] The method in Patent Document 1 can handle cases where the phase difference between the carrier and the voltage command is a fixed value. However, when the phase difference is not fixed, the amplitude and phase of the voltage output from the inverter will change depending on the phase difference between the carrier and the voltage command during overmodulation. Therefore, it is known that the method in Patent Document 1 cannot properly implement motor control during overmodulation.

[0009] Technical means to solve the problem

[0010] To solve the above problems, for example, the configuration described in the technical solution is adopted. This application includes various technical means to solve the above problems. As one example, a motor control device of the present invention is connected to a power converter that performs power conversion from DC power to AC power, and controls the drive of an AC motor driven by the AC power. The motor control device includes: a current control unit that generates a voltage command corresponding to a torque command; a carrier generation unit that generates a carrier wave; a carrier frequency adjustment unit that adjusts the frequency of the carrier wave; and a gate signal generation unit that uses the carrier wave to pulse-width modulate the voltage command to generate a gate signal for controlling the operation of the power converter. The carrier frequency adjustment unit adjusts the frequency of the carrier wave by changing the phase difference between the voltage command and the carrier wave. When the modulation rate corresponding to the voltage amplitude ratio of the DC power and the AC power exceeds a predetermined value, the current control unit corrects the amplitude and phase of the voltage command according to the phase of the carrier wave.

[0011] Furthermore, an electromechanical unit of the present invention includes: a motor control device; a power converter connected to the motor control device; an AC motor driven by the power converter; and a gear that transmits the rotational driving force of the AC motor, wherein the AC motor, the power converter, and the gear are integrated into a single structure.

[0012] Furthermore, an electric vehicle system according to the present invention includes: a motor control device; a power converter connected to the motor control device; and an AC motor driven by the power converter, which drives the vehicle using the rotational driving force of the AC motor.

[0013] Furthermore, a motor control method of the present invention controls the operation of a power converter that performs power conversion from DC power to AC power, and controls the drive of an AC motor driven by the AC power. In this motor control method, a voltage command corresponding to a torque command is generated, a carrier wave is generated, the frequency of the carrier wave is adjusted by changing the phase difference between the voltage command and the carrier wave, and a gate signal for controlling the operation of the power converter is generated by pulse width modulation of the voltage command using the carrier wave. In the generation of the voltage command, when the modulation rate corresponding to the voltage amplitude ratio of the DC power to the AC power exceeds a predetermined value, the amplitude and phase of the voltage command are corrected according to the phase of the carrier wave.

[0014] The effects of the invention

[0015] This invention enables proper motor control during overmodulation. Attached Figure Description

[0016] Figure 1This is an overall configuration diagram of a motor drive system with a motor control device according to an embodiment of the present invention.

[0017] Figure 2 This is a block diagram illustrating the functional configuration of the motor control device according to the first embodiment of the present invention.

[0018] Figure 3 This is a block diagram of the carrier frequency adjustment unit according to the first embodiment of the present invention.

[0019] Figure 4 This is a block diagram of the voltage phase error calculation unit according to the first embodiment of the present invention.

[0020] Figure 5 This is a conceptual diagram of the reference voltage phase calculation of the present invention.

[0021] Figure 6 This is a graph showing the relationship between the modulated wave and the carrier wave when the modulation rate is set to 1.25 (overmodulation).

[0022] Figure 7 This is a graph showing the relationship between the voltage command gain and the inverter output voltage.

[0023] Figure 8 This is a block diagram of the current control unit according to the first embodiment of the present invention.

[0024] Figure 9 This is a graph showing the relationship between the amplitude of the second voltage and the amplitude and phase of the first voltage when the phase of the first voltage is 30 degrees.

[0025] Figure 10 This is a graph showing the relationship between the amplitude of the second voltage and the amplitude and phase of the first voltage when the phase of the first voltage is 60 degrees.

[0026] Figure 11 This is a graph showing the relationship between the amplitude of the second voltage and the amplitude and phase of the first voltage when the phase of the first voltage is 90 degrees.

[0027] Figure 12 This is a block diagram illustrating the functional configuration of the motor control device according to the second embodiment of the present invention.

[0028] Figure 13 This is a block diagram of the carrier frequency adjustment unit according to the second embodiment of the present invention.

[0029] Figure 14 This is a block diagram of the current control unit according to the second embodiment of the present invention.

[0030] Figure 15 This is a perspective view of the mechatronic unit of the motor drive system using the present invention.

[0031] Figure 16 A schematic diagram of a hybrid vehicle system utilizing the motor drive system of the present invention. Detailed Implementation

[0032] (First Embodiment) Hereinafter, the first embodiment of the present invention will be described with reference to the accompanying drawings.

[0033] Figure 1 This is an overall configuration diagram of a motor drive system equipped with a motor control device according to an embodiment of the present invention. Figure 2 In the motor drive system 100, there are a motor control device 1, a permanent magnet synchronous motor (hereinafter referred to as "motor") 2, an inverter 3, a rotary position detector 41, and a high-voltage battery 5.

[0034] Motor control unit 1 generates a gate signal for controlling the drive of motor 2 based on a torque command T* corresponding to the target torque required by the vehicle for motor 2, and outputs it to inverter 3. Further details of motor control unit 1 will be explained later.

[0035] Inverter 3 includes an inverter circuit 31, a PWM signal drive circuit 32, and a smoothing capacitor 33. The PWM signal drive circuit 32 generates PWM signals for controlling the switching elements of the inverter circuit 31 based on a gate signal input from the motor control device 1, and outputs these signals to the inverter circuit 31. The inverter circuit 31 has switching elements corresponding to the upper and lower arms of the U-phase, V-phase, and W-phase, respectively. These switching elements are controlled according to the PWM signals input from the PWM signal drive circuit 32, thereby converting the DC power supplied from the high-voltage battery 5 into AC power and outputting it to the motor 2. The smoothing capacitor 33 smooths the DC power supplied from the high-voltage battery 5 to the inverter circuit 31.

[0036] Motor 2 is a synchronous motor driven by alternating current supplied from inverter 3, and has a stator and a rotor. When alternating current input from inverter 3 is applied to the armature coils Lu, Lv, and Lw located in the stator, three-phase alternating currents Iu, Iv, and Iw are energized in motor 2, thereby generating armature flux in each armature coil. Attraction and repulsion forces are generated between these armature fluxes and the magnetic flux of the permanent magnets located in the rotor, thus generating torque on the rotor and driving its rotation.

[0037] A rotational position sensor 4 for detecting the rotational position θ of the rotor is installed on the motor 2. The rotational position detector 41 calculates the rotational position θ based on the input signal from the rotational position sensor 4. The calculated rotational position θ from the rotational position detector 41 is input to the motor control device 1 and used in the phase control of the AC power generated by the motor control device 1 based on the phase of the induced voltage of the motor 2.

[0038] Here, the rotary position sensor 4 is preferably a rotary transformer consisting of an iron core and windings, but it can also be a sensor using magnetoresistive elements or Hall elements, such as a GMR sensor. Furthermore, the rotary position detector 41 can also estimate the rotary position θ using the three-phase AC currents Iu, Iv, Iw flowing to the motor 2 or the three-phase AC voltages Vu, Vv, Vw applied to the motor 2 from the inverter 3, instead of using the input signal from the rotary position sensor 4.

[0039] A current detection component 7 is disposed between the inverter 3 and the motor 2. The current detection component 7 detects the three-phase AC currents Iu, Iv, and Iw (U-phase AC current Iu, V-phase AC current Iv, and W-phase AC current Iw) flowing in the motor 2. The current detection component 7 is constructed, for example, using a Hall current sensor. The detection results of the three-phase AC currents Iu, Iv, and Iw from the current detection component 7 are input to the motor control device 1 for generating the gate signal performed by the motor control device 1. Furthermore, in Figure 2 The example shown is of a current detection component 7 consisting of three current detectors, but it is also possible to use two current detectors. The AC current of the remaining phase can be calculated using the fact that the sum of the three-phase AC currents Iu, Iv, and Iw is zero. Alternatively, the pulsed DC current flowing from the high-voltage battery 5 to the inverter 3 can be detected by using a shunt resistor inserted between the smoothing capacitor 33 and the inverter 3. The three-phase AC currents Iu, Iv, and Iw can be calculated based on this DC current and the three-phase AC voltages Vu, Vv, and Vw applied from the inverter 3 to the motor 2.

[0040] Next, details of the motor control device 1 will be explained. Figure 2 This is a block diagram illustrating the functional configuration of the motor control device 1 according to the first embodiment of the present invention. Figure 2 In this motor control device 1, there are functional blocks including a current command generation unit 11, a speed calculation unit 12, a three-phase / dq conversion current control unit 13, a current control unit 14, a dq / three-phase voltage command conversion unit 15, a carrier frequency adjustment unit 16, a triangular wave generation unit 17, and a gate signal generation unit 18. The motor control device 1 is, for example, composed of a microcomputer, and these functional blocks can be implemented by executing a predetermined program in the microcomputer. Alternatively, some or all of these functional blocks can be implemented using hardware circuits such as logic ICs or FPGAs.

[0041] The current command generation unit 11 calculates the d-axis current command Id* and the q-axis current command Iq* based on the input torque command T* and the power supply voltage Hvdc. Here, for example, a pre-set current command mapping or formula is used to calculate the d-axis current command Id* and the q-axis current command Iq* corresponding to the torque command T*.

[0042] The speed calculation unit 12 calculates the motor speed ωr, which represents the rotational speed (rotational speed) of the motor 2, based on the time change of the rotational position θ. Furthermore, the motor speed ωr can be a value expressed as either angular velocity (rad / s) or rotational speed (rpm). In addition, these values ​​can be converted between each other.

[0043] The three-phase / dq conversion current control unit 13 performs dq conversion on the three-phase AC currents Iu, Iv, and Iw detected by the current detection component 7 based on the rotational position θ obtained by the rotational position detector 41, and calculates the d-axis current value Id and the q-axis current value Iq.

[0044] The current control unit 14 calculates the d-axis voltage command Vd* and q-axis voltage command Vq* corresponding to the torque command T* based on the deviations between the d-axis current command Id* and q-axis current command Iq* output from the current command generation unit 11 and the d-axis current value Id and q-axis current value Iq output from the three-phase / dq conversion current control unit 13, ensuring that these values ​​are consistent. Here, for example, a control method such as PI control is used to determine the d-axis voltage command Vd* corresponding to the deviation between the d-axis current command Id* and the d-axis current value Id, and the q-axis voltage command Vq* corresponding to the deviation between the q-axis current command Iq* and the q-axis current value Iq.

[0045] Furthermore, in the motor control device 1 of this embodiment, the current control unit 14 is characterized by its calculation method for the d-axis voltage command Vd* and q-axis voltage command Vq* during overmodulation control when the amplitude of the output voltage of the inverter 3 becomes larger than the DC voltage of the high-voltage battery 5. During overmodulation control, the amplitude and phase of the d-axis voltage command Vd* and the q-axis voltage command Vq* are corrected respectively based on the carrier phase difference Δθcarr calculated by the carrier frequency adjustment unit 16. This will be explained in detail later.

[0046] The dq / three-phase voltage command conversion unit 15 performs a three-phase conversion on the d-axis voltage command Vd* and q-axis voltage command Vq* calculated by the current control unit 14 based on the rotational position θ obtained by the rotational position detector 41, and calculates the three-phase voltage commands Vu*, Vv*, and Vw* (U-phase voltage command value Vu*, V-phase voltage command value Vv*, and W-phase voltage command value Vw*). This generates the three-phase voltage commands Vu*, Vv*, and Vw* corresponding to the torque command T*.

[0047] The carrier frequency adjustment unit 16 calculates the carrier frequency fc, which represents the frequency of the carrier used in generating the gate signal, and the carrier phase difference Δθcarr, which represents the phase difference between the reference voltage phase θvb and the phase of the carrier, based on the d-axis voltage command Vd* and q-axis voltage command Vq* generated by the current command generation unit 11, the rotational position θ obtained by the rotational position detector 41, the rotational speed ωr obtained by the speed calculation unit 12, the torque command T*, and the power supply voltage Hvdc. The reference voltage phase θvb is the reference value of the phase of the carrier in synchronous PWM control, and is obtained when the carrier frequency adjustment unit 16 calculates the carrier frequency fc. That is, the carrier phase difference Δθcarr represents the phase of the carrier based on the reference voltage phase θvb. Furthermore, the details of the reference voltage phase θvb will be described later. The triangular wave generation unit 17 generates the carrier according to the carrier frequency fc, thereby adjusting the carrier frequency in a way that can suppress vibration and noise generated in the motor 2. Furthermore, during overmodulation control, the current control unit 14 corrects the d-axis voltage command Vd* and the q-axis voltage command Vq* based on the carrier phase difference Δθcarr. Moreover, details of the calculation method for the carrier frequency fc and the carrier phase difference Δθcarr by the carrier frequency adjustment unit 16 will be described later.

[0048] The triangular wave generation unit 17 generates a triangular wave signal (carrier signal) Tr based on the carrier frequency fc calculated by the carrier frequency adjustment unit 16.

[0049] The gate signal generation unit 18 uses the triangular wave signal Tr output from the triangular wave generation unit 17 to perform pulse width modulation on the three-phase voltage commands Vu*, Vv*, and Vw* output from the dq / three-phase voltage command conversion unit 15, respectively, to generate gate signals for controlling the operation of the inverter 3. Specifically, pulse-shaped voltages are generated for each of the U, V, and W phases based on the comparison result between the three-phase voltage commands Vu*, Vv*, and Vw* output from the dq / three-phase voltage command conversion unit 15 and the triangular wave signal Tr output from the triangular wave generation unit 17. Then, gate signals for the switching elements of each phase of the inverter 3 are generated based on the generated pulse-shaped voltages. At this time, the gate signals Gup, Gvp, and Gwp of the upper arm of each phase are logically inverted to generate the gate signals Gun, Gvn, and Gwn of the lower arm. The gate signal generated by the gate signal generation unit 18 is output from the motor control device 1 to the PWM signal drive circuit 32 of the inverter 3, and is converted into a PWM signal by the PWM signal drive circuit 32. As a result, the switching elements of the inverter circuit 31 are turned on / off, thereby adjusting the output voltage of the inverter 3.

[0050] Next, the operation of the carrier frequency adjustment unit 16 in the motor control device 1 will be explained. As described above, the carrier frequency adjustment unit 16 calculates the carrier frequency fc and carrier phase difference Δθcarr based on the d-axis voltage command Vd* and q-axis voltage command Vq*, rotational position θ, rotational speed ωr, torque command T*, and power supply voltage Hvdc. By successively controlling the frequency of the triangular wave signal Tr generated by the triangular wave generation unit 17 according to the carrier frequency fc, the period and phase of the carrier, i.e., the triangular wave signal Tr, are adjusted to a desired relationship relative to the voltage waveforms of the three-phase voltage commands Vu*, Vv*, and Vw* corresponding to the torque command T*. Furthermore, the desired relationship here refers, for example, to a relationship where the electromagnetic excitation force or torque pulsation generated in the motor 2 due to the high-order harmonic current caused by the switching operation of the inverter 3 based on the PWM signal is periodic but out of phase with the electromagnetic excitation force or torque pulsation generated by the fundamental current corresponding to the voltage command.

[0051] Figure 3 This is a block diagram of the carrier frequency adjustment unit 16 according to the first embodiment of the present invention. The carrier frequency adjustment unit 16 includes a synchronous PWM carrier number selection unit 161, a voltage phase calculation unit 162, a modulation rate calculation unit 163, a voltage phase error calculation unit 164, a synchronous carrier frequency calculation unit 165, and a carrier frequency setting unit 166.

[0052] The synchronous PWM carrier number selection unit 161 selects the synchronous PWM carrier number Nc, which represents the number of carriers corresponding to one cycle of the voltage waveform in synchronous PWM control, based on the rotational speed ωr. For example, the synchronous PWM carrier number selection unit 161 selects a number that is a multiple of 3 and satisfies the condition Nc = 3 × (2 × n - 1). In this condition, n represents any natural number, such as n = 1 (Nc = 3), n = 2 (Nc = 9), n = 3 (Nc = 15), etc. Alternatively, by using special carriers, numbers such as Nc = 6 or Nc = 12, which are multiples of 3 but do not satisfy the above condition, can be selected as the synchronous PWM carrier number Nc. Furthermore, the synchronous PWM carrier number selection unit 161 can also select the synchronous PWM carrier number Nc based not only on the rotational speed ωr but also on the torque command T*. Additionally, for example, a hysteresis can be set to change the selection criteria for the synchronous PWM carrier number Nc when the rotational speed ωr increases and decreases.

[0053] The voltage phase calculation unit 162 calculates the voltage phase θv according to the d-axis voltage command Vd* and the q-axis voltage command Vq*, the rotation position θ, the rotation speed ωr, and the carrier frequency fc using the following formulas (1) to (4).

[0054] θv=θ+φv+φdqv+0.5π···(1)

[0055] φv=ωr·1.5Tc···(2)

[0056] Tc=1 / fc···(3)

[0057] φdqv=atan(Vq / Vd)···(4)

[0058] Here, φv represents the operational delay compensation value for the voltage phase, Tc represents the carrier period, and φdqv represents the voltage phase at a distance from the d-axis. The operational delay compensation value φv is a value used to compensate for the operational delay of 1.5 control cycles that occurs during the period from when the rotational position detector 41 obtains the rotational position θ until the motor control device 1 outputs the gate signal to the inverter 3. Furthermore, in this embodiment, 0.5π is added to the fourth term on the right side of equation (1). The purpose of this calculation is to transform the viewpoint of the voltage phase calculated in the first to third terms on the right side of equation (1) from a cosine wave to a sinine wave.

[0059] The modulation rate calculation unit 163 calculates the modulation rate H according to the following formula (5) based on the d-axis voltage command Vd*, the q-axis voltage command Vq*, and the power supply voltage Hvdc. Furthermore, the modulation rate H represents the voltage amplitude ratio of the DC power supplied from the high-voltage battery 5 to the inverter 3 and the AC power output from the inverter 3 to the motor 2.

[0060] H=√(Vd^2+Vq^2) / (Hvdc / 2)···(5)

[0061] The voltage phase error calculation unit 164 calculates the voltage phase error Δθv and the carrier phase difference Δθcarr based on the synchronous PWM carrier number Nc selected by the synchronous PWM carrier number selection unit 161, the voltage phase θv calculated by the voltage phase calculation unit 162, the modulation rate H calculated by the modulation rate calculation unit 163, the speed ωr, and the torque command T*. The voltage phase error Δθv represents the phase difference between the voltage command for the inverter 3 (i.e., the three-phase voltage commands Vu*, Vv*, Vw*) and the carrier used for pulse width modulation (i.e., the triangular wave signal Tr). The voltage phase error calculation unit 164 calculates the voltage phase error Δθv according to a predetermined calculation cycle. Therefore, the frequency of the triangular wave signal Tr can be adjusted in the carrier frequency adjustment unit 16 by changing the phase difference between the voltage command for the inverter 3 and the carrier used for pulse width modulation. Furthermore, the carrier phase difference Δθcarr represents the phase difference between the reference voltage phase θvb and the triangular wave signal Tr. Moreover, as mentioned above, the reference voltage phase θvb is the reference value for the phase of the carrier in synchronous PWM control. Therefore, the carrier phase difference Δθcarr is equivalent to the phase of the triangular wave signal Tr during synchronous PWM control.

[0062] The synchronous carrier frequency calculation unit 165 calculates the synchronous carrier frequency fcs according to the following formula (6) based on the voltage phase error Δθv calculated by the voltage phase error calculation unit 164, the rotational speed ωr, and the synchronous PWM carrier number Nc selected by the synchronous PWM carrier number selection unit 161.

[0063] fcs=ωr·Nc·(1+Δθv·K) / (2π)···(6)

[0064] The synchronization carrier frequency calculation unit 165 can, for example, calculate the synchronization carrier frequency fcs based on equation (6) through PLL (Phase Locked Loop) control. Furthermore, in equation (6), the gain K can be set to a fixed value or can be varied according to conditions.

[0065] The carrier frequency setting unit 166 selects one of the synchronous carrier frequency fcs calculated by the synchronous carrier frequency calculation unit 165 and the asynchronous carrier frequency fcns based on the rotational speed ωr, and outputs it as the carrier frequency fc. The asynchronous carrier frequency fcns is a fixed value preset in the carrier frequency setting unit 166. Alternatively, multiple asynchronous carrier frequencies fcns can be prepared in advance, and one of them can be selected based on the rotational speed ωr. For example, the asynchronous carrier frequency fcns can be selected in the carrier frequency setting unit 166 in such a way that the larger the value of the rotational speed ωr, the larger the value of the asynchronous carrier frequency fcns, and output as the carrier frequency fc.

[0066] Next, details of the calculation method for the voltage phase error Δθv in the voltage phase error calculation unit 164 of the carrier frequency adjustment unit 16 will be explained.

[0067] Figure 4 This is a block diagram of the voltage phase error calculation unit 164 according to the first embodiment of the present invention. The voltage phase error calculation unit 164 includes a reference voltage phase calculation unit 1641, a carrier triangular wave phase table 1644, a voltage phase difference conversion unit 1645, an addition unit 1646, and a subtraction unit 1647.

[0068] The reference voltage phase calculation unit 1641 calculates the reference voltage phase θvb for the phase of the carrier in fixed synchronous PWM control based on the number of synchronous PWM carriers Nc and the voltage phase θv. By calculating the reference voltage phase θvb by the reference voltage phase calculation unit 1641, the period of the carrier corresponding to the voltage phase θv can be made consistent with the period of the electromagnetic excitation force and torque pulsation generated in the motor 2 due to the fundamental current.

[0069] Figure 5 This is a conceptual diagram of the reference voltage phase calculation performed by the reference voltage phase calculation unit 1641. The reference voltage phase calculation unit 1641 is, for example, like... Figure 5The reference voltage phase θvb is calculated as follows, varying in a stepwise manner between 0 and 2π, corresponding to the number of synchronous PWM carriers Nc. Furthermore, to make the explanation easier to understand, Figure 5 The example shown is when the number of synchronous PWM carriers Nc is 3, but in practice, the number of synchronous PWM carriers Nc is preferably set to Nc = 3, 9, or 15 as mentioned above. Alternatively, it can be set to Nc = 6 or 12.

[0070] In this embodiment, in order to reduce the processing load, for example, like Figure 5 As shown, the carrier frequency adjustment unit 16 can adjust the carrier frequency only within the trough segmentation interval where the triangular carrier rises from its minimum (valley) to its maximum (peak). In this case, the synchronization carrier frequency calculation unit 165 calculates the synchronization carrier frequency fcs sequentially within the trough segmentation interval of the carrier using the voltage phase error Δθv, as described later, thereby implementing synchronous PWM control. The reference voltage phase calculation unit 1641 calculates the reference voltage phase θvb used for calculating the voltage phase error Δθv, as a reference voltage phase... Figure 5 The discrete values ​​shown are varied in π / 3 intervals. Furthermore, the interval of the reference voltage phase θvb varies according to the number of synchronous PWM carriers Nc. The larger the number of synchronous PWM carriers Nc, the smaller the interval of the reference voltage phase θvb.

[0071] Specifically, the reference voltage phase calculation unit 1641 calculates the reference voltage phase θvb according to the following formulas (7) to (8) based on the voltage phase θv and the number of synchronous PWM carriers Nc.

[0072] θvb=int(θv / θs)·θs+0.5θs···(7)

[0073] θs=2π / Nc···(8)

[0074] Here, θs represents the change in voltage phase θv for each carrier wave, and int represents the rounding down operation below the decimal point.

[0075] Furthermore, in this embodiment, the reference voltage phase θvb is calculated in the reference voltage phase calculation unit 1641 according to equations (7) to (8) in such a way that the reference voltage phase θvb becomes 0 rad within the peak-to-minimum interval of the triangular carrier wave, i.e., the peak-to-valley interval. However, the period during which the reference voltage phase θvb becomes 0 rad is not limited to the peak-to-valley interval. As long as the voltage phase θvb can be calculated using the voltage phase θv, which varies in a stepwise manner between 0 and 2π in a series corresponding to the number of synchronous PWM carrier waves Nc, the reference voltage phase calculation unit 1641 can also perform the calculation of the reference voltage phase θvb using calculation methods other than equations (7) to (8).

[0076] The carrier triangular wave phase table 1644 is a table that represents the phase difference used to reduce electromagnetic excitation force and torque ripple in motor 2. Here, phase difference refers to the phase difference relative to the reference voltage phase θvb. The carrier triangular wave phase table 1644 is set separately for multiple values: rotational speed ωr, torque command T*, and modulation rate H. In the voltage phase error calculation unit 164, the carrier triangular wave phase table 1644 is referenced based on the rotational speed ωr, torque command T*, and modulation rate H, thereby determining the phase difference suitable for reducing electromagnetic excitation force and torque ripple.

[0077] For example, phase difference data relative to the reference voltage phase θvb, which reduces electromagnetic excitation force and torque ripple, is obtained in advance by simulation or actual measurement for each rotational speed ωr, torque command T*, and modulation rate H. The carrier triangular wave phase table 1644 is set based on this pre-obtained phase difference data. The reason for setting the carrier triangular wave phase table 1644 for each modulation rate H is to compensate for the change in the dominant order of electromagnetic excitation force and torque ripple caused by higher harmonic currents according to the modulation rate H. Furthermore, the phase difference output from the carrier triangular wave phase table 1644 can be either a current phase difference or a voltage phase difference. In this embodiment, the phase difference output from the carrier triangular wave phase table 1644 is a current phase difference, and the conversion from current phase difference to voltage phase difference is performed in the subsequent voltage phase difference conversion unit 1645.

[0078] The voltage phase difference conversion unit 1645 converts the current phase difference input from the carrier triangular wave phase meter 1644 into a voltage phase difference by adding 0.5π. The reason for adding 0.5π here is that higher harmonic currents are less affected by resistance than the fundamental current. Therefore, it is mainly the differential value (0.5π) of the higher harmonic current flowing to the inductive component of the motor 2 that affects the voltage of the motor 2. Furthermore, if the phase difference output from the carrier triangular wave phase meter 1644 is set as a voltage phase difference as described above, the voltage phase difference conversion unit 1645 is not necessary.

[0079] The voltage phase difference determined by referring to the carrier triangular wave phase table 1644 based on the rotational speed ωr, torque command T*, and modulation rate H is output from the voltage phase error calculation unit 164 as the carrier phase difference Δθcarr mentioned above. Therefore, in the carrier frequency adjustment unit 16, the carrier phase difference Δθcarr representing the phase of the triangular wave signal Tr can be calculated based on the reference voltage phase θvb and output to the current control unit 14.

[0080] The addition unit 1646 adds the voltage phase difference calculated by the voltage phase difference conversion unit 1645 to the reference voltage phase θvb calculated in the reference voltage phase calculation unit 1641, and calculates a corrected reference voltage phase θvb2 for reducing electromagnetic excitation force or torque pulsation caused by high-order harmonic current.

[0081] The subtraction unit 1647 subtracts the corrected reference voltage phase θvb2 from the voltage phase θv to calculate the voltage phase error Δθv.

[0082] As explained above, the voltage phase error calculation unit 164 calculates the voltage phase error Δθv and the carrier phase difference Δθcarr. Therefore, the voltage phase error Δθv can be determined based on the rotational speed ωr, torque command T*, and modulation rate H, in a manner where the torque ripple and electromagnetic excitation force caused by the fundamental current corresponding to the three-phase voltage commands Vu*, Vv*, and Vw* are canceled out by the torque ripple and electromagnetic excitation force caused by the carrier used in pulse width modulation. As a result, the carrier frequency fc can be set by changing the phase difference between the voltage command for the inverter 3 and the carrier used for pulse width modulation in a way that reduces the torque ripple and electromagnetic excitation force generated in the motor 2.

[0083] Furthermore, in the carrier frequency adjustment unit 16, the above-mentioned processing can be performed either during the power operation drive of the motor 2 or during regenerative drive. During power operation drive, the torque command T* becomes a positive value, and during regenerative drive, the torque command T* becomes a negative value. Therefore, in the carrier frequency adjustment unit 16, a determination can be made based on the value of the torque command T* to determine whether the motor 2 is in power operation drive or regenerative drive, and based on the result of this determination, the above-mentioned calculation processing is performed in the voltage phase error calculation unit 164. As a result, the carrier frequency fc is set by changing the voltage phase error Δθv in a way that reduces the torque pulsation and electromagnetic excitation force generated in the motor 2.

[0084] Next, the operation of the current control unit 14 in the motor control device 1 will be explained. As mentioned above, the motor control device 1 of this embodiment is characterized by the calculation method of the d-axis voltage command Vd* and q-axis voltage command Vq* performed by the current control unit 14 during overmodulation control, which will be described in detail below.

[0085] First, the modulation wave Vmod and the carrier Vcar are defined as in equations (9) and (10), respectively. In equation (9), the modulation wave Vmod is defined by injecting the third harmonic of the fundamental component with the third harmonic superimposed on it. Furthermore, the fundamental component of the modulation wave Vmod is equivalent to the three-phase voltage commands Vu*, Vv*, and Vw* output from the dq / three-phase voltage command conversion unit 15 and input to the gate signal generation unit 18. The modulation wave Vmod is compared with the carrier Vcar in the gate signal generation unit 18, thereby performing pulse width modulation. In addition, in equation (10), the triangular wave signal Tr generated by the triangular wave generation unit 17 is defined as the carrier Vcar.

[0086] Vmod=E×sin(ωt)+E / 6×sin(3ωt)···(9)

[0087] Vcar=sin(Nc×ωt+Δθcarr)···(10)

[0088] ※E: Gain of voltage command

[0089] ω: Electrical angular frequency

[0090] t: time

[0091] Figure 6 This is a graph showing the relationship between the modulated wave Vmod and the carrier wave Vcar when the modulation rate is set to 1.25 (overmodulation). Figure 6 In the diagram, (a) shows the relationship between the modulated wave Vmod and the carrier wave Vcar when Δθcarr = 0 degrees, (b) shows the relationship between the modulated wave Vmod and the carrier wave Vcar when Δθcarr = 90 degrees, (c) shows the relationship between the modulated wave Vmod and the carrier wave Vcar when Δθcarr = 180 degrees, and (d) shows the relationship between the modulated wave Vmod and the carrier wave Vcar when Δθcarr = 270 degrees.

[0092] Figure 7 This is a graph showing the relationship between the voltage command gain E and the output voltage of inverter 3. Figure 7 In the example, (a) shows the cases where Δθcarr = 0 degrees, 90 degrees, 180 degrees, and 270 degrees, which is... Figure 6 The relationship between the voltage command gain E and the output voltage phase of inverter 3 is shown in cases (a) to (d). Furthermore, (b) illustrates the relationships between Δθcarr = 0 degrees, 90 degrees, 180 degrees, and 270 degrees. Figure 6 The relationship between the voltage command gain E and the output voltage amplitude of inverter 3 in cases (a) to (d) is shown. Furthermore, Figure 7The output voltage phase of inverter 3 shown in (a) is based on the phase of the modulation wave Vmod, which is equivalent to the phase difference between the modulation wave Vmod and the output voltage of inverter 3. Furthermore, Figure 7 The output voltage amplitude of inverter 3 shown in (b) is based on the power supply voltage Hvdc, which is equivalent to the modulation rate.

[0093] according to Figure 7 It is understood that the relationship between the voltage command gain E and the output voltage (three-phase AC voltages Vu, Vv, Vw) of inverter 3 varies depending on the value of the carrier phase difference Δθcarr. Figure 7 As can be confirmed in (a), the output voltage phase of inverter 3 should be 0 degrees (unchanged) regardless of the value of the voltage command gain E, but it varies within a range of ±7 degrees when Δθcarr = 90 degrees and 270 degrees. This tendency becomes more pronounced when the modulation rate exceeds 1.15 for overmodulation. Furthermore, in Figure 7 As can be confirmed in (b), the output voltage amplitude of inverter 3 should change linearly with a certain slope in proportion to the gain E of the voltage command, but the slope changes when the modulation rate exceeds 1.15 for overmodulation, and the slope varies depending on the value of Δθcarr.

[0094] In the motor control device 1 of this embodiment, in order to reduce the amplitude and phase errors of the inverter 3's output voltage, which vary according to the carrier phase difference Δθcarr as described above, the current control unit 14 corrects the amplitude and phase of the d-axis voltage command Vd* and the q-axis voltage command Vq* based on the value of Δθcarr during overmodulation. This allows for proper motor control during overmodulation. Next, using... Figures 8 to 11 The details are described below.

[0095] Figure 8 This is a block diagram of the current control unit 14 according to the first embodiment of the present invention. The current control unit 14 includes a subtraction unit 141a, a subtraction unit 141b, a d-axis current control unit 142a, a q-axis current control unit 142b, a modulation rate calculation unit 143, an amplitude / phase calculation unit 144, an amplitude / phase correction unit 145, a correction voltage command calculation unit 146, and a switching unit 147.

[0096] The subtraction unit 141a calculates the deviation between the output of the current command generation unit 11 (i.e., the d-axis current command Id*) and the output of the three-phase / dq conversion current control unit 13 (i.e., the d-axis current Id). On the other hand, the subtraction unit 141b calculates the deviation between the output of the current command generation unit 11 (i.e., the q-axis current command Iq*) and the output of the three-phase / dq conversion current control unit 13 (i.e., the q-axis current Iq).

[0097] The d-axis current control unit (IdACR) 142a calculates the first d-axis voltage command Vd1* on the dq coordinate axis in such a way that the current deviation calculated by the subtraction unit 141a becomes zero. On the other hand, the q-axis current control unit (IqACR) 142b calculates the first q-axis voltage command Vq1* on the dq coordinate axis in such a way that the current deviation calculated by the subtraction unit 141b becomes zero.

[0098] The modulation rate calculation unit 143 calculates the modulation rate H according to the following formula (11) (=Formula (5)), based on the d-axis voltage command Vd*, the q-axis voltage command Vq*, and the power supply voltage Hvdc. Furthermore, as mentioned above, the modulation rate H represents the voltage amplitude ratio of the DC power supplied from the high-voltage battery 5 to the inverter 3 to the AC power output from the inverter 3 to the motor 2.

[0099] H=√(Vd^2+Vq^2) / (Hvdc / 2)···(11)

[0100] The amplitude / phase calculation unit 144 calculates the first voltage amplitude |V1*| and the first voltage phase θ1* according to the first d-axis voltage command Vd1* calculated by the d-axis current control unit 142a and the first q-axis voltage command Vq1* calculated by the q-axis current control unit 142b, according to the following equations (12) and (13).

[0101] |V1*|=√(Vd1*^2+Vq1*^2)···(12)

[0102] θ1*=atan(Vq1* / -Vd1*)···(13)

[0103] The amplitude / phase correction unit 145 corrects the first voltage amplitude |V1*| and the first voltage phase θ1* calculated by the amplitude / phase calculation unit 144 based on the carrier phase difference Δθcarr input from the carrier frequency adjustment unit 16, and calculates the second voltage amplitude |V2*| and the second voltage phase θ2*. For example, the amplitude / phase correction unit 145 stores the relationship between the first voltage amplitude |V1*| and the second voltage amplitude |V2*|, and the relationship between the first voltage phase θ1* and the second voltage phase θ2*, which are pre-calculated for various values ​​of the carrier phase difference Δθcarr, as correction mapping information. Specifically, correction mapping information is pre-created and stored in the amplitude / phase correction unit 145 in such a way that the amplitude and phase of the three-phase AC voltages Vu, Vv, Vw output from the inverter 3 according to the first d-axis voltage command Vd1* and the first q-axis voltage command Vq1*, when the voltage phase error Δθv mentioned above is set to a fixed value, are within a predetermined range compared with the amplitude and phase of the three-phase AC voltages Vu, Vv, Vw output from the inverter 3 according to the second d-axis voltage command Vd2* and the second q-axis voltage command Vq2*, determined according to the second voltage amplitude |V2*| and the second voltage phase θ2*, when the voltage phase error Δθv is changed in the voltage phase error calculation unit 164. Then, the pre-stored correction mapping information is mapped and retrieved according to the input carrier phase difference Δθcarr, the first voltage amplitude |V1*|, and the first voltage phase θ1*, thereby calculating the second voltage amplitude |V2*| and the second voltage phase θ2*.

[0104] Figure 9 , Figure 10 , Figure 11 The relationships between the second voltage amplitude |V2*| and the first voltage amplitude |V1*| and the second voltage phase θ2* are shown when the first voltage phase θ1* is 30 degrees, 60 degrees, and 90 degrees, respectively. Specifically, Figure 9 (a) shows the relationship between |V2*| and θ2* for the cases where θ1* = 30 degrees, Δθcarr = 0 degrees, 90 degrees, 180 degrees, and 270 degrees. Figure 9 (b) shows the relationship between |V2*| and |V1*| for the cases where θ1* = 30 degrees, Δθcarr = 0 degrees, 90 degrees, 180 degrees, and 270 degrees. Figure 10 (a) shows the relationship between |V2*| and θ2* for the cases where θ1* = 60 degrees, Δθcarr = 0 degrees, 90 degrees, 180 degrees, and 270 degrees. Figure 10 (b) shows the relationship between |V2*| and |V1*| for the cases where θ1* = 60 degrees, Δθcarr = 0 degrees, 90 degrees, 180 degrees, and 270 degrees. Figure 11(a) shows the relationship between |V2*| and θ2* for the cases where θ1* = 90 degrees, Δθcarr = 0 degrees, 90 degrees, 180 degrees, and 270 degrees. Figure 11 (b) shows the relationship between |V2*| and |V1*| for the cases where θ1* = 90 degrees, Δθcarr = 0 degrees, 90 degrees, 180 degrees, and 270 degrees. Furthermore, in Figure 9 , Figure 10 , Figure 11 In this context, both |V1*| and |V2*| are standardized according to the definition of modulation rate.

[0105] Amplitude / phase correction unit 145 can be based on Figures 9-11 The relationship between the second voltage amplitude |V2*| and the first voltage amplitude |V1*| and the second voltage phase θ2* is shown. For example, the second voltage amplitude |V2*| and the second voltage phase θ2* can be obtained as follows.

[0106] First, the amplitude / phase correction unit 145 selects based on the value of the input first voltage phase θ1*. Figures 9-11 One of them. That is, choosing when θ1*=30 degrees. Figure 9 When θ1*=60 degrees, choose Figure 10 When θ1*=90 degrees, choose Figure 11 Furthermore, here, the value of θ1* is 30 degrees, and the selection is based on this. Figures 9-11 One of them, but the same method can be used when the scale width of θ1* is more than 30 degrees. In this case, as long as the relationship between the second voltage amplitude |V2*|, the first voltage amplitude |V1*|, and the second voltage phase θ2* is pre-stored in the amplitude / phase correction unit 145 to a degree corresponding to the scale width of θ1*, the relationship corresponding to the value of θ1* can be selected.

[0107] When selected Figures 9-11 When one of them is selected, the amplitude / phase correction unit 145 then refers to the selected graph to calculate the values ​​of |V2*| and θ2* corresponding to the input values ​​of Δθcarr and |V1*|. For example, when θ1* = 30 degrees and the selected... Figure 9 In the case of, Figure 9 In (b), select the relationship between |V1*| and |V2*| corresponding to the value of Δθcarr, and calculate the value of |V2*| corresponding to the value of |V1*| based on this relationship. Then, in... Figure 9 In (a), select the relationship between θ2* and |V2*| corresponding to the value of Δθcarr, and use this relationship to calculate the relationship with... Figure 9The value of θ2* corresponding to the value of |V2*| obtained in (b) can be used to determine the second voltage amplitude |V2*| and the second voltage phase θ2*.

[0108] The amplitude / phase correction unit 145 calculates the values ​​of |V2*| and θ2* respectively using the method described above. Therefore, based on the carrier phase difference Δθcarr input from the carrier frequency adjustment unit 16, the first voltage amplitude |V1*| and the first voltage phase θ1* are corrected respectively, and the second voltage amplitude |V2*| and the second voltage phase θ2* are calculated. Furthermore, in Figures 9-11 The value of the carrier phase difference Δθcarr is set to a 90-degree scale to show the relationship between the second voltage amplitude |V2*|, the first voltage amplitude |V1*|, and the second voltage phase θ2*.

[0109] However, the scale width of the carrier phase difference Δθcarr can also be set to a value other than 90 degrees. By making the scale width more precise, the accuracy of the second voltage amplitude |V2*| and the second voltage phase θ2* can be improved.

[0110] The voltage command calculation unit 146 uses the second voltage amplitude |V2*| and the second voltage phase θ2* obtained by the amplitude / phase correction unit 145 to calculate the second d-axis voltage command Vd2* and the second q-axis voltage command Vq2* according to the following formulas (14) and (15).

[0111] Vd1*=-|V2*|sinθ2*···(14)

[0112] Vq1*=|V2*|cosθ2*···(15)

[0113] The switching unit 147 selects any combination of the first d-axis voltage command Vd1* and the first q-axis voltage command Vq1* calculated by the d-axis current control unit 142a and the q-axis current control unit 142b, or the second d-axis voltage command Vd2* and the second q-axis voltage command Vq2* calculated by the correction voltage command calculation unit 146, based on the value of the modulation rate H calculated by the modulation rate calculation unit 143. Then, the selected combination of the d-axis voltage command and the q-axis voltage command is output as the calculation result of the d-axis voltage command Vd* and the q-axis voltage command Vq* of the current control unit 14. Specifically, for example, when the value of the modulation rate H is 1.15 or less, the combination of the first d-axis voltage command Vd1* and the first q-axis voltage command Vq1* is selected and output; when the value of the modulation rate H exceeds 1.15, the combination of the first d-axis voltage command Vd1* and the first q-axis voltage command Vq1* is switched to the combination of the second d-axis voltage command Vd2* and the second q-axis voltage command Vq2*. At this time, the rate of change of the voltage command before and after the switching can also be limited to a certain value so that no switching shock is generated in the output voltage of the inverter 3. In addition, the modulation rate H used by the switching unit 147 to switch can be set to different values ​​when the modulation rate H rises and falls, thereby setting a hysteresis on the switching unit 147 to prevent oscillation.

[0114] Furthermore, even when the modulation rate H is less than 1.15, the amplitude and phase of the output voltage of inverter 3 will vary depending on the value of the carrier phase Δθcarr. Therefore, in this case, the above configuration can also reduce the error in voltage amplitude and phase.

[0115] As explained above, when the modulation rate H is a predetermined value, such as 1.15 or higher, the current control unit 14 outputs the second d-axis voltage command Vd2* and the second q-axis voltage command Vq2* calculated by the correction voltage command calculation unit 146 instead of the first d-axis voltage command Vd1* and the first q-axis voltage command Vq1* calculated by the d-axis current control unit 142a and the q-axis current control unit 142b, respectively. At this time, the correction voltage command calculation unit 146 uses the second voltage amplitude |V2*| and the second voltage phase θ2*, calculated by the amplitude / phase correction unit 145 based on the carrier phase difference Δθcarr representing the phase of the carrier wave signal Tr, to calculate the second d-axis voltage command Vd2* and the second q-axis voltage command Vq2*. Therefore, the amplitude and phase errors of the output voltage of inverter 3 when the voltage phase error Δθv is changed can be reduced to within the specified range, and the amplitude and phase of the d-axis voltage command Vd* and the q-axis voltage command Vq* can be corrected respectively according to the carrier phase difference Δθcarr.

[0116] According to the first embodiment of the present invention described above, the following effects are achieved.

[0117] (1) A motor control device 1 is connected to an inverter 3 that performs power conversion from DC power to AC power, and controls the drive of a motor 2 driven by the AC power. The motor control device includes: a current control unit 14 that generates a d-axis voltage command Vd* and a q-axis voltage command Vq* corresponding to a torque command T*; a triangular wave generation unit 17 that generates a carrier wave, i.e., a triangular wave signal Tr; a carrier frequency adjustment unit 16 that adjusts the carrier frequency fc, which represents the frequency of the triangular wave signal Tr; and a gate signal generation unit 18 that uses the triangular wave signal Tr to pulse-width modulate the three-phase voltage commands Vu*, Vv*, and Vw* to generate a gate signal for controlling the operation of the inverter 3. The carrier frequency adjustment unit 16 adjusts the carrier frequency fc by changing the voltage phase error Δθv, which represents the phase difference between the three-phase voltage commands Vu*, Vv*, and Vw* and the triangular wave signal Tr. When the modulation rate H, corresponding to the voltage amplitude ratio of the DC power supplied from the high-voltage battery 5 to the inverter 3 and the AC power output from the inverter 3 to the motor 2, exceeds a predetermined value, for example, 1.15, the current control unit 14 corrects the amplitude and phase of the d-axis voltage command Vd* and the q-axis voltage command Vq* based on the carrier phase difference Δθcarr, which represents the phase of the triangular wave signal Tr. This allows for appropriate motor control during overmodulation.

[0118] (2) The current control unit 14 corrects the amplitude and phase of the three-phase AC voltages Vu, Vv, and Vw output from the inverter 3 based on the first d-axis voltage command Vd1* and the first q-axis voltage command Vq1* before correction when the voltage phase error Δθv is fixed, and the difference between the amplitude and phase of the three-phase AC voltages Vu, Vv, and Vw output from the inverter 3 based on the second d-axis voltage command Vd2* and the second q-axis voltage command Vq2* after correction when the voltage phase error Δθv is changed, respectively, within a specified range. Therefore, even when overmodulation occurs with a modulation rate exceeding 1.15, the desired voltage amplitude and phase can be obtained in the output voltage of the inverter 3 even when the carrier frequency fc is adjusted by changing the voltage phase error Δθv. Thus, torque can be stably output in the motor 2.

[0119] (3) The carrier frequency adjustment unit 16 adjusts the carrier frequency fc by changing the voltage phase error Δθv according to the torque command T* and the speed ωr of the motor 2. Therefore, it can effectively suppress the vibration and noise generated in the motor 2.

[0120] (4) The carrier frequency adjustment unit 16 changes the voltage phase error Δθv based on the torque command T*, the speed ωr, and the modulation rate H, which represents the voltage amplitude ratio between the DC power supplied to the inverter 3 and the AC power output from the inverter 3. Therefore, even if the dominant order of the electromagnetic excitation force and torque pulsation caused by high-order harmonic current changes according to the modulation rate H, and thus the vibration and noise of the motor 2 change according to the modulation rate H, the change can be reliably compensated and the vibration and noise generated in the motor 2 can be effectively suppressed.

[0121] (5) The carrier frequency adjustment unit 16 selects the synchronous PWM carrier number Nc to a predetermined integer value through the synchronous PWM carrier number selection unit 161. Thus, the carrier frequency fc is adjusted so that it becomes an integer multiple of the frequencies of the three-phase voltage commands Vu*, Vv*, and Vw*. Therefore, by adjusting the period and phase of the carrier, i.e., the triangular wave signal Tr, relative to the voltage waveforms of the three-phase voltage commands Vu*, Vv*, and Vw*, respectively, a desired relationship is achieved, thereby enabling reliable synchronous PWM control.

[0122] (6) The current control unit 14 sets the above-mentioned specified value to 1.15. When the modulation rate H exceeds 1.15, it corrects the amplitude and phase of the d-axis voltage command Vd* and the q-axis voltage command Vq*.

[0123] Therefore, when adjusting the carrier frequency fc by changing the voltage phase error Δθv, the amplitude and phase of the d-axis voltage command Vd* and the q-axis voltage command Vq* can be reliably corrected during overmodulation when the amplitude and phase errors in the output voltage of inverter 3 are significantly increased.

[0124] (7) Alternatively, the predetermined value of the modulation rate H used for correcting the amplitude and phase of the d-axis voltage command Vd* and the q-axis voltage command Vq* in the current control unit 14 when the modulation rate H increases can be set to a different value than the predetermined value used for correcting the amplitude and phase of the d-axis voltage command Vd* and the q-axis voltage command Vq* in the current control unit 14 when the modulation rate H decreases. In this way, when the modulation rate H repeatedly increases and decreases between predetermined values, oscillations caused by frequent switching between the presence and absence of corrections can be prevented. Therefore, fluctuations in the output voltage of the inverter 3 can be suppressed.

[0125] (Second Embodiment) Next, a second embodiment of the present invention will be described using the accompanying drawings. In this embodiment, an example will be described regarding the presence or absence of corrections to the amplitude and phase of the d-axis voltage command Vd* and the q-axis voltage command Vq* based on the number of synchronous PWM carriers Nc.

[0126] Figure 12This is a block diagram illustrating the functional configuration of the motor control device 1 according to the second embodiment of the present invention. The configuration of the motor control device 1 in this embodiment is the same as that described in the first embodiment. Figure 2 Compared to the previous embodiment, the current control unit 14 and the carrier frequency adjustment unit 16 are replaced with a current control unit 14A and a carrier frequency adjustment unit 16A, respectively. All other aspects are the same as in the first embodiment, and therefore will be omitted from the following description.

[0127] In addition to the functions of the carrier frequency adjustment unit 16 described in the first embodiment, the carrier frequency adjustment unit 16A also has the function of outputting the synchronous PWM carrier number Nc. The synchronous PWM carrier number Nc output from the carrier frequency adjustment unit 16A is input to the current control unit 14A.

[0128] Similar to the current control unit 14 described in the first embodiment, the current control unit 14A performs corrections on the d-axis voltage command Vd* and q-axis voltage command Vq* based on the value of Δθcarr during overmodulation in order to reduce the amplitude and phase errors of the inverter 3's output voltage, which vary according to the value of the carrier phase difference Δθcarr. At this time, the number of synchronous PWM carriers Nc input from the carrier frequency adjustment unit 16A is used to switch between the presence and absence of corrections for the d-axis voltage command Vd* and q-axis voltage command Vq*.

[0129] Figure 13 This is a block diagram of the carrier frequency adjustment unit 16A according to the second embodiment of the present invention. The carrier frequency adjustment unit 16A is the same as that in the first embodiment... Figure 3 Compared to the carrier frequency adjustment unit 16 described above, the synchronous PWM carrier number Nc selected by the synchronous PWM carrier number selection unit 161 is output from the carrier frequency adjustment unit 16A, and otherwise has the same configuration.

[0130] Figure 14 This is a block diagram of the current control unit 14A according to the second embodiment of the present invention. The current control unit 14A is similar to that in the first embodiment... Figure 8 Compared to the current control unit 14 described above, the switching unit 147 is replaced by the switching unit 147A. All other aspects have the same configuration as in the first embodiment.

[0131] The modulation rate H calculated by the modulation rate calculation unit 143 and the number of synchronous PWM carriers Nc output from the carrier frequency adjustment unit 16A are input to the switching unit 147A. The switching unit 147A selects any combination of the first d-axis voltage command Vd1* and the first q-axis voltage command Vq1* calculated by the d-axis current control unit 142a and the q-axis current control unit 142b, or the second d-axis voltage command Vd2* and the second q-axis voltage command Vq2* calculated by the correction voltage command calculation unit 146, based on these values. Then, the selected combination of the d-axis voltage command and the q-axis voltage command is output as the calculation result of the d-axis voltage command Vd* and the q-axis voltage command Vq* of the current control unit 14.

[0132] Specifically, for example, if at least one of the following conditions is met: the modulation rate H is 1.15 or less, and the number of synchronous PWM carriers Nc is a predetermined value or more, a combination of the first d-axis voltage command Vd1* and the first q-axis voltage command Vq1* is selected and output. On the other hand, if neither of these conditions is met, i.e., if the modulation rate H exceeds 1.15 and the number of synchronous PWM carriers Nc is less than a predetermined value, the combination of the first d-axis voltage command Vd1* and the first q-axis voltage command Vq1* is switched to a combination of the second d-axis voltage command Vd2* and the second q-axis voltage command Vq2*. Thus, in addition to the modulation rate H, the number of synchronous PWM carriers Nc, which represents the number of carriers per cycle of the modulated wave Vmod (three-phase voltage commands Vu*, Vv*, Vw*), is also considered to switch the presence or absence of amplitude and phase corrections for the d-axis voltage command Vd* and the q-axis voltage command Vq* output from the current control unit 14A. Furthermore, as mentioned above, the number of synchronous PWM carriers Nc is preferably set to a multiple of 3.

[0133] Even in the current control unit 14A of this embodiment, similar to the current control unit 14 described in the first embodiment, the rate of change of the voltage command before and after switching can be limited to a certain value or a hysteresis can be set for the switching unit 147A when the modulation rate H rises and falls. Furthermore, when switching from a combination of the first d-axis voltage command Vd1* and the first q-axis voltage command Vq1* to a combination of the second d-axis voltage command Vd2* and the second q-axis voltage command Vq2* according to the change of the synchronous PWM carrier number Nc, or vice versa, the amplitude and phase of the voltage command before and after switching can be made to change continuously. In this way, switching shocks in the output voltage of the inverter 3 can be prevented, and smooth motor control can be achieved.

[0134] According to the second embodiment of the present invention described above, the current control unit 14A switches whether or not to perform amplitude and phase corrections for the d-axis voltage command Vd* and the q-axis voltage command Vq* based on the synchronous PWM carrier number Nc, which represents the number of carriers per cycle of the voltage command. For example, when the synchronous PWM carrier number Nc is a predetermined integer multiple of 3 or more, it is preferable not to perform amplitude and phase corrections for the d-axis voltage command Vd* and the q-axis voltage command Vq*. Therefore, when the synchronous PWM carrier number Nc is sufficiently large, even if the carrier frequency fc is adjusted by changing the voltage phase error Δθv, the amplitude and phase errors in the output voltage of the inverter 3 are sufficiently small, and the amplitude and phase corrections for the d-axis voltage command Vd* and the q-axis voltage command Vq* can be omitted. As a result, the load on the motor control device 1 can be reduced.

[0135] (Third Embodiment) Next, the third embodiment of the present invention will be described with reference to the accompanying drawings.

[0136] Figure 15 This is a perspective view of the mechatronic unit 71 according to the third embodiment of the present invention. The mechatronic unit 71 is configured to include the motor drive system 100 (motor control device 1, motor 2, and inverter 3) described in the first and second embodiments. The motor 2 and the inverter 3 are connected at the coupling section 713 via a bus 712. The output of the motor 2 is transmitted to the axle via a gear 711 to a differential gear (omitted from the illustration). Furthermore, in Figure 15 The illustration of motor control device 1 is omitted, but motor control device 1 can be configured in any position.

[0137] The electromechanical unit 71 is characterized in that the motor 2, inverter 3, and gear 711 are integrated into a single structure. In the electromechanical unit 71, miniaturization due to this integrated structure is strongly desired, along with the requirement for high efficiency performance comparable to conventional designs. Therefore, by using the motor control device 1 described in the first and second embodiments, the modulation rate can be increased and DC voltage can be effectively utilized while freely changing the voltage phase error Δθv. Thus, the motor size can be miniaturized, thereby realizing a compact and highly efficient electromechanical unit.

[0138] (Fourth Embodiment) Next, using Figure 16 An embodiment in which the motor drive system 100 described in the first and second embodiments is applied to a vehicle will be described.

[0139] Figure 16 This is a configuration diagram of a hybrid vehicle system according to the fourth embodiment of the present invention. Figure 16As shown, the hybrid vehicle system of this embodiment has a power system that uses motor 2 as a motor / generator.

[0140] exist Figure 16 In the hybrid vehicle system shown, a front axle 801 is rotatably supported at the front of the vehicle body 800, and front wheels 802 and 803 are provided at both ends of the front axle 801. A rear axle 804 is rotatably supported at the rear of the vehicle body 800, and rear wheels 805 and 806 are provided at both ends of the rear axle 804.

[0141] A differential gear 811, which serves as a power distribution mechanism, is provided in the center of the front axle 801 to distribute the rotational driving force transmitted from the engine 810 via the transmission 812 to the left and right front axles 801.

[0142] The pulley on the crankshaft of the engine 810 and the pulley on the rotating shaft of the motor 2 are mechanically connected together by a belt.

[0143] Thus, the rotational driving force of motor 2 can be transmitted to engine 810, and the rotational driving force of engine 810 can be transmitted to motor 2. In motor 2, the three-phase AC power output from inverter 3, controlled by motor control device 1, is supplied to the stator coils of the stator, thereby causing the rotor to rotate and generating a rotational driving force corresponding to the three-phase AC power.

[0144] That is, the motor 2 is controlled by the motor control device 1 to operate as an electric motor, and on the other hand, it operates as a generator that receives the rotational driving force of the engine 810 to rotate the rotor, thereby inducing an electromotive force in the stator coil of the stator to generate three-phase alternating current.

[0145] Inverter 3 is a power conversion device that converts the DC power supplied from the high-voltage battery 5, which is a high-voltage (42V or 300V) power source, into three-phase AC power. It controls the three-phase AC current flowing to the stator coil of motor 2 according to the operating command value and the magnetic pole position of the rotor.

[0146] The three-phase AC power generated by motor 2 is converted into DC power by inverter 3 to charge high-voltage battery 5. High-voltage battery 5 is electrically connected to low-voltage battery 823 via DC-DC converter 824. Low-voltage battery 823 constitutes the low-voltage (14V) power supply of the car, used to power starter motor 825 for initial starting (cold start) of engine 810, radio, lights, etc.

[0147] When the vehicle is stopped at a traffic light or other stationary position (idle stop mode), the engine 810 is stopped. When starting again, the engine 810 is restarted (hot start), driven by the inverter 3 and the motor 2, to restart the engine 810. Furthermore, in idle stop mode, if the high-voltage battery 5 is undercharged or the engine 810 is not fully warmed up, the engine 810 continues to operate without stopping. Additionally, in idle stop mode, it is necessary to ensure the operation of auxiliary equipment such as the air conditioning compressor, which is powered by the engine 810. In this case, the motor 2 drives the auxiliary equipment.

[0148] In acceleration mode or high-load operation mode, motor 2 is also driven to assist the engine 810. Conversely, in charging mode that requires charging of the high-voltage battery 5, the motor 2 generates electricity through the engine 810 to charge the high-voltage battery 5. That is, regenerative modes such as when braking or decelerating the vehicle.

[0149] This is achieved using the motor drive system 100 described in the first and second embodiments. Figure 16 In hybrid electric vehicle systems, even when the magnet temperature of motor 2 exceeds a specified value, by changing the effective value of the line-to-line voltage, the DC voltage (in the case of a boost system), and the motor speed (in the case of an engine-generator), the absolute voltage value will not fall within the specified range, and high-order harmonic voltages twice the switching frequency will not be generated. As a result, eddy current losses in the rotor magnets can be reduced, thereby increasing the continuous power of the motors used in environmentally friendly vehicles such as electric or hybrid electric vehicles. In other words, it can increase the torque required for continuous driving, such as driving on inclines at high speeds.

[0150] Furthermore, the present invention is not limited to the above-described embodiments, and various modifications can be made without departing from the spirit of the present invention.

[0151] Symbol Explanation

[0152] 1…Motor control device, 2…Permanent magnet synchronous motor (motor), 3…Inverter, 4…Rotary position sensor, 5…High voltage battery, 7…Current detection component, 11…Current command generation unit, 12…Speed ​​calculation unit, 13…Three-phase / dq conversion current control unit, 14, 14A…Current control unit, 15…dq / three-phase voltage command conversion unit, 16, 16A…Carrier frequency adjustment unit, 17…Triangle wave generation unit, 18…Gate signal generation unit, 31…Inverter circuit, 32…PWM signal drive circuit, 33…Smoothing capacitor, 41…Rotary position detector, 71…Mechatronics unit, 100…Motor drive system, 141a, 141b…Subtraction unit, 142a…d-axis current control unit (IdACR), 142b…q-axis current control unit (IqACR), 143…Modulation rate calculation unit, 144…Amplitude / phase calculation unit, 145…Amplitude / phase… 146… Correction voltage command calculation unit, 147… Switching unit, 161… Synchronous PWM carrier number selection unit, 162… Voltage phase calculation unit, 163… Modulation rate calculation unit, 164… Voltage phase error calculation unit, 165… Synchronous carrier frequency calculation unit, 166… Carrier frequency setting unit, 711… Gear, 712… Busbar, 713… Coupling unit, 800… Car body, 801… Front axle, 802… Front wheel, 8 03…Front wheel, 804…Rear wheel axle, 805…Rear wheel, 806…Rear wheel, 810…Engine, 810a…Pulley, 811…Differential gear, 812…Transmission, 823…Low-voltage battery, 824…DC-DC converter, 825…Starter, 1641…Reference voltage phase calculation unit, 1644…Carrier triangular wave phase meter, 1645…Voltage phase difference conversion unit, 1646…Adder, 1647…Subtractor.

Claims

1. A motor control device connected to a power converter that performs power conversion from direct current to alternating current, and controlling the drive of an AC motor driven using the alternating current, the motor control device being characterized by comprising: The current control unit generates a voltage command corresponding to the torque command; The carrier generation unit generates the carrier. Carrier frequency adjustment unit, which adjusts the frequency of the carrier; and The gate signal generation unit uses the carrier wave to pulse-width modulate the voltage command, generating a gate signal for controlling the operation of the power converter. The carrier frequency adjustment unit adjusts the carrier frequency by changing the phase difference between the voltage command and the carrier. When the modulation rate exceeds a predetermined value, the current control unit corrects the amplitude and phase of the voltage command based on the phase of the carrier wave. The modulation rate corresponds to the voltage amplitude ratio of the DC power to the AC power. The current control unit corrects the amplitude and phase of the voltage command by changing the difference between the amplitudes of the first AC voltage and the second AC voltage, and the difference between the phases of the first AC voltage and the second AC voltage, to within a predetermined range. The first AC voltage is output from the power converter according to the voltage command before correction, with the phase difference set to a fixed value. The second AC voltage is output from the power converter according to the voltage command after correction, with the phase difference changed. The carrier frequency adjustment unit adjusts the carrier frequency by changing the phase difference based on the torque command and the rotational speed of the AC motor.

2. The motor control device according to claim 1, characterized in that, The carrier frequency adjustment unit changes the phase difference according to the torque command, the rotational speed, and the voltage amplitude ratio.

3. The motor control device according to claim 1, characterized in that, The carrier frequency adjustment unit adjusts the carrier frequency in such a way that the carrier frequency becomes an integer multiple of the voltage command frequency.

4. The motor control device according to claim 3, characterized in that, The current control unit determines whether to switch to the amplitude and phase of the corrected voltage command based on the number of carrier waves in each cycle of the voltage command.

5. The motor control device according to claim 4, characterized in that, When the number of carrier waves in each cycle of the voltage command is a predetermined integer multiple of 3 or more, the current control unit does not perform the correction.

6. The motor control device according to claim 4, characterized in that, When the number of carrier waves changes in each cycle of the voltage command, the current control section causes the amplitude and phase of the voltage command to change continuously.

7. The motor control device according to claim 1, characterized in that, The specified value is 1.

15.

8. The motor control device according to claim 1, characterized in that, The predetermined value when the modulation rate increases is set to a different value than the predetermined value when the modulation rate decreases.

9. A mechatronic unit, characterized in that, have: The motor control device according to any one of claims 1 to 8; The power converter is connected to the motor control device; The AC motor is driven by the power converter; and Gears, which transmit the rotational driving force of the AC motor. The AC motor, the power converter, and the gear are integrated into a single structure.

10. An electric vehicle system, characterized in that, have: The motor control device according to any one of claims 1 to 8; The power converter is connected to the motor control device; and The AC motor is driven by the power converter. The electric vehicle system uses the rotational driving force of the AC motor to move.

11. A motor control method comprising controlling the operation of a power converter that performs power conversion from direct current to alternating current, and controlling the driving of an AC motor driven using the alternating current, characterized in that... Generate a voltage command corresponding to the torque command. Generate carrier wave, The frequency of the carrier is adjusted by changing the phase difference between the voltage command and the carrier. A gate signal for controlling the operation of the power converter is generated by pulse-width modulating the voltage command using the carrier wave. In the generation of the voltage command, when the modulation rate exceeds a predetermined value, the amplitude and phase of the voltage command are corrected according to the phase of the carrier wave. The modulation rate corresponds to the voltage amplitude ratio of the DC power to the AC power. In the generation of the voltage command, the amplitude and phase of the voltage command are corrected by changing the difference between the amplitudes of the first AC voltage and the second AC voltage, and the difference between the phases of the first AC voltage and the second AC voltage, to within a specified range. The first AC voltage is output from the power converter according to the voltage command before correction, with the phase difference set to a fixed value. The second AC voltage is output from the power converter according to the voltage command after correction, with the phase difference changed. The frequency of the carrier wave is adjusted by changing the phase difference based on the torque command and the rotational speed of the AC motor.

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