A Modulation Method for a Three-Phase Four-Leg Three-Level Inverter to Eliminate Midpoint Current

Abstract

The invention discloses a modulation method for a three-phase four-leg three-level inverter to eliminate the neutral point current, belonging to the field of power electronic conversion and control. The method includes: calculating the differential-mode component of the modulation waves of the three-phase bridge legs and the modulation wave of the fourth bridge leg before common-mode voltage compensation according to the voltages of the AC filter capacitors of each phase and the currents of the AC filter inductors of each phase; calculating the duty ratio of the 0 level before correction of the three-phase bridge legs according to the differential-mode component of the modulation waves of the three-phase bridge legs and determining the phase where the maximum duty ratio is located; calculating the common-mode component of the modulation waves of the three-phase bridge legs according to the phase; calculating the modulation waves of the three-phase bridge legs and the fourth bridge leg according to the common-mode component of the modulation waves of the three-phase bridge legs, and determining the actual duty ratios of the +1, -1, and 0 levels of each bridge leg; the modulation waves of the three-phase bridge legs and the modulation wave of the fourth bridge leg are respectively intercepted with the stacked carrier waves to generate the driving signals of the inverter. This method can achieve the neutral point voltage balance in the full power factor range and eliminate the low-frequency pulsating component in the neutral point voltage.

Description

Technical Field

[0001] The present invention belongs to the field of power electronic conversion and control, and more specifically, relates to a modulation method for a three-phase four-leg three-level inverter to eliminate the neutral point current. Background Art

[0002] In order to overcome the deficiency of being unable to be compatible with unbalanced / nonlinear AC loads, a three-phase three-wire (3P3W) inverter can be connected to a neutral line to form a three-phase four-wire (3P4W) inverter. The 3P4W inverter is mainly divided into a three-phase three-leg four-wire (3P3L4W) inverter and a three-phase four-leg four-wire (3P4L4W) inverter. Compared with the 3P3L4W inverter, the neutral line current of the 3P4L4W inverter flows through the fourth leg instead of the split DC-side capacitor, which greatly reduces the volume of the bus capacitors of the 3P4L4W inverter and makes it more advantageous in power density.

[0003] For a three-level inverter, the balance of the neutral point voltage is a prerequisite for its safe and stable operation. Due to the influence of factors such as uneven capacitance values of the bus capacitors, inconsistent switching actions of power devices, and parasitic parameters, the neutral point voltage will inevitably deviate from the rated value. If not treated in time, the neutral point voltage may further diverge, even leading to the collapse and shutdown of the entire system. Therefore, the neutral point voltage balance control is crucial for the safe and stable operation of the three-phase four-leg three-level inverter.

[0004] The basic modulation methods of the 3P4L4W inverter mainly include three-dimensional space vector modulation (3D-SVM) and carrier-based pulse width modulation (CB-PWM). Similar to the traditional space vector modulation, 3D-SVM also realizes the neutral point voltage balance control by adjusting the action time of positive and negative redundant small vectors. The three-dimensionally distributed space vectors make the implementation process of 3D-SVM very complex; at the same time, due to the large number of vectors, it is particularly difficult to select appropriate action vectors. Compared with 3D-SVM, CB-PWM is intuitive, simple, and easy to implement. The common-mode components in the three-phase modulation waves can be directly controlled through a voltage equalization loop to balance the voltages of the upper and lower split capacitors. Although this basic neutral point voltage equalization control is simple, when the inverter operates under different working conditions, the gain sign of the voltage equalization control needs to be manually modified to ensure the balance of the bus capacitor voltages at any time; at the same time, when the inverter operates under the zero power factor condition, due to the influence of control delay and parasitic parameters, there will inevitably be a small phase disturbance between the phase voltage and the phase current, which makes the actual output power factor of the system fluctuate around zero. However, because the disturbance is unknown and unobservable, the system cannot correctly modify the voltage equalization gain sign according to the output power factor, resulting in a risk of failure of the voltage equalization control. If the disturbance once forms a positive feedback logic in the voltage equalization loop, the voltages of the split capacitors will rapidly diverge in a short time, leading to the collapse of the system.

[0005] Although the fourth arm bypasses the neutral current, when the inverter operates under high unbalance and low power factor conditions, the voltage ripple of the split capacitor on the DC side is still very large, and a capacitor with a very large capacitance is required to suppress it. This capacitor generally uses an electrolytic capacitor. The service life of the electrolytic capacitor is about 5,000 hours. When affected by factors such as ambient temperature, current ripple, and operating frequency, its service life will be further shortened. Therefore, the electrolytic capacitor has become the main factor affecting the reliability and life of the three-level three-phase four-arm inverter; at the same time, the relatively large volume of the electrolytic capacitor also restricts the further improvement of the inverter power density, which is not conducive to the development of system light weight and miniaturization.

[0006] Therefore, for the three-level three-phase four-arm inverter, how to overcome the defect of the traditional voltage balancing control failing under zero power factor conditions on the basis of carrier modulation to achieve the balance of the neutral point voltage within the full power factor range; and further reduce the voltage ripple of the split capacitor on the DC side to improve the reliability and power density of the system has important research significance. Summary of the Invention

[0007] In view of the defects of the existing technology and the improvement requirements, the present invention provides a modulation method for a three-level three-phase four-arm inverter that eliminates the neutral point current, aiming to achieve the safe and stable operation of the three-phase four-arm inverter under full power factor and further reduce the voltage ripple of the split capacitor on the DC side to improve the reliability and power density of the system.

[0008] To achieve the above object, according to one aspect of the present invention, there is provided a modulation method for a three-level three-phase four-arm inverter that eliminates the neutral point current. The inverter includes a three-level three-phase arm, a three-level fourth arm, an LC AC filter, and a neutral line inductor. The output end of the three-level three-phase arm is connected to the LC AC filter and is connected to the output end of the three-level fourth arm through the neutral line inductor from the neutral point of the AC filter capacitor. The method includes: calculating the differential mode component v Cfx of the modulation wave of the three-phase arm and the modulation wave v Lfx of the fourth arm before common mode voltage compensation according to the voltage v mxDM of each phase AC filter capacitor and the current i md0 of each phase AC filter inductor; calculating the duty cycle d mxDM of the 0 level before correction of each phase according to v zx0 , and calculating the common mode component v zx0 of the modulation wave of the three-phase arm according to the phase corresponding to the maximum value in d mCM ; adding v mxDM and v mCM to obtain the modulation wave v mx of the three-phase arm, and using v mCM to correct v md0Perform open-loop compensation of the common-mode voltage to obtain the modulation wave v of the fourth bridge arm md ; According to v mx and v md Calculate the duty cycles of the actual +1, -1, and 0 levels of each bridge arm; For each bridge arm: Use the duty cycles of the actual +1, -1, and 0 levels of this bridge arm to calculate the positive and negative modulation waves that intersect with the upper and lower stacked carriers in real time for this bridge arm. The positive and negative modulation waves intersect with the corresponding carriers to generate drive signals to drive this bridge arm.

[0009] Furthermore, d zx0 is:

[0010] d zx0 = 1 - |v mxDM |

[0011] where x is the phase among the a, b, and c phases.

[0012] Furthermore, when the phase corresponding to the maximum value in d zx0 is the a phase, v mCM is:

[0013] v mCM = -(v mbDM + v mcDM ) / 2

[0014] where v mbDM is the differential-mode component of the modulation wave of the b-phase bridge arm, and v mcDM is the differential-mode component of the modulation wave of the c-phase bridge arm.

[0015] Furthermore, when the phase corresponding to the maximum value in d zx0 is the b phase, v mCM is:

[0016] v mCM = -(v maDM + v mcDM ) / 2

[0017] where v maDM is the differential-mode component of the modulation wave of the a-phase bridge arm, and v mcDM is the differential-mode component of the modulation wave of the c-phase bridge arm.

[0018] Furthermore, when the phase corresponding to the maximum value in d zx0 is the c phase, v mCM is:

[0019] v mCM = -(v maDM + v mbDM ) / 2

[0020] where v maDM is the differential-mode component of the modulation wave of the a-phase bridge arm, vmbDM It is the differential mode component of the modulation wave of the B-phase bridge arm.

[0021] Furthermore, in the three-level three-phase bridge arm, the duty ratios of the actual +1, -1, and 0 levels of each phase bridge arm are as follows:

[0022] d zx = min{1 + v ma , 1 + v mb , 1 + v mc , 1 - v ma , 1 - v mb , 1 - v mc}

[0023] d px = (1 - d zx + v mx ) / 2

[0024] d nx = (1 - d zx - v mx ) / 2

[0025] Among them, d px , d nx , d zx are the duty ratios of the actual +1, -1, and 0 levels of the x-phase in the current switching cycle respectively; v ma , v mb , v mc are the modulation waves of the a, b, and c phase bridge arms respectively, and x is the phase among the a, b, and c phases.

[0026] Furthermore, the positive and negative modulation waves of each phase bridge arm intersecting with the upper and lower stacked carriers in real time are:

[0027] v mxp = d px

[0028] v mxn = -d nx

[0029] Among them, v mxp is the positive modulation wave of the x-phase bridge arm intersecting with the upper and lower stacked carriers in real time, and v mxn is the negative modulation wave of the x-phase bridge arm intersecting with the upper and lower stacked carriers in real time.

[0030] Furthermore, the duty ratios of the actual +1, -1, and 0 levels of the three-level fourth bridge arm are:

[0031] d zd = min{1 + v ma , 1 + v mb , 1 + v mc , 1 - v ma , 1 - vmb , 1 - v mc}

[0032] d pd = (1 - d zd + v md ) / 2

[0033] d nd = (1 - d zd - v md ) / 2

[0034] Wherein, d zd , d pd , d nd are respectively the duty cycles of the actual +1, -1, and 0 levels of the fourth leg of the three-level in the current switching period of the three-phase inverter.

[0035] Furthermore, the positive and negative modulation waves of the fourth leg of the three-level intersecting with the upper and lower stacked carriers in real time are:

[0036] v mdp = d pd

[0037] v mdn = -d nd

[0038] Wherein, v mdp is the positive modulation wave of the fourth leg of the three-level intersecting with the upper and lower stacked carriers in real time, and v mdn is the negative modulation wave of the fourth leg of the three-level intersecting with the upper and lower stacked carriers in real time.

[0039] According to one aspect of the present invention, a controller is provided for performing the modulation method of the three-phase four-leg three-level inverter for eliminating the neutral point current as described above.

[0040] Generally speaking, through the above technical solutions conceived by the present invention, the following beneficial effects can be achieved:

[0041] (1) A modulation method of a three-phase four-leg three-level inverter for eliminating the neutral point current is provided. For an inverter including a three-level three-phase leg and a three-level fourth leg, by adding -1 level and +1 level respectively in each switching period of the positive / negative half cycle of the three-level leg, an independently adjustable duty cycle of the 0 level is constructed, effectively realizing the decoupling control of the neutral point current and the load voltage. And this method is based on carrier modulation, with the characteristics of simple control and easy implementation;

[0042] (2) By managing the duty cycles of the 0 levels of the three phases to achieve zero neutral point current control, the problem of the failure of the traditional voltage equalization control under zero power factor conditions is solved from the root, making the neutral point voltage always balanced within the full power factor range, and fully improving the reliability of the system;

[0043] (3) Provided a method for suppressing the low-frequency pulsation of the neutral point voltage of a three-phase four-leg three-level inverter. By managing the zero-level duty ratios of the three phases, zero neutral point current control is achieved to eliminate the low-frequency ripple component of the neutral point current, thereby completely suppressing the low-frequency pulsation of the neutral point voltage, greatly reducing the capacitance value of the split capacitor on the DC side, and further improving the power density of the entire system. Description of the Drawings

[0044] Figure 1 Flowchart of the modulation method for a three-phase four-leg three-level inverter for eliminating the neutral point current provided by an embodiment of the present invention;

[0045] Figure 2 Working principle diagram of the three-phase four-leg three-level inverter provided by an embodiment of the present invention;

[0046] Figure 3 Topological structure diagram of the three-phase four-leg three-level inverter provided by an embodiment of the present invention;

[0047] Figure 4 Modulation waveform diagram of phase a of the modulation method provided by an embodiment of the present invention;

[0048] Figure 5A Bus capacitor voltage waveform of the three-phase four-leg three-level inverter at zero power factor when using the traditional carrier modulation method;

[0049] Figure 5B Bus capacitor voltage waveform of the three-phase four-leg three-level inverter at zero power factor when using the modulation method provided by an embodiment of the present invention;

[0050] Figure 6 Control schematic diagram of using the traditional carrier modulation method;

[0051] Figure 7A Working waveform of the three-phase four-leg three-level inverter under balanced linear load when using the traditional carrier modulation method;

[0052] Figure 7B Working waveform of the three-phase four-leg three-level inverter under unbalanced linear load when using the traditional carrier modulation method;

[0053] Figure 7C Working waveform of the three-phase four-leg three-level inverter under balanced non-linear load when using the traditional carrier modulation method;

[0054] Figure 7D Working waveform of the three-phase four-leg three-level inverter under unbalanced non-linear load when using the traditional carrier modulation method;

[0055] Figure 8AWhen the modulation method provided by the embodiment of the present invention is adopted, it is the working waveform of the three-phase four-leg three-level inverter under a balanced linear load;

[0056] Figure 8B When the modulation method provided by the embodiment of the present invention is adopted, it is the working waveform of the three-phase four-leg three-level inverter under an unbalanced linear load;

[0057] Figure 8C When the modulation method provided by the embodiment of the present invention is adopted, it is the working waveform of the three-phase four-leg three-level inverter under a balanced non-linear load;

[0058] Figure 8D When the modulation method provided by the embodiment of the present invention is adopted, it is the working waveform of the three-phase four-leg three-level inverter under an unbalanced non-linear load. Detailed implementation manners

[0059] In order to make the objectives, technical solutions and advantages of the present invention clearer and more understandable, the present invention will be further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain the present invention and are not used to limit the present invention. In addition, the technical features involved in the various embodiments of the present invention described below can be combined with each other as long as they do not conflict with each other.

[0060] In the present invention, terms such as "first" and "second" in the present invention and the accompanying drawings (if any) are used to distinguish similar objects and do not necessarily need to describe a specific order or sequence.

[0061] Figure 1 It is a flowchart of the modulation method of the three-phase four-leg three-level inverter for realizing zero neutral point current provided by the embodiment of the present invention. Refer to Figure 1 , combined with Figures 2 - 8D , the modulation method of the three-phase four-leg three-level inverter for eliminating the neutral point current in this embodiment is described in detail, and the method includes operations S1 - S5.

[0062] The inverter includes a three-level three-phase leg, a three-level fourth leg, an LC AC filter, and a neutral line inductor. The output end of the three-level three-phase leg is connected to the LC AC filter and is connected to the midpoint of the fourth leg from the neutral point of the AC filter capacitor through the neutral line inductor.

[0063] Refer to Figure 2 , which shows the working principle diagram of the three-phase four-leg three-level inverter, including a main power circuit and a control circuit. The topology of the main power circuit is as Figure 3 shown. Refer to Figure 3 , the main power circuit adopts a three-level three-phase leg, a three-level fourth leg, an LC AC filter (including a filter inductor and a filter capacitor), and a neutral line inductor L NComposition. Among them, the three-level bridge arm adopts a T-type three-level structure. The input ends of the three-phase bridge arms are connected to the DC bus, and the output ends are externally connected to an LC AC filter; the input end of the fourth bridge arm is also connected to the DC bus and is connected to the midpoint o of the split capacitor through Q d3 and Q d2 and is connected to the neutral point N of the three-phase filter capacitor through the neutral line inductor L N .

[0064] Operation S1, according to the voltage v of each phase AC filter capacitor Cfx and the current i of each phase AC filter inductor Lfx , calculate the differential-mode component v of the modulation wave of the three-phase bridge arm mxDM and the modulation wave v of the fourth bridge arm before common-mode voltage compensation md0 .

[0065] The filter inductor current includes the current i of inductor L fa , the current i of inductor L Lfa , the current i of inductor L fb , the current i of inductor L Lfb , the current i of inductor L fc , and the current i of inductor L Lfc . The filter capacitor voltage includes the voltage v of capacitor C fa , the voltage v of capacitor C Cfa , the voltage v of capacitor C fb , the voltage v of capacitor C Cfb , the voltage v of capacitor C fc , and the voltage v of capacitor C Cfc .

[0066] Specifically, amplify the difference between the differential-mode control loop command value and the differential-mode component in the filter capacitor voltage v Cfx , and subtract the differential-mode component in the filter inductor current i Lfx from the amplification result to obtain the differential-mode component v of the modulation wave of each phase bridge arm mxDM ; amplify the difference between zero and the common-mode component in the filter capacitor voltage v Cfx , and subtract the common-mode component in the filter inductor current i Lfx from the amplification result to obtain the modulation wave v of the fourth bridge arm before common-mode voltage compensation md0 .

[0067] Refer to Figure 2 , the differential-mode / common-mode variable extraction module extracts the differential-mode component v in the filter capacitor voltage v Cfx and the common-mode component v CfxDMfb , as well as extracts the differential-mode component i in the filter inductor current i CfCMfb and the common-mode component i in the filter inductor current i Lfx , x ∈ {a, b, c}. Set the differential-mode control loop command value as a three-phase symmetric sine wave v LfxDMfb , v LfCMfb , v refx , vrefx and v CfxDMfb The difference between them passes through the differential mode regulator G DM (s) After amplification, it is subtracted from i LfxDMfb to obtain v mxDM ; Set the common mode control loop command value to 0, and the difference between 0 and v CfxCMfb passes through the differential mode regulator G CM (s) After amplification, it is subtracted from i LfCMfb to obtain v md0 .

[0068] Operation S2, calculate the duty cycle d of the 0 level before correction for each phase according to v mxDM , and according to the phase corresponding to the maximum value in d zx0 , calculate the common mode component v of the modulation wave of the three-phase bridge arm zx0 . mCM .

[0069] According to the embodiment of the present invention, d zx0 is:

[0070] d zx0 = 1 - |v mxDM |

[0071] where x is the phase among a, b, and c phases.

[0072] According to the phase corresponding to the maximum value in d zx0 , calculate the common mode component v of the modulation wave of the three-phase bridge arm mCM , specifically divided into the following three cases:

[0073] 1) The phase corresponding to the maximum value in d zx0 is the a phase (d zb0 , d zc0 are both less than d za0 ), v mCM is:

[0074] v mCM = -(v mbDM + v mcDM ) / 2

[0075] where v mbDM is the differential mode component of the modulation wave of the b-phase bridge arm, and v mcDM is the differential mode component of the modulation wave of the c-phase bridge arm.

[0076] 2) The phase corresponding to the maximum value in d zx0 is the b phase (d za0 , d zc0 are both less than d zb0 ), v mCM is:

[0077] v mCM= -(v maDM + v mcDM ) / 2

[0078] where v maDM is the differential mode component of the modulation wave of phase a bridge arm, and v mcDM is the differential mode component of the modulation wave of phase c bridge arm.

[0079] 3) The phase corresponding to the maximum value in d zx0 is phase c (d za0 , d zb0 are both less than d zc0 ), and v mCM is:

[0080] v mCM = -(v maDM + v mbDM ) / 2

[0081] where v maDM is the differential mode component of the modulation wave of phase a bridge arm, and v mbDM is the differential mode component of the modulation wave of phase b bridge arm.

[0082] Operation S3: Add v mxDM and v mCM to obtain the three-phase bridge arm modulation wave v mx , and use v mCM to perform open-loop compensation of the common-mode voltage on v md0 to obtain the fourth bridge arm modulation wave v md .

[0083] Operation S4: Calculate the duty cycles of the actual +1, -1, and 0 levels of each bridge arm according to v mx and v md .

[0084] According to the embodiments of the present invention, in the three-level three-phase bridge arm, the duty cycles of the actual +1, -1, and 0 levels of each bridge arm are:

[0085] d zx = min{1 + v ma , 1 + v mb , 1 + v mc , 1 - v ma , 1 - v mb , 1 - v mc}

[0086] d px = (1 - d zx + v mx ) / 2

[0087] d nx = (1 - d zx - v mx ) / 2

[0088] Among them, d px , d nx , d zx are respectively the duty cycles of the actual +1, -1, and 0 levels of the x-phase in the current switching period; v ma , v mb , v mc are respectively the modulation waves of the a, b, and c phase legs, and x is the phase among the a, b, and c three phases.

[0089] In a three-level three-phase leg, the positive and negative modulation waves where each phase leg intersects with the upper and lower stacked carrier waves in real time are:

[0090] v mxp = d px

[0091] v mxn = -d nx

[0092] Among them, v mxp is the positive modulation wave where the x-phase leg intersects with the upper and lower stacked carrier waves in real time, and v mxn is the negative modulation wave where the x-phase leg intersects with the upper and lower stacked carrier waves in real time.

[0093] According to the embodiments of the present invention, the duty cycles of the actual +1, -1, and 0 levels of the three-level fourth leg are:

[0094] d zd = min{1 + v ma , 1 + v mb , 1 + v mc , 1 - v ma , 1 - v mb , 1 - v mc}

[0095] d pd = (1 - d zd + v md ) / 2

[0096] d nd = (1 - d zd - v md ) / 2

[0097] Among them, d zd , d pd , d nd are respectively the duty cycles of the actual +1, -1, and 0 levels of the three-level fourth leg in the current switching period.

[0098] The positive and negative modulation waves where the three-level fourth leg intersects with the upper and lower stacked carrier waves in real time are:

[0099] v mdp = dpd

[0100] v mdn =-d nd

[0101] wherein, v mdp is the positive modulation wave of the three-level fourth bridge arm intersecting with the upper and lower stacked carriers in real time, and v mdn is the negative modulation wave of the three-level fourth bridge arm intersecting with the upper and lower stacked carriers in real time.

[0102] In this embodiment, the relationship between the neutral point current i mid and each phase load current i ox is as follows:

[0103] i mid =d za i oa +d zb i ob +d zc i oc

[0104] The current i NPd injected into the neutral point by the fourth bridge arm and the neutral line current i N are related as follows:

[0105] i NPd =d zd i N

[0106] It should be noted that each phase load current i ox and the neutral line current i N always satisfy i oa +i ob +i oc =i N . Therefore, when d za =d zb =d zc =d zd , i NPd =i mid , achieving complete elimination of the neutral point current.

[0107] Operation S5: For each bridge arm, calculate the positive and negative modulation waves of the bridge arm intersecting with the upper and lower stacked carriers in real time using the duty cycles of the actual +1, -1, and 0 levels of the bridge arm, and intersect the positive and negative modulation waves with the corresponding carriers to generate drive signals to drive the bridge arm.

[0108] Specifically, v mxp , v mxn , v mdp , v mdn intersect with the stacked carriers respectively to generate drive signals to drive the inverter.

[0109] Figure 4 Figure a-phase modulation waveform diagram of the three-phase four-leg three-level inverter modulation method for eliminating the midpoint current in the embodiment of the present invention. Different from the traditional carrier modulation method, the output voltage of each leg in one switching period is a three-level waveform.

[0110] Taking Figure 3 the main parameter settings in the shown test example as an example to verify the effectiveness of this method. Rated capacity S = 10 kVA, DC input voltage V dc = 800 V, AC output voltage V Cfx = 220 V (RMS), output frequency f o = 50 Hz, switching frequency f s of the three-phase inverter = 10 kHz, switching frequency f sd of the fourth leg = 10 kHz, split DC-side capacitor C busp = C busn = 25 μF, filter inductor L fx = 2.4 mF, neutral line inductor L N = 2.4 mF, filter capacitor C f = 10 μF.

[0111] It should be noted that when the traditional carrier modulation method is adopted, C busp = C busn = 250 μF.

[0112] Under the above parameter settings, when the traditional carrier modulation method is adopted, the bus capacitor voltage waveform of the three-phase four-leg three-level inverter at zero power factor is as shown in Figure 5A , and the control principle of the traditional carrier modulation method is as shown in Figure 6 ; when the modulation method provided by the embodiment of the present invention is adopted, the bus capacitor voltage waveform of the three-phase four-leg three-level inverter at zero power factor is as shown in Figure 5B .

[0113] Comparing Figure 5A and Figure 5B , it can be seen that: when the inverter operates at a power factor of 0, when the traditional method is adopted, the bus capacitor voltage diverges, seriously threatening the normal operation of the system; while when the modulation method provided by the embodiment is adopted, the bus capacitor voltage converges near 400 V, and the voltage sharing effect is good.

[0114] Figure 7A , Figure 7B , Figure 7C , Figure 7DThe working waveforms of a three-phase four-leg three-level inverter under different loads when using the traditional carrier modulation method are respectively shown. It can be seen that when the three-phase four-leg three-level inverter is externally connected to an unbalanced load, although the split capacitor used is very large, its voltage fluctuation is still very high, up to 48%. Figure 8A 、 Figure 8B 、 Figure 8C 、 Figure 8D The working waveforms of a three-phase four-leg three-level inverter under different loads when using the modulation method provided by the embodiment of the present invention are respectively shown. Compared with the traditional modulation method, this method not only greatly reduces the capacitance value of the split capacitor on the DC side, but also eliminates the low-frequency pulsating component in the capacitor voltage under any load.

[0115] The embodiment of the present invention also provides a controller for executing the modulation method of the three-phase four-leg three-level inverter for eliminating the neutral point current.

[0116] Those skilled in the art can easily understand that the above are only the preferred embodiments of the present invention and are not intended to limit the present invention. Any modifications, equivalent replacements, and improvements made within the spirit and principle of the present invention shall be included in the protection scope of the present invention.

Claims

1. A three-phase four-bridge-arm three-level inverter modulation method for eliminating midpoint current, characterized in that: The inverter includes a three-level three-phase bridge arm, a three-level fourth bridge arm, an LC AC filter and a neutral line inductor, the output end of the three-level three-phase bridge arm is connected to the LC AC filter, and the output end of the three-level fourth bridge arm is connected from the neutral point of the AC filter capacitor through the neutral line inductor, and the method includes: According to the AC filter capacitor voltage v of each phase Cfx And each phase AC filter inductor current i Lfx , calculate the differential mode component v of the three-phase bridge arm modulation wave mxDM The fourth bridge arm modulation wave v before common mode voltage compensation md0 ; According to v mxDM Calculate the 0 level duty cycle d of each phase before correction zx0 , according to d zx0 The phase corresponding to the maximum value in the calculation of the common mode component v of the three-phase bridge arm modulation wave mCM ; Among them, d zx0 =1-|v mxDM |, x is one of the three phases a, b, and c; v mxDM and v mCM Add together to get the three-phase bridge arm modulation wave v mx , using v mCM v md0 Perform open-loop compensation of the common-mode voltage to obtain the fourth bridge arm modulation wave v md ; According to v mx and v md Calculate the duty cycle of the actual +1, -1, and 0 levels of each bridge arm; For each bridge arm: the duty cycle of the actual +1, -1, 0 levels of the bridge arm is used to calculate the positive and negative modulation waves that intersect the bridge arm with the upper and lower stacked carriers in real time. The positive and negative modulation waves intersect with the corresponding carriers to generate a driving signal to drive the bridge arm.

2. The method according to claim 1, characterized in that When zx0 When the phase corresponding to the maximum value is phase a, v mCM for: v mCM =-(v mbDM +v mcDM ) / 2 Among them, v mbDM is the differential mode component of the b-phase bridge arm modulation wave, v mcDM is the differential mode component of the C-phase bridge arm modulation wave.

3. The method according to claim 1, characterized in that When zx0 When the phase corresponding to the maximum value is phase b, v mCM for: v mCM =-(v maDM +v mcDM ) / 2 Among them, v maDM is the differential mode component of the a-phase bridge arm modulation wave, v mcDM is the differential mode component of the C-phase bridge arm modulation wave.

4. The method according to claim 1, characterized in that When zx0 When the phase corresponding to the maximum value is phase c, v mCM for: v mCM =-(v maDM +v mbDM ) / 2 Among them, v maDM is the differential mode component of the a-phase bridge arm modulation wave, v mbDM It is the differential mode component of the modulation wave of the b-phase bridge arm.

5. The method according to any one of claims 1 to 4, characterized in that: In the three-level three-phase bridge arm, the actual duty ratio of +1, -1, and 0 levels of each phase bridge arm is: d zx =min{1+v ma ,1+v mb ,1+v mc ,1-v ma ,1-v mb ,1-v mc } d px =(1-d zx +v mx ) / 2 d nx =(1-d zx -v mx ) / 2 Among them, d px ,d nx ,d zx are the actual +1, -1, and 0 level duty ratios of x phase in the current switching cycle; v ma 、v mb 、v mc They are the a, b, and c phase bridge arm modulation waves respectively, and x is the phase among the three phases a, b, and c.

6. The method according to claim 5, characterized in that The positive and negative modulation waves of each phase bridge arm and the upper and lower stacked carriers in real time are: v mxp =d px v mxn =-d nx Among them, v mxp is the positive modulation wave that is intercepted by the x-phase bridge arm and the upper and lower stacked carriers in real time, v mxn It is a negative modulation wave that is intercepted in real time by the x-phase bridge arm and the upper and lower stacked carriers.

7. The method according to any one of claims 1 to 4, characterized in that: The actual duty ratio of +1, -1, 0 level of the fourth bridge arm of the three-level is: d zd =min{1+v ma ,1+v mb ,1+v mc ,1-v ma ,1-v mb ,1-v mc } d pd =(1-d zd +v md ) / 2 d nd =(1-d zd -v md ) / 2 Among them, d zd ,d pd ,d nd They are respectively the actual +1, -1, and 0 level duty cycles of the third-level fourth bridge arm in the current switching cycle.

8. The method according to claim 7, characterized in that The positive and negative modulation waves of the third-level fourth bridge arm and the upper and lower stacked carriers in real time are: v mdp =d pd v mdn =-d nd Among them, v mdp is the positive modulation wave that is intercepted by the fourth bridge arm of the three-level and the upper and lower stacked carriers in real time, v mdn It is a negative modulation wave that is intercepted in real time by the fourth bridge arm of the three-level and the upper and lower stacked carriers.

9. A controller, characterized in that: A three-phase four-bridge-arm three-level inverter modulation method for eliminating midpoint current as described in any one of claims 1 to 8.

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