Distributed position sensorless control method and system for dual three-phase permanent magnet synchronous motors

Abstract

The present invention discloses a distributed position sensorless control method and system for a dual three-phase permanent magnet synchronous motor. The method comprises the following steps: k :1 distribution, decoupling the dual three-phase permanent magnet synchronous motor into two equivalent independent three-phase permanent magnet synchronous motor units, and then, for the two equivalent independent three-phase permanent magnet synchronous motor units, using the position sensorless control method preset by the independent control unit to completely independently control each three-phase permanent magnet synchronous motor unit without the need for real-time data communication between the control units of the two three-phase permanent magnet synchronous motor units. The present invention aims to achieve distributed decoupling control, using the proposed adaptive selective harmonic elimination filter to replace the low-pass filter in the sliding mode observer to achieve position sensorless control of the dual three-phase permanent magnet synchronous motor, and to improve the performance of the traditional sliding mode observer and enhance system stability.

Description

Technical Field

[0001] The present invention relates to the technical field of control of permanent magnet synchronous motors, and in particular to a distributed position sensorless control method and system for dual three-phase permanent magnet synchronous motors. Background Art

[0002] In recent years, environmental pollution and energy crises have become increasingly prominent. Traditional fossil fuels can no longer meet the needs of sustainable development, leading to widespread interest in renewable energy. Among the many new energy generation methods, offshore wind power offers enormous development potential. Compared to onshore wind power, offshore wind power boasts more abundant resources, higher efficiency, and lower costs. However, offshore wind power systems operate in harsh marine environments such as high humidity, salt spray, and long days. This, coupled with significant operational and maintenance challenges, places extremely high demands on their reliability and fault tolerance. In offshore wind power systems, multi-phase permanent magnet motors offer numerous advantages over three-phase permanent magnet motors. First, they offer lower per-phase current and reduced torque ripple, which improves system stability and efficiency. Second, they offer greater fault tolerance, better adapting to harsh environments and emergencies. These advantages have led to their widespread adoption in low-voltage, high-power offshore wind power systems.

[0003] Traditional distributed control architectures use multiple controllers to control multiple converters. These controllers receive current and rotor position information from their respective windings, enabling decoupled control between the windings of a multi-phase permanent magnet synchronous motor. In this distributed architecture, rotor position information for each three-phase winding unit is obtained from a common position sensor. While traditional distributed control architectures offer strong redundancy and reliability, position sensor communication failures are more susceptible to these failures in the harsh operating environment of offshore wind turbines. Such failures can affect the decoupled control between the windings of the multi-phase permanent magnet synchronous motor. To improve system performance, high-performance control strategies rely on sensors to obtain accurate information about rotor position and speed. Position sensors, such as encoders or resolvers, are mounted on the mechanical shaft. However, the use of these additional position sensing devices and the associated signal processing equipment for this position information increases costs. Indeed, not only must the cost of the position sensor itself be considered, but also the wiring, appropriate connectors and interfaces, and all associated costs, as well as the harsh environment, which reduces system reliability. Summary of the Invention

[0004] The technical problem to be solved by the present invention is as follows: In response to the above-mentioned problems in the prior art, a distributed position sensorless control method and system for a dual three-phase permanent magnet synchronous motor are provided. The present invention aims to realize distributed decoupling control, utilize the proposed adaptive selective harmonic elimination filter to replace the low-pass filter in the sliding mode observer, realize position sensorless control of the dual three-phase permanent magnet synchronous motor, and improve the performance of the traditional sliding mode observer to enhance the system stability.

[0005] In order to solve the above technical problems, the technical solution adopted by the present invention is:

[0006] A distributed position sensorless control method for a dual three-phase permanent magnet synchronous motor, comprising:

[0007] S101, the current of each three-phase winding of the dual three-phase permanent magnet synchronous motor is calculated according to k :1 distribution, where k is a positive real number, thereby decoupling the dual three-phase permanent magnet synchronous motor into two equivalent independent three-phase permanent magnet synchronous motor units as shown in the following formula:

[0008] ,

[0009] ,

[0010] In the above formula, and is the winding dq axis voltage of the first three-phase permanent magnet synchronous motor unit, is the stator resistance, and is the winding dq axis current of the first three-phase permanent magnet synchronous motor unit, and is the dq-axis inductance of the winding of the first three-phase permanent magnet synchronous motor unit, and is the dq-axis mutual inductance of the windings of the two three-phase permanent magnet synchronous motor units, is the electrical angular velocity of the motor, is the permanent magnet flux, and is the winding dq axis voltage of the second three-phase permanent magnet synchronous motor unit, and is the winding dq axis current of the second three-phase permanent magnet synchronous motor unit, and is the dq-axis inductance of the winding of the second three-phase permanent magnet synchronous motor unit, and , ;

[0011] S102, for the two equivalent independent three-phase permanent magnet synchronous motor units, without the need for real-time data communication between the control units of the two three-phase permanent magnet synchronous motor units, each three-phase permanent magnet synchronous motor unit is completely independently controlled by a position sensorless control method preset by an independent control unit.

[0012] In step S102, the position sensorless control method preset by the independent control unit is used to independently control each three-phase permanent magnet synchronous motor unit, including obtaining the estimated rotor position and the estimated electrical angular velocity according to the following steps:

[0013] S201, using a current observer to observe and obtain an estimated current of the αβ axis and an estimated back electromotive force of the αβ axis;

[0014] S202, filtering the estimated back EMF along the αβ axes to obtain the extended back EMF along the αβ axes;

[0015] S203 , using the αβ-axis extended back electromotive force to obtain an estimated rotor position and an estimated electrical angular velocity using an inverse tangent function method, or using a phase-locked loop (PLL) to obtain an estimated rotor position and an estimated electrical angular velocity.

[0016] Step S201 includes:

[0017] S401, construct an improved sliding mode observer based on an adaptive selective harmonic elimination filter with a harmonic extractor as shown in the following formula:

[0018] ,

[0019] In the above formula, and are the αβ axis current observation errors of the i-th three-phase permanent magnet synchronous motor unit, and are the α and β axis currents of the i-th three-phase permanent magnet synchronous motor unit, and are the α and β axis current observation values ​​of the i-th three-phase permanent magnet synchronous motor unit, For time, is the stator resistance, and are the total inductance of the dq axis respectively. The total inductance is the sum of the inductance component of the axis and the mutual inductance of the two three-phase permanent magnet synchronous motor units. is the estimated value of the electrical angular velocity of the i-th three-phase permanent magnet synchronous motor unit, is the electrical angular velocity of the i-th three-phase permanent magnet synchronous motor unit, and are the αβ-axis extended back EMF of the i-th three-phase permanent magnet synchronous motor unit, and Estimate the back EMF of the αβ axes of the i-th three-phase permanent magnet synchronous motor unit, Extracted by adaptive selective harmonic elimination filter ASHEF with harmonic extractor Axis estimated back EMF; Extracted by adaptive selective harmonic elimination filter ASHEF with harmonic extractor The function expression of the transfer function of the adaptive selective harmonic elimination filter ASHEF with harmonic extractor is:

[0020] ,

[0021] In the above formula, is the transfer function of the adaptive selective harmonic elimination filter ASHEF with harmonic extractor, s is the complex frequency, z is the original signal, To observe the signal, is the damping ratio coefficient, is the resonant frequency of the adaptive selective harmonic elimination filter ASHEF with harmonic extractor, for h Subharmonic signal;

[0022] S402, select the current error sliding mode surface shown in the following formula:

[0023] ,

[0024] In the above formula, and are the current error sliding mode surfaces of the α and β axes, is the gain coefficient, >0;

[0025] In step S403, when the system reaches the sliding surface, the derivative of the current error sliding surface is 0, so the super-helical sliding mode control STA algorithm is selected, and the estimated back electromotive force of the αβ axis is obtained according to the following formula:

[0026] ,

[0027] In the above formula, and is the sliding mode coefficient, is a sign function, and hour ; hour .

[0028] Optionally, filtering the αβ-axis estimated back EMF to obtain the αβ-axis extended back EMF in step S202 means filtering the αβ-axis estimated back EMF using an adaptive selective harmonic elimination filter ASHEF with a harmonic extractor to obtain the αβ-axis extended back EMF, and the function expression for filtering the αβ-axis estimated back EMF using an adaptive selective harmonic elimination filter ASHEF with a harmonic extractor is:

[0029] ,

[0030] In the above formula, and For the i The αβ-axis extended back EMF of a three-phase permanent magnet synchronous motor unit, is the damping ratio coefficient, is the resonant frequency of the adaptive selective harmonic elimination filter ASHEF, and For the i The estimated back EMF of the αβ axes of a three-phase permanent magnet synchronous motor unit is s, and s is the complex frequency.

[0031] Optionally, filtering the estimated αβ-axis back EMF to obtain the αβ-axis extended back EMF in step S202 refers to filtering the estimated αβ-axis back EMF using a low-pass filter LPF to obtain the αβ-axis extended back EMF. The function expression for filtering the estimated αβ-axis back EMF using a low-pass filter LPF is:

[0032] ,

[0033] In the above formula, and For the i The αβ-axis extended back EMF of a three-phase permanent magnet synchronous motor unit, and For the i Estimated back EMF of the αβ axis of a three-phase permanent magnet synchronous motor unit, is the cutoff frequency of the low-pass filter LPF, and s is the complex frequency.

[0034] In addition, the present invention also provides a distributed position sensorless control system for a dual three-phase permanent magnet synchronous motor, comprising a microprocessor and a memory connected to each other, wherein the microprocessor is programmed or configured to execute the distributed position sensorless control method for the dual three-phase permanent magnet synchronous motor.

[0035] In addition, the present invention also provides a computer-readable storage medium, which stores a computer program / instruction, and the computer program / instruction is programmed or configured to execute the distributed position sensorless control method of the dual three-phase permanent magnet synchronous motor through a processor.

[0036] In addition, the present invention also provides a computer program product, including a computer program / instruction, which is programmed or configured to execute the dual three-phase permanent magnet synchronous motor distributed position sensorless control method through a processor.

[0037] Compared with the prior art, the present invention has the following advantages: the distributed position sensorless control method of the dual three-phase permanent magnet synchronous motor of the present invention comprises the following steps: k :1 distribution, decoupling the dual three-phase permanent magnet synchronous motor into two equivalent independent three-phase permanent magnet synchronous motor units, and then for the two equivalent independent three-phase permanent magnet synchronous motor units, without the need for real-time data communication between the control units of the two three-phase permanent magnet synchronous motor units, using the position sensorless control method preset by the independent control unit to fully independently control each three-phase permanent magnet synchronous motor unit. The present invention can realize distributed decoupling control for the dual three-phase permanent magnet synchronous motor, utilize the proposed adaptive selective harmonic elimination filter to replace the low-pass filter in the sliding mode observer, realize position sensorless control of the dual three-phase permanent magnet synchronous motor, and improve the performance of the traditional sliding mode observer, thereby improving system stability. BRIEF DESCRIPTION OF THE DRAWINGS

[0038] Figure 1 Schematic diagram of the basic flow of the method of embodiment 1 of the present invention.

[0039] Figure 2 Schematic diagram of the basic principle of the position sensorless control method in Example 1 of the present invention.

[0040] Figure 3 Schematic diagram of the principle of the adaptive selective harmonic elimination filter ASHEF in the first embodiment of the present invention.

[0041] Figure 4 1 is a Bode diagram of the adaptive selective harmonic elimination filter ASHEF in the first embodiment of the present invention.

[0042] Figure 5 Schematic diagram of the basic principle of the position sensorless control method in Example 1 of the present invention.

[0043] Figure 6 Schematic diagram of the principle of ASHEF with harmonic extractor in Example 1 of the present invention.

[0044] Figure 7 Schematic diagram of the principle of the harmonic extractor in Example 1 of the present invention.

[0045] Figure 8 Schematic diagram of the principle of the low-pass filter LPF in the second embodiment of the present invention. DETAILED DESCRIPTION

[0046] Example 1:

[0047] like Figure 1 As shown, the distributed position sensorless control method for dual three-phase permanent magnet synchronous motors in this embodiment includes:

[0048] S101, the current of each three-phase winding of the dual three-phase permanent magnet synchronous motor is calculated according to k :1 distribution, where k is a positive real number, thereby decoupling the dual three-phase permanent magnet synchronous motor into two equivalent independent three-phase permanent magnet synchronous motor units as shown in the following formula:

[0049] , (1)

[0050] , (2)

[0051] In the above formula, and is the winding dq axis voltage of the first three-phase permanent magnet synchronous motor unit, is the stator resistance, and is the winding dq axis current of the first three-phase permanent magnet synchronous motor unit, and is the dq-axis inductance of the winding of the first three-phase permanent magnet synchronous motor unit, and is the dq-axis mutual inductance of the windings of the two three-phase permanent magnet synchronous motor units, is the electrical angular velocity of the motor, is the permanent magnet flux, and is the winding dq axis voltage of the second three-phase permanent magnet synchronous motor unit, and is the winding dq axis current of the second three-phase permanent magnet synchronous motor unit, and is the dq-axis inductance of the winding of the second three-phase permanent magnet synchronous motor unit, and , ;

[0052] S102, for the two equivalent independent three-phase permanent magnet synchronous motor units, without the need for real-time data communication between the control units of the two three-phase permanent magnet synchronous motor units, each three-phase permanent magnet synchronous motor unit is completely independently controlled by a position sensorless control method preset by an independent control unit.

[0053] The voltage equation of the dual three-phase permanent magnet synchronous motor in the dq coordinate system is:

[0054] , (3)

[0055] , (4)

[0056] In the above formula, is the electrical angular velocity of the motor; 、 and 、 are the d-axis and q-axis voltages of each set of windings; 、 and 、 are the d-axis and q-axis currents of each set of windings; 、 and 、 is the stator flux of each set of windings along the d-axis and q-axis. The flux equation of the dual three-phase permanent magnet synchronous motor in the dq coordinate system is:

[0057] , (5)

[0058] , (6)

[0059] In the above formula, 、 is the mutual inductance of the d-axis and q-axis between the three-phase windings, is the permanent magnet flux.

[0060] It can be seen from equations (5) and (6) that the magnetic flux between each set of three-phase windings of the dual three-phase permanent magnet synchronous motor is coupled with each other. Therefore, the dq axis voltage equations of the dual three-phase permanent magnet synchronous motor in equations (3) and (4) will contain the mutual inductance voltage generated by each three-phase sub-unit. Therefore, the dq coordinate transformation cannot achieve complete decoupling of the multi-phase permanent magnet synchronous motor. Its three-phase sub-unit model is inconsistent with the three-phase permanent magnet synchronous motor model. Therefore, the control method of the three-phase permanent magnet synchronous motor cannot be universal with the multi-phase permanent magnet synchronous motor, and the algorithm has poor portability. In response to the problem of coupling between the three-phase sub-units of the dual three-phase permanent magnet synchronous motor, this embodiment compares and analyzes the mathematical models of the dual three-phase permanent magnet synchronous motor and the three-phase permanent magnet synchronous motor, and proposes a model transformation method. After the model transformation, the dual three-phase permanent magnet synchronous motor is decoupled into two equivalent independent three-phase permanent magnet synchronous motor units. The model transformation method is as follows:

[0061] make , ,in is the mutual inductance between the three-phase sub-units. According to equations (3) to (6), the voltage equation of the dual three-phase permanent magnet synchronous motor can be simplified as follows:

[0062] , (7)

[0063] , (8)

[0064] By transforming the equations (7) and (8), we can obtain:

[0065] , (9)

[0066] , (10)

[0067] The current of each three-phase winding of the dual three-phase permanent magnet synchronous motor is k :1 allocation( k is a positive real number), we can get 、 Then, equations (9) and (10) can be simplified to obtain equations (1) and (2). Equations (1) and (2) show that the dual three-phase permanent magnet synchronous motor is decoupled into two equivalent independent three-phase permanent magnet synchronous motor units through model transformation. The model of each three-phase sub-unit is consistent with that of an ordinary three-phase permanent magnet synchronous motor. After the model transformation, the dual three-phase permanent magnet synchronous motor does not require real-time data communication between control units and can continue to use the three-phase permanent magnet motor control algorithm. It has strong scalability and can achieve completely independent control of each three-phase winding unit.

[0068] The previous article decoupled the dual three-phase permanent magnet synchronous motor into two equivalent independent three-phase permanent magnet synchronous motor units through model transformation. However, the two three-phase sub-units share a position encoder, and the rotor position signal needs to be synchronized in real time. When the position encoder fails, each three-phase sub-unit cannot work normally, and the reliability of the distributed control method is reduced. Therefore, this embodiment is based on the distributed control architecture proposed in the previous article, and the position sensorless control method is applied to the three-phase sub-units of the dual three-phase permanent magnet synchronous motor. After adopting position sensorless control, each three-phase sub-unit can obtain position information without the help of a position sensor, which can effectively improve the reliability of the system. Figure 2 As shown, in step S102 of this embodiment, the position sensorless control method preset by the independent control unit is used to fully independently control each three-phase permanent magnet synchronous motor unit, including obtaining the estimated rotor position and the estimated electrical angular velocity according to the following steps:

[0069] S201, using a current observer to observe and obtain an estimated current of the αβ axis and an estimated back electromotive force of the αβ axis;

[0070] S202, filtering the estimated back EMF along the αβ axes to obtain the extended back EMF along the αβ axes;

[0071] S203 , using the αβ-axis extended back electromotive force to obtain an estimated rotor position and an estimated electrical angular velocity using an inverse tangent function method, or using a phase-locked loop (PLL) to obtain an estimated rotor position and an estimated electrical angular velocity.

[0072] Therefore, step S201 in this embodiment includes:

[0073] S301, construct an integral sliding mode observer based on STA as shown in the following formula:

[0074] , (11)

[0075] In the above formula, and are the αβ axis current observation errors of the i-th three-phase permanent magnet synchronous motor unit, and are the α and β axis currents of the i-th three-phase permanent magnet synchronous motor unit, and are the α and β axis current observation values ​​of the i-th three-phase permanent magnet synchronous motor unit, For time, is the stator resistance, and are the total inductance of the dq axis respectively. The total inductance is the sum of the inductance component of the axis and the mutual inductance of the two three-phase permanent magnet synchronous motor units. is the estimated value of the electrical angular velocity of the i-th three-phase permanent magnet synchronous motor unit, is the electrical angular velocity of the i-th three-phase permanent magnet synchronous motor unit, and are the αβ-axis extended back EMF of the i-th three-phase permanent magnet synchronous motor unit, and are the estimated back EMF of the αβ axis of the i-th three-phase permanent magnet synchronous motor unit respectively; the derivation process of formula (11) is as follows: Each three-phase winding unit in the dual three-phase permanent magnet synchronous motor is identical and symmetrically distributed, let , , then the voltage equation of each three-phase subunit in Equations (9) and (10) can be simplified as:

[0076] , (12)

[0077] In the above formula, and For the i The d-axis and q-axis voltages of a three-phase permanent magnet synchronous motor unit; and For the i The d-axis and q-axis currents of a three-phase permanent magnet synchronous motor unit ( i =1,2). To simplify the equation, let , , then formula (12) can be simplified to:

[0078] , (13)

[0079] The state equation in the two-phase stationary coordinate system can be obtained from formula (13):

[0080] , (14)

[0081] Constructing a traditional STA-based sliding mode observer:

[0082] , (15)

[0083] Where, 、 is the sliding mode control function.

[0084] Subtracting equations (15) and (14) yields the STA-based integral sliding mode observer shown in (11) to obtain the stator current error;

[0085] S302, select the current error sliding mode surface shown in the following formula:

[0086] , (16)

[0087] In the above formula, and are the current error sliding mode surfaces of the α and β axes, is the gain coefficient, >0; In order to improve the convergence speed of the observer, the above-mentioned current error sliding surface adds the integral of the state quantity on the basis of the traditional sliding surface. The idea of ​​the integral sliding mode observer is to design a high-frequency switching gain to force the state to reach the integral sliding surface. Then the integral action in the sliding surface pushes the state to the desired equilibrium position in the presence of mismatched uncertainty. Even in the presence of uncertainty, the switching function can remain on the sliding surface. Since robustness cannot be guaranteed in the reaching stage in traditional SMO, the integral sliding mode observer eliminates the reaching stage by enforcing the sliding stage in the entire system response, ensuring sliding mode invariance from the beginning, reducing chattering, and ensuring fast and accurate tracking;

[0088] S303, when the system reaches the sliding surface, the derivative of the current error sliding surface is 0 ( ), thus selecting the super-helical sliding mode control STA algorithm, and obtaining the estimated back electromotive force of the αβ axis according to the following formula:

[0089] , (17)

[0090] In the above formula, and is the sliding mode coefficient, is a sign function, and hour ; hour .

[0091] The stability of the sliding mode observer based on STA has been proven by the Lyapunov stability condition:

[0092] , (18)

[0093] In this embodiment, the sliding mode coefficient in step S303 satisfies the following constraints:

[0094] , (19)

[0095] , (20)

[0096] In the above formula, is a positive real number.

[0097] When the state variable of the observer reaches the sliding surface, that is, 、 , the estimated speed is equal to the actual speed Then formula (11) can be further simplified as:

[0098] ,(twenty one)

[0099] From Equation (21), we can see that the sign function contains the fundamental information of the motor's back EMF. However, the sign function contains a large number of harmonics, so the traditional sliding mode observer needs to add an LPF to filter it to obtain the estimated extended back EMF. 、 , as shown below:

[0100] ,(twenty two)

[0101] In the above formula, is the low-pass filter cutoff frequency, which is related to the fundamental frequency of the motor's electrical angular velocity. Here, the low-pass filter needs to select an appropriate cutoff frequency to minimize the high-frequency switching signals in the observed back-EMF information while minimizing the phase delay and amplitude attenuation caused by the low-pass filter. After obtaining the observed back-EMF, the rotor position information can be estimated using the inverse tangent method or a phase-locked loop.

[0102] In traditional sliding mode observers, in order to obtain a smooth back EMF and suppress the chattering generated by the sign function, a low-pass filter is often used to filter the observed value. However, the low-pass filter will produce phase delay and amplitude attenuation, requiring phase compensation. In order to reduce phase shift and improve position observation accuracy, this embodiment proposes to use an adaptive selective harmonic elimination filter instead of a low-pass filter to filter the estimated back EMF. Specifically, step S201 of this embodiment includes:

[0103] S401, construct an improved sliding mode observer based on an adaptive selective harmonic elimination filter as shown in the following formula:

[0104] ,(twenty three)

[0105] In the above formula, and are the αβ axis current observation errors of the i-th three-phase permanent magnet synchronous motor unit, and are the α and β axis currents of the i-th three-phase permanent magnet synchronous motor unit, and are the α and β axis current observation values ​​of the i-th three-phase permanent magnet synchronous motor unit, For time, is the stator resistance, and are the total inductance of the dq axis respectively. The total inductance is the sum of the inductance component of the axis and the mutual inductance of the two three-phase permanent magnet synchronous motor units. is the estimated value of the electrical angular velocity of the i-th three-phase permanent magnet synchronous motor unit, is the electrical angular velocity of the i-th three-phase permanent magnet synchronous motor unit, and are the αβ-axis extended back EMF of the i-th three-phase permanent magnet synchronous motor unit, and Estimate the back EMF of the αβ axes of the i-th three-phase permanent magnet synchronous motor unit, Extracted for the adaptive selective harmonic elimination filter ASHEF Axis estimated back EMF; Extracted for the adaptive selective harmonic elimination filter ASHEF The transfer function of the traditional adaptive selective harmonic elimination filter ASHEF is expressed as follows:

[0106] ,(twenty four)

[0107] In the above formula, is the transfer function of the adaptive selective harmonic elimination filter ASHEF, s is the complex frequency, z is the original signal, To observe the signal, is the damping ratio coefficient, is the resonant frequency of the adaptive selective harmonic elimination filter ASHEF. Figure 3Figure 1 is a schematic diagram of the principle of the adaptive selective harmonic elimination filter. In the figure, z is the original signal, which contains the DC offset, fundamental component, and each harmonic. x is the observed fundamental component, and d is the DC offset. The transfer function of the DC offset d is:

[0108] , (25)

[0109] In the above formula, is an estimate of the DC bias.

[0110] As can be seen from formula (24), ASHEF is equivalent to a bandpass filter, which is characterized by an amplitude-frequency characteristic of 1 at the center frequency and a phase-frequency characteristic of 0, that is, there is no amplitude attenuation and phase lag at the center frequency. Therefore, the use of an adaptive selective harmonic elimination filter can achieve distortion-free filtering. In addition, as can be seen from formula (25), a notch filter and a low-pass filter in cascade can be used to extract the DC component. Therefore, the use of the adaptive selective harmonic elimination filter ASHEF in this embodiment can not only filter out other harmonics but also effectively suppress the DC bias part in the signal;

[0111] S402, select the current error sliding mode surface shown in the following formula:

[0112] , (26)

[0113] In the above formula, and are the current error sliding mode surfaces of the α and β axes, is the gain coefficient, >0;

[0114] In step S403, when the system reaches the sliding surface, the derivative of the current error sliding surface is 0, so the super-helical sliding mode control STA algorithm is selected, and the estimated back electromotive force of the αβ axis is obtained according to the following formula:

[0115] , (27)

[0116] In the above formula, and is the sliding mode coefficient, is a sign function, and hour ; hour .

[0117] From the previous analysis, we can see that compared with the low-pass filter, the adaptive selective harmonic elimination filter (ASHEF) has no phase delay and amplitude attenuation at the center frequency point, and can extract the estimated back-EMF fundamental component without generating a phase difference. Therefore, in order to improve the accuracy of position observation, the extracted estimated back-EMF fundamental information is introduced into the traditional STA-based sliding mode observer shown in Equation (15), and the improved sliding mode observer based on the adaptive selective harmonic elimination filter is obtained as shown in Equation (23). In the control of permanent magnet synchronous motors, the back-EMF frequency changes in real time with the electrical angular velocity. When using the adaptive selective harmonic elimination filter, its center frequency should be the same as the back-EMF frequency.

[0118] Figure 4 is the Bode diagram of the adaptive selective harmonic elimination filter ASHEF in this embodiment. Figure 4 It can be seen that the coefficient The value of will affect the filtering effect of ASHEF, thus affecting the extraction of estimated back EMF information. ASHEF has unity gain and no phase delay at the fundamental frequency. At other frequencies, the farther away from the center frequency, the more obvious the amplitude attenuation and phase delay are, and the better the filtering effect is. Therefore, the coefficient The value of can directly affect the filtering effect. The smaller the value, the smaller the bandwidth, and thus the better the filtering effect on estimating the back EMF. However, the smaller the bandwidth, the worse the dynamic performance of the system. On the contrary, if The larger the value is, the worse the filtering effect on the estimated back EMF becomes, but the wider the bandwidth is, the better the dynamic response of the system becomes.

[0119] Although the adaptive selective harmonic elimination filter ASHEF can effectively filter out harmonics, when the harmonic frequency is close to the fundamental frequency, the harmonic suppression capability will be weakened, and low-order harmonics are difficult to be filtered out. In this embodiment, based on the first embodiment, the adaptive selective harmonic elimination filter ASHEF is further improved to an adaptive selective harmonic elimination filter ASHEF with a harmonic extractor, referred to as ASHEF with a harmonic extractor. Figure 5 shown.

[0120] In order to accurately filter out the harmonic components, the harmonic extractor used in this embodiment is as follows Figure 7 As shown in Figure 2, the harmonic extractor has infinite gain at the harmonic frequency and can effectively extract the harmonic components. Its transfer function is:

[0121] , (30)

[0122] In the above formula, h is the order of the harmonic.

[0123] The transfer function of ASHEF with harmonic extractor is:

[0124] , (29)

[0125] In the above formula, is the transfer function of the adaptive selective harmonic elimination filter ASHEF, s is the complex frequency, z is the original signal, To observe the signal, is the damping ratio coefficient, is the resonant frequency of the adaptive selective harmonic elimination filter ASHEF, where for h Subharmonic signal, where h The values ​​can be set as needed, for example, in this embodiment, the values ​​are 5 and 7. Figure 6 As shown in Figure 1, the harmonic extractor is connected to the input signal e of the feedback loop. The feedback loop of the harmonic extractor has the characteristics of a notch filter at the harmonic frequency, which can effectively filter out specific frequency harmonics. As can be seen from its transfer function, the harmonic extractor affects the denominator of the transfer function. First, it affects the fundamental frequency. The magnitude and phase response of the transfer function are not affected. However, the magnitude of the transfer function is affected at certain harmonic frequencies. The harmonic components are zero at 0 and have a band-stop response to the harmonic components. The working principle of ASHEF with harmonic extractor is as follows: Based on the adaptive selective harmonic elimination filter ASHEF, it uses the fifth harmonic extractor and the seventh harmonic extractor to extract the harmonic signals from the input signal e respectively, and then sums them to obtain the total harmonic signal extracted by the harmonic extractor. The first deviation between the original signal z and the total harmonic signal extracted by the harmonic extractor is calculated to replace the original original signal z as the input of the adaptive selective harmonic elimination filter ASHEF.

[0126] The above analysis shows that, compared to the traditional sliding-mode observer, the improved sliding-mode observer based on ASHEF with a harmonic extractor suppresses the harmonic components of the estimated back-EMF without causing phase delay or amplitude attenuation. Furthermore, ASHEF with a harmonic extractor utilizes a harmonic frequency extractor structure, effectively filtering out the fifth and seventh harmonics of the back-EMF, reducing position observation error and improving position observation accuracy.

[0127] In step S202 of this embodiment, filtering the estimated αβ-axis back EMF to obtain the extended αβ-axis back EMF means filtering the estimated αβ-axis back EMF using an adaptive selective harmonic elimination filter ASHEF with a harmonic extractor to obtain the extended αβ-axis back EMF. The function expression for filtering the estimated αβ-axis back EMF using an adaptive selective harmonic elimination filter ASHEF with a harmonic extractor is:

[0128] , (28)

[0129] In the above formula, and For the i The αβ-axis extended back EMF of a three-phase permanent magnet synchronous motor unit, is the gain coefficient, is the resonant frequency of the adaptive selective harmonic elimination filter ASHEF with harmonic extractor, and For the i The estimated back EMF of the αβ axes of a three-phase permanent magnet synchronous motor unit is s, and s is the complex frequency.

[0130] In addition, this embodiment also provides a dual three-phase permanent magnet synchronous motor distributed position sensorless control system, including a microprocessor and a memory connected to each other, and the microprocessor is programmed or configured to execute the dual three-phase permanent magnet synchronous motor distributed position sensorless control method.

[0131] In addition, this embodiment also provides a computer-readable storage medium, which stores a computer program / instruction, and the computer program / instruction is programmed or configured to execute the distributed position sensorless control method of the dual three-phase permanent magnet synchronous motor through a processor.

[0132] In addition, this embodiment also provides a computer program product, including a computer program / instruction, which is programmed or configured to execute the dual three-phase permanent magnet synchronous motor distributed position sensorless control method through a processor.

[0133] Example 2:

[0134] This embodiment is basically the same as the first embodiment, and the main differences are: Figure 8 As shown, in step S202 of this embodiment, filtering the estimated αβ-axis back EMF to obtain the extended αβ-axis back EMF means filtering the estimated αβ-axis back EMF using a low-pass filter LPF to obtain the extended αβ-axis back EMF. The function expression for filtering the estimated αβ-axis back EMF using a low-pass filter LPF is:

[0135] , (31)

[0136] In the above formula, and For the i The αβ-axis extended back EMF of a three-phase permanent magnet synchronous motor unit, and For the i Estimated back EMF of the αβ axis of a three-phase permanent magnet synchronous motor unit, is the cutoff frequency of the low-pass filter LPF, and s is the complex frequency.

[0137] In addition, this embodiment also provides a dual three-phase permanent magnet synchronous motor distributed position sensorless control system, including a microprocessor and a memory connected to each other, and the microprocessor is programmed or configured to execute the dual three-phase permanent magnet synchronous motor distributed position sensorless control method.

[0138] In addition, this embodiment also provides a computer-readable storage medium, which stores a computer program / instruction, and the computer program / instruction is programmed or configured to execute the distributed position sensorless control method of the dual three-phase permanent magnet synchronous motor through a processor.

[0139] In addition, this embodiment also provides a computer program product, including a computer program / instruction, which is programmed or configured to execute the dual three-phase permanent magnet synchronous motor distributed position sensorless control method through a processor.

[0140] Those skilled in the art will appreciate that the embodiments of the present application may be provided as methods, systems, or computer program products. Therefore, the present application may take the form of a completely hardware embodiment, a completely software embodiment, or an embodiment combining software and hardware aspects. Moreover, the present application may take the form of a computer program product implemented on one or more computer-readable storage media (including but not limited to disk storage, CD-ROM, optical storage, etc.) containing computer-usable program code. The present application is described with reference to the flowcharts and / or block diagrams of the methods, devices (systems), and computer program products according to the embodiments of the present application. It should be understood that each process and / or box in the flowchart and / or block diagram, as well as the combination of processes and / or boxes in the flowchart and / or block diagram, may be implemented by computer program instructions. These computer program instructions may be provided to a processor of a general-purpose computer, a special-purpose computer, an embedded processor, or other programmable data processing device to produce a machine, so that instructions executed by the processor of the computer or other programmable data processing device generate instructions for implementing the functions described in the process. Figure 1 a process or multiple processes and / or boxes Figure 1 These computer program instructions can also be stored in a computer-readable memory that can guide a computer or other programmable data processing device to work in a specific way, so that the instructions stored in the computer-readable memory produce a product including the instruction device, which implements the function specified in the process. Figure 1 a process or multiple processes and / or boxes Figure 1 These computer program instructions can also be loaded onto a computer or other programmable data processing device, so that a series of operation steps are executed on the computer or other programmable device to produce a computer-implemented process, thereby providing instructions for implementing the process in the process. Figure 1 a process or multiple processes and / or boxes Figure 1 A step that specifies a function in one or more boxes.

[0141] The above description is merely a preferred embodiment of the present invention. The scope of protection of the present invention is not limited to the above embodiment. All technical solutions based on the concept of the present invention are within the scope of protection of the present invention. It should be noted that for those skilled in the art, various improvements and modifications that do not depart from the principles of the present invention should also be considered within the scope of protection of the present invention.

Claims

1. A distributed position sensorless control method for dual three-phase permanent magnet synchronous motors, characterized in that: include: S101, the current of each three-phase winding of the dual three-phase permanent magnet synchronous motor is calculated according to k :1 distribution, where k is a positive real number, thereby decoupling the dual three-phase permanent magnet synchronous motor into two equivalent independent three-phase permanent magnet synchronous motor units as shown in the following formula: ， ， In the above formula, and is the winding dq axis voltage of the first three-phase permanent magnet synchronous motor unit, is the stator resistance, and is the winding dq axis current of the first three-phase permanent magnet synchronous motor unit, and is the dq-axis inductance of the winding of the first three-phase permanent magnet synchronous motor unit, and is the dq-axis mutual inductance of the windings of the two three-phase permanent magnet synchronous motor units, is the electrical angular velocity of the motor, is the permanent magnet flux, and is the winding dq axis voltage of the second three-phase permanent magnet synchronous motor unit, and is the winding dq axis current of the second three-phase permanent magnet synchronous motor unit, and is the dq-axis inductance of the winding of the second three-phase permanent magnet synchronous motor unit, and , ; S102: for the two equivalent independent three-phase permanent magnet synchronous motor units, independently controlling each three-phase permanent magnet synchronous motor unit using a position sensorless control method preset by the independent control unit without requiring real-time data communication between the control units of the two three-phase permanent magnet synchronous motor units; In step S102, the position sensorless control method preset by the independent control unit is used to independently control each three-phase permanent magnet synchronous motor unit, including obtaining the estimated rotor position and the estimated electrical angular velocity according to the following steps: S201, using a current observer to observe and obtain an estimated current of the αβ axis and an estimated back electromotive force of the αβ axis; S202, filtering the estimated back EMF along the αβ axes to obtain the extended back EMF along the αβ axes; S203, using the arc tangent function method to obtain the estimated rotor position and the estimated electrical angular velocity of the αβ-axis extended back EMF, or using a phase-locked loop (PLL) to obtain the estimated rotor position and the estimated electrical angular velocity; Step S201 includes: S401, construct an improved sliding mode observer based on an adaptive selective harmonic elimination filter as shown in the following formula: ， In the above formula, and are the αβ axis current observation errors of the i-th three-phase permanent magnet synchronous motor unit, and are the α and β axis currents of the i-th three-phase permanent magnet synchronous motor unit, and are the α and β axis current observation values ​​of the i-th three-phase permanent magnet synchronous motor unit, For time, is the stator resistance, and are the total inductance of the dq axis respectively. The total inductance is the sum of the inductance component of the axis and the mutual inductance of the two three-phase permanent magnet synchronous motor units. is the estimated value of the electrical angular velocity of the i-th three-phase permanent magnet synchronous motor unit, is the electrical angular velocity of the i-th three-phase permanent magnet synchronous motor unit, and are the αβ-axis extended back EMF of the i-th three-phase permanent magnet synchronous motor unit, and Estimate the back EMF of the αβ axes of the i-th three-phase permanent magnet synchronous motor unit, Extracted by adaptive selective harmonic elimination filter ASHEF with harmonic extractor Axis estimated back EMF; Extracted by adaptive selective harmonic elimination filter ASHEF with harmonic extractor The function expression of the transfer function of the adaptive selective harmonic elimination filter ASHEF with harmonic extractor is: ， In the above formula, is the transfer function of the adaptive selective harmonic elimination filter ASHEF with harmonic extractor, s is the complex frequency, z is the original signal, To observe the signal, is the damping ratio coefficient, is the resonant frequency of the adaptive selective harmonic elimination filter ASHEF with harmonic extractor, for h Subharmonic signal; S402, select the current error sliding mode surface shown in the following formula: ， In the above formula, and are the current error sliding mode surfaces of the α and β axes, is the gain coefficient, >0; In step S403, when the system reaches the sliding surface, the derivative of the current error sliding surface is 0, so the super-helical sliding mode control STA algorithm is selected, and the estimated back electromotive force of the αβ axis is obtained according to the following formula: ， In the above formula, and is the sliding mode coefficient, is a sign function, and hour ; hour .

2. The distributed position sensorless control method for dual three-phase permanent magnet synchronous motors according to claim 1, characterized in that: In step S202, filtering the estimated αβ-axis back EMF to obtain the αβ-axis extended back EMF means filtering the estimated αβ-axis back EMF using an adaptive selective harmonic elimination filter ASHEF with a harmonic extractor to obtain the αβ-axis extended back EMF. The function expression for filtering the estimated αβ-axis back EMF using an adaptive selective harmonic elimination filter ASHEF with a harmonic extractor is: ， In the above formula, and For the i The αβ-axis extended back EMF of a three-phase permanent magnet synchronous motor unit, is the damping ratio coefficient, is the resonant frequency of the adaptive selective harmonic elimination filter ASHEF with harmonic extractor, and For the i The estimated back EMF of the αβ axes of a three-phase permanent magnet synchronous motor unit is s, and s is the complex frequency.

3. The distributed position sensorless control method for dual three-phase permanent magnet synchronous motors according to claim 1, characterized in that: In step S202, filtering the estimated αβ-axis back EMF to obtain the extended αβ-axis back EMF means filtering the estimated αβ-axis back EMF using a low-pass filter LPF to obtain the extended αβ-axis back EMF. The function expression for filtering the estimated αβ-axis back EMF using a low-pass filter LPF is: ， In the above formula, and For the i The αβ-axis extended back EMF of a three-phase permanent magnet synchronous motor unit, and For the i Estimated back EMF of the αβ axis of a three-phase permanent magnet synchronous motor unit, is the cutoff frequency of the low-pass filter LPF, and s is the complex frequency.

4. A distributed position sensorless control system for a dual three-phase permanent magnet synchronous motor, comprising a microprocessor and a memory connected to each other, characterized in that: The microprocessor is programmed or configured to execute the distributed position sensorless control method for dual three-phase permanent magnet synchronous motors according to any one of claims 1 to 3.

5. A computer-readable storage medium having a computer program / instruction stored therein, characterized in that: The computer program / instruction is programmed or configured to execute the distributed position sensorless control method for dual three-phase permanent magnet synchronous motors according to any one of claims 1 to 3 through a processor.

6. A computer program product comprising a computer program / instructions, characterized in that The computer program / instruction is programmed or configured to execute the distributed position sensorless control method for dual three-phase permanent magnet synchronous motors according to any one of claims 1 to 3 through a processor.

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