Transmitter and receiver, and method thereof, for transmitting and receiving symbols on a time-varying channel with Doppler spread.

Abstract

A communication frame for an Orthogonal Time-Frequency Space (OTFS) transmission system includes at least one first type block and at least one second type block. The first type block comprises data signals arranged in a two-dimensional pattern along the delay and Doppler domains, wherein at least one data signal has a superimposed pilot signal. The second type block comprises data signals arranged in a two-dimensional pattern along the delay and Doppler domains, which may or may not have a superimposed pilot signal. At least one second type block is surrounded by a first type block in the delay domain, and one or more first type blocks surrounding the second type block have at least one identical data symbol and an associated identical superimposed pilot symbol at the same position in the two-dimensional pattern. An OTFS transmitter generates and transmits the communication frame, and a receiver utilizes its characteristics for oscillator frequency offset compensation and channel estimation.

Description

Technical Field

[0001] The present invention relates to a method for transmitting and receiving symbols on an orthogonal time-frequency space (OTFS) communication channel affected by Doppler spread, as well as a transmitter and receiver for implementing the method.

[0002] definition

[0003] Throughout this specification, bold symbols denote vectors or matrices. Superscripts T, H, and... These represent the transpose, complex conjugate transpose, and pseudoinverse of a vector or matrix, respectively. diag{a} is a diagonal matrix with vector a on its diagonal, while diag{A} is a vector whose elements come from the diagonal of matrix A. It is the Kronecker product. Background Technology

[0004] Sixth-generation (6G) wireless communication and subsequent technologies are expected to serve a large number of high-speed mobile users, such as vehicles, subways, highways, trains, drones, and low Earth orbit (LEO) satellites.

[0005] The preceding fourth-generation and fifth-generation (5G) wireless communications employed Orthogonal Frequency Division Multiplexing (OFDM) technology, which boasts high spectral efficiency and robustness against frequency-selective fading channels, and also allows for the use of low-complexity equalizers. However, high-speed mobile communications suffer from severe time and frequency dispersion due to velocity-dependent Doppler shift or spread and rapidly varying multipath reception. Both time and frequency dispersion contribute to receiver signal fading, hence this type of fading is also known as biselective channel fading. Biselective channel fading severely impairs the performance of OFDM communications.

[0006] As an alternative to OFDM, OTFS modulation has been proposed as a solution to address dual-selective fading channels.

[0007] OTFS modulation is a 2D modulation scheme that multiplexes information QAM symbols onto carrier waveforms. These waveforms correspond to local pulses in the signal representation, known as delayed-Doppler representation. OTFS waveforms are distributed in both time and frequency, but they remain approximately orthogonal to each other under the influence of the universal delayed-Doppler channel. Theoretically, OTFS combines the reliability and robustness of spread spectrum with the high spectral efficiency and low complexity of narrowband transmission.

[0008] Figure 1A block diagram of a general OTFS transmission system is shown. Transmitter 200 includes a first transmitter-side conversion unit 202 and a second transmitter-side conversion unit 204. Serial binary data is input to a signal mapper (not shown), which outputs a two-dimensional sequence x[k,l] of information symbols, where QAM symbols are arranged along the delay period and Doppler period of the delayed Doppler domain.

[0009] The use of delayed Doppler channel representation has advantages due to its compactness and sparsity. Since there are typically only a small number of physical reflectors and their associated reflected signals, the number of parameters required for channel modeling and estimation in the delayed Doppler domain is far less than the number required in the time-frequency domain.

[0010] A two-dimensional sequence of information symbols x[k,l] is input to the first transmitter-side transformation unit 202 and subjected to an inverse finite symmetric Fourier transform (iSFFT) to generate a matrix X[n,m], which represents the two-dimensional sequence of information symbols x[k,l] in the time-frequency domain. When the transmitter transmits in the time domain, further transformation is required in the second transmitter-side transformation unit 204 to generate a signal s[t] in the time domain, for example, using a Heisenberg transform. Then, the signal s[t] is transmitted over the communication channel via antenna 206.

[0011] In real-world environments, transmitted signals, traveling from the transmitter to the receiver via a communication channel, are subject to dual-selective fading with Doppler spread. The received signal is a superposition of a direct copy of the transmitted signal and multiple reflected copies, each delayed by a path delay that depends on the length of the signal path delay, and shifted by a Doppler shift that depends on the differential speed between the transmitter, reflectors, and receiver. Each signal copy is weighted according to its specific path delay and differential speed. Typical Doppler shifts are approximately 10 Hz–1 kHz, but larger values ​​can occur in highly mobile environments (e.g., high-speed trains) and / or high carrier frequencies. Because multiple reflectors and / or moving reflectors are likely to be present in real-world environments, the received superimposed signal is dispersed across a frequency range, not just a frequency shift; therefore, the signal distortion is also known as Doppler shift. In the following description, the real-world communication channel is also referred to as the actual communication channel.

[0012] exist Figure 1 In this context, the actual communication channel is represented by uninterrupted radio waves emitted by transmitter antenna 206 and various disordered radio waves from different directions and at different distances from each other at receiver antenna 302. Radio waves may arrive at the receiver antenna directly, or they may arrive at the receiver antenna after being reflected once or multiple times at one or more stationary and / or moving objects, which may introduce Doppler shift and different delays to the reflected radio waves.

[0013] Receiver 300 picks up the received signal r[t] in the time domain and provides it to the first receiver-side transformation unit 304, where it undergoes a Wigner transform to transform the received signal r[t] into a matrix Y[n,m] representing the received signal r[t] in the time-frequency domain. To enable signal detection in the delayed Doppler domain, the matrix Y[n,m] is then provided to the second receiver-side transformation unit 306, where it undergoes a finite sine Fourier transform (SFFT). The SFFT outputs a two-dimensional sequence of information symbols y[k,l] in the delayed Doppler domain. This two-dimensional sequence of information symbols y[k,l] is input to the channel estimation and equalization block 310, which performs channel estimation (CE) and signal detection (SD) and reconstructs the initially transmitted symbols. These symbols are ultimately input to a demapper to output the initially transmitted binary data (the demapper is not shown in the figure).

[0014] To perform channel estimation at the receiver, the transmitter may add pilots. These pilots are known at the receiver and are located at known positions in the two-dimensional sequence of information symbols ultimately transmitted. However, the pilots replace the positions of the data symbols and do not carry any data, which leads to a reduction in the system's spectral efficiency.

[0015] Some known OTFS receivers utilize the characteristics of delayed Doppler channel representation and employ a basis extended model (BEM) to parameterize the time-varying channel in OTFS as a weighted combination of multiple basis functions. The advantage of this approach is that BEM can help reduce the number of unknown channel coefficients that need to be estimated.

[0016] There are various types of BEMs, including complex exponential BEM (CE-BEM), generalized CE-BEM (GCE-BEM), noncritical sampling CE-BEM (NCS-CE-BEM), polynomial BEM, discrete prolate spheroidal (DPS) BEM, Karhunen-Loeve BEM (KL-BEM), and spatiotemporal BEM.

[0017] Among them, CE-BEM is the simplest model, but it suffers from significant modeling errors. On the other hand, CE-BEM and its variants GCE-BEM and NCS-CE-BEM do not rely on channel statistical properties. GCE-BEM is simple and analyzable. To approach optimal performance, its BEM order should be at least greater than 1, i.e., T≥2, where T is the modeling resolution parameter. Specifically, when T=1, GCE-BEM suffers from large modeling errors, while when T>1, although the modeling error is smaller, a higher BEM order and complexity are required.

[0018] In known OTFS receivers that use complex exponential basis spread modeling for channel estimation, pilot overhead must increase with the maximum channel delay and Doppler spread to achieve acceptable performance, thus reducing spectral efficiency. Although many OTFS channels may have known maximum channel delays, and even known maximum Doppler spreads, practical system designs account for even higher maximum delays and Doppler spreads to provide a safety margin. This further reduces the spectral efficiency of such practical systems.

[0019] Spectral efficiency can be improved by using superimposed pilots (SPs) and freeing up space for data symbols. The SPs are low-power pilots that are superimposed on the data symbols in the delayed Doppler domain. The data symbols and the superimposed pilots are then further transformed into the final transmitted OTFS signal vector x.

[0020] Figure 2 A general schematic diagram of pilot signals superimposed in a transmitted frame is shown. For example... Figure 2 As shown in the left portion, pilots can be arranged across the entire plane of a two-dimensional sequence of information symbols, which are arranged along the delay and Doppler periods of the delayed Doppler domain. Pilots with power much lower than that of the information symbols are represented by an ordered checkerboard pattern, indicating that the pilots are known in advance at the receiver. Data is represented by a random pattern, indicating the variability of the transmitted data. Power allocation is represented by the distance to the delayed Doppler plane. Figure 2 The right side shows an exemplary power allocation for pilot and data symbols. It can be seen that the power of the pilot is much lower than that of the data.

[0021] In practical applications, there are constraints on the transmit power, which needs to cover the needs of both data and pilot transmission. That is, the data symbols and pilots need to share the total available power of the transmitter.

[0022] In subsequent chapters, M and N represent the dimensions of the transmission frames in the delay grid and Doppler grid, respectively, where the symbol arrangement is located. The transmitted complex-valued OTFS vector x can be represented in the delayed Doppler domain as a superimposed pilot vector x. sp With data vector x d The sum of these is defined as:

[0023] x sp =[x sp [0,0],x sp [0,1],…,x sp [0,M-1],…,x sp [N-1,0],x sp [N-1,1],…,x sp [N-1,M-1]] T ,and

[0024] x d =[x d [0,0],x d [0,1],…,x d [0,M-1],…,x d [N-1,0],x d [N-1,1],…,x d [N-1,M-1]] T .

[0025] Define P T Let α be the total transmit power, α (α∈(0,1)) be the pilot power allocation ratio, and αP be the total transmit power. T and (1-α)P T These are used for transmitting pilot signals and data symbols, respectively. Therefore, the transmitted OTFS signal vector x can be represented as:

[0026]

[0027] Generally, if more power is allocated to pilot transmission, i.e., a larger α, better channel estimation performance can be expected. However, less power is allocated to data transmission, resulting in a lower signal-to-noise ratio (SNR) and thus lower reliability. Conversely, if less power is allocated to pilots, i.e., a smaller α, poorer channel and signal estimation will result. Therefore, appropriate power allocation between data and pilots is crucial for achieving high reliability.

[0028] In the delayed Doppler domain, the received signal vector y can be represented as:

[0029]

[0030] Among them, b BEM,q and c q These are the q-th BEM basis function and coefficient, respectively, where Q represents the relationship with f. D The relevant BEM order, F MN is the Discrete Fourier Transform (DFT) matrix, w is the additive white Gaussian noise vector, and z is the received OTFS signal error due to BEM modeling. It is worth noting that the above equation is equivalent to:

[0031]

[0032] The received signal vector y is represented as the data vector y. d The vector y represents the superimposed pilot frequencies. spThe combination of these factors, along with noise and errors, allows data symbols to be viewed as interference in channel estimation. Unlike pilot signals, data signals can be highly variable and may exhibit fairly random characteristics, which can be further enhanced by appropriately arranging data symbols in a two-dimensional OTFS transmission frame, especially when repetitive structures exist in the data signal.

[0033] While the superiority of OFTS modulation under ideal conditions is now undisputed, its practical limitations in cost-effective applications may hinder its widespread adoption, particularly with carrier frequency offset (CFO).

[0034] CFO, or carrier frequency mismatch, is caused by the Doppler effect and radio frequency (RF) equipment components. CFO caused by the Doppler effect is often called Doppler shift or Doppler spread, while CFO caused by the local crystal oscillator (XO) is called oscillator frequency offset (OFO).

[0035] There are three main types of crystal oscillators: free-running XO, temperature-controlled XO (TCXO), and oven-controlled XO (OCXO). Free-running XOs are the cheapest, but have the largest frequency error, for example, between ±10 and ±20 ppm, where ppm represents parts per million. OCXOs can reduce the frequency error to 0.0015 ppm, but the equipment cost is extremely high, about 2000 times that of free-running XOs, and also increases power costs. TCXOs strike a good balance between free-running XOs and OCXOs. They are much cheaper than OCXOs, approximately five times the cost, and can reduce the frequency error to ±1.5 ppm.

[0036] OFO is typically much larger than the Doppler shift or spread. For example, a TCXO with a frequency error of ±1.5ppm at a carrier frequency of 4GHz can result in an OFO of ±6kHz, which is more significant than the 0.5 to 2kHz Doppler shift at speeds between 125 and 500 km / h. It is worth noting that the Doppler shift added to the OFO further increases the maximum Doppler shift at the receiver.

[0037] Therefore, in the presence of OFO Doppler frequency shift spread, it is necessary to optimize the performance of the OTFS communication system in terms of spectral efficiency, performance, and reliability. Summary of the Invention

[0038] The present invention aims to solve this problem by providing a communication frame with dual-rate data, as defined in the appended claims, capable of estimating and compensating for large OFO (e.g., greater than 0.5 ppm) in an OTFS receiver, particularly in OTFS transmissions with low pilot overhead and high spectral efficiency. The term "dual-rate" refers to blocks of low-rate and high-rate data within the communication frame, where the low-rate data is particularly suitable for in-band signaling transmission in both the control and user planes. In-band signaling can be used to transmit UE-specific control information and common control information. Other claims of the invention relate to a Time-Frequency Distortion Resilient OTFS (TFDR-OTFS) transmitter and a receiver for a corresponding OTFS transmission system, respectively for transmitting and receiving binary data sequences in the communication frame according to the invention, for high-speed mobile communication over OTFS communication channels with long delay spread and Doppler spread, particularly in the presence of OFO. Further claims relate to methods for transmitting and receiving data symbols, wherein the receiving method includes using the proposed dual-rate data communication frame in an OTFS receiver to estimate and compensate for OFO. Further claims relate to a wireless device comprising an OTFS transmission system including the proposed receiver and / or transmitter, and a computer program product. Advantageous embodiments and extensions are given in the corresponding dependent claims.

[0039] In the following sections, an OTFS analysis system model considering OFO will be presented, followed by a communication frame, transmitter, and receiver for the OTFS transmission system used in the method of this invention. Finally, the proposed TFDR-OTFS receiver with dual-rate data and SP (superimposed pilot) will be discussed in detail.

[0040] Among the various types of BEMs described above, KL-BEM is considered the most accurate BEM model when there is a good understanding of the channel statistical characteristics. However, its performance is not optimal when the assumed channel characteristics do not match the actual channel, and it is also not optimal when there is residual OFO after OFO estimation and compensation. Therefore, this invention relies on GCE-BEM instead of KL-BEM. This invention further uses superimposed pilots for initial channel estimation and uses detected symbols as additional pseudo-pilots in repeated iterative channel estimation. Superimposed pilots can be used only for a portion of the communication frame or for the entire communication frame. The specific construction of the communication frame using first and second class blocks allows for OFO estimation at the receiver.

[0041] Similar to the OTFS system model discussed above, in the OTFS system model including OFO, N and M represent the dimensions of the delay grid and Doppler grid in which the symbols are arranged, respectively. Since the observation of OFO is also applicable to dedicated pilots and superimposed pilots, the following discussion adopts the general representation of the OTFS signal vector. The transmitted complex OTFS matrix x is defined as...

[0042] x=[x[0,0],x[0,1],...,x[0,M--1],...,x[N-1,0],x[N-1,1],...,x[N-1,M-1]] T

[0043] The received OTFS matrix y in the delayed Doppler domain is defined as

[0044] y=[y[0,0],y[0,1],..,y[0,M-1],...,y[N-1,0],y[N-1,1],...,,y[N-1,M-1]] T

[0045] H t It is an MN×MN time-varying channel matrix in the time domain, where the Jakes model is considered, and the maximum Doppler frequency is expressed as f. D . φ(φ∈[-e ofo e ofo )) is defined as OFO, where e ofo Let be the maximum frequency error of the XO, expressed in ppm. In the presence of OFO, the received OTFS signal vector y can be written as...

[0046]

[0047] in Let F be the OFO matrix. N Let I be the DFT (Discrete Fourier Transform) matrix. M Let H be an M×M identity matrix, and w be an additive white Gaussian noise vector. This is achieved through the channel model H... t Using GCE-BEM (Generalized Complex Exponential Basis Extended Modeling), for example, as described in German Patent Publication No. DE 10 2022 106 409A1, the entire contents of which are incorporated herein by reference, y is further denoted as

[0048]

[0049] Where b q and c q These are the q-th GCE-BEM basis function and coefficient, respectively, where Q represents the relationship with f. D The relevant GCE-BEM order, F MNIt is a DFT matrix, z mod It is a GCE-BEM modeling error.

[0050] Next, by considering Complex Exponential Basis Extended Modeling (CE-BEM)—with a BEM resolution of 1—the element-wise input-output relationship between x[k,l] and y[k,l] is derived. At the transmitter, after applying the inverse symplectic finite Fourier transform (iSFFT) and the Heisenberg transform, the time-domain symbol s[n,m] is written as...

[0051]

[0052] Let h[t, l′] be defined as the channel gain of the l′-th path (l′ = 0, 1, ..., L) at time t (t = 0, 1, ..., MN-1), where L represents the channel length. After propagation through a biselective fading channel, the received time-domain symbol r[n, l] is represented as...

[0053]

[0054] Where the assumption e[n,l] is the time-domain modeling error caused by GCE-BEM modeling. At the receiver, after performing SFFT and Wigner transform, the received symbol y[k,l] in the delayed Doppler domain is given by the following equation.

[0055]

[0056] Where z[k,l] represents the modeling error in the delayed Doppler domain caused by GCE-BEM modeling. Compared to an ideal system without OFO, this presents two problems:

[0057] -i) More symbols in the Doppler field interfere with each other, thus resulting in stronger inter-Doppler interference, and

[0058] ii) By introducing an additional exponential term This is used to change the phase of the received signal.

[0059] Therefore, without properly addressing the OFO problem, OTFS modulation will lose its superior performance in high-mobility communications. Furthermore, the presence of OFO makes OTFS channel estimation more challenging.

[0060] Reference Figure 3 Suppose a scenario where the carrier frequency f c Set to 4GHz, with a speed of v = 125km / h, the frequency error of XO is ±1.5ppm. The maximum Doppler frequency is calculated as f. D =500Hz, OFO frequency is f OFO = ±6KHz. Figure 3An example of a possible Doppler spectrum is depicted, where f D =500Hz, f OFO = -6kHz, 0, 6kHz. It's easy to see that OFO shifts the Doppler spectrum to the left or right, thus increasing the maximum Doppler frequency to f. D +f OFO Among known OTFS channel estimators, such as those discussed in German Patent Publication DE 10 2022 106 409A1 or German Patent Publication DE 10 2021 126 321A1, a new maximum Doppler frequency (i.e., f) is required. D +f OFO Regenerate subspace or basis extended modeling (BEM) basis functions. Since the frequency shift caused by OFO is much larger than that caused by the Doppler effect, the number of required subspace or BEM basis functions will increase significantly, resulting in high pilot overhead, higher pilot power, and high computational complexity in OTFS frames with dedicated pilots. Therefore, it is crucial to estimate and compensate for OFO before implementing the OTFS channel estimator.

[0061] The present invention achieves OFO estimation and final compensation by using a specific arrangement of first and second class blocks in the transmission frame, and by a specific arrangement of symbols in the first class blocks, especially by having at least some of the same symbols in the first class blocks immediately before and after the second class blocks.

[0062] Figure 4 Figure a) shows a first exemplary representation of data symbols in a first and second class blocks of a communication frame. The repetition of at least one data symbol at the same position in the leading and trailing first class (or low-rate) blocks is represented by a rectangular cross-shading pattern, while the completely random nature of the data symbols in the second class (or high-rate) blocks is represented by a pseudo-random pattern. The communication frame has a length of M in the delay domain and a "height" of N in the Doppler domain. It should be noted that the pilot symbols in the first class blocks are also identical. The distance between identical columns in subsequent first class blocks is D. At least two first class blocks and one or more second class blocks are provided within a communication frame. Therefore, the length of the second class blocks may vary. Figure 4 In a), the first type of blocks are arranged around the center of the transmission frame, while the second type of blocks are placed at the front, middle and back of the transmission frame.

[0063] Figure 4 b) illustrates a second exemplary arrangement of the first and second class blocks in a communication frame. Here, the first and second class blocks are placed alternately, i.e., a first class block is followed by a second class block, then another first class block, and finally another second class block.

[0064] Figure 4c) illustrates a third exemplary arrangement of the first and second class blocks in a communication frame. This arrangement corresponds to Figure 4 The arrangement in b) is reversed, but the positions of the first and second blocks are swapped.

[0065] Figure 4 d) illustrates a fourth exemplary arrangement of the first and second class blocks in a communication frame. Here, the first class blocks are placed at the beginning and end of the transmission frame, separated by a second class block.

[0066] The length L+1 of the first type of block depends on the longest path the signal travels in the channel. Channel matrix H t The channel length is indicated in the figure below.

[0067] As mentioned above, the received OTFS vector y in the delayed Doppler domain is defined as...

[0068] y=[y[0,0],y[0,1],…,y[0,M-1],…,y[N-1,0],y[N-1,1],…,y[N-1,M-1]] T .

[0069] Considering the use of superimposed pilots (SP) in the transmission frame, after propagation through a dual-selective fading channel with Doppler spread, the received signal vector y can be regarded as the sum of the vectors representing the received data and the superimposed pilots in the delayed Doppler domain, and can be expressed as:

[0070]

[0071] Among them, F N It is the Discrete Fourier Transform (DFT) matrix, I M It is an M×M identity matrix, w is an additive white Gaussian noise (AWGN) vector, and H t It is an MN×MN time-varying channel matrix in the time domain, defined as:

[0072]

[0073] Where h[t,l] represents the channel gain of the l-th path at time t, t=0,1,…,MN-1, l=0,1,…,L, and L represents the channel length. Maximum Doppler frequency. Where f c Here, v is the carrier frequency, v is the vehicle speed, and c is the speed of light. Based on the Jakes model and the U-shaped Doppler spectrum, the correlation function for the l-th path is defined as J0(2πnf). max T s ), where J0(·) represents the zeroth-order Bessel function, T s It is the sampling period.

[0074] The ratio between the spacing D between identical columns of the first type of block and the length M of the communication frame in the delay dimension separating the low-rate blocks and the high-rate blocks determines the estimable OFO range. Since the distance D can be adjusted according to the needs of the transmission system, and thanks to the use of SP, a wide and flexible OFO estimation range can be supported with only a moderate reduction in spectral efficiency. This allows for the use of lower-cost free-running crystal oscillators (XO) in both transmitters and receivers, thereby reducing equipment costs.

[0075] The oscillator frequency offset (OFO) can be estimated by utilizing the autocorrelation of the first type of block of low-rate data (which contains superimposed pilot SP) between two consecutive received low-rate data in an OTFS frame.

[0076] Define g k =[y[k,L],y[k,M-1]] T The autocorrelation matrix R is calculated as follows: Next, eigenvalue decomposition is performed on R to obtain a signal eigenvector u of length 2. Let u1 and u2 be the first and last elements of u, respectively. Due to the constant phase shift characteristic, u1 and u2 satisfy the following relationship: Therefore, OFO can be easily estimated as:

[0077]

[0078] Therefore, OFO estimation uses a closed-form solution and requires no additional pilot signal. It should be noted that the complexity of the OFO estimation process is 4N.

[0079] By using OFO estimation results Incorporating GCE-BEM basis functions b q The new GCE-BEM basis functions are obtained:

[0080]

[0081] Considering (Q+1) BEM basis functions, the OFO compensation method has a linear complexity of MN(Q+1). Therefore, the above system model can be remodeled as follows:

[0082]

[0083] Furthermore, in the delayed Doppler domain, the received signal vector y of the data and the superimposed pilot can be expressed as:

[0084]

[0085] After the initial OFO estimation and compensation, a small amount of OFO will remain due to BEM modeling errors and noise, which will be addressed in the joint estimation of the residual OFO and the communication channel.

[0086] Therefore, in a first aspect of the invention, a dual-rate communication frame for an OTFS transmission system is provided. The communication frame includes at least one first-class block and at least one second-class block. The one or more first-class blocks include data signals arranged in a two-dimensional manner along the delay and Doppler domains. At least one of the data signals in the one or more first-class blocks has a superimposed pilot signal. It should be noted that typically the entire block is superimposed with a pilot signal. The size of the one or more first-class blocks is N×(L+1). The one or more second-class blocks include data signals arranged in a two-dimensional manner along the delay and Doppler domains, or contain data signals arranged in a two-dimensional manner along the delay and Doppler domains, wherein at least one data signal has a superimposed pilot signal. In other words, the second-class blocks may or may not have superimposed pilots. In the delay domain, at least one second-class block of the communication frame is surrounded by a first-class block in front of and behind it. The first-class blocks surrounding the second-class blocks in front of and behind it have at least one identical data symbol and an associated identical superimposed pilot symbol at the same position in the two-dimensional arrangement. It should be noted that typically the first-class blocks surrounding the second-class blocks in front of and behind it are the same. The identical elements of the first-class blocks that surround the second-class blocks in front and behind allow or facilitate highly reliable initial oscillator frequency offset (OFO) estimation. Using the same symbols in the first-class blocks before and after the second-class blocks reduces the effective data rate of the first-class blocks; therefore, the first-class blocks can be referred to as low-rate blocks, and the second-class blocks as high-rate blocks.

[0087] Figure 4 e) illustrates the first example of superimposed pilots in the proposed dual-rate OTFS frame with superimposed pilots (SP), where both the data symbols of the first and second class blocks have superimposed pilot symbols. The arrangement / arrangement of the first and second class blocks corresponds to... Figure 4 The arrangement is shown in d). Low-rate data symbols are displayed with a crisscrossing shading pattern to emphasize their identical repetition in the preamble and trailing first-class blocks. High-rate data symbols are displayed with a pseudo-random pattern to emphasize the randomness of the data. Pilot symbols are displayed with a regular checkerboard pattern to emphasize the receiver's prior knowledge of the pilot symbols. Pilot symbols cover the entire communication frame, depending on the arrangement used here. Figure 4 The arrangement of the first and second type blocks shown in d) indicates that the frame contains two first type blocks (or low-rate blocks), which are placed at the front and back of the OTFS frame along the delay dimension, surrounding the second type blocks. As mentioned earlier, the first type blocks are used for initial OFO estimation.

[0088] Figure 4f) shows a second example of the proposed dual-rate OTFS frame with SP, where only the data symbols of the first class block have superimposed pilot symbols. Figure 4 d) Similarly, low-rate data symbols are displayed with a crisscrossing shading pattern to emphasize their identical repetition in the preamble and tail of the first-class blocks. High-rate data symbols are displayed with a pseudo-random pattern to emphasize the randomness of the data. Pilot symbols are displayed with a regular checkerboard pattern to emphasize the receiver's prior knowledge of the pilot symbols. Clearly, pilot symbols only cover the first-class blocks or low-rate blocks. Data symbols in the second-class blocks do not have superimposed pilots, so full power can be allocated to these data symbols.

[0089] It should be noted that, compared to the first example, the pilot symbols in the second example may require higher power, and the convergence performance may differ. Furthermore, the corresponding superimposed pilot arrangement can be used... Figure 4 Any of the other exemplary arrangements of the first and second class blocks shown in a) to 4c).

[0090] Therefore, in one or more embodiments, the length of the second block arranged between the leading and trailing first blocks is variable in the delay domain.

[0091] According to a second aspect of the invention, a transmitter of an OTFS transmission system includes a signal mapper disposed upstream of a first transmitter-side conversion unit and a second transmitter-side conversion unit. The signal mapper is adapted to receive a binary data sequence and output a two-dimensional arrangement / two-dimensional array of data symbols and / or data symbols with superimposed pilot symbols in a two-rate communication frame in the delayed Doppler domain according to the first aspect of the invention described above. The first transmitter-side conversion unit is adapted to receive the two-dimensional two-rate communication frame in the delayed Doppler domain output from the signal mapper at an input and output a two-dimensional arrangement of information symbols in the time-frequency domain. The two-dimensional arrangement of information symbols in the time-frequency domain includes and represents both data symbols and data symbols with pilot symbols (SP). The output of the first transmitter-side conversion unit is provided to the input of the second transmitter-side conversion unit, which is adapted to output a continuous time-domain signal representing the communication frame for transmission over a communication channel.

[0092] In one or more embodiments, the first transmitter-side transform unit is adapted to perform precoding and / or inverse symplectic finite Fourier transform.

[0093] In one or more embodiments, the second transmitter-side transform unit is adapted to perform the Heisenberg transform or the inverse finite Fourier transform (IFFT).

[0094] In one or more embodiments, the transmitter is arranged to allocate 50% to 99% (preferably 90% to 99%) of the total transmit power of the block with superimposed pilots to data symbols and allocate the remaining transmit power to pilot symbols.

[0095] In one or more embodiments, the transmitter is arranged to adapt the power allocated to data symbols and pilot symbols in the first type (or low-rate) block, respectively, based on the communication channel used, the carrier frequency used, and / or the speed difference between the transmitter and receiver. For example, if the communication channel, carrier frequency, and / or speed difference between the transmitter and receiver change, the adaptation can be dynamic for individual or grouped subsequent communication frames.

[0096] In one or more embodiments, the signal mapper is arranged to adapt the size of the first and / or second type of blocks, particularly their length in the delay domain, based on the communication channel used, the carrier frequency used, the maximum delay, and / or the speed difference between the transmitter and receiver. For example, if the communication channel, carrier frequency, and / or speed difference between the transmitter and receiver change, the adaptation can be dynamic for individual or grouped subsequent communication frames.

[0097] The static determination of the size and / or power allocation ratio of the first and / or second class blocks in the transmitter can be based on the assumption that the communication channel used, the carrier frequency, and / or the speed difference between the transmitter and receiver are static or change negligibly within tolerable limits. The static determination may also consider worst-case scenarios, such as the maximum expected or permissible speed difference, maximum expected delay, etc., between the transmitter and receiver when they are communicatively connected (e.g., within radio range). The expected maximum speed difference can be provided by external input data—such as speed limits of mobile entities (e.g., cars or trains) within the coverage area of ​​a fixed transmitter.

[0098] Dynamic adaptation of the size and / or power allocation ratio of the first and / or second class blocks in the transmitter can be based on the actual speed difference between the transmitter and receiver. Such information can be provided from the receiver, such as the receiver's velocity vector, or based on information available at the transmitter (e.g., the number of receivers within range of the transmitter). In a fixed transmitter (e.g., a base station), this number can correspond to the current or average number of receivers attached to or communicatively connected to the transmitter. Dynamic adaptation can also be based on information received from the receiver, such as channel state information, bit error rate, or the number of iterations required to decode previously received signals for channel estimation.

[0099] However, the power allocation in the transmitter and / or the size and / or power allocation ratio of the first and / or second class blocks can also be adjusted for specific requirements. For example, to achieve fast convergence performance in the receiver, the power allocation ratio α can be chosen to be less than the power allocation ratio when the data SNR and pilot SNR are equal, and / or the BEM resolution of the first channel estimation unit can be set to be greater than 1. Simulation experiments show that a power allocation ratio of approximately 95% achieves an optimal balance between bit error rate (BER) and convergence performance.

[0100] By performing training before actual transmission begins, a near-optimal power allocation for the ratio of pilot signals to data signals can be dynamically obtained. As further mentioned above, a favorable power allocation ratio can be obtained when the average data SNR and the average pilot SNR are equal.

[0101] According to a third aspect of the invention, a receiver for an OTFS transmission system is provided, comprising a first receiver-side conversion unit and a second receiver-side conversion unit. The receiver is configured to receive, at the input of the first receiver-side conversion unit, a time-domain signal representing a communication frame according to the first aspect of the invention, transmitted on an actual communication channel (i.e., a communication channel with Doppler spread), and the first receiver-side conversion unit outputs a two-dimensional representation of the received communication frame in the time-frequency domain. The output of the first receiver-side conversion unit is provided to the input of the second receiver-side conversion unit, which outputs a two-dimensional representation of the received communication frame, including first and second class blocks, in the delay-Doppler domain. The receiver includes an OFO estimator configured to perform initial OFO estimation and compensation using symbols carried in the first class blocks of the received communication frame, and further includes an iterative two-stage channel estimation and equalization block configured to perform joint estimation of residual OFO and the communication channel based on symbols carried in all blocks of the received communication frame.

[0102] Therefore, according to one or more embodiments, the initial OFO estimation includes performing autocorrelation processing on the received OTFS symbols (corresponding to low-rate data and SP) carried in a first type of block of the received communication frame. Initial OFO compensation includes providing the initial OFO estimate to an iterative two-stage channel estimation and equalization block for combination with a BEM basis function used for channel estimation. The oscillator frequency offset (OFO) estimation result is combined with the BEM basis function to generate a new BEM basis function. As discussed in detail above.

[0103] According to one or more embodiments, the joint estimation of the residual OFO and the channel uses an iterative two-stage channel estimation architecture: performing initial channel estimation and subsequent initial equalization and symbol estimation; and performing iterative channel estimation and subsequent corresponding equalization and symbol estimation.

[0104] According to one or more embodiments, at least the pilot signal output from the second receiver-side conversion unit is provided to the first channel estimation unit, which outputs a first estimation result of the time-domain channel matrix. First estimation result of the time-domain channel matrix At least the data signal output from the second receiver-side conversion unit, or the pilot signal and data signal output from the second receiver-side conversion unit, are provided to the equalizer unit, which outputs an estimated set of at least the data signal. The estimated set of at least the data signal, along with at least a first block or both first and second blocks output from the second receiver-side conversion unit, are provided to the second channel estimation unit, which outputs a second estimation result of the time-domain channel matrix. Output of the second channel estimation unit The data signal output from the second receiver-side conversion unit, or the pilot signal and data signal output from the second receiver-side conversion unit, are provided to the equalizer unit, which outputs a further estimated set of at least the data signal. The channel estimation in the second channel estimation unit and the estimation of the estimated set of at least the data signal in the equalizer unit are iteratively repeated until the termination criterion is met. In other words, the time-domain channel matrix is ​​repeatedly estimated in the second channel estimation unit. The process of estimating at least a set of data symbols in the equalizer unit, feeding back the latest output of the equalizer unit to the second receiver-side channel estimation unit, and at least a pilot signal output from the second receiver-side conversion unit, or a pilot signal and a data signal output from the second receiver-side conversion unit, continues until a termination criterion is met.

[0105] Termination criteria may include the convergence of the equalizer unit's output. For example, convergence can be assumed when the bit error rate of the decoded output of the equalizer units used for two subsequent iterations is below a predetermined threshold. This threshold could be, for example, less than 10. -6 The difference in bit error rate. Another conceivable termination criterion could be a predetermined number of iterations. Alternatively, a maximum number of iterations could be set, after which iterations would terminate, but if the bit error rate in two consecutive iterations falls below a predetermined threshold before reaching the maximum number of iterations, iterations would terminate prematurely.

[0106] In one or more embodiments of the receiver, the first receiver-side transformation unit is adapted to perform a finite Fourier transform, an inverse Heisenberg transform, or a Wigner transform.

[0107] In one or more embodiments of the receiver, the second receiver-side transformation unit is adapted to perform a symplectic finite Fourier transform.

[0108] In one or more embodiments of the receiver, the first channel estimation unit is adapted to perform channel estimation based on a base extension modeling of a first BEM order for the time-varying communication channel. The first BEM order refers to the order of the base extension used to model the communication channel. The first channel estimation is preferably pilot-aided channel estimation, i.e., estimation is performed using the known location and / or other properties of pilot signals in the communication frame.

[0109] In one or more embodiments of the receiver, the equalizer performs message passing, zero-forcing, and / or minimum mean square error equalization.

[0110] In one or more embodiments of the receiver, the second channel estimation unit is adapted to perform channel estimation based on a base extension modeling of a second BEM order for the time-varying communication channel. The second BEM order refers to the order of the base extension used to model the communication channel. The second channel estimation is preferably data-assisted channel estimation, i.e., estimation is performed using signals estimated in the equalizer unit in addition to the pilot signals in the communication frame.

[0111] The first and second BEM orders of the first and second channel estimation units can be the same or different. It is important to note that a smaller BEM order Q and a lower BEM resolution T can be used when lower pilot power is required. However, a smaller BEM order generally leads to slower convergence. A higher BEM order with higher resolution can provide excellent performance and faster convergence, but may require higher pilot power. For example, when the BEM order increases from Q = 2 to Q = 4, the resolution T can advantageously increase from 1 to 2.

[0112] Because the OFO is estimated in the receiver, the residual OFO is relatively small. This allows for the use of a smaller BEM order Q and a smaller resolution T in the initial channel estimation, while still achieving near-optimal performance and fast convergence.

[0113] One or more embodiments of the receiver further include a control unit adapted to receive information about the receiver's orientation and absolute velocity on the ground, the transmitter's orientation and absolute velocity on the ground, and / or the relative velocity between the receiver and the transmitter, and is also adapted to determine the BEM order Q. S And / or adapted to receive the BEM order Q used at the transmitter to compose the communication frame. S The received or determined BEM order Q SThe received information is transmitted to the first channel estimation unit and / or the second channel estimation unit to determine the appropriate order of the BEM to be applied or used. For example, when a mobile terminal attaches to a base station, or more generally, when a communication connection is established between a transmitter and a receiver, information about the BEM order Q to be used at the transmitter can be transmitted. S This information. Although such transmission only requires a few bytes, it can improve performance and / or spectral efficiency.

[0114] If the BEM order Q is not used S With dynamic adaptation, the receiver can use the specified default values.

[0115] The various components of the aforementioned transmitter and receiver can be implemented in hardware (i.e., hardware controlled and / or parameterized by software), software modules, or a combination thereof. Specifically, the first channel estimation unit and the second channel estimation unit of the receiver can rely on the same hardware or software module and can be parameterized for the corresponding pilot-aided or data-aided channel estimation by using the corresponding input data and GCE-BEM parameters.

[0116] A wireless device according to a fourth aspect of the present invention includes a transmitter and / or receiver for an OTFS transmission system as described above.

[0117] According to a fifth aspect of the invention, a method for transmitting a binary data sequence via an OTFS communication channel includes mapping a previously received binary data sequence in a signal mapper into a two-dimensional arrangement of data symbols and data symbols with superimposed pilot symbols in the delayed Doppler domain of a dual-rate communication frame as described in the first aspect of the invention. The two-dimensional arrangement forms a communication frame of the OTFS transmission system. The mapping may include receiving the binary data sequence at the input of the signal mapper and providing the two-dimensional arrangement of data symbols and data symbols with superimposed pilot symbols at the output of the signal mapper.

[0118] The method further includes mapping a two-dimensional communication frame in the delayed Doppler domain into a two-dimensional arrangement of information symbols in the time-frequency domain in a first transmitter-side conversion unit. The first conversion may include receiving the two-dimensional communication frame in the delayed Doppler domain at the input of the first transmitter-side conversion unit and providing a two-dimensional arrangement of information symbols at the output of the first transmitter-side conversion unit.

[0119] The method further includes transforming the two-dimensional arrangement of information symbols into a continuous time-domain signal representing a communication frame in a second transmitter-side conversion unit. The second transformation may include receiving the two-dimensional arrangement of information symbols in the time-frequency domain at the input of the second transmitter-side conversion unit and providing a continuous time-domain signal representing the communication frame at the output of the second transmitter-side conversion unit.

[0120] The method also includes transmitting a continuous time-domain signal representing a communication frame via a communication channel. Transmission may include steps known to conventional transmitters, such as amplification, beamforming, and pointing.

[0121] In one or more embodiments of the method, the first transformation step includes performing an inverse symplectic finite Fourier transform on the two-dimensional communication frame in the delayed Doppler domain.

[0122] In one or more embodiments of the method, the second transformation step includes performing a Heisenberg transform or an inverse finite Fourier transform (IFFT) on the two-dimensional arrangement of the information symbols.

[0123] In one or more embodiments, the method further includes setting the power allocation ratio between data symbols and pilot symbols to a range of 0.5 to 0.99, preferably to a range of 0.9 to 0.99.

[0124] In one or more embodiments, the method further includes adapting the power allocation ratio between data symbols and pilot symbols based on the communication channel used, the carrier frequency used, and / or the speed difference between the transmitter and the receiver.

[0125] In one or more embodiments, the method further includes adapting the number of data symbols with superimposed pilots according to the communication channel used, the carrier frequency used, and / or the speed difference between the transmitter and the receiver.

[0126] According to a sixth aspect of the invention, a method for receiving a binary data sequence via a practical OTFS communication channel includes receiving a continuous time-domain signal representing a communication frame according to a first aspect of the invention via the communication channel. The method further includes transforming the continuous time-domain signal representing the communication frame into a two-dimensional arrangement of information symbols in the time-frequency domain obtainable at the output of a first receiver-side transformation unit. In a next step of the method, the two-dimensional arrangement of information symbols in the time-frequency domain, including pilot signals and data signals, is transformed in a second receiver-side transformation unit into a two-dimensional communication frame in the delayed Doppler domain, including pilot signals and data signals, obtainable at the output of the second receiver-side transformation unit. Next, an OFO (Out-of-Flight) estimate is obtained from a first class of blocks of the communication frame, and the OFO is provided to a channel estimation unit to incorporate the OFO estimate into an applied channel estimation (CE) function.

[0127] In order to obtain the first estimation result of the time-domain channel matrix at the output of the first channel estimation unit The first type of block or the first and second type of blocks in the two-dimensional communication frame in the delay Doppler domain are provided to the first channel estimation unit. Then, the first estimation result of the time-domain channel matrix is ​​used. The communication frame (in which superimposed pilot symbols are removed) output from the second receiver-side conversion unit is provided to the equalizer unit to obtain at least an estimated set of data signals at the output of the equalizer unit.

[0128] Next, the estimated sets of the first and second type blocks output from the second receiver-side conversion unit and at least the data signal output from the equalizer unit are provided to the second channel estimation unit for estimating the time-domain channel matrix. Further estimation is then performed. The time-domain channel matrix is ​​then obtainable at the output of the second channel estimation unit. Further estimates and communication frames (with superimposed pilot symbols removed) output from the second receiver-side conversion unit are provided to the equalizer unit to obtain at least a further set of estimates for the data signal. The time-domain channel matrix is ​​iteratively estimated in the second channel estimation unit. The process involves estimating at least the data signal set in the equalizer unit until a termination criterion is met. During iteration, the time-domain channel matrix is ​​estimated using the latest estimates of the at least the data signal and the superimposed pilot signals. And use their latest estimated time-domain channel matrices And the communication frame output from the conversion unit on the second receiver side is used to perform at least the next step estimation of the data signal.

[0129] In one or more embodiments of the method, transforming a continuous time-domain signal representing a communication frame into a two-dimensional arrangement of information symbols in the time-frequency domain includes performing a finite Fourier transform, an inverse Heisenberg transform, or a Wigner transform on the continuous time-domain signal representing the communication frame.

[0130] In one or more embodiments of the method, transforming a two-dimensional arrangement of information symbols including data signals and SP data signals in the time-frequency domain into a two-dimensional communication frame including data signals and SP data signals in the delayed Doppler domain includes performing a symplectic finite Fourier transform on the two-dimensional arrangement of information symbols including data signals and SP data signals in the time-frequency domain.

[0131] In one or more embodiments of the method, estimating OFO from a first type of block includes separating a first type of block from a received communication frame, performing an autocorrelation calculation on at least the SP symbols contained therein, and extracting OFO information from the autocorrelation calculation result.

[0132] In one or more embodiments of the method, a first estimation result of the time-domain channel matrix is ​​obtained. Channel estimation is performed using basis extension modeling of time-varying communication channels based on the first BEM order.

[0133] In one or more embodiments of the method, the time-domain channel matrix in the second channel estimation unit The estimation includes performing channel estimation based on the base extension modeling of the time-varying communication channel of the second BEM order.

[0134] As further mentioned above, the first and second BEM orders Q and / or the resolution T of the first and second channel estimation units can be the same or different.

[0135] In one or more embodiments of the method, obtaining an estimated set of at least data signals in the equalizer unit includes message passing, zero-forcing, and / or minimum mean square error equalization processing on the at least data signals obtained after a second transformation in the second receiver-side conversion unit. This may include pre-removing any superimposed pilot signals from the transmitted frame.

[0136] In one or more embodiments, the method further includes receiving information in a control unit regarding the direction and absolute velocity of the receiver on the ground, the direction and absolute velocity of the transmitter on the ground, and / or the relative velocity between the receiver and the transmitter, and determining the BEM order Q. S And / or receive the BEM order Q used at the transmitter to compose the communication frame. S The received information can be used to determine the appropriate BEM order to be used in the first channel estimation unit and / or the second channel estimation unit. The corresponding received or determined BEM order is provided to the first channel estimation unit and / or the second channel estimation unit.

[0137] The method of sending and / or receiving can be represented by computer program instructions, which, when executed by a microprocessor, respectively cause the computer and / or control hardware components of the transmitter or receiver of the OFTS transmission system as described above to perform the sending or receiving method as described above.

[0138] Computer program instructions can be stored or transmitted in a retrieved manner on a computer-readable medium or data carrier. The medium or data carrier can physically take the form of, for example, a hard disk, a solid-state drive, or a flash memory device. However, the medium or data carrier can also include modulated electromagnetic, electrical, or optical signals that are received by the computer using a corresponding receiver and transmitted to and stored in the computer's memory.

[0139] The OTFS receiver described above effectively resists time-frequency distortion caused by the oscillator frequency offset (OFO) of the free-running crystal oscillator (XO) by employing dual-rate data and superimposed pilot (SP) technology.

[0140] An initial OFO estimate with only a small residual OFO is obtained by performing autocorrelation on two identical Class 1 blocks carrying data symbols and superimposed pilots (designed to support wide-range OFO estimation). This OFO estimate is then fused into the BEM basis function, and joint estimation of the residual OFO due to BEM modeling errors and noise, as well as the OTFS channel, is performed by using superimposed pilots and pseudo-pilots during the iterative process. Using superimposed pilots helps improve spectral efficiency, while OFO estimation allows for a reduction in the signal power allocated to the SP, leaving more signal power for the data signal, which improves the reliability of signal estimation.

[0141] It is important to note that OFO affects the bit error rate (BER) and mean squared error (MSE) of symbol detection. The BER deteriorates slightly with increasing residual OFO. This can be compensated for by increasing the signal-to-noise ratio (SNR) of data symbols in the first-class block.

[0142] XOs with very small OFOs are typically more expensive and require more power to operate. In transmitters and receivers implementing the corresponding method according to the invention, the proposed communication frames allow for processing of larger OFOs in cheaper XOs with larger OFOs at lower power requirements, thus also allowing for reduced power consumption and lower device costs. Due to the flexible design of the first and second class blocks, a wider range of OFO estimation can be adjusted to suit the needs of the system.

[0143] Using Class 1 and Class 2 blocks, and employing superimposed pilots (SP), OFO can be estimated and OTFS channel estimation can be performed while reducing frame data rate costs. By carefully designing the placement of the Class 1 blocks, OFO estimation does not require additional dedicated pilot signals.

[0144] Thanks to the carefully designed communication frames with Class I and Class II blocks, where the Class I block provides a low data rate and SP, the proposed OTFS receiver exhibits excellent performance in bit error rate (BER), mean square error of OFO estimation (MSE), and MSE of channel estimation, without requiring a dedicated pilot signal. Its BER performance is close to the lower bound of the BER assuming perfect estimation and compensation of OFO and the channel.

[0145] The communication frame, transmitter, receiver, and corresponding method according to the present invention can be advantageously used for high-speed mobile communication. Attached Figure Description

[0146] In the following sections, exemplary embodiments of the invention will be described in more detail with reference to the accompanying drawings. In the drawings,

[0147] Figure 1 A block diagram of a general OTFS transport system is shown.

[0148] Figure 2 The superimposed pilots and their power allocation are shown.

[0149] Figure 3 An exemplary visualization of the Doppler spectral shift in an OTFS communication channel in the presence of OFO is shown.

[0150] Figure 4 An exemplary OTFS frame pattern at the transmitter according to the present invention is shown.

[0151] Figure 5 A block diagram illustrating OFO estimation, channel estimation, and equalization of an exemplary receiver according to the present invention is shown.

[0152] Figure 6 A flowchart illustrating a method for transmitting binary data sequences via an OTFS communication channel is shown.

[0153] Figure 7 A flowchart illustrating a method for receiving binary data sequences through an OTFS communication channel susceptible to dual selective fading is shown.

[0154] Figure 8 An exemplary block diagram of an apparatus for performing a transmission method according to the present invention is shown, and

[0155] Figure 9 An exemplary block diagram of an apparatus for performing the receiving method according to the present invention is shown.

[0156] In all the accompanying drawings, the same reference numerals may be used to refer to the same or similar elements. Detailed Implementation

[0157] Figures 1 to 4 This has already been described above and will not be discussed further.

[0158] Figure 5 A schematic block diagram illustrating the initial OFO estimation and compensation, as well as the joint residual OFO and channel estimation, in an exemplary receiver 300 according to the present invention is shown. After performing SFFT and Wigner transform, the received symbols in the delayed Doppler domain y[k,l] can be used for further processing.

[0159] OFO estimation 312, channel estimation 320 with two-level iterative CE blocks 321 and 322, and equalization 326 replace [the previous methods]. Figure 1 The general channel estimation and equalization block 310 is shown. Figure 1 All other components of the receiver 300 shown, namely the first receiver-side conversion unit and the second receiver-side conversion units 304 and 306, are identical and are not shown in the figure.

[0160] First, the first type of blocks carrying data symbols and superimposed pilots, and the second type of blocks carrying data symbols and superimposed pilots or carrying only data symbols, from the two-dimensional arrangement (as a signal y[k,l] in the delayed Doppler domain) output from the second receiver-side conversion unit 306, are provided to the OFO estimation unit 312. The OFO estimation unit 312 includes a block separation unit 314, which separates the first type of blocks from the second type of blocks in the communication frame and provides the first type of blocks to the autocorrelation unit 316. The autocorrelation process may include eigenvalue decomposition (not shown in the figure), or eigenvalue decomposition may be performed after the autocorrelation process. The result of the autocorrelation process is provided to the OFO extraction unit 318, which determines the OFO and provides it to the BEM basis generation unit 319. Based on the OFO estimate output from the OFO extraction unit 318... BEM basis generation unit 319 determines the BEM basis functions to be used in channel estimation units 321 and 322, and forwards the corresponding information to the first channel estimation unit 321 and the second channel estimation unit 322 accordingly.

[0161] The first channel estimation unit 321 is provided with either the first type of block, which carries data symbols and superimposed pilots, output from the low-rate block extraction unit 314, or the entire frame represented by the signal y[k,l], which includes the first and second type blocks in the delayed Doppler domain. The first channel estimation unit 321 uses a first BEM order Q. S The GCE-BEM channel model performs a first channel estimation with superimposed pilot assistance and including OFO. The first BEM order Q... S It can be very small, using a low resolution T, although at the cost of slower convergence. However, the first BEM order Q... S It can also be quite large, using a higher resolution T, thus leading to faster convergence. The first channel estimation unit 321 outputs the first or initial channel estimation result. The estimation result It is provided to the input section of the pilot removal unit 324.

[0162] The communication frame represented by the signal y[k,l] includes a first type block and a second type block in the delayed Doppler domain. This communication frame is also provided to the input of the pilot removal unit 324 and the input of the second channel estimation unit 322.

[0163] Pilot removal unit 324 uses the latest channel estimate output by first channel estimation unit 321 or second channel estimation unit 322 to remove superimposed pilot signals from signal y[k,l] based on the known structure of the transmitted communication frames (i.e., their arrangement and power levels), retaining only the data signal. The output signal of pilot removal unit 324 is shown below. The data is provided to equalizer unit 326. Equalizer unit 326 determines and outputs the estimated results of the transmitted data symbols.

[0164] The first channel estimation unit 321 performs initial channel estimation and obtains initial estimates of the transmitted data symbols. Then, the latest estimated data symbols will be... The input group is provided to the input section of the second channel estimation unit 322 in an iterative manner. The second channel estimation unit 322 uses the GCE-BEM channel model (which has a second BEM order Q). L The second resolution T) is used to estimate the superimposed pilot signals in the signal y[k,l] and the transmission symbols fed back from the equalizer unit 326 to the second channel estimation unit 322. (As a pseudo-pilot) data-assisted channel estimation including OFO. The second channel estimation unit 322 can use a higher BEM order QL and a higher resolution T than the first channel estimation unit, although using the same BEM order Q. L And resolution T is also acceptable.

[0165] In each iteration, the latest output of the second channel estimation unit 322 (representing the channel estimate) The signal is input to the pilot removal unit 324. Based on this, the pilot removal unit 324 removes the superimposed pilot from the signal vector y[k,l] received in the delayed Doppler domain, and removes the signal representing only the received data signal. The estimation result is provided to the input section of the equalizer unit 326. The equalizer unit 326 outputs a more accurate estimation result of the transmitted data symbols than before. Repeat the process until the termination criteria are met.

[0166] Figure 6 A flowchart of a method 400 for transmitting a binary data sequence via an OTFS communication channel is shown. In step 402, the binary data sequence is mapped to a two-dimensional communication frame in the delayed Doppler domain, the two-dimensional communication frame comprising a first class block and a second class block according to a first aspect of the invention. In step 404, the two-dimensional communication frame in the delayed Doppler domain is transformed into a two-dimensional arrangement of information symbols in the time-frequency domain. In step 406, the two-dimensional arrangement of information symbols in the time-frequency domain is transformed into a continuous time-domain signal representing the communication frame, which is transmitted through the channel in step 408. Before transforming the two-dimensional arrangement of information symbols in the delayed Doppler domain into a two-dimensional arrangement of information symbols in the time-frequency domain, the power distribution ratio between the pilot signal and the data signal and / or the number of data symbols with superimposed pilots can be determined or adapted in optional step 410; these parameters are set in optional step 412.

[0167] Figure 7 A flowchart of a method 500 for receiving a binary data sequence carried in a communication frame according to a first aspect of the present invention via an OTFS communication channel susceptible to biselective fading is shown. In step 502, a continuous time-domain signal representing the communication frame is received via the communication channel. In step 504, the continuous time-domain signal representing the communication frame is transformed into a two-dimensional arrangement of information symbols in the time-frequency domain. In step 506, the two-dimensional arrangement of information symbols in the time-frequency domain, including pilot signals and data signals, is transformed into a two-dimensional communication frame in the delayed Doppler domain, the two-dimensional communication frame including a first type of block and a second type of block according to the first aspect of the present invention. In step 508, the OFO from the first type of block of the communication frame is estimated, and in step 510, the OFO is provided from the OFO estimator 312 to the channel estimation units 321, 322 so that the OFO estimation result is incorporated into the applied CE function. In step 512, the initial estimation result of the time-domain channel matrix is ​​obtained in the first channel estimation unit 320. The first channel estimation unit 321 performs channel estimation on the first type block or the first and second type blocks based on the basis extension modeling of the time-varying communication channel with a first BEM order and a first resolution. In step 514, pilot symbols are removed from the received communication frames represented by the first and second type blocks, and the resulting signal is provided to the equalizer unit 326. In step 516, based on the channel estimation and the communication frames in the delayed Doppler domain, at least an estimated set of data signals is determined in the equalizer unit 326. Step 518 checks whether the termination criterion is met. If it is met (the "yes" branch of step 518), then in step 522, the estimated received symbols are signaled to be output to the demapper, and finally, the received binary sequence can be output. If the termination criterion is not met (the "No" branch of step 518), a further estimate of the time-domain channel matrix is ​​obtained in the second channel estimation unit 322 in step 520. The second channel estimation unit 322 performs channel estimation based on the basis extension model of the time-varying communication channel at the second resolution and the second BEM order, using the previously estimated data signal in addition to the pilot signal. The result of the further channel estimation is provided to the pilot symbol removal step 514, and the equalization step 516 and the check step 518 to verify whether the termination criterion is met are repeated.

[0168] Optionally, in step 524, the BEM order Q used in the transmitter may be received. S Alternatively, it may allow information to determine the BEM order to be used in channel estimation. In step 526, the BEM order Q to be used is determined. S And in step 528, the order Q is... S Provided to the channel estimation unit.

[0169] Figure 8An exemplary block diagram of a transmitter 200 according to a second aspect of the present invention is shown, the transmitter being configured to perform a transmission method 400 according to a fourth aspect of the present invention. The transmitter 200 includes a microprocessor 220, a volatile memory 222, a non-volatile memory 224, and a communication interface 226 for transmitting signals to a receiver 300 via an antenna 206. The aforementioned components are communicatively connected via at least one data connection or bus 228. The non-volatile memory 224 stores computer program instructions that, when executed by the microprocessor 220, cause the transmitter 200 to perform the transmission method 400 of the fourth aspect of the present invention. It should be noted that, referring to… Figure 1 The various functional modules of the transmitter described (e.g., the first transmitter-side conversion unit 202 and the second transmitter-side conversion unit 204) can be implemented, in whole or in part, by software executed by the microprocessor 220.

[0170] Figure 9 An exemplary block diagram of a receiver 300 according to a third aspect of the present invention is shown, the receiver being configured to perform a receiving method 500 according to a fifth aspect of the present invention. The receiver 300 includes a microprocessor 330, a volatile memory 332, a non-volatile memory 334, and a communication interface 336 for receiving signals from a transmitter via an antenna 302. The aforementioned components are communicatively connected via at least one data connection or bus 338. The non-volatile memory 334 stores computer program instructions that, when executed by the microprocessor 330, cause the receiver 300 to perform the receiving method 500 of the fifth aspect of the present invention. It should be noted that, referring to… Figure 1 The various functional modules of the receiver described (e.g., the first receiver-side conversion unit 304, the second receiver-side conversion unit 306, the OFO estimation unit 312, the first channel estimation unit 321, the second channel estimation unit 322, the pilot removal unit 324, and the symbol estimation and detection functions in the equalizer unit 326) can be implemented in whole or in part by software executed by the microprocessor 330.

[0171] List of definitions and figure references (part of the specification)

[0172] f c carrier frequency

[0173] Δf Subcarrier spacing

[0174] L Channel Length

[0175] The number of delay bins in the M communication frame

[0176] The number of Doppler cells in N communication frames

[0177] P T Total transmit power

[0178] α pilot power allocation ratio

[0179] Q S BEM order in initial channel estimation

[0180] Q L The BEM order in subsequent iterative channel estimation

[0181] AWGN Additive White Gaussian Noise

[0182] BEM base extension model

[0183] CE-BEM Complex Exponential Basis Extended Model

[0184] GCE-BEM Generalized Complex Exponential Basis Extended Model

[0185] DFT (Discrete Fourier Transform)

[0186] KL-BEM Karhunen-Loeve BEM

[0187] Mean Squared Error

[0188] OTFS Orthogonal Time-Frequency Space

[0189] SNR (Signal-to-Noise Ratio)

[0190] BER (Bit Error Rate)

[0191] OFDM (Orthogonal Frequency Division Multiplexing)

[0192] MP Message Passing

[0193] SFFT (Symphitic Finite Fourier Transform)

[0194] 200 transmitters

[0195] 202 First Transmitter Side Transformation Unit

[0196] 204 Second Transmitter Side Conversion Unit

[0197] 206 antennas

[0198] 220 microprocessor

[0199] 222 Volatile Memory

[0200] 224 Non-volatile memory

[0201] 226 Communication Interface

[0202] 228 Data Connection / Bus

[0203] 300 receiver

[0204] 302 antenna

[0205] 304 First Receiver Side Conversion Unit

[0206] 306 Second Receiver Side Conversion Unit

[0207] 310 Channel Estimation and Equalization Box

[0208] 312 OFO Estimator

[0209] 314 Low-rate block extraction

[0210] 316 Autocorrelation

[0211] 318 OFO Extraction

[0212] 319 Generation of BEM base containing OFO

[0213] 320 Two-level CE and EQ

[0214] 321 First Channel Estimation Unit

[0215] 322 Second Channel Estimation Unit

[0216] 324 Pilot Removal Unit

[0217] 326 Equalizer Units

[0218] 326 Pilot Extraction Unit

[0219] 330 microprocessor

[0220] 332 Volatile Memory

[0221] 334 Non-volatile memory

[0222] 336 Communication Interface

[0223] 338 Data Connection / Bus

[0224] 500 receiving method

[0225] 502 Receives continuous time-domain signals

[0226] 504 Transforms a continuous time-domain signal into a two-dimensional arrangement of information symbols in the time-frequency domain.

[0227] 506 Transforms the two-dimensional arrangement of information symbols in the time-frequency domain into a two-dimensional communication frame in the delayed Doppler domain.

[0228] 508 Estimate OFO

[0229] 510 Provide the OFO estimation results to the channel estimation

[0230] 512 Initial Channel Estimation

[0231] 514 Remove pilot signal

[0232] 516 Estimating the sign in the equalizer unit

[0233] Does 518 meet the termination criteria?

[0234] 522 Output the latest estimation results to the demapping unit.

[0235] 520 Estimate the channel in the second channel estimation unit

[0236] 524 BEM order used in receivers and transmitters

[0237] 526 Determine the BEM order to be used in the receiver

[0238] 528. The BEM order is provided to the channel estimation unit.

Claims

1. A method of generating a communication frame for an orthogonal time and frequency space transmission system, the method comprising generating at least one first type block comprising a plurality of data signals arranged two-dimensionally along a delay domain and a Doppler domain, at least one of which has an overlay pilot signal, and generating at least one second type block comprising data signals arranged two-dimensionally along a delay domain and a Doppler domain, or comprising a plurality of data signals arranged two-dimensionally along a delay domain and a Doppler domain, at least one of which has an overlay pilot signal, the at least one second type block being surrounded in the delay domain by first type blocks, the first type blocks surrounding the second type blocks having at least one identical data symbol and an associated identical overlay pilot symbol at the same position in the two-dimensional arrangement.

2. The method of generating a communication frame for an orthogonal time frequency space transmission system of claim 1, wherein, The data symbols in the first type blocks surrounding the second type blocks in the communication frame carry only data for controlling a communication connection between a transmitter and a receiver.

3. A method of generating a communication frame for an orthogonal time frequency space transmission system as claimed in claim 1 or 2, wherein, The spacing between at least two first type blocks in the delay domain is variable.

4. A transmitter (200) for an orthogonal time and frequency space transmission system, the transmitter comprising a first transmitter side transformation unit (202), a second transmitter side transformation unit (204) and a signal mapper arranged upstream of the first transmitter side transformation unit and the second transmitter side transformation unit, wherein the signal mapper being adapted to receive a binary data sequence and to output a two-dimensional bi- rate communication frame in a delay Doppler domain generated using a method according to claim 1, the communication frame comprising data symbols and overlay pilot symbols, wherein the first transmitter side transformation unit (202) is adapted to receive at an input the two-dimensional bi-rate communication frame in the delay Doppler domain output from the signal mapper and to output a two-dimensional arrangement of information symbols in a time frequency domain, and wherein the output of the first transmitter side transformation unit (202) is provided to an input of the second transmitter side transformation unit (204), the second transmitter side transformation unit being adapted to output a continuous time domain signal representing the communication frame for transmission over a communication channel.

5. The transmitter (200) for orthogonal time frequency space transmission systems according to claim 4, wherein The first transmitter side transformation unit (202) is adapted to perform a pre- coding and / or an inverse sine finite Fourier transform.

6. The transmitter (200) for orthogonal time frequency space transmission systems according to claim 4 or 5, wherein The second transmitter side transformation unit (204) is adapted to perform a Heisenberg transform or an inverse finite Fourier transform (IFFT).

7. The transmitter (200) for orthogonal time frequency space transmission systems according to claim 4 or 5, wherein The transmitter (200) is arranged to allocate 50% to 99% of the total transmit power of a block with overlay pilot to data symbols and the remaining transmit power to pilot symbols.

8. The transmitter (200) for orthogonal time frequency space transmission systems according to claim 4 or 5, wherein The transmitter (200) is arranged to allocate 90% to 99% of the total transmit power of a block with overlay pilot to data symbols and the remaining transmit power to pilot symbols.

9. The transmitter (200) for orthogonal time frequency space transmission systems according to claim 4 or 5, wherein The transmitter (200) is arranged to adapt the power allocated to data symbols and overlay pilot symbols, respectively, in dependence of the used communication channel, the used carrier frequency, and / or the speed difference between transmitter and receiver.

10. The transmitter (200) for orthogonal time frequency space transmission systems according to claim 4 or 5, wherein The signal mapper is arranged to adapt the size of the first type of block and / or the second type of block depending on the used communication channel, the used carrier frequency, and / or the speed difference between the transmitter and the receiver.

11. A wireless device for an orthogonal time frequency space transmission system, the wireless device comprising a transmitter (200) according to any one of claims 4 to 10.

12. A method (400) of transmitting a sequence of binary data over an orthogonal time frequency space communication channel, the method comprising: - mapping (402) the sequence of binary data into a two-dimensional communication frame in delay-Doppler domain generated using the method according to any one of claims 1 to 3 in a signal mapper, - transforming (404) the two-dimensional communication frame in delay-Doppler domain into a two-dimensional arrangement of information symbols in time-frequency domain in a first transmitter-side transformation unit (202), - transforming (406) the two-dimensional arrangement of information symbols in time- frequency domain into a continuous time domain signal representing the communication frame in a second transmitter-side transformation unit (204), and - transmitting (408) the continuous time domain signal representing the communication frame over the communication channel.

13. The method (400) of claim 12, wherein The first transformation step (404) comprises performing an inverse symplectic finite Fourier transform on the two-dimensional communication frame in delay-Doppler domain.

14. The method (400) of claim 12 or 13, wherein The second transformation step (406) comprises performing a Heisenberg transform or an inverse finite Fourier transform on the two-dimensional arrangement of information symbols.

15. The method (400) according to claim 12 or 13, further comprising setting (410) a power allocation ratio between data symbols and pilot symbols to be in the range of 0.5 to 0.

99.

16. The method (400) according to claim 12 or 13, further comprising setting (410) a power allocation ratio between data symbols and pilot symbols to be in the range of 0.9 to 0.

99.

17. The method (400) according to claim 12 or 13, further comprising adapting (412) a power allocation ratio between data symbols and pilot symbols depending on the used communication channel, the used carrier frequency, and / or the speed difference between the transmitter and the receiver.

18. The method (400) according to claim 12 or 13, further comprising adapting (412) the number of data symbols with superimposed pilots depending on the used communication channel, the used carrier frequency, and / or the speed difference between the transmitter and the receiver.

19. A computer program product comprising computer program instructions which, when executed by a microprocessor, cause the computer and / or control hardware components of a transmitter (200) for an orthogonal time frequency space transmission system according to any one of claims 4 to 10 to perform the method (400) according to any one of claims 12 to 18, respectively.

20. A computer readable medium retrievably transmitting or storing the computer program product according to claim 19.

21. A data carrier retrievably transmitting or storing the computer program product according to claim 19.

Images