Quasi-Z-source inverter control method and device, medium and equipment

Abstract

The invention discloses a control method and device of a quasi-Z-source inverter, a medium and equipment, and relates to the technical field of control of the quasi-Z-source inverter. According to an error between the inductive current sampling value and the inductive current reference value, determining an action sequence of applying a direct-through voltage vector and a non-direct-through voltage vector and a duty ratio of applying the direct-through voltage vector in a control period of the quasi-Z-source inverter; applying a plurality of reference non-through voltage vectors to the Z-source inverters respectively; predicting a capacitor voltage predicted value and an output current predicted value at a future moment under each reference non-through voltage vector; according to the capacitor voltage predicted value and the output current predicted value at the future moment and the corresponding reference value, determining a target non-direct-connection voltage vector from the multiple reference non-direct-connection voltage vectors; and controlling the Z-source inverter according to the action sequence of the through voltage vector and the non-through voltage vector, the duty ratio and the target non-through voltage vector. According to the method, the control precision of the qZSI on the output current is improved.

Description

Technical Field

[0001] The present invention relates to the technical field of control of a quasi-Z-source inverter, and in particular to a control method, device, medium and equipment for a quasi-Z-source inverter. Background Art

[0002] The quasi-Z-source inverter (qZSI) is a new type of power converter with a unique buck-boost mechanism, which has important practical significance for improving the overall performance of power converters.

[0003] Taking motor drive as an example, in motor drive applications, the output current of the qZSI is used to control motor speed and torque. Inaccurate output current can lead to problems such as unstable motor speed, insufficient torque, or overload, affecting the motor's service life and the reliability of the entire drive system. The inductor current, capacitor voltage, and output current are the three controlled variables of the qZSI. On the DC side of the qZSI, the inductor current is primarily affected by the duration of the through-voltage vector; on the AC side of the qZSI, the output current is affected by the duration of the effective voltage vector and the zero voltage vector. Existing technologies reduce output current control errors by continuously improving the voltage vector's action method.

[0004] However, the control accuracy of the output current of the qZSI in the prior art is low. Summary of the Invention

[0005] Based on this, it is necessary to provide a control method, device, medium and equipment for a quasi-Z-source inverter to address the above technical problems. The control method can improve the control accuracy of the qZSI output current.

[0006] The present invention adopts the following technical solutions: The present invention provides a control method for a quasi-Z-source inverter, comprising: Obtaining an inductor current sampling value of the quasi-Z-source inverter, and determining, based on an error between the inductor current sampling value and an inductor current reference value, an action sequence of applying a through-voltage vector and a non-through-voltage vector within a control period of the quasi-Z-source inverter and a duty cycle of applying the through-voltage vector; Applying a variety of reference non-through voltage vectors to the quasi-Z-source inverter respectively; For any reference non-through voltage vector, obtaining an initial inductor current sampling value, an initial capacitor voltage sampling value, and an initial output current sampling value of the quasi-Z-source inverter under the reference non-through voltage vector; determining a capacitor voltage prediction value at a future time based on the initial inductor current sampling value and the initial capacitor voltage sampling value, and determining an output current prediction value at a future time based on the initial output current sampling value; Quantifying the difference between the output current prediction value and the output current reference value under each benchmark non-through voltage vector, and screening multiple candidate non-through voltage vectors from multiple benchmark non-through voltage vectors based on the quantification results; Quantify the difference between the capacitor voltage prediction value and the capacitor voltage reference value under each candidate non-through voltage vector to determine the target non-through voltage vector; The quasi-Z-source inverter is controlled by applying a through voltage vector and a non-through voltage vector in a control period of the quasi-Z-source inverter, a duty cycle of applying the through voltage vector, and a target non-through voltage vector.

[0007] Preferably, determining the order of applying the through-voltage vector and the non-through-voltage vector within the control period of the quasi-Z-source inverter according to the error between the inductor current sampling value and the inductor current reference value includes: When the error is less than or equal to 0, it is determined that the through voltage vector is applied first and then the non-through voltage vector is applied within the control period of the quasi-Z-source inverter; When the error is greater than 0, it is determined that the non-through voltage vector is applied first and then the through voltage vector is applied within the control period of the quasi-Z-source inverter.

[0008] Preferably, determining the duty cycle of applying the through voltage vector within the control period of the quasi-Z-source inverter according to the error between the inductor current sampling value and the inductor current reference value includes: Determine the action time of the through voltage vector within the control cycle according to the error between the inductor current sampling value and the inductor current reference value; According to the action time of the through voltage vector in the control period, the duty cycle of the through voltage vector applied in the control period of the quasi-Z-source inverter is determined.

[0009] Preferably, determining the action time of the through voltage vector within the control period according to the error between the inductor current sampling value and the inductor current reference value includes: When the error is less than or equal to 0, the calculation method for determining the action time of the through voltage vector within the control period is: ; When the error is greater than 0, the calculation method for determining the action time of the through voltage vector within the control period is: ; in, is the action time of the through voltage vector within the control period, is the voltage gain of the quasi-Z-source inverter, is the control cycle length of the quasi-Z source inverter, is the inductor, is the DC supply voltage of the Z-source inverter, is the error between the inductor current sampling value and the inductor current reference value.

[0010] Preferably, quantifying the gap between the output current prediction value and the output current reference value under each reference non-shoot-through voltage vector includes: For any benchmark non-through voltage vector, the gap between the output current prediction value and the output current reference value is quantified by the output current cost function to obtain the output current cost function value; The output current cost function is: ; in, is the output current cost function value, They are Coordinate axes and Output current reference value on the coordinate axis, They are time Coordinate axes and The output current prediction value on the coordinate axis, The moment is the future moment, is the initial moment, The control cycle duration.

[0011] Preferably, the quantization result is an output current cost function value; and multiple candidate non-through voltage vectors are screened out from multiple reference non-through voltage vectors according to the quantization result, including: The reference non-indirect voltage vectors corresponding to a preset number of minimum output current cost function values ​​are determined as candidate non-indirect voltage vectors.

[0012] Preferably, quantifying the difference between the capacitor voltage prediction value and the capacitor voltage reference value under each candidate non-through voltage vector to determine the target non-through voltage vector includes: For any candidate non-through voltage vector, the gap between the capacitor voltage prediction value and the capacitor voltage reference value is quantified using the capacitor voltage cost function to obtain a capacitor voltage cost function value; Determine the candidate non-through voltage vector corresponding to the minimum capacitor voltage cost function value as the target non-through voltage vector; The capacitor voltage cost function is: ; in, is the capacitor voltage cost function value, is the capacitor voltage reference value, for The predicted value of the capacitor voltage at time , The moment is the future moment, is the initial moment, The control cycle duration.

[0013] The present invention provides a control device for a quasi-Z-source inverter, comprising: an acquisition module, configured to acquire an inductor current sampling value of the quasi-Z-source inverter and, based on an error between the inductor current sampling value and an inductor current reference value, determine an action sequence of applying a through-voltage vector and a non-through-voltage vector within a control period of the quasi-Z-source inverter and a duty cycle of applying the through-voltage vector; An applying module, for respectively applying a plurality of reference non-through voltage vectors to the quasi-Z-source inverter; A prediction module is configured to obtain, for any reference non-through voltage vector, an initial inductor current sampling value, an initial capacitor voltage sampling value, and an initial output current sampling value of the quasi-Z-source inverter under the reference non-through voltage vector; determine a predicted capacitor voltage value at a future time based on the initial inductor current sampling value and the initial capacitor voltage sampling value, and determine a predicted output current value at a future time based on the initial output current sampling value; A first screening module is configured to quantify the difference between the output current prediction value and the output current reference value under each reference non-through voltage vector, and screen multiple candidate non-through voltage vectors from the multiple reference non-through voltage vectors according to the quantization results; The second screening module is used to quantify the difference between the capacitor voltage prediction value and the capacitor voltage reference value under each candidate non-through voltage vector, and determine the target non-through voltage vector; The control module is used to control the quasi-Z-source inverter by applying the action sequence of the through voltage vector and the non-through voltage vector within the control period of the quasi-Z-source inverter, the duty cycle of applying the through voltage vector, and the target non-through voltage vector.

[0014] The present invention provides a computer-readable storage medium, wherein the storage medium stores a computer program, and when the computer program is executed by a processor, the control method of the quasi-Z-source inverter is implemented.

[0015] The present invention provides a computer device, comprising a memory, a processor, and a computer program stored in the memory and executable on the processor, wherein the processor implements the control method of the quasi-Z-source inverter when executing the program.

[0016] At least one of the above technical solutions adopted by the present invention can achieve the following beneficial effects: Since the inductor current error affects the output current control accuracy, the action sequence of applying the through-voltage vector and the non-through-voltage vector and the duty cycle of applying the through-voltage vector within the control period of the quasi-Z-source inverter are determined based on the inductor current error. Applying the obtained action sequence and duty cycle to the quasi-Z-source inverter can improve the control accuracy of the quasi-Z-source inverter over the output current. Furthermore, priority screening is performed on multiple reference non-through-voltage vectors based on the gap between the output current predicted value and the output current reference value and the gap between the capacitor voltage predicted value and the capacitor voltage reference value to determine the target non-through-voltage vector, so that the target non-through-voltage vector has a priority effect on the output current control accuracy, thereby further improving the control accuracy of the output current of the quasi-Z-source inverter. BRIEF DESCRIPTION OF THE DRAWINGS

[0017] The drawings described herein are used to provide a further understanding of the present invention and constitute a part of the present invention. The exemplary embodiments of the present invention and their descriptions are used to explain the present invention and do not constitute an improper limitation of the present invention. In the drawings:

[0018] Figure 1 A schematic flow chart of a control method for a quasi-Z-source inverter provided by the present invention; Figure 2 A topological structure diagram of a quasi-Z-source inverter provided by the present invention; Figure 3 A structural diagram of a quasi-Z-source inverter provided by the present invention in a through state and a non-through state; Figure 4 The schematic diagram of the deadbeat control of the inductor current of a quasi-Z-source inverter is shown in FIG. Figure 5 is an inductor current waveform diagram with an initial error greater than zero; Figure 6 An inductor current waveform diagram with an initial error less than zero; Figure 7 A schematic diagram of a feasible region of initial error of inductor current provided by the present invention; Figure 8 A waveform diagram of the inductor current when the initial error of the inductor current is less than zero and the method of the present invention; Figure 9 A waveform diagram of the inductor current when the initial error of the inductor current is greater than zero and the method provided by the present invention; Figure 10 A flowchart of sequential model predictive control implementation of qZSI provided by the present invention; Figure 11 A control structure diagram of a control method for a quasi-Z-source inverter provided by the present invention; Figure 12A schematic diagram of a control device for a quasi-Z-source inverter provided by the present invention; Figure 13 A schematic diagram of a computer device for implementing a control method for a quasi-Z-source inverter provided by the present invention. DETAILED DESCRIPTION

[0019] To make the objectives, technical solutions, and advantages of the present invention more clear, the technical solutions of the present invention will be clearly and completely described below in conjunction with specific embodiments of the present invention and corresponding drawings. Obviously, the embodiments described are only some embodiments of the present invention, not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without making creative efforts are within the scope of protection of the present invention.

[0020] The qZSI (qZSI) is a new type of power converter with a unique buck-boost mechanism, which has important practical implications for improving overall power converter performance. First, the qZSI allows for direct conduction of the bridge arm, reducing the risk of inverter damage. Second, the qZSI eliminates the need for dead time in the pulse-width modulation (PWM) signal, reducing output voltage distortion. Finally, the qZSI topology integrates DC buck-boost and AC-DC conversion, reducing hardware and software costs. Since its introduction in 2008, the qZSI has rapidly developed, encompassing applications in AC motor speed regulation, fuel cell power generation systems, distributed generation, and microgrids.

[0021] The development of microprocessors has promoted the application of high-performance industrial control methods in power electronics. Among the many control methods, model predictive control (MPC) has been a hot topic in recent years. Based on the discrete nature of the inverter, MPC uses a predictive model that includes switching states to estimate the future state of the system. A cost function is used to evaluate the system state to select the set of switching states with optimal performance. Consequently, without the need for a PWM modulator or PI regulator, MPC can simplify the multi-stage cascade control structure and eliminate the non-minimum phase problem in qZSI control. Furthermore, MPC applies mathematical optimization theory to solve nonlinear and multivariable constrained control problems, offering advantages such as simple principles and fast dynamic response.

[0022] While using optimization theory in MPC can improve the performance of qZSI, overreliance on optimization algorithms can extend microprocessor execution time. Consequently, computational complexity, control performance, and control parameter optimization are mutually constrained. Complicated weight coefficient tuning, large steady-state errors, and high computational complexity are the main challenges facing MPC. Many researchers have leveraged methods from vector control, deadbeat control, and direct torque control to improve MPC.

[0023] The weight coefficient is a key parameter that determines the performance of MPC control. The MPC method of qZSI matches three weight coefficients and requires a tedious trial-and-error method to achieve the best parameter matching. For example, the model prediction algorithm of the qZSI single-stage topology is improved by eliminating the weight coefficient by using multi-objective optimization and defining a sub-cost function. A sequential model predictive control method is proposed, which can balance the weight of each control variable without adjusting the weight coefficient, and has good steady-state performance and dynamic characteristics. A qZSI model predictive control method based on logical operations is proposed. The control of the qZSI inductor current, capacitor voltage and output current is implemented in two special logic modules respectively, with good decoupling control capability and no need to adjust the weight coefficient. An improved MPC method for photovoltaic power generation system based on qZSI is proposed. By calling the prediction model and cost function in one control loop, the steps of implementing multivariable collaborative control are simplified.

[0024] The ultimate result of the MPC operation is an optimal voltage vector. However, current MPC research indicates that using more voltage vectors can effectively improve MPC control accuracy. For example, a method based on an established discrete-time average model of the circuit is proposed to predict the required through-duty cycle and modulation signal for qZSI, achieving fast dynamic response and precise steady-state tracking performance. A multi-vector model predictive power control for grid-connected qZSI is proposed. This method uses an improved sliding-mode control method to constrain the capacitor voltage and inductor current near a reference value, achieving weight-free coefficient control of the grid and inductor currents and reducing current ripple. A high-precision model predictive power control method for qZSI with battery energy storage is proposed, which improves the control accuracy of the inductor and output currents by overcoming the shortcomings of deadbeat control and dual-vector model predictive control. The through-voltage vector, zero voltage vector, and effective voltage vector are combined into a joint voltage vector to approach the target voltage vector. By calculating the duty cycle, deadbeat control of the inductor current and effective voltage vector amplitude control of the output current are achieved, reducing inductor current ripple and output current error.

[0025] Building on existing research, this paper further explores the mechanism by which the inductor current of a qZSI (quasi-Z-source inverter) affects the output current control accuracy. It identifies a common problem with existing control methods: the coupling and mutual influence between the inductor current and the output current are ignored. This phenomenon manifests itself as inductor current errors being converted into the output current through the power balance relationship, increasing the output current error. Output current errors, in turn, cause input power variations, which in turn affect the inductor current. To address this issue, this paper proposes a control method for a quasi-Z-source inverter. Unlike the deadbeat control principle of inductor current, this method employs the area method and the current error symmetry principle to minimize inductor current ripple and limit the average inductor current. Therefore, while maintaining inductor current control performance, the proposed method indirectly improves output current control accuracy. Furthermore, based on the control characteristics of the qZSI AC and DC sides, a sequential MPC (Multi-Purpose Control) method is designed to balance the capacitor voltage and output current. Experimental results validate the correctness and feasibility of the proposed method.

[0026] The technical solutions provided by various embodiments of the present invention are described in detail below with reference to the accompanying drawings.

[0027] Figure 1 The figure is a flow chart of a control method for a quasi-Z-source inverter in the present invention, which specifically includes the following steps: S101, obtaining an inductor current sampling value of the quasi-Z-source inverter, and determining, based on an error between the inductor current sampling value and an inductor current reference value, an action sequence of applying a through-voltage vector and a non-through-voltage vector within a control period of the quasi-Z-source inverter and a duty cycle of applying the through-voltage vector.

[0028] like Figure 2 As shown, Figure 2 The topology diagram of a quasi-Z-source inverter is shown below. The quasi-Z-source inverter consists of a quasi-Z-source network and a three-phase inverter. Unlike conventional inverters, the qZSI not only controls the three-phase output current but also the inductor current and capacitor voltage on the DC side. The qZSI's boost function is achieved by adding a shoot-through duty cycle to the PWM.

[0029] qZSI has two working states, namely direct-through state and non-direct-through state. Figure 3 As shown in (a), in the non-through state (including zero state and effective state), the current of inductor L1 and the voltage of capacitor C1 can be expressed as:

[0030] (1); (2); in, and are the capacitance and inductance of the quasi-Z source network respectively; and They are DC power supply and capacitor voltage; It is an inductor Current; is the output current of the quasi-Z source network, which can be obtained by solving the switch state function of formula (3).

[0031] (3); in, 、 and is the three-phase output current, 、 、 The switch state can be 0 or 1, "0" means off, "1" means on. The equivalent circuit diagram of the through state of qZSI is as follows: Figure 3 According to the symmetry principle of qZSI, inductors with the same inductance and The current and voltage are the same. Therefore, the inductor The current and voltage can be measured using the inductor Instead of the current and voltage, their voltage and current equations can be expressed as:

[0032] (4); (5).

[0033] Output voltage in αβ coordinate system It can be expressed as: (6); in, is the DC link voltage, For a three-phase resistance-inductance (RL) load, formula (6) can be expressed as:

[0034] (7); in, yes 、 Output current in the coordinate system; R and L RL is the resistance and inductance of the load.

[0035] The inductor current, capacitor voltage, and output current are the three controlled quantities of the qZSI. The inductor current is primarily affected by the duration of the through-voltage vector on the DC side of the qZSI, while the output current is affected by the duration of the effective voltage vector and the zero voltage vector on the AC side of the qZSI. Analysis of the qZSI model predictive control in the prior art indicates that there is no absolute correlation between the inductor current and the output current in terms of control strategy, so the output current control strategy does not need to consider the impact of the inductor current. This embodiment, by analyzing the relationship between the input and output power of the qZSI, demonstrates that an inductor current control error will cause an amplitude deviation of the output current, i.e., there is a coupling relationship between the inductor current and the output current. To this end, the average value and error of the inductor current under different initial states are calculated and comprehensively analyzed in detail.

[0036] Output current error analysis: According to the working principle of the qZSI topology, the input current on the DC side is the inductor current, and the output current on the AC side flows through the load. The input power and output power of the qZSI can be expressed as:

[0037] (8); in, is the input power, is the output power, is the inductor current of qZSI, which is also the input current. Its average value and effective value are equal. is the DC supply voltage. is the effective value of the output current, Z is the load impedance, and after coordinate transformation, It can be expressed as:

[0038] (9).

[0039] According to the power conservation principle of qZSI, the input power and output power in formula (8) are equal. Therefore, it can be derived that and The relationship between them is:

[0040] (10).

[0041] From formula (10), we can see that the square of the load current is proportional to the inductor current. If the inductor current control error is large, the output current performance will be affected. Assume that the inductor current error and output current error are and , substitute the inductor current and output current containing the error, as well as the reference inductor current and output current into formula (10) for comparison, and we get

[0042] (11); in, is the reference value of the inductor current, is the instantaneous value of the output current. From formula (11), we can derive for:

[0043] (12).

[0044] From formula (12), we can see that as long as the inductor current generates a The output current will get an error error. This will cause the output power to change, which in turn affects the input power and inductor current, forming a relationship in which the inductor current and output current are coupled and influence each other.

[0045] Prior art has reduced output current control errors by continuously improving the voltage vectoring mechanism. While this significantly improves output current performance, the above analysis demonstrates that prior art overlooks a key factor: the inductor current control method also affects output current error. Therefore, the present invention analyzes inductor current control errors and demonstrates that output current errors caused by inductor current control cannot be directly eliminated by improving the output current control method.

[0046] In an exemplary embodiment, analyzing the inductor current error specifically includes: Among them, deadbeat control is a method that is frequently used to reduce inductor current errors. The prior art proposes an inductor current deadbeat control method based on the idea of ​​space voltage vector synthesis. This method uses a combination of direct and non-direct voltage vectors to control the variation range of the inductor current. The principle of inductor current deadbeat control is as follows: Figure 4 As shown, by calculating the action time of the two voltage vectors, the inductor current At the end of the control cycle, the The inductor current error varies continuously as the inductor current changes dynamically. When the direct voltage vector switches to a non-direct voltage vector, the inductor current error reaches its maximum value; at the end of the control cycle, the inductor current error reaches its minimum value. εi L1-A is the maximum error of the inductor current 。

[0047] However, the existing technology overlooks a key issue: the inductor current error is determined by its maximum value, not its minimum value. This is primarily due to the fact that the impact of two voltage vectors on the inductor current is not equivalent to the impact of a single composite voltage vector on the inductor current. This fundamentally differs from the principle of space voltage vector synthesis applicable to power converters. Therefore, the impact of each voltage vector on the inductor current must be analyzed separately. Furthermore, the initial inductor current error during the dynamic process of qZSI is uncertain, which in turn affects the maximum inductor current error over the entire control cycle. This embodiment divides the initial error into three cases, analyzing how the inductor current varies over the entire control cycle.

[0048] (1) The initial error of the inductor current is zero When the initial error of the inductor current ε When 0 is zero, it indicates that qZSI is in steady state and the inductor current is equal to its reference command at the beginning and end of the control cycle. The maximum error of the inductor current occurs at the moment of voltage vector switching. Please continue to refer to Figure 4 In one control cycle T s The action time of the through voltage vector is T sh , the non-through voltage vector action time is T s - T sh According to formula (1), the maximum control error of the inductor current is Export as:

[0049] (13); in, , ， ; d sh is the duty cycle of the shoot-through voltage vector, G is the voltage gain of the Z-source inverter. v C1 and T sh Substituting into formula (13), It can be expressed as

[0050] (14).

[0051] (2) The initial error of the inductor current is greater than zero When the initial error of the inductor current ε When 0 is greater than zero, it indicates that qZSI is in a transient state, such as Figure 5 As shown, Figure 5is the inductor current waveform when the initial error is greater than zero. Since the inductor current is greater than its reference command at the beginning of the control cycle, the switching time of the direct and non-direct voltage vectors is compared with ε 0=0 is advanced. Therefore, ε Maximum inductor current error when 0>0 εi L1-B Greater than ε Maximum inductor current error when 0=0 εi L1-A In order to ensure the effectiveness of deadbeat control, ε The minimum value of 0 is equal to 0, and the maximum value is equal to the current ripple value generated by applying the non-through voltage vector during the entire control cycle, ε The range of 0 can be derived as:

[0052] (15).

[0053] Will v C1 Substituting the expression into formula (15), we can get: (16).

[0054] Since the maximum error of the inductor current εi L1-B Appears at the switching moment of the two vectors, using auxiliary lines to construct similar triangles to obtain εi L1-A and εi L1-B The relationship between them is: (17); Among them, Δ t yes i L1,ref The reverse extension line at time zero. According to formula (17), εi L1-B It can be deduced as:

[0055] (18).

[0056] Will εi L1-A and T sh Substituting into formula (18), εi L1-B It can be deduced as: (19).

[0057] (3) The initial error of the inductor current is less than zero When the initial error of the inductor current ε When 0 is less than zero, it indicates that qZSI is in a transient state, such as Figure 6 As shown, Figure 6 The inductor current waveform is shown when the initial error is less than zero. Since the inductor current is less than its reference command at the beginning of the control cycle, the switching time between the direct and non-direct voltage vectors is shorter than that of the inductor current. ε 0 = 0 will lag. Therefore, ε Maximum inductor current error when 0<0 εi L1-C Less than ε Maximum inductor current error when 0=0 εi L1-A In order to ensure the effectiveness of deadbeat control, ε The minimum value of 0 is equal to 0, and the maximum value is equal to the current ripple value generated by applying the through voltage vector during the entire control cycle, ε The range of 0 can be derived as:

[0058] (20).

[0059] Will v C1 Substituting the expression into formula (20), we can get: (twenty one).

[0060] exist Figure 6 Draw auxiliary lines to construct similar triangles, and we get εi L1-A and εi L1-C Relationship: (twenty two).

[0061] According to formula (22), εi L1-C It can be expressed as: (twenty three).

[0062] Will εi L1-A and T sh Substituting into formula (23), εi L1-C It can be expressed as: (twenty four).

[0063] Based on the above analysis, it can be seen that if the average value of the inductor current in the control cycle does not coincide with its reference value, the resulting error will be converted into the output current through the power conservation relationship. The deviation of the output current from its reference instruction will cause the input power to change, which in turn affects the inductor current. To this end, the present invention starts from the source of the problem and adopts an area method and error symmetry principle to eliminate the average value error of the inductor current. The principle is to make the areas enclosed by the inductor currents on both sides of the reference instruction equal on the time axis, and to ensure that the positive and negative current errors are symmetrical, thereby offsetting the inductor current error. Because the method proposed by the present invention is essentially different from the goals and principles of zero-beat control, the action order and duration of their voltage vectors are completely different.

[0064] In order to ensure the effectiveness of the voltage vector combination, the initial error of the inductor current must satisfy equations (15) and (21), which can be expressed as: (25).

[0065] The feasible domain of the initial error of the inductor current of the method proposed in the present invention is as follows: Figure 7 If the initial error of the inductor current is ε 0 exceeds the feasible region of the formula, then only the through voltage vector or the non-through voltage vector will be applied in one control cycle. ε 0 In the feasible region, the action order of the through-voltage vector and the non-through-voltage vector is interchangeable.

[0066] Specifically, according to the error between the inductor current sampling value and the inductor current reference value, the order of applying the through-voltage vector and the non-through-voltage vector within the control period of the quasi-Z-source inverter is determined, including: when the error is less than or equal to 0, determining that the through-voltage vector is applied first and then the non-through-voltage vector is applied within the control period of the quasi-Z-source inverter; when the error is greater than 0, determining that the non-through-voltage vector is applied first and then the through-voltage vector is applied within the control period of the quasi-Z-source inverter.

[0067] It should be noted that, when the error is less than or equal to 0, it can also be determined that the non-through voltage vector is applied first and then the through voltage vector is applied within the control period of the quasi-Z-source inverter.

[0068] Furthermore, a duty cycle of a through-voltage vector applied within a control period of the quasi-Z-source inverter is determined based on an error between a sampled inductor current value and a reference inductor current value, including: determining an action time of the through-voltage vector within the control period based on the error between the sampled inductor current value and the reference inductor current value; and determining a duty cycle of the through-voltage vector applied within the control period of the quasi-Z-source inverter based on the action time of the through-voltage vector within the control period.

[0069] The inductor current of the method proposed in the present invention will be analyzed below based on the initial error.

[0070] (1) The initial error of the inductor current is less than zero like Figure 8 As shown, Figure 8 The initial error of the inductor current is given by ε When 0 is less than zero, the inductor current waveforms of the deadbeat control and the method of the present invention are represented by dotted and solid lines respectively. It can be seen that the voltage vector switching moment of the method of the present invention is earlier than that of the deadbeat control. S abc The area of ​​the triangle is equal to S cde , the average value of the inductor current during the entire control cycle is close to the reference value of the inductor current, and the average error of the inductor current is zero. Therefore, the relationship between the inductor current at the time of voltage vector switching and the end of the control cycle can be derived from the similar triangle: the difference between the inductor current and its reference command at the time of voltage vector switching is equal to the difference between the inductor current and its reference command at the end of the control cycle, that is, Figure 9 In the equation ba=ed, it can be expressed as:

[0071] (26); in, is the action time of the through voltage vector within the control period, is the voltage gain of the quasi-Z-source inverter, is the control cycle length of the quasi-Z source inverter, is the inductor, is the DC supply voltage of the Z-source inverter, is the error between the inductor current sampling value and the inductor current reference value.

[0072] Will Substituting the expression into formula (26), the switching time of the voltage vector can be obtained as: (27).

[0073] Substituting formula (27) into formula (26), the maximum inductor current error can be calculated as: (28).

[0074] (2) The initial error of the inductor current is greater than zero Figure 9 The initial error of the inductor current is given by εThe inductor current waveforms of the deadbeat control and the method provided by the present invention when 0 is greater than zero are represented by dotted lines and solid lines respectively. It can be seen that the order of action of the voltage vector in the proposed method is completely different from that of the deadbeat control. By swapping the order of action of the direct voltage vector and the non-direct voltage vector, the control error can be avoided from being accumulated on the basis of the initial error of the inductor current. Therefore, the non-direct voltage vector is applied first to reduce the inductor current, and then the direct voltage vector is applied to ensure that the error of the inductor current within the control cycle is symmetrical with respect to the inductor current reference instruction. When the triangle S abc The area of ​​the triangle is equal to S cde The average error of the inductor current is zero. This relationship can be expressed as:

[0075] (29).

[0076] Will v C1 Substituting the expression into formula (29), the switching time of the voltage vector can be obtained as: (30).

[0077] Substituting formula (30) into formula (29), the maximum error of the inductor current can be calculated as: (31).

[0078] Therefore, according to the error between the inductor current sampling value and the inductor current reference value, the action time of the through-voltage vector within the control cycle is determined, including: when the error is less than or equal to 0, the calculation method for determining the action time of the through-voltage vector within the control cycle is shown in formula (27); when the error is greater than 0, the calculation method for determining the action time of the through-voltage vector within the control cycle is shown in formula (30).

[0079] Then, according to the action time of the through voltage vector in the control cycle, the duty cycle can be determined by calculate.

[0080] S102 , applying a plurality of reference non-through voltage vectors to the quasi-Z-source inverter respectively.

[0081] Combining the characteristics of the qZSI's shoot-through and non-shoot-through states with the advantages of sequential MPC, this paper designs a sequential MPC method suitable for qZSI to control capacitor voltage and output current. When the qZSI is in the non-shoot-through state, the inductor current decreases, while in the shoot-through state, the inductor current increases. Therefore, the shoot-through voltage vector and the non-shoot-through voltage vector are applied sequentially within a control cycle. The non-shoot-through voltage vectors include six effective voltage vectors and one zero voltage vector, which are used when performance requires balancing capacitor voltage and output current.

[0082] Therefore, based on the order of applying the through-voltage vector and the non-through-voltage vector within the control cycle of the quasi-Z-source inverter and the duty cycle of applying the through-voltage vector obtained in the above embodiment, a reference through-voltage vector and a non-through-voltage vector can be applied to the quasi-Z-source inverter. Since there are multiple types of non-through-voltage vectors, multiple reference non-through-voltage vectors can be applied to the quasi-Z-source inverter in sequence, and the inductor current, capacitor voltage, and output current under each reference non-through-voltage vector can be collected.

[0083] S103, for any reference non-through voltage vector, obtain the initial inductor current sampling value, initial capacitor voltage sampling value and initial output current sampling value of the quasi-Z-source inverter under the reference non-through voltage vector; determine the capacitor voltage prediction value at a future time based on the initial inductor current sampling value and the initial capacitor voltage sampling value, and determine the output current prediction value at a future time based on the initial output current sampling value.

[0084] The inductor current, capacitor voltage and output current collected under multiple benchmark non-through voltage vectors are used as the initial inductor current sampling value, initial capacitor voltage sampling value and initial output current sampling value respectively. Then, a prediction model can be used to combine the initial inductor current sampling value and the initial capacitor voltage sampling value to determine the capacitor voltage prediction value at a future moment, and based on the initial output current sampling value, the output current prediction value at a future moment can be determined.

[0085] Specifically, the process of building a prediction model includes: The Euler method is used to discretize formulas (1) and (2), and the predicted values ​​of the inductor current and capacitor voltage in the non-shoot-through state are obtained as follows: (32); (33); in, They are time Coordinate axes and The output current prediction value on the coordinate axis, for The predicted value of the capacitor voltage at time , The moment is the future moment, is the initial moment, The control cycle duration.

[0086] Similarly, after discretization of formulas (4) and (5), the predicted values ​​of the inductor current and capacitor voltage in the through state are obtained as follows: (34); (35).

[0087] The discrete-time model of the output current in formula (7) can be derived as: (36); Where, k 、 T s and k - T s Represent the current moment, control cycle and previous moment respectively. Recursively forward the time by one moment, and the predicted value of the output current is

[0088] (37).

[0089] Therefore, the capacitor voltage prediction value and output current prediction value at the future moment can be predicted according to formulas (32), (33) and (36).

[0090] S104, quantifying the difference between the output current prediction value and the output current reference value under each reference non-through voltage vector, and screening multiple candidate non-through voltage vectors from the multiple reference non-through voltage vectors based on the quantization results; and quantifying the difference between the capacitor voltage prediction value and the capacitor voltage reference value under each candidate non-through voltage vector, to determine a target non-through voltage vector.

[0091] According to the sequential MPC principle, the cost functions of capacitor voltage and output current need to be designed separately, and the optimal voltage vector is screened out in turn. The capacitor voltage cost function can be expressed as:

[0092] (38); in, is the capacitor voltage cost function value, is the capacitor voltage reference value.

[0093] The output current cost function can be expressed as: (39); in, is the output current cost function value, They are Coordinate axes and Output current reference value on the coordinate axis.

[0094] Optionally, the gap between the output current prediction value and the output current reference value for each benchmark non-through voltage vector is quantified, including: for any benchmark non-through voltage vector, quantizing the gap between the output current prediction value and the output current reference value using an output current cost function to obtain an output current cost function value; the quantization result is the output current cost function value; and then determining the benchmark non-through voltage vectors corresponding to a preset number of minimum output current cost function values ​​as candidate non-through voltage vectors. The gap between the capacitor voltage prediction value and the capacitor voltage reference value for each candidate non-through voltage vector is quantified, and a target non-through voltage vector is determined, including: for any candidate non-through voltage vector, quantizing the gap between the capacitor voltage prediction value and the capacitor voltage reference value using a capacitor voltage cost function to obtain a capacitor voltage cost function value; and then determining the candidate non-through voltage vector corresponding to the minimum capacitor voltage cost function value as the target non-through voltage vector.

[0095] Specifically, first calculate the output current cost function in formula (39), and select three candidate voltage vectors with smaller cost function values ​​from the seven benchmark non-through voltage vectors. Then, substitute these three candidate voltage vectors into the capacitor voltage cost function in formula (38) to select the optimal voltage vector. Another way is to swap the order of selecting voltage vectors by the capacitor voltage and output current cost functions. Obviously, the first order MPC is better than the second order MPC in improving the output current performance. The flowcharts of the two methods are shown in Figure 2. Figure 10 Regardless of the order, MPC does not require the use of weight coefficients.

[0096] S105 , controlling the quasi-Z-source inverter by applying the through voltage vector and the non-through voltage vector in a control period of the quasi-Z-source inverter, the duty cycle of applying the through voltage vector, and the target non-through voltage vector.

[0097] In an exemplary embodiment, the steady-state inductor current is analyzed: regardless of the initial error of the inductor current ε Regardless of the value of 0, the control error at the end of the current cycle will be used as the initial inductor current error of the next control cycle. ε 0. After a transient state, ε 0 will eventually satisfy ε 0= = Substituting this relationship into equations (28) and (31), the steady-state error of the inductor current is:

[0098] (40).

[0099] From formula (40), it can be seen that the inductor current error is a constant value and is proportional to the initial inductor current error. ε Based on the above analysis, it can be concluded that the differences between the effects of the proposed method and the deadbeat control on the inductor current are: 1) the proposed method can completely eliminate the average error of the inductor current; 2) the maximum error of the inductor current in the proposed method is much lower than that in the deadbeat control.

[0100] In an exemplary embodiment, the zero-beat control can reduce the inductor current ripple of the qZSI, but ignores the impact of the average error of the inductor current in the control cycle on the output current. The prior art has not yet conducted in-depth analysis and improvement on this issue. The present invention proposes a control method for qZSI, which is a qZSI AC and DC side current decoupling control method based on model prediction, which can minimize the error of the inductor current and its negative impact on the output current, thereby improving the control accuracy of the output current. At the same time, the method proposed in the present invention has the same advantages as the existing control method, such as small amount of calculation and no need for weight coefficient adjustment. The control structure of the method proposed in the present invention is as follows: Figure 11 As shown in the figure, the proposed method includes qZSI state sampling and prediction, initial inductor current error determination, voltage vector combination, the area method and error symmetry principle, and priority control of output current and capacitor voltage. First, the inductor current, capacitor voltage, and output current on the DC and AC sides of the qZSI are sampled, and the state at the next moment is predicted. Then, using the area method and error symmetry principle, and based on the relationship between the inductor current reference value and the inductor current sample value, the duty cycle and action sequence of the through-voltage vector and non-through-voltage vector combination are calculated. Finally, the sequential MPC principle is used to select a voltage vector that balances the capacitor voltage and output current performance.

[0101] It should be noted that this method collects the inductor current sampling value of the quasi-Z-source inverter at the current moment, and then generates a control strategy for the quasi-Z-source inverter based on the sampled inductor current sampling value (the order of applying the through-voltage vector and the non-through-voltage vector within the control period of the quasi-Z-source inverter, the duty cycle of applying the through-voltage vector, and the target non-through-voltage vector applied), and controls the quasi-Z-source inverter according to the control strategy.

[0102] To fully compare the differences in inductor current control performance between the proposed method and deadbeat control, Table 1 lists their calculation formulas for the maximum inductor current error. Table 1 plots the maximum inductor current error for different voltage gains and control periods. It can be concluded that the maximum inductor current error increases with increasing control period and DC voltage gain. When the initial inductor current error is greater than zero, the maximum inductor current error of the proposed method is much smaller than that of deadbeat control. When the initial inductor current error is less than zero, the maximum inductor current error of the proposed method is similar to that of deadbeat control. In steady state, the maximum inductor current error of the proposed method is only half that of deadbeat control. Therefore, the proposed method can not only significantly reduce the maximum inductor current error, but also eliminate the average inductor current error, which is of great significance for eliminating the output current error caused by the inductor current control error.

[0103] Table 1 In an exemplary embodiment, a three-phase qZSI experimental platform was constructed to verify the impact of the inductor current control method analyzed in this invention on output current error and the effectiveness of the proposed method in resolving this issue. The experimental platform consists of a power supply, a three-phase qZSI control system, and an RL load. The qZSI system control algorithm was implemented in a TMS320F28335 digital signal processor. The sampling frequency was set to 15 kHz to meet the requirements of the proposed method and the deadbeat control execution time. The inductance and capacitance of the qZSI DC side were 2 mH and 560 μF, respectively. The three-phase resistive-inductive load had a resistance of 10 Ω and an inductance of 7.7 mH.

[0104] Verification of the inductor current control problem: In a steady-state experiment, the supply voltage was 100V, the output power was 500W, and the voltage gain was 2, which means the DC link voltage was increased to 200V. The reference commands for the DC link voltage and capacitor voltage were set to 200V and 150V, respectively, and the reference value for the output current was set to 5.7A. The proposed method and deadbeat control were used to control the qZSI, and the errors in the inductor current and output current were observed.

[0105] Steady-state experimental results show that the capacitor voltage amplitude is 148V and the DC link voltage is 197V. The reference value of the inductor current is 5A, and the inductor current ripple is 1.2A. Regardless of the method used, the qZSI exhibits stable inductor current, capacitor voltage, output current, and DC link voltage. However, the main differences between the experimental results of the two methods lie in the maximum, minimum, and average values ​​of the inductor current, as well as the peak output current. The inductor current of the proposed method fluctuates around its reference value, ranging from 4.4A to 5.6A. Therefore, the average value of the inductor current is almost equal to its reference value, and there is no error in the average inductor current. However, the inductor current of the deadbeat control method is higher than its reference value, ranging from 4.8A to 6.1A, and the average value of the inductor current is 0.6A higher than the reference value. On the other hand, the output current peak values ​​of the proposed method and the deadbeat control method are approximately 5.75A and 6.2A, respectively, with output current errors of 0.05A and 0.5A, respectively. The steady-state experimental results show that the deadbeat control cannot accurately control the average value of the inductor current, which leads to output current error. The method proposed in the present invention can effectively solve this problem.

[0106] Dynamic experiments comparing tracking performance: To compare the tracking capabilities of the proposed method with those of deadbeat control, dynamic experiments involving output current and DC link voltage step changes were designed. In the output current step change experiments, the reference output current value was reduced from 8.2 A to 4.5 A. In the experiments involving simultaneous step changes in output current and DC link voltage, the reference DC link voltage and output current values ​​were reduced from 200 V to 150 V and 8 A to 5.5 A, respectively.

[0107] The experimental results for an output current step show that, according to the power conservation principle, when the output current on the AC side decreases from 8.2 A to 4.5 A, the inductor current reference value on the DC side decreases from 9.7 A to 3.2 A. Throughout the dynamic process, the capacitor voltage and DC link voltage remain constant at 150 V and 200 V, respectively, and the inductor current ripple remains constant at 1.2 A. It can be seen that both the proposed method and deadbeat control exhibit fast response speeds during the dynamic process. The inductor current and output current of the proposed method both track the reference values ​​well, with a steady-state error of less than 0.1 A. However, after the transition process, the inductor current error of the deadbeat control is approximately 0.6 A, and the output current error is approximately 0.3 A, both of which are higher than those of the proposed method.

[0108] The experimental results of simultaneous step changes in the output current and DC link voltage show that when the reference values ​​of the DC link voltage and output current change, the capacitor voltage and DC link voltage drop from 150V to 125V and 200V to 150V, respectively. Simultaneously, the reference value of the inductor current decreases from 9.7A to 4.5A. Due to the change in the DC link voltage, the inductor current ripple decreases from 1.3A to 1A. In terms of dynamic performance, the output current and inductor current of the proposed method and deadbeat control can quickly follow their reference instructions, but the DC link voltage and capacitor voltage change more slowly. In terms of steady-state performance, the inductor current error of the deadbeat control is 0.5A, resulting in a 0.27A deviation in the output current; the steady-state errors of the output current and inductor current of the proposed method are both less than 0.1A. The dynamic experiments show that the proposed method has the same dynamic performance as deadbeat control and better control accuracy than deadbeat control.

[0109] This paper analyzes the output power and inductor current errors of a qZSI (qZSI) circuit, pointing out the coupling and mutual influence between the DC and AC side currents. To address this coupling and mutual influence between the inductor current and output current in qZSI deadbeat control, the paper first analyzes the mechanism that influences the output current during inductor current control. It then calculates the maximum inductor current error generated by deadbeat control under different initial errors. Finally, based on the area method and the principle of current error symmetry, a model-predicted decoupling control method for the AC and DC side currents of a qZSI circuit is proposed. This method ensures that the average inductor current within a control cycle is equal to its reference value, ensuring that the output current closely follows the reference value, thereby improving the control accuracy of both the inductor and output currents. The proposed method not only reduces the maximum inductor current error within a control cycle but also eliminates the average inductor current error and the resulting output current amplitude deviation. Experimental results demonstrate that the proposed method improves the control accuracy of both the inductor and output currents, effectively improving the steady-state performance of deadbeat control while maintaining low computational complexity and implementation complexity.

[0110] When applying the control method of the quasi-Z source inverter provided by the present invention, it is not necessary to Figure 1 The steps are executed in the order shown. The specific execution order of the steps can be determined according to needs, and the present invention does not limit this.

[0111] The above is a control method for a quasi-Z source inverter provided by one or more embodiments of the present invention. Based on the same idea, the present invention also provides a corresponding control device for a quasi-Z source inverter, such as Figure 12 shown.

[0112] Figure 12 A schematic diagram of a control device for a quasi-Z-source inverter provided by the present invention, the device 1200 includes: An acquisition module 1201 is configured to acquire an inductor current sampling value of the quasi-Z-source inverter and determine, based on an error between the inductor current sampling value and an inductor current reference value, an order of applying a through-voltage vector and a non-through-voltage vector within a control period of the quasi-Z-source inverter and a duty cycle of applying the through-voltage vector. An applying module 1202 is configured to apply a plurality of reference non-through voltage vectors to the quasi-Z-source inverter respectively; The prediction module 1203 is configured to obtain, for any reference non-through voltage vector, an initial inductor current sampling value, an initial capacitor voltage sampling value, and an initial output current sampling value of the quasi-Z-source inverter under the reference non-through voltage vector; determine a predicted capacitor voltage value at a future time based on the initial inductor current sampling value and the initial capacitor voltage sampling value, and determine a predicted output current value at a future time based on the initial output current sampling value; A first screening module 1204 is configured to quantify the difference between the output current prediction value and the output current reference value under each reference non-through voltage vector, and screen multiple candidate non-through voltage vectors from the multiple reference non-through voltage vectors according to the quantization results; The second screening module 1205 is configured to quantify the difference between the capacitor voltage prediction value and the capacitor voltage reference value under each candidate non-through voltage vector, and determine a target non-through voltage vector; The control module 1206 is used to control the quasi-Z-source inverter by applying the through voltage vector and the non-through voltage vector in the control period of the quasi-Z-source inverter, the duty cycle of the through voltage vector, and the target non-through voltage vector.

[0113] For specific definitions of the control device for a quasi-Z-source inverter, please refer to the definitions of the control method for a quasi-Z-source inverter described above and will not be repeated here. Each module in the control device for the quasi-Z-source inverter described above can be implemented in whole or in part through software, hardware, or a combination thereof. Each of the modules described above can be embedded in or independent of a processor in a computer device in hardware form, or can be stored in a memory in a computer device in software form, so that the processor can call and execute the corresponding operations of each of the modules described above.

[0114] The present invention also provides a computer-readable storage medium, which stores a computer program, which can be used to execute the above Figure 1 A control method for a quasi-Z-source inverter is provided.

[0115] The present invention also provides Figure 13 The structural diagram of the computer equipment shown in FIG. Figure 13As shown in the figure, at the hardware level, the computer device includes a processor, an internal bus, a network interface, a memory, and a non-volatile memory. Of course, it may also include other hardware required for the business. The processor reads the corresponding computer program from the non-volatile memory into the memory and then runs it to achieve the above Figure 1 A control method for a quasi-Z-source inverter is provided.

[0116] Those skilled in the art will appreciate that all or part of the processes in the above-described method embodiments can be implemented by instructing the relevant hardware using a computer program. The computer program can be stored in a non-volatile computer-readable storage medium. When executed, the computer program can include the processes in the above-described method embodiments. Any reference to memory, storage, database, or other media used in the various embodiments provided herein may include at least one of non-volatile and volatile memory. Non-volatile memory may include read-only memory (ROM), magnetic tape, floppy disk, flash memory, or optical storage. Volatile memory may include random access memory (RAM) or external cache memory. By way of illustration and not limitation, RAM can take various forms, such as static random access memory (SRAM) or dynamic random access memory (DRAM).

[0117] The technical features of the above embodiments can be combined arbitrarily. In order to make the description concise, not all possible combinations of the technical features in the above embodiments are described. However, as long as there is no contradiction in the combination of these technical features, they should be considered to be within the scope of the present invention.

Claims

1. A control method for a quasi-Z-source inverter, characterized in that: include: Obtaining an inductor current sampling value of the quasi-Z-source inverter, and determining, based on an error between the inductor current sampling value and an inductor current reference value, an action order of applying a through-voltage vector and a non-through-voltage vector within a control period of the quasi-Z-source inverter and a duty cycle of applying the through-voltage vector; applying a plurality of reference non-through voltage vectors to the quasi-Z-source inverter respectively; For any reference non-through voltage vector, obtaining an initial inductor current sampling value, an initial capacitor voltage sampling value, and an initial output current sampling value of the quasi-Z-source inverter under the reference non-through voltage vector; Determining a predicted capacitor voltage value at a future time according to the initial inductor current sampling value and the initial capacitor voltage sampling value, and determining a predicted output current value at a future time according to the initial output current sampling value; quantifying the difference between the output current prediction value and the output current reference value under each reference non-through voltage vector, and selecting multiple candidate non-through voltage vectors from the multiple reference non-through voltage vectors according to the quantification results; Quantify the difference between the capacitor voltage prediction value and the capacitor voltage reference value under each candidate non-through voltage vector to determine the target non-through voltage vector; The quasi-Z-source inverter is controlled by applying a through voltage vector and a non-through voltage vector in a control period of the quasi-Z-source inverter, a duty cycle of applying the through voltage vector, and the target non-through voltage vector.

2. The method according to claim 1, characterized in that Determining, according to an error between the inductor current sampling value and the inductor current reference value, an action order of applying a through-voltage vector and a non-through-voltage vector within a control period of the quasi-Z-source inverter, including: When the error is less than or equal to 0, determining to first apply a through voltage vector and then apply a non-through voltage vector within a control period of the quasi-Z-source inverter; When the error is greater than 0, it is determined that a non-through voltage vector is applied first and then a through voltage vector is applied within a control period of the quasi-Z-source inverter.

3. The method according to claim 2, characterized in that Determining a duty cycle of applying a through voltage vector within a control period of the quasi-Z-source inverter according to an error between the inductor current sampling value and the inductor current reference value includes: determining an action time of a through voltage vector within the control period according to an error between the inductor current sampling value and the inductor current reference value; The duty cycle of the through-voltage vector applied within the control period of the quasi-Z-source inverter is determined according to the action time of the through-voltage vector within the control period.

4. The method according to claim 3, characterized in that The step of determining the action time of the through voltage vector within the control period according to the error between the inductor current sampling value and the inductor current reference value includes: When the error is less than or equal to 0, the calculation method for determining the action time of the through voltage vector within the control period is: ； When the error is greater than 0, the calculation method for determining the action time of the through voltage vector within the control period is: ； in, is the action time of the through voltage vector within the control period, is the voltage gain of the quasi-Z-source inverter, is the control cycle length of the quasi-Z source inverter, is the inductor, is the DC supply voltage of the Z-source inverter, is the error between the inductor current sampling value and the inductor current reference value.

5. The method according to claim 1, characterized in that The step of quantifying the difference between the output current prediction value and the output current reference value under each reference non-shoot-through voltage vector comprises: For any benchmark non-through voltage vector, the gap between the output current prediction value and the output current reference value is quantified by the output current cost function to obtain the output current cost function value; The output current cost function is: ； in, is the output current cost function value, They are Coordinate axes and Output current reference value on the coordinate axis, They are time Coordinate axes and The output current prediction value on the coordinate axis, The moment is the future moment, is the initial moment, The control cycle duration.

6. The method according to claim 5, characterized in that The quantified result is the output current cost function value; The step of selecting a plurality of candidate non-through voltage vectors from the plurality of reference non-through voltage vectors according to the quantization result includes: The reference non-indirect voltage vectors corresponding to a preset number of minimum output current cost function values ​​are determined as candidate non-indirect voltage vectors.

7. The method according to claim 1, characterized in that The step of quantifying the difference between the capacitor voltage prediction value and the capacitor voltage reference value under each candidate non-through voltage vector to determine the target non-through voltage vector includes: For any candidate non-through voltage vector, the gap between the capacitor voltage prediction value and the capacitor voltage reference value is quantified using the capacitor voltage cost function to obtain a capacitor voltage cost function value; Determine the candidate non-through voltage vector corresponding to the minimum capacitor voltage cost function value as the target non-through voltage vector; The capacitor voltage cost function is: ； in, is the capacitor voltage cost function value, is the capacitor voltage reference value, for The predicted value of the capacitor voltage at time , The moment is the future moment, is the initial moment, The control cycle duration.

8. A control device for a quasi-Z-source inverter, characterized in that: include: an acquisition module, configured to acquire an inductor current sampling value of the quasi-Z-source inverter, and determine, based on an error between the inductor current sampling value and an inductor current reference value, an action sequence of applying a through-voltage vector and a non-through-voltage vector within a control period of the quasi-Z-source inverter and a duty cycle of applying the through-voltage vector; an applying module, configured to apply a plurality of reference non-through voltage vectors to the quasi-Z-source inverter respectively; A prediction module, configured to obtain, for any reference non-through voltage vector, an initial inductor current sampling value, an initial capacitor voltage sampling value, and an initial output current sampling value of the quasi-Z-source inverter under the reference non-through voltage vector; Determining a predicted capacitor voltage value at a future time according to the initial inductor current sampling value and the initial capacitor voltage sampling value, and determining a predicted output current value at a future time according to the initial output current sampling value; a first screening module, configured to quantify the difference between the output current prediction value and the output current reference value under each reference non-through voltage vector, and screen multiple candidate non-through voltage vectors from the multiple reference non-through voltage vectors according to the quantization results; The second screening module is used to quantify the difference between the capacitor voltage prediction value and the capacitor voltage reference value under each candidate non-through voltage vector, and determine the target non-through voltage vector; A control module is used to control the quasi-Z-source inverter by applying a through voltage vector and a non-through voltage vector in a control period of the quasi-Z-source inverter, a duty cycle of applying the through voltage vector, and the target non-through voltage vector.

9. A computer-readable storage medium, characterized in that The storage medium stores a computer program, and when the computer program is executed by a processor, the method according to any one of claims 1 to 7 is implemented.

10. A computer device, characterized in that: The method comprises a memory, a processor and a computer program stored in the memory and capable of running on the processor, wherein when the processor executes the program, the method according to any one of claims 1 to 7 is implemented.

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