Integrated magnetic part suitable for low-voltage large-current DC / DC converter and design method thereof

Abstract

The invention relates to an integrated magnetic part suitable for a low-voltage large-current DC / DC converter and a design method of the integrated magnetic part. The integrated magnetic part comprises a transformer IM and a half-bridge current doubling rectifying circuit, the half-bridge current-doubling rectifying circuit comprises a half-bridge circuit and a current-doubling rectifying circuit. The invention mainly solves the problems of low conversion efficiency, more inductive components and large volume during high-step-down-ratio low-voltage large-current DC / DC (Direct Current / Direct Current) conversion. According to the low-voltage large-current power supply requirements of high-computing-power chips such as a GPU and an FPGA in an intelligent power supply system, the number of large-current windings and large-current connecting points can be effectively reduced, a plurality of discrete magnetic parts in a topological structure are integrated on one magnetic core, packaging and circuit board design are simplified, and the size can be reduced by 30% compared with that of a traditional circuit architecture. Meanwhile, by effectively transforming and utilizing the space magnetic field of a plurality of inductors, not only is the winding alternating current loss generated by the excitation current reduced, but also the magnetic core loss and the switching loss are reduced, and the power supply conversion efficiency is improved by more than 5%.

Description

TECHNICAL FIELD

[0001] The present application relates to the technical field of integrated magnetic components of power electronic converters, and particularly relates to an integrated magnetic component suitable for a low-voltage and high-current DC / DC converter and a design method thereof. BACKGROUND

[0002] The magnetic integration technology refers to winding two or more separate magnetic components (such as inductors, transformers, etc.) on a magnetic core, and realizing the integration of multiple magnetic components by reasonably designing the structure of the magnetic core and the layout of the winding. With the wide application of power electronic devices in the fields of new energy, electric vehicles, communication, etc., the demand for high-efficiency and compact magnetic components increases. Therefore, integration is the trend of the development of power electronic technology, and the magnetic integration technology is an important part of the research content of passive power electronic integration. The separate magnetic components such as filter inductors, power inductors and transformers in traditional switching power supplies usually account for more than 30% of the total volume of the power supply and have high loss, which restricts the improvement of power density and high-power conversion efficiency. Through the magnetic integration technology, multiple magnetic components can be integrated, which can significantly reduce the overall volume of the power supply, reduce the connection loss between components, reduce the material and assembly cost, and improve the electrical and thermal performance.

[0003] In a switching power supply, the magnetic components such as transformers, energy storage inductors and filter inductors work according to a certain switching time sequence, and the current and magnetic flux of these magnetic components have a certain phase relationship. By analyzing the magnetic flux distribution characteristics of each magnetic component and the magnetic flux coupling relationship between each winding, a simple magnetic circuit structure is adopted to construct a complex multi-magnetic circuit coupled inductor, and various functional magnetic components are integrated on a simple magnetic core structure, so as to achieve the purposes of reducing the number of magnetic components, reducing the size, reducing the weight, improving the power density, improving the electrical performance and reducing the cost. Therefore, it is of great practical significance to study a simple and accurate integrated magnetic component design method.

[0004] For the demand of low-voltage and high-current, the prior art scheme adopts a half-bridge combined with a current doubling rectifier circuit topology as shown in Figure 1 As shown in Figure 1 , the primary side of the converter is composed of two complementary conducting power tubes and an energy storage voltage dividing capacitor; the secondary side is composed of a synchronous rectifier tube, an output filter inductor and an output filter capacitor. The magnetic components in the scheme are all separate devices, which are composed of a transformer and two output filter inductors. However, the magnetic components in the prior art scheme include a power transformer and two large-current energy storage filter inductors, which have many magnetic components, large size and large weight. When applied in products, they will occupy a large amount of layout and wiring space, resulting in very limited available layout space. At the same time, the welding interconnection stress and loss of large-current leads are large, which limits the realization of high efficiency and high power density of the product.

[0005] Therefore, it is urgent to realize an integrated magnetic component suitable for a low-voltage large-current DC / DC converter and a design method thereof, so as to reduce the number and size of magnetic components as much as possible, reduce the cost and loss of magnetic components, and improve the efficiency and power density of the power converter under the premise of ensuring efficiency. SUMMARY

[0006] In order to solve the problems of large number, large size, large weight, large stress and large loss of discrete magnetic components in low-voltage large-current power supply application scenarios, the application provides an integrated magnetic component suitable for a low-voltage large-current DC / DC converter and a design method thereof.

[0007] To achieve the above object, the application adopts the following technical scheme:

[0008] In a first aspect of the application, an integrated magnetic component suitable for a low-voltage large-current DC / DC converter is disclosed. The integrated magnetic component comprises a transformer IM and a half-bridge current doubling rectifier circuit; the half-bridge current doubling rectifier circuit comprises a half-bridge circuit located at the primary side of the transformer IM and a current doubling rectifier circuit located at the secondary side of the transformer IM.

[0009] The transformer IM comprises a first E-shaped magnetic core and a second E-shaped magnetic core which are integrally formed and symmetrically arranged; the first E-shaped magnetic core and the second E-shaped magnetic core enclose a rectangle; the first E-shaped magnetic core and the second E-shaped magnetic core each comprise a connecting column and a first side column, a middle column and a second side column which are sequentially and spacedly arranged on the inner side of the connecting column from top to bottom; the first side column of the first E-shaped magnetic core is wound with a primary winding N p1 , and the second side column is wound with a primary winding N p2 ; the first side column of the second E-shaped magnetic core is wound with a secondary winding N L1 , and the second side column is wound with a secondary winding N L2 .

[0010] Further, the half-bridge circuit comprises an input voltage source V i , an energy storage voltage dividing capacitor C1, an energy storage voltage dividing capacitor C2, a power switch tube S1 and a power switch tube S2; the positive electrode of the input voltage source V i is connected with the first end of the energy storage voltage dividing capacitor C1, the second end of the energy storage voltage dividing capacitor C1 is connected with the first end of the energy storage voltage dividing capacitor C2, and the second end of the energy storage voltage dividing capacitor C2 is connected with the negative electrode of the input voltage source V i ; the drain electrode of the power switch tube S1 is connected with the first end of the energy storage voltage dividing capacitor C1, the source electrode of the power switch tube S1 is connected with the drain electrode of the power switch tube S2; the source electrode of the power switch tube S2 is connected with the second end of the energy storage voltage dividing capacitor C2; the first end of the primary winding N p1 is connected with the source electrode of the power switch tube S1, and the primary winding N p1The second end is connected to the primary winding N p2 The second end is connected, and the primary winding N p2 The first end is connected to the node between the energy storage voltage divider capacitor C1 and the energy storage voltage divider capacitor C2.

[0011] Furthermore, the current-doubling rectifier circuit includes: synchronous rectifier diode SR1, synchronous rectifier diode SR2, and output filter capacitor C. The drain of synchronous rectifier diode SR1 is connected to the secondary winding N. L1 The first end is connected, and the secondary winding N L1 The second end and the secondary winding N L2 The second terminal is connected to the second terminal of the output filter capacitor C, and the secondary winding N L2 The first terminal of the output filter capacitor C is connected to the drain of the synchronous rectifier SR2, the source of the synchronous rectifier SR2 is connected to the source of the synchronous rectifier SR1, and the first terminal of the output filter capacitor C is connected to the source of the synchronous rectifier SR1.

[0012] In a second aspect of the invention, a design method for integrated magnetic components suitable for low-voltage, high-current DC / DC converters is disclosed, the method comprising the following steps:

[0013] S1. Determine the core shape and circuit topology of the integrated magnetic component, and establish the functional relationship between the maximum AC magnetic flux of each magnetic column of the integrated magnetic component core and the input voltage.

[0014] S2. Determine the DC flux of each magnetic column in the integrated magnetic component and establish the functional relationship between DC flux and magnetic reluctance;

[0015] S3. Based on the properties of the magnetic core material, determine the minimum cross-sectional area and actual cross-sectional area of ​​each magnetic column in the integrated magnetic core.

[0016] S4. Obtain the maximum magnetic flux density limit of each magnetic column in the integrated magnetic core, calculate the maximum DC magnetic flux density and magnetic reluctance of each magnetic column, and determine the length of the air gap in the magnetic column containing the air gap.

[0017] S5. Based on the actual cross-sectional area of ​​each magnetic column in the integrated magnetic core and the length of the air gap in the magnetic column containing the air gap, determine the core dimensions and winding structure of the integrated magnetic component.

[0018] Furthermore, step S1 specifically includes the following steps:

[0019] S11. Determine the core shape and circuit topology of the integrated magnetic components according to the technical specifications.

[0020] S12. Perform AC flux analysis based on circuit topology, and calculate the AC flux change ΔΦ of each magnetic column in the core under different operating modes. ac ;

[0021] The change in alternating magnetic flux ΔΦ ac The calculation formula is as follows:

[0022]

[0023] Where N represents the number of coil turns, V represents the voltage across the coil, D represents the duty cycle, and T represents the converter's operating cycle;

[0024] S13. Based on the principle of magnetic flux continuity, the changes in AC magnetic flux under each mode are superimposed to determine the input voltage V. in and output voltage V out The relationship with the turns ratio n;

[0025] S14. Use the following formula to establish the maximum AC magnetic flux Φ of each magnetic column in the integrated magnetic core. ac_max With input voltage V in Functional relationship:

[0026]

[0027] Where, Φ ac_max V represents the maximum alternating magnetic flux; in Indicates input voltage; N L Φ represents the number of turns of the secondary coil of the magnetic core, Vo represents the voltage across the secondary coil of the magnetic core, D represents the duty cycle, and T represents the operating period of the converter; ac_max (V in The expression represents the functional relationship between the maximum AC magnetic flux and the input voltage.

[0028] Furthermore, step S2 specifically includes the following steps:

[0029] S21. Based on the technical specifications, calculate the DC current I of each winding of the integrated magnetic component. dc and ampere-turns NI;

[0030] S22. Transform the magnetic circuit of the integrated magnetic component into an equivalent magnetoresistive network to generate a DC equivalent magnetic circuit diagram;

[0031] S23. Based on Ampere's circuital law and combined with the DC equivalent magnetic circuit diagram, the DC magnetic flux Φ of each magnetic column is determined using formulas (4) and (5). dc and establish DC magnetic flux Φ dc With magnetic reluctance R m The functional relationship Φ dc (R m );

[0032] ∑i·N=∑U m =∑H·l e (4);

[0033]

[0034] wherein i represents the current flowing through the coil; U m represents the magnetomotive force, determined by the current I dc flowing through the coil and the number of turns N; H represents the magnetic field strength, l e represents the length of the closed integration path.

[0035] Further, the step S3 specifically comprises the following steps:

[0036] S31, according to the characteristics of the magnetic core material, setting the maximum variation of the alternating magnetic flux density of each magnetic column as ΔB max , which is determined by the relationship B = Φ / A e between the magnetic flux density B and the magnetic flux Φ, the cross-sectional area A e ; max ΔB ac = ΔΦ e_min / A

[0037] S32, calculating the minimum cross-sectional area A e_min of each magnetic column of the integrated magnetic component by using formula (6):

[0038]

[0039] wherein ΔΦ ac (V in_max ) represents the maximum variation of the alternating magnetic flux in the magnetic column when the input voltage is the highest;

[0040] S33, determining the actual cross-sectional area A e_real of each magnetic column of the integrated magnetic component by using formula (7):

[0041] A e_real ≥ A e_min (7);

[0042] wherein A e_min represents the minimum cross-sectional area of each magnetic column in the magnetic core of the integrated magnetic component, and A e_real represents the actual cross-sectional area of each magnetic column.

[0043] Further, the step S4 specifically comprises the following steps:

[0044] S41, obtaining the maximum magnetic flux density limit value B max of each magnetic column in the magnetic core of the integrated magnetic component by using formula (8), and calculating the maximum direct-current magnetic flux density B dc_max of each magnetic column in the magnetic core of the integrated magnetic component based on the maximum magnetic flux density limit value B max by using formula (9):

[0045]

[0046] wherein B total is the total magnetic flux density in the magnetic circuit, ΔB is the magnetic flux density change in the magnetic circuit, B max is the maximum magnetic flux density limit of each magnetic column, B dc_max is the maximum direct current magnetic flux density in the magnetic circuit;

[0047] S42, using formula (10)-(12), obtaining the magnetic resistance R m of each magnetic column in the integrated magnetic core;

[0048] Φ dc = B dc · A e_real (10);

[0049] Φ dc_max (R m ) = B dc_max · A e_real (11);

[0050]

[0051] wherein B dc_max is the maximum direct current magnetic flux density in the magnetic circuit, Φ dc is the direct current magnetic flux of each magnetic column, A e_real is the actual cross-sectional area of each magnetic column;

[0052] S43, based on the magnetic resistance R m and the actual cross-sectional area A e_real of each magnetic column in the integrated magnetic core, estimating the length l g of the air gap in the magnetic column containing the air gap using formula (13):

[0053] l g = μ· A e_real · R m (13);

[0054] In formula (13), R m is the magnetic resistance of the magnetic column without air gap, R m ≈0; μ is the magnetic permeability.

[0055] Further, the step S5 specifically comprises the following steps:

[0056] S51, according to the actual cross-sectional area A e_real and the length l g of the air gap in the magnetic column containing the air gap of each magnetic column in the integrated magnetic core, designing the geometric dimensions of each magnetic column of the magnetic core;

[0057] S52, calculating the effective value I rmsAccording to the maximum current density J of the winding, the cross-sectional area S of the single-turn winding is determined w ;

[0058] S53, the cross-sectional area S of each turn of the winding is determined by constraining formula (14) and formula (15) w :

[0059] S w >J·I rms (14);

[0060] N·S w <A w_min (15);

[0061] Wherein, A w_min is the minimum window area of the magnetic column where the winding is located, and N is the total number of turns of the coil of the magnetic column.

[0062] Further, in the step S31, the magnetic core material characteristic is the saturation magnetic density of ferrite.

[0063] Compared with the prior art, the application has the following advantages:

[0064] The integrated magnetic component suitable for a low-voltage large-current DC / DC converter and the design method thereof disclosed by the application mainly solve the problems of low conversion efficiency, many inductive components and large volume in the conversion of a high step-down ratio low-voltage large-current DC / DC converter. In view of the low-voltage large-current power supply demand of GPU and FPGA high-performance chips in an intelligent power supply system, the magnetic integrated design method disclosed by the application is combined with a half-bridge current doubling rectifier circuit topology, the number of large-current windings and large-current connection points can be effectively reduced, a plurality of discrete magnetic components (inductors and transformers) in the topology structure are integrated into one magnetic core, the packaging and circuit board design are simplified, and the volume can be reduced by 30% compared with the traditional circuit architecture. At the same time, through effective modification and utilization of the spatial magnetic field of a plurality of inductors, not only the winding alternating current loss generated by the excitation current is reduced, but also the magnetic core loss and switching loss are reduced, so that the power conversion efficiency is improved by more than 5%. BRIEF DESCRIPTION OF DRAWINGS

[0065] Figure 1 is a half-bridge current doubling rectifier circuit principle diagram in the prior art;

[0066] Figure 2 is a circuit principle diagram of the integrated magnetic component suitable for a low-voltage large-current DC / DC converter in the application;

[0067] Figure 3 is a working waveform diagram of four time periods of the half-bridge current doubling rectifier circuit in a steady state;

[0068] Figure 4is a flow chart of the design method of the integrated magnetic component suitable for the low-voltage large-current DC / DC converter in the application;

[0069] Figure 5 is a direct-current equivalent magnetic circuit diagram of the integrated magnetic component suitable for the low-voltage large-current DC / DC converter in the application;

[0070] Figure 6 is a magnetic core structure schematic diagram of the integrated magnetic component suitable for the low-voltage large-current DC / DC converter in the application;

[0071] Figure 7 is a simulation circuit diagram of the integrated magnetic component suitable for the low-voltage large-current DC / DC converter in the application;

[0072] Figure 8 is a primary and secondary winding current waveform diagram of the integrated magnetic component suitable for the low-voltage large-current DC / DC converter in the application;

[0073] Figure 9 is a magnetic column magnetic density change waveform diagram of the integrated magnetic component suitable for the low-voltage large-current DC / DC converter in the application;

[0074] Figure 10 is a PCB winding structure diagram of the integrated magnetic component suitable for the low-voltage large-current DC / DC converter in the application;

[0075] Figure 11 is a magnetic core thermal simulation diagram of the integrated magnetic component suitable for the low-voltage large-current DC / DC converter in the application. DETAILED DESCRIPTION

[0076] The application will be further described below in combination with the drawings:

[0077] The application aims to realize an integrated magnetic component suitable for a low-voltage large-current DC / DC converter and a design method thereof, and integrates multiple discrete magnetic elements. Under the premise of ensuring efficiency, the application can not only reduce the number of magnetic elements as much as possible, reduce the cost of magnetic elements, and reduce the size of elements, and it is expected that the inductive element size is reduced by 40% to 50%; but also can reduce the loss of magnetic elements, improve the efficiency and power density of the power converter. The purpose of magnetic integration is to reduce the number of magnetic elements, reduce the cost of magnetic elements; reduce the size of magnetic elements, reduce the loss of magnetic elements, and improve the efficiency and power density of the power converter.

[0078] For the demand of low-voltage large-current, the application designs an integrated magnetic element based on a half-bridge current doubling rectifier circuit topology, which is used to replace the transformer and two output filter inductors in the prior art. The E-type magnetic core has the advantages of multiple magnetic columns, regular shape, good stability, etc., and is the preferred integrated magnetic component. Figure 2A circuit schematic of an integrated magnetic component for a low-voltage high-current DC / DC converter according to the present application is drawn based on the topology and the E-type magnetic core.

[0079] As shown in Figure 2 , the integrated magnetic component for a low-voltage high-current DC / DC converter according to the present application comprises a transformer IM and a half-bridge current-boosting rectifier circuit; the half-bridge current-boosting rectifier circuit comprises a half-bridge circuit on the primary side of the transformer IM and a current-boosting rectifier circuit on the secondary side of the transformer IM.

[0080] Specifically, the transformer IM comprises a first E-type magnetic core and a second E-type magnetic core which are integrally formed and symmetrically arranged; the first E-type magnetic core and the second E-type magnetic core enclose a rectangle; the first E-type magnetic core and the second E-type magnetic core each comprise a connecting column and a first side column, a middle column and a second side column which are sequentially and spacedly arranged on the inner side of the connecting column from top to bottom; the first side column of the first E-type magnetic core is wound with a primary winding N p1 , and the second side column is wound with a primary winding N p2 ; the first side column of the second E-type magnetic core is wound with a secondary winding N L1 , and the second side column is wound with a secondary winding N L2 .

[0081] As a further improvement of the above technical solution, the half-bridge circuit comprises an input voltage source V i , an energy storage voltage dividing capacitor C1, an energy storage voltage dividing capacitor C2, a power switch tube S1 and a power switch tube S2; the positive electrode of the input voltage source V i is connected with the first end of the energy storage voltage dividing capacitor C1, the second end of the energy storage voltage dividing capacitor C1 is connected with the first end of the energy storage voltage dividing capacitor C2, and the second end of the energy storage voltage dividing capacitor C2 is connected with the negative electrode of the input voltage source V i ; the drain electrode of the power switch tube S1 is connected with the first end of the energy storage voltage dividing capacitor C1, the source electrode of the power switch tube S1 is connected with the drain electrode of the power switch tube S2; the source electrode of the power switch tube S2 is connected with the second end of the energy storage voltage dividing capacitor C2; the first end of the primary winding N p1 is connected with the source electrode of the power switch tube S1, the second end of the primary winding N p1 is connected with the second end of the primary winding N p2 , and the first end of the primary winding N p2 is connected to a node between the energy storage voltage dividing capacitor C1 and the energy storage voltage dividing capacitor C2.

[0082] As a further improvement of the above technical solution, the current-boosting rectifier circuit comprises a synchronous rectifier SR1, a synchronous rectifier SR2 and an output filter capacitor C. The drain electrode of the synchronous rectifier SR1 is connected with the secondary winding N L1The first end is connected, and the secondary winding N L1 The second end and the secondary winding N L2 The second terminal is connected to the second terminal of the output filter capacitor C, and the secondary winding N L2 The first terminal of the output filter capacitor C is connected to the drain of the synchronous rectifier SR2, the source of the synchronous rectifier SR2 is connected to the source of the synchronous rectifier SR1, and the first terminal of the output filter capacitor C is connected to the source of the synchronous rectifier SR1.

[0083] like Figure 3 As shown, the steady-state operation of the half-bridge current multiplier rectifier circuit includes four time periods. The working process of the present invention during these four time periods will be described in detail below.

[0084] (1) [t0~t1] time period

[0085] Primary winding N p Voltage V on p If positive, the primary winding N p There is a positive current i p The secondary winding N flows through. L1 The voltage induced on the upper winding is positive, and the synchronous rectifier diode SR1 is turned on. The secondary winding N... L2 The induced voltage on the circuit is negative, and the synchronous rectifier diode SR2 is cut off. The secondary winding N... L1 The entire load current flows through the secondary winding N. L2 The current flowing through it is zero, that is, i1 = i o i2 = 0. Where i1 is the current flowing through the secondary winding N. L1 The current i2 is the current flowing through the secondary winding N. L2 The current, i o It is the load current.

[0086] (2) [t1~t2] time period

[0087] Primary winding N p Voltage V on p The primary winding N is zero. p Current i on p =0, the secondary side is in the freewheeling stage, synchronous rectifier tubes SR1 and SR2 are turned on, and the secondary winding N L1 Secondary winding N L2 The current flowing through it is always half of the output current, i.e., i1 = i2 = i o / 2.

[0088] (3) [t2~t3] time period

[0089] Primary winding N p Voltage V on p If negative, the primary winding N p There is a negative current ip flow through, secondary winding N L1 the induced voltage is negative, the synchronous rectifier SR1 is off. Secondary winding N L2 the induced voltage is positive, the synchronous rectifier SR2 is on. Secondary winding N L1 the current flowing through is zero, secondary winding N L2 the total load current flows through, i.e. i1=0, i2=i0.

[0090] (4) [t3-t4] period

[0091] primary winding N p the voltage V p is zero, primary winding N p the current i p =0, the secondary is in the freewheeling stage, the synchronous rectifier SR1 and the synchronous rectifier SR2 are on, secondary winding N L1 , secondary winding N L2 the currents flowing through are both half of the output current, i.e. i1=i2=i o / 2.

[0092] As shown in Figure 4 , the application also includes a design method of an integrated magnetic component suitable for a low-voltage large-current DC / DC converter, which comprises the following steps:

[0093] S1, determine the shape of the magnetic core of the integrated magnetic component and the circuit topology, and establish the functional relationship between the maximum alternating magnetic flux of each magnetic column of the magnetic core of the integrated magnetic component and the input voltage. Step S1 is the initial link of the design method of the integrated magnetic component suitable for the low-voltage large-current DC / DC converter, and the core purpose is to lay a foundation for the subsequent magnetic component design.

[0094] The step S1 specifically comprises the following steps:

[0095] S11, according to the technical index requirements, determine the shape of the magnetic core of the integrated magnetic component and the circuit topology.

[0096] According to the technical index requirements, the shape of the integrated magnetic core and the circuit topology structure are selected to ensure that the selected magnetic core can meet the performance requirements in the high-current and high-frequency working scenario, and the determined circuit topology can adapt to the low-voltage and high-current conversion scenario. According to the project technical index (input voltage, output voltage, transformer turns, switching frequency, etc.), combined with the magnetic properties (permeability, saturation magnetic induction intensity) of the magnetic core, the structural characteristics (number of magnetic columns, shape) and the heat dissipation capacity, the type of the magnetic core suitable for the integrated magnetic component is selected. In this embodiment, an E-shaped magnetic core is selected because it has multiple magnetic columns (middle column + side column), stable structure, is suitable for integrated transformer and filter inductor, and is convenient for winding layout. By selecting the appropriate shape of the magnetic core of the integrated magnetic component, the foundation is laid for subsequent magnetic flux calculation and magnetic circuit design, ensuring that the magnetic core can meet the performance requirements in the high-current and high-frequency working scenario.

[0097] In step S12, the alternating magnetic flux analysis is performed based on the circuit topology structure, and the alternating magnetic flux change amount ΔΦ of each magnetic column in the magnetic core under different working modes is calculated. ac .

[0098] Step S12 provides data support for subsequent determination of the relationship between input voltage, output voltage and turns ratio. In a switching power supply, the product of the voltage V and the duration t (volt-second product Vt) at both ends of the winding under each working mode is equal to the product of the number of turns N and the magnetic flux change amount ΔΦ, that is: V*t=N*ΔΦ. For a periodic working transformer, by analyzing the volt-second product under different modes, the alternating magnetic flux Φ of each magnetic column under each mode can be determined. ac During the conduction period of the power tube, the primary winding voltage is the input voltage V in , the conduction time is t on , and the number of turns is N p , then the alternating magnetic flux of the corresponding magnetic column is: ΔΦ ac =V in *t on / N p . Combined with the working timing of the circuit topology (such as the on / off state of the power tube in a half-bridge circuit), the alternating magnetic flux change amount ΔΦ ac under each mode is calculated.

[0099] The calculation formula of the alternating magnetic flux change amount ΔΦ ac is as follows:

[0100]

[0101] In formula (1), N represents the number of turns, V represents the voltage at both ends of the coil, D represents the duty cycle, and T represents the working period of the transformer.

[0102] In step S13, according to the magnetic flux continuity principle, the alternating magnetic flux change amount under each mode is superimposed to determine the input voltage Vin and output voltage V out The relationship between the number of turns and the turns ratio n. This relationship is a key basis for designing the number of turns in the winding of an integrated magnetic component.

[0103] The principle of magnetic flux continuity states that the sum of the total changes in AC magnetic flux of each magnetic column within one cycle is zero. The formula for the principle of magnetic flux continuity is as follows:

[0104] ∑ΔΦ ac =0 (2);

[0105] In equation (2), ΔΦ ac This represents the change in AC magnetic flux in each working mode of each magnetic column.

[0106] S14. Use the following formula to establish the maximum AC magnetic flux Φ of each magnetic column in the integrated magnetic core. ac_max With input voltage V in Functional relationship:

[0107]

[0108] In equation (3), Φ ac_max V represents the maximum alternating magnetic flux; in Indicates input voltage; N L Φ represents the number of turns of the secondary coil of the magnetic core, Vo represents the voltage across the secondary coil of the magnetic core, D represents the duty cycle, and T represents the operating period of the converter; ac_max (V in The expression represents the functional relationship between the maximum AC magnetic flux and the input voltage.

[0109] Step S14 is used to determine the magnetic flux distribution of each magnetic column in the magnetic core under different working conditions, providing a basis for the design of the magnetic core cross-sectional area and the optimization of the magnetic circuit, and ensuring that the magnetic core is unsaturated and has the lowest loss.

[0110] S2. Determine the DC flux of each magnetic column in the integrated magnetic component and establish the functional relationship between DC flux and magnetic reluctance.

[0111] Based on the technical specifications, determine the DC magnetic flux Φ of each magnetic column in the integrated magnetic core. dc and establish DC magnetic flux Φ dc With magnetic reluctance R m The functional relationship Φ dc (R m ).

[0112] Step S2 specifically includes the following steps:

[0113] S21. Based on the technical specifications, calculate the DC current I of each winding of the integrated magnetic component. dcAnd the ampere-turns NI. The technical specifications required in step S21 include output current and transformer turns.

[0114] S22. Transform the magnetic circuit of the integrated magnetic component into an equivalent magnetoresistive network to generate a DC equivalent magnetic circuit diagram.

[0115] like Figure 5 As shown, Φ 1dc Φ 0dc Φ 2dc R represents the magnitude and direction of the magnetic flux in the three magnetic pillars, respectively. m1 R m0 R m2 These represent the magnetic reluctance of the three magnetic pillars.

[0116] S23. According to Ampere's circuital law, the DC flux of each magnetic column is determined by the DC ampere-turns and magnetic reluctance of the winding. Combining the DC equivalent magnetic circuit diagram, the DC flux Φ of each magnetic column is determined using formulas (4) and (5). dc and establish DC magnetic flux Φ dc With magnetic reluctance R m The functional relationship Φ dc (R m );

[0117] ∑i·N=∑U m =∑H·l e (4);

[0118]

[0119] In equations (4) and (5), i represents the current flowing through the coil; U m The magnetomotive force is represented by the current I in the coil. dc The number of turns N determines the magnetic field strength; H represents the magnetic field strength, l e This indicates the length of the closed integral path. In a DC (static) magnetic field, the magnetic flux is constant, and there is no induced electromotive force. At this time, the magnetomotive force is provided entirely by the coil current and the number of turns.

[0120] S3. Based on the properties of the magnetic core material, determine the minimum cross-sectional area A of each magnetic column in the integrated magnetic core. e_min and actual cross-sectional area A e_real .

[0121] Step S3 specifically includes the following steps:

[0122] S31. Based on the characteristics of the magnetic core material, the maximum change in AC magnetic flux density of each magnetic column is set to ΔB. max The magnetic flux density B and magnetic flux Φ, and the cross-sectional area A e The relationship B = Φ / A e We obtain: ΔB max =ΔΦac / A e_min The magnetic core material property is the saturation magnetic flux density of ferrite. This relationship is an important basis for subsequent calculation of the minimum cross-sectional area of the magnetic column, and by specifying the maximum allowable variation of the alternating magnetic flux, combined with the alternating magnetic flux variation, the minimum cross-sectional area that meets the unsaturation requirement of the magnetic core can be derived.

[0123] S32, calculating the minimum cross-sectional area A of each magnetic column of the integrated magnetic component by formula (6) e_min :

[0124]

[0125] In formula (6), ΔΦ ac (V in_max ) represents the maximum variation of the alternating magnetic flux in the magnetic column when the input voltage is the highest.

[0126] S33, determining the actual cross-sectional area A of each magnetic column of the integrated magnetic component according to actual requirements by formula (7) e_real ;

[0127] A e_real ≥A e_min (7);

[0128] In formula (7), A e_min represents the minimum cross-sectional area of each magnetic column in the magnetic core of the integrated magnetic component, and A e_real represents the actual cross-sectional area of each magnetic column.

[0129] S4, obtaining the maximum magnetic flux density limit value of each magnetic column in the magnetic core of the integrated magnetic component, calculating the maximum direct-current magnetic flux density and magnetic resistance of each magnetic column in the magnetic core of the integrated magnetic component, and further determining the length of the air gap in the magnetic column containing the air gap.

[0130] The step S4 specifically includes the following steps:

[0131] S41, obtaining the maximum magnetic flux density limit value B max of each magnetic column in the magnetic core of the integrated magnetic component by formula (8) max , and calculating the maximum direct-current magnetic flux density B dc_max of each magnetic column in the magnetic core of the integrated magnetic component based on the maximum magnetic flux density limit value B total by formula (9) :

[0132]

[0133] In formula (8) and formula (9), B total is the total magnetic flux density in the magnetic circuit, which is obtained by adding the direct-current magnetic flux density B dc and the alternating magnetic flux density B ac , ΔB is the magnetic flux density variation in the magnetic circuit, B max is the maximum magnetic flux density limit value of each magnetic column, and Bdc_max Bmaxis the maximum DC magnetic flux density in the magnetic circuit, because the DC magnetic flux density is generated by a constant current and does not change with time, and is completely determined by B max and the maximum magnetic flux density variation ΔB max ( i.e. the peak-to-peak value of the AC magnetic flux density B ac ).

[0134] S42, using formula (10) to (12), obtaining the magnetic reluctance R m of each magnetic column in the integrated magnetic component magnetic core:

[0135] Φ dc = B dc · A e_real (10);

[0136] Φ dc_max (R m ) = B dc_max · A e_real (11);

[0137]

[0138] In formula (10) to (12), B dc_max is the maximum DC magnetic flux density in the magnetic circuit, Φ dc is the DC magnetic flux of each magnetic column, and A e_real is the actual cross-sectional area of each magnetic column.

[0139] S43, based on the magnetic reluctance R m and the actual cross-sectional area A e_real of each magnetic column in the integrated magnetic component magnetic core, further estimating the length l g of the air gap in the magnetic column containing the air gap using formula (13):

[0140] l g = μ· A e_real · R m (13);

[0141] In formula (13), R m is the magnetic column reluctance without air gap, R m ≈0; and μ is the magnetic permeability.

[0142] S5, according to the actual cross-sectional area of each magnetic column in the integrated magnetic component magnetic core and the length of the air gap in the magnetic column containing the air gap, determining the core shape size and winding structure of the integrated magnetic component.

[0143] The step S5 specifically comprises the following steps:

[0144] S51, according to the actual cross-sectional area A e_real of each magnetic column in the integrated magnetic component magnetic core and the length l g of the air gap in the magnetic column containing the air gap., the geometric size of each magnetic column of the magnetic core (such as the width of the middle column and the height of the side column) is designed to ensure that the overall size does not exceed the limit value.

[0145] S52, the effective value I of the primary and secondary winding currents of the integrated magnetic component is calculated rms , according to the maximum current density J of the winding, the cross-sectional area S of the single-turn winding is determined w .

[0146] S53, the cross-sectional area S of each turn of the winding is constrained by using formula (14) and formula (15) to determine the winding structure: w

[0147] S w >J·I rms (14);

[0148] N·S w <A w_min (15);

[0149] In formula (14) and formula (15), A w_min is the minimum window area of the magnetic column where the winding is located, and N is the total number of turns of the coil of the magnetic column.

[0150] Taking an E-shaped magnetic core as an example, the design method of the integrated magnetic component is designed by using the design method of the integrated magnetic component. The design input is as follows:

[0151] (1) Input voltage: 22V-36V;

[0152] (2) Output voltage: 1.0V;

[0153] (3) Output current: maximum output current 50A;

[0154] (4) Switching frequency: 200kHz;

[0155] (5) Maximum magnetic core size: 20x18x10mm.

[0156] Based on the above design input, the design method of the integrated magnetic component includes the following steps:

[0157] (1) Select EE type magnetic core, transformer turn ratio is 4:1, according to formula (1) and (2) to calculate the alternating current magnetic flux change of each mode of each magnetic column to obtain the input voltage V i and the relationship between the output voltage V o and the turn ratio n is as follows:

[0158]

[0159] In the formula, since the windings on the primary two magnetic columns are connected in series, N p is N​p1 and N p2 The sum; since the windings on the two magnetic pillars on the secondary side are connected in parallel, therefore, N p1 equals N p2 Unified representation as N L .

[0160] The maximum AC magnetic flux Φ of each magnetic column is obtained according to formula (3). ac_max (V in ):

[0161]

[0162] (2) According to the following Figure 5 The DC equivalent magnetic circuit diagram and formulas (4) to (5) shown establish the relationship between the DC flux and reluctance of each magnetic column. Since the air gap is set to open on the middle column and the side columns do not have air gaps, R m1 =R m2 Much smaller than R m0 This can be ignored. After simplification, the DC magnetic flux Φ of each magnetic column... dc (R m ):

[0163]

[0164] (3) The minimum cross-sectional area A of each magnetic column is calculated according to formulas (6) and (7). e_min :

[0165]

[0166]

[0167] Design the actual cross-sectional area A of each magnetic post. e_real :

[0168] A e1_real =A e2_real =3.52×10 -5 m 2 ;

[0169] A e0_real =7.49×10 -5 m 2 ;

[0170] (4) Set the maximum change in AC magnetic flux density of the side column as ΔB. 1_max =ΔB 2_max =0.2T, and the maximum change in the AC magnetic flux density of the middle column is set to ΔB. 0_max =0.075T. The maximum magnetic flux density is set to B. max =0.3T, and according to formulas (8) and (9), the maximum DC magnetic flux density is obtained as And according to the simplification result of step (2) and formulas (10)-(12), the magnetic resistance R of the middle column is calculated m0 :

[0171]

[0172] The magnetic resistance R of the magnetic column without air gap m ≈0, and the length l of the air gap in the magnetic column with air gap is calculated according to formula (13) g :

[0173] l g0 = μ0·A e0_real ·R m0 = 1.496 × 10 -4 m;

[0174] (5) According to the calculation results and the maximum magnetic core size limit, the integrated component size is 19.6 × 16.2 × 9.8 mm, the side column size is 16.2 × 2.5 × 5.4 mm, and the middle column size is 15.9 × 5.6 × 5.4 mm, and the appearance is shown in Figure 6 .

[0175] Figure 7 is the simulation circuit diagram of the integrated magnetic component suitable for the low-voltage large-current DC / DC converter in the application. The simulation circuit diagram is used for simulation analysis of the integrated magnetic component suitable for the low-voltage large-current DC / DC converter. Through the circuit diagram, the electrical performance of the integrated magnetic component in different working states can be simulated, which provides a basis for subsequent analysis of the primary and secondary winding current waveforms, the magnetic flux density change waveforms of each magnetic column, etc. It is an important tool for verifying the rationality and effectiveness of the integrated magnetic component design, which can help researchers understand the electrical behavior of the integrated magnetic component in actual work, and then evaluate whether it meets the needs of low-voltage large-current DC / DC conversion.

[0176] Figure 8 is the primary and secondary winding current waveform of the integrated magnetic component suitable for the low-voltage large-current DC / DC converter in the application. In Figure 8 , the horizontal coordinate t represents time (unit: millisecond, ms); the vertical coordinate i represents current (unit: ampere, A); the waveform corresponding to i1 is the current waveform of the secondary side coil L N1 ; the waveform corresponding to i2 is the current waveform of the secondary side coil L N2 ; the waveform corresponding to ip is the current waveform of the primary side coil L p ; the upper right corner of each waveform is a legend indicating the measured waveform. From Figure 8It can be seen that the peak-to-peak current of the two coils on the secondary side is 62A, and the phase difference is 180°; the peak-to-peak current of the primary measurement coil is 8A. This result conforms to the working principle of the half-bridge current doubling rectifier circuit. In different working periods, the two windings on the secondary side alternately bear the load current, and each bears half of the load current in the freewheeling stage. The rationality of the winding current distribution in the integrated magnetic component is verified, which shows that it can adapt to the working scene of low voltage and large current, and the current waveform meets the design expectation.

[0177] Figure 9 is the magnetic flux density variation waveform diagram of each magnetic column of the integrated magnetic component suitable for the low-voltage and large-current DC / DC converter in the application. Figure 9 In the application, the horizontal coordinate t represents time (unit: millisecond, ms); the vertical coordinate B represents magnetic induction intensity (unit: Tesla, T); the waveform diagram corresponding to B1 is the magnetic flux density waveform of the upper magnetic column; the waveform diagram corresponding to B0 is the magnetic flux density waveform of the middle magnetic column; the waveform diagram corresponding to B2 is the magnetic flux density waveform of the lower magnetic column; the upper right corner of each waveform diagram is a legend, indicating the measured waveform. It can be seen from the simulation waveform shown in the application that the magnetic flux density variation of the two side columns is less than the set maximum value 0.2T, and the magnetic flux density variation of the middle column is less than the set maximum value 0.075T. This shows that the magnetic core design of the integrated magnetic component is reasonable, the magnetic flux density variation is within a safe range, the magnetic core will not be saturated due to excessively high magnetic flux density, thereby ensuring the normal work of the magnetic core and reducing the magnetic core loss, and further verifying the accuracy of the calculation of the magnetic core cross-sectional area, magnetic flux density and other parameters in the integrated magnetic component design method. Figure 9

[0178] Figure 10 is the PCB winding structure diagram of the integrated magnetic component suitable for the low-voltage and large-current DC / DC converter in the application. Figure 10 The winding structure diagram is drawn by Altium Designer software, and the primary and secondary winding turns ratio is 4:1. According to the simulation result in the application, since a large current needs to pass through the secondary side, the planar winding structure is designed as 4 turns and 1 wire on the primary side, and 1 turn and 8 wires in parallel on the secondary side. The structure design is determined according to the working requirements and current conditions of the integrated magnetic component. The primary winding turns and wire diameter design meet the current transmission requirements of the primary side, and the secondary side adopts a multi-wire parallel mode, which effectively reduces the resistance of the winding and reduces the winding loss, and can adapt to the working conditions of the large current on the secondary side. At the same time, reasonable winding layout is also helpful to optimize the magnetic field distribution and improve the overall performance of the integrated magnetic component, which provides structural guarantee for the efficient work of the integrated magnetic component in the low-voltage and large-current scene. Figure 8

[0179] Figure 11 is the magnetic core thermal simulation diagram of the integrated magnetic component suitable for the low-voltage and large-current DC / DC converter in the application. The diagram is obtained by finite element simulation of the magnetic component by Maxwell software, in combination with the application. Figure 10 ​​The simulation results show that, compared with the traditional separated inductive element, the power density and the converter efficiency are greatly improved after the integrated magnetic element is used in the designed PCB winding structure. This shows that the integrated magnetic element can handle higher power in the same space volume, effectively solves the problem of large size and large space occupation of the traditional separated magnetic element, and improves the space utilization. The improvement of the converter efficiency is mainly due to the effective modification and utilization of the multiple inductance space magnetic fields by the integrated magnetic element, which reduces the winding alternating current loss generated by the excitation current, and also reduces the magnetic core loss and the switching loss, so that the waste in the energy conversion process is reduced, and the overall efficiency is improved.

[0180] The above-described embodiments are merely preferred embodiments of the present application, and are not intended to limit the scope of the present application. Various modifications and improvements to the technical solutions of the present application made by those of ordinary skill in the art without departing from the design spirit of the present application shall fall within the scope of protection of the present application as defined by the claims.

Claims

1. An integrated magnetic component suitable for low-voltage, high-current DC / DC converters, characterized in that, The integrated magnetic component includes: a transformer IM and a half-bridge current multiplier rectifier circuit; the half-bridge current multiplier rectifier circuit includes a half-bridge circuit located on the primary side of the transformer IM and a current multiplier rectifier circuit located on the secondary side of the transformer IM. The transformer IM includes a first E-type magnetic core and a second E-type magnetic core integrally formed and symmetrically arranged; the first E-type magnetic core and the second E-type magnetic core form a rectangle; both the first E-type magnetic core and the second E-type magnetic core include a connecting post and a first side post, a middle post, and a second side post arranged sequentially from top to bottom on the inner side of the connecting post; the first side post of the first E-type magnetic core is wound with a primary winding N. p1 The primary winding N is wound on the second side post. p2 The second type E magnetic core has a secondary winding N wound on its first side post. L1 The secondary winding N is wound on the second side post. L2 .

2. The integrated magnetic component for low-voltage, high-current DC / DC converters according to claim 1, characterized in that, The half-bridge circuit includes: an input voltage source V i Energy storage voltage divider capacitor C1, energy storage voltage divider capacitor C2, power switch S1 and power switch S2; the input voltage source V i The positive terminal of the capacitor is connected to the first terminal of the energy storage voltage divider capacitor C1. The second terminal of the energy storage voltage divider capacitor C1 is connected to the first terminal of the energy storage voltage divider capacitor C2. The second terminal of the energy storage voltage divider capacitor C2 is connected to the input voltage source V. i The negative terminal of the power switch S1 is connected to the first terminal of the energy storage voltage divider capacitor C1; the source of the power switch S1 is connected to the drain of the power switch S2; the source of the power switch S2 is connected to the second terminal of the energy storage voltage divider capacitor C2; the primary winding N p1 The first terminal is connected to the source of the power switch S1, and the primary winding N p1 The second end is connected to the primary winding N p2 The second end is connected, and the primary winding N p2 The first end is connected to the node between the energy storage voltage divider capacitor C1 and the energy storage voltage divider capacitor C2.

3. The integrated magnetic component for low-voltage, high-current DC / DC converters according to claim 1, characterized in that, The current-doubling rectifier circuit includes: synchronous rectifier diode SR1, synchronous rectifier diode SR2, and output filter capacitor C; the drain of synchronous rectifier diode SR1 is connected to the secondary winding N. L1 The first end is connected, and the secondary winding N L1 The second end and the secondary winding N L2 The second terminal is connected to the second terminal of the output filter capacitor C, and the secondary winding N L2 The first terminal of the output filter capacitor C is connected to the drain of the synchronous rectifier SR2, the source of the synchronous rectifier SR2 is connected to the source of the synchronous rectifier SR1, and the first terminal of the output filter capacitor C is connected to the source of the synchronous rectifier SR1.

4. A design method for integrated magnetic components suitable for low-voltage, high-current DC / DC converters, characterized in that, The method includes the following steps: S1. Determine the core shape and circuit topology of the integrated magnetic component, and establish the functional relationship between the maximum AC magnetic flux of each magnetic column of the integrated magnetic component core and the input voltage. S2. Determine the DC flux of each magnetic column in the integrated magnetic component and establish the functional relationship between DC flux and magnetic reluctance; S3. Based on the properties of the magnetic core material, determine the minimum cross-sectional area and actual cross-sectional area of ​​each magnetic column in the integrated magnetic core. S4. Obtain the maximum magnetic flux density limit of each magnetic column in the integrated magnetic core, calculate the maximum DC magnetic flux density and magnetic reluctance of each magnetic column, and determine the length of the air gap in the magnetic column containing the air gap. S5. Based on the actual cross-sectional area of ​​each magnetic column in the integrated magnetic core and the length of the air gap in the magnetic column containing the air gap, determine the core dimensions and winding structure of the integrated magnetic component.

5. The design method according to claim 4, characterized in that, Step S1 specifically includes the following steps: S11. Determine the core shape and circuit topology of the integrated magnetic components according to the technical specifications. S12. Perform AC flux analysis based on circuit topology, and calculate the AC flux change ΔΦ of each magnetic column in the core under different operating modes. ac ; The change in alternating magnetic flux ΔΦ ac The calculation formula is as follows: Where N represents the number of coil turns, V represents the voltage across the coil, D represents the duty cycle, and T represents the converter's operating cycle; S13. Based on the principle of magnetic flux continuity, the changes in AC magnetic flux under each mode are superimposed to determine the input voltage V. in and output voltage V out The relationship with the turns ratio n; S14. Use the following formula to establish the maximum AC magnetic flux Φ of each magnetic column in the integrated magnetic core. ac_max With input voltage V in Functional relationship: Where, Φ ac_max V represents the maximum alternating magnetic flux; in Indicates input voltage; N L Φ represents the number of turns of the secondary coil of the magnetic core, Vo represents the voltage across the secondary coil of the magnetic core, D represents the duty cycle, and T represents the operating period of the converter; ac_max (V in The expression represents the functional relationship between the maximum AC magnetic flux and the input voltage.

6. The design method according to claim 4, characterized in that, Step S2 specifically includes the following steps: S21. Based on the technical specifications, calculate the DC current I of each winding of the integrated magnetic component. dc and ampere-turns NI; S22. Transform the magnetic circuit of the integrated magnetic component into an equivalent magnetoresistive network to generate a DC equivalent magnetic circuit diagram; S23. Based on Ampere's circuital law and combined with the DC equivalent magnetic circuit diagram, the DC magnetic flux Φ of each magnetic column is determined using formulas (4) and (5). dc and establish DC magnetic flux Φ dc With magnetic reluctance R m The functional relationship Φ dc (R m ); ∑i·N=∑U m <∑H·l e (4); Where i represents the current flowing through the coil; U m The magnetomotive force is represented by the current I in the coil. dc The number of turns N determines the magnetic field strength; H represents the magnetic field strength, l e This indicates the length of the closed integration path.

7. The design method according to claim 4, characterized in that, Step S3 specifically includes the following steps: S31. Based on the characteristics of the magnetic core material, the maximum change in AC magnetic flux density of each magnetic column is set to ΔB. max The magnetic flux density B and magnetic flux Φ, and the cross-sectional area A e The relationship B = Φ / A e We obtain: ΔB max =ΔΦ ac / A e_min ; S32. Calculate the minimum cross-sectional area A of each magnetic post in the integrated magnetic component using formula (6). e_min : Where, ΔΦ ac (V in_max This represents the maximum change in AC magnetic flux in the magnetic column when the input voltage is at its highest. S33. Use equation (7) to determine the actual cross-sectional area A of each magnetic post in the integrated magnetic component. e_real ; A e_real ≥A e_min (7); Among them, A e_min A represents the minimum cross-sectional area of ​​each magnetic post in the integrated magnetic core. e_real This represents the actual cross-sectional area of ​​each magnetic column.

8. The design method according to claim 4, characterized in that, Step S4 specifically includes the following steps: S41. Use equation (8) to obtain the maximum magnetic flux density limit B of each magnetic column in the integrated magnetic core. max And based on the maximum magnetic flux density limit B max The maximum DC magnetic flux density B of each magnetic column in the integrated magnetic core is calculated using equation (9). dc_max : Among them, B total Let B be the total magnetic flux density in the magnetic circuit, and ΔB be the change in magnetic flux density in the magnetic circuit. max B represents the maximum magnetic flux density limit for each magnetic column. dc_max This is the maximum DC magnetic flux density in the magnetic circuit; S42. Using equations (10) to (12), obtain the magnetic reluctance R of each magnetic column in the integrated magnetic core. m : F dc =B dc ·A e_real (10); F dc_max (R m )=B dc_max ·A e_real (11); Among them, B dc_max Φ is the maximum DC magnetic flux density in the magnetic circuit. dc For each magnetic column, A is the DC magnetic flux. e_real The actual cross-sectional area of ​​each magnetic column; S43, Based on the magnetic reluctance R of each magnetic column in the integrated magnetic core. m and actual cross-sectional area A e_real The length l of the air gap in the magnetic column containing the air gap is estimated using equation (13). g : l g =μ·A e_real ·R m (13); In equation (13), R m For a magnetic reluctance column without an air gap, R m ≈0; μ is the magnetic permeability.

9. The design method according to claim 4, characterized in that, Step S5 specifically includes the following steps: S51, based on the actual cross-sectional area A of each magnetic column in the integrated magnetic core. e_real and the length l of the air gap in the magnetic column containing the air gap g Design the geometric dimensions of each magnetic column in the magnetic core; S52. Calculate the effective value I of the primary and secondary winding currents of the integrated magnetic component. rms And based on the maximum current density J of the winding, determine the cross-sectional area S of a single-turn winding. w ; S53. Using equations (14) and (15), the cross-sectional area S of each turn of the winding is calculated. w Apply constraints to determine the winding structure: S w >J·I rms (14); N·S w <A w_min (15); Among them, A w_min N represents the minimum window area of ​​the magnetic post containing the winding, and N is the total number of turns of the coil in the magnetic post.

10. The design method according to claim 7, characterized in that, In step S31, the magnetic core material has the characteristic of saturated magnetic flux density of ferrite.

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