Weak magnetic motor system control method, device and equipment based on embedded repeated active disturbance rejection and storage medium

Abstract

The invention discloses a weak magnetic motor system control method based on embedded repeated active disturbance rejection, and the method comprises the steps: firstly, building a quantitative model of a harmonic distortion rate and an overmodulation index through the analysis of an inverter overmodulation waveform; secondly, constructing an embedded control structure, and embedding a discrete repetitive controller into an active disturbance rejection current loop; designing repetitive controller parameters based on the error signal and the frequency domain relation between observation disturbance and actual disturbance; and repeating the controller parameter inverter so as to control the permanent magnet synchronous motor. The invention further discloses a weak magnetic motor system control device based on embedded repeated active disturbance rejection, corresponding equipment and a storage medium. According to the invention, the advantages of strong disturbance rejection of active disturbance rejection control and accurate suppression of periodic harmonic waves by repeated control are fully played through an embedded structure, and meanwhile, the problem of non-integer delay is effectively solved by a double fractional order approximation method, so that the operation stability and reliability of the permanent magnet synchronous motor in a weak magnetic region are remarkably improved.

Description

Technical Field

[0001] This application relates to the field of motor control technology, and more specifically, to a control method, apparatus, device, and storage medium for a weak field motor system based on embedded repetitive self-rejection. Background Technology

[0002] Currently, traditional field-oriented control and linear modulation techniques are insufficient to meet the requirements of motor field weakening control. In contrast, overmodulation techniques exhibit stronger dynamic performance and higher DC bus voltage utilization, but the introduction of additional harmonics during control significantly limits the reliability and stability of the motor under high-speed field weakening conditions. Repetitive control not only possesses the ability to track periodic signals with zero steady-state error but also enables control of all integer harmonics of the reference periodic signal, making it an ideal choice for motor harmonic current control strategies. To address the harmonic suppression problem, an embedded discrete repetitive controller can be introduced, utilizing an observer to extract harmonic signals and improve the performance of the repetitive controller.

[0003] However, this approach of controlling harmonics through repetition still has drawbacks. Since the repetitive controller is essentially an integrator with an integration period of N sampling cycles, its dynamic performance is poor. Furthermore, implementing repetitive control in a practical digital controller requires storing the integration results of the past N sampling cycles, placing a significant storage burden on the digital chip. Therefore, this method faces challenges in rapidly improving the repetitive control, and in obtaining suitable integer order beats for the improved period delay and phase lead compensation stages. Summary of the Invention

[0004] To address at least one defect or improvement requirement in the prior art, this invention provides a control method, apparatus, device, and storage medium for a weak magnetic motor system based on embedded repetitive self-disturbing, which will improve dynamic performance and effectively solve the frequency offset problem caused by the number of non-integer sampling points.

[0005] To achieve the above objectives, according to a first aspect of the present invention, a control method for a field-weakening motor system based on embedded repetitive active disturbance rejection is provided, the method comprising: S1: Analyze the output phase voltage waveform of the inverter during overmodulation operation in the weak magnetic region to obtain a quantitative relationship model between harmonic distortion rate and overmodulation index; S2: Based on the quantitative relationship model, the discrete repetitive controller is embedded into the active disturbance rejection current loop controller to construct an embedded discrete repetitive active disturbance rejection control structure; wherein, the current tracking error signal of the active disturbance rejection current loop controller is the input signal of the discrete repetitive controller; S3: Based on the frequency domain transfer relationship between the current tracking error signal of the active disturbance rejection current loop controller, the observed total disturbance, and the actual total disturbance of the system, determine the open-loop gain coefficient and phase compensation coefficient of the discrete repetitive controller; S4: Based on the open-loop gain coefficient and the phase compensation coefficient, a composite control voltage signal is generated using a double fractional approximation method; wherein, the double fractional approximation method is applied to the periodic delay element and the phase lead element of the discrete repetitive controller respectively; S5: Apply the composite control voltage signal to the inverter to control the permanent magnet synchronous motor, thereby effectively suppressing multiple harmonics in the weak magnetic field region.

[0006] Furthermore, the above-mentioned control method for a field-weakening motor system based on embedded repetitive self-rejection also includes: the step of obtaining a quantitative relationship model between harmonic distortion rate and overmodulation index in step S1 includes: Based on the value of the overmodulation index MI, the modulation interval is divided into the linear modulation region, the overmodulation I region, and the overmodulation II region; For overmodulation region I and overmodulation region II, the corresponding piecewise functions of the phase voltage are determined respectively; By performing Fourier analysis on the piecewise function, the fundamental amplitude and total voltage amplitude of the output voltage are determined, so as to establish the corresponding relationship curve between harmonic distortion rate and overmodulation index.

[0007] Furthermore, the above-mentioned control method for a field-weakening motor system based on embedded repetitive active disturbance rejection also includes: step S2 further includes: Based on the observer of the active disturbance rejection current loop controller, the current tracking error signal of the active disturbance rejection current loop controller is determined, and the current tracking error signal is input into the repetitive controller; The output signal of the repetitive controller is added as an additional compensation signal to the control signal of the self-disruption current loop controller.

[0008] Furthermore, the above-mentioned control method for a field-weakening motor system based on embedded repetitive active disturbance rejection also includes: in step S3: Based on the embedded discrete repetitive active disturbance rejection control structure, the suppression equation for the harmonics is determined; Based on the frequency domain transfer relationship between the current tracking error signal of the self-disturbance rejection current loop controller, the observed total disturbance, and the actual total disturbance of the system, the harmonic suppression equation is solved to obtain the open-loop gain coefficient and the phase compensation coefficient. The suppression equation is an equation established based on the frequency domain transfer relationship, which relates to a specific harmonic angular frequency, the observer bandwidth, and the system sampling angular frequency.

[0009] Furthermore, the above-mentioned control method for a field-weakening motor system based on embedded repetitive self-rejection also includes: in step S4, the double fractional-order approximation method specifically includes: The parameters of the periodic delay element are decomposed into integer and fractional parts, and the fractional part is approximated by Lagrange interpolation under the first interpolation order condition. The fractional part of the parameters of the phase-leading element is approximated using Lagrange interpolation under the second interpolation order condition.

[0010] Furthermore, the above-mentioned control method for a field weakening motor system based on embedded repetitive self-rejection also includes: the harmonic currents of the field weakening motor system in the synchronous rotating coordinate system are all converted to the 6kth order, and the number of period delay points of the discrete repetitive controller is N / 6. Wherein, k is a positive integer, and N is the sampling period of the repetitive controller.

[0011] Furthermore, the above-mentioned control method for a field weakening motor system based on embedded repetitive active disturbance rejection also includes: the field weakening motor system is applied to at least one of an electric vehicle drive system, a CNC machine tool spindle drive system, or a wind power converter system.

[0012] According to a second aspect of the present invention, a control device for a field-weakening motor system based on embedded repetitive self-rejection is also provided, comprising: The quantitative relationship model determination module is configured to analyze the output phase voltage waveform of the inverter during overmodulation operation in the weak magnetic region to obtain a quantitative relationship model between harmonic distortion rate and overmodulation index. An embedded control structure construction module is configured to embed a discrete repetitive controller into an active disturbance rejection current loop controller according to the quantitative relationship model, thereby constructing an embedded discrete repetitive active disturbance rejection control structure; wherein, the current tracking error signal of the active disturbance rejection current loop controller is the input signal of the discrete repetitive controller; The parameter determination module is configured to determine the open-loop gain coefficient and phase compensation coefficient of the discrete repetitive controller based on the frequency domain transfer relationship between the current tracking error signal of the active disturbance rejection current loop controller, the observed total disturbance, and the actual total disturbance of the system. The control signal generation module is configured to generate a composite control voltage signal based on the open-loop gain coefficient and the phase compensation coefficient using a double fractional approximation method; wherein the double fractional approximation method is applied to the periodic delay element and the phase lead element of the discrete repetitive controller respectively. A composite control module is configured to apply the composite control voltage signal to the inverter to control the permanent magnet synchronous motor, thereby effectively suppressing multiple harmonics in the field weakening region.

[0013] According to a third aspect of the present invention, a control device for a weak magnetic motor system based on embedded repetitive self-disruption is also provided, which includes at least one processing unit and at least one storage unit, wherein the storage unit stores a computer program, and when the computer program is executed by the processing unit, the processing unit performs the steps of any of the methods described above.

[0014] According to a fourth aspect of the invention, a storage medium is also provided that stores a computer program executable by an access authentication device, which, when run on the access authentication device, causes the access authentication device to perform the steps of any of the methods described above.

[0015] According to a fifth aspect of the present invention, a computer program product is also provided, comprising a computer program / instructions, characterized in that, when executed by a processor, the computer program / instructions implement the steps of any of the methods described above.

[0016] In summary, compared with the prior art, the above-described technical solutions conceived by this invention can achieve the following beneficial effects: This invention provides a control method for a field-weakening motor system based on embedded repetitive active disturbance rejection (ADDPR). To address the harmonic problem introduced by overmodulation, a discrete repetitive controller is introduced into the ADDPR control current loop to construct an embedded discrete repetitive ADDPR structure. A linear extended state observer is used to acquire harmonic signals and send them to the repetitive controller, resulting in high coupling between the ADDPR and the repetitive controller. To address the problems of high order and non-integer period delay in traditional repetitive controllers, this invention proposes a double fractional-order fast embedded repetitive control based on Lagrange interpolation, improving dynamic performance and resolving the frequency offset problem caused by the number of non-integer sampling points. Attached Figure Description

[0017] To more clearly illustrate the technical solutions in the embodiments of this application, the accompanying drawings used in the embodiments will be briefly introduced below. Obviously, the drawings described below are only some embodiments of this application. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0018] Figure 1 This is a schematic diagram of the spatial voltage vector trajectory and voltage waveform of the linear modulation region in an embodiment of this application; Figure 2 This is a schematic diagram of the spatial voltage vector trajectory and voltage waveform of the overmodulation region 1 in this application embodiment; Figure 3 This is a schematic diagram of the spatial voltage vector trajectory and voltage waveform of the overmodulation region 2 in this application embodiment; Figure 4This is a schematic diagram illustrating the relationship between harmonic distortion rate (THD) and overmodulation index (MI) in an embodiment of this application. Figure 5 Bode plot of the discrete repetitive controller DTRC in the embodiments of this application; Figure 6 The difference between the current command and the feedback in the embodiments of this application. e 1c With total system disturbance f c The Bode plot corresponding to the transfer function; Figure 7 This is a block diagram of the inner current control based on double fractional-order fast repetitive control in an embodiment of this application; Figure 8 A flowchart illustrating a control method for a field-weakening motor system based on embedded repetitive active disturbance rejection provided in an embodiment of this application; Figure 9 This diagram illustrates the relationship between the action path of the composite control voltage signal and the theoretical model in this embodiment. Detailed Implementation

[0019] To make the objectives, technical solutions, and advantages of this invention clearer, the invention will be further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative and not intended to limit the invention. Furthermore, the technical features involved in the various embodiments of this invention described below can be combined with each other as long as they do not conflict with each other.

[0020] The terms "first," "second," "third," etc., in the specification, claims, and accompanying drawings of this application are used to distinguish different objects, not to describe a specific order. Furthermore, the terms "comprising" and "having," and any variations thereof, are intended to cover non-exclusive inclusion. For example, a process, method, system, product, or apparatus that includes a series of steps or units is not limited to the listed steps or units, but may optionally include steps or units not listed, or may optionally include other steps or units inherent to these processes, methods, products, or apparatuses.

[0021] This invention proposes a control method for a field-weakening motor system based on embedded repetitive active disturbance rejection, including: analysis of overmodulation harmonic content in the field-weakening region, design of an embedded discrete repetitive active disturbance rejection current loop, parameter design of the repetitive controller, and establishment of a fast embedded repetitive control based on a double fractional order. Specifically, as shown... Figure 8 As shown, it includes the following steps: S1: Analysis of overmodulation harmonic content in the weak magnetic region, that is, analyzing the output phase voltage waveform of the inverter when it is overmodulated in the weak magnetic region, and obtaining a quantitative relationship model between harmonic distortion rate and overmodulation index.

[0022] In the medium-to-high speed range of motor field weakening operation, the output voltage is not a standard sine wave due to inverter overmodulation. As the speed increases, the voltage distortion and harmonic content also increase. This harmonic content can be analyzed.

[0023] To clearly distinguish the different stages of overmodulation control, the overmodulation index MI is defined as follows: (1) in Reference voltage vector amplitude, DC voltage The fundamental amplitude of the maximum output phase voltage.

[0024] According to equation (1), it can be determined that... MI Different values ​​of divide the entire modulation process into different stages.

[0025] Simultaneously, voltage utilization rate is introduced. m Concept: (2) in The amplitude of the fundamental voltage of the reference voltage vector line.

[0026] In this embodiment, the steps for obtaining a quantitative relationship model between harmonic distortion rate and overmodulation index include: dividing the modulation interval into a linear modulation region, an overmodulation I region, and an overmodulation II region based on the value of the overmodulation index MI; determining the piecewise functions of the corresponding phase voltages for the overmodulation I region and the overmodulation II region; and determining the fundamental amplitude and total voltage amplitude of the output voltage by performing Fourier analysis on the piecewise functions, so as to establish a corresponding relationship curve between harmonic distortion rate and overmodulation index.

[0027] Specifically, based on the overmodulation index, the modulation interval can be divided into the linear modulation region (constant torque region), overmodulation 1 region (constant power region), and overmodulation 2 region (constant voltage region).

[0028] (1) When the reference voltage vector is inside the inscribed circle, it is called the linear modulation region. When the reference voltage vector falls on the inscribed circle, the limit of the linear modulation region is reached, and the maximum undistorted output voltage is achieved. At this time, , MI= 0.9069, m =1, reference trajectory as follows Figure 1 As shown.

[0029] (2) When the reference voltage vector lies between the inscribed circle and the hexagonal trajectory, the trajectory of the reference voltage vector is constrained by the hexagonal trajectory and is no longer a standard circle. This is called overmodulation zone 1. Figure 2As shown, the blue dashed line represents the inscribed circle of the hexagonal trajectory and the corresponding maximum undistorted sine wave in the polar coordinate system. The circle slightly larger than the inscribed circle represents the reference voltage vector trajectory, while the bold colored line representing this circle constrained by the hexagon represents the actual voltage trajectory, corresponding to a slightly distorted approximate sine wave in the polar coordinate system. The angle between the actual voltage vector and each vertex of the spatial hexagon is now defined as the reference angle. This angle represents the time the actual voltage vector spends within the hexagonal boundary. During the voltage vector's rotation, constrained by the hexagonal boundary, the amplitude of the actual voltage vector is not always equal to the reference voltage vector, but their phase angle remains consistent. At each vertex, the two angles to the left and right... Within the specified interval, the actual voltage vector trajectory coincides with the reference voltage vector trajectory, and The maximum angle is ,when = When , it represents the limit of the linear modulation region. When =0, the limit of the overmodulation region 1 is reached. At this point, it can be obtained through area equivalence. , MI= 0.9514, m= 1.05.

[0030] when At this time, it is in the linear modulation region and no harmonics are generated, but when When harmonics exist and are related to the voltage vector amplitude, the piecewise function expression of the phase voltage is obtained as follows: (3) in , The angular velocity is the reference voltage vector of the motor.

[0031] Considering symmetry, the harmonics in the output voltage can be obtained through Fourier analysis in the first quadrant: (4) in n For harmonic orders, solve for the fundamental frequency. n Taking 1, substituting equation (3) into equation (4) and solving, we obtain the fundamental amplitude of the output voltage under over-modulation region 1: (5) (3) When the overmodulation zone 1 reaches its limit, the voltage utilization rate is increased to 1.05 times that of linear modulation. However, it is still desirable to further improve the voltage utilization rate. As the reference voltage vector rotates, we can choose to continue controlling the time of the basic voltage vector so that it remains at the vertex of the hexagonal trajectory for a long time. This is the proposed holding angle. The concept states that when the reference voltage vector is within the holding angle range on either side of the hexagon's vertex, the actual voltage reference vector remains at the position of the basic voltage vector without rotation. When the reference voltage vector exceeds the holding angle range, the actual voltage vector begins to rotate at a higher angular frequency than the reference voltage vector, and the two voltage vectors coincide when they reach the hexagon's vertex in each cycle. At this point, neither the amplitude nor the phase angle can remain consistent. Within a sector, one can obtain... .when At this time, the inverter output voltage trajectory cycles through the six basic voltage vectors. The inverter operates in a square wave control state, known as the "six-step" working mode, achieving maximum output voltage, deepest overmodulation, and maximum waveform distortion. , MI= 1, m= 1.1027. Maintaining Angle The following is a schematic diagram of the trajectory. Figure 3 As shown.

[0032] The piecewise function expression for the phase voltage is obtained as follows: (6) Fourier analysis was performed to obtain the fundamental amplitude of the overmodulated region 2 output voltage: (7) The harmonic distortion rate (THD) is defined as: (8) F The magnitude of the output voltage. F 0 represents the fundamental amplitude of the output voltage. Based on the area equivalence principle, the output voltage amplitudes of overmodulation region 1 and overmodulation region 2 can be obtained as follows: (9) (10) Equation (8) gives the THD and overmodulation index. MI Relationship such as Figure 4 As shown.

[0033] More precisely, step S1 in this embodiment analyzes the actual phase voltage waveform output by the inverter when it operates in different overmodulation regions. f(θ) How does an ideal sine wave become distorted? In a motor control system, the voltage waveform on the motor windings is generated by the upstream inverter (usually a three-phase full-bridge circuit). The inverter operates according to the switching signals generated by the control algorithm, changing the DC bus voltage (… U dcThe inverter converts the DC voltage into a three-phase AC drive motor. During high-speed, field-weakening operation of the motor, to fully utilize the DC bus voltage, the inverter needs to enter an "overmodulation" state. This means that the voltage vector amplitude required by the controller exceeds the maximum value that the inverter can linearly output (i.e., the radius of the inscribed circle of the hexagon). Overmodulation forces the inverter's output voltage vector trajectory to deviate from the ideal circle, confining it to the hexagonal boundary. This trajectory distortion directly affects the phase voltage waveform output by the inverter. f(θ) It is no longer a smooth sine wave, thus generating rich harmonic components. Therefore, the analysis of S1 is essentially to predict and quantify the harmonic interference that will be injected into the motor by studying the output voltage characteristics of the inverter in the overmodulation state.

[0034] S2: Design an embedded discrete repetitive active disturbance rejection current loop. Based on the quantitative relationship model, embed the discrete repetitive controller into the active disturbance rejection current loop controller to construct an embedded discrete repetitive active disturbance rejection control structure. The current tracking error signal of the active disturbance rejection current loop controller is the input signal of the discrete repetitive controller. Specifically, in this embodiment, the current tracking error signal of the active disturbance rejection current loop controller is determined based on the observer of the active disturbance rejection current loop controller, and the current tracking error signal is input into the repetitive controller. Then, the output signal of the repetitive controller is added as an additional compensation signal to the control signal of the active disturbance rejection current loop controller. The specific model design of this embodiment is as follows: (1) Design of self-disturbance rejection ADRC current loop In this embodiment, disturbances from both inside and outside the system are normalized into a total disturbance, therefore the current loop model can be written as: (11) In the formula, , , These represent the actual values ​​of the inductance along the d-axis and q-axis, respectively. , , These represent the total disturbances along the d-axis and q-axis, respectively. This is a reference value for the output voltage; To control the cycle.

[0035] Will As an extended state, and assuming Differentiable, its derivative If it is bounded, then equation (11) can be written as: (12) The linear state observer LESO for the current loop is constructed as shown in equation (13): (13) in, for i Observed values; for Observed values; This represents the error between the sampled current value and the observed value. This represents the error between the actual and observed values ​​of the current loop interference. and This is the gain for LESO.

[0036] The current tracking error signal in this embodiment e 1c (k) It does not come directly from the simple subtraction of the measured value and the command value of the current sensor; its direct source is the linear state observer, the core component of the active disturbance rejection controller.

[0037] For an active disturbance rejection current loop controller, its core component, the linear state observer, combines all the internal and external uncertainties experienced by the permanent magnet synchronous motor current loop (the controlled object) into a single "total disturbance" and estimates it. This estimated value... z 2c The disturbance is what the controller "sees," and it includes changes in motor parameters, external load disturbances, model nonlinearity, and the periodic harmonic disturbances caused by overmodulation, which this invention aims to suppress. Its goal is to approximate the real disturbance infinitely, as shown in equation (13) above.

[0038] For the actual total disturbance of the system, in this embodiment, it is the real total disturbance actually borne by the current loop of the permanent magnet synchronous motor, i.e., in the above formula. f c (k) This parameter is actually an objectively existing physical quantity that cannot be directly and precisely measured. It is a variable used in the system's mathematical model to characterize all uncertainties. It is also a comprehensive manifestation of the aforementioned disturbances (including parameter mismatch, load changes, harmonics, etc.), as shown in equation (12) above. Define the current reference value in the dq coordinate system. The self-disruption current loop control law is designed as follows: (14) in, This represents the closed-loop bandwidth of the current loop.

[0039] (2) Embedded discrete repetitive self-rejection current loop To further improve the controller's anti-disturbance capability and cope with system periodic disturbances, a discrete repetitive controller (DTRC) is introduced into the self-disturbance rejection current loop structure. This controller can simultaneously track the fundamental frequency and its integer multiples, making it ideal for suppressing motor disturbances. kSecond harmonic. The expression for DTRC is: (15) in N The order of the delay element, K rc For the system open-loop gain, K This is the phase adjustment coefficient. Q ( z ) is the stability coefficient, which is generally chosen as a constant less than 1. Q ( z When the DTRC value is 0.95, the Bode plot of DTRC is as follows. Figure 5 As shown.

[0040] For the current loop of ADRC, the difference between the command signal and the feedback signal can be used to extract the current after it is sent to the repetitive controller. However, this method cannot avoid the step interference and noise caused by sudden changes in the command signal, thus reducing the dynamic performance of DTRC. At the same time, the overly simple parallel structure results in no obvious coupling relationship between the parameters of ADRC and DTRC, often requiring judgment based on human experience.

[0041] If the low-frequency harmonics caused by sudden changes in the command current can be filtered out, the problem of sudden changes in command values ​​due to control algorithm switching can be effectively solved. To this end, the difference between the current command and the feedback can be... e 1c With total system disturbance f c The transfer function is expressed as: (16) When the observer bandwidth When the change occurs, the Bode plot of equation (16) is drawn as follows: Figure 6 As shown in the figure. It can be seen that the deviation is used. e 1c Harmonic signal extraction can filter out low-frequency harmonics while retaining mid- to high-frequency harmonics, thus eliminating the impact of command step.

[0042] Therefore, based on the designed current loop ADRC, it can be used e 1c After extracting the harmonic signal and embedding the repetitive controller, the embedded discrete repetitive active disturbance rejection structure EDTR-ADRC is obtained. The control law is then rewritten as: (17) In this embodiment, this signal selection method is designed based on the system's frequency domain characteristics. It is not simply a matter of piecing together two controllers, but rather using the aforementioned current tracking error signal. e 1c (k)The optimal coupling point for the "cooperative operation" of the control structure of EDTR-ADRC was determined.

[0043] S3: Repetitive controller parameter design, specifically, based on the frequency domain transfer relationship between the current tracking error signal and the observed total disturbance of the active disturbance rejection current loop controller and the actual total disturbance of the system, to determine the open-loop gain coefficient and phase compensation coefficient of the discrete repetitive controller. In this embodiment, it further includes the following steps: determining the harmonic suppression equation based on the embedded discrete repetitive active disturbance rejection control structure; solving the harmonic suppression equation based on the frequency domain transfer relationship between the current tracking error signal and the observed total disturbance of the active disturbance rejection current loop controller and the actual total disturbance of the system to obtain the open-loop gain coefficient and phase compensation coefficient; wherein, the suppression equation is an equation established based on the frequency domain transfer relationship, concerning a specific harmonic angular frequency, the observer bandwidth, and the system sampling angular frequency.

[0044] Specifically, the process in this embodiment is as follows. Definition: N=f s / f , N For the order of the repetitive controller, fs The system sampling frequency, f The repetitive controller base frequency is the frequency of the given input signal. In motor control, this frequency can be selected as the fundamental frequency of the motor current.

[0045] In the EDTR-ADRC control structure, the system's disturbance compensation is handled by... The output of the embedded repetitive controller is used to achieve perfect harmonic suppression. Therefore, the following equation needs to be satisfied. In this embodiment, the following equation is defined as the harmonic suppression equation: (18) Considering that the discretization period is short enough, the above equation can be analyzed in the frequency domain. By rewriting equations (12) and (13) in the frequency domain, we can obtain , and Transfer function: (19) Ideally, the estimated value z 2c Able to be exactly equal to the actual value f c (k) Referring to equation (19) above and Figure 6 This indicates that by measuring error e 1c To reflect the actual disturbance f c transfer function Ge1cfc It possesses high-pass characteristics and can effectively extract harmonic signals. Meanwhile, the observed disturbance... z 2c For actual disturbance f c of response It then exhibits low-pass characteristics.

[0046] In this embodiment, the amplitude-frequency response of the above equation (19) is: (20) (twenty one) z -1 In the discrete domain, it represents the delay per unit sampling time; in the frequency domain, it can be viewed as the delay of an angle. ,in To select the angular frequency of the harmonics, is the sampling angular frequency. When... Q ( z Once determined, the DTRC gain for the selected frequency can be determined. Q ( z When )=0.95, the gain is 20. K rc Phase adjustment section of DTRC z K In the frequency domain, it can be represented separately as the lead angle. Equation (18) can be reasonably simplified to: (twenty two) Solving the magnitude and phase parts of the above equation, we can obtain the following information regarding the open-loop gain. K rc and phase adjustment coefficient K Formula for selecting parameters: (twenty three) (twenty four) In summary, the parameters of the repetitive controller in S3 ( K rc and K The design of ) is to enable the power of at a specific frequency. e 1c The compensation signal generated by the driven repetitive controller, and z 2c Working together, they can ultimately perfectly offset the actual total disturbance of the system. f c Therefore, the analysis and design of S3 are based on the fundamental concept of active disturbance rejection control, which is to approximate the "observed disturbance" with the "actual disturbance".

[0047] S4: Establishing a fast embedded repetitive control based on a two-fractional-order method. In this embodiment, a composite control voltage signal is generated using a two-fractional-order approximation method based on the open-loop gain coefficient and phase compensation coefficient. The two-fractional-order approximation method is applied to the periodic delay element and the phase lead element of the discrete repetitive controller, respectively. Therefore, in this embodiment, the two-fractional-order approximation method specifically includes: decomposing the parameters of the periodic delay element into integer and fractional parts, approximating the fractional part using Lagrange interpolation under the first interpolation order condition; and approximating the fractional part of the parameters of the phase lead element using Lagrange interpolation under the second interpolation order condition. In a preferred embodiment, the harmonic currents of the field-weakening motor system in the synchronous rotating coordinate system are all converted to 6k orders, and the number of periodic delay points of the discrete repetitive controller is N / 6; where k is a positive integer and N is the sampling period of the repetitive controller.

[0048] The repetitive controller is equivalent to a... N An integrator with an integral period equal to its sampling period has poor dynamic performance. Furthermore, repetitive control, when implemented in a practical digital controller, must consider past data... N The integration results of each sampling period are stored, which puts a significant storage burden on the digital chip. This is because the harmonic currents of the three-phase permanent magnet synchronous motor are all converted into 6... k Therefore, the number of cycle delay points for repetitive control is... N / 6, to reduce the controller order, thereby improving dynamic performance and reducing storage pressure.

[0049] Specifically, in the three-phase stator windings (A, B, C) of the motor, due to factors such as inverter overmodulation, the generated phase currents are not ideal sine waves, but rather distorted waveforms containing abundant harmonics. These harmonics are mainly odd-order, such as the 5th, 7th, 11th, and 13th harmonics. Physically, this means that in the stationary A, B, C coordinate system, these harmonic currents intersect spatially at different frequencies. To simplify control, we use mathematical transformations (Clark transform + Park transform) to convert the stationary ABC coordinate system to a dq coordinate system that rotates synchronously with the rotor poles. The transformation rule is that in the ABC coordinate system, harmonics with a frequency of (6k±1)f1 (where k=1,2,3..., f1 is the fundamental frequency) will all appear as AC quantities with a frequency of 6k*f1 after transformation to the dq coordinate system. This means that, in the dq coordinate system, the originally dispersed harmonics of different orders (5th, 7th, 11th, 13th...) are "aggregated" and "transformed" into regular 6k harmonics (6th, 12th, 18th...). The control objective is simplified from a broad "suppress multiple harmonics" to a precise "suppress 6k harmonics".

[0050] The repetitive controller has an internal memory unit that stores the system's error information for the past complete fundamental frequency cycle. In each control cycle, it retrieves the error information from "one cycle ago," calculates it, and uses it to compensate for the current disturbance. N represents the number of control cycles corresponding to one fundamental frequency cycle. For example, if the motor's fundamental frequency is 50Hz and the control system's sampling frequency is 10kHz, then N = 10000 / 50 = 200. That is, within one fundamental frequency cycle, the controller samples and performs calculations 200 times. The cycle delay point refers to this delay "one cycle ago," which is used in digital controllers. z -N This represents a delay of N sampling points. Traditional repetitive controllers use this... z -N The process generates an integral effect on periodic errors.

[0051] As mentioned above, in the dq coordinate system, all harmonics that need to be suppressed are of the 6kth order. The fundamental frequency of these 6kth harmonics is 6. f 1, instead of f 1. In other words, the variation period of the harmonic signal is 1 / 6 of the fundamental period. Shortening the repetitive control period from point N to point N / 6 brings two direct benefits: greatly improved dynamic performance and significantly reduced storage pressure.

[0052] Therefore, in this embodiment, the transfer function corresponding to the basic fast repeating controller is defined as follows: (25) However, in many applications, without properly designing the sampling frequency beforehand, it is impossible to guarantee [the desired performance]. N / 6 is an integer, but digital controllers can only implement integers. N of z -N This leads to a shift in the ideal resonant frequency of the repetitive controller and the harmonic frequency of the actual motor current; and the phase compensation link calculated by equation (24) z K It is also difficult to obtain a positive integer. K Meeting the ideal phase compensation requirements of the system can lead to the system phase being either undercompensated or overcompensated.

[0053] To address the aforementioned problems, a fractional-order concept is introduced, and a six-fold frequency-multiplying fast repetitive control based on a fractional-order periodic delay element and a fractional-order phase-lead element is proposed. For the delay element... z -N / 6 The decimal part: (26) In the formula, Ni yes N The integer part of / 6 F = N – Ni ( F >0) is N The decimal part of / 6. Therefore Middle fraction delay part This can be expressed using the Lagrange interpolation method as follows: (27) (28) Similarly, for the phase compensation stage: (29) (30) In the formula, N 1. N 2 represents the interpolation order of the fractional delay stage and the phase compensation stage, respectively. , These are the interpolation coefficients.

[0054] The current inner loop control block diagram based on double fractional-order fast repetitive control is shown below. Figure 7 As shown, where P ( z () is the controlled object; R ( z () is the system input; E ( z () represents the difference between the input and the output; Y (z This is the system output. It can be deduced from... Y ( z )to E ( z The transfer function of ) is as follows: (31) S5: Harmonic suppression. In this embodiment, the composite control voltage signal is applied to the inverter to control the permanent magnet synchronous motor, thereby effectively suppressing multiple harmonics in the field weakening region. Preferably, the above method in this embodiment can be applied to at least one of an electric vehicle drive system, a CNC machine tool spindle drive system, or a wind power converter system.

[0055] The composite control voltage signal in step S4 does not directly act on the quantitative relationship model established in step S1. The model in step S1 is a theoretical tool for analysis, understanding, and prediction. The ultimate target of the composite control voltage signal is the inverter in the actual physical system, with the aim of suppressing harmonic currents generated in the permanent magnet synchronous motor.

[0056] To more clearly understand the complete path of the entire control flow from signal generation to the final target, Figure 9 The paper demonstrates the relationship between the action path of the composite control voltage signal S4 in a real-world electric drive system and the theoretical model of S1.

[0057] This application also provides a computer-readable storage medium having a computer program stored thereon, which, when executed by a processor, implements the steps of the above-described method. The computer-readable storage medium may include, but is not limited to, any type of disk, including floppy disks, optical disks, DVDs, CD-ROMs, microdrives, as well as magneto-optical disks, ROMs, RAMs, EPROMs, EEPROMs, DRAMs, VRAMs, flash memory devices, magnetic cards or optical cards, nanosystems (including molecular memory ICs), or any type of medium or device suitable for storing instructions and / or data.

[0058] It should be noted that, for the sake of simplicity, the foregoing method embodiments are all described as a series of actions. However, those skilled in the art should understand that this application is not limited to the described order of actions, as some steps may be performed in other orders or simultaneously according to this application. Furthermore, those skilled in the art should also understand that the embodiments described in the specification are preferred embodiments, and the actions and modules involved are not necessarily essential to this application.

[0059] In the above embodiments, the descriptions of each embodiment have different focuses. For parts not described in detail in a certain embodiment, please refer to the relevant descriptions in other embodiments.

[0060] In the several embodiments provided in this application, it should be understood that the disclosed apparatus can be implemented in other ways. For example, the apparatus embodiments described above are merely illustrative; for instance, the division of units is only a logical functional division, and in actual implementation, there may be other division methods. For example, multiple units or components may be combined or integrated into another system, or some features may be ignored or not executed. Furthermore, the coupling or direct coupling or communication connection shown or discussed may be through some service interface; the indirect coupling or communication connection between devices or units may be electrical or other forms.

[0061] The units described as separate components may or may not be physically separate. The components shown as units may or may not be physical units; that is, they may be located in one place or distributed across multiple network units. Some or all of the units can be selected to achieve the purpose of this embodiment according to actual needs.

[0062] Furthermore, the functional units in the various embodiments of this application can be integrated into one processing unit, or each unit can exist physically separately, or two or more units can be integrated into one unit. The integrated unit can be implemented in hardware or as a software functional unit.

[0063] If the integrated unit is implemented as a software functional unit and sold or used as an independent product, it can be stored in a computer-readable storage device (CMD). Based on this understanding, the technical solution of this application, in essence, or the part that contributes to the prior art, or all or part of the technical solution, can be embodied in the form of a software product. This computer software product is stored in a memory and includes several instructions to cause a computer device (which may be a personal computer, server, or network device, etc.) to execute all or part of the steps of the methods described in the various embodiments of this application. The aforementioned memory includes various media capable of storing program code, such as USB flash drives, read-only memory (ROM), random access memory (RAM), portable hard drives, magnetic disks, or optical disks.

[0064] Those skilled in the art will understand that all or part of the steps in the various methods of the above embodiments can be implemented by a program instructing related hardware. The program can be stored in a computer-readable storage medium, which may include: a flash drive, a read-only memory (ROM), a random access memory (RAM), a magnetic disk, or an optical disk, etc.

[0065] The foregoing description is merely an exemplary embodiment of this disclosure and should not be construed as limiting the scope of this disclosure. Any equivalent changes and modifications made in accordance with the teachings of this disclosure shall still fall within the scope of this disclosure. Those skilled in the art will readily conceive of embodiments of this disclosure upon considering the specification and practicing the disclosure herein. This application is intended to cover any variations, uses, or adaptations of this disclosure that follow the general principles of this disclosure and include common knowledge or customary techniques in the art not described herein. The specification and embodiments are to be considered exemplary only, and the scope and spirit of this disclosure are defined by the claims.

[0066] The technical features of the above embodiments can be combined in any way. For the sake of brevity, not all possible combinations of the technical features in the above embodiments are described. However, as long as there is no contradiction in the combination of these technical features, they should be considered to be within the scope of this specification.

[0067] Those skilled in the art will readily understand that the above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included within the scope of protection of the present invention.

Claims

1. A control method for a field-weakening motor system based on embedded repetitive self-disturbance rejection, wherein the field-weakening motor system includes an inverter and a permanent magnet synchronous motor, characterized in that, The method includes: S1: Analyze the output phase voltage waveform of the inverter during overmodulation operation in the weak magnetic region to obtain a quantitative relationship model between harmonic distortion rate and overmodulation index; S2: Based on the quantitative relationship model, the discrete repetitive controller is embedded into the active disturbance rejection current loop controller to construct an embedded discrete repetitive active disturbance rejection control structure; wherein, the current tracking error signal of the active disturbance rejection current loop controller is the input signal of the discrete repetitive controller; S3: Based on the frequency domain transfer relationship between the current tracking error signal of the active disturbance rejection current loop controller, the observed total disturbance, and the actual total disturbance of the system, determine the open-loop gain coefficient and phase compensation coefficient of the discrete repetitive controller; S4: Based on the open-loop gain coefficient and the phase compensation coefficient, a composite control voltage signal is generated using a double fractional approximation method; wherein, the double fractional approximation method is applied to the periodic delay element and the phase lead element of the discrete repetitive controller respectively; S5: Apply the composite control voltage signal to the inverter to control the permanent magnet synchronous motor, thereby effectively suppressing multiple harmonics in the weak magnetic field region.

2. The method as described in claim 1, characterized in that, Step S1, the step of obtaining a quantitative relationship model between harmonic distortion rate and overmodulation index, includes: Based on the value of the overmodulation index MI, the modulation interval is divided into the linear modulation region, the overmodulation I region, and the overmodulation II region; For overmodulation region I and overmodulation region II, the corresponding piecewise functions of the phase voltage are determined respectively; By performing Fourier analysis on the piecewise function, the fundamental amplitude and total voltage amplitude of the output voltage are determined, so as to establish the corresponding relationship curve between harmonic distortion rate and overmodulation index.

3. The method as described in claim 1, characterized in that, Step S2 further includes: Based on the observer of the active disturbance rejection current loop controller, the current tracking error signal of the active disturbance rejection current loop controller is determined, and the current tracking error signal is input into the repetitive controller; The output signal of the repetitive controller is added as an additional compensation signal to the control signal of the self-disruption current loop controller.

4. The method as described in claim 1, characterized in that, In step S3: Based on the embedded discrete repetitive active disturbance rejection control structure, the suppression equation for the harmonics is determined; Based on the frequency domain transfer relationship between the current tracking error signal of the self-disturbance rejection current loop controller, the observed total disturbance, and the actual total disturbance of the system, the harmonic suppression equation is solved to obtain the open-loop gain coefficient and the phase compensation coefficient. The suppression equation is an equation established based on the frequency domain transfer relationship, which relates to a specific harmonic angular frequency, the observer bandwidth, and the system sampling angular frequency.

5. The method as described in claim 1, characterized in that, In step S4, the double fractional approximation method specifically includes: The parameters of the periodic delay element are decomposed into integer and fractional parts, and the fractional part is approximated by Lagrange interpolation under the first interpolation order condition. The fractional part of the parameters of the phase-leading element is approximated using Lagrange interpolation under the second interpolation order condition.

6. The method as described in claim 1, characterized in that, The harmonic currents of the field weakening motor system in the synchronous rotating coordinate system are all converted to the 6kth order, and the number of period delay points of the discrete repetitive controller is N / 6. Wherein, k is a positive integer, and N is the sampling period of the repetitive controller.

7. The method according to any one of claims 1 to 7, characterized in that, The weak magnetic motor system is applied to at least one of an electric vehicle drive system, a CNC machine tool spindle drive system, or a wind power converter system.

8. A control device for a field-weakening motor system based on embedded repetitive self-disturbing rejection, characterized in that, include: The quantitative relationship model determination module is configured to analyze the output phase voltage waveform of the inverter during overmodulation operation in the weak magnetic region to obtain a quantitative relationship model between harmonic distortion rate and overmodulation index. An embedded control structure construction module is configured to embed a discrete repetitive controller into an active disturbance rejection current loop controller according to the quantitative relationship model, thereby constructing an embedded discrete repetitive active disturbance rejection control structure; wherein, the current tracking error signal of the active disturbance rejection current loop controller is the input signal of the discrete repetitive controller; The parameter determination module is configured to determine the open-loop gain coefficient and phase compensation coefficient of the discrete repetitive controller based on the frequency domain transfer relationship between the current tracking error signal of the active disturbance rejection current loop controller, the observed total disturbance, and the actual total disturbance of the system. The control signal generation module is configured to generate a composite control voltage signal based on the open-loop gain coefficient and the phase compensation coefficient using a double fractional approximation method; wherein the double fractional approximation method is applied to the periodic delay element and the phase lead element of the discrete repetitive controller respectively. A composite control module is configured to apply the composite control voltage signal to the inverter to control the permanent magnet synchronous motor, thereby effectively suppressing multiple harmonics in the field weakening region.

9. A control device for a field-weakening motor system based on embedded repetitive self-disturbance rejection, characterized in that, The method includes at least one processing unit and at least one storage unit, wherein the storage unit stores a computer program that, when executed by the processing unit, causes the processing unit to perform the steps of the method according to any one of claims 1 to 7.

10. A storage medium, characterized in that, It stores a computer program executable by an access authentication device, which, when run on the access authentication device, causes the access authentication device to perform the steps of the method according to any one of claims 1 to 7.

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