Power factor correction in power converters

Abstract

The invention relates to power factor correction in a power converter. An apparatus, such as a resonant power converter, as discussed herein may include: a first transformer winding; sensing circuitry operable to sense a first energy supplied from an input voltage to the first transformer winding; and switching circuitry operable to apply a power factor correction associated with converting an input voltage to an output voltage derived from an output of a second transformer, the second transformer winding being magnetically coupled to the first transformer winding, the applied power factor correction is operable to control a flow of first energy from the input voltage to the first transformer winding.

Description

Technical Field

[0001] This disclosure relates to power converters, and more particularly to power factor correction in power converters. Background Technology

[0002] As the name suggests, a conventional power converter converts the received input voltage into an output voltage. One type of conventional power converter receives an alternating current (AC) voltage and converts it into a corresponding DC output voltage. Summary of the Invention

[0003] Devices such as power sources or other suitable entities discussed herein can be configured to include: a first winding; a first bidirectional switching circuit device; and a second bidirectional switching circuit device connected in series with the first bidirectional switching circuit device. The combination of the first and second bidirectional switching circuit devices can be configured to control the amplitude of the current through the first winding to generate an output voltage.

[0004] As discussed further here, the device may also include a capacitor disposed in a series circuit path including the first winding. The series circuit path is operable to support resonance of current through / to the first winding. More specifically, the series circuit path including the capacitor disposed in series with the first winding may be a resonant circuit.

[0005] Furthermore, note that the first winding can be located in the transformer. The second winding of the transformer can be magnetically coupled to the first winding. In this case, the flow of the (first) current through the first winding induces a (second) current flowing through the second winding. The transformer windings support the conversion of the received input voltage into an output voltage and the output of the output voltage from the second winding.

[0006] Furthermore, as discussed herein, the first bidirectional switching circuit device is operable to block current / voltage from passing through the first bidirectional switching circuit device in both the first and second directions; the second bidirectional switching circuit device is operable to block current / voltage from passing through the second bidirectional switching circuit device in both the first and second directions.

[0007] Based on other examples discussed herein, the first and second bidirectional switching circuit devices can be implemented in any suitable manner. In one example, the first bidirectional switching circuit device may be a first GaN (Gallium Nitride) switch; and the second bidirectional switching circuit device may be a second GaN (Gallium Nitride) switch.

[0008] Furthermore, the first bidirectional switching circuit device may include a first switch connected in series with the second switch; the second bidirectional switching circuit device may include a third switch connected in series with the fourth switch; the combination of the first switch and the second switch may be set in a first series circuit path between the first node and the second node of the power converter; the combination of the third switch and the fourth switch may be set in a second series circuit path between the second node and the third node of the power converter.

[0009] Additionally, the apparatus discussed herein can be configured to include a controller operable to: during a first mode in which the input voltage between the first and third nodes is positive: i) activate both the first and third switches to the ON state, and ii) alternately switch between activating the second switch to the ON state and activating the fourth switch to the ON state, to control the magnitude of the current through the first winding. The controller can also be operable to: during a second mode in which the input voltage between the first and third nodes is negative: i) activate both the second and fourth switches to the ON state, and ii) alternately switch between activating the first switch to the ON state and activating the third switch to the ON state, to control the magnitude of the current through the first winding.

[0010] Furthermore, the device, such as the circuit arrangement discussed herein, can be configured to include a transformer comprising a first winding and a second winding. The second winding can be magnetically coupled to the first winding in the transformer. In this case, the flow of current through the first winding is operable to generate an output voltage, wherein the output voltage is output from the second winding. The controller can be configured to regulate the amplitude of the output voltage by switching between operation in a first mode and a second mode, and by implementing switching control in each of the first and second modes.

[0011] Other examples discussed herein include configurations in which the first bidirectional switching circuitry can be a first dual-gate switch comprising a first drain node, a first source node, a first gate node, and a second gate node. The second switching circuitry can be a second dual-gate switch comprising a second drain node, a second source node, a third gate node, and a fourth gate node.

[0012] Another example discussed here includes a controller. This controller can be configured to: during a first mode where the input voltage between the first and third nodes is positive: i) apply an on-control signal to the first and third gate nodes, and ii) switch between applying an on-control signal to the second and fourth gate nodes to control the magnitude of the current through the first winding. Alternatively, the controller can also be configured to: during a second mode where the input voltage between the first and third nodes is negative: i) apply an on-control signal to the second and fourth gate nodes, and ii) switch between applying an on-control signal to the first and third gate nodes to control the magnitude of the current through the first winding.

[0013] In another example, the controller is operable to adjust the magnitude of the output voltage by switching between operation in a first mode and a second mode.

[0014] Other examples discussed herein include: the device further comprising: a first capacitor connected in parallel with the combination of the first bidirectional switching circuit device and the first winding; and a second capacitor connected in parallel with the combination of the second bidirectional switching circuit device and the first winding.

[0015] The apparatus discussed herein may also include a first capacitor. The first bidirectional switching circuit may be directly connected to the second bidirectional switching circuit via a first node, and the first capacitor may be connected in series between the first node and the first winding. The apparatus may also include a controller operable to control the first and second bidirectional switching circuits based on a feedback signal generated at a second node directly connecting the first capacitor and the first winding.

[0016] Another example discussed herein includes a method comprising: controlling the operation of a first bidirectional switching circuit device; controlling the operation of a second bidirectional switching circuit device connected in series with the first bidirectional switching circuit device; and wherein the controlled operation of the first bidirectional switching circuit device and the second bidirectional switching circuit device controls the amplitude of the current through a first winding to generate an output voltage.

[0017] As previously described, the first bidirectional switching circuit device can be configured to include a first switch connected in series with a second switch; the second bidirectional switching circuit device can be configured to include a third switch connected in series with a fourth switch. A combination of the first and second switches can be configured in a first series circuit path between the first and second nodes. A combination of the third and fourth switches can be configured in a second series circuit path between the second and third nodes. In this case, the method may further include, via a controller, during a first mode in which the input voltage between the first and third nodes is positive: i) activating both the first and third switches to an ON state, and ii) alternately switching between activating the second switch to an ON state and activating the fourth switch to an ON state to control the amplitude of the current through the first winding. For example, in the first mode, when the first switch is ON, the second switch is OFF; and when the first switch is OFF, the second switch is ON.

[0018] Furthermore, examples as discussed herein include: via a controller, during a second mode in which the input voltage between the first and third nodes is negative: i) both the second and fourth switches are activated to the ON state, and ii) alternating between activating the first switch to the ON state and activating the third switch to the ON state to control the magnitude of the current through the first winding, for example, in the second mode, when the first switch is ON, the second switch is OFF; and when the first switch is OFF, the second switch is ON.

[0019] These and other more specific examples are disclosed in more detail below.

[0020] Another example discussed herein includes an apparatus comprising: a first transformer winding; sensing circuitry operable to sense a first energy supplied from an input voltage to the first transformer winding; and switching circuitry operable to apply a power factor correction associated with converting the input voltage into an output voltage derived from an output of a second transformer winding magnetically coupled to the first transformer winding, the applied power factor correction operable to control the flow of the first energy from the input voltage to the first transformer winding.

[0021] The output voltage can supply the second energy to the corresponding load. The device may also include a controller operable to: i) monitor the magnitude of the first energy via feedback from the sensing circuitry, and ii) apply a power factor correction via controlled operation of the switching circuitry, such that the average magnitude of the first energy supplied from the input voltage to the first transformer winding is substantially equal to the average magnitude of the second energy supplied from the second transformer winding to the corresponding load.

[0022] Furthermore, the sensing circuitry may include a first capacitor. The device may also include a series circuit path comprising the first capacitor coupled in series with a first transformer winding. The device may further include a controller operable to control the flow of resonant current through the series circuit path via control of the switching circuitry, the flow of the resonant current being controlled based on an applied power factor correction.

[0023] In other examples discussed here, the sensing circuitry may be configured to include a capacitor operable to sense the magnitude of a first energy supplied from an input voltage to a first transformer winding via a switching circuitry. An output voltage derived from the output of a second transformer winding may be operable to supply a second energy to a corresponding load. The device may also include a controller operable to: i) monitor feedback received from the sensing circuitry indicating the magnitude of the first energy, and ii) control the operation of the switching circuitry via power factor correction such that the average magnitude of the first energy over a plurality of control cycles is substantially equal to the average magnitude of the second energy over a plurality of control cycles.

[0024] In yet another example, the sensing circuit device as discussed herein can be configured to include a series circuit path comprising a first capacitor connected in series with a first switch and connected in parallel with a first transformer winding.

[0025] In another example, the sensing circuitry may include a sensing capacitor operable to store a voltage value that indicates the integral of the current supplied by the input voltage through the windings of a first transformer.

[0026] The apparatus discussed herein may further include: a signal generator circuit operable to generate a threshold signal based at least in part on the magnitude of the output voltage relative to a setpoint reference voltage; and a controller operable to control the switching circuitry and the flow of first energy from the input voltage to the first transformer winding based on the threshold signal to apply power factor correction.

[0027] Another device discussed here includes a controller. This controller can be configured to: control the flow of first energy received from an input voltage through magnetic coupling to a second transformer winding via control of a switching circuit device, the controlled flow of the first energy through the first transformer winding to the second transformer winding being operable to generate an output voltage based on a second energy supplied to a load from the output of the second transformer winding; receive feedback indicating the magnitude of the first energy; and adjust the operation of the switching circuit device over time, at least based on the feedback and the magnitude of the output voltage, the adjusted operation of the switching circuit being operable to adjust the magnitude of the first energy supplied to the first transformer winding.

[0028] Another example discussed herein includes a method comprising: controlling, via a switching circuit device, the flow of a first energy received from an input voltage through a first transformer winding, the first transformer winding being magnetically coupled to a second transformer winding, the controlled flow of the first energy through the first transformer winding to the second transformer winding generating an output voltage based on a second energy supplied from the output of the second transformer winding to a load; receiving feedback indicating the magnitude of the first energy; and applying a power factor correction via time-varying control of the switching circuit device, the control of the switching circuit device including: adjusting the magnitude of the first energy supplied to the first transformer winding.

[0029] Controlling the flow of the first energy may include controlling the resonant current supplied by the input voltage through the windings of the first transformer.

[0030] Furthermore, the time-varying control of the switching circuit device can be configured to substantially equalize the average amplitude of the first energy and the average amplitude of the second energy.

[0031] As discussed further herein, time-varying control of the switching circuit device may include adjusting the operation of the switching circuit device over time based on one or more of the following: i) feedback indicating the magnitude of a first energy, ii) the magnitude of the output voltage, iii) the magnitude of the input voltage, iv) the capacitance associated with the sensing circuit device that generates the feedback, and iv) controlling the switching cycle of the switching circuit device.

[0032] Furthermore, control of the switching circuit device may include: generating a threshold signal; comparing a feedback indicating the amplitude of a first energy with the threshold signal; and terminating the flow of a first current through the first transformer winding based on the comparison. The amplitude of the threshold signal may be based on the variation of the input voltage over time.

[0033] In addition, the methods discussed herein may include generating a threshold signal based on a combination of the following: i) the magnitude of the output voltage relative to a setpoint reference voltage, ii) the magnitude of the input voltage, and iii) the capacitance associated with the sensing circuitry that generates the feedback.

[0034] In another example, the threshold signal is a threshold signal TS. Generating the threshold signal TS may include setting the threshold signal TS = OSV – [(tsw1 / (K*C))*Vin] - cmp, where OSV is the offset value, tsw1 is a measure of the period of the control switching circuit device, C is the capacitance associated with the sensing circuit device that generates feedback, K is a value of the error voltage derived by comparing the amplitude of the output voltage with the setpoint reference voltage, Vin is the amplitude of the input voltage, and cmp is an optional compensation factor for the current induced by the input voltage AC line in the sensing circuit device.

[0035] Furthermore, the method discussed herein may include receiving feedback from a sensing circuit device. This feedback may be generated by the sensing circuit device based on the integral of the amplitude of a first current supplied from the input voltage through the windings of a first transformer. The first transformer may be arranged in a resonant circuit, wherein the first current is a resonant current flowing through the windings of the first transformer.

[0036] It should also be noted that while the examples discussed herein can be applied to the operation of a power converter, the concepts disclosed herein can be advantageously applied to any other suitable topology.

[0037] Furthermore, it should be noted that although each of the different features, techniques, configurations, etc., described herein may be discussed in different places within this disclosure, it is desirable that each concept may optionally be practiced independently of or in combination with each other where appropriate. Therefore, one or more of the inventions described herein can be practiced and observed in many different ways.

[0038] Furthermore, it should be noted that this preliminary discussion of the examples (Summary of the Invention) herein does not intentionally designate every example and / or incremental aspect of the novelty of this disclosure or the claimed invention(s). Rather, this brief description merely presents general examples of conventional art and corresponding points of novelty. For additional details and / or possible aspects (arrangements) of the invention(s), the reader may refer to the detailed description of the disclosure (which is an overview of the examples) and the corresponding drawings discussed further below. Attached Figure Description

[0039] Figure 1 This is an example diagram illustrating the configuration of the power converter discussed in this article.

[0040] Figure 2A These are diagrams illustrating different example implementations of the bidirectional switching circuit device discussed in this article.

[0041] Figure 2B This is a diagram illustrating an example implementation of the bidirectional switching circuit device discussed herein.

[0042] Figure 3 This is an example detailed diagram illustrating a power converter circuit as discussed in this article, which implements multiple instances of a bidirectional switching circuit device to convert an input voltage into an output voltage.

[0043] Figure 4 This is an example timing diagram illustrating the control of a corresponding bidirectional switching circuit device in a power converter that converts input voltage to output voltage, as discussed herein.

[0044] Figure 5 These are example methods for power converter circuits that operate as discussed herein, corresponding to the conversion of input voltage to output voltage. These power converter circuits include several examples of bidirectional switching circuit devices.

[0045] Figure 6 This is an example circuit diagram of a power converter as discussed in this article.

[0046] Figure 7 This is an example circuit diagram illustrating the implementation of the power converter and corresponding control as discussed in this article.

[0047] Figure 8 This is an example diagram illustrating the derivation of the control method using the generated threshold level discussed in this paper.

[0048] Figure 9 This is a timing diagram illustrating the control of the corresponding power converter over multiple AC input voltage cycles, as discussed in this article.

[0049] Figure 10 This is an example timing diagram illustrating the control of the corresponding power converter as discussed here.

[0050] Figure 11 This is an example diagram illustrating the control of the corresponding power converter as discussed here.

[0051] Figure 12 This is an example circuit diagram illustrating power flow sensing and control as discussed in this article.

[0052] Figure 13 This is an example diagram illustrating a sensing circuit as discussed herein, which is configured to monitor the magnitude of the current (first power) passing through the corresponding primary transformer winding.

[0053] Figure 14 This is an example diagram illustrating a sensing circuit as discussed herein, which is configured to monitor the magnitude of the current (first power) passing through the corresponding primary transformer winding.

[0054] Figure 15This is an example diagram illustrating a sensing circuit as discussed herein, which is configured to monitor the magnitude of the current (first power) passing through the corresponding primary transformer winding.

[0055] Figure 16 This is an example method for controlling the corresponding resonant / flyback power converter to provide power factor correction, as discussed in this paper.

[0056] The foregoing and other purposes, features, and advantages of the examples herein will become apparent from the following more specific description as illustrated in the accompanying drawings, in which the same reference characters refer to the same parts in different views. The drawings are not necessarily drawn to scale, but rather focus on illustrating examples, principles, concepts, etc. Detailed Implementation

[0057] Now, referring to the attached diagram, Figure 1 This is a diagram illustrating a bridgeless hybrid flyback power source using bidirectional circuitry, as illustrated in the examples provided herein.

[0058] like Figure 1 As shown, the power converter 100 includes a power supply 120, a bidirectional switching circuit device 131, a bidirectional switching circuit device 132, a transformer 161, capacitors C1, C2, C3, and C4, a diode D1, and a load 118.

[0059] The bidirectional switching circuit device 131 includes switches Q11 and Q12. The bidirectional switching circuit device 132 includes switches Q21 and Q22.

[0060] Transformer 161 includes a primary winding 161-1 and a secondary winding 161-2. The secondary winding 161-2 is magnetically coupled (inductively coupled) to the primary winding 161-1. The primary winding 161-1 is connected between nodes N3 and N6. The secondary winding 161-2 is connected between nodes N7 and N8.

[0061] A power supply 120, providing an input voltage Vin (such as AC voltage or input voltage 121), is connected between nodes N1 and N2. Therefore, the voltage across nodes N1 and N2 is Vin. A capacitor C1 (such as a standard EMI capacitor or electromagnetic interference capacitor) is connected in parallel with the power supply 120 between nodes N1 and N2 to filter high-frequency currents and store the small charge associated with the input voltage Vin.

[0062] Note that the bidirectional switching circuit device 131 or 132 can be configured in any suitable manner. Figure 2 shows a non-limiting example of one implementation of the bidirectional switching circuit device associated with the power converter 100. The bidirectional switching circuit device shown in Figure 2 is discussed in more detail below.

[0063] Refer again Figure 1 Note that the source node S of switch Q11 is connected to node N1. The drain node D of switch Q11 is connected to node N4 and the corresponding drain node D of switch Q12. The source node S of switch Q12 is connected to node N3 and the corresponding source node S of switch Q21. The drain node D of switch Q21 is connected to the drain node D of switch Q22. The source node S of switch Q22 is connected to node N2.

[0064] Note that the bidirectional switching circuit device discussed here can be configured in any suitable manner. For example, in Figure 2A The diagram shows a first option #1 of the bidirectional switching circuit device discussed here, such as having a common drain connection; Figure 2A The image shows a second option #2 of the bidirectional switching circuit device discussed here, such as having a common source connection; Figure 2A The BDS switch option #3 is shown in the image.

[0065] Notice, Figure 1 The diagram illustrates the implementation of the first option #1; Figure 3 The diagram illustrates the implementation of option #3. Below is... Figure 2B Another option #4 is discussed in the discussion of bidirectional switching circuit devices (such as so-called IGBTs or insulated gate bipolar transistors).

[0066] In addition, such as Figure 1 As shown, capacitor C2 is connected between node N1 and node N6. Capacitor C3 is connected between node N6 and node N2.

[0067] Each of capacitors C2 and C3 can have any suitable capacitance value.

[0068] Therefore, the combination of bidirectional switching circuit device 131 and bidirectional switching circuit device 132 is connected in series between node N1 and node N2. The series circuit path including bidirectional switching circuit device 131 and bidirectional switching circuit device 132 is arranged in parallel with power supply 120 and corresponding capacitor C1.

[0069] Also note that, Figure 1 The power converter 100 shown can be configured to include a controller 140.

[0070] In one example, controller 140 generates a control signal S11 (also known as LSN) to drive the gate node G of switch Q11. The control signal S11 is used to control switch Q11 between an on state and an off state.

[0071] The controller 140 generates a control signal S12 (also known as HSP) to drive the gate node G of the switch Q12. The control signal S12 is used to control the switch Q12 between the on and off states.

[0072] The controller 140 generates a control signal S21 (also known as HSN) to drive the gate node G of the switch Q21. The control signal S21 is used to control the switch Q21 between the on and off states.

[0073] The controller 140 generates a control signal S22 (also known as LSP) to drive the gate node G of the switch Q22. The control signal S22 is used to control the switch Q22 between the on and off states.

[0074] As shown in this example, and as discussed above, each instance of the bidirectional switching circuit device discussed here is configured to selectively block or conduct current independently of voltage direction.

[0075] For example, the bidirectional switching circuit 131 can be controlled to prevent voltage / current at node N1 from being transmitted to node N3.

[0076] The bidirectional switching circuit device 131 can be controlled to prevent voltage / current at node N3 from being transmitted to node N1.

[0077] In a similar manner, the bidirectional switching circuit 132 can be controlled to prevent current from flowing from node N2 to node N3.

[0078] The bidirectional switching circuit device 132 can be controlled to prevent current from flowing from node N3 to node N2.

[0079] Note that the flow of current 151 through the primary winding 161-1 (such as a change in amplitude) can cause a corresponding current 152 to flow through the secondary winding 161-2. The flow of current 152 through the secondary winding 161-2 across capacitor C4 and load 118 produces an output voltage 123 (also known as Vout).

[0080] It should also be noted that there are two stages: one is used only for the intake current 151, and the other is used for both the transfer currents 151 and 152.

[0081] Diode D1 prevents current 152 from flowing in the negative direction through secondary winding 161-2. In other words, secondary current 152 flows through secondary winding 161-2 in the direction from node N7 to node N8. When the diode D1 module is in the energy intake phase, diode D1 prevents current from flowing from node N7 through diode D1 to node N9.

[0082] Notice, Figure 1The power converter 100 is shown by way of non-limiting example. As discussed further below, the power converter 140 discussed herein can also be... Figure 3 This can be achieved as shown, or through other configurations.

[0083] Therefore, see again Figure 1 As discussed herein, devices such as power sources, power converters, or other suitable entities may be configured to include: a first winding 161-1; a first bidirectional switching circuit device 131; and a second bidirectional switching circuit device 132 connected in series with the first bidirectional switching circuit device 131. The combination of the first bidirectional switching circuit device 131 and the second bidirectional switching circuit device 132 may be configured to control the amplitude of the current 151 through the first winding 161-1 of the transformer 161 to generate an output voltage 123 (also referred to as Vout).

[0084] Furthermore, the second winding 161-2 of transformer 161 can be magnetically coupled to the first winding 161-1 of transformer 161. In this case, the flow of current 151 (also known as iPRI) through the first winding 161-1 is operable to convert the received input voltage Vin and corresponding current 121 into output voltage 153 (also known as Vout) via the output of output voltage 153 and the corresponding current 152 from the second winding 161-2 to the load 118.

[0085] Furthermore, as previously described, the first bidirectional switching circuit device 131 is operable to block voltage from passing through the first bidirectional switching circuit device 131 in both the first and second directions; the second bidirectional switching circuit device 132 is operable to block voltage from passing through the second bidirectional switching circuit device 132 in both the first and second directions.

[0086] In addition, the first bidirectional switching circuit device 131 may include a first switch Q11 connected in series with the second switch Q12; the second bidirectional switching circuit device 132 may include a third switch Q21 connected in series with the fourth switch Q22.

[0087] The combination of the first switch Q11 and the second switch Q12 can be set in the first series circuit path between node N1 and node N3; the combination of the third switch Q21 and the fourth switch Q22 can be set in the second series circuit path between node N3 and node N2.

[0088] In addition, such as Figure 1As shown, the example discussed here includes a power converter 100, which further includes: a capacitor C2, connected in parallel with a combination of a first bidirectional switching circuit device 131 and a first winding 161-1; and a capacitor C3, connected in parallel with a combination of a second bidirectional switching circuit device 132 and a first winding 161-1.

[0089] The first circuit path, including capacitor C2 and winding 161-1, is the first resonant circuit; the second circuit path, including capacitor C3 and winding 161-1, is the second resonant circuit. The first resonant circuit path and the second resonant circuit path are connected in parallel.

[0090] Furthermore, as previously mentioned, bidirectional switching circuit devices, such as bidirectional switching circuit device 131 and bidirectional switching circuit device 132, can be implemented in any suitable manner. An example of bidirectional switching circuit device 131 and bidirectional switching circuit device 132 is shown in Figure 2.

[0091] Figure 2B This is a diagram illustrating an example implementation of a bidirectional switch as discussed in this article.

[0092] In this example, the implementation of the bidirectional switching circuit device 131 (such as bidirectional switching circuit device 131-1) may include transistor 211, transistor 212, diode D11, and diode D12.

[0093] Transistor 211 and diode D11 can be connected to each other at node N4. A combination of transistor 211 and diode D11 can be connected in series between nodes N1 and N3.

[0094] Diode D12 and transistor 212 can be connected to each other at node N4 and are connected in series between nodes N1 and N3.

[0095] The implementation of the bidirectional switching circuit device 132 (such as bidirectional switching circuit device 132-1) may include transistor 221, transistor 222, diode D21, and diode D22. Transistor 221 and diode D21 may be connected to each other at node N5 and may be connected in series between nodes N2 and N3. Diode D22 and transistor 222 may be connected to each other at node N5 and may be connected in series between nodes N2 and N3.

[0096] Figure 3 Another example of implementing a first bidirectional switching circuit device 131 and a second bidirectional switching circuit device 132 in an AC-DC power converter is shown. Note that the output voltage 153 can be a DC voltage.

[0097] Figure 3The example power converter circuit diagram discussed in this article illustrates the implementation of multiple dual-gate bidirectional switches (such as gallium nitride (GaN) switches) in a power converter to convert the input voltage to the output voltage.

[0098] As previously stated, the first bidirectional switching circuit device 131 and the second bidirectional switching circuit device 132 can be implemented in any suitable manner.

[0099] In one example, the first bidirectional switching circuit device 131 can be implemented as a dual-gate bidirectional switching circuit device 131-2, such as a first GaN (gallium nitride) switch. The second bidirectional switching circuit device 132 can be implemented as a dual-gate bidirectional switching circuit device 132-2, such as a second GaN (gallium nitride) switch.

[0100] like Figure 3 As further shown, the first bidirectional switching circuit device 131-2 may be a first dual-gate switch including a first drain node D31, a first source node S31, a first gate node GF1 (such as a floating gate), and a second gate node G31.

[0101] The second bidirectional switching circuit device 132-1 may be a second dual-gate switch including a drain node D32, a source node S32, a gate node GF2 (e.g., a floating gate), and a gate node G32.

[0102] The power converter 100-1 also includes an EMI cancellation circuit 301, such as capacitor C31, capacitor C32 (similar to...). Figure 1 The capacitors 311, 311, and 312 are included. If needed, the controller 140 or other suitable entity provides power factor correction.

[0103] In this example, the power converter 100-1 also includes a capacitor C39, which is connected in series with the transformer winding 161-1 between nodes N3 and N6. This series circuit path can be configured to support current sensing.

[0104] As further shown, the power converter 100-1 can be configured to include a controller 140-1 (similar to controller 140 operation) to control a first bidirectional switching circuit device 131-2 and a second bidirectional switching circuit device 132-2 based on a feedback signal 399 (such as voltage) generated at node N14, node N14 being directly connected to capacitor C39 and first winding 161-1.

[0105] Note that the controller can be configured to monitor feedback from any appropriate node to control the first and second bidirectional switching circuit devices. For example, an additional example discussed herein could include monitoring node N6 or N8 to control the switching circuit devices.

[0106] Figure 4 This is an example timing diagram illustrating the control of a corresponding bidirectional switching circuit device in a power converter that converts input voltage to output voltage, as discussed herein.

[0107] refer to Figure 1 The power converter 100 and Figure 4 In the timing diagram 400, during the first mode (when the polarity of the input voltage Vin is positive), such as between time T31 and time T41, where the input voltage Vin between node N1 and node N2 is positive as detected by controller 140 or other suitable entity, controller 140: i) activates switch Q11 (via driving control signal S11 to logic high between time T31 and time T41) and switch Q21 to the ON state (via setting control signal S21 to logic high between time T31 and time T41), and ii) alternately switches between activating the second switch Q12 to the ON state and activating the fourth switch Q22 to the ON state to control the amplitude of the current 151 through the first winding 161-1.

[0108] In other words, during the first mode between time T31 and time T41, when controller 140 activates switch Q12 to the ON state, controller 140 deactivates switch Q22 to the OFF state. Conversely, when controller 140 deactivates switch Q12 to the OFF state, controller 140 activates switch Q22 to the ON state.

[0109] Note that the switching frequency at which switches Q12 and Q22 are turned on and off between time T31 and time T41 is substantially greater than the line frequency associated with the input voltage Vin (as defined by the period of the input voltage Vin between time T31 and time T51).

[0110] Between time T51 and time T61, between time T71 and time T81, etc., the controller 140 controls the operation of the corresponding bidirectional switching circuit device 131 and bidirectional switching circuit device 132 in a manner similar to that described above (first mode) during the duration between time T31 and time T41.

[0111] Between time T21 and time T31, between time T41 and time T51, between time T61 and time T71, etc., the controller 140 operates in the second mode (when the polarity of the input voltage Vin is negative).

[0112] During the second mode when the input voltage Vin between nodes N1 and N2 is negative, controller 140: i) activates bidirectional switching circuit devices Q12 and Q22 to the ON state between time T41 and time T51, and ii) switches between activating bidirectional switching circuit device Q11 to the ON state and activating bidirectional switching circuit device Q21 to the ON state to control the amplitude of the current 151 through the first winding 161-1.

[0113] In other words, during the second mode between time T41 and time T51, when controller 140 activates switch Q11 to the ON state, controller 140 deactivates switch Q21 to the OFF state. Conversely, when controller 140 deactivates switch Q11 to the OFF state, controller 140 activates switch Q21 to the ON state.

[0114] Note that the switching frequency at which switches Q11 and Q12 are turned on and off between time T41 and time T51 is substantially greater than the line frequency associated with the input voltage Vin (as defined by the period of the input voltage Vin between time T31 and time T51).

[0115] Between time T21 and time T31, between time T61 and time T71, etc., the controller 140 controls the operation of the corresponding bidirectional switching circuit device 131 and bidirectional switching circuit device 132 in a manner similar to that described above (first mode) during the duration between time T41 and time T51.

[0116] As previously described, the second winding 161-2 can be magnetically coupled to the first winding 161-1 in the transformer 161. In this case, the flow of current 151 through the first winding 161-1 is operable to induce the flow of current 152 through the secondary winding 161-2 to generate an output voltage 153, which is output from the second winding 161-2 to the load 118 and the corresponding capacitor C4. The controller 140 can be configured to adjust the amplitude of the output voltage 153 by switching between operation in a first mode and a second mode.

[0117] refer to Figure 3 The power converter 100-1 and Figure 4In the timing diagram 400, during the first mode (when the polarity of the input voltage Vin is positive), such as between time T31 and time T41, wherein the input voltage Vin between node N1 and node N2 is positive as detected by controller 140-1 or other suitable entity, controller 140-1: i) generates a control signal S11 applied to gate node GF1, making it logic high between time T31 and time T41, and generates a control signal S21 supplied to gate node G32, making it logic high between time T31 and time T41, and ii) switches between generating control signal S12 and control signal S22 between logic high and logic low states in a manner similar to that described above.

[0118] More specifically, during the first mode, when controller 140-1 generates a logic high control signal S12 applied to gate node G31, controller 140-1 generates a logic low control signal S22 applied to gate node GF2. Conversely, when controller 140-1 generates a logic low control signal S12 applied to gate node G31, controller 140-1 generates a logic high control signal S22 applied to gate node GF2.

[0119] Note that the switching frequency at which gate nodes G31 and GF2 are turned on and off between time T31 and time T41 is substantially greater than the line frequency associated with the input voltage Vin (as defined by the period of the input voltage Vin between time T31 and time T51).

[0120] Between time T51 and time T61, between time T71 and time T81, etc., the controller 140 controls the operation of the corresponding bidirectional switching circuit device 131-2 and bidirectional switching circuit device 132-2 in a manner similar to that described above (first mode) during the duration between time T31 and time T41.

[0121] Between time T21 and time T31, between time T41 and time T51, between time T61 and time T71, etc., the controller 140 operates in the second mode (when the polarity of the input voltage Vin is negative).

[0122] During the second mode (when the polarity of the input voltage Vin is negative), such as between time T41 and time T51, where the input voltage Vin between node N1 and node N2 is negative as detected by controller 140-1 or other suitable entity, controller 140-1: i) generates a control signal S22 applied to gate node GF2, making it logic high between time T41 and time T51, and generates a control signal S12 supplied to gate node G31, making it logic high between time T41 and time T51, and ii) switches between logic high and logic low states between generating control signal S11 and control signal S21.

[0123] More specifically, during the second mode, when controller 140-1 generates a logic high control signal S21 applied to gate node G32, controller 140-1 generates a logic low control signal S11 applied to gate node GF1. Conversely, when controller 140-1 generates a logic low control signal S21 applied to gate node G32, controller 140-1 generates a logic high control signal S11 applied to gate node GF1.

[0124] Note that the switching frequency at which gate nodes G32 and GF1 are turned on and off between time T41 and time T51 is substantially greater than the line frequency associated with the input voltage Vin (as defined by the period of the input voltage Vin between time T31 and time T51).

[0125] Between time T21 and time T31, between time T61 and time T71, etc., the controller 140 controls the operation of the corresponding bidirectional switching circuit device 131-2 and bidirectional switching circuit device 132-2 in a manner similar to that described above (first mode) during the time period between time T41 and time T51. In other words, between time T21 and time T31, between time T41 and time T51, between time T61 and time T71, etc., the controller 140 operates in the second mode (when the polarity of the input voltage Vin is negative).

[0126] Figure 5 This is an example method of operating a corresponding power converter circuit as discussed herein, which includes multiple instances of bidirectional switching circuit devices to convert an input voltage into an output voltage.

[0127] In this example, during processing operation 510, the controller 140 of the power converter 100 controls the operation of the first bidirectional switching circuit device 131.

[0128] In processing operation 520, controller 140 controls the operation of second bidirectional switching circuit device 132, which is connected in series with first bidirectional switching circuit device 131.

[0129] In processing operation 530, via controlled operation of the first bidirectional switching circuit device 131 and the second bidirectional switching circuit device 132, the controller 140 can be configured to control the amplitude of the current 151 through the first winding 161-1 to generate an output voltage Vout supplied to the load 118.

[0130] Note again that the techniques described here are well-suited for power source applications. However, it should be noted that the examples in this article are not limited to such applications, and the techniques discussed herein are also well-suited for other applications.

[0131] Figure 6 This is an example circuit diagram of a power converter as discussed in this article.

[0132] As mentioned above, and again note that the implementation of the example circuit here can vary depending on the application. For example, Figure 6 The diagram illustrates a circuit example including a two-way switch. Note that... Figures 13 to 15 The diagram illustrates a unidirectional circuit option for implementing the techniques discussed herein. Therefore, the techniques described herein can be implemented in both unidirectional and bidirectional circuit options. In this case, the previously defined signals LSN and HSP are combined into a single HS signal driving the unidirectional transistor. Similarly, LSP and HSN can be combined into a single LS signal.

[0133] Additionally, note that the control concept discussed here, using the generated threshold signal TS, can be referred to as a charging control technique such as dq or DeltaQ-based, since control can be implemented based on charging differences. Therefore, the circuit arrangement discussed here can be configured to include the generation of a bypass capacitor voltage (e.g., feedback 725) generated by sensing circuit 145 and the generation of a control threshold (such as the threshold signal TS) generated by threshold level generator 715 (such as an integrated circuit or semiconductor chip), as... Figure 7 As shown.

[0134] exist Figure 6 In this example, the power converter 100-6 includes a power supply 120 (input voltage source), a bidirectional switching circuit device 131, a bidirectional switching circuit device 132, a transformer 161, capacitors C1, C2, C3, and C4, a diode D1, and a load 118.

[0135] The bidirectional switching circuit device 131 includes switches Q11 and Q12. The bidirectional switching circuit device 132 includes switches Q21 and Q22.

[0136] It should also be noted that circuit device 131 may include a single field-effect transistor switch instead of multiple switches Q11 and Q12; circuit device 132 may include a single field-effect transistor switch instead of multiple switches Q21 and Q22. Therefore, it is not necessary to include a bidirectional switching circuit device.

[0137] Transformer 161 includes a primary winding 161-1 and a secondary winding 161-2. The secondary winding 161-2 is magnetically coupled (inductively coupled) to the primary winding 161-1. In this example, the combination of the primary winding 161-1 and capacitor C19 is connected in series between nodes N3 and N6. The secondary winding 161-2 is connected between nodes N7 and N8.

[0138] A power supply 120 providing an input voltage Vin (such as AC voltage, input voltage Vin, or input current 121) is connected between node N1 and node N2. Therefore, the voltage across node N1 and node N2 is Vin.

[0139] Also note that, Figure 6 The power converter 100-6 shown can be configured to include a controller 140 that generates corresponding control signals.

[0140] In one example, controller 140 generates a control signal S11 (also known as LSN) to drive the gate node G of switch Q11. The control signal S11 is used to control switch Q11 between an on state and an off state.

[0141] The controller 140 generates a control signal S12 (also known as HSP) to drive the gate node G of the switch Q12. The control signal S12 is used to control the switch Q12 between the on and off states.

[0142] The controller 140 generates a control signal S21 (also known as HSN) to drive the gate node G of the switch Q21. The control signal S21 is used to control the switch Q21 between the on and off states.

[0143] The controller 140 generates a control signal S22 (also known as LSP) to drive the gate node G of the switch Q22. The control signal S22 is used to control the switch Q22 between the on and off states.

[0144] Note that the flow of current 151 through the primary winding 161-1 (such as changes in amplitude) causes a corresponding current 152 to flow through the secondary winding 161-2. The flow of current 152 through the secondary winding 161-2 across capacitor C4 and load 118 produces an output voltage 123 (also known as Vout).

[0145] Diode D1 prevents current 152 from flowing in the negative direction through the secondary winding 161-2. In other words, the secondary current 152 through the secondary winding 161-2 flows in the direction from node N7 to node N8. Diode D1 prevents current from flowing from node N7 through diode D1 to node N9.

[0146] Notice, Figure 6 The power converter 100-6 in the example is shown in a non-limiting manner.

[0147] In this example, the power supply 100-6 includes a sensing circuit device 145, such as a capacitor C19, a resistor R1, a capacitor C21, and a switch 325.

[0148] The sensing circuit 145 is configured to sense a first power supplied from the input voltage Vin through the first transformer winding 161-1 to the second transformer winding 161-2. As previously described, the second transformer winding 161-2 is magnetically coupled to the first transformer winding 161-1.

[0149] As discussed further here, the controller 140 and the switching circuitry (Q11, Q12, Q21 and Q22) together provide power factor correction, which is associated with converting the input voltage Vin to the output voltage Vout from the output of the second transformer winding 161-2 to the load 118.

[0150] The power factor correction discussed here can be configured to control the flow (and corresponding magnitude) of the first power from the input voltage Vin through the first transformer winding 161-1.

[0151] Additional details regarding the implementation of the sensing circuitry 145 and the corresponding control of the corresponding switches to provide power factor correction are discussed in the following figures and corresponding descriptive text.

[0152] Figure 7 This is an example circuit diagram illustrating the implementation of the power converter and corresponding control as discussed in this article.

[0153] In this example, any power converter discussed herein includes a corresponding switching circuit device 710 (such as switch Q11, switch Q12, etc.) to control the corresponding magnitude of the current (and power P1) supplied from the input voltage Vin to the transformer winding 161-1.

[0154] As previously described, transformer winding 161-2 is magnetically coupled to transformer winding 161-1 to receive power P1 from transformer winding 161-1. In other words, transformer 161 facilitates the transfer of power P1 from transformer winding 161-1 to transformer winding 161-2. The energy P1 received by secondary transformer winding 161-2 serves as the basis for generating the output voltage that supplies power P2 to load 118. The techniques employed here include power factor correction and a fundamental equalization of the average values ​​of power P1 and power P2.

[0155] Furthermore, in this example, note that the controller 140 or other suitable entity may be configured to include circuitry to control the transmission of current 151 through the corresponding winding 161-1 based on a corresponding control signal (threshold signal TS) generated by the threshold level generator 715. Current 151 may or may not pass through the sensing circuitry 145.

[0156] As further shown, note that the circuit arrangement discussed herein can be configured to include one or more of the following: error voltage signal generator 735, filter function 738, threshold level generator 715, and comparator 720.

[0157] As shown in this example, the error voltage signal generator 735 generates a corresponding error signal 731 (such as error voltage, error current, etc.) based on the difference between the amplitude of the output voltage Vout and the setpoint reference voltage 729. In one example, the setpoint reference voltage 729 represents a target amplitude value, wherein the controller 140 adjusts the amplitude of the output voltage Vout via implementing a control signal (threshold signal TS) generated by the generator 715.

[0158] Furthermore, note that the filter function 738 of a PID controller (where P = Proportional, I = Integral, D = Derivative), PI controller, or other suitable entity converts the received error signal 731 into a signal K (such as a filtered error signal) supplied to the threshold level generator 715.

[0159] The threshold level generator 715 receives a signal K, a capacitor C associated with the sensing circuit 145, a switching period Tsw associated with operating the corresponding switch 710, and an input voltage Vin.

[0160] In one example, controller 140 and / or the corresponding circuitry associated with controller 140 generate the control threshold signal TS as follows:

[0161] Set the threshold signal TS = OSV – [(tsw1 / (K*C))*Vin] - cmp, (note that K can be in the denominator or numerator, depending on whether a PID is thus established. The negative sign can also be positive, depending on the sensing configuration. In other words, if the sensing reference is not AC_HB, but capacitor C19 (see...) Figure 6 )).

[0162] Where OSV is the selected offset value; note that OSV can be negative, positive, or zero. In one example, OSV is constant for one switching cycle (not AC), but can vary over time.

[0163] The switching period Tsw1 is a measure of the period of the control switching circuit device around the time of generating the corresponding threshold signal TS. The switching period can be generated based on one or more current or previous switching periods controlling the switch, or potentially based on the current switching period controlling the switch in the current control period.

[0164] Where C is the capacitance associated with the sensing circuit device 145 (there is one of several different configurations associated with the sensing circuit device 145. For some configurations, C is the equivalent capacitance of the bypass capacitor).

[0165] Where K is the feedback (filtered error voltage value), and

[0166] Vin is a sample instance of the amplitude of the input voltage near the time when the corresponding threshold signal TS is generated.

[0167] Here, cmp is an optional compensation factor for the current induced by the AC line (input voltage) in the sensing circuitry. In one example, line current (@50 / 60Hz) from the input voltage flows through capacitors C91 and C92, which disrupts the sensing network. When the current circulates only locally, it is assumed that a portion of that current is transferred to the secondary side through the sensing network. Compensating cmp improves the power factor. The compensation (Cmp) can be proportional to dVin / dt.

[0168] As further shown, comparator 720 is used to generate trigger signal 755, which (by controller 140) serves as the basis for terminating activation of one or more switches Q11 and Q12 during a positive cycle when the input voltage is greater than 0, or terminating activation of one or more switches Q21 and Q22 during a negative cycle when the input voltage is less than 0. Both operations limit the amplitude of the power supplied to transformer winding 161-1 by input voltage Vin to provide power factor correction.

[0169] In one example, in response to the comparator 720 detecting a transition in the feedback 725 (such as the current-based integral indication power P1) across the corresponding threshold signal TS, the comparator 720 generates a corresponding trigger signal 755. As discussed further here, the trigger signal 755 can be used as a basis for terminating the activation of the corresponding switching circuit device 710, so that the input voltage Vin no longer supplies the corresponding power P1 to the transformer winding 161-1.

[0170] Therefore, the controller 140 and the corresponding sensing circuit 145, as discussed herein, can be configured to: i) monitor the magnitude of the first power (the integral of the current 151) via feedback 725 (signal) from the sensing circuit 145, and ii) provide power factor correction via controlled operation of the switching circuit devices (Q11 and Q12) such that the average magnitude of the first power P1 supplied from the input voltage Vin to the first transformer winding 161-1 over time is substantially equal to the average magnitude of the second power P2 supplied from the second transformer winding 161-2 to the corresponding load 118.

[0171] Refer again Figure 6 The circuit path between nodes N3 and N6 can be configured to include a first capacitor C19 coupled in series with the first transformer winding 161-1. The controller 140 is operable to control the flow of resonant current through the series circuit path via control of the switching circuitry. The controlled flow of the resonant current is operable to provide power factor correction as discussed herein.

[0172] Furthermore, in this example, the sensing circuit device 145 includes a series circuit path that includes a capacitor C21 connected in series with the switch 325, wherein the series circuit path (the combination of C21 and the switch 325) is connected in parallel with the capacitor C19.

[0173] Furthermore, in this example, as previously described, the sensing circuit 145 includes a sensing capacitor C19 connected in series with the first transformer winding 161-1 to sense a first charge quantity (or charge amount or charge amplitude) supplied from the input voltage Vin to the first transformer winding 161-1. Note that a bypass capacitor may not sense power; it produces the arithmetic sum of the amount of charge flowing through the bypass capacitor (i.e., the capacitor is based on the integral of the current flowing through it). The sensing capacitor C19 is initially set to an initial state, such as 0 volts or other suitable value for a measurement phase to monitor the amplitude of power P1. The output voltage Vout is generated by the secondary winding of power supply 100-6 based on the power received by transformer winding 161-2 (i.e., the second transformer winding 161-2 receives power from the first transformer winding 161-1). The second transformer winding 161-2 uses the received power P1 to generate the output voltage Vout to power load 118. Load 118 consumes power P2.

[0174] Examples here include controlling the magnitude of the power input to the first winding 161-1 and transmitted to the second winding 161-2 to provide power factor correction. For example, sensing circuit 145 generates a corresponding feedback signal 725 (also referred to as, for example, the voltage V19 across capacitor C19). Controller 140: i) monitors the magnitude of the voltage across capacitor C19 (such as feedback 725) to determine the magnitude of the power P1 supplied to transformer winding 161-1, and ii) controls the operation of switching circuit devices (switches Q11, Q12, Q21, and Q22) such that the average magnitude of the first power transmitted from winding 161-1 to winding 161-2 over multiple control cycles is substantially equal to the average magnitude of the second power consumed by the corresponding dynamic load 118 over multiple control cycles. In other words, on average, the magnitude of the power supplied to transformer winding 161-1 and transmitted to second transformer winding 161-2 should generally be equal to the magnitude of the power supplied by secondary winding 161-2 and consumed by load 118.

[0175] As previously discussed, as a non-limiting example, sensing circuit 145 includes sensing capacitor C19, which is operable to store a voltage value (also referred to as feedback 725) indicating the integral (and also power) of the current 151 supplied through the first transformer winding 161-1.

[0176] As further discussed below, the threshold level signal generator circuit 715 generates a threshold signal TS (threshold level) based on one or more parameters, such as the signal K, the capacitance associated with the sensing circuit 145, the amplitude of the switch Tsw, and the input voltage Vin. The controller 140 controls the switching circuitry based on the threshold signal TS to provide power factor correction. Figure 8The details related to the generation threshold signal TS are further discussed in the following section.

[0177] Figure 8 This is an example diagram illustrating the derivation of the control method discussed in this article.

[0178] In this example, the threshold signal TL (such as the threshold of [Tsw / (K*C)]*Vin) is derived based on multiple received values ​​(such as the input voltage Vin, the switching period Tsw, the filtered error signal K, and the capacitance C associated with the sensing circuit 145).

[0179] The relationship between the average periodic current, the resonant capacitor, and Δ(delta)V discussed here is useful.

[0180] As previously described, comparator 720 (along with the corresponding threshold signal TS and feedback 725) can be used to determine the exact point in time when iPR1 or input current 121 (or current 151) and the corresponding first power P1 supplied to transformer winding 161-1 have reached a certain threshold level. In one example, the average value of current 151 is integrated over time via the corresponding sensing circuit 145 to detect the first power P1 supplied to transformer winding 161-1 by input voltage Vin.

[0181] In one example, the actual control is based on the stored charge q, which represents the amount of charge transferred; and the charge transferred over the duration is the average current.

[0182] In another example, if the input current iPRI remains proportional to the input voltage, the input behaves like PFC (Power Factor Correction).

[0183] Here is an example definition of PFC: if iIPRI = k*Vin, then it is a PFC.

[0184] The only thing left is to ensure that ΔV is proportional to Vin.

[0185] For example, when the PFC requirement is met, Δu = Δu = T_sw / KC*V_in.

[0186] Tsw is the switching cycle.

[0187] In one example, signal K is a scaling factor representing the system load resistance.

[0188] C is the equivalent capacitance of the sensing network associated with sensing circuit 145.

[0189] For closed-loop control, K can be used as a feedback input.

[0190] Any regulator (PI, PID, P...) or filter (such as low-pass filter, notch filter...) with low gain at AC frequencies such as approximately 50 to 100 Hz can be used to generate signal K.

[0191] The input voltage Vin to the threshold signal generator 715 can be measured during operation (instantaneous voltage) because the input voltage Vin changes over time, and the magnitude of the first power stored in the transformer winding 161-1 over time depends on the magnitude of the input voltage Vin (such as a sine wave).

[0192] Any suitable circuit can be used to generate and / or measure Tsw, such as a so-called bang-bang circuit that determines the most recent or last value of the switching cycle, the slope (the actual value of the current cycle), etc.

[0193] Figure 9 This is a timing diagram illustrating the control of the corresponding power converter over multiple AC input voltage cycles, as discussed in this article.

[0194] As shown in timing diagram 800, the input voltage Vin is essentially a sine wave (AC signal, AC current, or AC voltage), with positive polarity between times T31 and T41, and between times T51 and T61. The input voltage Vin is negative polarity between times T21 and T31, and between times T41 and T51.

[0195] Voltage V4 is the voltage at node N6.

[0196] Signal S21 (HSN) controls the operation of switch Q21; signal S12 (HSP) controls the operation of switch Q12; signal S11 (LSN) controls the operation of switch Q11; and signal S22 (LSP) controls the operation of switch Q22.

[0197] When the input voltage Vin is positive: Controller 140 generates a logic high control signal S11 to activate the corresponding switch Q11. Controller 140 also generates a logic high control signal S21 to activate switch Q21. Controller 140 alternately switches between activating switch Q12 and switch Q22 via signals S12 and S22. As discussed herein, controlling switches Q21 and Q22 supplies the desired amount of power from the input voltage Vin through transformer winding 161-1, thereby providing power factor correction.

[0198] When the input voltage is negative: Controller 140 generates a logic high control signal S12 to activate the corresponding switch Q12. Controller 140 also generates a logic high control signal S22 to activate switch Q22. Controller 140 alternately switches between activating switch Q11 and switch Q21 via signals S11 and S21. As discussed herein, switches Q11 and Q21 are controlled to supply the desired amount of power from the input voltage Vin through transformer winding 161-1.

[0199] As previously stated, feedback 725 is generated by sensing circuit 145 (monitor circuit), which monitors the magnitude of the current 151 (and therefore the power) supplied to transformer winding 161-1.

[0200] As further shown and as previously discussed, timing diagram 900 also illustrates how the amplitude of the threshold signal TL varies over time to provide power factor correction. For example, as previously described, controller 140 receives feedback signal 725 (shown as envelope and timing diagram 900) from sensing circuit 145. By comparing feedback signal 725 with the threshold signal TL (also referred to as the control signal), controller 140 provides power factor correction based on controlled operation of the switching circuitry. In one example, controlled operation of the switching circuitry results in the average amplitude of a first power (as indicated by sensing circuit 145) supplied from input voltage Vin to first transformer winding 161-1 being substantially equal to the average amplitude of a second power (such as the output current supplied by output voltage Vout) output from second transformer winding 161-2 to the corresponding load 118.

[0201] Figure 10 This is an example timing diagram illustrating the control of the corresponding power converter as discussed here.

[0202] In this example, at time T33 in timing diagram 1000, controller 140 or other suitable entity precharges the voltage of bypass capacitor C19 (as associated with sensing circuitry 145) to a known value. Any suitable precharge value can be used, which may include shorting node N22 to node N3.

[0203] As shown in timing diagram 1000, because the AC input voltage Vin is positive, controller 140 activates both switches Q11 and Q21 to the ON state. Controller 140 alternately switches between activating switches Q12 and Q22 to provide power factor correction as discussed here.

[0204] Between time T32 and time T33, controller 140 sets switch Q22 to the ON state and switch Q12 to the OFF state. In this state, the energy stored in transformer winding 161-1 is discharged based on the generation of output voltage Vout. In other words, the energy in transformer 161-1 decreases.

[0205] Just after or near time T33, controller 140 generates control signal S12 to activate switch Q12 to the ON state. Controller 140 generates control signal S22 to deactivate switch Q22 to the OFF state. Therefore, at or near time T33, energy intake begins (to the transformer winding 161-1), with the amplitude of the current 151 supplied from the input voltage Vin through capacitor C19 and transformer winding 161-1 increasing.

[0206] Note that the primary current is also the magnetizing current 167, because transformer 161 is used as an inductor.

[0207] Due to the increased flow of current 151 (resonant current) from input voltage source 120 through the combination of switches Q11 and Q12, and through capacitor C19 and transformer winding 161-1, starting at time T33, the voltage V19 across bypass capacitor C19 (feedback 725) decreases from an initial pre-charge value of 1 volt or other suitable value. In other words, the amplitude of feedback signal 725 decreases proportionally to the amount of initial energy stored in transformer winding 161-1.

[0208] As previously described, sensing circuit device 145 ( Figure 7 The controller 140 implements a comparator 720 to compare feedback 725 (or voltage V19 from capacitor C19) with a threshold signal TS generated by threshold level generator 715. When comparator 720 detects at or near time T34 that the amplitude of feedback 725 has decreased below the threshold signal TS (such as -40mV or other suitable amplitude below the precharge voltage), comparator 720 generates a trigger signal 755 to notify controller 140 of a trigger event (the amplitude of feedback 725 has decreased below the threshold signal TS). In response to the trigger event indicated by trigger signal 755, controller 140 deactivates switch Q12 and activates switch Q22 at or near time T34.

[0209] At or near time T34, the deactivation of switch Q12 ends the energy intake phase (such as receiving energy from the input voltage Vin from the transformer winding 161-1) and begins the energy transfer phase (energy in the transformer winding 161-1 is transferred to the transformer winding 161-2 to generate the corresponding output voltage Vout).

[0210] Due to the blocking capacitors C2 and C3, most of the current flowing out of the bypass capacitor flows back into the interior. At the end of the energy transfer cycle (where the cycle is between time T33 and time T36) (such as near time T36), the voltage V19 across the bypass capacitor C19 returns almost to its pre-charge value of about 1 volt.

[0211] In one example, resistor R1 provides drift cancellation relative to capacitor C19, although any suitable drift capacitor circuit can be used.

[0212] Therefore, one concept discussed here is that voltage V19 (such as feedback 725) can be used to determine ΔQ, which can be converted to the average value of current 151 and the corresponding first energy during the intake period (such as between time T33 and time T34), and (for hybrid flyback) can also be converted to the average current during the period (Tsw) such as between time T33 and time T36. The power converter discussed here can be configured to match the average current and average voltage by a constant (depending on feedback, no gain at 50 / 60Hz) during a switching cycle, resulting in power factor correction.

[0213] Therefore, the feedback 725 (a high-frequency signal relative to the frequency of the input voltage Vin) generated by the sensing circuit device 145 is the sensing signal used for DeltaQ control. The threshold signal TS is a threshold value compared with signal 725. When both are the same (crossed) as detected by comparator 720, controller 140 terminates the intake of energy from the input voltage Vin to transformer winding 161-1 at or near times T34, T37, T38, etc. DeltaQ can be higher or lower than the threshold, especially during the energy transfer phase.

[0214] It should also be noted that the amplitude of the feedback signal 725 can be proportional to the amplitude of the energy transferred from the input voltage Vin to the transformer winding 161-1, wherein the amplitude of the feedback signal 725 is configured to increase over time rather than decrease over time. In this case, the comparator 720 or other suitable entity can be configured to generate a trigger signal 755 in response to detecting that the amplitude of the feedback signal 725 is greater than the threshold signal TS.

[0215] Therefore, as previously discussed, the controller 140, as described herein, can be configured to continuously adjust (cycle after cycle, as needed) the operation of the switching circuitry (i.e., Q11, Q12, Q21, Q22) by generating a threshold signal TS based at least on the amplitude of the output voltage Vout and possibly other parameters. The comparator 720 compares the feedback signal 725 (indicating the amplitude of the first energy delivered to the transformer winding 161-1) generated by the sensing circuit 145 with the threshold signal TS. Based on the comparison and the detection of a trigger event, such as the amplitude of the feedback signal 725 decreasing below the threshold signal TS, the controller 140 operates the switching circuitry to terminate the flow of the first current 151 through the first transformer winding 161-1.

[0216] In one example, the amplitude of the threshold signal TS varies over time. Figure 11 Another example is shown in more detail below.

[0217] See again Figure 10 Furthermore, as previously discussed, the threshold level signal generator 715 can be configured to generate a threshold signal TS based on a combination of one or more of the following: i) the amplitude of the input voltage Vin, ii) the capacitor CSC associated with the sensing circuit device 145 that generates the feedback signal 725, iii) a filtered error voltage signal K derived based on the difference in amplitude between the output voltage Vout and the setpoint reference voltage 729 (i.e., the filtered error voltage signal K derived by the error voltage signal generator 785 based on the error signal 736 generated based on the difference in amplitude between the output voltage Vout and the setpoint reference voltage 729), and iv) the switching period Tsw of the corresponding switching circuit device.

[0218] Note that the switching frequencies associated with generating the corresponding control signals S11, S12, S21, and S22 can vary over time. As previously mentioned, the duty cycle associated with activating the corresponding high-side switching circuit device Q12 can also vary over time based on the amplitude of the threshold signal TS (which varies to provide a threshold level for control and power factor correction).

[0219] In one example, as previously described, in response to detecting that the amplitude of the feedback signal 725 has decreased below the threshold level TS, the controller 140 sets the corresponding switch Q12 to the off state, so that the input current iIN from the voltage source 120 no longer flows through the switch and transformer winding 161-1 (the amplitude of current 151 stops increasing). The remaining energy stored in the primary winding 161-1 is transferred to the secondary transformer winding 161-2 to generate the output voltage Vout.

[0220] As previously discussed, the threshold signal TS can change slowly with dependence on the input voltage (Vin) and simultaneously with the change in the compensation period (Tsw).

[0221] In one example, the offset value OSV associated with the threshold signal TS is used to keep the value TS within the range of comparator 720. In other words, the offset OSV can be implemented to prevent the threshold signal TS (threshold level) from exceeding the comparator (720) rail (1V in analog, which has no effect on control), where the value K is the feedback input (e.g., it could be an equivalent input resistance forced by control – changing it alters the average energy intake), and the capacitance C is the gain of the sensing capacitor network in coulombs.

[0222] Figure 11 This is an example timing diagram illustrating the control of the corresponding power converter as discussed in this article.

[0223] Timing diagram 1100 illustrates an example of sensed voltage (Vsence or feedback 725 generated by sense circuit 145) and Δ(delta)U control. In this example, the voltage Vsence (also referred to as feedback 725) is a sensed voltage feedback generated by a bypass capacitor or capacitive voltage divider circuit (such as sense circuit 145), which monitors the magnitude of current 151 or the magnitude of energy supplied to or through primary winding 161-1.

[0224] In one example, at the start of inhalation (also known as the end of transfer) at a time such as approximately T92, the voltage signal Vsense and the corresponding sensing circuit 145 are initialized to known values.

[0225] Note that some drift in the initial conditions (pre-charge) associated with sensing circuit 145 is expected. Without resetting sensing circuit 145 and the corresponding capacitor, the U1 value will be lost over time.

[0226] As further shown, the next cycle begins when Vsense (feedback 725) (such as at time T91, time T93, etc.) is equal to, crosses, or falls below the threshold control line (threshold signal TS) at voltage U2. For convenience, as previously discussed, an offset OSV can be added to the control signal (TS). When the offset is canceled out, the offset value OSV associated with the threshold signal TS has no effect on ΔU. As previously discussed, the use of the offset value OSV reduces the need for an additional set of comparator voltage rails for operating comparator 720.

[0227] therefore, Figure 11Timing diagram 1100 illustrates how the amplitude of the threshold signal TS changes over time from one cycle to the next, and how the amplitude of the feedback 725 generates the corresponding trigger signal to start a new cycle, as previously discussed.

[0228] Figure 12 This is an example circuit diagram illustrating the power converter circuit discussed in this article and the corresponding primary winding power / energy flow sensing.

[0229] Depending on the control position, as shown in power source 600-12, additional windings 1110 and capacitor C93 can be implemented to bring the sensed voltage (feedback signal 725 generated by sensing circuit 145-1) to the same reference as controller 140. This is useful for bridgeless HBF (hybrid flyback) configurations due to the floating arrangement of controller 140.

[0230] Figure 13 This is an example diagram illustrating a sensing circuit as discussed herein, which is configured to monitor the magnitude of the current (first energy) passing through a corresponding primary transformer winding.

[0231] In this example, power source 600-13 is essentially the same as power source 600 discussed earlier. However, controller 140 controls the operation of switches 131 and 132 to transfer first energy from input voltage Vin (from node N1) to the first transformer winding 161-1, which is magnetically coupled to transformer winding 161-2.

[0232] In this example, the sensing circuit 1301 (such as a capacitor divider circuit) is implemented to monitor the amplitude of the energy supplied to the transformer winding 161-1, including multiple capacitors C201 and C202 connected in series between node N6 and node N2. The combination of capacitors C201 and C202 is connected in parallel with the resonant capacitor C3.

[0233] Furthermore, in this example, node N210 of coupling capacitors C201 and C202 generates a corresponding feedback signal 725, which indicates the magnitude of the first energy P1 supplied to transformer winding 161-1 via the control of switches 131 and 132.

[0234] In one example, as previously described, power source 600 (and any corresponding instances such as 600-1, 600-2, 600-3, etc.) is implemented as a flyback power converter operating in resonant mode via controlled switching of corresponding switches 131 and 132 implemented by controller 140.

[0235] In one example, controller 140 controls the flow of first energy P1 through first winding 161-1 by controlling the resonant current (such as current 151) flowing from input voltage Vin through first transformer winding 161-1. As discussed herein, the controlled flow of the resonant current (such as current 151) through first transformer winding 161-1 provides power factor correction such that the average magnitude of the first energy supplied from input voltage to first transformer winding (such as over one or more cycles of input voltage) is substantially equal to the average magnitude of the second energy supplied from second winding to the corresponding load.

[0236] Therefore, charging control and power factor correction via transformer winding 161-1, as discussed herein, are options for any resonant power converter topology that links the resonant capacitor voltage to the energy transfer from one transformer winding 161-1 to the transformer winding 161-2.

[0237] During the high-side turn-on time (such as the activation of switch 131), the voltage difference across the capacitor is proportional to the input current.

[0238] Δu=U_end-U_start=1 / C∫_(T_intk)^〖I_intk dt〗=I_avg / C

[0239] By controlling the ΔV of the transresonant capacitor C3, the average input current is controlled, providing most of the advantages of current-mode control with resilience against reactive / circulating current.

[0240] Figure 14 This is an example diagram illustrating a sensing circuit as discussed herein, which is configured to monitor the magnitude of the current (first energy) passing through a corresponding primary transformer winding.

[0241] In this example, power source 600-14 is essentially the same as power source 600 discussed earlier. However, in this example, controller 140 controls the operation of switches 131 and 132 to transfer first energy from input voltage Vin to the first transformer winding 161-1, which is magnetically coupled to transformer winding 161-2.

[0242] Sensing circuit 1401 (such as a bypass capacitor sensing circuit) is implemented to monitor the magnitude of energy supplied to transformer winding 161-1, and includes capacitor C211 connected in series with capacitor C3 between node N6 and node N2. Node N220, which couples capacitors C3 and C211, generates a corresponding energy feedback signal (such as feedback 725) indicating the magnitude of a corresponding first energy supplied to transformer winding 161-1 by input voltage Vin via control of switches 131 and 132.

[0243] Figure 15 This is an example diagram illustrating a sensing circuit as discussed herein, which is configured to monitor the magnitude of the current (first energy) passing through a corresponding primary transformer winding.

[0244] In this example, power source 600-15 is essentially the same as power source 600 discussed earlier. However, in this example, controller 140 controls the operation of switches 131 and 132 to transfer first energy from input voltage Vin to the first transformer winding 161-1, which is magnetically coupled to transformer winding 161-2.

[0245] As shown in the figure, the sensing circuit 1501 (such as via a floating reference sensing circuit) is implemented to monitor the amplitude of the power source 600-3 of the transformer winding 161-1.

[0246] Figure 16 This is an example method for controlling a corresponding resonant power converter to provide power factor correction, as discussed in this paper.

[0247] In processing operation 1810, controller 140 controls the flow of first energy (such as based on resonant current 151 or iPRI) received from input voltage Vin and transmitted through first transformer winding 161-1 via magnetic coupling to second transformer winding 161-2 via switching circuit devices such as switches Q11, Q12, Q21, Q22. The controlled flow of first energy (such as resonant current 151) from first transformer winding 161-1 to second transformer winding 161-2 results in the generation of output voltage Vout based on second energy supplied from second transformer winding 161-2 to load 118 (to the first power converter of the second energy).

[0248] In processing operation 1820, controller 140 receives feedback such as 725 (such as voltage V19 across capacitor C19 or feedback 725 from any of the different sensing circuit devices 145 discussed herein). In one example, feedback 725 multiplied by the input voltage indicates the magnitude of the first energy transmitted through the first winding 161-1 during a duration such as between time T33 and time T34.

[0249] In processing operation 1830, controller 140 adjusts the operation of switching circuit devices (switches Q11, Q12, Q21, Q22) over time based at least on feedback (indicating the first energy P1) and the amplitude of the output voltage Vout. The adjusted operation of the switching circuit devices (such as activating the termination of the high-side switching circuit device Q12) adjusts the amplitude of the first energy P1 supplied to the first transformer winding 161-1 each cycle. More specifically, as previously described, the switching circuit devices are activated at time T34 to prevent further energy P1 from being supplied to the transformer winding 161-1.

[0250] In one example, as previously described, the controlled flow of the first energy through the first winding 161-1 includes controlling a resonant current (such as current 151) flowing from the input voltage Vin through the first transformer winding 161-1. The controlled flow of the resonant current through the first transformer winding provides power factor correction associated with converting the input voltage Vin to the output voltage Vout via a flyback power converter configuration as shown in power converter 600 and other power converters as previously discussed.

[0251] Note again that the techniques described here are well-suited for power source applications. However, it should be noted that the examples in this article are not limited to such applications, and the techniques discussed herein are also well-suited for other applications.

[0252] While the invention has been specifically shown and described with reference to preferred embodiments thereof, those skilled in the art will understand that various changes in form and detail may be made without departing from the spirit and scope of the invention as defined by the appended claims. These changes are intended to be covered by the scope of the invention. Therefore, the foregoing description of the examples of this application is not restrictive. Rather, any limitations of the invention are set forth in the following claims.

Claims

1. An apparatus comprising: First transformer winding; A sensing circuit device operable to sense a first energy supplied from an input voltage to the first transformer winding; as well as A switching circuit device operable to apply a power factor correction associated with converting the input voltage into an output voltage derived from the output of a second transformer winding, the second transformer winding being magnetically coupled to the first transformer winding, the applied power factor correction operable to control the flow of the first energy from the input voltage to the first transformer winding.

2. The apparatus of claim 1, wherein the output voltage is operable to supply a second energy to a corresponding load, the apparatus further comprising: The controller is operable to: i) monitor the magnitude of the first energy via feedback from the sensing circuit device, and ii) apply the power factor correction via controlled operation of the switching circuit device such that the average magnitude of the first energy supplied from the input voltage to the first transformer winding is substantially equal to the average magnitude of the second energy supplied from the second transformer winding to the corresponding load.

3. The apparatus of claim 1, wherein the sensing circuit device includes a first capacitor, and the apparatus further includes: The series circuit path includes the first capacitor coupled in series with the first transformer winding.

4. The apparatus according to claim 3, further comprising: The controller is operable to control the flow of resonant current through the series circuit path via control of the switching circuit device, the flow of the resonant current being controlled based on the applied power factor correction.

5. The apparatus of claim 1, wherein the sensing circuitry includes a capacitor operable to sense the magnitude of the first energy supplied from the input voltage to the first transformer winding via the switching circuitry.

6. The apparatus of claim 5, wherein the output voltage derived from the output of the second transformer winding is operable to supply second energy to a corresponding load, the apparatus further comprising: The controller is operable to: i) monitor feedback received from the sensing circuit device, the feedback indicating the amplitude of the first energy, and ii) control the operation of the switching circuit device via the power factor correction such that the average amplitude of the first energy over a plurality of control cycles is substantially equal to the average amplitude of the second energy over the plurality of control cycles.

7. The apparatus of claim 1, wherein the sensing circuit includes a series circuit path, the series circuit path including a first capacitor connected in series with the first switch, and the series circuit path being connected in parallel with the first transformer winding.

8. The apparatus of claim 1, wherein the sensing circuitry includes a sensing capacitor operable to store a voltage value indicating the integral of the current supplied by the input voltage through the first transformer winding.

9. The apparatus according to claim 1, further comprising: A signal generator circuit operable to generate a threshold signal based at least in part on the magnitude of the output voltage relative to a setpoint reference voltage; as well as The controller is operable to control the switching circuitry and the flow of the first energy from the input voltage to the first transformer winding based on the threshold signal, in order to apply the power factor correction.

10. An apparatus comprising: The controller is operable to: By controlling the switching circuit device, the flow of first energy received from the input voltage through magnetic coupling to the first transformer winding of the second transformer winding is controlled. The controlled flow of the first energy through the first transformer winding to the second transformer winding is operable to generate an output voltage based on the second energy supplied to the load from the output of the second transformer winding. Receive feedback indicating the magnitude of the first energy; as well as The operation of the switching circuit device is adjusted over time based at least on the feedback and the amplitude of the output voltage. The adjusted operation of the switching circuit device is operable to adjust the amplitude of the first energy supplied to the first transformer winding.

11. A method comprising: The flow of first energy received from the input voltage through a first transformer winding is controlled via a switching circuit device. The first transformer winding is magnetically coupled to a second transformer winding. The controlled flow of the first energy through the first transformer winding to the second transformer winding generates an output voltage based on the second energy supplied from the output of the second transformer winding to the load. Receive feedback indicating the magnitude of the first energy; as well as Based at least on the received feedback and the magnitude of the output voltage, a power factor correction is applied via time-varying control of the switching circuit device, the control of the switching circuit device including: adjusting the magnitude of the first energy supplied to the first transformer winding.

12. The method of claim 11, wherein controlling the flow of the first energy comprises: Control the resonant current supplied by the input voltage through the winding of the first transformer.

13. The method of claim 11, wherein the control of the switching circuit device over time substantially balances the average amplitude of the first energy and the average amplitude of the second energy.

14. The method of claim 11, wherein the control of the switching circuit device over time comprises: The operation of the switching circuit device over time is adjusted based on one or more of the following: i) the feedback indicating the amplitude of the first energy, ii) the amplitude of the output voltage, iii) the amplitude of the input voltage, iv) the capacitance associated with the sensing circuit device that generates the feedback, and iv) controlling the switching cycle of the switching circuit device.

15. The method of claim 11, wherein the control of the switching circuit device comprises: Generate a threshold signal; The feedback indicating the magnitude of the first energy is compared with the threshold signal; as well as The flow of the first current through the first transformer winding is terminated based on the comparison.

16. The method of claim 15, wherein the amplitude of the threshold signal varies over time based on the amplitude of the input voltage.

17. The method of claim 15, further comprising: The threshold signal is generated based on a combination of the following: i) The magnitude of the output voltage relative to the setpoint reference voltage. ii) the amplitude of the input voltage, and iii) A capacitor associated with the sensing circuitry that generates the feedback.

18. The method of claim 15, wherein the threshold signal is a threshold signal TS; The generation of the threshold signal TS includes: Set the threshold signal TS = OSV – [(tsw1 / (K*C))*Vin] – cmp. Where OSV is the offset value. Where tsw1 is a measure of the period controlling the switching circuit device. Where C is the capacitor associated with the sensing circuit device that generates the feedback. Where K is an error voltage derived by comparing the amplitude of the output voltage with a setpoint reference voltage. Where Vin is the amplitude of the input voltage, and Where cmp is an optional compensation factor for the current induced by the input voltage in the sensing circuit device.

19. The method of claim 11, further comprising: The feedback received is generated by the sensing circuitry based on the integral of the magnitude of a first current supplied from the input voltage through the first transformer winding.

20. The method of claim 19, wherein the first transformer winding is disposed in a resonant circuit, and the first current is a resonant current flowing through the first transformer winding.

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