Independent winding linear motor control method, system and electronic equipment

Abstract

This invention relates to a control method, system, and electronic device for an independent winding linear motor, belonging to the field of motor control technology. The method includes: driving a mover; real-time detection of the mover's position and generating a position feedback signal; based on the position feedback signal, performing gating control on the stator windings and dynamically dividing the stator windings into multiple vector control combinations, synchronously acquiring the three-phase current signals of each vector control combination; calculating the three-phase current vector sum for each three-phase current signal; obtaining the zero-sequence current based on the three-phase current vector sum and performing closed-loop control on it to generate a voltage compensation quantity; performing field-oriented control on each vector control combination to generate a three-phase voltage control signal; and superimposing the voltage compensation quantity and the three-phase voltage control signal to obtain a control signal, thereby achieving independent closed-loop control. This invention improves voltage utilization and operational stability, and solves the problems of insufficient motor output and overheating.

Description

Technical Field

[0001] This invention relates to the field of motor control technology, and in particular to a control method, system and electronic equipment for an independent winding linear motor. Background Technology

[0002] In recent years, the rapid iteration of intelligent manufacturing and precision conveying technologies has driven the upgrading of flexible conveying systems towards higher efficiency and precision. Flexible conveying systems driven by permanent magnet linear motors, with their unique advantages such as fast response speed, high positioning accuracy, and flexible layout, have gained widespread attention and application in automated production lines, logistics sorting, and precision manufacturing, becoming a core component of intelligent manufacturing equipment. Among them, the independent coil permanent magnet linear motor system, due to the physical isolation characteristics of the distributed coils, allows each coil unit to form an independent current path, possessing stronger modular expansion capabilities and motion control flexibility, further meeting the diverse application needs of flexible conveying systems.

[0003] However, the structural characteristics of independent coil permanent magnet linear motor systems also present technical bottlenecks, especially the series of problems caused by zero-sequence current, which severely restricts the improvement of system performance. In current technical solutions, all coils generally adopt a common bus voltage power supply method, making the phase voltage, line voltage and bus voltage of each independent coil directly equivalent. When a space vector control strategy is adopted, the phase voltage waveform exhibits typical saddle wave characteristics. This modulation method inevitably introduces third harmonic components, thereby exciting significant zero-sequence current. At the same time, when the mover enters and exits different coil regions, the abrupt change in magnetic field distribution will trigger boundary effects, further aggravating the generation and uncontrollable fluctuations of zero-sequence current. More importantly, the abrupt change effect of the end magnetic field unique to linear motors will cause the amplitude and fluctuation of zero-sequence current to deteriorate sharply in the motion boundary region, forming a superposition effect of third harmonic, boundary effect and end effect.

[0004] The presence of zero-sequence current presents three core challenges to the system: First, voltage resources are ineffectively occupied by zero-sequence circulating current, failing to be converted into effective work to drive linear motion, resulting in a significant reduction in bus voltage utilization and directly impacting system operating efficiency and output capacity. Second, multi-path zero-sequence current forms high-frequency circulating eddy currents in the dispersed coils, causing a surge in winding copper losses and core eddy current losses, resulting in continuous high-temperature heating of the motor body and power devices, which not only shortens equipment lifespan but also severely restricts the system's continuous operating capability. Third, uncontrollable fluctuations in zero-sequence current lead to a decrease in motor output thrust and a deterioration in operational stability, further affecting the positioning accuracy and motion smoothness of the flexible conveyor system. These intertwined problems cause independent coil permanent magnet linear motor systems to face high energy consumption, low efficiency, and thermal runaway issues, becoming key technical obstacles restricting their application to higher performance and wider scenarios.

[0005] To address the aforementioned issues, existing control technologies have yet to provide effective solutions. How to optimize control strategies, develop precise and real-time zero-sequence current compensation methods, suppress the generation and fluctuations of zero-sequence current, overcome voltage utilization bottlenecks, and fundamentally solve the overheating problem has become a core technical challenge that urgently needs to be addressed in the current research and development of flexible conveying systems using independent coil permanent magnet linear motors. Summary of the Invention

[0006] Therefore, the technical problem to be solved by the present invention is to overcome the defects of low bus voltage utilization, high temperature heating of windings and power devices, low motor output thrust and poor operation stability in the prior art. At the same time, the existing control technology lacks an effective zero-sequence current compensation scheme and cannot overcome the defects of high energy consumption, low efficiency and thermal runaway.

[0007] In a first aspect, to solve the above-mentioned technical problems, the present invention provides a control method for an independent winding linear motor, comprising:

[0008] S1. The moving part is driven by the driving force generated by the current excitation of the stator coil winding; the position of the moving part is detected in real time, and a position feedback signal is generated.

[0009] S2. Based on the position feedback signal, perform gating control on the stator coil winding, and dynamically divide the stator coil winding into multiple vector control combinations, and synchronously acquire the three-phase current signal of each vector control combination;

[0010] S3. Perform sign number addition on each of the three-phase current signals to obtain the three-phase current vector sum; obtain the zero-sequence current based on the three-phase current vector sum; perform closed-loop control on the zero-sequence current to generate voltage compensation amount;

[0011] S4. Perform field-oriented control on each of the vector control combinations to generate a three-phase voltage control signal; superimpose the voltage compensation amount and the three-phase voltage control signal to obtain a control signal;

[0012] S5. According to the control signal, control the magnitude and direction of the current in the stator coil winding to complete the independent closed-loop control of the current in each phase stator coil winding.

[0013] In one embodiment of the present invention, in step S3, the expression for the voltage compensation amount is:

[0014] ;

[0015] in, , , and This indicates the corresponding zero-sequence voltage compensation amount; , , and These represent the zero-sequence currents of the four vector control combinations, respectively. , , and These represent the integral gains of the four vector control combinations; Represents the complex frequency variable in the Laplace transform; Indicated based on zero-sequence current and Dynamically adjustable proportional gain; Indicated based on zero-sequence current and Dynamically adjustable proportional gain; Indicated based on zero-sequence current and Dynamically adjustable proportional gain; Indicated based on zero-sequence current and Dynamically adjustable proportional gain.

[0016] In one embodiment of the present invention, in step S3, the process of performing closed-loop control on the zero-sequence current and generating the voltage compensation amount includes transforming the zero-sequence current using a transformation matrix, wherein the expression of the transformation matrix is:

[0017] ;

[0018] in, , , and These represent the zero-sequence currents of the four vector control combinations, respectively. , and They represent the first Three-phase current of a vector control combination.

[0019] In one embodiment of the present invention, the method for synchronously acquiring the three-phase current signal of each vector control combination in step S2 is as follows: the three-phase current signal of each vector control combination is synchronously acquired using an architecture of cascaded three-stage integrators and three-stage differentials.

[0020] In one embodiment of the present invention, step S4, which involves performing field-oriented control on each of the vector control combinations to generate a three-phase voltage control signal, is as follows:

[0021] Obtain the three-phase current in each of the vector control combinations;

[0022] The three-phase current is converted into a first current and a second current by Clark;

[0023] The first current and the second current are transformed using Park to obtain the third current and the fourth current in the rotating coordinate system;

[0024] The first error between the actual value and the reference value of the third current and the second error between the actual value and the reference value of the fourth current are obtained respectively; the first voltage vector and the second voltage vector are calculated based on the first error and the second error.

[0025] The first voltage vector and the second voltage vector are transformed into a two-phase stationary coordinate system by Park inverse transformation to obtain the first orthogonal voltage value and the second orthogonal voltage value.

[0026] SVPWM control is applied to the first quadrature voltage value and the second quadrature voltage value to obtain a three-phase voltage control signal.

[0027] In one embodiment of the present invention, the step of performing SVPWM control on the first quadrature voltage value and the second quadrature voltage value to obtain a three-phase voltage control signal is as follows:

[0028] Determine the sectors where the first orthogonal voltage value and the second orthogonal voltage value are located;

[0029] Calculate the duration of action of the basic vector corresponding to each sector, and calculate the sum of the durations of the action times;

[0030] Determine whether the total action time exceeds the modulation period. If so, reduce the action time by a preset ratio to obtain a new action time; otherwise, do not reduce the action time, and the action time is the new action time.

[0031] Based on the new operating time, the three-phase voltage switching time point is calculated, and a three-phase voltage control signal is generated.

[0032] Secondly, to solve the above-mentioned technical problems, the present invention provides an independent winding linear motor control system for implementing the above-mentioned independent winding linear motor control method, comprising:

[0033] The stator winding module is equipped with multiple stator coil windings;

[0034] A mover drive module includes a mover and a permanent magnet array; wherein the permanent magnet array generates a driving force to drive the mover by means of current excitation from the stator coil winding;

[0035] The motor drive module includes multiple H-inverter bridge drive circuits, each of which includes multiple MOSFET devices; each phase of the stator coil winding is connected to the H-inverter bridge drive circuit.

[0036] A position detection module is used to detect the movement position of the mover drive module and generate a position feedback signal;

[0037] The control module is used to control the signal duty cycle of the MOSFET device according to the position feedback signal and the feedback signal of the stator coil winding current; and to perform independent closed-loop control of the winding current of each phase of the stator coil winding according to the signal duty cycle.

[0038] In one embodiment of the present invention, the H inverter bridge drive circuit includes a current sensor, which is used to acquire the current of the stator coil winding and generate a feedback signal.

[0039] In one embodiment of the present invention, the control module includes a filter module, which includes a cascaded architecture of a three-stage integrator and a three-stage differential.

[0040] Thirdly, to solve the above-mentioned technical problems, the present invention provides an electronic device, including a processor and a memory, wherein the memory stores computer-readable instructions, and when the computer-readable instructions are executed by the processor, the steps in the above-mentioned method are performed.

[0041] Compared with the prior art, the above-described technical solution of the present invention has the following advantages:

[0042] (1) The independent winding linear motor control method, system, and electronic equipment described in this invention can achieve accurate real-time acquisition and efficient processing of multiple sets of three-phase currents. This invention provides reliable and accurate data support for subsequent control by monitoring the current status. At the same time, this invention designs a proportional-integral (PI) control method with parameter self-tuning function. This control method can dynamically adjust the control parameters according to the real-time operating status of the system, effectively solving key problems such as low voltage utilization, insufficient output, and severe heat generation under common-mode interference and high-order harmonic phenomena, thereby avoiding the spectral inaccuracy problem caused by speed fluctuations due to sensor errors in proportional resonant (PR) controllers at high sampling rates.

[0043] (2) The multi-inverter collaborative modulation strategy designed in this invention includes a switching timing optimization method and zero-sequence voltage injection logic. This strategy ensures efficient coordination among the inverters and improves the overall efficiency of the system.

[0044] (3) This invention designs the architecture of the entire closed-loop system and develops a full-operating-domain adaptive control strategy. This strategy enables the system to maintain optimal performance under various operating conditions such as low speed, high speed, light load, and heavy load, ensuring the efficient and stable operation of the system. Attached Figure Description

[0045] To make the content of this invention easier to understand, the invention will be further described in detail below with reference to specific embodiments and accompanying drawings.

[0046] Figure 1 This is a flowchart of a control method for an independent winding linear motor according to a preferred embodiment of the present invention;

[0047] Figure 2 This is a schematic diagram of the H-type inverter bridge drive circuit in a preferred embodiment of the present invention;

[0048] Figure 3 This is a schematic diagram of the stator coil winding control circuit in a preferred embodiment of the present invention;

[0049] Figure 4 This is a flowchart of the magnetic field orientation control for each vector control combination in a preferred embodiment of the present invention;

[0050] Figure 5 This is a schematic diagram of vector PI control in a preferred embodiment of the present invention. Detailed Implementation

[0051] The present invention will be further described below with reference to the accompanying drawings and specific embodiments, so that those skilled in the art can better understand and implement the present invention. However, the embodiments described are not intended to limit the present invention.

[0052] Example 1:

[0053] Reference Figure 1 As shown, this embodiment of the invention provides a control method for an independent winding linear motor, including but not limited to the following steps:

[0054] S1. The moving part is driven by the driving force generated by the current excitation of the stator coil winding; the moving part's position is detected in real time, and a position feedback signal is generated.

[0055] S2. Based on the position feedback signal, perform gating control on the stator coil winding and dynamically divide the stator coil winding into multiple vector control combinations, and synchronously acquire the three-phase current signal of each vector control combination.

[0056] S3. Perform sign-number addition on each three-phase current signal to obtain the three-phase current vector sum; obtain the zero-sequence current based on the three-phase current vector sum; perform closed-loop control on the zero-sequence current to generate voltage compensation quantity;

[0057] S4. Perform field-oriented control on each vector control combination to generate a three-phase voltage control signal; superimpose the voltage compensation amount and the three-phase voltage control signal to obtain the control signal;

[0058] S5. Based on the control signal, control the magnitude and direction of the current in the stator coil winding to complete the independent closed-loop control of the current in each phase of the stator coil winding.

[0059] This invention discloses a control method for an independent winding linear motor, which improves the motion control accuracy and dynamic response capability of the linear motor system. First, the stator winding current generates a driving force to propel the mover, while simultaneously detecting the mover's position in real time and generating position feedback signals. These feedback signals provide accurate information about the mover's current position. Based on the position feedback signals, the stator winding is gating control, dynamically dividing it into multiple vector control combinations. This step allows for the synchronous acquisition of the three-phase current signals for each vector control combination, thereby controlling the motor's operation more precisely. By performing sign-number addition on each three-phase current signal, the three-phase current vector sum is obtained, leading to the zero-sequence current. Closed-loop control of the zero-sequence current generates a voltage compensation amount, which helps suppress the zero-sequence current, reduce the motor's common-mode voltage, and improve the system's stability and reliability. Furthermore, field-oriented control is performed on each vector control combination to generate three-phase voltage control signals. The voltage compensation amount and the three-phase voltage control signals are superimposed to obtain the final control signal. This control signal is used to control the magnitude and direction of the current in the stator windings, completing independent closed-loop control of the current in each phase of the stator windings. This independent closed-loop control makes the control of current magnitude and direction more precise, which helps to improve the operating efficiency of the motor and the stability of torque output. Compared with the common proportional resonant (PR) control scheme, this method avoids the spectral inaccuracy problem caused by speed fluctuations due to sensor errors in the PR controller at high sampling rates by adopting a multi-closed-loop control architecture. Through the proportional-integral (PI) optimization compensation algorithm, key problems such as low voltage utilization, insufficient output, and severe heat generation under common-mode interference and high-order harmonic phenomena are effectively solved.

[0060] Specifically, this embodiment of the invention includes a mover drive module, which comprises a mover and a permanent magnet array. In step S1, the permanent magnet array generates electromagnetic driving force under the current excitation of multiple stator coil windings in the stator winding module, driving the trolley of the mover drive module to move smoothly along the guide rail via guide rollers. Furthermore, in this embodiment of the invention, an encoder is used to detect the movement position of the mover in real time and generate an output position feedback signal to provide a position reference for subsequent control.

[0061] Furthermore, the stator-side controlled coil winding adopts an independent winding architecture, with each phase winding Lr connected one-to-one with the drive circuit of the motor driver (i.e., the H-type inverter bridge). There is no electrical coupling between the phase windings, thus achieving independent driving.

[0062] The embodiments of the present invention employ a linear motor with a moving secondary permanent magnet structure. By laying multiple independent coil units throughout the entire motion stroke, high thrust density, high dynamic response, and micron-level positioning accuracy are achieved.

[0063] Furthermore, referring to Figure 2 As shown, in this embodiment, the H-type inverter bridge includes four MOSFET devices (represented as Q1, Q2, Q3, and Q4, respectively) and at least one current sensor. This current sensor collects the winding current in real time and outputs a detection signal I_fd to the control unit of the motor driver, forming a current closed-loop control circuit. The motor driver achieves independent control of the current amplitude and phase in the corresponding coil windings by precisely adjusting the duty cycle of the pulse width modulation (PWM) signals of the four MOSFET devices, ultimately achieving flexible and controllable control of the current amplitude and phase of any winding.

[0064] Furthermore, the control circuit for the stator coil windings, such as Figure 3 As shown, the control circuit includes 12 MOSFET devices (Q1~Q12) in a three-phase full-bridge inverter structure. The three-phase bridge includes Q1 / Q2 / Q9 / Q10, Q3 / Q4 / Q11 / Q12, and Q5 / Q6 / Q7 / Q8; each bridge arm (such as Q1 and Q2) is a complementary switching structure with the upper and lower MOSFETs, and its conduction / turn-off is controlled by a PWM signal to invert DC voltage into AC voltage output. By controlling the PWM switching sequence of Q1~Q12, this circuit can output two independent sets of three-phase AC power to drive a dual three-phase winding motor.

[0065] Specifically, for step S2, based on the position feedback signal from the encoder, gating control of the stator coil windings is performed, followed by optimized selection of the vector combination. The specific steps are as follows:

[0066] S210. Based on the number of poles and winding distribution of the motor, map the position feedback signal to the gating control logic of the stator coil windings. For example, for a three-phase motor, each electrical cycle (360° electrical angle) may correspond to multiple physical position intervals, and each interval corresponds to a specific set of coil winding on / off states.

[0067] S220. Dynamically divide a preset number (e.g., 6 to 12) of stator coil windings into multiple vector control combinations (e.g., 2 to 4). Each vector control combination contains several coils, which form an independent control unit through a specific connection method. The number and method of division are optimized according to the specific structure of the motor and control requirements to ensure that each vector control combination can work independently and efficiently.

[0068] S230. Synchronously acquire the three-phase current signals of each vector control combination. For the synchronous acquisition method, a cascaded architecture of a three-stage integrator and a three-stage differential converter is used to acquire the three-phase current signals of each vector control combination in real time and synchronously. This architecture can effectively filter out high-frequency noise, improve signal stability and accuracy, and ensure that the acquired current signals can truly reflect the operating status of each vector control combination. The specific steps of this process are as follows:

[0069] S231, the three-stage integrator includes the first integrator. Second integrator and the third integrator The recursive expression for each integrator is:

[0070] ;

[0071] ;

[0072] ;

[0073] in, This represents the input signal at the current sampling moment, i.e., the original sampled value of the three-phase current signal for each vector control combination; This represents each sampling time.

[0074] S232, a three-stage differential circuit including a first differential circuit. Second differential and the third differential The recursive expression for each differencer is:

[0075] ;

[0076] ;

[0077] ;

[0078] in, The over-sampling ratio (OSR) is preferred in this embodiment to be 128. This design further filters out high-frequency noise through step-by-step differential operations, ensuring signal smoothness and accuracy.

[0079] S233, According to the third differential... The output results yield the three-phase current signal for each vector control combination. The calculation formula is:

[0080] ;

[0081] in, This indicates a floor operation, ensuring the result is an integer. The result is guaranteed to be within the range of a 16-bit signed integer to avoid data overflow, while retaining sufficient precision to reflect the actual changes in the current signal.

[0082] Furthermore, based on step S230, this embodiment of the invention designs an integrator-comb (CIC) filter module. This module adopts a three-stage cascaded integrator-comb structure design, which can efficiently process the downsampling of the Sigma-Delta modulator output signal. The advantages of the CIC filter module are mainly reflected in the following four aspects:

[0083] First, this module employs a three-stage integrator ( , , ) and three-stage differential ( , , The architecture employs a cascaded structure. At an oversampling rate (OSR) of 128x, it maintains 27-bit cumulative arithmetic precision, representing a 6-fold increase in effective bits compared to traditional single-stage structures. This high-precision architecture significantly improves the accuracy and reliability of signal processing.

[0084] Secondly, the CIC filter module features a timing control mechanism. Through the flag signal chain (flag_d1, flag_d2, flag_d3), this mechanism achieves precise phase synchronization, ensuring strict alignment between the downsampling process and the data acquisition window. This feature is crucial for maintaining signal integrity and accuracy.

[0085] Furthermore, this module features a dynamic reset function triggered by the sample_start signal, coupled with a pipelined processing structure. This design enables the system to maintain stable filtering characteristics even in bursty operating modes, and reduces the transition time to only 3 clock cycles, significantly improving the system's response speed and stability.

[0086] Finally, the CIC filter module implements intelligent output enable control through a multi-level state machine. The accurate generation of data validity flags (data_valid, data_valid_single) effectively eliminates metastability issues, ensuring the stability and reliability of data output and further improving the overall system performance.

[0087] Specifically, for step S3, based on the three-phase current signals acquired in step S2, a sign-number addition calculation is performed in real time to obtain the three-phase current vector sum. Based on the three-phase current vector sum, proportional-integral (PI) control is applied to the zero-sequence current. The specific steps are as follows:

[0088] S310. The three-phase current signal acquired in step S2 Perform sign-number addition to obtain the three-phase current vector sum. Among them, the three-phase current signal include and The specific calculation formula is as follows:

[0089] .

[0090] This vector sum reflects the overall state of the three-phase currents, providing fundamental data for subsequent zero-sequence current control.

[0091] S320, Based on the calculated three-phase current vector sum This refers to the zero-sequence current. Closed-loop control is applied to the zero-sequence current to generate voltage compensation. Specifically, a variable proportional PI control is used to handle the zero-sequence current, with the proportional coefficient depending on the magnitude of the zero-sequence current and... ( The shaft current is dynamically adjusted. This control strategy can effectively compensate for the effects of back electromotive force and end effect under high acceleration and deceleration, and the actual values ​​of its parameters are determined based on experimental experience.

[0092] Furthermore, the process of closed-loop control of the zero-sequence current and generation of voltage compensation involves transforming the zero-sequence current using a transformation matrix. Taking the zero-sequence current of a combination of four vector control methods as an example, the expression for the transformation matrix is:

[0093] ;

[0094] in, , , and These represent the zero-sequence currents of the four vector control combinations, respectively. , , , respectively represent the first Three-phase current of a vector control combination .

[0095] Furthermore, the expression for the voltage compensation amount is:

[0096] ;

[0097] in, , , and These represent the corresponding zero-sequence voltage compensation amounts; , , and These represent the zero-sequence currents of the four vector control combinations, respectively. , , and These represent the integral gains of the four vector control combinations; Represents the complex frequency variable in the Laplace transform. Indicated based on zero-sequence current and Dynamically adjustable proportional gain; Indicated based on zero-sequence current and Dynamically adjustable proportional gain; Indicated based on zero-sequence current and Dynamically adjustable proportional gain; Indicated based on zero-sequence current and Dynamically adjustable proportional gain.

[0098] Specifically, in step S4, the field-programmable gate array (FPGA) will synchronously execute multiple sets of field-oriented control (FOC). For each vector control combination, the FPGA performs field-oriented control, generating three-phase voltage control signals. The operation steps of a single FOC control are illustrated below: Three-phase voltage control signals in a vector control combination The expression is:

[0099] ;

[0100] in, This indicates Space Vector Pulse Width Modulation (SVPWM). This represents the inverse Park transform. This indicates current loop PI control. This represents the Park transform. This represents the Clarke transform. Indicates the first The three-phase current signal vector in a vector control combination Specifically .

[0101] Furthermore, referring to Figure 4 As shown, Figure 4This demonstrates a DSP (Digital Controller)-based FOC architecture. Three-phase current is acquired via a CIC filter module, converted to DQ-axis current through Clark / Park coordinate transformation, compared with a given value, and then output as a voltage command via a PI controller. This command is then converted into a PWM signal to drive a voltage source inverter (VSI) through inverse Park transformation (INPark) and SVPWM modulation (non-saddle wave modulation). Simultaneously, an encoder provides electrical angle feedback for field orientation, and zero-sequence current compensation and non-saddle wave modulation are introduced to optimize control performance. This architecture supports high-precision servo drives for 6 / 9 / 12 motor configurations. For the... The specific steps for generating three-phase voltage control signals using a vector control combination and FPGA-executed field-oriented control are as follows:

[0102] S410. Obtain the electrical angle from the encoder. Within each FOC execution cycle, compensate for the delay lag of the electrical angle to obtain a new electrical angle. The corresponding sine and cosine values ​​are calculated. Furthermore, the stator current is measured in real time during motor operation, typically acquiring 6 to 12 current signals. These current signals reflect the motor's operating state at different phases, providing fundamental data for precise control.

[0103] S420, Obtain the three-phase current in each vector control combination. .

[0104] S430, convert the three-phase current The first current is converted into a two-phase stationary coordinate system using Clark transformation. Second current .

[0105] S440, the first current in the two-phase stationary coordinate system Second current The third current in the rotating coordinate system is obtained through the Park transformation. and the fourth current The specific expression is:

[0106] ;

[0107] in, This represents the zero-sequence component, which is usually neglected in a balanced system. The electrical angle of the motor is obtained through the electrical angle compensation step S410.

[0108] S450, Calculate the error between the actual and reference values ​​of the D-axis and Q-axis currents. Obtain the third current respectively. The first error between the actual value and the reference value The fourth current The second error between the actual value and the reference value The specific expression is:

[0109] ;

[0110] ;

[0111] in, and These represent the reference values ​​for the D-axis and Q-axis currents, respectively. This indicates the current time point. The reference value of the D-axis current determines the magnetic flux of the motor rotor, while the reference value of the Q-axis current determines the magnitude of the motor's torque output.

[0112] Furthermore, the first error Second error As the input to the current loop PI controller, the first voltage vector is calculated through PI control. Second voltage vector First voltage vector Second voltage vector This refers to the voltage vector to be applied to the motor windings.

[0113] Furthermore, referring to Figure 5 As shown, based on the first error Second error The first voltage vector is calculated through PI control. Second voltage vector The specific steps are as follows:

[0114] S451. Calculate the proportional control output, the specific expression is:

[0115] ;

[0116] ;

[0117] in, This represents the proportional gain in D-axis current control. This represents the proportional gain in Q-axis current control.

[0118] S452. Calculate the integral control output, the specific expression is:

[0119] ;

[0120] ;

[0121] in, This represents the integral gain in D-axis current control. This represents the integral gain in Q-axis current control. This represents the integral variable.

[0122] S453. Calculate the anti-saturation compensation output. The specific expression is as follows:

[0123] ;

[0124] ;

[0125] in, Represents a saturation function. This represents the D-axis compensation scaling factor. This represents the Q-axis compensation ratio coefficient.

[0126] S454. Calculate the final control output. The specific expression is as follows:

[0127] ;

[0128] .

[0129] S455. Limit the output to ensure it remains within the allowable range. The expression is:

[0130] ;

[0131] in, Indicates control output. ; Indicates the maximum allowed output value. This indicates the minimum allowed output value.

[0132] Furthermore, based on the final control output The first voltage vector is obtained. Second voltage vector .

[0133] S460, the first voltage vector Second voltage vector The first orthogonal voltage value is obtained by inverse Park transformation onto the two-phase stationary coordinate system. Second orthogonal voltage value The specific expression is:

[0134] ;

[0135] in, It represents the zero-sequence voltage component.

[0136] S470, for the first quadrature voltage value Second orthogonal voltage value Space vector pulse width modulation (SVPWM) control is performed, including determining which sector the synthesized voltage vector is located in, calculating the conduction time of each bridge arm switch in the three phases, and finally outputting the three-phase voltage control signal required by the motor through the three-phase inverter drive module. The specific steps are as follows:

[0137] S471, Determine the first quadrature voltage value Second orthogonal voltage value The sector in question.

[0138] S472. Calculate the duration of the fundamental vector corresponding to each sector (using symbols). , (e.g., ), and calculate the total duration of action.

[0139] S473. Determine whether the total action time exceeds the modulation period. If so, reduce the action time by a preset ratio to obtain a new action time (using symbols). , (e.g., indicating the opposite); otherwise, no reduction processing is performed, and the action time is the new action time.

[0140] S474. Calculate the three-phase voltage switching time point based on the new action time, and generate the three-phase voltage control signal.

[0141] Furthermore, the following exemplary steps are given for steps S471 to S474:

[0142] Step 1: Determine the first quadrature voltage value Second orthogonal voltage value The specific calculation formula for the sector is as follows:

[0143] ;

[0144] ;

[0145] ;

[0146] ;

[0147] Among them, when ,but ,otherwise ;when ,but ,otherwise ;when ,but ,otherwise .also, , and Represents binary bits. Indicates a sector.

[0148] Step 2: Set three parameters , , Its expression is:

[0149] ;

[0150] ;

[0151] ;

[0152] in, Indicates the sampling period; This indicates the bus voltage.

[0153] For example, the durations of T0 (T7), T4, and T6 in each sector are obtained, as shown in Table 1. and This indicates the duration of action of the basic vector corresponding to the sector.

[0154] Table 1:

[0155]

[0156] Step 3: Determine whether the total duration of action exceeds the modulation period. The action time is reduced to obtain a new action time. Specifically, if... Then these two time values ​​need to be adjusted to ensure that their sum does not exceed the sampling period. The specific adjustment method is as follows:

[0157] .

[0158] Step 4: Calculate the three-phase voltage switching time based on the new operating time. Specifically, based on the adjusted... and Calculate the three-phase voltage switching time point , and The formula for calculating the three-phase voltage switching time is as follows:

[0159] ;

[0160] in, This indicates the duration of the zero-voltage vector.

[0161] Step 5: Based on the calculated three-phase voltage switching time points , and The switching states of the three-phase inverter can be determined by analyzing the relationships between these relationships and the various sectors. The specific relationships are shown in Table 2.

[0162] Table 2:

[0163]

[0164] By following the steps above, it can be ensured that, under any circumstances, the sum of the three-phase voltage switching times does not exceed the sampling period. And correctly control the switching state of the inverter according to the sector it belongs to.

[0165] Specifically, step S4 superimposes the voltage compensation amount and the three-phase voltage control signal (i.e., the three-phase voltage control signal obtained based on the three-phase voltage switching time point) to obtain the final control signal. The following are exemplary steps for this process:

[0166] For example, in order to improve the accuracy of voltage compensation, it is necessary to... , and The corresponding compensation voltages are superimposed at these three time points. Based on the voltage compensation amounts mentioned above and Table 2, the compensated voltage vector is obtained. Its expression is:

[0167] .

[0168] Similarly, for all The expression for the compensated voltage vector in the vector control combination is:

[0169] ;

[0170] in, , The symbol for the transpose of a matrix; Indicates the first Group vector control combined compensation voltage vector; Indicates the first The group vector control combines the original voltage vectors (k=1,2,3,4).

[0171] Specifically, in step S5, the duty cycle of the PWM signal of the MOSFET device is controlled according to the control signal, thereby realizing independent closed-loop control of the current of each phase winding.

[0172] This invention improves the performance of an independent winding permanent magnet synchronous linear motor system through a multi-closed-loop control architecture and intelligent modulation strategy. Its core advantages are reflected in the following four aspects:

[0173] First, distributed zero-sequence current parallel detection technology is adopted to achieve real-time synchronous acquisition and processing of multiple sets of three-phase currents. This technology can efficiently monitor the current status and provide accurate data support for subsequent control.

[0174] Secondly, a dynamic parameter self-tuning PI controller was developed, which maintains a stable zero-sequence current suppression rate by adjusting control parameters in real time. This controller also effectively solves the problem of spectrum inaccuracy in traditional PR controllers during sudden speed changes, thus improving the dynamic response capability of the system.

[0175] Third, the multi-inverter collaborative modulation algorithm optimizes the switching timing and zero-sequence voltage injection phase, enabling efficient collaborative operation among inverters. This algorithm significantly improves the overall efficiency and stability of the system.

[0176] Fourth, the full-condition adaptive control strategy ensures stable operation of the system in low-speed and overload conditions. This strategy can automatically adjust control parameters according to different operating states, keeping the system in optimal operating condition at all times.

[0177] This invention proposes a multi-closed-loop control strategy for real-time monitoring of the vector sum of three-phase currents to accurately extract the zero-sequence component. This strategy utilizes a PI controller to generate corresponding compensation voltages, which are then directly superimposed onto the SVPWM modulation signal. This design cleverly leverages the fast response capability of the current loop to compensate for potential deficiencies in position detection. In applications with high common-mode interference, high sampling rates, and high control frequencies, this method exhibits significant advantages over traditional proportional-resonant (PR) control methods. This method avoids the sensitivity of the PR controller to specific frequencies, thereby reducing the impact of speed fluctuations. Furthermore, this method maintains stable zero-sequence suppression over a wide frequency range, improving the system's anti-interference capability. These characteristics make this method particularly suitable for precision motion control applications with extremely high requirements for dynamic response and anti-interference capabilities. In summary, this method not only overcomes the shortcomings of traditional PR controllers but also significantly improves the dynamic performance and stability of the system, providing an efficient and reliable solution for precision motion control.

[0178] Example 2:

[0179] Based on the same inventive concept, this embodiment provides an independent winding linear motor control system. The principle of solving the problem is similar to that of the independent winding linear motor control method provided in Embodiment 1, and the repeated parts will not be described again.

[0180] This embodiment provides an independent winding linear motor control system for implementing the independent winding linear motor control method described in Embodiment 1, including:

[0181] The stator winding module is equipped with multiple stator coil windings;

[0182] The mover drive module includes a mover and a permanent magnet array; the permanent magnet array uses the current excitation of the stator coil winding to generate a driving force to drive the mover.

[0183] The motor drive module includes multiple H-inverter bridge drive circuits, each of which includes multiple MOSFET devices; each phase stator coil winding is connected to the H-inverter bridge drive circuit.

[0184] The position detection module is used to detect the movement position of the mover drive module and generate a position feedback signal;

[0185] The control module is used to control the signal duty cycle of the MOSFET devices based on the position feedback signal and the stator coil winding current feedback signal; and to perform independent closed-loop control of the winding current of each phase stator coil winding based on the signal duty cycle.

[0186] Specifically, the H inverter bridge drive circuit includes a current sensor, which is used to acquire the current of the stator coil winding and generate a feedback signal.

[0187] Specifically, the position detection module includes an encoder, which is used to detect the movement position of the mover drive module.

[0188] Specifically, the control module includes a control unit for receiving position feedback signals from the encoder and winding current (I_f / d) feedback signals from the current sensor.

[0189] Furthermore, the control module also includes a filter module and a zero-sequence current calculation module. The filter module comprises a cascaded architecture of a three-stage integrator and a three-stage differential. The zero-sequence current calculation module is used to calculate the zero-sequence current in each vector control combination.

[0190] Compared to traditional PR control schemes, this embodiment achieves precise closed-loop control of zero-sequence current, making it more suitable for systems with high sampling rates and wide speed ranges. This provides a breakthrough solution for the field of high-precision motor drives, especially in applications requiring high-precision linear motion control, such as precision manufacturing, automated equipment, and robotics. This embodiment significantly improves bus voltage utilization, enhances system temperature rise performance, strengthens output thrust stability, and achieves high-precision suppression of zero-sequence current across a wide speed range and all operating conditions, providing a breakthrough solution for high-precision linear motion control.

[0191] This embodiment provides an active zero-sequence current suppression system for an independent winding permanent magnet synchronous linear motor. The system employs a multi-closed-loop control architecture to effectively suppress zero-sequence current. The specific implementation process includes: real-time acquisition of multiple sets of three-phase output currents, accurate calculation of the zero-sequence component of each module, and comparison with a preset zero reference value. Based on the comparison result, the PI controller generates a corresponding compensation voltage, which is then used by the modulation module to generate a coordinated signal to drive the power converter. This process not only improves the system's response speed but also enhances its ability to suppress current fluctuations. The hardware implementation structure and real-time algorithm of the zero-sequence current calculation module in the control module ensure the accuracy and real-time performance of the calculation. This embodiment designs a zero-sequence current controller with parameter self-tuning function, particularly its variable gain adjustment mechanism, which can dynamically adjust control parameters according to the actual operating state of the system, improving the flexibility and adaptability of the control. A multi-inverter coordinated modulation strategy, including a switching timing optimization method and zero-sequence voltage injection logic, ensures coordinated operation between the inverters and improves the overall efficiency of the system. The architecture design of the entire closed-loop system and its adaptive control method in the entire operating domain (low speed / high speed, light load / heavy load) enable the system to maintain optimal performance under different operating conditions.

[0192] Example 3:

[0193] This embodiment provides an electronic device, including a processor and a memory. The memory stores computer-readable instructions. When the computer-readable instructions are executed by the processor, the steps in the method described in Embodiment 1 are performed.

[0194] Example 4:

[0195] This embodiment provides a motor, including the independent winding linear motor control system described in Embodiment 2. The motor can be a high-precision linear motor or a servo motor.

[0196] Those skilled in the art will understand that embodiments of this application can be provided as methods, systems, or computer program products. Therefore, this application can take the form of a completely hardware embodiment, a completely software embodiment, or an embodiment combining software and hardware aspects. Furthermore, this application can take the form of a computer program product embodied on one or more computer-usable storage media (including but not limited to disk storage, CD-ROM, optical storage, etc.) containing computer-usable program code.

[0197] This application is described with reference to flowchart illustrations and / or block diagrams of methods, apparatus (systems), and computer program products according to embodiments of this application. It will be understood that each block of the flowchart illustrations and / or block diagrams, and combinations of blocks in the flowchart illustrations and / or block diagrams, can be implemented by computer program instructions. These computer program instructions can be provided to a processor of a general-purpose computer, special-purpose computer, embedded processor, or other programmable data processing apparatus to produce a machine, such that the instructions, which execute via the processor of the computer or other programmable data processing apparatus, generate instructions for implementing the flowchart... Figure 1 One or more processes and / or boxes Figure 1 A device that provides the functions specified in one or more boxes.

[0198] These computer program instructions may also be stored in a computer-readable storage medium that can direct a computer or other programmable data processing device to function in a particular manner, such that the instructions stored in the computer-readable storage medium produce an article of manufacture including instruction means, which are implemented in a process Figure 1 One or more processes and / or boxes Figure 1 The function specified in one or more boxes.

[0199] These computer program instructions may also be loaded onto a computer or other programmable data processing equipment to cause a series of operational steps to be performed on the computer or other programmable equipment to produce a computer-implemented process, thereby providing instructions that execute on the computer or other programmable equipment for implementing the process. Figure 1 One or more processes and / or boxes Figure 1 The steps of the function specified in one or more boxes.

[0200] Obviously, the above embodiments are merely illustrative examples for clear explanation and are not intended to limit the implementation. Those skilled in the art will recognize that other variations or modifications can be made based on the above description. It is neither necessary nor possible to exhaustively list all possible implementations here. However, obvious variations or modifications derived therefrom are still within the scope of protection of this invention.

Claims

1. A control method for an independent winding linear motor, characterized in that, include: S1. The moving part is driven by the driving force generated by the current excitation of the stator coil winding. The motion position of the moving part is detected in real time, and a position feedback signal is generated; S2. Based on the position feedback signal, perform gating control on the stator coil winding, and dynamically divide the stator coil winding into multiple vector control combinations, and synchronously acquire the three-phase current signal of each vector control combination; S3. Perform sign-number addition on each of the three-phase current signals to obtain the three-phase current vector sum; obtain the zero-sequence current based on the three-phase current vector sum; perform closed-loop control on the zero-sequence current to generate a voltage compensation quantity; wherein, the expression for the voltage compensation quantity is: ； in, , , and This indicates the corresponding zero-sequence voltage compensation amount; , , and These represent the zero-sequence currents of the four vector control combinations, respectively. , , and These represent the integral gains of the four vector control combinations; Represents the complex frequency variable in the Laplace transform; Indicated based on zero-sequence current and Dynamically adjustable proportional gain; Indicated based on zero-sequence current and Dynamically adjustable proportional gain; Indicated based on zero-sequence current and Dynamically adjustable proportional gain; Indicated based on zero-sequence current and Dynamically adjustable proportional gain; S4. Perform field-oriented control on each of the vector control combinations to generate a three-phase voltage control signal; superimpose the voltage compensation amount and the three-phase voltage control signal to obtain a control signal; S5. According to the control signal, control the magnitude and direction of the current in the stator coil winding to complete the independent closed-loop control of the current in each phase stator coil winding.

2. The independent winding linear motor control method according to claim 1, characterized in that, In step S3, the zero-sequence current is subjected to closed-loop control. The process of generating the voltage compensation quantity includes transforming the zero-sequence current using a transformation matrix, wherein the expression of the transformation matrix is: ； in, , , and These represent the zero-sequence currents of the four vector control combinations, respectively. , and They represent the first Three-phase current of a vector control combination.

3. The independent winding linear motor control method according to claim 1, characterized in that, The method for synchronously acquiring the three-phase current signal of each vector control combination in S2 is as follows: a three-stage integrator and a three-stage differential converter are cascaded to synchronously acquire the three-phase current signal of each vector control combination.

4. The independent winding linear motor control method according to claim 1, characterized in that... The step S4, which involves performing field-oriented control on each vector control combination to generate a three-phase voltage control signal, is as follows: Obtain the three-phase current in each of the vector control combinations; The three-phase current is converted into a first current and a second current by Clark; The first current and the second current are transformed using Park to obtain the third current and the fourth current in the rotating coordinate system; The first error between the actual value and the reference value of the third current and the second error between the actual value and the reference value of the fourth current are obtained respectively; based on the first error and the second error, the first voltage vector and the second voltage vector are calculated. The first voltage vector and the second voltage vector are transformed into a two-phase stationary coordinate system by Park inverse transformation to obtain the first orthogonal voltage value and the second orthogonal voltage value. SVPWM control is applied to the first quadrature voltage value and the second quadrature voltage value to obtain a three-phase voltage control signal.

5. The independent winding linear motor control method according to claim 4, characterized in that, The steps for obtaining three-phase voltage control signals by performing SVPWM control on the first quadrature voltage value and the second quadrature voltage value are as follows: Determine the sectors where the first orthogonal voltage value and the second orthogonal voltage value are located; Calculate the duration of action of the basic vector corresponding to each sector, and calculate the sum of the durations of the action times; Determine whether the total action time exceeds the modulation period. If so, reduce the action time by a preset ratio to obtain a new action time. Conversely, if no reduction is performed, the action time becomes the new action time. Based on the new operating time, the three-phase voltage switching time point is calculated, and a three-phase voltage control signal is generated.

6. A control system for an independent winding linear motor, used to implement the independent winding linear motor control method according to any one of claims 1 to 5, characterized in that, include: The stator winding module is equipped with multiple stator coil windings; The mover drive module includes a mover and a permanent magnet array; The permanent magnet array uses the current excitation of the stator coil windings to generate a driving force to propel the mover. The motor drive module includes multiple H-inverter bridge drive circuits, each of which includes multiple MOSFET devices; each phase of the stator coil winding is connected to the H-inverter bridge drive circuit. A position detection module is used to detect the movement position of the mover drive module and generate a position feedback signal; A control module is used to control the signal duty cycle of the MOSFET device based on the position feedback signal and the feedback signal of the stator coil winding current. Based on the signal duty cycle, the winding current of each phase of the stator coil winding is independently controlled in a closed loop.

7. A control system for an independent winding linear motor according to claim 6, characterized in that, The H inverter bridge drive circuit includes a current sensor, which is used to collect the current of the stator coil winding and generate a feedback signal.

8. A control system for an independent winding linear motor according to claim 6, characterized in that, The control module includes a filter module, which has a cascaded architecture of a three-stage integrator and a three-stage differential.

9. An electronic device, characterized in that, It includes a processor and a memory, the memory storing computer-readable instructions that, when executed by the processor, perform the steps of the method as described in any one of claims 1 to 5.

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