Frequency spectrum notch coded modulation method and system

Abstract

The invention relates to a frequency spectrum trapped wave coded modulation method and a frequency spectrum trapped wave coded modulation system. Comprising the following steps: dividing a symbol sequence mapped by an input bit sequence according to code blocks with the length of 1; constructing two groups of candidate symbol sequences for each code block to be coded, wherein the first group is an original sequence; the second group is an inverse sequence; setting identification symbols for each group of symbol sequences; the WRDS of each group of candidate symbol sequences is calculated; and selecting a group of candidate symbol sequences with smaller WRDS amplitude as final coding output of the current code block. According to the method, the power of a specific frequency point is suppressed by constraining the weighted operation digital sum of the signal sequence, so that the purpose of generating a low power point in the whole frequency spectrum is achieved, the original signal carrier still has the property of a single carrier, DSP (Digital Signal Processor) processing on a plurality of subcarriers is not needed, and the complexity of the system is reduced. According to the invention, the signal spectrum has a low power point in a coding mode, so that performance damage caused by narrowband filtering operation is avoided.

Description

Technical Field

[0001] This application relates to the field of optical signal processing technology, and more specifically, to a spectrum notch coding modulation method and system. Background Technology

[0002] With the rapid development of 6G networks, real-time high-definition video broadcasting, and cloud services, the application of coherent technology in short-distance optical interconnects will become a future trend to support high-capacity data transmission. However, the blind digital signal processing (DSP) algorithms of traditional coherent optical communication systems are highly complex. While frequency-domain pilot-based DSPs can simplify these algorithms, for single-carrier systems, frequency-domain pilots can only be inserted outside the signal band, and the attenuation of the system at high frequencies will affect the pilots.

[0003] In traditional single-carrier systems, frequency-domain pilots are typically inserted outside the signal bandwidth to avoid their influence on the signal while ensuring accurate pilot information extraction. However, out-of-band pilots are often highly dependent on system bandwidth, and high high-frequency attenuation necessitates a high pilot signal power ratio (PTSPR), affecting the quantization bit depth and leading to performance degradation. In multi-carrier systems, digital multicarrier multiplexing (DSCM) provides a guard interval between two subcarriers for pilot insertion, but each subcarrier has a lower baud rate and longer symbol duration compared to a single carrier, making the system more susceptible to laser phase noise. Furthermore, DSCM requires separate processing for each subcarrier in the receiver's DSP, adding further complexity. Alternatively, narrowband filters can be used to filter the signal, creating a dip in the signal spectrum, but this inevitably impacts signal performance. Therefore, to achieve pilot insertion within the signal band while maintaining single-carrier characteristics without introducing additional performance degradation, a spectral notch technique needs to be explored to generate low-power points in the spectrum for pilot insertion. Summary of the Invention

[0004] The purpose of this invention is to overcome the shortcomings of existing technologies, such as the easy impact of pilot insertion within the signal band on the properties of a single carrier and the complexity of the processing system. This invention provides a spectrum notch coding modulation method and system that can ensure that the original signal carrier still has the properties of a single carrier, effectively reducing the complexity of the system.

[0005] To solve the above-mentioned technical problems, the technical solution adopted by the present invention is as follows: A spectrum notch coded modulation method is provided, comprising the following steps: S1. Map the input bit sequence to the symbol sequence of length . l The code blocks are divided into blocks, each containing... l- One payload symbol and one identification symbol; S2. For each code block to be encoded, construct at least two sets of candidate symbol sequences. The first set is the original sequence with the original symbols unchanged; the second set is the inverse sequence of all symbols in the same code block after inverting the symbols. S3. Set an identifier for each group of symbol sequences; S4. For each candidate symbol sequence, calculate the weighted running numbers and WRDS respectively; S5. Select a set of candidate symbol sequences with smaller WRDS amplitudes as the final encoded output of the current code block.

[0006] This invention discloses a spectrum notch coding modulation method that suppresses power at specific frequency points by constraining the weighted running digital sum of a signal sequence. This achieves the goal of creating low-power points across the entire spectrum, allowing the original signal carrier to retain its single-carrier nature. This eliminates the need for DSP processing of multiple subcarriers, reducing system complexity. Furthermore, the encoding method can induce low-power points in the signal spectrum, thereby avoiding the performance degradation caused by narrowband filtering operations.

[0007] Further, in step S4, after calculating the WRDS of each group of candidate symbol sequences, the calculated WRDS result of each group is added to the cumulative WRDS value of all previously encoded code blocks to obtain the cumulative WRDS value corresponding to each group of candidate sequences; then, the group with the smaller cumulative WRDS value is selected as the final encoded output of the current code block.

[0008] Furthermore, the WRDS is calculated using the following formula:

[0009] In the formula, This represents the symbol sequence mapped from the input bit sequence. For the specified normalized frequency, k Indicates the previous k WRDS is calculated for each symbol sequence. j The imaginary unit, Indicates the symbol sequence index.

[0010] Further, in step S3, when adding an identifier to each code block: firstly, arbitrarily specify one in-phase component from the preset identifiers as the initial identifier; then, according to preset rules, assign different identifiers to the original sequence and the reverse sequence for each new code block. The identifiers distinguish states only through the in-phase components, and the quadrature components are not constrained. The preset identifiers include -A, +A, -B, and +B, where A and B are non-zero integers, and the Euclidean distance between -A and +B and between +A and -B is at least greater than 4. When the identifier of the previous code block is -A or -B, the identifier of the original sequence of the current code block is -A, and the identifier of the reverse sequence is +B. When the identifier of the previous code block is +A or +B, the identifier of the original sequence of the current code block is +A, and the identifier of the reverse sequence is -B.

[0011] Furthermore, during the decoding operation, based on the received identifier, the current code block is determined according to the same identifier selection rule as the encoded segment. If it is the original sequence, a hold operation is performed; if it is the reversed sequence, a reverse operation is performed.

[0012] The present invention also provides a spectrum notch coded modulation system, comprising: Division unit: used to divide the symbol sequence after mapping the input bit sequence into units of length 1. l The code blocks are divided into blocks, each containing... l - One payload symbol and one identification symbol; Candidate sequence construction unit: used to construct at least two sets of candidate symbol sequences for each code block to be encoded. The first set is the original sequence with the original symbols unchanged; the second set is the inverse sequence of all symbols in the same code block after inverting the symbols. Identifier addition unit: used to set an identifier for each group of symbol sequences; WRDS calculation unit: used to calculate the weighted running numbers and WRDS for each group of candidate symbol sequences; Output unit: Used to select a set of candidate symbol sequences with smaller WRDS amplitudes as the final encoded output of the current code block.

[0013] Furthermore, after calculating the WRDS of each group of candidate symbol sequences, the WRDS calculation unit adds the calculated WRDS result of each group to the cumulative WRDS value of all previously encoded code blocks to obtain the cumulative WRDS value corresponding to each group of candidate sequences.

[0014] Furthermore, when adding identifiers to each code block, the identifier adding unit first arbitrarily assigns an in-phase component from the preset identifiers as the initial identifier. Then, according to preset rules, it assigns different identifiers to the original sequence and the reverse sequence for each new code block. The identifiers are distinguished only by the in-phase component, and the orthogonal component is not constrained. The preset identifiers include -A, +A, -B, and +B, where A and B are non-zero integers, and the Euclidean distance between -A and +B and between +A and -B is at least greater than 4. When the identifier of the previous code block is -A or -B, the original sequence identifier of the current code block is -A, and the reverse sequence identifier is +B. When the identifier of the previous code block is +A or +B, the original sequence identifier of the current code block is +A, and the reverse sequence identifier is -B.

[0015] Furthermore, it also includes a decoding unit: used to decode the encoded code block. During the decoding operation, it determines whether the current code block has been inverted according to the received identifier and the same identifier selection rule as the encoded segment. If it is the original sequence, it performs a hold operation; if it is the reverse sequence, it performs a reverse operation.

[0016] The present invention also provides a digital signal processing method, comprising: During digital signal processing at the transmitter, the pseudo-random bit sequence is first mapped to 16QAM symbols to generate two 35 Gbuad 16QAM signals. Then, the signals are encoded using the spectrum notch coding modulation method described above, and the QAM symbols are shaped by the Nyquist filter using a root raised cosine filter. After resampling the signals, pilot signals are inserted to estimate the frequency shift and laser phase noise. During digital signal processing at the receiving end, dispersion compensation is first performed on the received signal. Then, pilot-assisted frequency offset estimation is used. Next, the pilot signal is down-converted to baseband and extracted through a low-pass filter to estimate phase noise. Subsequently, a cascaded multimode algorithm is used for polarization demultiplexing and to suppress inter-symbol interference. Then, a decision-oriented least mean square algorithm is used to eliminate the IQ offset introduced at the transmitting end and to compensate for residual phase noise. Finally, decoding, demapping, and bit error rate calculation are performed sequentially.

[0017] Compared with the prior art, the beneficial effects of the present invention are: This invention discloses a spectrum notch coding modulation method and system that suppresses power at specific frequency points by constraining the weighted running digital sum of a signal sequence, thereby achieving a low-power point across the entire spectrum. Compared to DSCM systems, this invention ensures that the original signal carrier retains its single-carrier nature, allowing pilot signals to be inserted within the signal spectrum without requiring DSP processing of multiple subcarriers, thus reducing system complexity. Furthermore, this invention achieves a low-power point in the signal spectrum through coding, thereby avoiding the performance degradation caused by narrowband filtering operations. Attached Figure Description

[0018] Figure 1 This is a schematic diagram of the spectrum notch coding modulation method in one embodiment; Figure 2 Here is an example diagram of spectral notch coded modulation in one embodiment; Figure 3 This is a schematic diagram of the decoding operation in one embodiment; Figure 4 This is a schematic diagram of a simulation system in one embodiment; Figure 5 This is a schematic diagram of the power spectral density of a 16QAM signal under different overhead and notch positions in one embodiment; Figure 6 This is a schematic diagram of phase noise and error values ​​using pilot estimation in one embodiment; Figure 7 This is a schematic diagram illustrating the bit error rate performance of a 35 Gbaud 16QAM signal at OBTB and 40 km in one embodiment. Figure 8 This is a schematic diagram illustrating the relationship between bit error rate and normalized spectrum notch bandwidth in one embodiment, as well as a comparison under different coding overheads; Figure 9 This is a schematic diagram of the bit error rate curve of a 35 Gbaud 16QAM signal using a DFB laser in one embodiment. Detailed Implementation

[0019] The present invention will be further described below with reference to specific embodiments. The accompanying drawings are for illustrative purposes only, representing schematic diagrams rather than actual physical objects, and should not be construed as limiting the invention. To better illustrate the embodiments of the invention, some components in the drawings may be omitted, enlarged, or reduced, and do not represent the actual dimensions of the product. It is understandable to those skilled in the art that some well-known structures and their descriptions may be omitted in the drawings.

[0020] In the accompanying drawings of the embodiments of the present invention, the same or similar reference numerals correspond to the same or similar components. In the description of the present invention, it should be understood that if terms such as "upper," "lower," "left," "right," etc., indicate the orientation or positional relationship based on the orientation or positional relationship shown in the drawings, they are only for the convenience of describing the present invention and simplifying the description, and do not indicate or imply that the device or element referred to must have a specific orientation, or be constructed and operated in a specific orientation. Therefore, the terms used to describe positional relationships in the drawings are only for illustrative purposes and should not be construed as limiting the present invention. For those skilled in the art, the specific meaning of the above terms can be understood according to the specific circumstances.

[0021] Example 1 This embodiment is the first embodiment of the spectrum notch coding modulation method, which achieves spectrum notch filtering at any normalized frequency position by minimizing the weighted running digital sum (WRDS). Figure 1 and Figure 2 As shown, the specific steps include: S1. Map the input bit sequence to the symbol sequence of length . l The code blocks are divided into blocks, each containing... l - 1 payload symbol and 1 identifier symbol; the corresponding coding overhead is 1 / l .

[0022] S2. For each code block to be encoded, construct two sets of candidate symbol sequences. The first set is the original sequence with the original symbols unchanged; the second set is the inverse sequence of all inverted symbols within the same code block. For example... Figure 2 As shown in (a) of the diagram.

[0023] S3. Set an identifier for each group of symbol sequences; add the corresponding identifier to the end of each of the two groups of candidate symbol sequences to record whether the current code block uses the "original sequence" or the "reverse sequence".

[0024] S4. For each candidate symbol sequence, calculate the weighted running number and WRDS; then add the result to the cumulative WRDS of all previously encoded code blocks to obtain the corresponding cumulative WRDS value. WRDS is calculated using the following formula:

[0025] In the formula, This represents the symbol sequence mapped from the input bit sequence. For the specified normalized frequency, k Indicates the previous k WRDS is calculated for each symbol sequence. j The imaginary unit, Indicates the symbol sequence index.

[0026] S5. Select a set of candidate symbol sequences with smaller WRDS amplitudes as the final encoded output of the current code block. This suppresses the signal power at the specified frequency as much as possible throughout the transmission sequence, ensuring a stable dip at the target frequency.

[0027] like Figure 2 As shown in (b) of the figure, the selection criteria for the identifier symbols are given using a 16QAM signal as an example. In the figure, the arrows indicate the direction of transition of the in-phase component of the identifier symbol from the previous symbol to the current symbol. The identifier symbol selection rules are shown in Table 1 below: Table 1. Identifier Symbol Selection Rules

[0028] First, arbitrarily choose one of the following values—-3, -1, +1, +3—as the in-phase component of the initial identifier. Then, assign different identifiers to the original sequence and the reverse sequence for each new code block. The identifiers distinguish states only by the in-phase component; the quadrature components are unconstrained and can be randomly selected. For example, if the identifier of the previous code block has a value of -1 in the in-phase component, then in the current code block, if the original sequence is selected, the in-phase component of the identifier will still be -1; if the reverse sequence is selected, the in-phase component will be +3. Figure 2 As shown in (b) and Table 1, the current combination of in-phase components for the identifier is only (-1, +3) and (+1, -3). Compared to using only -1 and +1 as flags, the Euclidean distance is only half that of this embodiment, making it more prone to misjudgment. Furthermore, since each symbol in -3, -1, 1, and 3 appears the same number of times, the final encoded output symbol distribution exhibits the same probability distribution as the input sequence. That is, through this identifier construction method, on the one hand, the inversion state of the current code block can be explicitly recorded, thereby avoiding error propagation between code blocks; on the other hand, since identifiers are introduced only at a finite number of symbol positions, and according to the state transition diagram of the selection criterion, the selection probability of each flag is equal, thus not changing the probability distribution of the original input sequence symbols. This preserves the statistical characteristics of the input sequence while obtaining the required spectral notch characteristics.

[0029] In this embodiment, during the decoding operation, such as Figure 3 As shown. Simply select the identifier symbol according to the same rules as the encoding end, based on the received flag symbol (e.g., ...). Figure 2 As shown in (b) in the diagram, determine whether the current code block has been inverted, and perform a "hold / invert" operation on the corresponding symbol to restore the original code. lThe decoding process involves discarding one payload symbol and then removing the identifier symbols from each code block. If the identifier symbol of the previous code block is +1 on the in-phase component, then according to the state transition table, the identifier symbol state of the current code block can only be between +1 and -3. The distances of the in-phase component of the current code block's identifier symbol to both +1 and -3 are compared. The smaller distance is selected for a "hold / invert" operation. If the distance to +1 is smaller, "hold" is performed; if the distance to -3 is smaller, "invert" is performed. It is important to note that the constellation points obtained through this operation are still standard constellation points and do not introduce additional signal impairment.

[0030] This embodiment of a spectrum notch coding modulation method suppresses power at specific frequency points by constraining the weighted running digital sum of the signal sequence, thereby achieving the goal of creating low-power points throughout the spectrum. This allows the original signal carrier to retain the characteristics of a single carrier, eliminating the need for DSP processing of multiple subcarriers and reducing system complexity. Furthermore, the low-power points in the signal spectrum can be encoded to avoid the performance degradation caused by narrowband filtering operations.

[0031] This embodiment provides a spectral notch coding modulation method. Compared to multi-carrier systems such as DSCM, this embodiment utilizes coding technology to create a spectral notch in the signal spectrum for pilot insertion while maintaining a single carrier. Compared to single-carrier systems that insert pilots out of band, this embodiment can insert pilots within the signal band, reducing the pilot power requirements due to system bandwidth limitations. Compared to using narrowband filters, this embodiment generates spectral notches through coding, avoiding performance degradation caused by filtering effects. Furthermore, this embodiment uses a state transition diagram as the selection criterion for identifiers, balancing the needs of spectrum shaping, avoiding error propagation, and maintaining the original signal probability distribution.

[0032] Example 2 This embodiment is a second embodiment of a spectrum notch coding modulation method. This embodiment analyzes and verifies the method of the first embodiment through simulation and time delay.

[0033] The figure shows a schematic diagram of the simulation experimental system in this embodiment. At the transmitting end, a digital-to-analog converter (DAC) with a sampling rate of 80 GSa / s provides electrical signal drive for the dual-biased IQ modulator (DP-IQM). The optical carrier is generated by a laser operating at a wavelength of 1549.984 nm and then injected into the DP-IQM for signal modulation. The output optical signal is amplified by an erbium-doped fiber amplifier (EDFA) and then injected into a 40 km long optical fiber for transmission, with the transmit power set to 8 dBm. At the receiving end, a variable optical attenuator (VOA) is used to adjust the received optical power (ROP) to measure sensitivity. The received optical signal and the local oscillator (LO) are connected to an integrated coherent receiver (ICR) for coherent detection, and the output electrical signal is sampled by an 80 GSa / s analog-to-digital converter (ADC).

[0034] Transmitter Digital Signal Processing (DSP): In the transmitter DSP, the pseudo-random bit sequence is first mapped to 16QAM symbols to generate two 35Gbuad 16QAM signals. Then, the signals are encoded using the spectral notch coding modulation method provided in Example 1, and the QAM symbols are Nyquist shaped using a root-raised cosine filter with a roll-off factor of 0.1. After resampling the signals, pilot signals are inserted at 273.43MHz, where the pilot signal power ratio (PTSPR) is -26 dB, to estimate the frequency offset and laser phase noise.

[0035] Receiver DSP: At the receiver, dispersion compensation is first performed on the received signal. Then, pilot-assisted frequency offset estimation is used. The pilot signal is then down-converted to baseband and extracted using a low-pass filter to estimate phase noise. Subsequently, a cascaded multimode algorithm is used for polarization demultiplexing and to suppress inter-symbol interference. Then, a decision-guided least mean square algorithm is used to eliminate the IQ offset introduced at the transmitter and compensate for residual phase noise. Finally, decoding, demapping, and bit error rate (BER) calculation are performed sequentially.

[0036] Results Analysis: First, the spectral notch coding modulation method proposed in Example 1 was used to encode the 16QAM signal under different redundancy levels and normalization frequencies, and the power spectral density of the encoded signal was estimated. The results are as follows: Figure 5 As shown in the figure, the power spectrum curves of notch filtering at the normalized frequency of 0 are given under conditions of 5% and 10% redundancy overhead, respectively, and the power spectrum curve of notch filtering at the normalized frequency of -1 / 4 is given under condition of 10% redundancy overhead. It can be seen from the figure that, while maintaining the overall spectral shape essentially unchanged, this embodiment can generate a narrowband notch with a depth exceeding 20 dB at the target normalized frequency, and the depth and bandwidth of the notch can be flexibly controlled by adjusting the redundancy and constraint position.

[0037] based on Figure 4 The simulation system diagram shows that, firstly, numerical simulations were performed on pilot-based phase noise estimation, and the results are as follows. Figure 6 As shown, the influence of the common rotating phase is ignored. The results show that the error between the actual phase noise and the estimated value is very small, which verifies the feasibility of the notch filter generated by the present invention for inserting pilot signals.

[0038] For ease of comparison, this embodiment transmits a DSCM signal with a 0.5 GHz guard interval. Figure 7 Figures (a) and (b) show the BER performance versus ROP curves for a 35 Gbaud 16 QAM signal after optical back-to-back (OBTB) transmission and after transmission over 40 km of fiber, respectively. The performance is similar in both cases. In the receiver DSP, the BER performance using the blind phase search (BPS) algorithm and without the BPS algorithm was evaluated. The proposed spectral notch coding modulation method maintains the same performance at shorter symbol periods, indicating improved tolerance to phase noise. Without the BPS algorithm, the method of this invention achieves a 0.83 dB receive optical power gain compared to DSCM in 40 km fiber transmission, with a power budget of 35.59 dB, under a 15% soft decision error correction coding (SD-FEC) threshold of 1.0 × 10⁻². Furthermore, experimental results show that the BER curves with 10% overhead and 5% overhead almost completely overlap.

[0039] Figure 8 The relationship between BER performance and the normalized bandwidth of the notch filter as a function of overhead in 40 km fiber transmission is shown. Clearly, the bandwidth increases with increasing redundancy overhead. When the coding overhead is 1 / 64, the normalized notch filter 3dB and 6dB bandwidths remain at 0.0059 and 0.0029, respectively, corresponding to equivalent frequency bandwidths of 227.15 MHz and 111.65 MHz. With fixed overhead, only a small performance difference exists between the scheme using and not using the BPS algorithm.

[0040] Figure 9 This paper presents a comparison between a scheme using a 2 MHz linewidth distributed feedback (DFB) laser with a 5% overhead after 40 km transmission and a scheme using a DSCM, along with the corresponding electrical spectra. Experimental results show that the proposed scheme has stronger robustness to large linewidth lasers.

[0041] Example 3 This embodiment is an example of a spectrum notch coded modulation system, including: Division unit: used to divide the symbol sequence after mapping the input bit sequence into units of length 1.l The code blocks are divided into blocks, each containing... l - One payload symbol and one identification symbol; Candidate sequence construction unit: used to construct at least two sets of candidate symbol sequences for each code block to be encoded. The first set is the original sequence with the original symbols unchanged; the second set is the inverse sequence of all symbols in the same code block after inverting the symbols. Identifier addition unit: used to set an identifier for each group of symbol sequences; WRDS calculation unit: used to calculate the weighted running numbers and WRDS for each group of candidate symbol sequences; Output unit: Used to select a set of candidate symbol sequences with smaller WRDS amplitudes as the final encoded output of the current code block.

[0042] Furthermore, after calculating the WRDS of each group of candidate symbol sequences, the WRDS calculation unit adds the calculated WRDS result of each group to the cumulative WRDS value of all previously encoded code blocks to obtain the cumulative WRDS value corresponding to each group of candidate sequences.

[0043] Specifically, when adding identifiers to each code block, the identifier adding unit first arbitrarily assigns an in-phase component from the preset identifiers as the initial identifier. Then, according to preset rules, it assigns different identifiers to the original sequence and reverse sequence for each new code block. Identifiers distinguish states only through in-phase components; quadrature components are unconstrained. The preset identifiers include -A, +A, -B, and +B, where A and B are non-zero integers, and the Euclidean distance between -A and +B, and between +A and -B, is at least greater than 4. When the identifier of the previous code block is -A or -B, the original sequence identifier of the current code block is -A, and the reverse sequence identifier is +B. Similarly, when the identifier of the previous code block is +A or +B, the original sequence identifier of the current code block is +A, and the reverse sequence identifier is -B. Figure 2 As shown in (b) and Table 1, the selection rules for the identifiers in this embodiment are the same as those in Embodiment 1, and will not be repeated here.

[0044] It also includes a decoding unit: used to decode the encoded code blocks. During the decoding operation, based on the received identifier, it determines whether the current code block has been inverted according to the same identifier selection rule as the encoded segment. If it is the original sequence, a hold operation is performed; if it is the reversed sequence, a reverse operation is performed. Figure 3 As shown, in this embodiment, the decoding operation is the same as in Embodiment 1, and will not be described again here.

[0045] In the specific implementation of the above embodiments, the technical features can be combined in any non-contradictory way. For the sake of brevity, not all possible combinations of the above technical features are described. However, as long as the combination of these technical features is not contradictory, it should be considered to be within the scope of this specification.

[0046] Obviously, the above embodiments of the present invention are merely examples for clearly illustrating the present invention, and are not intended to limit the implementation of the present invention. Those skilled in the art can make other variations or modifications based on the above description. It is neither necessary nor possible to exhaustively describe all embodiments here. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included within the scope of protection of the claims of the present invention.

Claims

1. A spectral notch coded modulation method, characterized by, Includes the following steps: S1. Map the input bit sequence to the symbol sequence of length . l The code blocks are divided into blocks, each containing... l - One payload symbol and one identification symbol; S2. For each code block to be encoded, construct at least two sets of candidate symbol sequences. The first set is the original sequence with the original symbols unchanged; the second set is the inverse sequence of all symbols in the same code block after inverting the symbols. S3. Set an identifier for each group of symbol sequences; S4. For each candidate symbol sequence, calculate the weighted running numbers and WRDS respectively; S5. Select a set of candidate symbol sequences with smaller WRDS amplitudes as the final encoded output of the current code block.

2. The spectral notch coding modulation method according to claim 1, characterized in that, In step S4, after calculating the WRDS of each group of candidate symbol sequences, the calculated WRDS result of each group is added to the cumulative WRDS value of all previously encoded code blocks to obtain the cumulative WRDS value corresponding to each group of candidate sequences. Then, the group with the smaller cumulative WRDS value amplitude is selected as the final encoded output of the current code block.

3. The spectrum notch coding modulation method according to claim 2, characterized in that, The WRDS is calculated using the following formula: In the formula, This represents the symbol sequence mapped from the input bit sequence. For the specified normalized frequency, k Indicates the previous k WRDS is calculated for each symbol sequence. j The imaginary unit, Indicates the symbol sequence index.

4. The spectrum notch coding modulation method according to claim 2, characterized in that, In step S3, when adding an identifier to each code block: first, arbitrarily specify one in-phase component from the preset identifiers as the initial identifier; then, according to preset rules, assign different identifiers to the original sequence and the reverse sequence for each new code block. The identifiers distinguish states only through the in-phase components, and the quadrature components are not constrained. The preset identifiers include -A, +A, -B, and +B, where A and B are non-zero integers, and the Euclidean distance between -A and +B and between +A and -B is at least greater than 4. When the identifier of the previous code block is -A or -B, the identifier of the original sequence of the current code block is -A, and the identifier of the reverse sequence is +B. When the identifier of the previous code block is +A or +B, the identifier of the original sequence of the current code block is +A, and the identifier of the reverse sequence is -B.

5. The spectral notch coding modulation method according to claim 4, characterized in that, During the decoding operation, based on the received identifier, the current code block is determined according to the same identifier selection rule as the coded segment. If it is the original sequence, a hold operation is performed. If it is a reverse sequence, then perform the reverse operation.

6. A spectrum notch coded modulation system, characterized in that, include: Division unit: used to divide the symbol sequence after mapping the input bit sequence into units of length 1. l The code blocks are divided into blocks, each containing... l - One payload symbol and one identification symbol; Candidate sequence construction unit: used to construct at least two sets of candidate symbol sequences for each code block to be encoded, the first set being the original sequence that keeps the original symbols unchanged; The second group is the reverse sequence of all inverted symbols within the same code block; Identifier addition unit: used to set an identifier for each group of symbol sequences; WRDS calculation unit: used to calculate the weighted running numbers and WRDS for each group of candidate symbol sequences; Output unit: Used to select a set of candidate symbol sequences with smaller WRDS amplitudes as the final encoded output of the current code block.

7. The spectrum notch coded modulation system according to claim 6, characterized in that, After calculating the WRDS of each group of candidate symbol sequences, the WRDS calculation unit adds the calculated WRDS result of each group to the cumulative WRDS value of all previously encoded code blocks to obtain the cumulative WRDS value corresponding to each group of candidate sequences.

8. The spectrum notch coded modulation system according to claim 6, characterized in that, When the identifier adding unit adds an identifier to each code block: first, it arbitrarily assigns an in-phase component from the preset identifiers as the initial identifier; then, according to preset rules, it assigns different identifiers to the original sequence and the reverse sequence for each new code block. The identifiers are distinguished only by the in-phase component, and the quadrature component is not constrained. The preset identifiers include -A, +A, -B, and +B, where A and B are non-zero integers, and the Euclidean distance between -A and +B and between +A and -B is at least greater than 4. When the identifier of the previous code block is -A or -B, the identifier of the original sequence of the current code block is -A, and the identifier of the reverse sequence is +B. When the identifier of the previous code block is +A or +B, the identifier of the original sequence of the current code block is +A, and the identifier of the reverse sequence is -B.

9. The spectrum notch coded modulation system according to claim 8, characterized in that, It also includes a decoding unit: used to decode the encoded code block. During the decoding operation, it determines whether the current code block has been inverted according to the received identifier and the same identifier selection rule as the encoded segment. If it is the original sequence, it performs a hold operation. If it is a reverse sequence, then perform the reverse operation.

10. A digital signal processing method, characterized in that, include: During digital signal processing at the transmitting end, the pseudo-random bit sequence is first mapped to 16QAM symbols to generate two 35Gbuad 16QAM signals. Then, the signals are encoded using the spectrum notch coding modulation method described in any one of claims 1 to 5, and the QAM symbols are shaped by the Nyquist filter using a root raised cosine filter. After resampling the signals, pilot signals are inserted to estimate the frequency shift and laser phase noise. During digital signal processing at the receiving end, dispersion compensation is first performed on the received signal. Then, pilot-assisted frequency offset estimation is used. Next, the pilot signal is down-converted to baseband and extracted through a low-pass filter to estimate phase noise. Subsequently, a cascaded multimode algorithm is used for polarization demultiplexing and to suppress inter-symbol interference. Then, a decision-oriented least mean square algorithm is used to eliminate the IQ offset introduced at the transmitting end and to compensate for residual phase noise. Finally, decoding, demapping, and bit error rate calculation are performed sequentially.

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