System and method for processing radio frequency signals

Abstract

The present disclosure relates to systems and methods for processing radio frequency signals. A method for processing an RF signal is provided herein. The method includes receiving the signal with a chip sequence, sampling the signal to generate a sampled signal, generating a cancellation signal based on the chip sequence, subtracting the cancellation signal from the sampled signal to generate a cancelled signal, and performing accumulation based on the cancelled signal to generate a channel impulse response (CIR) estimate.

Description

Technical Field

[0001] This disclosure relates to channel estimation in ultra-wideband (UWB) communications, and more particularly to systems and methods for processing radio frequency (RF) signals. Background Technology

[0002] Ultra-wideband (UWB) generally refers to a wireless communication technology that uses a wide bandwidth, typically around 500 MHz or greater, or a bandwidth of 10 dB greater than 20% of the center frequency. Impulse UWB (IR-UWB) is a specific case of UWB where signals are transmitted via extremely short pulses (approximately in nanoseconds). IR-UWB is particularly well-suited for ranging or sensing applications because the pulses are robust to multipath propagation. Another advantage of IR-UWB is its ability to transmit data with extremely low power consumption.

[0003] Ranging is the process of determining the distance between two devices using UWB technology. In current wireless communication systems, Time-of-Flight (TOF) is commonly used to determine the distance between two devices / nodes in UWB communication. Pseudo-random cryptographic codes (e.g., scrambled timestamp sequences or STS) can be generated and transmitted between the two devices to calculate the Channel Impulse Response (CIR), thus preventing the TOF results from being influenced by attackers. However, with STS, there may be a trade-off between signal security and detectability. When a receiver raises its security threshold to block risky signals (e.g., attack signals), useful signals may be filtered out. Therefore, methods and systems that can mitigate this trade-off are desired. Summary of the Invention

[0004] This disclosure provides a method for processing a signal. The method includes: receiving the signal comprising a sequence of chips; sampling the signal to generate a sampled signal; generating a canceled signal based on the chip sequence; subtracting the canceled signal from the sampled signal to generate a canceled signal; and performing an accumulation to generate a channel impulse response (CIR) estimate based on the canceled signal.

[0005] In some embodiments, subtracting the cancellation signal from the sampled signal includes subtracting the real component of the cancellation signal from the real component of the sampled signal, and subtracting the imaginary component of the cancellation signal from the imaginary component of the sampled signal.

[0006] In some embodiments, generating the cancellation signal includes applying a cancellation function to the chip sequence to generate an initial cancellation signal, the initial cancellation signal including a real component and an imaginary component, and applying a rotation operation to the real component and the imaginary component of the initial cancellation signal to apply a phase shift to the initial cancellation signal and generate the cancellation signal such that the phase of the cancellation signal matches the phase of the sampled signal.

[0007] In some embodiments, sampling the signal to generate the sampled signal includes sampling the signal to generate an initial sampled signal comprising real and imaginary components, and performing a derotation operation on the real and imaginary components of the initial sampled signal to remove carrier frequency offset (CFO) and generate the sampled signal.

[0008] In some embodiments, the method further includes applying a saturation operation to the eliminated signal to generate a saturated signal, wherein the amplitude of the saturated signal is equal to or lower than the saturation level.

[0009] In some embodiments, applying the saturation operation includes applying a monotonic function to the eliminated signal within the saturation level.

[0010] In some embodiments, the monotonic function includes a clipping function.

[0011] In some embodiments, the method further includes performing a cross-correlation operation between the saturated signal and the chip sequence to generate a correlated signal.

[0012] In some embodiments, the method further includes performing a derotation operation to remove the carrier frequency offset (CFO) and generating a derotated signal as input to an accumulation operation, thereby generating the CIR estimate.

[0013] In some embodiments, the method further includes using the correlated signal as input to perform an accumulation to generate the CIR.

[0014] In some embodiments, the chip sequence includes a scrambled time sequence (STS).

[0015] In some embodiments, the cancellation function includes at least one of a single finite impulse response (FIR) filter, a dual FIR filter, or an infinite impulse response (IIR) filter. In some embodiments, the single FIR filter, the dual FIR filter, and / or the IIR filter may each individually serve as the cancellation function. In some embodiments, one or more single FIR filters, dual FIR filters, and / or IIR filters may be cascaded to serve as the cancellation function.

[0016] In some embodiments, the method further includes generating the cancellation function by: performing an accumulation operation on the synchronization header (SYNC) field of the signal to generate a second CIR estimate; performing a tap selection operation to select taps above a predetermined threshold; and determining the coefficients of the CF based on the index of the selected taps. In some embodiments, the generation of the cancellation function may be completed at the end of the SYNC or during the reception of the SYNC, when the CIR quality is considered sufficient.

[0017] In some embodiments, the method further includes generating the cancellation function by: performing an accumulation operation on the same STS field of the signal to generate an initial CIR estimate; performing a tap selection operation to select taps above a predetermined threshold; and determining the coefficients of the cancellation function based on the index of the taps selected during STS reception when the CIR quality is considered sufficient. That is, some STS symbols can be used to generate the cancellation function, and the cancellation function can be applied to the remaining STS symbols.

[0018] In some embodiments, the method further includes generating the cancellation function by performing a cross-correlation operation between the synchronization header (SYNC) field of the signal and a predetermined SYNC field to generate a second correlated signal; and applying an adaptive filter (AF) to the second correlated signal to generate the coefficients of the CF. [00XX] In some embodiments, the method further includes generating the cancellation function by applying an adaptive filter (AF) to the sample synchronization header (SYNC) field of the signal to generate the coefficients of the CF. That is, the AF can also be used in the time domain before the correlation. The AF can be adjusted until the chip in the SYNC symbol is eliminated.

[0019] In some embodiments, the AF is based on an adaptive algorithm including the Least Mean Square (LMS) algorithm.

[0020] In some embodiments, the method further includes generating the elimination function by performing a cross-correlation operation between the received chip sequence and the original chip sequence to generate a third correlated signal; and applying an adaptive filter (AF) to the third correlated signal to generate the coefficients of the CF. In some embodiments, the method further includes generating the elimination function by applying an adaptive filter (AF) to the received chip sequence to generate the coefficients of the CF. That is, the AF can also be used in the time domain before the correlation. The AF can be adjusted until the chips in the STS symbol are eliminated.

[0021] In some embodiments, the method further includes: generating a reference channel impulse response (CIR) estimate; and comparing the CIR estimate with the reference CIR estimate to obtain a first path (FP) signal in the reference CIR estimate.

[0022] In some embodiments, the generation of the reference CIR estimate includes applying the cancellation function to the chip sequence to generate the initial cancellation signal including the real component and the imaginary component; applying a rotation operation to the real component and the imaginary component of the initial cancellation signal to apply a phase shift to the initial cancellation signal and generate a rotated signal; scaling the rotated signal to a factor equal to the total number of chips in the sequence to generate a scaled signal; performing a derotation operation on the scaled signal to generate a second derotated signal; and accumulating the second derotated signal to generate the reference CIR estimate.

[0023] In some embodiments, generating the reference CIR estimate includes applying the cancellation function to the chip sequence to generate the initial cancellation signal including the real component and the imaginary component; scaling the initial cancellation signal to a factor equal to the total number of chips in the sequence to generate a scaled signal; and summing the scaled signal to generate the reference CIR estimate.

[0024] This disclosure provides an ultra-wideband (UWB) receiver. The UWB receiver includes an analog-to-digital converter (ADC), wherein the receiver is configured to: receive a signal comprising a chip sequence at the ADC; sample the signal by the ADC to generate a sampled signal; generate a canceled signal based on the chip sequence; subtract the canceled signal from the sampled signal to generate a canceled signal; and perform an accumulation to generate a channel impulse response (CIR) estimate based on the canceled signal.

[0025] In some embodiments, subtracting the cancellation signal from the sampled signal includes subtracting the real component of the cancellation signal from the real component of the sampled signal, and subtracting the imaginary component of the cancellation signal from the imaginary component of the sampled signal.

[0026] In some embodiments, generating the cancellation signal includes applying a cancellation function to the chip sequence to generate an initial cancellation signal, the initial cancellation signal including a real component and an imaginary component, and applying a rotation operation to the real component and the imaginary component of the initial cancellation signal to apply a phase shift to the initial cancellation signal and generate the cancellation signal such that the phase of the cancellation signal matches the phase of the sampled signal.

[0027] In some embodiments, sampling the signal to generate the sampled signal includes sampling the signal to generate an initial sampled signal comprising real and imaginary components, and performing a derotation operation on the real and imaginary components of the initial sampled signal to remove carrier frequency offset (CFO) and generate the sampled signal.

[0028] In some embodiments, the operation further includes applying a saturation operation to the eliminated signal to generate a saturated signal, wherein the amplitude of the saturated signal is equal to or lower than the saturation level.

[0029] In some embodiments, applying the saturation operation includes applying a monotonic function to the eliminated signal within the saturation level.

[0030] In some embodiments, the monotonic function includes a clipping function.

[0031] In some embodiments, the method further includes performing a cross-correlation operation between the saturated signal and the chip sequence to generate a correlated signal.

[0032] In some embodiments, the operation further includes performing a derotation operation to remove the carrier frequency offset (CFO) and generating a derotated signal as input to an accumulation operation, thereby generating the CIR estimate.

[0033] In some embodiments, the operation further includes using the correlated signal as input to perform an accumulation to generate the CIR estimate.

[0034] In some embodiments, the chip sequence includes a scrambled time sequence (STS).

[0035] In some embodiments, the elimination function includes at least one of a single finite impulse response (FIR) filter, a dual FIR filter, or an infinite impulse response (IIR) filter.

[0036] In some embodiments, the operation further includes generating the cancellation function by: performing an accumulation operation on the synchronization header (SYNC) field of the signal to generate a second CIR estimate; performing a tap selection operation to select taps above a predetermined threshold; and determining the coefficients of the CF based on the index of the selected taps.

[0037] In some embodiments, the operation further includes generating the cancellation function by performing a cross-correlation operation between the synchronization header (SYNC) field of the signal and a predetermined SYNC field to generate a second correlated signal; and applying an adaptive filter (AF) to the second correlated signal to generate the coefficients of the CF.

[0038] In some embodiments, the AF is based on an adaptive algorithm including the Least Mean Square (LMS) algorithm.

[0039] In some embodiments, the operation further includes generating the elimination function by performing a cross-correlation operation between the received chip sequence and the original chip sequence to generate a third correlated signal; and applying an adaptive filter (AF) to the third correlated signal to generate the coefficients of the CF.

[0040] In some embodiments, the operation further includes: generating a reference channel impulse response (CIR) estimate; and comparing the CIR estimate with the reference CIR estimate to obtain a first path (FP) signal in the reference CIR estimate.

[0041] In some embodiments, the generation of the reference CIR estimate includes applying the cancellation function to the chip sequence to generate the initial cancellation signal including the real component and the imaginary component; applying a rotation operation to the real component and the imaginary component of the initial cancellation signal to apply a phase shift to the initial cancellation signal and generate a rotated signal; scaling the rotated signal to a factor equal to the total number of chips in the sequence to generate a scaled signal; performing a derotation operation on the scaled signal to generate a second derotated signal; and accumulating the second derotated signal to generate the reference CIR estimate.

[0042] In some embodiments, generating the reference CIR estimate includes applying the cancellation function to the chip sequence to generate the initial cancellation signal including the real component and the imaginary component; scaling the initial cancellation signal to a factor equal to the total number of chips in the sequence to generate a scaled signal; and summing the scaled signal to generate the reference CIR estimate.

[0043] This disclosure provides a non-transitory computer-readable medium (CRM) having program code recorded thereon. The program code includes code for causing an ultra-wideband (UWB) device to receive a signal having a chip sequence; code for causing the UWB device to sample the signal to generate a sampled signal; code for causing the UWB device to generate a canceled signal based on the chip sequence; code for causing the UWB device to subtract the canceled signal from the sampled signal to generate a canceled signal; and code for causing the UWB device to perform an accumulation to generate a channel impulse response (CIR) estimate based on the canceled signal.

[0044] In some embodiments, the code for causing the UWB device to subtract the cancellation signal from the sampled signal includes code for causing the UWB device to subtract the real component of the cancellation signal from the real component of the sampled signal; and code for causing the UWB device to subtract the virtual component of the cancellation signal from the virtual component of the sampled signal.

[0045] In some embodiments, the code for causing the UWB device to generate the cancellation signal includes code for causing the UWB device to apply a cancellation function (CF) to the chip sequence to generate an initial cancellation signal containing real and imaginary components; and code for causing the UWB device to apply a rotation operation to the real and imaginary components of the initial cancellation signal to apply a phase shift to the initial cancellation signal and generate the cancellation signal, such that the phase of the cancellation signal matches the phase of the sampled signal.

[0046] In some embodiments, the code for causing the UWB device to sample the signal to generate the sampled signal includes code for causing the UWB device to sample the signal to generate an initial sampled signal containing real and imaginary components; and code for causing the UWB device to perform a de-rotation operation on the real and imaginary components of the initial sampled signal to remove carrier frequency offset (CFO) and generate the sampled signal.

[0047] In some embodiments, the non-transient CRM further includes code for causing the UWB device to apply a saturation operation to the eliminated signal to generate a saturated signal, wherein the amplitude of the saturated signal is equal to or below the saturation level.

[0048] In some embodiments, the code for causing the UWB device to apply the saturation operation includes code for causing the UWB device to apply a monotonic function to the canceled signal, the monotonic function being within the saturation level.

[0049] In some embodiments, the monotonic function includes a clipping function.

[0050] Those skilled in the art will recognize the scope of this disclosure and understand its other aspects after reading the following detailed description of preferred embodiments and the accompanying drawings. Attached Figure Description

[0051] The accompanying drawings, which are incorporated in and form a part of this specification, illustrate several aspects of this disclosure and, together with the description, serve to explain the principles of this disclosure.

[0052] Figure 1 A block diagram of a communication system with an exemplary receiver according to some embodiments of the present disclosure is shown.

[0053] Figure 2A A block diagram of an exemplary receiver having a cancellation function according to some embodiments of the present disclosure is shown.

[0054] Figure 2B A block diagram of another exemplary receiver with a cancellation function according to some embodiments of the present disclosure is shown.

[0055] Figure 2C Experimental results are shown for CIR estimates using existing and exemplary receivers according to some embodiments of this disclosure.

[0056] Figure 2D A block diagram of another exemplary receiver with a cancellation function according to some embodiments of the present disclosure is shown.

[0057] Figures 3A-3E Block diagrams of different exemplary elimination function generation modules according to some embodiments of the present disclosure are shown.

[0058] Figure 3F The coefficients of exemplary elimination functions according to some embodiments of this disclosure are shown.

[0059] Figure 4A and 4B Block diagrams of different exemplary path recovery modules according to some embodiments of the present disclosure are shown.

[0060] Figure 4C Experimental results are shown for CIR estimates using existing and exemplary receivers according to some embodiments of this disclosure.

[0061] Figure 5 A method for processing signals using an exemplary receiver according to some aspects of this disclosure is shown. Detailed Implementation

[0062] The embodiments described below illustrate the information necessary to enable those skilled in the art to practice the embodiments and demonstrate the best mode of practicing the embodiments. After reading the following description with reference to the accompanying drawings, those skilled in the art will understand the concepts of this disclosure and will appreciate the application of these concepts, even those not specifically set forth herein. It should be understood that these concepts and applications fall within the scope of this disclosure and the appended claims.

[0063] It will be understood that while the terms first, second, etc., may be used herein to describe various elements, these elements should not be limited by these terms. These terms are used only to distinguish one element from another. For example, without departing from the scope of this disclosure, a first element may be referred to as a second element, and similarly, a second element may be referred to as a first element. As used herein, the term "and / or" includes any and all combinations of one or more of the associated listed items.

[0064] The terminology used herein is for the purpose of describing particular embodiments only and is not intended to limit this disclosure. As used herein, unless the context clearly indicates otherwise, the singular forms “a / an” and “the” are intended to include the plural forms as well. It will also be understood that, when used herein, the terms “comprises,” “comprising,” “includes,” and / or “including” specify the presence of the stated features, integers, steps, operations, elements, and / or components, but do not exclude the presence or addition of one or more other features, integers, steps, operations, elements, components, and / or groups thereof.

[0065] Unless otherwise defined, all terms used herein (including technical and scientific terms) shall have the same meaning as commonly understood by one of ordinary skill in the art to which this disclosure pertains. It will be further understood that, unless expressly defined herein, the terms used herein shall be interpreted as having the same meaning as they have in the context of this specification and related art, and shall not be interpreted in an idealized or overly formal sense. Additionally, similar element symbols denote similar features throughout the specification and figures.

[0066] It should be understood that blocks in each signaling diagram or flowchart, and combinations of signaling diagrams or flowcharts, can be executed by computer program instructions. Since computer program instructions can be configured in the processor of a general-purpose computer, a special-purpose computer, or other programmable data processing apparatus, the instructions executed by the processor of the computer or other programmable data processing apparatus produce means for performing the functions described in the blocks of each signaling diagram or flowchart. Since computer program instructions can be stored in computer-usable or computer-readable storage, and can be directed to the computer or other programmable data processing apparatus to implement functions in a particular manner, the instructions stored in computer-usable or computer-readable storage can produce a product that includes instructions for performing the functions described in the blocks of each signaling diagram or flowchart. Since computer program instructions can be configured in a computer or other programmable data processing apparatus, instructions that produce a process executed by the computer as a series of operational steps are executed by the computer or other programmable data processing apparatus, and the instructions operating the computer or other programmable data processing apparatus can provide steps for performing the functions described in the blocks of each signaling diagram or flowchart.

[0067] Each block can represent a module, section, or code segment, which includes one or more executable instructions for performing a specific logical function. Furthermore, it should be noted that in some alternative execution instances, the functions mentioned in a block may occur in a different order. For example, two consecutively shown blocks may be executed substantially simultaneously or in reverse order, depending on the corresponding functions.

[0068] In the following description, embodiments are illustrated in detail with reference to the accompanying drawings. Furthermore, although a communication system using Ultra-Wideband (UWB) has been described in conjunction with embodiments, as examples, the embodiments can also be applied to other communication systems with similar technical backgrounds or features. For example, this may include communication systems using Bluetooth or ZigBee. Moreover, it will be determined by those skilled in the art that modifications can be made to such embodiments within the scope of this disclosure without significantly departing from it, and such modifications can be applied to other communication systems.

[0069] UWB can refer to a short-range, high-speed wireless communication technology that uses a wide bandwidth of several GHz or more, low spectral density, and short pulse width (e.g., 1 nanosecond to 4 nanoseconds) in baseband mode. UWB can also refer to the band itself used for UWB communication. UWB enables secure and accurate ranging between devices. Therefore, UWB achieves relative position estimation based on the distance between two devices, or accurate position estimation based on the distance to a fixed device (whose location is known, also referred to as an anchor device). This disclosure assumes that the user carries a device capable of communicating via UWB (referred to as a "UWB-enabled device" or simply a "UWB device").

[0070] In this disclosure, symbols and sequences may each refer to multiple chips (e.g., -1s and 1s). In some embodiments, a sequence includes one or more symbols.

[0071] In some communication systems, a cryptographically generated scrambled timestamp sequence (STS) can be used to generate a CIR that reflects the conditions of the UWB channel between the receiver and transmitter. However, STS typically cannot guarantee a normalized Effective False Acceptance (NEFA) rate at the physical (PHY) layer. To guarantee the desired NEFA (NEFA*), the detection threshold usually needs to be set as a fraction of the full-scale range of the analog-to-digital converter (ADC). As an example, the fraction could depend on the NEFA* and the total number (P) of UWB pulses in the STS sequence. The detection threshold defined in this way can be the minimum secure detection threshold that guarantees NEFA <= NEFA*. However, legitimate signals below the secure threshold are not detected or are rejected. Therefore, receiver performance is limited, especially under non-line-of-sight (NLOS) conditions.

[0072] A STS can be represented as a sequence of pseudo-random values, where each value is +1 or -1 with equal probability. After sharing a cryptographic key, two UWB-enabled devices can generate the same sequence (e.g., STS). A transmitter (e.g., a transmitting device) transmits a sequence of UWB pulses with a specific polarity according to the STS. The signal propagates from the transmitter to the receiver (e.g., a receiving device) and can attenuate, be delayed, and reflect after different paths. In some embodiments, the receiver cross-correlates the incoming signal with a known STS to separate and distinguish different paths. The first path (FP) to the receiver carries information about the distance between the two devices (D).

[0073] Several proximity-based services require an accurate estimate of D. Overestimating D can lead to denial of service (the service is denied when it should be allowed), while underestimating D can lead to a lack of security (the service is allowed when it should be denied).

[0074] FP Dynamic Range (FPDR) is a measure of receiver performance and indicates the weakest FP detected with a probability > 99%. FPDR can be expressed as the dB ratio between the FP and the strongest path in the received UWB signal, defined as the primary path or multipath (MP). NEFA is a measure of receiver security and indicates the probability of erroneous FP reception within the critical search window (CSW). The CSW typically precedes the MP.

[0075] Without loss of generality, the upper and lower limits of the ADC are assumed to be encoded as +1 and -1, respectively. Therefore, in this scenario, the magnitude of the maximum signal after cross-correlation can be equal to P, where P is the total number of UWB pulses in the STS sequence.

[0076] Suppose that an attacker attempting to infiltrate a receiver using an attack sequence (e.g., an attack signal) has no information about the legitimate STS; therefore, the cross-correlation between the attack sequence and the legitimate STS is random and independent of zero-mean and finite-variance concatenation. After the ADC, the attack sequence pulses are bounded by the ADC limit, and it can be proven that the probability distribution of the attack sidelobe magnitudes is bounded by the upper bound of the Rayleigh distribution with the scaling parameter S = sqrt(2 × P).

[0077] In other words, and without being bound by theory, NEFA <= exp(-T) 2 / (2 × S 2 And it is possible to calculate the minimum safety threshold T from the required NEFA*. For example, if NEFA* = 2 -48 Then T = sqrt(48 × 2 × log2 × S) 2 ) ≈ 8.16 × S. In summary, attack sidelobes are proportional to S, while legitimate signals are proportional to P, and the higher the required security level, the higher the security threshold T. FPDR may also be important in performance analysis.

[0078] A FP with a magnitude equal to the safety threshold has a 50% probability of being detected and accepted. This is because sidelobes and noise from other paths add to the FP, thus interfering constructively or destructively with equal probability. Generally, the sidelobes of the main path or multipath (MP) are dominant. In the case of destructive interference, the MP sidelobes interfering with the FP have equal and opposite phases.

[0079] To ensure a FP detection rate > 99%, the target path must typically be at least 2.33 × S higher than the threshold T. For example, if P = 4096, S = sqrt(2 × P) = 90.51, and the required security level is NEFA* = 2. -48 The safety threshold is calculated as T = sqrt(48 × 2 × log2) × S ≈ 738, and the 1% FP detection is calculated as T + 2.33 × S ≈ 949.

[0080] If the ADC full-scale range is suitable for MP, then the maximum dynamic range (DR) is obtained. In this case, after the example above, MP value = 4096, and DR = dB(949 / 4096) ≈ -12.7 dB. Similarly, for NEFA* = 2 -20 And NEFA* = 2 -10 The DR values ​​were ≈ -15.5 dB and DR values ​​were ≈ -17.5 dB, respectively.

[0081] If the received signal is allowed to exceed the ADC's limits, the DR can be improved because this will result in a gain for weaker signals, not the gain from an attack pulse that is already saturating the ADC. For example, if the ADC's full-scale range is suited to half of the MP, the FPDR is improved by 6 dB. However, this performance cannot be guaranteed: if the FP and MP pulses collide, the FP signal will disappear due to ADC saturation.

[0082] In summary, the higher the required safety level, the lower the guaranteed FPDR. FPDR can be improved, but not guaranteed by ADC saturation, and FPDR can be improved by about 3 dB when the number of pulses P is doubled.

[0083] This disclosure proposes a more efficient method for improving FPDR and detecting FP. The method includes estimating interference paths and eliminating these paths from the signal. In this way, performance loss due to destructive interference can be avoided, and receiver performance can be improved through signal saturation even if FP and MP pulses collide. This disclosure provides a technical solution that can lower the detection threshold to improve receiver performance without compromising security. The technical solution presented herein facilitates easier detection of desired signals, such as FP signals, which can be used to calculate the direct distance between the transmitter and receiver. This technical solution is achieved by eliminating unwanted signals. In ultra-wideband (UWB) security, the solution allows for a reduction in the effective full-scale range of the ADC, thus lowering the security threshold. In UWB sensing, the solution helps remove leaky signals without requiring post-processing techniques.

[0084] In this disclosure, a signal sampled by the receiver's ADC (e.g., a UWB signal) can be canceled by a cancellation signal. Cancellation can remove certain unwanted signals, such as MP signals. In some embodiments, the canceled signal is then clipped by undergoing a saturation operation to limit any strong unwanted signals (e.g., attack signals) that were not canceled. The saturated signal can then be cross-correlated with a STS to generate a correlated signal, which is a copy of a signal without canceled unwanted signals and with clipped strong signals. The correlated signals can then be accumulated to generate a channel impulse response (CIR) estimate, which can be compared with a reference CIR estimate (e.g., without cancellation and clipping / saturation) to obtain information about the FP signal. In various embodiments, the reference CIR estimate is generated using a receiver without cancellation or clipping or using the path recovery module of this disclosure.

[0085] In some embodiments, elimination is achieved by first applying an elimination function (CF) to the STS to generate a time-domain signal. The time-domain signal, which may or may not be processed, can be used to combine with or eliminate the sampled signal. The elimination function may have a filter architecture in which coefficients are generated using various methods provided in this disclosure, such as packet-based or STS-based synchronization (SYNC) fields.

[0086] Some elements of a technical solution may include the following:

[0087] First, the CF can be generated or estimated from the SYNC CIR. Digital cancellation and / or analog cancellation can be performed before cross-correlation. Methods for analog cancellation are provided. Second, adaptive algorithms can be used to estimate the CF. The CF can include a single finite impulse response (FIR) filter, a dual FIR filter, or an infinite impulse response (IIR) filter. Adaptive analog cancellation of stronger signals (e.g., noise and / or high-power attack signals) can improve / optimize the ADC range for weaker signals, thereby increasing the effective number of ADC bits.

[0088] Furthermore, processing operations such as de-rotation, resampling, dithering, and / or quantization of the signal can be applied before accumulation. Alternatively, rotation and resampling of digital samples can be applied to the signal before accumulation. Processing operations such as rotation and de-rotation can be applied to individual samples, sample blocks, or cells. Real-time saturation is used for saturated signals with strong amplitudes.

[0089] The recovery of the eliminated signal can be used to obtain FP information from the signal. Alternatively, additional delay lines, additional correlators, and / or additional CIRs can be used.

[0090] In addition, it reduces sidelobes caused by carrier frequency offset (CFO).

[0091] Figure 1 An exemplary communication system 100, including a first device 104 and a second device 106, is illustrated according to some embodiments. In some embodiments, the communication system 100 may include location determination features for determining a distance 102 between the first device 104 and the second device 106. In some embodiments, the first device 104 may be a source, while the second device 106 may include a mobile device. The first device 104 and the second device 106 may communicate with each other wirelessly, such as via UWB. The first device 104, provided at a fixed or known location, may be part of another device and / or coupled to an external device 108 via a network 110, such as the Internet, the Public Switched Telephone Network (PSTN), etc. The external device 108 may include any suitable device, such as an application server, that may be communicatively coupled to the first device 104 via the network 110. In some embodiments, location-based services may be provided by the external device 108 when the location of the second device 106 is determined. In some embodiments, the first device 104 is part of another device that may provide location-based services based on measurements of distance 102.

[0092] The first device 104 may include control circuitry 112, memory 114, and receiver (RX) 116, as well as antenna 120. Control circuitry 112 may be communicatively coupled to memory 114 and receiver 116. Antenna 120 may be communicatively coupled to receiver 116. Receiver 116 may receive RF signal 118 via the antenna. In some embodiments, RF signal 118 is a UWB signal and includes an RF packet with SYNC and STS fields. Control circuitry 112 may include any suitable software and / or hardware for controlling the functionality of the first device 104. For example, control circuitry 112 may include a processor, such as a central processing unit (CPU), graphics processing unit (GPU), and / or microprocessor. Receiver 116, controlled by control circuitry 112, may generate a CF (Current Flow Counter) and use the CF to receive and process RF signal 118 to generate a CIR estimate. In some embodiments, receiver 116 may perform path recovery to obtain a reference CIR estimate. Alternatively, the first device 104 may include a receiver (not shown) without a CF for generating the reference CIR estimate. Control circuit 112 can compare the CIR estimate with a reference CIR estimate to obtain FP information. Memory 114 may include random access memory (RAM), read-only memory (ROM), electrically erasable programmable ROM (EEPROM), optical disc storage device, magnetic disk storage device, etc. Operating software, data, and / or signal symbols can be stored / buffered in memory 114 for use. For example, STS can be stored / buffered in memory 114. Memory 114 can also store / buffer other data together for processing, such as SYNC in RF signal 118.

[0093] The second device 106 may include a portable computing device, such as a smartphone, laptop computer, tablet computer, wearable device, etc. Although not shown, the second device 106 may include a transmitter and an antenna 122 communicatively coupled together for transmitting signals such as RF signal 118. In some embodiments, the second device 106 may also include control circuitry and a memory communicatively coupled to the transmitter. The control circuitry may include any suitable software and / or hardware for controlling the functions of the second device 106. For example, the control circuitry may include a processor, such as a CPU, GPU, and / or microprocessor. The memory may include RAM, ROM, EEPROM, optical disc storage, disk storage, etc. Operating software, data, and / or signal symbols may be stored in the memory for use. For example, the transmitter of the second device 106, controlled by the control circuitry, may generate an RF signal 118 with STS and SYNC fields and transmit it to the first device 104.

[0094] Figure 2AThe architecture of an exemplary receiver 200 of this disclosure is shown. Receiver 200 may be an instance of receiver 116. Receiver 200 may include an ADC 202, a saturation module 204 (“SAT”), a correlator 206 (“CRR”), a de-rotation module 208 (“DE-ROT”), and an accumulator 210 (“ACC”). Receiver 200 may also include a cancellation module 207, which includes a cancellation function 212 (CF) and a rotation module 214 (ROT). Cancellation module 207 may generate a cancellation signal to cancel the sampled signal output by ADC 202. In some embodiments, receiver 200 may include analog and RF and baseband (BB) processing circuitry (not shown) that acquires a received RF signal 118 (modulated with a carrier frequency and a BB frequency) and provides front-end processing (e.g., amplification, filtering, down-conversion to the BB frequency) to generate a signal 203 (BB signal).

[0095] ADC 202 may include in-phase / quadrature (I / Q) sampling circuitry 107. Using the input signal 203, ADC 202 can output a sampled signal 218, which includes an in-phase component 218a (or I component, for ease of description) and a quadrature component 218b (or Q component). The I component 218a and Q component 218b are 90 degrees out of phase and can vary continuously according to the carrier frequency. In this disclosure, it is assumed that the sampled signal 218 may include actual signals containing FP and / or MP signals (e.g., BB signals including STS), random noise from the environment, any potentially attacking signals (possibly strong signals with high power / amplitude), etc.

[0096] Cancellation module 207 can generate cancellation signal 222, which includes I component 222a and Q component 222b configured to cancel the sampled signal 218. Cancellation signal 222 can be generated by performing a rotation operation on CF output signal 220 (including I component 220a and Q component 220b) generated by CF 212 by rotation module 214. CF output signal 220 can be the output of CF 212 with sequence 216 (e.g., an STS sequence shared by receiver 200 and transmitter) as input. In various embodiments, sequence 216 includes any suitable sequence with pseudo-random polarity for secure ranging. For example, sequence 216 may include ranging integrity fragments (RIFs) of IEEE 4z packets or 4ab multi-millisecond (MMS) packets. In some embodiments, sequence 216 may typically include any UWB sequence (even insecure ones such as SYNC or 4ab ranging sequence fragments (RSFs)) to increase dynamic range and / or improve ranging under NLOS conditions.

[0097] The coefficients of CF 212 can make the I component 220a and Q component 220b similar in amplitude to the I component 218a and Q component 218b, respectively. The rotation operation may include applying a phase shift to the CF output signal 220, wherein the phase shift matches the phase difference between the local oscillators of the transmitter and receiver 200. The cancellation signal 222 can then have a phase and amplitude that match the actual signal. In this disclosure, it is assumed that the phase shift applied by the rotation module 214 is known. In some embodiments, the cancellation signal 222 can cancel any MP signals in the sampled signal 218 that may potentially interfere with the CIR estimate of FP. In some embodiments, the cancellation signal 222 can also cancel FP signals in the sample signal 218.

[0098] like Figure 2A As shown, I component 222a can be applied / subtracted from I component 218, and Q component 222b can be applied / subtracted from Q component 218b to eliminate the actual signal, or eliminate as much signal as possible from the sampled signal 218. Although random noise and / or a high-power attack signal may be present, the eliminated signal 224, including the subtraction operation of I component 224a and Q component 224b, can ideally be zero. In some embodiments, random noise and / or high-power attack signals in the eliminated signal 224 are not eliminated by the eliminated signal 222 due to their randomness and / or unpredictability.

[0099] The eliminated signal 224 can then undergo a saturation operation by the saturation module 204. The saturation operation saturates the eliminated signal 224, thus limiting the intensity / amplitude of any uneliminated attack signals and / or noise in the eliminated signal 224. Signals below and / or above the limits of the saturation level are removed. In some embodiments, the saturation operation includes applying a saturation function to the eliminated signal 224. In some embodiments, the saturation function includes a suitable monotonic function within the saturation level. In some embodiments, the monotonic function includes a clipping function.

[0100] The saturated signal 226, having I components 226a and Q components 226b and a sequence 216 (e.g., STS), is then cross-correlated by correlator 206 to generate a correlated signal 228 having I components 228a and Q components 228b. The correlated signal 228 may include an actual signal similar to the sampled signal 218, but with reduced / clipped noise, reduced / clipped attack signals, removed MP signals, and / or removed FP signals.

[0101] The correlated signal 228 is then subjected to a derotation operation by the derotation module 208 to remove the CFO from the correlated signal 228, thereby generating a derotated signal 230 (with I component 230a and Q component 230b). Therefore, the phase of the derotated signal 230 can be aligned with the local oscillator. The derotated signal 230 can then be fed to the accumulator 210 for accumulation, and a CIR estimate 205 can be generated. In some embodiments, the CIR estimate 205 includes a channel response of sequence 216 having a clipped attack signal (if present), a down-shaving / clipped noise signal, a canceled FP signal, and a canceled MP signal.

[0102] Figure 2B The architecture of another exemplary receiver 201 of this disclosure is shown. Receiver 201 may be an instance of receiver 116. Receiver 201 may include ADC 202, saturation module 204, correlator 206, derotation module 208, and accumulator 210. Receiver 201 may also include cancellation module 209, which includes CF 212. Cancellation module 209 may generate a cancellation signal to cancel the sampled signal output by ADC 202. In some embodiments, receiver 201 may include analog and RF and baseband (BB) processing circuitry (not shown) that acquires the received RF signal 118 (modulated at the carrier frequency) and provides front-end processing (e.g., amplification, filtering, downconversion to the BB frequency) to generate signal 203 (BB signal).

[0103] Similar to receiver 200, receiver 201 can output a sampled signal 218 having I components 218a and Q components 218b. Unlike receiver 200, the sampled signal 218 undergoes a de-rotation operation via de-rotation module 208 to remove the CFO from the sampled signal 218, aligning the phase of the sampled signal 218 with the local oscillator of receiver 201. De-rotation module 208 can generate a de-rotated signal 223 having I components 223a and Q components 223b.

[0104] Simultaneously, the cancellation module 209 can generate a cancellation signal equal to the CF output signal 220, which includes the I component 220a and Q component 220b configured to cancel the derotated signal 223. Unlike receiver 200, since the sampled signal 218 has already been derotated to remove the CFO, the cancellation signal may not need to undergo a rotation operation. The cancellation signal (e.g., CF output signal 220) can be the direct output of CF 112, with sequence 216 (e.g., an STS sequence shared by receiver 200 and transmitter) as input. The cancellation signal 220 can then have a matching phase and amplitude with the derotated signal 223. In some embodiments, the cancellation signal can cancel any MP signals in the sampled signal 218 that may potentially interfere with the CIR estimate of FP. In some embodiments, the cancellation signal 220 can also cancel FP signals in the sampled signal 218.

[0105] Subtracting the cancellation signal (e.g., CF output signal 220) from the sampled signal 218 generates the cancelled signal 225 (comprising I component 225a and Q component 225b). Specifically, I component 220a can cancel I component 218a, and Q component 220b can cancel Q component 218b, thus eliminating the actual signal. While random noise and / or a high-power attack signal may be present, the cancelled signal 225 can ideally be zero. In some embodiments, due to their randomness and / or unpredictability, any random noise and / or high-power attack signal retained in the cancelled signal 225 will not be eliminated by the cancellation signal.

[0106] The eliminated signal 225 can then undergo a saturation operation by the saturation module 204. The saturation operation saturates the eliminated signal 225 and thus limits the intensity / amplitude of any uneliminated attack signals and / or noise in the eliminated signal 225. Signals below and / or above the saturation level limits are removed. In some embodiments, the saturation operation includes applying a saturation function to the eliminated signal 225. In some embodiments, the saturation function may be similar to the saturation function in receiver 200 and will not be described in detail again herein.

[0107] The saturated signal 227, having I components 227a and Q components 227b and a sequence 216 (e.g., STS), is then cross-correlated by correlator 206 to generate a correlated signal 229, having I components 229a and Q components 229b. The correlated signal 229 may include an actual signal similar to the sampled signal 218, but with down- / clipping noise, down- / clipping attack, removed MP, and removed FP signals.

[0108] The correlated signal 229 can then be transmitted to accumulator 210 for accumulation, and a CIR estimate 215 can be generated. In some embodiments, the CIR estimate 215 includes a channel response of sequence 216 having a clipped attack signal (if any), a down-sampling / clipped noise signal, a canceled FP signal, and a canceled MP signal.

[0109] Figure 2C The CIR estimates obtained using the disclosed receiver (“New”) and a conventional receiver (“Original”) are shown, in cases where (i) there is no attack signal and (ii) there is an attack signal. The New security threshold is lower when using the disclosed receiver. (Shown in...) Figure 2C In (i) and (ii), the FP signal is approximately 30 dB weaker than the MP signal in the RF channel. Using the disclosed receiver, a saturation level (SL) of 0.2 is employed. For NEFA* = 2 -48 The new FPDR improves by approximately 2 + dB(1 / 0.2) ≈ 16 dB from -17.5 dB, -15.5 dB, and -12.7 dB to -33.5 dB, -31.5 dB, and -28.7 dB, respectively. (i) shows the removal of MP sidelobes, and (ii) shows the clipping of stacked sidelobes. It should be noted that in the presence of an attack signal (ii), the attack sidelobes are clipped to ensure safety.

[0110] Figure 2D The architecture of another exemplary receiver 231 of this disclosure is shown. Receiver 231 can be configured to perform both time shift and phase shift.

[0111] The sampled signal 218 can undergo a phase derotation operation performed by the derotation module 208 to remove the phase rotation introduced by the CFO in the sampled signal 218 and form a derotated signal 223 (with I component 223a and Q component 223b) so that the phase of the derotated signal 223 is aligned with the local oscillator of the receiver 231.

[0112] The cancellation signal 222 (having I components 222a and Q components 222b) can be generated by performing a rotation operation on the CF output signal 220 (including I components 220a and Q components 220b) generated by CF 212 by the rotation module 214. The rotation operation may include applying a time drift / time shift to the CF output signal 220, wherein the time drift is matched with the time drift / time shift between the local oscillators of the transmitter and receiver.

[0113] Due to the aforementioned de-rotation and rotation operations, the eliminated signal 222 can then have a phase and time that match the received signal 203.

[0114] The eliminated signal 235, comprising I component 235a and Q component 235b, can be formed between the de-rotated signal 223 and the eliminated signal 222. Although random noise and / or a high-power attack signal may be present, the eliminated signal 235 can ideally be zero. In some embodiments, random noise and / or the high-power attack signal in the eliminated signal 235 are not eliminated due to their randomness and / or unpredictability.

[0115] The eliminated signal 235 can then undergo a saturation operation by the saturation module 204 to generate a saturated signal 237. The saturation operation saturates the eliminated signal 225, and thus limits the intensity / amplitude of any uneliminated attack signals and / or noise in the eliminated signal 225. Signals below and / or above the limits of the saturation level are removed. In some embodiments, the saturation operation includes applying a saturation function to the eliminated signal 225. In some embodiments, the saturation function includes a suitable monotonic function, such as a clipping function.

[0116] The saturated signal 237, having I components 237a and Q components 237b and sequence 216 (e.g., STS), is then cross-correlated by correlator 206 to generate a correlated signal 239, having I components 239a and Q components 239b. The correlated signal 239 is then de-rotated by de-rotation module 208 to remove the CFO from the correlated signal 239, resulting in a de-rotated signal 241 (having I components 241a and Q components 241b). Therefore, the phase and time of the de-rotated signal 241 can be aligned with a local oscillator. The de-rotated signal 241 can then be fed to accumulator 210 for accumulation, and a CIR estimate 235 can be generated. In some embodiments, the CIR estimate 235 includes the channel response of sequence 216, which has a clipped attack signal (if present), a down-shaving / clipped noise signal, a canceled FP signal, and a canceled MP signal.

[0117] Figure 3A , 3B Figures 3C, 3D, and 3E respectively illustrate exemplary CF 310 generated by corresponding CF generation modules 300, 301, 302, 303, and 309 according to some embodiments. CF 310 may be an instance of CF 212 in receivers 200 and 201. Figure 3A , 3B3C and 3C demonstrate that CF 310 can be estimated while receiving an RF signal (e.g., 118). In some embodiments, CF generation modules 300, 301, and 302 can automatically adapt to update the coefficients of CF 310 while continuing to receive the RF signal 118, thereby maximizing signal cancellation. Figure 2A 218 or Figure 2B (223 in the original text). In some embodiments, CF 310 is estimated in real time (e.g., when RF signal 118 is received via a UWB channel) using one of CF generation modules 300, 301, and 302 to generate coefficients reflecting the current UWB channel. Therefore, CF 310 can be tuned to achieve optimal signal cancellation. In CF generation modules 300, 301, and 302, coefficients can be generated and applied to suitable filters, such as FIR filters, dual FIR filters, and / or IIR filters, to form CIR 310. In some embodiments, real coefficients refer to the real part of the complex number used in the transfer function of CF 310 and are configured to modify the amplitude of the input signal (e.g., sequence 216). In some embodiments, imaginary coefficients refer to the imaginary part of the complex number in the transfer function of CF 310 and are configured to affect the phase of different frequency components of the input signal (e.g., sequence 216). Figure 3F Examples of real coefficient sets and imaginary coefficient sets are shown.

[0118] The proposed receiver avoids performance loss by subtracting samples above the saturation level before saturating the signal. This involves characterizing the magnitude, delay, and phase of these samples. Although not mandatory, samples below the saturation level can also be eliminated.

[0119] Typically, the strongest samples are grouped around the MP (Maximum Transformer) and can be captured by the coefficients of a relatively short FIR filter. However, in an MP environment, multiple strong paths can be received. Therefore, longer filters or multiple filters, or any other functional or filter architecture that can generate a time-domain signal from a known STS (Synchronous Transformer), can be used. For simplicity, all these solutions may be referred to as unique CF (Cyclic Fibre Channel) in this disclosure.

[0120] Several filter architectures can be designed. Complex finite impulse response (FIR) filters can directly represent the magnitude, delay, and phase of the sample to be eliminated. Filters can be formed as polyphase filters because the sequence rate is typically 1 / 8 or 1 / 16 of the sampling rate. The oscillatory nature of downmixed UWB signals makes them suitable for modeling by IIR filters. CF can be divided into blocks configured with known pulse shapes and blocks encoding the magnitude, delay, and phase of the MP or multiple samples to be eliminated. The pulse shape can be approximated by square or trigonometric functions. Pulse shape blocks can be IIR filters, and path block FIR filters.

[0121] The STS field is transmitted after the SYNC field. Therefore, the CF can be generated from the SYNC CIR, such as... Figure 3A As shown. For example, SYNC CIR samples above a certain threshold or within a window centered on the MP can be fed into the CF. This can be done at the end of SYNC or during its reception when the CIR quality is considered sufficient. Alternatively, adaptive algorithms can be used to estimate the quality during SYNC reception (e.g., Figure 3B (as shown) or during STS reception (such as Figure 3C The example shown is the CF algorithm for generating FIR or IIR filters. SYNC CIR can be used to initialize filters for faster convergence.

[0122] In some embodiments, additive random noise or other jitter techniques may be used during the generation or estimation of the CF. Alternatively or additionally, additive random noise or other jitter techniques may be used at the output of the CF, at the input of the ADC (analog domain), or at the output of the ADC (digital domain).

[0123] In some embodiments, CF 310 can be generated based on SYNC CIR. For example... Figure 3AAs shown, a receiver (e.g., 200 or 201) can receive SYNC 304 (e.g., archived SYNC / synchronization) in a received RF packet (e.g., RF signal 118) and can accumulate SYNC 304 in an accumulator 306 (“ACC”) to generate a SYNCCIR estimate 312 including I component 312a and Q component 312b. A tap selection (“TS”) operation 308 can then be performed based on the SYNC CIR estimate to select an appropriate number of taps as coefficients 314 for CF 310. In some embodiments, any tap greater than 0.5 × SL is selected. For example, if SL equals 0.2, taps with an absolute value equal to or greater than 0.1 × P, where P is the number of UWB pulses in the SYNC sequence, are selected. In some embodiments, if a predetermined number of X taps are to be selected, the X strongest taps are selected regardless of their magnitude. In some embodiments, the strongest taps refer to those samples with the highest complexity magnitude. When processing I and Q taps independently, the magnitude refers to the absolute value of the real / virtual sample. In some embodiments, since energy is typically concentrated in bursts, X taps are selected around the strongest tap. In some embodiments, the selected taps are above a predetermined threshold. In some embodiments, coefficient 324 includes virtual coefficient 314a and real coefficient 314b. Coefficient 314 can then be applied to one or more of an FIR filter, a dual FIR filter, and / or an IIR filter to form CIR 310. In some embodiments, since SYNC is received before STS, CF 310 can be generated / estimated before or during STS reception (e.g., 216). In some embodiments, the CF generator module 300 involves minimal modifications to a conventional receiver.

[0124] In some embodiments, CF 310 can be generated based on the cross-correlation between SYNC 304 (received) and the original SYNC 305 (known to both the transmitter and receiver). In some embodiments, CF 310 can be generated during the reception of SYNC 304. Figure 3BAs shown, a receiver (200 or 201) can receive SYNC 304 (e.g., an archived SYNC) in an RF packet (e.g., a portion of RF signal 118) and cross-correlate it with the original SYNC 305 in correlator 318. Correlator 318 can output a correlated signal 322, which serves as I component 322a and Q component 322b. The correlated signal 322 can then be applied with an adaptive filter (AF) 320, which includes an adaptive algorithm such as least mean square (LMS). AF 320 can generate coefficients 324 of CF 310. In some embodiments, the inputs of correlator 318 (a sequence of pulses in the time domain, e.g., 304 and 305) can be used as error inputs to the adaptive filter loop (e.g., 320). AF 320 is tuned until the signal is correctly eliminated. In some embodiments, adaptive filtering should be performed before the saturation module (e.g., 204). In some embodiments, the output of correlator 318 (a compressed pulse in the accumulator domain) can also be used to drive an adaptive filter (e.g., 320) in a similar manner. In this case, the saturation module (e.g., 204) should be turned off during filter tuning. In some embodiments, coefficient 324 includes imaginary coefficients 324a and real coefficients 324b. Coefficient 324 can then be applied to one or more of an FIR filter, a dual FIR filter, and / or an IIR filter to form CIR 310.

[0125] In some embodiments, CF 310 can be generated based on the cross-correlation between STS 326 (received) and the original STS 307 (known to both the transmitter and receiver). In some embodiments, CF 310 can be generated during the reception of STS 326. Figure 3CAs shown, a receiver (e.g., 200 or 201) can receive STS 326 in an RF packet (e.g., a portion of RF signal 118) and can perform a cross-correlation between STS 326 and the original STS 307 in a cross-correlator 318. The correlator 318 can output a correlated signal 328, which is presented as I-component 328a and Q-component 328b. The correlated signal 328 can then be applied with an adaptive filter (AF) 320, which includes an adaptive algorithm such as Least Mean Square (LMS). The AF 320 can generate coefficients 330 of CF 310. In some embodiments, coefficients 330 include imaginary coefficients 330a and real coefficients 330b. Coefficients 330 can then be applied to one or more of an FIR filter, a dual FIR filter, and / or an IIR filter to form CF 310. In some embodiments, the CFO generator module 302 is independent of SYNC, making SYNC and STS reception independent. The CFO generator module 302 can automatically correct and eliminate errors. For example, if carrier recovery is imperfect, derotation may be inaccurate, but an adaptive filter (e.g., 320) can compensate for this, resulting in better cancellation.

[0126] In some embodiments, the time-domain signal, rather than the cross-correlated signal, can also be used as input to AF 320 to generate coefficients 332 of CF 310. Figure 3D and 3E Each example is shown. For example... Figure 3D As shown, in the CF generation module 303, the output of the ADC 202 (e.g., a sampled signal 218 including I component 218a and Q component 218b, a time-domain signal) can be the input of the AF 320, which generates coefficients 332 (including imaginary coefficients 320a and real coefficients 320b) based on the received SYNC 304. In some embodiments, the AF 320 includes a feedback mechanism that adjusts coefficients 332 based on the difference between the sampled signal 218 of the CF 310 and the output signal 220. For example, the AF 320 can adjust coefficients 332 generated based on continuously received SYNC 304 until the difference is substantially zero or below a predetermined threshold. Figure 3E As shown, in the CF generation module 309, the sampled signal 218 can also be the input to the AF 320, which generates coefficients 334 (including imaginary coefficients 320a and real coefficients 320b) based on continuously received STS 326. In some embodiments, the feedback mechanism of the AF 320 adjusts coefficients 334 based on the difference between the sampled signal 218 of the CF 310 and the output signal 220 until the difference is substantially zero or below a predetermined threshold.

[0127] In some embodiments, the transmitter and receiver (e.g., 200 or 201) have a residual carrier frequency offset (CFO) after downconversion from the RF frequency. Appropriate corrections can be implemented to achieve better cancellation of the I / Q components. At lower CFOs, cancellation may stop at a fraction of the STS field to avoid adding signals rather than canceling them. At higher CFOs, de-rotation and resampling techniques can be applied.

[0128] Rotation can modify the CF function or be applied to the CF output, such as... Figure 2A (For example, 220) As shown. Alternatively, derotation can be applied directly to the ADC output, as... Figure 2B (For example, as shown in 218). In this case, rotation after the correlator is not necessary. It can be proven that if FP has a phase equal to 0, then S = sqrt(P). Therefore, Figure 2B The proposed scheme can be used to guarantee an FP (FP) phase equal to 0 and improve the dynamic range (DR) by 3 dB. De-rotation can include resampling to follow the time shift between the two streams (ADC output and CF output).

[0129] Rotation and / or derotation can be performed on a 512-chip cell basis, with each chip taking approximately 2 nanoseconds. Elimination can be stopped at a fraction of the cell to avoid adding a signal rather than eliminating it. The fraction can be selected based on the estimated CFO. Rotation and / or derotation can be performed on a sample or a block of fractions of cells. If as Figure 2A After the correlator shown undergoes derotation, the correlator can also be divided into fractions of units. The fraction can be a power of 2, and higher powers can be dynamically selected based on the estimated CFO. For this purpose, derotation and rotation can be applied to each complex sample.

[0130] Although this disclosure focuses on elimination in the digital domain, the same approach can also be applied in the analog domain, before the mixer, or in the baseband.

[0131] Methods for performing cancellation in the analog domain may include, but are not limited to, using one or more digital-to-analog converters (DACs), reusing the transmitter side of a transceiver, quickly adjusting the DC offset of the ADC input to follow the signal to be cancelled, and using multiple ADCs with different DC offsets to select the correct DC offset for each sample.

[0132] In various embodiments, the residual signal at the input of the saturation module can be equal to, less than, or greater than the saturation level (SL). Residual signals above, equal to, or less than SL are not destructive to the method because safety is still guaranteed.

[0133] The residual signal level (SL) can be dynamically selected based on the residual signal. For example, if SL = 0.2, clipping / saturation of samples up to 0.3 may not be necessary. The maximum number of samples allowed across the SL and the maximum level that cannot be exceeded (0.3 in this example) can be parameters of the SL, which can be configured according to NEFA*. Of course, in some embodiments, the final safety threshold must be adjusted based on the number of samples saturated at 0.3. The same idea can be extended to multiple levels (0.2, 0.3, 0.4, ...) where the associated maximum number of samples allows access to this level.

[0134] In some embodiments, quantization may be applied before elimination to match the quantization level of the ADC output. In some embodiments, quantization may be applied after elimination to match the correlator input. In some embodiments, quantization may be applied both before and after elimination. Depending on the derotation and / or rotation scheme used, quantization may be applied before or after the derotator (or rotation), or both.

[0135] If cancellation is performed in the analog domain, saturation can be performed in the analog domain by reducing the ADC full-scale range and / or by increasing the signal gain. Alternatively, cancellation can be performed in the digital domain as described in this disclosure. If cancellation is performed in the digital domain, saturation can be performed in the digital domain as described in this disclosure. Any saturation or loss of linearity in the analog domain, or typically prior to cancellation, may result in performance degradation if the clipped sample collides with the FP.

[0136] The saturation level (SL) can be selected based on the desired NEFA level (NEFA*). The lower the NEFA*, the more DR can be recovered, and therefore the lower the SL. A higher SL relaxes the constraint on elimination, which may not be optimal. Similarly, the saturation limit can be fixed to + / -1 or any other fixed value, and the signal can be digitally scaled by 1 / SL or a similar scaling factor, or scaled by applying a 1 / SL gain to the analog chain.

[0137] The saturation function (SF) can include any monotonic function not exceeding SL. For example, SF can be a clipping function for any positive sample whose SL is equal to or greater than SL and for any negative sample whose -SL is equal to or less than -SL.

[0138] While keeping the information on unsaturated samples is not a strict requirement, it allows the published steps to be applied only at those lags within the Critical Search Window (CSW). At lags outside the CSW, the original correlator CIR block can remain unchanged. For example, if the CF is calculated based on the SYNC field, as... Figure 3AAs shown, the CF can be easily attacked to introduce false MPs. Eliminating non-existent MPs during the STS field will introduce erroneous features. However, since the CSW always precedes the MP, the only effect within the CSW is the injection of sidelobes. If the SF is also applied to those hysteresis sequences corresponding to the CF, careful channel impulse analysis (CIA) is required to eliminate false MPs injected by erroneous CFs.

[0139] The unsaturated sample is typically stored in a buffer or digital delay line. The SF (Self-Saturation) can be applied "on the go" when the correlator reads the stored sample. In this way, information about the sample before saturation can be maintained in the buffer. For example, if the sample is stored as a binary number, the SF can be an OR between all bits, effectively setting each number to the least significant bit (LSB) above and equal to the LSB. If the correlator processes the sample bits independently, saturation can be achieved by considering only the LSB. Alternatively, the unsaturated sample can be stored in a second buffer or delay line, or... Figure 2A and 2B The complete architecture shown can be an additional block where the original correlator CIR block works on unsaturated samples (e.g., for reference CIR estimates).

[0140] Figure 4A and 4B Two exemplary path recovery modules 400 and 401 according to embodiments of this disclosure are shown. Path recovery modules 400 and 401 can recover any signals that have been eliminated or clipped / saturated in receivers 200 and 201, and can be used as additional portions / branches of receivers 200 / 201 communicatively coupled to accumulator 210. Accumulator 210 can then output a reference CIR estimate containing information about the recovery of noise, attack signals, MP signals, and FP signals. The reference CIR estimate can be compared with CIR 205 / 215 to extract more accurate information about the FP signal. In some embodiments, in path recovery modules 400 and 401, CF 402 can be an instance of CF 212.

[0141] In the path recovery module 400, the CF 402 can output a CF output signal 220 with a sequence 216 (STS) as input. The CF output signal 220 can undergo a rotation operation via the rotation module 404 (similar to rotation module 214) to align its phase with the input signal 203, such as... Figure 2AThe rotated signal 222 can then undergo quantization and / or resampling (to generate a quantized signal including I components 414a and Q components 414b) and be cross-correlated with a known STS sequence 216 to generate a correlated signal 416 (including I components 416a and Q components 416b). The correlated signal 416 (with I components 416a and Q components 416b) can then undergo a derotation operation via a derotation module 406 (similar to derotation module 208) to generate a derotated signal 418 including I components 418a and Q components 418b, to remove the CFO and align the phase of the correlated signal 416 with the receiver's local oscillator (e.g., 200 or 201). The derotated signal 418 can then be fed as input to accumulator 210. Accumulator 210 can accumulate the derotated signal 418 to generate a reference CIR estimate 407, which can be compared with CIR estimate 205 to extract the FP signal and / or other information from the accumulated post-processing.

[0142] In path recovery module 401, unlike the path recovery module of path recovery module 400, the CF output signal 220 may not undergo rotation or de-rotation operations. The CF output signal 220 may be scaled by P (e.g., P is the number of UWB pulses in the STS sequence) to form a scaled signal 418 (with I component 418a and Q component 418b), which is transmitted as input to accumulator 210. Accumulator 210 may then accumulate the scaled signal 418 and output a reference CIR estimate 407, which may be compared with CIR estimate 205 to extract FP signal and / or other post-accumulation processing information.

[0143] In various embodiments, path recovery modules 400 and 401 may be part of receivers 200 / 201 and may share the same accumulator (e.g., 210). For example, the inputs to accumulator 210 may include signals that have undergone clipping (e.g., 230, 229) and signals that have not undergone clipping (e.g., 416, 418). The output of accumulator 210 may include CIR estimates (e.g., 205, 207) with some of the clipped / reduced / canceled signals, and a reference CIR estimate with all the retained / recovered signals. The CIR estimates can be compared with the reference CIR estimates to extract more accurate FP signals and other information about the RF channel.

[0144] In some embodiments, receivers 200 / 201 may not include a path recovery module, and accumulator 210 may not output any reference CIR estimate. Instead, another receiver, such as a conventional receiver without a signal cancellation function, may be computed in parallel to generate a reference CIR estimate.

[0145] In some embodiments, it may be desirable to recover information removed by elimination. For example, if no FP path exists, it may be desirable to detect MP. In the absence of the methods disclosed in this section, careful CIA is required to detect eliminated MP.

[0146] Uneliminated samples (ADC output, such as 218) can be stored in a second buffer or delay line, or Figure 2A and 2B The complete architecture shown can be an additional block where the original correlator CIR block works on unremoved samples to obtain a reference CIR estimate.

[0147] Alternatively, the CF coefficients can be directly added to the corresponding hysteresis in the CIR after scaling by the total number of UWB pulses in the STS sequence P, such as... Figure 4A and 4B As shown. If the solution involves rotation and / or rotation, it is similar to... Figure 2A Furthermore, if quantization is applied after the rotator or before the correlator, then it can be used. Figure 4A The illustrated path recovery (PR) solution involves applying quantization after the rotation module. Alternatively, one could use... Figure 4B The PR solution is shown. In both cases, if Figure 2A and 2B The solution shown involves quantization prior to CIR, then Figure 4A and 4B The solution shown can also be quantized before CIR to obtain better PR results.

[0148] If the CF is determined from the SYNC field, such as Figure 3A As shown, it can be easily attacked to introduce false MPs. During the STS field, eliminating non-existent MPs will introduce erroneous features. In this case, PR will automatically remove the erroneous features. However, if saturation is applied to those hysteretic samples corresponding to CF, then... Figure 4A and 4B The PR solution shown can have an additional saturation step. Otherwise, PR will introduce the opposite unsaturation characteristics and will require careful CIA to exclude spurious MP injected by PR.

[0149] Figure 4CThe path recovery results are shown for a path without FP and with one MP and an attack signal. (a) shows that after the proposed path recovery, NEFA* = 2. -48 The new security threshold is lower, and MP is unaffected. (b) is a magnified image of (a) and shows that the side lobes of the attack signal are clipped. It can be noted from (a) that after the method is applied, one of the back lobes of the MP back lobes is clipped.

[0150] Figure 5 This is a flowchart of a method 500 for processing signals by a receiver (e.g., 200 and / or 201) according to some embodiments of the present disclosure. Method 500 is merely one example and is not intended to limit the present disclosure beyond what is expressly stated in the claims. Additional operations may be provided before, during, and after method 500, and some of the described operations may be replaced, eliminated, or moved around for other embodiments of method 500. For ease of illustration, Figure 5 It is a combination Figure 2A and 2B Described.

[0151] At step 502, a signal having a chip sequence is received (e.g., 203).

[0152] At step 504, the signal is sampled to generate a sampled signal (e.g., 218).

[0153] At step 506, a cancellation signal (e.g., 222 or 220) is generated based on the chip sequence.

[0154] At step 508, the cancellation signal is subtracted from the sampled signal to generate the cancellation signal (e.g., 224 or 225).

[0155] At step 510, accumulation (e.g., 210) is performed based on the eliminated signal to generate a channel impulse response (CIR) estimate (e.g., 205 or 215).

[0156] It should be noted that, in this disclosure, at least one of the operations (e.g., derotation, saturation, cross-correlation, accumulation, rotation, etc.) performed by the receiver or module (e.g., 200, 201, 400 and 401) is implemented by dedicated hardware (such as application-specific integrated circuits (ASICs) and / or digital signal processors (DSPs)) and / or general-purpose processors.

[0157] Those skilled in the art will recognize improvements and modifications to the preferred embodiments of this disclosure. All such improvements and modifications are considered to be within the scope of the concepts disclosed herein and the claims below.

Claims

1. A method for processing a signal, the method comprising: Receive the signal containing the chip sequence; The signal is sampled by an analog-to-digital converter (ADC) to generate a sampled signal; An elimination signal is generated based on the chip sequence; The eliminated signal is generated by subtracting the eliminated signal from the sampled signal; as well as Based on the eliminated signal, an accumulation is performed to generate a channel impulse response (CIR) estimate.

2. The method of claim 1, wherein subtracting the cancellation signal from the sampled signal comprises: Subtract the real component of the canceled signal from the real component of the sampled signal; and Subtract the virtual component of the canceled signal from the virtual component of the sampled signal.

3. The method of claim 1, wherein generating the cancellation signal comprises: An elimination function (CF) is applied to the chip sequence to generate an initial elimination signal, which includes real and imaginary components; and A rotation operation is applied to the real and imaginary components of the initial cancellation signal to apply a phase shift to the initial cancellation signal and generate the cancellation signal such that the phase of the cancellation signal matches the phase of the sampled signal.

4. The method of claim 2, wherein sampling the signal to generate the sampled signal comprises: The signal is sampled to generate an initial sampled signal, the initial sampled signal comprising real and imaginary components; and A derotation operation is performed on the real and imaginary components of the initially sampled signal to remove the carrier frequency offset (CFO) and generate the sampled signal.

5. The method of claim 1, further comprising applying a saturation operation to the eliminated signal to generate a saturated signal, wherein the amplitude of the saturated signal is equal to or below the saturation level.

6. The method of claim 5, wherein applying the saturation operation comprises applying a monotonic function to the eliminated signal, the monotonic function being within the saturation level.

7. The method of claim 6, wherein the monotonic function comprises a clipping function.

8. The method of claim 5, further comprising performing a cross-correlation operation between the saturated signal and the chip sequence to generate a correlated signal.

9. An ultra-wideband (UWB) device comprising a UWB receiver, the UWB receiver including an analog-to-digital converter (ADC), wherein the receiver is configured to: The ADC receives a signal containing the chip sequence. The ADC samples the signal to generate a sampled signal; An elimination signal is generated based on the chip sequence; The eliminated signal is generated by subtracting the eliminated signal from the sampled signal; and Based on the eliminated signal, an accumulation is performed to generate a channel impulse response (CIR) estimate.

10. A non-transitory computer-readable medium (CRM) having program code recorded thereon, the program code comprising: Code used to enable ultra-wideband (UWB) devices to receive signals containing chip sequences; Code used to enable the UWB device to sample the signal using an analog-to-digital converter (ADC) to generate a sampled signal; Code used to enable the UWB device to generate a cancellation signal based on the chip sequence; Code used to cause the UWB device to subtract the cancellation signal from the sampled signal to generate the cancelled signal; as well as Code for enabling the UWB device to perform an accumulation to generate a channel impulse response (CIR) estimate based on the eliminated signal.

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