Circularly polarized stacked antenna element and beam steering method

Abstract

This application provides a circularly polarized stacked antenna element and a beam control method. The antenna element includes first to third dielectric substrates arranged from top to bottom. The first and second dielectric substrates are respectively provided with coaxially aligned parasitic and main radiating patches. The third dielectric substrate is provided with a metal ground plane and a feed network configured to output orthogonal signals. The ground plane has orthogonal coupling slots corresponding to the main radiating patches. The first and second dielectric substrates are provided with a metallized via array surrounding the patches, and the array is connected to the ground plane to form a shielded cavity. The beam control method determines the amplitude and phase compensation values ​​based on a multidimensional error compensation matrix and temperature characteristics, and injects the RF signal into the feed network after pre-distortion processing. This application utilizes the shielded cavity formed by the metallized via array to cut off surface waves, reduce spatial mutual coupling during large-angle scanning, and, combined with amplitude and phase pre-distortion and temperature compensation, maintains high-purity circularly polarized radiation, broadens the operating bandwidth, and improves polarization stability.

Description

Technical Field

[0001] This application relates to the field of satellite communications, and in particular to a circularly polarized stacked antenna element and a beam control method. Background Technology

[0002] The Ka band, with its abundant spectrum resources, has become one of the main operating frequency bands for high-throughput satellite communication systems. Spaceborne phased array antennas electronically control the feed phase of each antenna element, enabling rapid beam pointing without mechanical rotation, and are therefore widely used in communication links between satellites and ground terminals. To meet the requirements for ground coverage, spaceborne phased array antennas typically need wide-angle scanning capability, meaning beam deflection angles of ±45 degrees or even ±60 degrees or more. Meanwhile, satellite communication links generally employ circular polarization transmission to reduce polarization mismatch losses caused by satellite attitude changes and Faraday rotation effects. Axial ratio is a core indicator for measuring circular polarization purity; the closer the axial ratio is to 0 dB, the higher the circular polarization purity.

[0003] Under conditions of beam orthogonal beam or small-angle deflection, existing Ka-band circularly polarized microstrip patch antenna elements can maintain relatively ideal axial ratio and sidelobe level specifications. However, when the scanning angle increases to ±45 degrees or more, surface waves propagating laterally along the substrate are excited within the dielectric substrate containing the antenna elements. The electromagnetic energy carried by these surface waves forms coupling channels between adjacent antenna elements, causing distortion of the active radiation pattern of each element relative to the isolated radiation pattern. Due to the difference in propagation characteristics between surface waves and the main radiated wave, the energy components introduced by surface wave coupling disrupt the equal amplitude orthogonal relationship established by orthogonal feeding during spatial synthesis, leading to a break in the amplitude-phase balance between the main polarization component and the cross-polarization component in the radiation field. The direct consequence is a significant increase in the axial ratio of the antenna array in the wide-angle scanning direction and a decrease in circular polarization purity; simultaneously, the cross-polarization components superimpose in phase in a specific spatial direction, resulting in abnormal uplift of the cross-polarization sidelobes, and in severe cases, even cross-polarization grating lobes.

[0004] The impact of the aforementioned deterioration in cross-polarization performance on spaceborne communication systems is not limited to the radiation performance of a single antenna. In high-throughput satellite systems employing frequency reuse and polarization reuse technologies, signal isolation between adjacent beams relies on orthogonal polarization. Once the level of cross-polarization sidelobes rises or cross-polarization grating lobes appear, the polarization isolation between adjacent beams will decrease, leading to co-channel interference and consequently compressing the system's available spectrum reuse efficiency and communication capacity. Therefore, how to suppress surface wave mutual coupling under wide-angle scanning conditions at the antenna element level, enabling antenna elements to maintain stable circular polarization radiation characteristics and low cross-polarization sidelobe levels even under large-angle deflection, is a critical technical problem that needs to be solved in the design of Ka-band spaceborne phased array antennas. Summary of the Invention

[0005] To address the issues of polarization degradation and cross-polarized sidelobe elevation caused by surface wave mutual coupling under wide-angle scanning conditions, this application provides a circularly polarized stacked antenna element and a beam control method.

[0006] In a first aspect, the circularly polarized stacked antenna element provided in this application adopts the following technical solution: A circularly polarized stacked antenna element includes a first dielectric substrate, a second dielectric substrate, and a third dielectric substrate arranged sequentially from top to bottom. The upper surface of the first dielectric substrate is provided with a parasitic radiation patch; The upper surface of the second dielectric substrate is provided with a main radiating patch, and the main radiating patch and the parasitic radiating patch are coaxially aligned in the vertical direction; The upper surface of the third dielectric substrate is provided with a metal ground plane, and the lower surface of the third dielectric substrate is provided with a power supply network. An orthogonal coupling gap is opened on the metal ground plane at the vertical projection center of the main radiating patch. The first dielectric substrate and the second dielectric substrate are provided with a through-hole array of metallized vias. The array of metallized vias is arranged around the periphery of the parasitic radiating patch and the main radiating patch. The array of metallized vias is electrically connected to the metal ground plane to form a shielding cavity around the main radiating patch and the parasitic radiating patch. The power supply network is configured to output two orthogonal signals with equal amplitude and a 90-degree phase difference, which are coupled to the main radiating patch through the orthogonal coupling gap.

[0007] By employing the above technical solution, a physically isolated shielded cavity is constructed by setting an array of metallized vias surrounding the radiating patch in the first and second dielectric substrates. This shielded cavity can effectively cut off and suppress surface waves excited inside the dielectric substrate under wide-angle scanning conditions, thereby significantly reducing the spatial mutual coupling effect between adjacent antenna elements. The combination of orthogonal coupling slots and orthogonal signal excitation from the feed network not only ensures high-purity circularly polarized radiation but also maintains the amplitude-phase balance of the main polarization and cross-polarization components when the antenna deflects at large angles, effectively suppressing the abnormal rise of cross-polarization sidelobes. Furthermore, the stacked design of the main radiating patch and the parasitic radiating patch introduces dual-resonance characteristics, greatly broadening the impedance bandwidth and axial ratio bandwidth of the antenna element.

[0008] Optionally, the metallized via array includes a plurality of metallized vias arranged in a ring at equal intervals, wherein the spacing between adjacent metallized vias is less than one-quarter of the wavelength in the medium corresponding to the highest operating frequency of the circularly polarized stacked antenna element.

[0009] By adopting the above technical solution and utilizing the quarter-wavelength electromagnetic shielding principle in microwave engineering, the spacing between adjacent vias is less than one-quarter of the wavelength in the high-frequency dielectric. This allows for the formation of a continuous and electromagnetically closed metal shielding wall within the dielectric substrate, preventing high-frequency electromagnetic waves from leaking outwards through the via gaps. This maximizes the cutting off of the lateral propagation path of surface waves, further enhancing the isolation of the shielding cavity and ensuring extremely low mutual coupling levels in the array environment.

[0010] Optionally, both the main radiating patch and the parasitic radiating patch are centrally symmetrical, and the area of ​​the parasitic radiating patch is smaller than the area of ​​the main radiating patch.

[0011] By adopting the above technical solution, the centrally symmetrical shape ensures the absolute symmetry of the radiation field structure in the orthogonal microwave mode, which is conducive to maintaining excellent circular polarization purity (i.e., low axial ratio); while the small-area parasitic radiation patch can introduce a resonant point slightly higher than the resonant frequency of the main radiation patch above it. The two frequency points are coupled and superimposed on each other, thereby significantly widening the working frequency band, making it better able to meet the high bandwidth transmission requirements of high-throughput satellite communication.

[0012] Optionally, the power supply network includes a 3dB bridge and two orthogonally arranged microstrip feed lines. The two output terminals of the 3dB bridge are respectively connected to the two microstrip feed lines, and the two microstrip feed lines correspond to the two orthogonal directions of the orthogonal coupling gap.

[0013] By adopting the above technical solution and using a 3dB bridge as the core feeding component, two orthogonal excitation signals with strictly equal amplitude and a precise 90-degree phase difference can be output stably and intrinsically, ensuring the initial excitation quality of the circularly polarized wave from the RF input source; the physical spatial alignment of the orthogonal microstrip feed line and the orthogonal coupling gap realizes efficient mode coupling of microwave energy and reduces the risk of polarization distortion caused by the asymmetric structure.

[0014] Secondly, the circularly polarized stacked phased array antenna provided in this application adopts the following technical solution: A circularly polarized stacked phased array antenna includes: Multiple circularly polarized stacked antenna elements as described above are arranged in an array according to a triangular lattice. The center-to-center spacing of adjacent circularly polarized stacked antenna elements is less than 0.54 times the free-space wavelength corresponding to the highest operating frequency of the circularly polarized stacked phased array antenna.

[0015] By adopting the above technical solution, the triangular lattice arrangement combined with a compact spacing of less than 0.54 times the wavelength can achieve wide-angle scanning of ±60 degrees and above, while pushing the grating lobes outside the visible field of view, thus avoiding the gain reduction and electromagnetic interference caused by spatial grating lobes.

[0016] Thirdly, this application provides a beam control method for a circularly polarized stacked phased array antenna, which adopts the following technical solution: A beam control method for a circularly polarized stacked phased array antenna, applied to the circularly polarized stacked phased array antenna as described above, includes the following steps: S1. Obtain the scanning angle and current operating frequency of the target beam; S2. Determine the amplitude and phase compensation value corresponding to each of the circularly polarized stacked antenna elements based on the scanning angle and the current operating frequency. S3. Obtain the initial radio frequency signal to be transmitted, and use the amplitude and phase compensation value to perform amplitude modulation and phase shifting on the initial radio frequency signal input to each of the circularly polarized stacked antenna elements to generate a compensated radio frequency signal; S4. The compensated radio frequency signal is injected into the feed network of each of the circularly polarized stacked antenna elements. The compensated radio frequency signal is converted into two orthogonal signals with equal amplitude and a 90-degree phase difference by the feed network, and then coupled to the main radiating patch of the circularly polarized stacked antenna element through the orthogonal coupling gap.

[0017] By adopting the above technical solution, this application proposes a "digital predistortion" control strategy that combines the physical structural characteristics of the antenna. During beam scanning, the compensation weight is dynamically determined based on the scanning angle and frequency, and active amplitude and phase adjustment is performed before the RF signal is injected into the feed network. This can accurately offset the feed imbalance caused by factors such as array mutual coupling and spatial dispersion at the system control level. After the compensated signal enters the antenna element, it can re-establish perfect orthogonal amplitude and phase balance in the specified wide-angle radiation direction, enabling the phased array antenna to maintain excellent circular polarization axial ratio and extremely low cross-polarization level even during large-angle scanning.

[0018] Optionally, S2 includes: S21. Query the preset multidimensional error compensation matrix to obtain the amplitude and phase predistortion compensation value corresponding to the scanning angle and the current working frequency, and use the amplitude and phase predistortion compensation value as the amplitude and phase compensation value; wherein, the multidimensional error compensation matrix records the amplitude and phase predistortion compensation value of each of the circularly polarized stacked antenna elements under different scanning angles and different frequency combinations.

[0019] By adopting the above technical solution, and through offline calibration and the establishment of a lookup table-based multidimensional error compensation matrix, the system only needs to directly call the pre-stored compensation weights during actual beam switching, avoiding the complicated real-time polarization coupling matrix calculation process, greatly shortening the response time of beam pointing switching, and improving the real-time beam scheduling efficiency of the system.

[0020] Optionally, the multidimensional error compensation matrix is ​​constructed through the following pre-calibration steps: S01. Control the circularly polarized stacked phased array antenna to traverse through a preset set of scanning angles and frequency points one by one, and obtain the active standing wave ratio data of each circularly polarized stacked antenna element under each traversal node and the spatial mutual coupling characteristic parameters between adjacent circularly polarized stacked antenna elements; wherein, each combination of the scanning angle and the frequency point constitutes a traversal node. S02. Based on the active VSWR data and the spatial mutual coupling characteristic parameters, construct a mutual coupling scattering matrix, calculate the array active element pattern function under each of the traversal nodes, and perform polarization decomposition on the array active element pattern function to extract the main polarization component and the cross polarization component. Generate a polarization distortion vector based on the deviation between the main polarization component and the preset ideal circular polarization reference vector and the cross polarization component. S03. Based on the polarization distortion vector, solve for the inverse amplitude and phase weights that minimize the overall axial ratio of the array, and write the inverse amplitude and phase weights as the amplitude and phase pre-distortion compensation values ​​under the corresponding traversal nodes into the multidimensional error compensation matrix.

[0021] By adopting the above technical solution, a closed-loop high-precision calibration mechanism based on the real electromagnetic environment is provided. By extracting the active VSWR and spatial coupling parameters, the actual boundary conditions of the antenna array under large-angle deflection are restored; then, based on the polarization decomposition of the active element pattern, the inverse weight is solved with the minimization of the overall array axial ratio as the direct optimization objective. This can accurately capture and compensate for the underlying error sources that cause polarization degradation, ensuring the scientific and physical validity of the error compensation matrix data.

[0022] Optionally, after S21, step S2 further includes a step of temperature correction of the amplitude-phase pre-distortion compensation value: S22. Obtain the real-time temperature characteristic value of the circularly polarized stacked phased array antenna; S23. Based on the real-time temperature characteristic value and the thermal change dielectric constant curve of the third dielectric substrate in the circularly polarized stacked antenna unit, calculate the wavelength drift of the current operating frequency point inside the third dielectric substrate. S24. Calculate the phase delay change in the feed network based on the wavelength drift; S25. The phase delay change is converted into a compensation factor, and the compensation factor is added to the amplitude-phase predistortion compensation value to obtain the final amplitude-phase compensation value.

[0023] By employing the above technical solution, specialized adaptive optimization was performed for the extreme temperature fluctuations faced by spaceborne or outdoor base station systems. Temperature changes cause the dielectric constant of the third dielectric substrate to drift, thereby altering the electrical length of the microstrip feeder and disrupting the original 90-degree phase difference. This solution establishes a mapping relationship between temperature and phase delay for real-time factor compensation, effectively eliminating phase dispersion errors caused by environmental temperature fluctuations and ensuring the polarization stability and all-weather communication capability of the antenna system under harsh thermophysical environments.

[0024] Optionally, step S23 includes the following sub-steps: S231. Calculate the temperature difference between the real-time temperature characteristic value and the preset nominal reference temperature; S232. Obtain the temperature coefficient of dielectric constant of the third dielectric substrate, and calculate the real-time equivalent dielectric constant of the third dielectric substrate based on the temperature difference and the temperature coefficient of dielectric constant; S233. Calculate the real-time dielectric waveguide wavelength for microwave signal transmission inside the third dielectric substrate based on the current operating frequency and the real-time equivalent dielectric constant. S234. The wavelength drift is obtained by subtracting the real-time dielectric waveguide wavelength from the nominal dielectric waveguide wavelength at the nominal reference temperature.

[0025] By adopting the above technical solution, the specific calculation steps rigorously derive the analytical result of wavelength drift through material physical parameters (temperature coefficient), thereby improving the accuracy and engineering feasibility of temperature compensation calculation.

[0026] In summary, this application includes at least one of the following beneficial technical effects: 1. Effectively suppresses surface wave mutual coupling under wide-angle scanning conditions, maintaining high-purity circular polarization and broadening the bandwidth. This application constructs an electromagnetic shielding cavity by setting a metallized via array surrounding the radiating patch within the dielectric substrate, maximally cutting off the lateral propagation path of surface waves and significantly reducing the spatial mutual coupling effect between adjacent antenna elements. Combined with the stacked dual-resonant design of the main and parasitic radiating patches and orthogonal slot coupling, it not only maintains the amplitude and phase balance of the main polarization and cross-polarization components during large-angle deflections and suppresses the abnormal rise of cross-polarization sidelobes, but also significantly broadens the impedance and axial ratio bandwidth of the antenna element, meeting the requirements of high-throughput communication.

[0027] 2. Achieve high-precision, low-latency beam amplitude and phase predistortion compensation to ensure polarization isolation during wide-angle scanning. This application proposes a multi-dimensional error compensation strategy combined with real electromagnetic mutual coupling calibration. By constructing the multi-dimensional error compensation matrix offline and directly querying it during beam switching, precise active amplitude and phase adjustment is performed before the RF signal is injected into the feed network. This method offsets the orthogonal feed imbalance caused by array mutual coupling at the system control level, enabling the phased array antenna to maintain excellent axial ratio during wide-angle scanning, avoiding co-channel interference, and significantly improving the real-time response efficiency of beam scheduling.

[0028] 3. Possesses temperature adaptive correction capability, significantly improving system polarization stability under extreme thermophysical environments. This application addresses the challenge of drastic temperature variations faced by spaceborne or outdoor systems by establishing a wavelength drift compensation model based on the temperature coefficient of the dielectric constant of the substrate. By calculating and compensating for phase delay errors in the feed network caused by ambient temperature fluctuations in real time, it effectively eliminates phase dispersion caused by thermal expansion and contraction of materials and changes in dielectric properties, ensuring the antenna system's all-weather communication capability and polarization consistency under harsh temperature environments. Attached Figure Description

[0029] Figure 1 This is a schematic diagram of the cross-sectional structure of a circularly polarized stacked antenna element in one embodiment of the present invention.

[0030] Figure 2 This is a top view schematic diagram of a circularly polarized stacked antenna element using a circular patch in one embodiment of the present invention.

[0031] Figure 3 This is a top view schematic diagram of a circularly polarized stacked antenna element using a chamfered square patch in one embodiment of the present invention.

[0032] Figure 4 This is a planar schematic diagram of the power supply network in one embodiment of the present invention.

[0033] Figure 5 This is a top view schematic diagram of the array layout of a circularly polarized stacked phased array antenna in one embodiment of the present invention.

[0034] Figure 6 This is a flowchart of the beam control method in one embodiment of the present invention.

[0035] Explanation of reference numerals in the attached figures: 100. Circularly polarized stacked antenna element; 110. First dielectric substrate; 111. Parasitic radiating patch; 120. Second dielectric substrate; 121. Main radiating patch; 130. Third dielectric substrate; 131. Metallic ground plane; 132. Feed network; 133. Orthogonal coupling slot; 140. Metallized via array; 1321. 3dB bridge; 1322a. Microstrip feed line (corresponding to the horizontal arm direction of the orthogonal coupling slot); 1322b. Microstrip feed line (corresponding to the vertical arm direction of the orthogonal coupling slot). Detailed Implementation

[0036] The present application will be further described in detail below with reference to the accompanying drawings. It should be understood that the specific embodiments described herein are merely illustrative of the present application and are not intended to limit the scope of the application.

[0037] Circular polarization refers to the electromagnetic wave polarization in which the electric field vector rotates with equal amplitude over time in a plane perpendicular to the direction of propagation.

[0038] Axis ratio is an indicator of the purity of circular polarization. It is defined as the ratio of the major axis to the minor axis of the polarization ellipse. The axial ratio of ideal circular polarization is 0 dB. The larger the axial ratio, the lower the purity of circular polarization.

[0039] Orthogonal coupling slots are a pair of mutually perpendicular narrow slots opened on a metal ground plane to couple electromagnetic energy in the power supply network to the radiating patch above in a non-contact manner.

[0040] Metallized via array refers to a collection of vias that penetrate a dielectric substrate and have a conductive metal layer plated on the inner wall. Multiple metallized vias are arranged at a specific interval and electrically connected to the ground plane to form a substrate-integrated waveguide cavity structure.

[0041] Surface waves are electromagnetic wave modes that propagate laterally along the interface between a dielectric substrate and air. In phased array antennas, surface waves are one of the main ways in which mutual coupling occurs between adjacent antenna elements.

[0042] An active radiation pattern is the far-field radiation pattern measured in an array environment when only one antenna element is excited and the remaining elements are terminated with matched loads. The active radiation pattern includes the perturbation of the radiation characteristics by array mutual coupling.

[0043] The mutual scattering matrix is ​​an N×N matrix, where N is the total number of antenna elements in the array. The element in the i-th row and j-th column of the matrix represents the amplitude and phase of the coupled signal generated at the port of the i-th element when the j-th element is excited.

[0044] The polarization distortion vector is a vector that describes the deviation of an antenna element from the ideal circular polarization state at a specific scanning angle and frequency. It includes three components: amplitude deviation, phase deviation, and cross-polarization crosstalk.

[0045] The amplitude and phase pre-distortion compensation value is the amplitude and phase adjustment amount applied in advance before the radio frequency signal is fed into the antenna element, which is used to offset the polarization distortion introduced by mutual coupling and manufacturing errors.

[0046] A traversal node is a specific combination of scanning angle and operating frequency. During calibration, the antenna array works on all traversal nodes one by one to collect complete error data.

[0047] The thermal dielectric constant curve describes the relationship between the relative dielectric constant of a dielectric substrate material and temperature. The wavelength of a dielectric waveguide is the actual wavelength of a microwave signal propagating inside the dielectric substrate, which is determined by the operating frequency and the equivalent dielectric constant of the substrate material.

[0048] A typical application scenario for this application is a Ka-band spaceborne high-throughput satellite communication system. As an example, the operating frequency range is 26.5 GHz to 40 GHz, the required scanning angle of the antenna array is ±60°, and the system's requirement for the circular polarization axial ratio is no more than 3 dB across the entire scanning range.

[0049] Figure 1 A schematic cross-sectional view of a circularly polarized stacked antenna element 100 according to some embodiments of this application is shown. Figure 1 As shown, the circularly polarized stacked antenna element 100 includes a first dielectric substrate 110, a second dielectric substrate 120, and a third dielectric substrate 130 arranged sequentially from top to bottom. A parasitic radiating patch 111 is provided on the upper surface of the first dielectric substrate 110. A main radiating patch 121 is provided on the upper surface of the second dielectric substrate 120, and the main radiating patch 121 and the parasitic radiating patch 111 are coaxially aligned in the vertical direction. A metal ground plane 131 is provided on the upper surface of the third dielectric substrate 130, and a feed network 132 is provided on the lower surface of the third dielectric substrate 130. An orthogonal coupling slot 133 is formed on the metal ground plane 131 at the center of the vertical projection of the main radiating patch 121. A through-hole array of metallized vias 140 is provided in the first dielectric substrate 110 and the second dielectric substrate 120. The metallized via array 140 is arranged around the periphery of the parasitic radiating patch 111 and the main radiating patch 121. The bottom of the metallized via array 140 is electrically connected to the metal ground plane 131, thereby forming a shielding cavity around the main radiating patch 121 and the parasitic radiating patch 111. The feed network 132 is configured to output two orthogonal signals with equal amplitude and a 90-degree phase difference. The orthogonal signals are coupled to the main radiating patch 121 in a non-contact manner through the orthogonal coupling gap 133.

[0050] The reason for adopting a three-layer substrate stacked structure is that the impedance bandwidth of a single-layer microstrip patch antenna in the Ka band is typically only 3%-5% of the operating frequency, which is insufficient to cover the bandwidth required for Ka band broadband communication. By adding a parasitic radiating patch 111 above the main radiating patch 121, with the main radiating patch 121 acting as an active resonator and the parasitic radiating patch 111 acting as a parasitic resonator, the two form two closely spaced resonant frequencies under electromagnetic coupling, which significantly broadens the impedance bandwidth and axial ratio bandwidth of the antenna element 100, enabling it to cover the broadband operating requirements of the Ka band.

[0051] As an example, the first dielectric substrate 110, the second dielectric substrate 120, and the third dielectric substrate 130 can be microwave dielectric materials, such as Rogers RO3003 (relative permittivity approximately 3.0, loss tangent approximately 0.0013) or Rogers RO4350B (relative permittivity approximately 3.48, loss tangent approximately 0.0037). The thickness of the first dielectric substrate 110 and the second dielectric substrate 120 can be selected in the range of 0.127 mm to 0.508 mm, and the thickness of the third dielectric substrate 130 can be selected in the range of 0.203 mm to 0.508 mm. It should be understood that the material and thickness of the dielectric substrate are not limited to the examples above, and LTCC (Low Temperature Co-fired Ceramics) material systems can also be used; no limitation is made here.

[0052] The shielding cavity formed by the metallized via array 140 and the metal ground plane 131 suppresses surface waves through the following physical mechanism: In the Ka band, TM0 and TE1 mode surface waves propagating laterally along the substrate are easily excited inside the dielectric substrate. When the antenna array performs a wide-angle scan, a portion of the electromagnetic energy radiated by each element couples into the dielectric substrate to form surface waves. These surface waves propagate laterally along the substrate and establish coupling channels between adjacent elements. The inner walls of each via in the metallized via array 140 are coated with a conductive metal layer, and the bottom of the via is electrically connected to the metal ground plane 131. When the spacing between adjacent vias is sufficiently small, the via array is electromagnetically equivalent to a continuous conductive wall. When the laterally propagating surface waves reach the conductive boundary formed by the via array, they are forced to short-circuit to the metal ground plane 131 and cannot continue to propagate to adjacent elements. Therefore, the radiation field of each antenna element 100 is confined within the shielding cavity, and the deviation between the active radiation pattern and the isolated radiation pattern of each element is suppressed, thereby maintaining the amplitude and phase balance of circularly polarized radiation in the wide-angle scan state.

[0053] In some embodiments, the first dielectric substrate 110, the second dielectric substrate 120, and the third dielectric substrate 130 are bonded and pressed together into an integral structure using a prepreg. In other embodiments, the three-layer dielectric substrate can also be prepared using an LTCC (Low-Temperature Ceramic Co-firing) process, in which multiple ceramic green bodies are co-fired at a temperature of 850°C to 900°C to form a dense, integrated multilayer structure. Both processes can meet the high-precision alignment requirements between the layers on the vertical projection plane.

[0054] In some embodiments, the metallized via array 140 includes a plurality of metallized vias arranged in a ring at equal intervals, the spacing between adjacent metallized vias being less than one-quarter of the wavelength within the dielectric corresponding to the highest operating frequency of the circularly polarized stacked antenna element 100. For example, assuming the highest operating frequency is 40 GHz and the relative permittivity of the second dielectric substrate 120 is 3.0, then the wavelength within the dielectric is... The wavelength within a quarter of the dielectric is approximately 1.08 mm. Therefore, the center-to-center spacing of adjacent metallized vias should be controlled within 1.08 mm. As an example, the diameter of the metallized via can be 0.2 mm, the center-to-center spacing of adjacent vias can be 0.5 mm, and the total number of vias can be determined based on the annular circumference and spacing. For example, for a square arrangement with a side length of approximately 4 mm, approximately 8 vias are needed on each side. When the spacing between adjacent vias exceeds a quarter of the dielectric wavelength, the gap between adjacent vias is no longer equivalent to a continuous conductive wall in electromagnetic characteristics. Surface wave energy can penetrate the gap and leak to adjacent cells, significantly reducing the surface wave cutoff capability of the shielding cavity.

[0055] Besides the circular arrangement, in other embodiments, the metallized via array 140 can also be arranged in a rectangular ring. The rectangular ring arrangement is suitable for array layouts where the antenna elements 100 have a square outer contour, aligning the contour of the via array with the boundary of the square elements, which helps improve the fill rate of the array arrangement. Both the circular and rectangular ring arrangements are implementations where the metallized via array 140 is arranged around the periphery of the radiating patch.

[0056] Figure 2 and Figure 3Top views of antenna elements 100 according to some embodiments of this application are shown. In some embodiments, both the main radiating patch 121 and the parasitic radiating patch 111 are centrally symmetrical, with the area of ​​the parasitic radiating patch 111 being smaller than that of the main radiating patch 121. The requirement for a centrally symmetrical shape stems from the physical conditions of circular polarization excitation—the feed network 132 excites two orthogonal degenerate modes (TM10 and TM01 modes) on the main radiating patch 121 through orthogonal coupling slots 133, and the resonant frequencies of the two degenerate modes must be strictly equal. If the shape of the radiating patch deviates from central symmetry, the equivalent electrical lengths in the two orthogonal directions will be inconsistent, causing the resonant frequencies of the two degenerate modes to split, violating the equal-amplitude orthogonality condition, and deteriorating the axial ratio. The main radiating patch 121 and the parasitic radiating patch 111 can be circular or chamfered squares, with the chamfer used to fine-tune the coupling degree between the two orthogonal modes.

[0057] The area of ​​the parasitic radiating patch 111 is smaller than that of the main radiating patch 121, causing the resonant frequency of the parasitic radiating patch 111 to be slightly higher than that of the main radiating patch 121. The two adjacent resonant frequencies together cover the target operating frequency band, forming a broadband dual-resonance characteristic. As an example, the ratio between the side length (or diameter) of the parasitic radiating patch 111 and the side length (or diameter) of the main radiating patch 121 can be selected within the range of 0.85 to 0.92.

[0058] Figure 4 A plan view of a feed network 132 according to some embodiments of this application is shown. In some embodiments, the feed network 132 includes a 3dB bridge 1321 and two orthogonally arranged microstrip feed lines 1322a and 1322b. The input of the 3dB bridge 1321 receives radio frequency signals, and the two outputs of the 3dB bridge 1321 are respectively connected to microstrip feed lines 1322a and 1322b. Microstrip feed line 1322a corresponds to the horizontal arm direction of the orthogonal coupling slot 133, and microstrip feed line 1322b corresponds to the vertical arm direction of the orthogonal coupling slot 133. After the radio frequency signal is divided by the 3dB bridge 1321, it forms two orthogonal signals with equal amplitude and a 90-degree phase difference. The two orthogonal signals are transmitted along the microstrip feed line 1322a and the microstrip feed line 1322b to the horizontal and vertical arms of the orthogonal coupling slot 133, respectively. The electromagnetic energy is transferred to the main radiating patch 121 through the slot coupling, and two orthogonal degenerate modes are excited on the main radiating patch 121. The two degenerate modes become circularly polarized radiation waves in the far field.

[0059] The 3dB bridge 1321 can be implemented using a branch-line coupler. A branch-line coupler consists of a rectangular loop structure formed by two quarter-wavelength parallel transmission lines and two quarter-wavelength series transmission lines, with the input signal distributed equally in amplitude and 90 degrees out of phase at the two output ports. In other embodiments, the 3dB bridge 1321 can also be implemented using a combination of a Wilkinson power divider and a 90-degree phase-shift line. The Wilkinson power divider distributes the input signal equally in amplitude and phase to the two output ports, and an additional 90-degree phase-shift line applies a 90-degree phase delay to one of the signals, thus obtaining two equally orthogonal outputs. Compared to the branch-line coupler, the combination of the Wilkinson power divider and the 90-degree phase-shift line offers more flexibility in layout, but its bandwidth is slightly narrower. Both implementations are methods of implementing the 3dB bridge 1321.

[0060] The orthogonal coupling slot 133 can be a cross-shaped slot, that is, a straight narrow slot is formed on the metal ground plane 131 along two orthogonal directions, and the two narrow slots intersect at the center. In other embodiments, the orthogonal coupling slot 133 can also be two mutually perpendicular H-shaped slots, with a transverse stub loaded at the end of the H-shaped slot. The transverse stub increases the equivalent electrical length of the slot, so that the slot can achieve effective coupling at a lower frequency, thereby widening the coupling bandwidth.

[0061] Figure 5 A top-view schematic diagram of the array layout of a circularly polarized stacked phased array antenna according to some embodiments of this application is shown. Figure 4 As shown, the circularly polarized stacked phased array antenna includes multiple circularly polarized stacked antenna elements 100, which are arranged in an array according to a triangular lattice. The center-to-center spacing d between adjacent circularly polarized stacked antenna elements 100 is less than 0.54 times the free-space wavelength λ0 corresponding to the highest operating frequency of the circularly polarized stacked phased array antenna. For example, when the highest operating frequency is 40 GHz, the free-space wavelength λ0 = c / f = 3 × 10⁻⁶. 8 / (40×10 9 =7.5mm, and 0.54 times the free space wavelength is 7.5 × 0.54 = 4.05mm. Therefore, the center-to-center distance d between adjacent antenna elements 100 should be controlled within 4.05mm.

[0062] Triangular lattice arrangements offer advantages in grating lobe suppression compared to rectangular lattice arrangements. With a rectangular lattice arrangement, grating lobes of the array factor enter the visible region earlier when the beam scans to large angles. The triangular lattice arrangement allows adjacent rows of antenna elements to be staggered, pushing the critical scanning angle for grating lobe appearance to a larger angle under the same element spacing. Combined with the surface wave suppression capability of the SIW via array 140 of each antenna element 100, the circularly polarized stacked phased array antenna can eliminate grating lobes within a scanning range of ±60°. The spacing upper limit of 0.54 times the free space wavelength is derived from the grating lobe equation of the triangular lattice—when the element spacing exceeds this threshold, grating lobes will appear within a scanning range of ±60°. Furthermore, the triangular lattice arrangement allows for approximately 15% more antenna elements per unit area than the rectangular arrangement, which is beneficial for improving array gain.

[0063] The triangular lattice can be an equilateral triangular lattice, where the rows are equally spaced and staggered, and the row spacing and column spacing satisfy the geometric relationship of an equilateral triangle. In other embodiments, the triangular lattice can also be an isosceles triangular lattice, where the row spacing and column spacing are not equal, and can be adjusted according to the outer contour shape of the antenna array and the installation space constraints. Both lattice forms are implementations of triangular lattice arrangements.

[0064] Figure 6 A general flowchart of a beam control method for a circularly polarized stacked phased array antenna according to some embodiments of this application is shown. Figure 5 As shown, the beam control method is applied to a circularly polarized stacked phased array antenna, including steps S1 to S4.

[0065] In step S1, the scanning angle and current operating frequency of the target beam are obtained. As an example, the scanning angle can be given by the beam pointing control command of the satellite platform, represented in spherical coordinates as a combination of elevation angle θ and azimuth angle φ. The current operating frequency can be determined by frequency planning in the communication system. For example, in a specific operating scenario, the scanning angle of the target beam is θ=50°, φ=30°, and the current operating frequency is 30GHz.

[0066] In step S2, the amplitude and phase compensation values ​​corresponding to each circularly polarized stacked antenna element 100 are determined based on the scanning angle and the current operating frequency. In conventional phased array beam scanning, the beam control computer calculates the required phase gradient for each element based solely on the scanning angle, and applying the phase gradient to each element achieves beam pointing deflection. However, the phase gradient only controls the beam pointing and cannot compensate for polarization distortion caused by inter-element coupling and manufacturing errors. Under wide-angle scanning conditions, the amplitude and phase imbalance caused by mutual coupling will disrupt the equal amplitude orthogonality condition of the feed network output, leading to a decrease in circular polarization purity. Therefore, the amplitude and phase compensation values ​​determined in step S2 not only include the phase gradient required for beam scanning but also include additional amplitude and phase pre-distortion amounts used to compensate for polarization distortion.

[0067] In step S3, the initial radio frequency (RF) signal to be transmitted is acquired. The amplitude and phase compensation values ​​are used to modulate the initial RF signal input to each circularly polarized stacked antenna element 100 and perform phase shifting to generate a compensated RF signal. As an example, the amplitude modulation and phase shifting are performed by the digitally controlled attenuator and digitally controlled phase shifter integrated in the T / R (transmit / receive) components corresponding to each antenna element 100. The beam control computer converts the amplitude and phase compensation values ​​determined in step S2 into attenuator control codes and phase shifter control codes, which are then sent to each T / R component. The digitally controlled attenuator adjusts the amplitude of the RF signal according to the amplitude compensation amount, and the digitally controlled phase shifter adjusts the phase of the RF signal according to the phase compensation amount. The two work together to output the compensated RF signal.

[0068] In step S4, the compensated radio frequency (RF) signal is injected into the feed network 132 of each circularly polarized stacked antenna element 100. The compensated RF signal is split into two orthogonal signals with equal amplitude and a 90-degree phase difference by a 3dB bridge in the feed network 132. The two orthogonal signals are transmitted along microstrip feed lines 1322a and 1322b to the horizontal and vertical arms of the orthogonal coupling slot 133, respectively, and electromagnetic energy is transferred to the main radiating patch 121 through slot coupling. Two orthogonal degenerate modes are excited on the main radiating patch 121, and the two degenerate modes become circularly polarized radiation waves at a distance. The electromagnetic coupling between the parasitic radiating patch 111 and the main radiating patch 121 further broadens the radiation bandwidth. The circularly polarized waves radiated by each antenna element 100 are vector-synthesized in space according to the amplitude and phase distribution applied in steps S2 and S3 to form a high-purity circularly polarized beam pointing towards the target scanning angle.

[0069] During the execution of steps S1 to S4 above, the metallized via array 140 of each antenna element 100 continuously plays a surface wave suppression role. When the beam deflects to a large angle, the surface wave component coupled into the dielectric substrate in the radiated energy of each element increases. The shielding cavity formed by the metallized via array 140 short-circuits the laterally propagating surface waves to the metal ground plane 131, blocking the mutual coupling channel of surface waves between adjacent elements, maintaining the stability of the active radiation pattern of each element, thereby preventing the abnormal rise of cross-polarized sidelobes at the spatial synthesis level.

[0070] In some embodiments, step S2 includes sub-step S21. In sub-step S21, a preset multidimensional error compensation matrix is ​​queried to obtain the amplitude and phase predistortion compensation value corresponding to the scanning angle and the current operating frequency, and the amplitude and phase predistortion compensation value is used as the amplitude and phase compensation value. The multidimensional error compensation matrix records the amplitude and phase predistortion compensation values ​​of each circularly polarized stacked antenna element 100 under different scanning angles and different frequency combinations.

[0071] The data structure of the multidimensional error compensation matrix can be described as follows. The matrix uses the scanning angle (θ, φ) and frequency f as index dimensions. Each index node corresponds to a specific combination of scanning angle and frequency. Each index node stores N sets of amplitude and phase predistortion compensation values, where N is the total number of antenna elements 100 in the circularly polarized stacked phased array antenna. Each set of amplitude and phase predistortion compensation values ​​includes an amplitude compensation amount and a phase compensation amount.

[0072] For example, assuming the scanning angle set covers the range of 0° to 60° in the θ direction with a step size of 5° (a total of 13 discrete values), and covers the range of 0° to 345° in the φ direction with a step size of 15° (a total of 24 discrete values), and the frequency point set covers the range of 26.5GHz to 40GHz with a step size of 500MHz (a total of 28 discrete values), then the multidimensional error compensation matrix contains a total of 13×24×28=8736 traversal nodes, and each traversal node stores N sets of amplitude and phase predistortion compensation values. During the query process in step S21, the beam control computer locates the corresponding traversal node in the matrix based on the scanning angle θ=50°, φ=30° and the operating frequency f=30GHz obtained in step S1, and reads the amplitude compensation and phase compensation values ​​of each of the N antenna elements under that node.

[0073] In some embodiments, when the actual scanning angle and frequency do not precisely hit the discrete traversal nodes in the matrix, bilinear interpolation between adjacent traversal nodes can be used to obtain the amplitude and phase predistortion compensation value. For example, if the actual scanning angle θ=52° is located between two discrete nodes θ=50° and θ=55° in the matrix, the beam control computer reads the amplitude and phase predistortion compensation values ​​under these two nodes respectively, and performs linear interpolation according to the position of θ=52° in the interval between 50° and 55° to obtain the amplitude and phase predistortion compensation value corresponding to θ=52°.

[0074] In some embodiments, the multidimensional error compensation matrix is ​​constructed through the following pre-calibration steps.

[0075] In step S01, the circularly polarized stacked phased array antenna is controlled to sequentially traverse within a preset set of scanning angles and frequencies, acquiring the active VSWR data of each circularly polarized stacked antenna element 100 at each traversal node and the spatial mutual coupling characteristic parameters between adjacent circularly polarized stacked antenna elements 100. Each combination of scanning angle and frequency constitutes a traversal node. As an example, the calibration process is performed in a microwave anechoic chamber or a near-field antenna test system. At each traversal node, the beam control computer sequentially excites each antenna element 100, and measures the port reflection coefficient of the excited element using a vector network analyzer (thereby calculating the active VSWR), while simultaneously measuring the amplitude and phase of the coupled signals received at the ports of the remaining elements (thereby obtaining the spatial mutual coupling characteristic parameters).

[0076] In step S02, a cross-coupling scattering matrix is ​​constructed based on the active VSWR data and spatial cross-coupling characteristic parameters. The cross-coupling scattering matrix is ​​an N×N matrix, where the element in the i-th row and j-th column is... This represents the amplitude and phase of the coupled signal generated at the port of the i-th antenna element 100 when the j-th antenna element 100 is excited. Diagonal elements The port reflection coefficients of each unit are directly related to the active VSWR data. Off-diagonal elements. Spatial mutual coupling characteristic parameters between adjacent and non-adjacent units.

[0077] Based on the mutual coupling scattering matrix, the radiation pattern function of the array active elements under each traversal node is calculated. The difference between the radiation pattern of an active element and that of an isolated element is that the radiation pattern of an active element includes the perturbation of the radiation pattern by mutual coupling in the array environment. Specifically, when the k-th antenna element 100 is excited, although the other elements are terminated with matched loads, due to the existence of mutual coupling, some energy is radiated into space again from the radiating patches of adjacent elements through the scattering path, superimposed on the direct radiation of the k-th element, forming the active element radiation pattern.

[0078] Polarization decomposition is performed on the radiation pattern function of the array active cells to extract the principal polarization components and cross-polarization components. The polarization decomposition operation involves projecting the far-field vector of the active pattern onto two orthogonal circular polarization basis vectors—the projection along the left-hand circular polarization (LHCP) basis vector is the principal polarization component, and the projection along the right-hand circular polarization (RHCP) basis vector is the cross-polarization component (taking the system with left-hand circular polarization as the desired polarization as an example).

[0079] The polarization distortion vector is generated based on the deviation between the main polarization component and the preset ideal circular polarization reference vector, as well as the cross-polarization components. The ideal circular polarization reference vector is the main polarization far-field vector that the antenna element should output at the corresponding scanning angle and frequency point under conditions of no mutual coupling and no manufacturing errors. The amplitude difference between the main polarization component and the ideal circular polarization reference vector is the amplitude deviation. The phase difference is the phase deviation. The normalized amplitude of the cross-polarization component is the cross-polarization crosstalk. Amplitude deviation Phase deviation and cross-polarization crosstalk Combine the vectors to generate the polarization distortion vector of the k-th antenna element 100 under the traversal node.

[0080] In step S03, based on the polarization distortion vector, the inverse amplitude and phase weights that minimize the overall axial ratio of the array are solved. The solution process can be described as an optimization problem—using the amplitude and phase compensation amounts of each of the N antenna elements as optimization variables, and the axial ratio of the array's far-field synthetic radiation in the target scanning direction as the objective function, the combination of amplitude and phase compensation amounts that minimizes the objective function is solved. This optimization problem can be solved using the least squares method or a convex optimization method. The solved inverse amplitude and phase weights are written into the multidimensional error compensation matrix as the amplitude and phase pre-distortion compensation values ​​for the corresponding traversal nodes.

[0081] The above describes an implementation method for constructing a multidimensional error compensation matrix using calibration based on measured data. In other embodiments, the multidimensional error compensation matrix can also be constructed based on full-wave electromagnetic simulation data. Specifically, a complete array simulation model of a circularly polarized stacked phased array antenna is established using three-dimensional electromagnetic simulation software (such as HFSS or CST). The scanning angle set and frequency point set are traversed in the simulation environment to extract the active VSWR data and mutual scattering parameters obtained from the simulation. The same polarization decomposition and optimization solution process as steps S02 and S03 is executed, and the inverse amplitude and phase weights obtained from the simulation are written into the multidimensional error compensation matrix. The advantage of simulation calibration is that it does not depend on the physical testing environment and is suitable for verifying pre-compensation schemes during the antenna design stage. The compensation matrix obtained from simulation calibration can be used as the initial compensation matrix after the physical fabrication is completed, and then refined through measured calibration. Both the calibration method based on measured data and the calibration method based on simulation data belong to the implementation methods of constructing a multidimensional error compensation matrix through pre-calibration steps.

[0082] Step S2, following sub-step S21, also includes sub-steps S22 to S25 for temperature correction of the amplitude-phase pre-distortion compensation value.

[0083] In the spaceborne operating environment, the temperature of the antenna array surface fluctuates within the range of approximately -40°C to +80°C, influenced by the solar illumination angle and the satellite orbital period. The dielectric constant of the third dielectric substrate 130 drifts with temperature; the temperature coefficient of the dielectric constant of a typical microwave dielectric substrate is approximately ±50 ppm / °C. This dielectric constant drift causes changes in the electrical lengths of the microstrip feed lines 1322a and 1322b in the feed network 132, disrupting the 90-degree phase difference accuracy of the 3dB bridge output. This results in the orthogonal signals reaching the two arms of the orthogonal coupling slot 133 deviating from the equal amplitude orthogonality condition, leading to a deterioration in the axial ratio. Therefore, after retrieving the static amplitude and phase predistortion compensation value from the multidimensional error compensation matrix, this compensation value needs to be dynamically corrected based on the real-time temperature of the antenna array surface.

[0084] In sub-step S22, the real-time temperature characteristic value of the circularly polarized stacked phased array antenna is obtained. As an example, multiple temperature sensors are integrated on the antenna array surface, and the real-time temperature characteristic value is obtained by weighted averaging of the readings from each sensor. For example, at a certain moment, the readings of the various temperature sensors are 63°C, 65°C, 67°C, and 66°C, respectively. A weighted average is then used to obtain the real-time temperature characteristic value. .

[0085] In sub-step S23, based on the real-time temperature characteristic value The thermal change dielectric constant curve of the third dielectric substrate 130 in the circularly polarized stacked antenna unit 100 is used to calculate the wavelength drift of the current operating frequency point inside the third dielectric substrate 130.

[0086] In sub-step S24, the phase delay change in the feed network 132 is calculated based on the wavelength drift. The phase delay of the microstrip feed and the microstrip feed line in the feed network 132 is directly related to the wavelength of the dielectric waveguide—when the wavelength of the dielectric waveguide drifts due to temperature changes, the phase shift generated by the microstrip feed line with a fixed physical trace length also changes accordingly. The phase delay change can be calculated based on the wavelength drift and the physical trace length that generates a 90-degree phase difference in the microstrip feed line. For example, the additional physical trace length of microstrip feed line 1322b compared to microstrip feed line 1322a is... (This length corresponds to a 90-degree phase shift at the nominal reference temperature), when the wavelength of the dielectric waveguide drifts. At that time, the phase shift deviation of this segment of the trace from 90 degrees is the phase delay change. .

[0087] In sub-step S25, the phase delay change amount The values ​​are converted into compensation factors and then superimposed on the amplitude-phase pre-distortion compensation values ​​to obtain the final amplitude-phase compensation values. The physical meaning of the compensation factor is—an additional phase adjustment applied to the digitally controlled phase shifter of the T / R component, ensuring that the feed network 132, after temperature drift, can still restore the equal-amplitude orthogonal signal relationship at both arms of the orthogonal coupling gap 133. The final amplitude-phase compensation value simultaneously includes static polarization distortion compensation (from the multidimensional error compensation matrix) and dynamic temperature drift compensation (from the temperature correction step). The superposition of these two components is then uniformly executed by the digitally controlled attenuator and digitally controlled phase shifter in step S3.

[0088] The following example illustrates the complete calculation process of sub-steps S22 to S25 using a set of specific numerical values. Assume a nominal reference temperature. Real-time temperature characteristic value: 25°C At 65°C, the nominal relative permittivity of the third dielectric substrate 130 The dielectric constant is 3.0, the temperature coefficient of dielectric constant α is +50ppm / °C, and the current operating frequency f is 30GHz.

[0089] In some embodiments, sub-step S23 includes sub-steps S231 to S234.

[0090] In sub-step S231, the real-time temperature characteristic value is calculated. With the preset nominal reference temperature The temperature difference ΔT between them. Continuing with the above numerical example, .

[0091] In sub-step S232, the temperature coefficient of dielectric constant α of the third dielectric substrate 130 is obtained, and the real-time equivalent dielectric constant of the third dielectric substrate 130 is calculated based on the temperature difference ΔT and the temperature coefficient of dielectric constant α. Real-time equivalent dielectric constant The calculation method is as follows: [The text abruptly ends here, likely due to an incomplete sentence or a formatting error. The dielectric constant increment is added on top of the increase caused by temperature. Continuing with the numerical example above, the dielectric constant increment is... Real-time equivalent dielectric constant .

[0092] In sub-step S233, based on the current operating frequency f and the real-time equivalent dielectric constant... Calculate the real-time dielectric waveguide wavelength of the microwave signal propagating inside the third dielectric substrate 130. Continuing with the numerical examples above, .

[0093] In sub-step S234, the real-time dielectric waveguide wavelength is... The nominal dielectric waveguide wavelength at the nominal reference temperature T_0 By subtraction, the wavelength shift can be obtained. Nominal dielectric waveguide wavelength Wavelength shift A negative value indicates that the dielectric constant increases with increasing temperature, and the wavelength of the guided wave in the dielectric shortens.

[0094] It should be understood that the linear approximation model used in sub-step S232 is applicable to operating conditions where the temperature difference ΔT is within approximately 80°C. When the temperature difference exceeds 80°C, a nonlinear relationship may exist between the dielectric constant and temperature. In this case, piecewise linear interpolation or a lookup table method can be used instead of linear calculation of a single temperature coefficient. Specifically, the measured dielectric constant values ​​of the third dielectric substrate 130 material at multiple temperature points are pre-measured, and the measurement results are stored as a lookup table. In sub-step S232, the model is applied based on the real-time temperature characteristic value. The real-time equivalent dielectric constant is obtained by interpolation in the lookup table. Both the linear approximation model and the lookup table method are implementation methods for calculating the real-time equivalent dielectric constant based on the temperature difference and the temperature coefficient of dielectric constant.

[0095] It should be understood that the sequence number of each step in the above embodiments does not imply the order of execution. The execution order of each process should be determined by its function and internal logic, and should not constitute any limitation on the implementation process of the embodiments of the present invention.

[0096] Those skilled in the art will clearly understand that, for the sake of convenience and brevity, the above-described division of functional units and modules is used as an example. In practical applications, the above functions can be assigned to different functional units and modules as needed, that is, the internal structure of the device can be divided into different functional units or modules to complete all or part of the functions described above.

[0097] The above embodiments are only used to illustrate the technical solutions of the present invention, and are not intended to limit it. Although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of the embodiments of the present invention, and should all be included within the protection scope of the present invention.

Claims

1. A circularly polarized stacked antenna element, characterized in that, It includes a first dielectric substrate, a second dielectric substrate, and a third dielectric substrate arranged from top to bottom; The upper surface of the first dielectric substrate is provided with a parasitic radiation patch; The upper surface of the second dielectric substrate is provided with a main radiating patch, and the main radiating patch and the parasitic radiating patch are coaxially aligned in the vertical direction; The upper surface of the third dielectric substrate is provided with a metal ground plane, and the lower surface of the third dielectric substrate is provided with a power supply network. An orthogonal coupling gap is opened on the metal ground plane at the vertical projection center of the main radiating patch. The first dielectric substrate and the second dielectric substrate are provided with a through-hole array of metallized vias. The array of metallized vias is arranged around the periphery of the parasitic radiating patch and the main radiating patch. The array of metallized vias is electrically connected to the metal ground plane to form a shielding cavity around the main radiating patch and the parasitic radiating patch. The power supply network is configured to output two orthogonal signals with equal amplitude and a 90-degree phase difference, which are coupled to the main radiating patch through the orthogonal coupling gap.

2. The circularly polarized stacked antenna element according to claim 1, characterized in that, The metallized via array includes multiple metallized vias arranged in a ring at equal intervals, and the spacing between adjacent metallized vias is less than one-quarter of the wavelength in the medium corresponding to the highest operating frequency of the circularly polarized stacked antenna element.

3. The circularly polarized stacked antenna element according to claim 1, characterized in that, Both the main radiating patch and the parasitic radiating patch are centrally symmetrical, and the area of ​​the parasitic radiating patch is smaller than the area of ​​the main radiating patch.

4. The circularly polarized stacked antenna element according to claim 1, characterized in that, The power supply network includes a 3dB bridge and two orthogonally arranged microstrip feed lines. The two output terminals of the 3dB bridge are respectively connected to the two microstrip feed lines, and the two microstrip feed lines correspond to the two orthogonal directions of the orthogonal coupling gap.

5. A circularly polarized stacked phased array antenna, characterized in that, include: A plurality of circularly polarized stacked antenna elements as described in any one of claims 1 to 4, wherein the plurality of circularly polarized stacked antenna elements are arranged in an array according to a triangular lattice; The center-to-center spacing of adjacent circularly polarized stacked antenna elements is less than 0.54 times the free-space wavelength corresponding to the highest operating frequency of the circularly polarized stacked phased array antenna.

6. A beam control method for a circularly polarized stacked phased array antenna, characterized in that, The method applied to the circularly polarized multilayer phased array antenna as described in claim 5 includes the following steps: S1. Obtain the scanning angle and current operating frequency of the target beam; S2. Determine the amplitude and phase compensation value corresponding to each of the circularly polarized stacked antenna elements based on the scanning angle and the current operating frequency. S3. Obtain the initial radio frequency signal to be transmitted, and use the amplitude and phase compensation value to perform amplitude modulation and phase shifting on the initial radio frequency signal input to each of the circularly polarized stacked antenna elements to generate a compensated radio frequency signal; S4. The compensated radio frequency signal is injected into the feed network of each of the circularly polarized stacked antenna elements. The compensated radio frequency signal is converted into two orthogonal signals with equal amplitude and a 90-degree phase difference by the feed network, and then coupled to the main radiating patch of the circularly polarized stacked antenna element through the orthogonal coupling gap.

7. The beam control method according to claim 6, characterized in that, S2 includes: S21. Query the preset multidimensional error compensation matrix to obtain the amplitude and phase predistortion compensation value corresponding to the scanning angle and the current working frequency, and use the amplitude and phase predistortion compensation value as the amplitude and phase compensation value; wherein, the multidimensional error compensation matrix records the amplitude and phase predistortion compensation value of each of the circularly polarized stacked antenna elements under different scanning angles and different frequency combinations.

8. The beam control method according to claim 7, characterized in that, The multidimensional error compensation matrix is ​​constructed through the following pre-calibration steps: S01. Control the circularly polarized stacked phased array antenna to traverse through a preset set of scanning angles and frequency points one by one, and obtain the active standing wave ratio data of each circularly polarized stacked antenna element under each traversal node and the spatial mutual coupling characteristic parameters between adjacent circularly polarized stacked antenna elements; wherein, each combination of the scanning angle and the frequency point constitutes a traversal node. S02. Based on the active VSWR data and the spatial mutual coupling characteristic parameters, construct a mutual coupling scattering matrix, calculate the array active element pattern function under each of the traversal nodes, and perform polarization decomposition on the array active element pattern function to extract the main polarization component and the cross polarization component. Generate a polarization distortion vector based on the deviation between the main polarization component and the preset ideal circular polarization reference vector and the cross polarization component. S03. Based on the polarization distortion vector, solve for the inverse amplitude and phase weights that minimize the overall axial ratio of the array, and write the inverse amplitude and phase weights as the amplitude and phase pre-distortion compensation values ​​under the corresponding traversal nodes into the multidimensional error compensation matrix.

9. The beam control method according to claim 7, characterized in that, Following S21, step S2 further includes a step of temperature correction for the amplitude-phase pre-distortion compensation value: S22. Obtain the real-time temperature characteristic value of the circularly polarized stacked phased array antenna; S23. Based on the real-time temperature characteristic value and the thermal change dielectric constant curve of the third dielectric substrate in the circularly polarized stacked antenna unit, calculate the wavelength drift of the current operating frequency point inside the third dielectric substrate. S24. Calculate the phase delay change in the feed network based on the wavelength drift; S25. The phase delay change is converted into a compensation factor, and the compensation factor is added to the amplitude-phase predistortion compensation value to obtain the final amplitude-phase compensation value.

10. The beam control method according to claim 9, characterized in that, S23 includes the following sub-steps: S231. Calculate the temperature difference between the real-time temperature characteristic value and the preset nominal reference temperature; S232. Obtain the temperature coefficient of dielectric constant of the third dielectric substrate, and calculate the real-time equivalent dielectric constant of the third dielectric substrate based on the temperature difference and the temperature coefficient of dielectric constant; S233. Calculate the real-time dielectric waveguide wavelength for microwave signal transmission inside the third dielectric substrate based on the current operating frequency and the real-time equivalent dielectric constant. S234. The wavelength drift is obtained by subtracting the real-time dielectric waveguide wavelength from the nominal dielectric waveguide wavelength at the nominal reference temperature.

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