A broadband filter dielectric resonator antenna with wide harmonic suppression characteristics and a method for controlling its radiation characteristics.

Abstract

This invention belongs to the field of microwave antenna and RF front-end circuit technology, specifically relating to a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics and a method for controlling its radiation characteristics. The method includes the following steps: obtaining an initial electromagnetic state parameter set and performing fundamental mode field distribution decoupling calculations; constructing an open-loop feeder partially coupled topology to generate a hybrid broadband response signal set; parsing the passband envelope and attenuation determination to map a folded open-loop resonator to generate lower stopband zeros; determining the harmonic frequency band energy density threshold to trigger a multi-topology notch filter to generate an upper stopband zero array; and compensating for phase group delay and gain flatness to output the converged final radiation state parameters of the antenna. This invention overcomes the problems of multimode interference between the feed network and the radiator, the inability to independently generate stopband radiation zeros, and nonlinear phase group delay shift caused by the lack of distributed notch filter node cooperative configuration.

Description

Technical Field

[0001] This invention belongs to the field of microwave antenna and radio frequency front-end circuit technology, specifically relating to a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics and a method for controlling its radiation characteristics. Background Technology

[0002] Broadband dielectric resonator antennas typically use a microstrip feed network on a dielectric substrate to excite a stepped dielectric pillar. Frequency-selective transmission is achieved by loading parasitic resonant elements or cascading external filter networks at both ends of the passband. Conventional designs rely on impedance matching between independent filters and radiating elements, using coupled slot lines to transfer electromagnetic energy to the bottom of the dielectric resonator, and leveraging the fundamental mode splitting mechanism to extend the operating bandwidth. Existing harmonic suppression schemes often involve etching periodic structures on the ground plane or adding microstrip bandstop elements around the antenna to alter the surface current distribution path.

[0003] Problems with existing technology: In broadband and multi-band coexistence scenarios, the existing architecture lacks a coordinated excitation topology between the open-loop feeder and the coupled slotted line, as well as the cascaded configuration of folded open-loop resonators, lateral stubs, spur lines, and double "H"-shaped bandstop structures on the main transmission path. This leads to multimode interference in the electromagnetic energy flow between the feeder network and the radiator. The lack of distributed notch nodes prevents the independent generation of upper and lower stopband radiation nulls within the feeder link, resulting in a nonlinear shift in the phase group delay at the passband edge. The additional distributed parameters introduced by the external cascaded network disrupt the symmetry of the fundamental mode electric field splitting, causing current phase reversal failure in higher harmonic bands, resulting in insufficient stopband attenuation slope and fluctuations in the main lobe width of the far-field radiation pattern. Summary of the Invention

[0004] The purpose of this invention is to provide a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics and a method for controlling its radiation characteristics. This method can overcome the problems of multimode interference between the feed network and the radiator, the inability to independently generate stopband radiation nulls, and the nonlinear shift of phase group delay caused by the lack of distributed notch node cooperative configuration.

[0005] The specific technical solution adopted by this invention is as follows: A method for controlling the radiation characteristics of a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics includes the following steps: Obtain the initial electromagnetic state parameter set of the dielectric resonator and the overall feed line, perform fundamental mode field distribution decoupling operation, and output impedance matching eigenvector and target frequency band ground state energy distribution; An open-loop feeder partial coupling topology is constructed, and the electromagnetic excitation signal is transmitted through the open-loop feeder part to the coupling slot line region etched on the ground plane. The coupling field is injected into the bottom of the dielectric resonator to generate a hybrid broadband response signal set containing the fundamental mode split state and the slot line transmission mode. The passband envelope analysis and edge attenuation slope determination are performed on the hybrid broadband response signal set. When it is found that the attenuation characteristics of the upper stopband frequency domain do not meet the first preset condition, a folded open-loop resonator is mapped at the symmetrical node of the open-loop feed section to generate the first lower stopband radiation zero and the extended resonant frequency domain interval. When the energy density distribution of the harmonic frequency band in the mixed broadband response signal set exceeds the second preset threshold, the multi-topology notch control process is triggered. A transverse stub, a spur line and a double "H" type bandstop structure are sequentially configured at the specified coupling position of the main transmission path of the overall feeder to generate an upper stopband radiation zero-point array. Based on the frequency domain superposition relationship between the first lower stopband radiation null and the upper stopband radiation null array, a linearization compensation operation is performed on the phase group delay and gain flatness in the passband, and the final radiation state parameters of the antenna after convergence are output.

[0006] According to another aspect of the present invention, the step of performing the fundamental mode field distribution decoupling operation and outputting the impedance matching eigenvector and the target frequency band ground state energy distribution specifically includes the following steps: Extract the stepped geometric profile parameters of the dielectric resonator and the spatial coordinate set of the cuboid groove dug at the bottom to construct a three-dimensional dielectric constant gradient distribution model; The three-dimensional dielectric constant gradient distribution model is mapped to the excitation port of the full-wave electromagnetic field iterative algorithm to calculate the electric field distortion rate of the TE fundamental mode under the perturbation of the cuboid slot. When the electric field distortion rate reaches the preset splitting threshold, the splitting resonant frequency combination is locked, and the impedance matching feature vector and the target frequency band ground state energy distribution are output.

[0007] According to another aspect of the present invention, the generation of a hybrid broadband response signal set including a fundamental mode split state and a slotted line transmission mode specifically includes the following steps: Extract the physical tilt trend of the two branches of the open-loop feeder extending towards the ground plane and the characteristic impedance of the transmission line. The overlap range between the open-loop resonant frequency and the transmission cutoff frequency of the coupled slot line is calculated based on the physical tilt trend and characteristic impedance. The electromagnetic energy distribution within the overlapping interval is vector superimposed to eliminate the phase cancellation blind zone and output the hybrid broadband response signal set.

[0008] According to another aspect of the present invention, a folded open-loop resonator is mapped at the symmetrical node of the open-loop feed section to generate a first lower stopband radiation zero and an extended resonant frequency domain interval, specifically including the following steps: Obtain the current standing wave node positions on both sides of the open-loop feed section to determine the electrical connection anchor point of the folded open-loop resonator; Adjust the number of folds and coupling gap parameters of the folded open-loop resonator to establish a parameter mapping table of branch equivalent capacitance and equivalent inductance; Based on the parameter mapping table, the branch resonant frequency is adjusted to the target frequency point of the lower stopband. When the branch reflection phase and the main feed path form an antiphase superposition, the first lower stopband radiation zero point is solidified.

[0009] According to another aspect of the present invention, a transverse stub, a spur line, and a double "H"-shaped bandstop structure are sequentially configured at a designated coupling position on the main transmission path of the overall feeder to generate an upper stopband radiation null array, specifically including: The spur line is etched in the high energy concentration region of the main transmission path of the overall feeder, and the equivalent series inductance parameter of the spur line is extracted. The lateral stubs are positioned in the upstream coupling region of the spur line, and the stub arm length is adjusted to match the quarter-wavelength resonance condition of the target harmonic frequency. The double "H"-shaped bandstop structure is configured in the downstream isolation zone of the spur line. By adjusting the bridging width and transverse slot depth of the double "H"-shaped bandstop structure, the attenuation depth of different order harmonic frequencies is independently tuned, and the upper stopband radiation zero-point array is output.

[0010] According to another aspect of the present invention, the method further includes a resonance overlap anomaly handling procedure: Real-time monitoring of the frequency domain distribution of the first lower stopband radiation null and the upper stopband radiation null array; When the calculated frequency domain spacing between adjacent zeros is less than the third preset threshold, or the passband edge insertion loss fluctuation exceeds the fourth preset threshold, it is determined to be an abnormal resonance overlap. The topology compensation mechanism is triggered to adaptively fine-tune the end load impedance of the lateral branches or the width of the center connecting bridge of the double "H" type band-stop structure until the isolation of adjacent zeros is restored to the allowable range, and the hybrid broadband response signal set is regenerated.

[0011] According to another aspect of the present invention, the step of performing linearization compensation operation on the phase group delay and gain flatness within the passband and outputting the converged final radiation state parameters of the antenna specifically includes the following steps: Within the target wideband operating range, sample the far-field radiation phase values ​​and the main lobe width of the radiation pattern at multiple discrete frequency points; Construct a radiation pattern distortion correction matrix and perform spatial weighted smoothing on the main lobe width of the radiation pattern; Combining the root mean square error of gain flatness with the linearity index of phase group delay, multi-objective optimization iteration is performed. When all indices fall within the preset tolerance window, the converged antenna final radiation state parameters are output.

[0012] According to another aspect of the present invention, an antenna electromagnetic control and layout generation system is also provided, including a storage module and a processing module, wherein the processing module is configured to execute the radiation characteristic control method described in any one of the preceding embodiments and output the etching layout parameter file of the overall feed line and ground plane.

[0013] According to another aspect of the present invention, a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics is also provided, comprising a dielectric resonator, a ground plane, a dielectric substrate and an integral feed line stacked sequentially from top to bottom; The ground plane facing the dielectric resonator has a corresponding coupling groove line etched on it, and the projection area of ​​the coupling groove line covers the bottom of the dielectric resonator. The integral feed line is printed on the side of the dielectric substrate away from the ground plane, and includes an open-loop feed line portion. The two ends of the open-loop feed line portion extend toward both sides of the ground plane and are electromagnetically coupled to the dielectric resonator through the coupling groove line. The overall feed line also integrates a folded open-loop resonator, a spur wire, a lateral stub, and a double "H" type band-stop structure; the folded open-loop resonator is connected to both sides of the open-loop feed line section, and the lateral stub, spur wire, and double "H" type band-stop structure are arranged sequentially on the main feed branch along the signal transmission direction of the overall feed line.

[0014] According to another aspect of the present invention, the dielectric resonator has a stepped structure, with the cuboid groove formed by the inward indentation on both sides of its bottom surface; The two arm segments of the open-loop feed line extend toward the double "H"-shaped band-stop structure and are symmetrically distributed with the folded open-loop resonator on the overall feed line plane. The spur line maintains a preset isolation gap with the lateral branch at one end, and the double "H"-shaped band stop structure is electrically connected to the main transmission conductor of the overall feeder through the central bridging section. The etching trajectory of the spur line and the edge of the main transmission conductor have a preset coupling distance.

[0015] The technical effects achieved by this invention are as follows: This invention utilizes the coordinated vertical projection of the bottom cuboid slot of the stepped dielectric resonator, the open-loop feed section, and the ground plane coupling slot line to cause directional distortion of the boundary conditions of the fundamental mode TE electric field and split it into multi-frequency resonant states. At the same time, it promotes the energy overlap between the feed microstrip mode and the slot line transmission mode in the frequency domain, thereby realizing the autonomous expansion of impedance bandwidth and the in-situ generation of zero radiation points at the passband edge under a single-stage feed structure. This avoids the additional distributed parameters and insertion loss introduced by the cascading of external filter networks and reduces the cascading complexity of the RF front-end system.

[0016] This invention cuts off the DC path and forms a controllable capacitive coupling node by pre-setting an isolation gap between the spur line and the lateral branch. Combined with the distributed cascade configuration of the folded open-loop resonator, lateral branch, spur line and double "H" type bandstop structure on the main transmission path of the overall feeder, the surface current path of the target harmonic frequency band undergoes an equivalent series inductance abrupt change and reflection phase reversal. Then, multiple frequency-controllable radiation null arrays are independently synthesized in the lower stopband and upper stopband, realizing distributed interception and high-selectivity out-of-band suppression of high-order harmonic energy in a wide frequency range, and widening the spectrum anti-interference tolerance without increasing the physical size of the antenna.

[0017] This invention uses passband envelope analysis and edge attenuation slope determination logic to trigger topology mapping, and combines iterative linearization compensation of phase group delay and gain flatness to dynamically correct and converge the frequency domain spacing deviation of adjacent resonant zeros and passband phase nonlinear residuals to a preset tolerance range. This eliminates far-field pattern main lobe distortion and cross-polarization level rise caused by multimode resonance overlap, ensuring the axial stability of the radiating main lobe and port impedance matching consistency of the antenna in the broadband operating range. Attached Figure Description

[0018] Figure 1 This is a three-dimensional structural schematic diagram of the antenna of the present invention; Figure 2 This is a front view of the antenna of the present invention; Figure 3 This is a side view of the antenna of the present invention; Figure 4 This is a top view of the antenna of the present invention; Figure 5 The diagram shows the S11 parameters and gain performance of the antenna of this invention. Figure 6 This is a schematic diagram of the XOZ and YOZ radiation pattern distribution of the antenna of the present invention at a typical resonant point in the passband; Figure 7 This is a flowchart illustrating the radiation characteristic modulation method of the present invention. Figure 8 This is a diagram of the broadband filter dielectric resonator antenna stacked topology of the present invention.

[0019] The attached diagram lists the components represented by each number as follows: 1. Dielectric resonator; 2. Ground plane; 3. Dielectric substrate; 4. Integral feed line; 5. Cuboid slot; 6. Coupled slot line; 7. Open-loop feed line section; 8. Folded open-loop resonator; 9. Spur line; 10. Lateral stub; 11. Double "H" type band-stop structure. Detailed Implementation

[0020] To make the objectives and advantages of this invention clearer, the invention will be specifically described below with reference to embodiments. It should be understood that the following text is merely used to describe one or more specific embodiments of the invention and does not strictly limit the scope of protection specifically claimed by the invention.

[0021] It should be noted that the terms "first," "second," etc., used in this invention are used to distinguish similar objects and are not necessarily used to describe a specific order or sequence. It should be understood that such data can be interchanged where appropriate so that the embodiments of the invention described herein can be implemented in orders other than those illustrated or described herein. Furthermore, the terms "comprising" and "having," and any variations thereof, are intended to cover non-exclusive inclusion; for example, a process, method, system, product, or apparatus that comprises a series of steps or units is not necessarily limited to those steps or units explicitly listed, but may include other steps or units not explicitly listed or inherent to such processes, methods, products, or apparatus.

[0022] According to an embodiment of the present invention, a method embodiment for controlling the radiation characteristics of a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics is provided. It should be noted that the steps shown in the flowchart in the accompanying drawings can be executed in a computer system such as a set of computer-executable instructions. Furthermore, although a logical order is shown in the flowchart, in some cases, the steps shown or described may be executed in a different order than that shown here.

[0023] like Figure 7 As shown, a method for controlling the radiation characteristics of a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics includes the following steps: S1. Obtain the initial electromagnetic state parameter set of dielectric resonator 1 and overall feeder 4, perform fundamental mode field distribution decoupling operation, and output impedance matching characteristic vector and target frequency band ground state energy distribution.

[0024] In step S1 above, during execution, the control unit reads the three-dimensional geometric parameters of the dielectric resonator 1 and the microstrip line shape parameters of the overall feed line 4, and compiles them into an initial electromagnetic state parameter set. The antenna electromagnetic control and layout generation system then calls the three-dimensional full-wave solver engine, inputs the parameter set into the electromagnetic field iterative model, extracts eigenvalues ​​of the fundamental mode electric field distribution at the dielectric boundary, performs fundamental mode field distribution decoupling operations, generates an impedance matching eigenvector characterizing the port matching state, and outputs a spatial distribution map of the ground state energy within the target frequency band. Based on this, if there are manufacturing tolerances in the dielectric constant of the dielectric, the system can introduce multiple sets of equivalent dielectric parameters for Monte Carlo perturbation, ensuring that the output parameter set covers the actual manufacturing tolerance range, thereby providing a robust benchmark for subsequent topology determination.

[0025] S2. Construct an open-loop feeder section 7 coupling topology, transmit the electromagnetic excitation signal through the open-loop feeder section 7 to the coupling slot line 6 region etched on the ground plane 2, inject the coupling field into the bottom of the dielectric resonator 1, and generate a mixed broadband response signal set containing the fundamental mode split state and the slot line transmission mode.

[0026] Furthermore, based on the decoupled energy distribution, the system generates the trace data of the open-loop feeder section 7, establishes a near-field coupling gap with the etched area of ​​the ground plane 2, and transmits the radio frequency signal along the trace to the projection area of ​​the coupling groove line 6, injecting a vertical coupling field into the bottom of the dielectric resonator 1 through edge field penetration.

[0027] The transmission characteristics and open-loop resonant characteristics of the coupled slot line 6 overlap in the frequency domain, causing the field vectors of the fundamental mode split state and the slot line transmission mode to be superimposed, eliminating the phase cancellation blind zone and outputting a hybrid broadband response signal set. By using coupled topology construction and field vector superposition, the electromagnetic response of the main feed channel and the slot line cutoff frequency forms a continuous coverage in the frequency domain, thereby realizing the complete generation of the broadband signal set.

[0028] S3. Perform passband envelope analysis and edge attenuation slope determination on the hybrid broadband response signal set. When it is found that the attenuation characteristics in the upper stopband frequency domain do not meet the first preset condition, map the folded open-loop resonator 8 at the symmetrical node of the open-loop feed section 7 to generate the first lower stopband radiation zero point and the extended resonant frequency domain interval.

[0029] Furthermore, the algorithm module performs spectral envelope fitting on the signal set, extracts the power attenuation rate curve transitioning from the passband edge to the stopband, and compares the slope of this curve with a first preset condition. If the absolute value of the attenuation slope is lower than a threshold, it indicates insufficient out-of-band suppression. At this time, the system generates topology mapping instructions at the symmetrical current standing wave nodes on both sides of the open-loop feed section 7, configures the electrical connection anchor points of the folded open-loop resonator 8, and generates the first lower stopband radiation zero and extended resonance frequency domain interval. By using envelope analysis and condition judgment, the out-of-band suppression state is quantized in real time and triggers topology mapping, thereby realizing the directional generation of the lower stopband zero and resonance interval.

[0030] S4. When the energy density distribution of the harmonic frequency band in the mixed broadband response signal set exceeds the second preset threshold, the multi-topology notch control process is triggered. At the designated coupling position of the main transmission path of the overall feeder 4, the transverse stub 10, the spur line 9 and the double "H" type bandstop structure 11 are configured in sequence to generate the upper stopband radiation zero array.

[0031] Furthermore, the control unit performs high-frequency energy integration on the signal set to extract the peak power density in the harmonic frequency domain.

[0032] When the peak value exceeds the second preset threshold, the system determines the risk of harmonic leakage and sends a multi-topology notch control command to the layout generation module. The multi-topology notch control command drives the sequential layout of horizontal branches 10, spur lines 9, and double "H"-shaped bandstop structures 11 on the main transmission path of the overall feeder 4. Through the spatial cascading electrical interconnection characteristics, multiple destructive interference points are generated in the target harmonic frequency band, which are combined to form an upper stopband radiation zero-point array. By employing energy density integration and sequential topology layout, high-frequency harmonic energy is distributedly intercepted, thereby realizing the synthesis of a broadband notch array.

[0033] S5. Based on the frequency domain superposition relationship between the first lower stopband radiation null and the upper stopband radiation null array, perform linearization compensation operation on the phase group delay and gain flatness in the passband, and output the final radiation state parameters of the antenna after convergence.

[0034] Furthermore, the optimization engine extracts the frequency domain distribution coordinates of the upper and lower stopband zeros, constructing a nonlinear impact model of the zero-phase abrupt change on the passband group delay. The system performs least-squares fitting on the phase group delay curve and the gain fluctuation envelope, calculating the residual value deviating from the ideal linear state. By iteratively adjusting the coupling gap and topology size parameters, the residual value gradually converges to a preset tolerance window, ultimately outputting the antenna's final radiation state parameters. By employing residual fitting and parameter iterative compensation, the phase nonlinearity and gain fluctuation at the passband edges are synchronously corrected, thereby achieving a stable convergence output of the final radiation state parameters.

[0035] As an optional embodiment, performing fundamental mode field distribution decoupling operation and outputting impedance matching eigenvectors and target frequency band ground state energy distribution specifically includes the following steps: Extract the stepped geometric profile parameters of dielectric resonator 1 and the spatial coordinate set of the cuboid groove 5 dug at the bottom to construct a three-dimensional dielectric constant gradient distribution model. The three-dimensional dielectric constant gradient distribution model is mapped to the excitation port of the full-wave electromagnetic field iterative algorithm to calculate TE. 111 Electric field distortion rate of the fundamental mode under perturbation of cuboid slot 5; When the electric field distortion rate reaches the preset splitting threshold, the splitting resonant frequency combination is locked, and the output impedance matches the characteristic vector and the ground state energy distribution of the target frequency band.

[0036] Based on the above steps, during step S1, the system calls the layout parsing interface to extract the cross-sectional dimension sequence of the stepped dielectric layer and the center offset and depth data of the cuboid groove 5 on the bottom surface, and maps them into a three-dimensional dielectric constant gradient distribution model. The three-dimensional dielectric constant gradient distribution model is then loaded onto the excitation boundary of the full-wave solver, and the eigenmode sweep algorithm is run to trace the TE. 111 The direction of the electric field vector of the fundamental mode changes at the boundary of the groove. The position offset of the maximum magnitude of the electric field before and after the introduction of the groove and the rate of change of the phase angle are calculated to obtain the value of the electric field distortion rate.

[0037] When the distortion rate is detected to cross a preset splitting threshold, the algorithm automatically marks the corresponding multiple resonant peak frequencies and their phase relationships, combining them into a split resonant frequency set. Based on this frequency set, the S-parameter curve is reconstructed, the port matching state and spatial field strength envelope are extracted, and the impedance matching feature vector and the ground state energy distribution of the target frequency band are output. By employing contour parameter extraction and distortion rate calculation, the perturbation effect of the groove on the fundamental mode electric field is quantified and characterized, thereby achieving precise locking of the splitting frequency combination and accurate output of the energy distribution spectrum.

[0038] As an optional embodiment, generating a hybrid broadband response signal set containing fundamental mode split states and slotted line transmission modes specifically includes the following steps: Extract the physical tilt trend and transmission line characteristic impedance of the two branches of the open-loop feeder section 7 extending towards the two sides of the ground plane 2. The overlap range between the open-loop resonant frequency and the transmission cutoff frequency of the coupled slot line 6 is calculated based on the physical tilt trend and characteristic impedance. The electromagnetic energy distribution within the overlapping region is vector superimposed to eliminate the phase cancellation blind zone and output a hybrid broadband response signal set.

[0039] Further, during step S2, the system obtains the geometric orientation angle and dielectric layer thickness of the two branches through the microstrip line analysis module, calculates their characteristic impedance value based on transmission line theory, quantifies the tilt trend of the branch's extension to ground, substitutes the tilt trend parameter into the resonant frequency estimation model, and corrects the edge field coupling coefficient; it also calculates the slot line's equivalent transmission constant based on the characteristic impedance. The open-loop resonant curve and the slot line's cutoff frequency response are plotted, and the overlapping segment of the two curves in the frequency domain is identified as the overlapping region. The electric and magnetic field vectors of the open-loop mode and the slot line mode are extracted at the overlapping frequency point, and complex domain addition is performed. By adjusting the coupling spacing of the open-loop feed section 7, the phase difference between the two modes approaches the matching angle, avoiding the blind zone formed by field strength cancellation. The superimposed field distribution forms a continuous and smooth energy response, transforming into a set of hybrid broadband response signals in the broadband frequency domain. Through joint modeling of the tilt trend and characteristic impedance, the frequency domain overlapping region of the open-loop mode and the slot line mode is dynamically locked, thereby achieving continuous coverage of the broadband energy response and elimination of the blind zone.

[0040] As an optional embodiment, a folded open-loop resonator 8 is mapped at the symmetrical node of the open-loop feed section 7 to generate a first lower stopband radiation zero and an extended resonant frequency domain interval, specifically including the following steps: Obtain the current standing wave node positions on both sides of the open-loop feeder section 7, and determine the electrical connection anchor point of the folded open-loop resonator 8; Adjust the number of folds and coupling gap parameters of the folded open-loop resonator 8 to establish a parameter mapping table of branch equivalent capacitance and equivalent inductance; Based on the parameter mapping table, the branch resonant frequency is adjusted to the target frequency of the lower stopband. When the branch reflection phase and the main feed path form an antiphase superposition, the first lower stopband radiation zero point is solidified.

[0041] Based on the above steps, during step S3, the high-frequency electromagnetic simulation engine plots the current vector distribution map of the microstrip surface and identifies current amplitude extrema or phase abrupt change points as standing wave nodes. The system anchors the feed pin positioning logic of the folded open-loop resonator 8 to the coordinates of this node. The parameter scanning module iterates through different combinations of folding layers and gaps, recording the equivalent L and C values ​​for each combination, and constructs a two-dimensional parameter mapping table. The parameter set that aligns the LC resonant frequency with the target stopband frequency is retrieved from the mapping table, and the layout dimensions of the folded structure are modified.

[0042] At the target frequency, the phase difference between the reflection coefficient of the folded branch and the phase difference of the main path is measured. When the difference is close to 180 degrees, the signal undergoes destructive interference at the connection point, and a deep valley appears on the S21 curve, which means the first lower stopband radiation zero point is solidified. By using standing wave node positioning and mapping table retrieval tuning, the resonant frequency of the folded branch is accurately matched, thereby realizing the stable solidification of the lower stopband zero point and closed-loop control of impedance matching.

[0043] As an optional embodiment, a transverse stub 10, a spur line 9, and a double "H"-shaped bandstop structure 11 are sequentially configured at designated coupling positions along the main transmission path of the overall feeder 4 to generate an upper stopband radiation null array, specifically including: The spur line 9 is etched in the high energy concentration area of ​​the main transmission path of the overall feeder 4, and the equivalent series inductance parameter of the spur line 9 is extracted. The lateral stub 10 is positioned in the upstream coupling region of the spur line 9, and the stub arm length is adjusted to match the quarter-wavelength resonance condition of the target harmonic frequency. The double "H" type bandstop structure 11 is configured in the downstream isolation area of ​​the spur line 9. By adjusting the bridging width and transverse slot depth of the double "H" type bandstop structure 11, the attenuation depth of different order harmonic frequencies is independently tuned, and the upper stopband radiation zero-point array is output.

[0044] Based on the above, during step S4, on the signal input side of the main transmission path of the overall feeder 4, a horizontal stub 10 pattern perpendicular to the main transmission path is first generated; the stub arm length is adjusted to satisfy the quarter-wavelength resonance condition of the open transmission line at the target harmonic frequency. At a preset isolation gap downstream of the horizontal stub 10, a spur line 9 pattern is etched, ensuring that the end of the spur line 9 facing the horizontal stub 10 maintains a preset isolation gap with the horizontal stub 10, through which edge field coupling impedance is introduced. Subsequently, in the isolation region downstream of the spur line 9, a symmetrical microstrip slot structure is drawn to form a double "H"-shaped bandstop structure 11.

[0045] The width of the central connecting bridge and the depth of the transverse slots on both sides are independently adjusted, allowing the distributed capacitance and parasitic inductance to form depth-controllable notch responses at multiple higher harmonic frequencies. Multiple tuning results are superimposed in the frequency domain to form a wideband upper stopband radiation null array. By employing the sequential arrangement and independent parameter adjustment of these three components, the notch depth and center frequency at different harmonic frequencies are decoupled and controlled, thereby achieving distributed attenuation of wideband harmonic energy.

[0046] As an optional embodiment, a method for controlling the radiation characteristics of a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics further includes a process for handling resonance overlap anomalies: Real-time monitoring of the frequency domain distribution of the first lower stopband radiation null and the upper stopband radiation null array; When the calculated frequency domain spacing between adjacent zeros is less than the third preset threshold, or the passband edge insertion loss fluctuation exceeds the fourth preset threshold, it is determined to be an abnormal resonance overlap. Trigger the topology compensation mechanism to adaptively fine-tune the end load impedance of the lateral branch 10 or the width of the central connecting bridge of the double "H" type band-stop structure 11 until the isolation of adjacent zeros is restored to the allowable range, and regenerate the hybrid broadband response signal set.

[0047] According to the above steps, during the transition between steps S4 and S5, the system continuously collects S-parameter curve data and tracks the drift trajectory of the zero-point frequency of the lower stopband and the center frequency of each zero point in the upper stopband. The system calculates the difference between adjacent zero-point frequencies. If the difference is lower than the set isolation safety distance, it determines that there is a risk of resonance overlap. At the same time, the system extracts the insertion loss curve of the passband boundary frequency band and calculates its standard deviation. If the fluctuation amplitude exceeds the allowable threshold, it determines that the impedance matching has deteriorated and confirms the abnormal state.

[0048] When an anomaly is detected, the algorithm sends a topology compensation command to the layout parameter pool, adaptively adjusting the size of the micro-variable capacitor patch loaded at the end of the lateral branch 10, or etching the fine-tuning slot of the central bridging section of the double "H"-shaped band-stop structure 11 to change the local equivalent impedance; it observes the frequency domain spacing change in real time, and when the spacing expands to the safe range and the insertion loss fluctuation falls back to the allowable range, it locks the new parameter configuration, reruns the electromagnetic solution using the corrected geometric model, and outputs an updated hybrid broadband response signal set to complete the closed-loop correction.

[0049] Based on the above, by monitoring the frequency domain spacing and adaptively adjusting the micro-impedance, the impedance mismatch caused by the overlap of multi-mode resonances is dynamically corrected, thereby achieving stable zero-point distribution and concentrated signal generation.

[0050] As an optional embodiment, linearization compensation is performed on the phase group delay and gain flatness within the passband to output the final radiation state parameters of the converged antenna, specifically including the following steps: Within the target wideband operating range, sample the far-field radiation phase values ​​and the main lobe width of the radiation pattern at multiple discrete frequency points; Construct a pattern distortion correction matrix and perform spatial weighted smoothing on the main lobe width of the pattern; Combining the root mean square error of gain flatness and the linearity index of phase group delay, multi-objective optimization iteration is performed. When all the indices fall within the preset tolerance window, the final radiation state parameters of the antenna after convergence are output.

[0051] According to the above steps, during the execution of step S5, the post-processing module sets the frequency scanning sequence, reads the phase distribution data of the electric field of the main radiation plane and the 3dB beamwidth value corresponding to each discrete frequency point, and forms a discrete sampling set.

[0052] Based on the sample set, deviation vectors of beamwidths at different frequencies from the ideal value are established, and a weighted coefficient matrix is ​​constructed. The weighted coefficient matrix is ​​applied to perform low-pass filtering and interpolation smoothing on the main lobe width data to suppress high-frequency pattern broadening. The root mean square error between the smoothed gain curve and the reference level is calculated, and the fitting residual of the group delay curve to the linear trend line is also calculated. Using error minimization and residual convergence as objective functions, the height difference of the stepped dielectric layer and the chamfer dimensions of the coupling slot line 6 are finely adjusted. When both the error and residual decrease to within a preset tolerance window, the iteration terminates, and the structural parameters and electromagnetic response data at this point are recorded as the final radiation state parameters after convergence. By employing discrete sampling and spatial weighted smoothing, the phase nonlinear transitions and gain fluctuations within the passband are synchronously converged, thereby achieving accurate output of the final radiation state parameters.

[0053] As an optional embodiment, an antenna electromagnetic control and layout generation system includes a storage module and a processing module. The processing module is configured to execute the radiation characteristic control method of any of the foregoing and output the etching layout parameter file of the overall feed line 4 and the ground plane 2.

[0054] When constructing this system, the storage module uses non-volatile media to store the 3D field solution algorithm library, parameter mapping table, and linear compensation iteration logic code. After loading the code, the processing module initializes the electromagnetic simulation environment and loads the physical property configuration file. The processing module instantiates execution threads S1 to S5 sequentially, receives antenna geometric input data, and performs fundamental mode decoupling, topology mapping, anomaly detection, and group delay compensation calculations. When the calculation converges and triggers the termination condition, the processing module converts the microstrip linewidth, slot coordinates, stub length, and bridging dimensions in the final radiation state parameters into vector graphic data recognizable by the lithography machine. By employing parametric modeling-driven and vector conversion logic, the optimized electromagnetic topology is accurately mapped into manufacturing instructions, thereby achieving automated data stream output from the electromagnetic control algorithm to the physical etching pattern.

[0055] It should be noted that the present application will be further described in detail below with reference to the accompanying drawings and specific embodiments. The following embodiments are only used to explain the present invention and should not be construed as limiting the scope of protection of the present invention.

[0056] In summary: First, when implementing the radiation characteristic modulation method, the system first obtains the initial electromagnetic state parameter set of the dielectric resonator 1 and the overall feed line 4 based on the antenna physical architecture.

[0057] Combination Figure 1 3D structural diagram and Figure 2 As can be seen from the front view, the antenna entity adopts a four-layer physical architecture stacked from top to bottom: The top layer is a stepped dielectric resonator 1, which is made of a composite ceramic material with a dielectric constant of 10; The second layer is a copper-etched ground plane 2, with a rectangular geometrically contoured coupling groove 6 in its central area. The third layer is a dielectric substrate 3 made of FR4 material, with an example thickness of 0.8 mm; The fourth layer is the integral feed line 4 printed on the lower surface of the substrate.

[0058] The lower surface of the dielectric resonator 1 is tightly bonded to the ground plane 2 via a low-loss adhesive layer. Its bottom surface has inwardly shaped cuboid grooves 5 on both sides. These grooves can be formed by methods including, but not limited to, milling, 3D printing, laser processing, ion beam milling, mechanical cutting, photolithography, and sintering. Spatially, these grooves correspond to the projection area of ​​the coupling groove line 6 on the upper surface of the ground plane 2. The system calls a three-dimensional full-wave solving engine to map the stepped profile dimensions (length and width a=20.8 mm, b=20.8 mm, total height d=16 mm), the cuboid groove 5 cut dimensions (a1=20.8 mm, b1=3 mm, d1=5 mm), and the coupling groove line 6 dimensions (Ls1=0.6 mm, Ws1=15.5 mm) to a dielectric constant gradient distribution model. After the radio frequency signal is injected through the 50 Ω SMA connector from the input end of the integral feeder 4, the electromagnetic excitation signal is transmitted along the open-loop feeder section 7 to the edge of the coupling slot line 6, and a vertical electric field is injected into the bottom of the dielectric resonator 1 through the near-field edge coupling mechanism.

[0059] Combination Figure 3 Side view and Figure 4 As seen in the top view, the two ends of the open-loop feeder section 7 extend inclined towards the sides of the ground plane 2, causing the open-loop resonant frequency to overlap with the transmission cutoff frequency of the coupled slot line 6 in the frequency domain. By performing a decoupling operation on the fundamental mode field distribution, the abrupt boundary change of the cuboid slot 5 forces the TE111 fundamental mode electric field to undergo directional distortion, splitting into multiple secondary resonant states, which are then vector-superimposed with the electromagnetic energy of the slot line transmission mode, eliminating the phase cancellation blind zone and outputting a hybrid broadband response signal set. By adopting a spatially coordinated layout of stepped profile slotting and inclined open-loop routing, the frequency domain response of the fundamental mode splitting state and the slot line cutoff characteristics are continuously covered, thereby achieving the initial construction of the broadband energy distribution.

[0060] Furthermore, when the passband envelope is analyzed on the mixed broadband response signal set and it is determined that the upper stopband attenuation characteristics need to be enhanced, the system maps a folded open-loop resonator 8 at the symmetrical node of the open-loop feed section 7.

[0061] Combination Figure 2 and Figure 4 As can be seen, the folded open-loop resonator 8 is symmetrically connected to the left and right sides of the open-loop feed section 7 and extends in a direction away from the coupling slot line 6.

[0062] In a preferred embodiment, the folded structure employs a triple-fold topology, with example segment dimensions of Lf21=0.1mm / Wf21=3 mm, Lf22=10 mm / Wf22=0.1 mm, Lf23=0.1 mm / Wf23=1 mm, and Lf24=9 mm / Wf24=0.1 mm, where Lf represents the x-axis direction and Wf represents the y-axis direction. This representation will be used for subsequent meanings. Adjusting the number of folded layers and coupling gaps causes the branch reflection phase to be superimposed with the main feed path at a 180-degree out-of-phase ratio, generating a first radiation null point in the lower stopband. Simultaneously, an additional parasitic resonant mode is introduced to widen the impedance matching range at the low-frequency end. When the harmonic frequency band energy density exceeds a threshold, multi-topology notch filtering is triggered, sequentially configuring a transverse stub 10, a spur line 9, and a double "H"-shaped bandstop structure 11 on the main transmission path of the overall feeder 4.

[0063] Combination Figure 4 As can be seen from the layout, the spur line 9 is directly etched above the main feed strip, for example, Lf31=0.5 mm / Wf31=0.125 mm and Lf32=7.5 mm / Wf32=0.125 mm, introducing an increase in series inductance by changing the current flow path; the lateral branch 10 is arranged at a position slightly below the upstream of the spur line 9, for example, Lf4=0.3 mm / Wf4=11 mm; the double "H" type band-stop structure 11 is placed at a position slightly above the downstream of the spur line 9, with the lateral part Lf51=0.2 mm / Wf51=6.1 mm and the vertical bridging part Lf52=0.5 mm / Wf52=0.2 mm, and the higher-order harmonic attenuation depth is independently tuned by the asymmetrical configuration of the central bridging width and the lateral slot depth.

[0064] The cascaded arrangement and electrical interconnection of the three components in space cause the notch frequencies generated by each component to superimpose, forming an upper stopband radiation null array. By employing a sequential arrangement of a third-order bandstop topology and independently adjusting the equivalent LC parameters, multi-band harmonic energy is distributedly intercepted along the transmission path, thereby achieving precise synthesis of broadband notch characteristics.

[0065] Combining the resonance overlap anomaly handling process with linearization compensation calculations for phase group delay and gain flatness within the passband: Based on this, the system monitors the frequency domain distribution of the lower and upper stopband radiating null arrays in real time. When the spacing between adjacent nulls is less than the preset isolation threshold or the insertion loss fluctuation at the passband edge exceeds the standard, it is determined to be an abnormal resonance overlap. The control unit triggers adaptive fine-tuning logic, which fine-tunes the size of the load capacitor at the end of the lateral branch 10 or the width of the central bridging section of the double "H"-shaped bandstop structure 11 through etching process, changing the local equivalent impedance distribution until the frequency domain spacing is restored to the safe range, and then regenerating the response signal set.

[0066] Combination Figure 4As can be seen from the symmetrical layout, the overall antenna structure maintains strict geometric symmetry in the X and Y axes. The mirror arrangement of the feed branch, resonant parasitic unit, and band-stop structure causes the cross-polarized field components to cancel each other out, ensuring the axial symmetry of the main polarization radiation. In the optimization iteration stage, the system samples the far-field radiation phase and 3dB beamwidth of discrete frequency points in the target wideband range, constructs a pattern distortion correction matrix, and performs spatial weighted smoothing on the main lobe width. The root mean square error of gain flatness and the phase group delay linearity residual are calculated. When the indicators converge to the preset tolerance window, the final radiation state parameters are locked. By adopting the frequency domain spacing dynamic compensation and spatial weighted smoothing algorithm, the impedance mismatch caused by multimode resonance overlap is corrected in real time, thereby achieving synchronous convergence of phase group delay and gain fluctuation in the passband.

[0067] Combining antenna electromagnetic control and layout generation system: Furthermore, when constructing the antenna electromagnetic control and layout generation system, the storage module loads a code library containing a full-wave solution kernel, parameter mapping tables, and linear compensation iteration logic. The processing module instantiates the aforementioned control process, converting the converged antenna's final radiation state parameters, including microstrip linewidth, slot coordinates, stub length, and bridging dimensions, into vector graphics data in Gerber, DXF, and other formats. The system outputs an etched layout parameter file containing the etching trajectory of the ground plane 2 coupling slot and the bottom metallized trace of the overall feed line 4, driving photolithography or PCB processing equipment to achieve a data closed loop from electromagnetic topology optimization to physical manufacturing.

[0068] Combined with broadband filter dielectric resonator antenna structure: Furthermore, during the assembly of the antenna, a multilayer dielectric lamination process is used to place the stepped dielectric resonator 1 on the top layer. The ground plane 2 is electrically connected to the copper-clad layer on the upper surface of the dielectric substrate 3 through conductive adhesive. The overall feed line 4 is formed on the lower surface of the substrate using a copper foil etching process. The open-loop feed line section 7 adopts a double-folded topology with dimensions of Lf11=26 mm / Wf11=0.5 mm, Lf12=1 mm / Wf12=4 mm, and Lf13=7.5 mm / Wf13=0.5 mm. The two ends of the open-loop feed line section 7 extend obliquely towards the sides of the ground plane 2 and generate near-field coupling with the coupling slot line 6 through vertical slots.

[0069] Combination Figure 1 3D layering relationship and Figure 4 As can be seen from the top view projection, the central axes of each structural layer coincide. Energy is input from the bottom SMA connector, and after being fed through an open loop and coupled through the slot line, it penetrates into the top dielectric resonator 1, exciting the main radiation mode.

[0070] Combination Figure 5As can be seen from the S11 and gain simulation curves, through the distributed radiated null and multimode fusion architecture, the antenna center operating frequency is stabilized at 3.48 GHz, the impedance bandwidth covers 2.88-4.14 GHz with S11 < -10 dB, the fractional bandwidth reaches 37.9%, the average gain in the passband is greater than 6.4 dBi, and the peak gain reaches 6.9 dBi. Radiated nulls are formed at the upper and lower stopband edges of the passband, respectively, and four frequency-controllable deep valleys are generated at 6.34 GHz, 7.78 GHz, 8.38 GHz, and 8.58 GHz in the stopband, achieving wide harmonic suppression.

[0071] Combination Figure 6 As can be seen from the XOZ and YOZ radiation patterns, at typical resonant points of 2.88 GHz, 3.16 GHz, 3.70 GHz, and 4.08 GHz within the passband, the radiation patterns all exhibit stable axial main lobe distributions. The cross-polarization level is approximately 30 dB lower than the main polarization, and the radiation pattern shows no significant distortion. By adjusting the physical lengths of the open-loop feed section 7 and the folded open-loop resonator 8, the zeros on both sides of the passband can be shifted. Adjusting the dimensions of the lateral stub 10, the spur line 9, and the double "H"-shaped bandstop structure 11 allows for independent control of the stopband zero frequency position. Through the deep integration of the four-layer compact stacked architecture and the distributed feed notch topology, broadband radiation, filter selectivity, and harmonic suppression functions are integrated within a single physical plane, thereby achieving high gain, low loss, and stable radiation pattern output without introducing an external filter network.

[0072] As an optional embodiment, refer to the appendix Figure 8 A broadband filter dielectric resonator antenna with wide harmonic suppression characteristics includes a dielectric resonator 1, a ground plane 2, a dielectric substrate 3 and an integral feed line 4 stacked from top to bottom. The surface of the ground plane 2 facing the dielectric resonator 1 is etched with a coupling groove line 6, and the projection area of ​​the coupling groove line 6 covers the bottom of the dielectric resonator 1. The overall feed line 4 is printed on the side of the dielectric substrate 3 away from the ground plane 2, and includes an open-loop feed line portion 7. The two ends of the open-loop feed line portion 7 extend toward both sides of the ground plane 2 and are electromagnetically coupled to the dielectric resonator 1 through the coupling slot line 6. The overall feeder 4 also integrates a folded open-loop resonator 8, a spur wire 9, a lateral stub 10, and a double "H" type band-stop structure 11. The folded open-loop resonator 8 is connected to both sides of the open-loop feeder section 7, and the lateral stub 10, the spur wire 9, and the double "H" type band-stop structure 11 are arranged sequentially on the main feeder branch along the signal transmission direction of the overall feeder 4.

[0073] As described above, when assembling the antenna, a multilayer dielectric lamination process is used to place the dielectric resonator 1 on the top layer of the physical structure. Its bottom surface is electrically and mechanically connected to the ground plane 2 through a low-loss adhesive. The ground plane 2 covers the upper surface of the dielectric substrate 3, forming a complete metal shielding reference plane.

[0074] Furthermore, the integral feed line 4 is formed on the lower surface of the dielectric substrate 3 using photolithography etching, forming the bottom signal input network. Metal material is removed from the ground plane 2 corresponding to the bottom region of the dielectric to form a rectangular opening, constituting the coupling groove line 6. The vertical projection of this groove line falls entirely within the bottom contour boundary of the dielectric resonator 1. The radio frequency signal is injected through the input port of the integral feed line 4 and transmitted along the microstrip conductor to the bifurcation point at the end of the open-loop feed line section 7. The two ends of the open-loop feed line section 7 extend outwards in an open shape, generating near-field electromagnetic induction through the vertical gap formed by the thickness of the dielectric substrate 3 and the coupling groove line 6. Energy penetrates the ground plane and is injected into the dielectric resonator 1 to excite a radiation mode.

[0075] Furthermore, along the main transmission path, a folded open-loop resonator 8 is symmetrically printed on both sides of the open-loop arm, while lateral branches 10, spur lines 9, and double "H"-shaped band-stop structures 11 are sequentially etched onto the feed line conductor. These structures are connected by transmission line segments with matched characteristic impedances. Through a stacked architecture and projection overlap design, a vertical energy tunnel is formed between the ground plane coupling slot and the underlying microstrip open loop, thereby achieving low-loss signal penetration and multimode resonant excitation from the feed layer to the radiation layer.

[0076] As an optional embodiment, refer to the appendix Figures 1 to 8 The dielectric resonator 1 has a stepped structure, with its bottom surface recessed inward on both sides to form a cuboid groove 5; The two arm sections of the open-loop feed line 7 extend toward the double "H" type band-stop structure 11 and are symmetrically distributed with the folded open-loop resonator 8 on the plane of the overall feed line 4. The upstream end of the spur line 9 (i.e. the end facing the transverse branch 10) maintains a preset isolation gap with the transverse branch 10. The double "H"-shaped band-stop structure 11 is electrically connected to the main transmission conductor of the overall feeder 4 through the central bridging section. The etching trajectory of the spur line 9 and the edge of the main transmission conductor have a preset coupling distance.

[0077] Based on the above, in the detailed implementation of the structure, the dielectric resonator 1 is composed of multiple layers of dielectric blocks with decreasing cross-sectional dimensions coaxially stacked to form a stepped profile. In the central region of the bottom layer, a portion of the material is removed through milling to form an inwardly recessed cuboid groove 5, disrupting the symmetry boundary of the fundamental mode electric field and forcing the TE... 111The mode splits directionally at the bottom. The two arm segments of the open-loop feed section 7 are laid out with a slightly inclined trace trajectory towards the ground plane 2. The parasitic folded structure of the folded open-loop resonator 8 is symmetrically arranged on the outer side of the two arms to maintain the balance of the equivalent impedance on both sides and avoid field polarization rotation caused by parasitic coupling on one side. The spur line 9 is etched on the overall feed line 4. The lateral branch 10 is a lateral branch. The lateral branch 10 and the spur line 9 are on the same plane, and the spur line 9 faces the end of the lateral branch 10 without contacting the lateral branch 10. There is a certain gap, forming a micron-level air isolation gap. The middle crossbeam of the double "H"-shaped band-stop structure 11 is directly connected to the main microstrip line to achieve low-resistance electrical connection. The metal edge of the spur line 9 is arranged parallel to the edge of the main microstrip line and maintains a constant gap. The series inductance increment is introduced through the mutual inductance of the gap magnetic field. By spatially arranging the bottom slots and capacitor gaps, the frequency domain response of the fundamental mode split state and the resonant path of the high-frequency parasitic parameters are physically decoupled, thereby achieving independent control of broadband impedance matching and multi-band harmonic notch filtering.

[0078] The above description is merely a preferred embodiment of the present invention. It should be noted that those skilled in the art can make various improvements and modifications without departing from the principles of the present invention, and these improvements and modifications should also be considered within the scope of protection of the present invention. Structures, devices, and operating methods not specifically described or explained in this invention are implemented according to conventional methods in the art unless otherwise specified or limited.

Claims

1. A method for controlling the radiation characteristics of a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics, characterized in that, Includes the following steps: Obtain the initial electromagnetic state parameter set of the dielectric resonator (1) and the overall feed line (4), perform the fundamental mode field distribution decoupling operation, and output the impedance matching characteristic vector and the target frequency band ground state energy distribution; Construct an open-loop feed section (7) coupling topology, transmit the electromagnetic excitation signal through the open-loop feed section (7) to the coupling slot line (6) region etched on the ground plane (2), inject the coupling field into the bottom of the dielectric resonator (1), and generate a mixed broadband response signal set containing the fundamental mode split state and the slot line transmission mode; The passband envelope analysis and edge attenuation slope determination are performed on the hybrid broadband response signal set. When it is found that the attenuation characteristics of the upper stopband frequency domain do not meet the first preset condition, a folded open-loop resonator (8) is mapped at the symmetrical node of the open-loop feed section (7) to generate the first lower stopband radiation zero point and the extended resonant frequency domain interval. When the energy density distribution of the harmonic frequency band in the mixed broadband response signal set exceeds the second preset threshold, the multi-topology notch control process is triggered. A transverse stub (10), a spur line (9) and a double "H" type bandstop structure (11) are sequentially configured at the designated coupling position of the main transmission path of the overall feeder (4) to generate an upper stopband radiation zero array. Based on the frequency domain superposition relationship between the first lower stopband radiation null and the upper stopband radiation null array, a linearization compensation operation is performed on the phase group delay and gain flatness in the passband, and the final radiation state parameters of the antenna after convergence are output.

2. The method for controlling the radiation characteristics of a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics according to claim 1, characterized in that, The execution of the fundamental mode field distribution decoupling operation, and the output impedance matching eigenvector and the target frequency band ground state energy distribution, specifically includes the following steps: Extract the stepped geometric contour parameters of the dielectric resonator (1) and the spatial coordinate set of the cuboid groove (5) dug at the bottom, and construct a three-dimensional dielectric constant gradient distribution model. The three-dimensional dielectric constant gradient distribution model is mapped to the excitation port of the full-wave electromagnetic field iterative algorithm to calculate TE. 111 The electric field distortion rate of the fundamental mode under the perturbation of the cuboid slot (5); When the electric field distortion rate reaches the preset splitting threshold, the splitting resonant frequency combination is locked, and the impedance matching feature vector and the target frequency band ground state energy distribution are output.

3. The method for controlling the radiation characteristics of a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics according to claim 1, characterized in that, The generation of the hybrid broadband response signal set, which includes the fundamental mode split state and the slotted line transmission mode, specifically includes the following steps: Extract the physical tilt trend of the two branches of the open-loop feeder section (7) extending toward the two sides of the ground plane (2) and the characteristic impedance of the transmission line; The overlap range between the open-loop resonant frequency and the transmission cutoff frequency of the coupling slot line (6) is calculated based on the physical tilt trend and characteristic impedance. The electromagnetic energy distribution within the overlapping interval is vector superimposed to eliminate the phase cancellation blind zone and output the hybrid broadband response signal set.

4. The method for controlling the radiation characteristics of a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics according to claim 1, characterized in that, A folded open-loop resonator (8) is mapped at the symmetrical node of the open-loop feed section (7) to generate the first lower stopband radiation zero and the extended resonant frequency domain interval, specifically including the following steps: Obtain the current standing wave node positions on the left and right sides of the open-loop feeder section (7) and determine the electrical connection anchor point of the folded open-loop resonator (8); Adjust the number of folds and coupling gap parameters of the folded open-loop resonator (8) to establish a parameter mapping table of branch equivalent capacitance and equivalent inductance; Based on the parameter mapping table, the branch resonant frequency is adjusted to the target frequency point of the lower stopband. When the branch reflection phase and the main feed path form an antiphase superposition, the first lower stopband radiation zero point is solidified.

5. The method for controlling the radiation characteristics of a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics according to claim 1, characterized in that, At a designated coupling position on the main transmission path of the overall feed line (4), a transverse stub (10), a spur line (9), and a double "H" type bandstop structure (11) are sequentially configured to generate an upper stopband radiation null array, specifically including: The spur line (9) is etched in the energy high concentration area of ​​the main transmission path of the overall feeder (4), and the equivalent series inductance parameter of the spur line (9) is extracted. The lateral stub (10) is positioned in the upstream coupling region of the spur line (9), and the stub arm length is adjusted to match the quarter-wavelength resonance condition of the target harmonic frequency. The double "H" type bandstop structure (11) is placed in the downstream isolation area of ​​the spur line (9). By adjusting the bridging width and transverse slot depth of the double "H" type bandstop structure (11), the attenuation depth of different order harmonic frequencies is independently tuned, and the upper stopband radiation zero-point array is output.

6. The method for controlling the radiation characteristics of a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics according to claim 1, characterized in that, The method also includes a resonance overlap anomaly handling procedure: Real-time monitoring of the frequency domain distribution of the first lower stopband radiation null and the upper stopband radiation null array; When the calculated frequency domain spacing between adjacent zeros is less than the third preset threshold, or the passband edge insertion loss fluctuation exceeds the fourth preset threshold, it is determined to be an abnormal resonance overlap. Trigger the topology compensation mechanism to adaptively fine-tune the end load impedance of the lateral branch (10) or the width of the center connecting bridge of the double "H" type band-stop structure (11) until the isolation of the adjacent zero points is restored to the allowable range, and regenerate the hybrid broadband response signal set.

7. The method for controlling the radiation characteristics of a broadband filter dielectric resonator antenna with wide harmonic suppression characteristics according to claim 1, characterized in that, The linearization compensation operation on the phase group delay and gain flatness within the passband, and the output of the converged final radiation state parameters of the antenna, specifically includes the following steps: Within the target wideband operating range, sample the far-field radiation phase values ​​and the main lobe width of the radiation pattern at multiple discrete frequency points; Construct a radiation pattern distortion correction matrix and perform spatial weighted smoothing on the main lobe width of the radiation pattern; Combining the root mean square error of gain flatness with the linearity index of phase group delay, multi-objective optimization iteration is performed. When all indices fall within the preset tolerance window, the converged antenna final radiation state parameters are output.

8. An antenna electromagnetic control and layout generation system, comprising a storage module and a processing module, wherein the processing module is configured to execute the radiation characteristic control method as described in any one of claims 1 to 7, and output the etched layout parameter file of the overall feed line (4) and the ground plane (2).

9. A broadband filter dielectric resonator antenna with wide harmonic suppression characteristics, characterized in that, The structural parameters of the antenna are optimized and determined by the radiation characteristic control method as described in any one of claims 1 to 7, including a dielectric resonator (1), a ground plane (2), a dielectric substrate (3), and an overall feed line (4) stacked sequentially from top to bottom. The ground plane (2) facing the dielectric resonator (1) has a corresponding coupling groove line (6) etched on its surface, and the projection area of ​​the coupling groove line (6) covers the bottom of the dielectric resonator (1). The integral feed line (4) is printed on the side of the dielectric substrate (3) away from the ground plane (2) and includes an open-loop feed line portion (7). The two ends of the open-loop feed line portion (7) extend toward both sides of the ground plane (2) and are electromagnetically coupled to the dielectric resonator (1) through the coupling groove line (6). The overall feed line (4) also integrates a folded open-loop resonator (8), a spur wire (9), a transverse stub (10), and a double "H" type band-stop structure (11); the folded open-loop resonator (8) is connected to both sides of the open-loop feed line section (7), and the transverse stub (10), the spur wire (9), and the double "H" type band-stop structure (11) are arranged sequentially on the main feed branch along the signal transmission direction of the overall feed line (4).

10. A broadband filter dielectric resonator antenna with wide harmonic suppression characteristics according to claim 9, characterized in that, The dielectric resonator (1) has a stepped structure, with the bottom surface recessed inward on both sides to form the cuboid groove (5). The two arm segments of the open-loop feed line section (7) extend toward the double "H" type band-stop structure (11) and are symmetrically distributed with the folded open-loop resonator (8) on the plane of the overall feed line (4); The spur line (9) has a preset isolation gap between one end facing the transverse branch (10) and the transverse branch (10). The double "H" type band stop structure (11) is electrically connected to the main transmission guide of the overall feed line (4) through the central bridging section. The etching trajectory of the spur line (9) is at a preset coupling distance with the edge of the main transmission guide.

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