Method for estimating the characteristics of a pulsed ultra-wideband signal emitted by multiple antennas

Description

[0001] The invention relates to the field of pulsed ultra-wideband communication systems, i.e. for which the signals are modulated in the form of pulses of very short duration.

[0002] More specifically, the invention relates to systems in which several antennas are used for transmission and relates to a method for estimating phase shifts between signals received from different antennas. The phase shifts thus estimated make it possible to calculate phase differences and then estimate the angular direction of transmission of the signal. One application of the invention relates to the location of a transmitter or the measurement of the distance between a transmitter and a receiver.

[0003] Estimating phase shifts also helps improve signal demodulation.

[0004] The method also covers the estimation of a carrier frequency offset between a receiver and a transmitter as well as the estimation of an impulse response of the propagation channel.

[0005] In the field of radio communication systems, the signals received in baseband by the receivers are disturbed by multiple defects linked to desynchronizations between the transmitter and the receiver or to the conditions of signal propagation. Indeed, in practice, the reception conditions are not ideal.

[0006] More precisely, a carrier frequency offset appears between the received signal and the transmitted signal when the respective local oscillators of the transmitter and receiver are not synchronized.

[0007] In the case where the transmitter has several transmitting antennas, a phase shift specific to each antenna also exists between the received signal and the transmitted signal, due to different propagation times between each transmitting antenna and the receiving antenna.

[0008] These phase shifts must be compensated to ensure good demodulation performance but they can also be useful for estimating an angular direction of the transmitted signal.

[0009] Furthermore, ultra-wideband communication systems are subject to interference due to multiple reflections of the signal from environmental obstacles. These disturbances affect the received signals in the form of intersymbol interference due to multiple signal paths received simultaneously.

[0010] A technical problem to be solved in this context consists of estimating the phase shifts between the signals received from different antennas of a multi-antenna transmitter but also of estimating the carrier frequency shift and the impulse response of the propagation channel.

[0011] US patent application 20200252101 describes a method for determining an angular direction for ultra-wideband communication systems based on a direct estimation of the departure angles of the signals coming from each antenna. This document does not describe a method for estimating, at reception, the phase differences between signals coming from different transmitting antennas.

[0012] Patent documents US2017 / 0227623, US9048979 and US9048980 describe phase and frequency synchronization methods for ultra-wideband systems. These documents propose phase and frequency estimation methods based on a correlation between the square of the reception channel I and the reception channel Q. These methods have the disadvantage of being inefficient at low signal-to-noise ratio due to the squaring of the reception signals.

[0013] We also know the document US 2018 / 254870 A1, relating to a method for estimating phase difference at the receiver, based on STS transmission by a transmitter with 2 antennas.

[0014] The invention proposes a new method for estimating phase and frequency shifts for ultra-wideband communication systems in which the transmitters comprise several antennas. The invention is defined by the subject matter of claim 1, relating to the estimation method, and by the subject matter of claim 8, relating to the corresponding receiver. The preferred embodiments are defined by the subject matter of the dependent claims.

[0015] An advantage of the invention is that it exhibits good estimation performance even at low signal-to-noise ratio.

[0016] The subject of the invention is a method for estimating at least one characteristic of a signal received by a receiver, the signal having been emitted successively by several antennas in successive time segments, each segment being dedicated to a separate antenna, the signal being modulated in the form of pulses according to an ultra-wideband modulation, the method comprising the steps of: Receiving and digitizing said signal, Calculating the product of each symbol of the received signal with the complex conjugate of the corresponding symbol transmitted, For each segment and for each symbol of the signal received for this segment, estimating a phase error by means of a phase-locked loop applied to said product, For each segment, determining a reference phase by means of a linear regression applied to the phase errors estimated on all the segments, Determining, for at least one pair of antennas, a phase difference between the signals transmitted by the antennas of the pair, from the difference between the reference phases calculated for the segments associated with said antennas.

[0017] According to a particular aspect of the invention, the determination of the phase difference, the determination of a carrier frequency offset between the signal emitted by all of the antennas and the signal received by means of said linear regression applied to the phase errors estimated on all of the segments.

[0018] According to a particular aspect of the invention, the successive time segments are separated by guard intervals and the method further comprises the suppression, in the received and digitized signal, of these guard intervals.

[0019] According to a particular embodiment, the method further comprises a step of determining an angular direction of emission of the signal from said phase differences.

[0020] According to a particular embodiment, the method further comprises the steps of: Correct each symbol of the received signal by the phase error calculated by means of the phase-locked loop, Estimate an impulse response of the propagation channel from a cross-correlation calculation between the sequence of corrected received symbols and a sequence of symbols transmitted in the segments.

[0021] According to a particular embodiment, the method first comprises an initial synchronization phase specific to each segment comprising the steps of: Determine, on a preamble sequence prior to the first segment, a first estimate of a carrier frequency offset between the received signal and the digitized signal, Correct the digitized received signal by this first estimate, Estimate, on each start of segment, an impulse response of the propagation channel from a calculation of intercorrelation between a sequence of corrected received symbols and a sequence of corresponding transmitted symbols, Determine, from the estimated impulse response for each start of segment, a symbol corresponding to the start of the segment.

[0022] According to a particular aspect of the invention, the signal complies with the IEEE802.15.4z standard.

[0023] According to a particular aspect of the invention, the sequences of symbols transmitted in the time segments are secured by means of an encryption algorithm.

[0024] The invention also relates to a receiver of a signal modulated in the form of pulses according to an ultra-wideband modulation comprising an antenna and a computer configured to execute the steps of the method for estimating at least one characteristic of the signal received according to the invention.

[0025] The invention also relates to a communication system comprising a transmitter of a signal modulated in the form of pulses according to an ultra-wideband modulation comprising several antennas, the signal being emitted successively by several antennas in successive time segments, each segment being dedicated to a separate antenna, the system further comprising a receiver according to the invention.

[0026] Other features and advantages of the present invention will become more apparent upon reading the following description in relation to the following appended drawings. [ Fig. 1 ] represents a theoretical diagram of the evolution of the phase of an ultra-wideband signal emitted by two antennas and received by one antenna, [ Fig. 2 ] represents several examples of frame format configurations from the IEEE802.15.4z standard, [ Fig. 3 ] represents an example of an ultra-wideband pulsed signal, [ Fig. 4 ] represents a diagram of a phase-locked loop, [ Fig. 4b ] represents a diagram of variation of a phase error, [ Fig. 5 ] represents a block diagram of a method for estimating phase and frequency shifts and estimating impulse response according to different embodiments of the invention, [ Fig. 6 ] represents a diagram illustrating a step of determining a transmission angle, [ Fig. 7 ] represents a performance diagram of the results of the method according to the invention as a function of different signal-to-noise ratios and signal emission configuration, [ Fig. 8 ] represents a synopsis of a variant of the method according to the invention, [ Fig. 9 ] represents a comparative performance diagram of the method according to the invention, [ Fig. 10 ] represents a simplified diagram of a communication system according to the invention.

[0027] The invention is now described in the context of an application to the IEEE802.15.4z standard which is an amendment to the IEEE802.15.4 standard. This application is given as an illustrative and non-limiting example. The invention applies to any standard or to any compatible waveform of an ultra-wideband pulsed modulation (IR-UWB) and which respects the conditions illustrated in figure 1 .

[0028] The invention applies to IR-UWB communications systems comprising a multi-antenna transmitter (comprising at least two antennas) and a single-antenna receiver.

[0029] According to the invention, several characteristics of the signal received by the receiver are estimated. For this, it is considered that the signal is emitted successively by each antenna of the transmitter over time intervals called segments, each segment being separated by a guard interval.

[0030] On the figure 1 , an example of a transmission format for a transmitter having two antennas is shown. The two antennas transmit successively and respectively in segments S1 and S2, separated by guard intervals G1, G2, G3.

[0031] On the figure 1 The theoretical shape of the temporal evolution of the phase of the signal received by the receiver is also represented. On each of the segments, the phase of the signal evolves linearly with a slope common to all the segments which corresponds to a carrier frequency deviation. Here we consider that all the transmitting antennas are controlled by the same oscillator. The carrier frequency deviation or "carrier frequency offset" CFO in English, on the received signal is due to the fact that the respective oscillators of the transmitter and receiver are not synchronized.

[0032] On each segment, the phase has an ordinate at the origin β 1 , β 2 which represents a common reference at a reference instant. The difference β 2 -β 1 contains the information of phase deviations between the signals coming from the two antennas.

[0033] The invention aims in particular to estimate the aforementioned phase deviations and frequency deviation. Estimating the phase deviations makes it possible to calculate phase differences in order to deduce, for example, a starting angle of the signal, i.e. an angular direction of the signal, which makes it possible to locate the transmitter or the receiver.

[0034] The invention also makes it possible to maintain phase coherence to carry out coherent demodulation of any data transmitted after segments S1, S2 which are dedicated to frequency and phase estimations.

[0035] There figure 2 schematizes several frame configurations according to the IEEE 802.15.4z standard. This standard offers several frame formats including a field dedicated to measuring distance and / or angle of arrival, this field being called STS (Scrambled Timestamp Sequence).

[0036] The C0 configuration presents a frame consisting of the following elements: a preamble P consisting of a repetition of a given sequence and used for initial time and frequency synchronization and frame detection, an SFD (Start of Frame Delimiter) field which is a repetition of the same sequence as the preamble but with a weighting of 1.0 or -1. This sequence is used to confirm frame detection and detect the end of the preamble, a PHR (PHY header) field which contains header information, a PSDU (PHY service data unit) field which contains the data from the physical layer point of view.

[0037] The C1 configuration also includes an STS field between the SFD field and the PHR header.

[0038] The C2 configuration has an STS field after the PSDU data.

[0039] The C3 configuration does not include PSDU data (nor PHR header).

[0040] Configurations C1, C2, C3 can be used to implement the invention.

[0041] The STS field is formed by segments S1,S2 separated by non-transmitting guard intervals G1,G2,G3 as illustrated in figure 1 .

[0042] There figure 1 represents two segments, but the invention is applicable regardless of the number of segments being at least equal to 2.

[0043] The signal emitted in a time interval corresponding to a segment S1,S2 is composed of separate pulses of a fixed duration T PRP . Each pulse being emitted with a known complex amplitude cn . Without departing from the scope of the invention, the duration between two successive pulses can be variable to the extent that this duration is known to the receiver.

[0044] On the figure 3 , an example of a symbol sequence transmitted for a segment is shown. In this example, the symbols are real but they can be complex, in which case there are two channels I and Q, each channel being modulated in the way shown in figure 3 .

[0045] The symbols transmitted for a segment S1,S2 are thus modulated with IR-UWB modulation. The typical duration of a pulse is of the order of a nanosecond, for example equal to 2ns.

[0046] We now explain the form of the signal received by a receiver, the signal having been emitted according to the principles previously explained by a transmitter having N segment antennas, corresponding to N segment segments.

[0047] We define the signal received in baseband z(t) for a segment: z t = exp 2 iπ Δ ft + iϕ 0 ∑ k = 0 N − 1 c k ∑ v = 0 N path − 1 h v p t − kT PRP − τ v + b t Or : N is the number of complex symbols per segment, ϕ 0 is a phase difference between the phase of the receiver's local oscillator and the phase of the signal emitted by each antenna. This difference does not include the phase difference due to the signal propagation time, Δf is a frequency difference between the receiver and the transmitter (common to all antennas), p(t) is the shape of the basic pulse, b(t) is a noise whose real and imaginary parts are assumed to be centered white Gaussian and of the same variance, N path is the number of paths in the propagation channel, hv and τ v are respectively the complex amplitude of the signal path v and the delay of the path v,

[0048] the channel impulse response, h ( t ) , is therefore defined by: h t = ∑ v = 0 N path − 1 h v δ t − τ v

[0049] We ask: p h t = ∑ v = 0 N path − 1 h v p t − τ v

[0050] And we get: z t = exp 2 iπ Δ ft + iϕ 0 ∑ k = 0 N − 1 c k p h t − kT PRP + b t

[0051] We extend the notation to the entire STS field by considering that the symbols c k are zero in the guard intervals G1,G2,G3 between segments S1,S2.

[0052] If we consider that a guard interval G1,G2,G3 has a duration equal to N g T PRP , the symbols c N ...c N+N g -1 "transmitted" during a guard interval following the first segment are zero.

[0053] Similarly, if the STS field has three segments, the symbols c 2N+Ng ... c 2N+2N g -1 of the guard interval between the second and third segment are zero and so on.

[0054] The propagation channel varies between segments since each segment corresponds to a different transmitting antenna. We note h s the channel on the segment with index s. We can then write (with N seg the number of segments, T s = N × T PRP et T g = N g × T PRP : z t = exp 2 iπ Δ ft + iϕ 0 ∑ s = 0 N seg − 1 ∑ k = 0 N − 1 c k + s N + N g p h s t − kT PRP − s T s + T g + b t

[0055] On reception, this signal is sampled with a sampling period T ech (respecting a sampling frequency greater than twice the bandwidth) to obtain the discrete signal z n : z n = z nT ech

[0056] A time synchronization step performed on the preamble P makes it possible to determine the index n 0 corresponding to a particular path (for example the strongest path or the first) for the first pulse of the STS field. If necessary, ns is determined for each segment s.

[0057] We then define the signal u n (with M = N + N g is the number of symbols in a segment followed by a guard interval): u n = exp 2 iπΔf n o T ech + nT PRP + iΦ 0 Σ k = 0 N − 1 c k p h o n 0 T ech + n − k T PRP + b n pourlesegment 0 u n = exp 2 iπΔf n 1 T ech + nT PRP + iΦ 0 Σ k = 0 N − 1 c k + M p h 1 n 1 T ech + n − k − M T PRP + b n pourlesegment 1 u n = exp 2 iπΔf n N seg − 1 T ech + nT PRP + iϕ 0 ∑ k = 0 N − 1 c k + N seg − 1 M p h N s − 1 n N seg − 1 T ech + n − k − N seg − 1 M T PRP + b n pour the last segment 0 between segments

[0058] If the temporal support of the function p hs is less than the duration between two pulses, then the sum on each of the lines disappears to leave only the nth term (no interference between pulses).

[0059] Each symbol of the received signal is then multiplied by the complex conjugate of the symbols c k emitted during an STS field: w n = u n × c n *

[0060] w n is zero in the guard intervals (i.e. for n ∈ N ; M − 1 ∪ N + M ; 2 M − 1 ∪ … ∪ N + N seg − 2 M ; N s − 1 M − 1 )

[0061] If there is no interference between pulses, only the modulus of the symbols c n then appears. In the general case, the other terms correspond to interference between pulses which can be assimilated to noise.

[0062] The steps of the method according to the invention are illustrated in the diagram of the figure 5 .

[0063] In step 501, the signal zn which has been previously digitized is received and the signal is corrected by an estimate of a sampling frequency offset (SFO). This SFO is obtained from a prior frequency synchronization step which is carried out to estimate the carrier frequency offset. Δf ^ between the transmitter and the receiver. This shift Δf ^ is also called "Carrier Frequency Offset" or CFO in English. The sampling frequency deviation SFO is the equivalent, in the time domain, of the carrier frequency deviation CFO which has, relatively, the same value because it is generated by the same frequency oscillators in transmission. This preliminary frequency synchronization step is carried out on the preamble P of the frame.

[0064] In step 502, one of the paths n 0 of the signal is selected from a prior time synchronization carried out on the preamble P of the frame to determine the start of the STS field. For example, the selected path is the path with the strongest amplitude. In other words, step 502 consists of selecting the samples of the signal corresponding to the selected path from the result of the time synchronization.

[0065] According to a first embodiment of the invention, this step 502 is common to all segments. This first mode is applicable when the transmitter antennas are close enough that the propagation channels between each transmitting antenna and the receiver can be assumed to be identical.

[0066] In step 503, each symbol of the STS field (in other words each symbol of each segment) is multiplied by the complex conjugate of the corresponding emitted symbol: w n = u n × c n ∗ .

[0067] In step 504, a digital phase-locked loop is applied to the symbols wn in order to extract the phase of the signal.

[0068] An example of a phase-locked loop is shown in figure 4 and consists of several stages.

[0069] At step 401, the symbol wn is corrected by the last phase estimate yn provided at the output of the loop at the previous iteration.

[0070] In step 402, the phase of the corrected symbol or a value which varies in the same direction of variation as this phase over at least one predetermined phase interval is calculated, for example by means of an arctangent calculation or the following simplified calculation: e n = I w n ˜ − sign I w n ˜ ∗ 1 − sign R w n ˜ ∗ R w n ˜

[0071] I and R denote respectively the imaginary part and the real part of the symbol.

[0072] There figure 4b illustrates an example of phase variation as calculated via the above formula.

[0073] In step 403, a filter proportional to the error in is applied, i.e. a gain gp .

[0074] In step 404, an integrating filter is applied to the error in , for example, which performs the calculation ψ n ˜ = g 1 e n + ψ n − 1 ˜ .

[0075] The outputs of the two filters 403,404 are summed to produce a proportional-integral system.

[0076] At step 405, a loop filter is finally applied to provide the final estimate of the phase y n = K 0 ψ n -1 + y n -1 .

[0077] Without departing from the scope of the invention, other implementations are possible for producing a phase-locked loop, in particular other types of filters can be implemented.

[0078] We then resume the sequence of steps of the method described in figure 5 .

[0079] In step 505, the phases yn calculated for all the symbols of all the segments are provided as input to a linear regression step 505 which aims to determine the slope and the ordinates at the origin of the lines described in figure 1 .

[0080] In other words, we are looking for the parameters α and β = [β 0 ... β NS-1 ] T< which minimize the function: f α β = ∑ s = 0 N seg − 1 ∑ n = sM sM + N − 1 y n − αn − β s 2

[0081] This optimization can be solved by a numerical resolution algorithm. An example of a possible resolution is presented here.

[0082] By deriving with respect to each of the parameters: ∂ f ∂ α = − 2 ∑ s = 0 N seg − 1 ∑ n = sM sM + N − 1 n y n − αn − β s ∂ f ∂ β s = − 2 ∑ n = sM sM + N − 1 y n − αn − β s pour s ∈ 0 ; N seg − 1

[0083] We define S 1,s as the sum of the symbols yn on the segment s: S 1 , s = ∑ n = sM sM + N − 1 y n

[0084] We define S 2 as the double sum of all the symbols yn including the guard interval symbols: S 2 = ∑ m = 0 N seg − 1 M + N − 1 ∑ n = 0 m y n

[0085] We can also write: S 2 = ∑ n = 0 N seg − 1 M + N − 1 N seg − 1 M + N − n y n = N seg − 1 M + N ∑ s = 0 N seg − 1 S 1 , s − ∑ n = 0 N seg − 1 M + N − 1 ny n

[0086] The resolution of the following system ∂ f ∂ α = 0 ∂ f ∂ β s = 0 pour s ∈ 0 ; N seg − 1 then gives us the following results: α = 6 N seg N + 1 S 1 , N S − 1 + 2 M + N + 1 S 1 , N S − 2 + ⋯ + 2 N seg − 1 M + N + 1 S 1 , 0 − 2 S 2 N N + 1 N − 1 β 0 = S 1 , 0 N − α 2 N − 1 β 1 = S 1 , 1 N − α 2 2 M + N − 1 β s = S 1 , s N − α 2 2 sM + N − 1 β N seg − 1 = S 1 , N s − 1 N − α 2 2 N seg − 1 M + N + 1

[0087] The computational complexity to obtain these values ​​is low. For example, in a case with two segments we obtain: α = 3 N + 1 S 1 , 1 + 2 M + N + 1 S 1 , 0 − 2 S 2 N N + 1 N − 1 β 1 = S 1 , 0 N − α 2 N − 1 β 2 = S 1 , 1 N − α 2 2 M + N − 1

[0088] The unit of the parameters β s is the same as the unit of the symbols yn . Thus, the values ​​of the parameters β s give the estimates of the respective phase deviations between the signals emitted by each antenna and the received signal.

[0089] The slope α is an estimate of the term 2πΔfT PRP and therefore makes it possible to obtain an estimate of the term Δf which is the carrier frequency difference between the received signal and the transmitted signal.

[0090] The phase and frequency deviation estimates are thus calculated in step 506. In particular, the calculation of the differences β j - β i gives the phase deviations, in reception, between signals emitted by two different antennas.

[0091] From the phase differences β j - β i , an additional step (not shown in the figure 5 ) allows the direction of signal transmission to be determined according to geometric principles illustrated in figure 6 .

[0092] On the figure 6 , we have schematically represented two antennas T 1 , T 2 separated by a distance d and transmitting a signal in a transmission direction θ.

[0093] By geometry, with the notations of the figure 6 , we have: p = d sin θ

[0094] Noting δ the difference of the phase deviations estimated in step 506 (i.e. β 2 - β 1 converted to radians) and λ the wavelength, knowing that the phase difference is caused by the difference in antenna path length T 1 to the receiver and antenna T 2 to the receiver, we have: δ = 2 π λ p

[0095] Equality is modulo 2 π and there is no ambiguity if p / λ < 0.5 so if d < λ / 2 (because p ≤ d in all situations).

[0096] We deduce the relationship between the arrival phase difference δ and the starting angle θ : θ = arcsin δλ 2 πd

[0097] If d ≥ λ / 2, other solutions appear by adding multiples of 2 π has δ .

[0098] There figure 7 represents a performance diagram of the method according to the invention obtained by simulation.

[0099] A simulation is carried out by choosing the following parameters: Number of pulses per segment: N = 4096 or N = 32768 Number of segments: N seg = 4 The signal-to-noise ratio is given by the energy per pulse divided by the power spectral density of the noise: E p / N 0 = -5 dB or E p / N 0 = -1 dB Carrier Frequency Offset (CFO) = 400kHz

[0100] An entire frame is simulated in configuration 1. Initial synchronization is performed on the packet preamble by correlating the received signal with the known preamble sequence. The preamble sequence (defined by the IEEE802.15.4z standard) has zero autocorrelation when the sequences are not aligned, so both an estimate of the channel response to an impulse and time synchronization can be obtained. Thus, the index can be determined n 0 mentioned above.

[0101] We assume that the antennas are separated by less than half a wavelength, so the arrival time differences between antennas are less than 143 ps (for a central frequency at 3.5 GHz) so we can consider that the synchronization instant n 0 is constant (The sampling period T ech being of the order of 500 ps).

[0102] The preamble synchronization step also allows the CFO carrier frequency deviation to be estimated (for example by looking at the phase variation between two correlation peaks). The symbol rate is also subject to a transmitter-receiver deviation (SFO) but since the different frequencies come from the same oscillator (receiver side and transmitter side) the relative CFO and SFO deviations are identical in this example.

[0103] The performances obtained in simulation are given on the figure 7 The signal-to-noise ratio of -5dB corresponds to a very good sensitivity for a receiver compatible with the IEEE802.15.4z standard.

[0104] The estimation error (in absolute value) on the phase difference between two antennas corresponding to two segments identified in the legend is represented on the abscissa and the ratio of estimates which have an error greater than the value on the abscissa (empirical inverse distribution function) is represented on the ordinate. The legend also gives the standard deviation of the error σ.

[0105] In an alternative embodiment of the invention, the additional steps 507 and 508 of the figure 5 are applied to reconstruct the impulse response of the transmission channel.

[0106] For this, in step 507, the phase information provided by the phase-locked loop 504 is used to correct the received signal: z n ˜ = z n ⋅ exp − i y n ˜ y n ˜ is obtained by oversampling the phase estimate y n provided by the phase-locked loop 504. Indeed, the phase estimates y n are provided at the rate of one sample per pulse period and it is necessary to return to the sampling rate of the signal zn.

[0107] Then, in step 508, the impulse response of the channel is estimated by performing a cross-correlation between the symbols corrected in step 507 and the corresponding symbols that have been transmitted c n * : R s k = ∑ n = 0 N − 1 Z ˜ n 0 + n × N PRP + k + s × M × N PRP × c n *

[0108] The impulse response R s is determined for each segment with k varying within a predetermined range of sample index values ​​depending on the desired extent of the impulse response.

[0109] The method described in the figure 5 assumes that the propagation channel does not vary or varies little depending on the transmitting antennas, because these antennas are sufficiently close.

[0110] In the case where this hypothesis is no longer verified, for example because the antennas are far apart, an alternative embodiment of the invention is proposed, as illustrated in figure 8 In this variant, we consider that the instant n 0 corresponding to the start of the STS field and determined by the prior time synchronization is no longer identical for each segment.

[0111] This new embodiment variant is represented by the diagram of the figure 8 which covers all steps 501 to 508 of the figure 5 and adds two additional steps 801,802.

[0112] In the general case where the channel response varies significantly between antennas, a synchronization must be repeated to determine the indices n i for each segment.

[0113] Starting from the received signal z n , after step 501 and via step 801, compensation for the carrier frequency deviation CFO is carried out Δf ^ obtained when synchronizing on the P preamble to obtain the signal z̃ n corrected: z ˜ n = z n × exp − i 2 π Δf ^ n T ech

[0114] Note that if the frequency difference between the transmitter and the receiver is sufficiently small, that is to say if the phase rotation is sufficiently small in front of π on the correlation length which will be executed for synchronization in step 802, thanks to the precision of the oscillators, this compensation carried out in step 801 is unnecessary.

[0115] Then, in step 802, a cross-correlation calculation is performed to determine the impulse response of the propagation channel on the segment s.

[0116] This 802 estimation is carried out at the beginning of each segment in order to estimate the channel corresponding to each transmitting antenna. Once again, it is assumed here that all the antennas have the same CFO carrier frequency deviation with the receiver (which is the case if the antennas are on the same device and are fed in turn by the same signal source).

[0117] In addition, synchronization on the preamble makes it possible to know in which window the first pulse of each STS field is located, knowing the distance between the transmitting antenna of the preamble and the transmitting antenna of the segment considered. If we call d this distance, the pulse (compared to that which would have been emitted if the antenna had not been changed) will be shifted by a maximum of ± d / c Or c is the propagation speed of the wave.

[0118] At the start of each segment, step 802 aims to calculate the intercorrelation between the symbols corrected in step 801 and the symbols emitted (same principle as in step 508). R s k = ∑ n = 0 N part − 1 z ˜ n pre + n × N PRP + k + s × M × N PRP × c n *

[0119] The answer R s [ k ] is calculated for k such that the set of k includes at least the interval − d / c × T ech ; d / c × T ech , n pre is the position index of the first pulse of the STS field if it had been transmitted with the antenna which transmitted the preamble on which this value n pre is estimated, N PRP is the number of samples per period, N part is the number of pulses of each segment of the STS field on which synchronization is carried out (this number being chosen according to the targeted sensitivity).

[0120] The chosen criterion is then applied to determine the synchronization times. n s , for each segment s, for example if we choose the strongest path: n s = argmax k R s k + n pre

[0121] Once the moments n s determined, we apply the same treatment as previously (where we had n s = n pre ), namely steps 501 to 508 described previously.

[0122] In the case where the invention is applied for a C1 configuration of the IEEE802.15.4z standard, that is to say an STS field located between the SFD frame start field and the PHR physical layer header, it is necessary to be able to start the coherent demodulation of the data at the start of the PHR field.

[0123] This assumes that the symbols are aligned in phase. The antenna that transmits the PHR header is the one that transmits the preamble and the SFD field, so it is necessary to be able to compensate for the phase variations (mainly due to the CFO carrier frequency deviation) that occur during the time interval corresponding to the STS field. To achieve this, the invention exploits the phase tracking that is performed on the STS field and then subtracts the phase jumps caused by antenna changes.

[0124] The invention thus gives good performance as illustrated in the figure 9 which represent two packet error rate curves as a function of a signal-to-noise ratio respectively without the presence of an STS field and therefore without the invention (curve 901) and with a configuration C2 for which the invention is implemented on STS fields present in the transmitted frames.

[0125] The conditions for carrying out the simulation illustrated in figure 9 are as follows: When the STS field is present (curve 902) Number of pulses per segment: N = 32768 Number of segments: N seg = 4 The propagation channel is of the residential type. We have 8 pulses per bit, so we must subtract 9 dB to have a signal-to-pulse-to-noise ratio E p / N 0 from the signal-to-noise ratio at the bit level E b / N 0 The carrier frequency deviation is equal to CFO = 400 kHz, N part = 1024 pulses dedicated to initial synchronization.

[0126] There figure 9 , shows that for the same packet error rate (PER) a loss limited to 0.5dB is observed when the invention is applied (curve 902) compared to an ideal case (901).

[0127] The invention can be implemented by means of an ultra-wideband receiver REC which comprises an antenna ANT-R1, a reception and digitization channel NUM and a computer CALC configured to execute the steps of the method according to the invention. Such a receiver is shown diagrammatically in figure 10 .

[0128] The CALC calculator can be implemented in software and / or hardware form, in particular by using one or more processor(s) and one or more memory(s). The processor can be a generic processor, a specific processor, an application-specific integrated circuit (also known as ASIC for “Application-Specific Integrated Circuit”) or an in situ programmable gate network (also known as FPGA for “Field-Programmable Gate Array”).

[0129] More broadly, the invention can be implemented within an ultra-wideband communications system, as described in figure 10, comprising a receiver REC of the type described above and at least one ultra wideband EM transmitter having at least two transmitting antennas ANT-E1, ANT-E2.

[0130] The invention is compatible with the IEEE 802.15.4z standard or any other standard involving IR-UWB modulation and for which segments dedicated to each transmission antenna are provided in the format of the transmission frames and comprise symbols known to the transmitter and the receiver, for example pilot symbols.

[0131] Advantageously, the sequences of symbols used in the planned segments are secured by means of an encryption algorithm or cryptography, so as to secure their access so that only the transmitter and the receiver have access to these sequences.

[0132] For example, a symmetric encryption protocol such as AES can be implemented between the transmitter and receiver of the system to pre-exchange the values ​​of the sequences used in the segments.

[0133] The invention notably provides assistance with radiolocation and secure access control by making it possible to locate a transmitter from its angular transmission direction.

Claims

1. A method for estimating at least one characteristic of a signal received by a receiver, the signal having been transmitted in succession by a plurality of antennas in successive time segments (S1, S2), each segment (S1, S2) being dedicated to one separate antenna, the signal being modulated into the form of pulses according to ultra-wideband modulation, the method comprising steps of: - receiving and digitizing the signal, - computing (503) the product of multiplication of each symbol of the received and digitized signal by the complex conjugate of the corresponding transmitted symbol, which is known to the receiver, - for each segment and for each symbol of the signal received for this segment, estimating a phase error by means of a phase-locked loop (504) applied to said product, - for each segment, determining (505) a reference phase by means of a linear regression applied to the phase errors estimated for all of the segments, - determining (506), for at least one pair of antennas, a phase difference between the signals transmitted by the antennas of the pair, on the basis of the difference between the reference phases computed for the segments associated with said antennas.

2. The estimating method according to Claim 1, further comprising, conjointly with the determination (506) of the phase difference, determining a carrier frequency offset between the signal transmitted by all of the antennas and the received signal by means of said linear regression (505) applied to the phase errors estimated for all of the segments.

3. The estimating method according to either one of the preceding claims, wherein the successive time segments (S1, S2) are separated by guard intervals (G1, G2, G3) and the method further comprises removing these guard intervals from the received and digitized signal.

4. The estimating method according to any one of the preceding claims, further comprising a step of determining an angular direction of transmission of the signal on the basis of said phase differences.

5. The estimating method according to any one of the preceding claims, further comprising steps of: - correcting (507) each symbol of the received and digitized signal for the computed phase error by means of the phase-locked loop, - estimating (508) an impulse response of the propagation channel by means of a computation of an intercorrelation between the sequence of the corrected received symbols and a sequence of the symbols transmitted in the segments.

6. The estimating method according to any one of the preceding claims, comprising beforehand an initial synchronizing phase specific to each segment comprising steps of: - determining, in a preamble sequence prior to the first segment, a first estimation of a carrier frequency offset between the received and digitized signal and the transmitted signal, - correcting (801) the digitized received signal using this first estimation, - estimating (802), for the start of each segment, an impulse response of the propagation channel by means of a computation of an intercorrelation between a sequence of corrected received symbols and a sequence of the corresponding transmitted symbols, - determining, on the basis of the impulse response estimated for the start of each segment, a symbol corresponding to the start of the segment.

7. The estimating method according to any one of the preceding claims, wherein the sequences of symbols transmitted in the time segments are secured by means of an encryption algorithm.

8. A receiver of a signal modulated into the form of pulses according to ultra-wideband modulation, comprising an antenna and a computer configured to execute the steps of the method for estimating at least one characteristic of the received signal according to any one of the preceding claims.

9. A communication system comprising a transmitter of a signal modulated into the form of pulses according to ultra-wideband modulation comprising a plurality of antennas, the signal being transmitted in succession by a plurality of antennas in successive time segments, each segment being dedicated to one separate antenna, the system further comprising a receiver according to Claim 8.