Circuit for detecting complex alternating current resistance

Abstract

The invention relates to a circuit for detecting complex alternating current resistance, said circuit comprising: a signal source (14) which is designed to generate an alternating current excitation signal; and a first signal path (20), a second signal path (22), and a third signal path (24), the alternating current excitation signal from the signal source (14) being fed into each of said signal paths in parallel with each other. The alternating current excitation signal is fed into the first signal path (20) or into the second and third signal paths (22, 24) at least temporarily with a phase offset relative to the alternating current excitation signal generated by the signal source (14). The first signal path (20) has a high-pass filter (30) and, in series therewith, the alternating current resistance to be detected. A measurement signal tapped at a measurement point (32) between the high-pass filter (30) and the alternating current resistance to be detected is mixed with the alternating current excitation signal in the second signal path (22) and furthermore demodulated in order to obtain a first demodulated measurement signal. The measurement signal tapped at the measurement point (32) is also mixed with the alternating current excitation signal in the third signal path (24) and furthermore demodulated in order to obtain a second demodulated measurement signal. A signal evaluation unit (56), which receives the demodulated measurement signals, determines, on the basis of the demodulated measurement signals, the imaginary part and real part of the alternating current resistance to be detected.

Description

Circuit for detecting a complex AC resistance

[0001] The invention relates to a circuit for detecting a complex AC resistance.

[0002] Such a circuit can be used for an impedance sensor in various applications.

[0003] Document EP 3 512 099 B1 discloses a capacitive sensor comprising a sensor electrode, wherein the sensor electrode is connected to a signal generation circuit for generating an output signal to the sensor electrode, and wherein the sensor electrode is further connected to a signal evaluation circuit configured to evaluate an input signal from the sensor electrode. The signal evaluation circuit includes a synchronous rectifier, wherein the synchronous rectifier has a first and a second switch connected to the signal generator and configured for alternating switching in phase with the input signal. are designed. This known sensor can only detect the imaginary part of a complex AC resistance.

[0004] Further capacitive sensors are known from US 2015 / 0323372 A1 and DE 102012 201 226 B4. The latter document discloses a probe for a capacitive level sensor with a measuring impedance, a reference impedance, a first rectifier and a second rectifier, two measuring resistors, a ground connection, and a connection designed for attaching a multi-core connecting cable, wherein the connecting cable has a first core for transmitting a signal and four further cores for transmitting DC voltage signals. This circuit is comparatively complex.

[0005] DE 102018209 904 A1 discloses a level sensor or limit level sensor with temperature compensation. The sensor has a processing unit for processing a measurement signal generated by the sensor and a reference unit for generating a reference signal, wherein the processing unit and the reference unit each have a signal conversion unit with temperature-dependent signal conversion.

[0006] The invention is based on the objective of providing a circuit for detecting an unknown complex AC resistance, which is simple in design and with which it is particularly possible to detect the AC resistance with respect to the imaginary part and the real part.

[0007] To solve this problem, a circuit for detecting a complex AC resistance is provided, comprising a signal source designed to generate an AC excitation signal, a first signal path, a second signal path, and a third signal path, into each of which the AC excitation signal from the signal source is fed in parallel, wherein the AC excitation signal is fed into the first signal path or into the second and third signal paths at least temporarily with a phase shift relative to the AC excitation signal generated by the signal source, wherein the The first signal path comprises a high-pass filter and, in series with it, the AC resistance to be measured, wherein a measurement signal tapped at a measuring point between the high-pass filter and the AC resistance to be measured is mixed with the AC excitation signal in the second signal path and further demodulated to obtain a first demodulated measurement signal, and is mixed with the AC excitation signal in the third signal path and further demodulated to obtain a second demodulated measurement signal, and with a signal evaluation unit that receives the demodulated measurement signals and determines the AC resistance to be measured from the demodulated measurement signals according to the imaginary and real parts.

[0008] The circuit according to the invention can detect an unknown AC resistance, which is connected, for example, to a measuring electrode against a reference potential, where the reference potential can be the reference potential of the signal source. The circuit includes a signal source that generates an AC excitation signal, the spectrum of which is preferably selected to utilize a broad frequency spectrum or signal band. The AC excitation signal is fed into a first signal path, which includes a high-pass filter and, in series with this, the AC resistance to be detected. The high-pass filter and the AC resistance to be detected form an AC voltage divider. Parallel to the injection of the AC excitation signal into the first signal path, the AC excitation signal is also fed into a second signal path and a third signal path.The measurement signal used to determine the complex AC resistance is tapped between the high-pass filter and the AC resistance to be measured. The circuit according to the invention further provides for the setting of a phase shift, at least temporarily. When the phase shift is activated, an AC excitation signal, phase-inverted relative to the AC excitation signal generated by the signal source, is fed into the first signal path or into the second and third signal paths. The AC resistance measurement can be performed alternately with the phase shift activated and deactivated. The phase shift can be, for example, 0° (deactivated), 90°, 180°, or 270°, including -90°, -180°, or -270°. The measurement signal tapped at the measuring point between the high-pass filter and the AC resistance to be measured is fed into the second signal path. The signal path and the third signal path are mixed with the AC excitation signal. With phase shift enabled, the phase-shifted measurement signal is mixed with the non-phase-shifted AC excitation signal and demodulated, or the non-phase-shifted measurement signal is mixed with the phase-shifted AC excitation signal, depending on whether the phase shift is present in the first signal path or in the second and third signal paths. The phase shift can be achieved using a phase shifter. The mixing and demodulation process is comparable to amplitude demodulation. The signal evaluation unit receives the signal, if any.low-pass filtered demodulated measurement signals and determines the AC resistance to be measured based on these signals according to the imaginary and real parts, which is made possible by the temporarily switched-on phase shift, which enables not only an amplitude determination but also a phase determination of the AC resistance.

[0009] The circuit according to the invention has a simple design and thus enables the simple detection of a complex AC resistance with respect to the reactive and active components.

[0010] Preferred embodiments are specified in the dependent claims or are described below.

[0011] In a preferred embodiment, the second signal path can have a first diode and a first measuring capacitor, wherein the measuring capacitor is connected to the measuring point between the high-pass filter and the AC resistance to be measured, and the third signal path can have a second diode and a second measuring capacitor, wherein the second measuring capacitor is connected to the measuring point between the high-pass filter and the AC resistance to be measured, wherein the first and the second diode are arranged to alternately pass through a half-wave of the AC voltage excitation signal.

[0012] The two measuring capacitors, together with the two diodes, form a simple arrangement for mixing the measurement signal taken at the measuring point with the respective signal. The circuit is designed to separate the AC excitation signal from the second and third signal paths and to demodulate the resulting mixed signals. The diodes are connected to the measurement point between the high-pass filter and the complex AC impedance via the measuring capacitors. The circuit is designed such that during the positive half-cycle of the AC excitation signal from the signal source, one diode is conducting and a current flows through the associated measuring capacitor, charging it to a measurement voltage proportional to the AC voltage divider. During the negative half-cycle of the AC excitation signal from the signal source, the other diode is conducting, so a charging current then flows through the associated measuring capacitor. The high-pass filter and the two measuring capacitors separate the measurement signal at the measurement point from the DC component of the AC excitation signal.

[0013] Preferably, the first measuring capacitor and / or the second measuring capacitor have a capacitance in the picofarad range.

[0014] Furthermore, an inverter can be arranged in the second or third signal path, which is connected to the signal source and is connected in front of the first or second diode, and which inverts the AC voltage excitation signal coming from the signal source, i.e. rotates it by 180°.

[0015] In this configuration, the two diodes with the same polarity can be arranged in the second and third signal paths. However, it is also possible to connect the two diodes with opposite polarities to the signal source and, if necessary, couple in a bias voltage, thus eliminating the need for an inverter.

[0016] Furthermore, the high-pass filter can contain a capacitor, in particular only a capacitor.

[0017] This further simplifies the circuit in terms of its complexity. The capacitance of the high-pass capacitor can be selected depending on the expected range of values ​​of the AC resistance to be measured. It is preferred if the largest possible voltage swing is achieved between the measurement signal level at a low measurement impedance and a high measurement impedance.

[0018] It is understood that further filter and protection elements may be arranged between the high-pass filter and the measuring electrode to which the complex AC resistance is connected, in order to optimize the EMC characteristics of the circuit.

[0019] The frequency spectrum of the AC excitation signal generated by the signal source preferably encompasses frequencies within a broad frequency band. Preferably, the achievable frequencies lie in the range of 100 kHz to 200 MHz, depending on the signal source used. The circuit can, for example, be operated at frequencies between 5 and 50 MHz. A voltage-dependent oscillator (VCO) controlled by a microcontroller can be used as the signal source, with the control voltage being determined by the input range of the voltage-dependent oscillator and serving to generate a broadband output spectrum at the output of the voltage-dependent oscillator. Alternatively, instead of a voltage-dependent oscillator, an oscillating signal source contained within a microcontroller can be modulated accordingly to output a broadband frequency spectrum.

[0020] Preferably, the signal source is designed to generate the AC excitation signal with a time-varying frequency. For this purpose, the signal source can, for example, be designed as a sweep generator. This allows for a simple improvement in the EMC characteristics of the circuit.

[0021] Preferably, the circuit can include a first low-pass filter and a second low-pass filter into which the demodulated measurement signals are fed. The AC signal component can be separated from the demodulated measurement signals by means of the low-pass filters. An analog-to-digital converter (ADC) can be connected downstream of each low-pass filter to digitize the measurement signals. Alternatively, only one ADC can be used, in which case the optional subtraction stage mentioned above is performed via an analog stage before digitization.

[0022] The signal evaluation unit can preferably be designed to subtract the demodulated measurement signals from each other. This has the advantage that, firstly, the measurement signal is amplified, and secondly, external interference signals are filtered out. The complex AC resistance can then be determined from the difference between the measurement signals.

[0023] Furthermore, according to the invention, an impedance sensor is provided which has a measuring electrode to which the complex AC resistance to be detected is connected against a reference potential, and has a circuit according to one or more of the aforementioned embodiments.

[0024] The impedance sensor according to the invention can be used in a wide variety of applications. In particular, the impedance sensor is capable of detecting not only the capacitance between the measuring electrode and the reference potential, especially ground potential, but also the conductive coupling between the measuring electrode and the reference potential, which is determined by the real part of the complex AC resistance. This makes it possible, for example, to differentiate objects or to detect contamination.

[0025] For example, the impedance sensor can be used as a level sensor to continuously monitor liquid media or bulk solids in a tank. The measuring electrode can be designed as a rod. A conductive casing or the conductive wall of the tank can serve as the counter electrode. The measuring electrode and the counter electrode together form a capacitor. The capacitance value is continuously changed by the fill level in the tank and can be measured by the circuit according to the invention. Conductive contaminants such as biofilms or other deposits in the tank can be detected via the conductive measurement (conductance) of the circuit according to the invention. Changes in the media properties, which are reflected in the conductivity of the process medium, can also be detected via the conductive measurement.

[0026] The impedance sensor can also be used as a level switch for a process medium. In this case, the measuring electrode can be designed, for example, as a cap. When the process medium contacts the cap of the level switch, the measured capacitance changes significantly, triggering a switching action. This makes it possible, for example, to implement dry-run protection for pumps or overflow protection when filling open containers. Here, too, the conductive measurement can be used to detect contamination or changes in the process medium.

[0027] Furthermore, the impedance sensor can be used as a proximity sensor, for example in the automation industry. Similar to a limit switch, a switching action can be triggered. As soon as an object approaches the measuring electrode, the capacitance between the measuring electrode and the reference potential, e.g., ground potential, changes. If the signal change due to this capacitance is sufficiently large, a switching action can be triggered. The conductive measurement can be used to distinguish objects. It can also be used to detect conductive deposits on the measuring electrode of the proximity sensor. This makes it possible to continue detecting objects in processes where, for example, water films or puddles form on the proximity sensor.

[0028] Further possible uses of the impedance sensor according to the invention are in the field of human-machine interaction in the form of pushbuttons, slide or rotary encoders. In addition, the impedance sensor according to the invention can be used as a flow monitor, in which the properties of the process medium with regard to capacitive or conductive coupling to the sensor are monitored.

[0029] Further advantages and features will become apparent from the following description and the attached drawings.

[0030] It is understood that the aforementioned and subsequently explained characteristics exist not only in the combinations specified, but also in other combinations. They can be used in combination or on their own without leaving the scope of the present invention.

[0031] Exemplary embodiments of the invention are shown in the drawing and are described in more detail below with reference to them. The drawing shows: Fig. 1 shows a basic circuit diagram of an AC voltage divider with an AC resistor; Fig. 2 shows a circuit diagram of a first embodiment of a circuit for detecting a complex AC resistance; Fig. 3 shows a circuit diagram of another embodiment of a circuit for detecting a complex AC resistance; Fig. 4 shows a circuit diagram of another embodiment of a circuit for detecting a complex AC resistance; Fig. 5 shows a circuit diagram of another embodiment of a circuit for detecting a complex AC resistance; Fig. 6 shows a circuit diagram of another embodiment of a circuit for detecting a complex AC resistance; Fig. 7 shows a circuit diagram of another embodiment of a circuit for detecting a complex AC resistance; Fig. 8 shows a circuit diagram of another embodiment of a circuit for detecting a complex AC resistance; and Fig. 9 shows a schematic sketch of a possible use of an impedance sensor.

[0032] Fig. 1 shows a schematic diagram of an AC voltage divider with an unknown, in particular variable, AC resistance Z, which is designated by reference numeral 1 in Fig. 1. The unknown AC resistance Z is connected to a measuring electrode 2 against a reference potential 3. To measure the unknown AC resistance Z, the measuring electrode 2 is connected to a signal source 4 via an internal coupling capacitor CK, as shown by reference numeral 5 in Fig. 1. The reference potential 3 can be the reference potential of the signal source 4. The coupling capacitor CK, together with the unknown AC resistance Z, forms an AC voltage divider for the voltage of the signal source 4. Ideally, the spectrum of the signal source 4 should be selected to utilize a wide signal band.This serves two purposes: firstly, to distribute the radiation pattern across many frequencies, and secondly, to reduce the sensitivity of the measurement to external signal sources. The measurement voltage V of the AC voltage divider, consisting of the coupling capacitor CK and the AC resistance Z to be determined, is tapped at a measuring point between the coupling capacitor CK and the AC resistance Z. In Fig. 1, the measurement voltage V is tapped directly at the measuring electrode 2 in parallel with the AC resistance Z.

[0033] With reference to Figs. 2 to 4, exemplary embodiments of circuits are described below with which the complex AC resistance Z can be determined in a simple way.

[0034] Fig. 2 shows a circuit 10 for detecting a complex AC resistance connected to a measuring electrode 12 against a reference potential, as described with reference to Fig. 1. The AC resistance Z is not shown in Fig. 2.

[0035] Circuit 10 includes a signal source 14 designed to generate an AC excitation signal. In the embodiment shown in Fig. 2 Signal source 14 includes a voltage-dependent oscillator (VCO) 16. A microcontroller (PC) 18 generates a control voltage for the voltage-dependent oscillator 16. The control voltage is determined by the input range of the voltage-dependent oscillator and serves to generate a broadband output spectrum at the output of the voltage-dependent oscillator 16. The control voltage can be output via a digital-to-analog converter integrated in the microcontroller 18. Alternatively, a pulse-width modulation (PWM) signal can be generated with the microcontroller 18 and this signal converted into a low-frequency control voltage for the voltage-dependent oscillator, for example, using a low-pass filter. The control signal for the voltage-dependent oscillator can be varied during a sweep time.The timing of the control voltage can be selected so that the frequency spectrum generated by the voltage-dependent oscillator has optimal EMC characteristics. The sweep time, for example, can range from a few milliseconds to several seconds.

[0036] The frequency spectrum of the AC excitation signal output by the voltage-dependent oscillator 16 can range from a few kHz to several MHz. For example, the frequency spectrum can start at 70 MHz and extend up to a frequency of 150 MHz. This broadband AC signal spectrum serves to improve the EMC characteristics of the entire circuit 10. Firstly, the radiated energy is distributed across many frequencies, and secondly, the susceptibility to interference from external signal frequencies is reduced.

[0037] Starting from the signal source 14, the circuit 10 has a first signal path 20, a second signal path 22 and a third signal path 24, into which the AC voltage excitation signal output by the signal source 14 is fed in parallel to each other.

[0038] The AC excitation signal is fed into the first signal path, at least temporarily, with a phase shift. For this purpose, a phase shifter 26 (Arp) is arranged in the first signal path 20. The phase shifter 26 causes a phase shift of its output signal relative to the injected AC excitation signal. The phase shifter 26 can be controlled by the microcontroller 18, as indicated by a dashed line 28. The microcontroller 18 can specify the phase offset between the input signal fed into the signal path 20 and the output signal at the output of the phase shifter 26. Typically, the phase offset is set to 0°, 90°, 180°, or 270°. However, other phase offsets can also be used.

[0039] The output signal of the phase shifter 26 is coupled into a high-pass filter 30. As will be shown in embodiments described later, the high-pass filter 30 can be implemented as a capacitor, corresponding to the coupling capacitor 5 in Fig. 1. The high-pass filter 30, together with the externally measurable complex AC resistance connected to the measuring electrode 12, forms an AC voltage divider, as described above with reference to Fig. 1. At a measuring point 32, which, as described with reference to Fig. 1, can be located directly at the measuring electrode 12, the measuring voltage, from which the complex AC resistance is determined as described below, is tapped relative to the reference potential, e.g., the reference potential of the signal source 14.

[0040] If the high-pass filter 30 is implemented as a capacitor, its capacitance is selected based on the expected range of values ​​for the complex AC resistance to be measured. The aim is to achieve the largest possible voltage swing between the AC voltage level of the measurement signal at a low measured impedance and a high measured impedance.

[0041] It is understood that further filter and protection elements may be arranged between the high-pass filter 30 and the measuring electrode 12, for example to optimize the EMC properties of the circuit 10.

[0042] The AC excitation signal output by signal source 14 is fed in parallel to the first signal path 20 into the second signal path 22 and the third signal path 24. The AC excitation signal is shown in the In this embodiment, the signal paths 22 and 24 are not phase-shifted relative to the output signal of the signal source 14.

[0043] The AC voltage measurement signal tapped at measuring point 32 is mixed with the AC voltage excitation signal fed into the second signal path 22 in a first mixer-demodulator 34, which is arranged in the second signal path 22, and the mixed signal is then demodulated. Likewise, the AC voltage measurement signal tapped at measuring point 32 is mixed with the AC voltage excitation signal fed into the third signal path 24 in a second mixer-demodulator 36, and the mixed signal is demodulated. The first mixer-demodulator 34 comprises a first mixer 38 and a first demodulator 40. The second mixer-demodulator 36 comprises a second mixer 42 and a second demodulator 44. The first and second mixer-demodulator 34 are preferably designed similarly to an amplitude demodulator and an envelope demodulator, respectively.They can have a diode-capacitor arrangement, as shown in embodiments described later. In the case of an embodiment of the mixer-demodulator elements 34, 36 as a diode-capacitor arrangement, an inverter 46 can be arranged in one of the signal paths 22 or 24, here in signal path 24, which inverts the phase of the AC excitation signal fed into signal path 24 by 180°. Thus, the diodes in the first and second mixer-demodulator elements 34, 36 can be arranged with the same polarity to each other. In the case of an embodiment of the mixer-demodulator elements 34, 36 as a diode-capacitor arrangement, the respective diode has the lowest possible junction capacitance, and the respective capacitor also typically has a capacitance in the picofarad range.

[0044] The respective demodulated measurement signals output by demodulators 40 and 44 are fed to low-pass filters 48 and 50, respectively, which separate the measurement signals from RF components. Low-pass filters 48 and 50 are dimensioned according to the desired step response of circuit 10 to changes in the complex AC resistance being measured. The low-pass filtered AC voltage measurement signals are then fed to analog-to-digital converters 52 and 54, respectively, which digitize the measurement signals. When dimensioning low-pass filters 48, 50 The antialiasing criterion of the analog-to-digital converters 52, 54 can also be taken into account.

[0045] The digitized measurement signals are evaluated in a signal evaluation unit 56, which can be integrated into the microcontroller 18, to calculate the imaginary and real parts of the AC resistance to be determined. Preferably, the digitized measurement signals supplied to the signal evaluation unit 56 are subtracted from each other. The measurement is performed temporarily, for example alternately, with and without a phase shift, and, depending on the phase shift set on the phase shifter 26, the imaginary or real signal component of the complex AC resistance to be measured can be selectively determined from these measurements.

[0046] Figure 3 shows a modified embodiment of circuit 10a compared to Figure 2. Elements of circuit 10a that are identical or comparable to elements of circuit 10 are provided with the same reference numerals as in Figure 2.

[0047] The following describes only the differences between circuit 10a and circuit 10.

[0048] In circuit 10a, the phase shifter 26 is not located in the first signal path 20, but in signal paths 22 and 24. In circuit 10a, the measurement signal tapped at measuring point 32, which is not phase-shifted relative to the AC excitation signal output by signal source 14, is thus mixed with the phase-shifted AC excitation signals in signal paths 22 and 24 when a phase offset is set via the phase shifter 26.

[0049] With circuit 10a, as with circuit 10, an unknown AC resistance Z, which is connected to the measuring electrode 12 against a reference potential, can be determined according to its real and imaginary parts.

[0050] Figure 4 shows another embodiment of a circuit 10b for detecting a complex AC resistance. Elements of the circuit 10b, which are connected to Elements of circuit 10 in Fig. 2 that are identical or comparable are provided with the same reference numerals as in Fig. 2. Only the differences between circuit 10b and circuit 10 are described below.

[0051] One difference concerns the design of the signal source 14. Instead of a voltage-dependent oscillator (VCO) as shown in Fig. 2, circuit 10b uses an oscillating signal generator 16 integrated into the microcontroller 18 as the signal source 14. This generator is modulated to output a broadband frequency spectrum. Depending on the microcontroller 18, the output AC excitation signal frequencies are in the kHz to lower MHz range. For example, the frequency spectrum output by the oscillating signal generator 16 can start at 150 kHz and extend up to a frequency of 2 MHz. Integrating the signal source 14 into the microcontroller 18 is particularly advantageous for compact circuit designs, such as in human-machine interfaces in the form of pushbuttons and the like.

[0052] Furthermore, in circuit 10b the analog-to-digital converters 52, 54, the phase shifter 26 and the inverter 46 are also integrated into the microcontroller 18.

[0053] With reference to Figures 5 to 8, further embodiments of circuits are described that operate according to the same basic principle as circuits 10, 10a, and 10b. Elements of the circuits described below that are identical or comparable to elements of circuits 10, 10a, and 10b are designated with the same reference numerals as in Figures 2 to 4.

[0054] Fig. 5 shows a circuit 100 for detecting a complex AC resistance Z, which is connected to a measuring electrode 12 relative to the reference potential of the signal source 14. The measurement signal is tapped at a measuring point 32. The signal source 14 can be configured as described with reference to Fig. 2 or with reference to Fig. 4.

[0055] In circuit 100, the high-pass filter 30 is implemented as a capacitor 60 in Figures 2 to 4. The mixer-demodulator elements 34 and 36 in Figures 2 to 4 are specifically implemented as diode-capacitor arrangements, with a diode (D1) 62 and a measuring capacitor 64, through which the diode 62 is connected to the measuring point 32, and with a diode (D2) 66, which is connected to the measuring point 32 via a measuring capacitor 68. The AC excitation signal from the signal source 14 is fed in parallel to the coupling into the first signal path 20, which contains the AC voltage divider consisting of the capacitor 60 and the AC resistance Z to be measured, into the diode 62 and into the diode 66 in the second and third signal paths 22 and 24. The inverted AC voltage excitation signal of the signal source 14 is connected to the diode D1 via the inverter 46, which is located here in the second signal path 22.Diodes 62 and 66 are connected to signal source 14 with the same polarity to each other.

[0056] During the positive half-cycle of the AC excitation signal from signal source 14, diode 66 (D2) is conducting and a current flows through the measuring capacitor 68. The current through capacitor 68 charges it to a measuring voltage proportional to the AC voltage divider formed by capacitor 60 and the AC resistance Z. During the negative half-cycle of the AC excitation signal from signal source 14, diode 62 is conducting due to inverter 46. While diode 62 is conducting, a charging current flows through the measuring capacitor 64.

[0057] The voltage at measuring point 32 of the measuring electrode 12 is separated from the DC component of the signal source 14 via capacitors 60, 64, and 68. Thus, the respective measurement voltage at the input of the low-pass filters 48 and 50 is equal to the charge of capacitors 64 and 68 plus an AC signal component. The low-pass filters 48 and 50 separate the AC signal component from the measurement signals of the capacitor charges of capacitors 64 and 68. The low-pass filtered measurement signals can then be digitized in analog-to-digital converters ADC1 and ADC2. A phase shifter, indicated by "±90°" in Fig. 5, can be connected in the second and third signal paths 22 and 24. The measurement signals digitized by the analog-to-digital converters ADC1 and ADC2 can then be evaluated in an evaluation unit as described above in order to determine the complex AC resistance Z with respect to the real part and imaginary part.

[0058] In the evaluation circuit, as described above, the difference between the digitized measurement signals is preferably calculated. This increases the resulting measurement signal and also filters out external interference signals.

[0059] The diodes 62 and 66 are preferably housed in a common package to reduce temperature differences between the diodes 62 and 66.

[0060] The circuit 100 in Fig. 5 has, among other advantages, the fact that the resulting measurement voltages have a positive sign, and can therefore be easily processed further, e.g., using the analog-to-digital converter ADC1.

[0061] Resistors R1 and R2 in Fig. 5 serve to divert the bias current through diodes 62 and 66 to a reference potential. The reference potential can be a reference voltage source or, as shown in Fig. 5, the reference potential of the signal source 14. Resistors R1 and R2 also reduce or prevent signal drift caused by the bias currents.

[0062] Fig. 6 shows a circuit 100a, which is a variant of the circuit 100 in Fig. 5. Elements of circuit 100a that are identical or comparable to elements of circuit 100 in Fig. 5 are provided with the same reference numerals as in Fig. 5. Only differences from circuit 100 are described below.

[0063] In circuits 100a and 100, the two diodes 62 and 66 are connected to the signal source 14 with opposite polarity compared to Fig. 5, but with the same polarity between them. A bias voltage may also be necessary. The bias voltage ensures that diodes 62 and 66 are switched to a conducting state when the voltage level at the cathode of the respective diode is sufficiently far below the bias voltage. The current through diodes 62 and 66 is further limited by resistors R1 and R2, respectively.

[0064] The capacitors 64 and 68 of circuits 100 and 100a can have essentially the same capacitance. However, they can also have different capacitances.

[0065] The resistors R1, R2 or the input resistances of the low-pass filters 48, 50 can be many times larger than the AC resistance of the capacitor 66 of the high-pass filter 30 over the frequency range of the signal source 14.

[0066] Diodes 62 and 66 should have the smallest possible junction capacitance.

[0067] In Fig. 6, a phase shifter can be provided at position 70.

[0068] To determine the complex AC impedance Z, in addition to measuring the voltage at measuring point 32 in phase with the AC excitation signal from signal source 14, a phase-shifted measurement signal is required. For this purpose, as shown in Fig. 5, the AC excitation signal is coupled into signal paths 22, 24, for example, with a 90° phase shift. The measurement with and without the phase shift can be performed alternately by switching the phase shift on or off. The voltage at measuring point 32 can be determined for a specific period, for example, the period required to traverse a specific frequency spectrum (Wöbbel period). The measurement is then repeated with a phase shift of, for example, ±90°. The two measurement signals obtained in this way can be used as measured quantities for the complex AC impedance to calculate the amplitude and phase of the AC impedance Z.

[0069] Figures 7 and 8 show further embodiments of circuit 100b and circuit 100c, where again the same reference numerals are used for elements that are identical or comparable to elements of circuit 100 in Figure 5 as in Figure 5. Only differences from circuit 100 are described.

[0070] In Fig. 7, diodes 62 and 66 are arranged with opposite polarity to each other in the second and third signal paths 22 and 24, respectively. Inverter 46 can therefore be omitted.

[0071] The same applies to the circuit 100c in Fig. 8, in which the two diodes 62, 66 are arranged with reversed polarity in the corresponding signal paths 22 and 24 with reference to Fig. 7.

[0072] Fig. 9 shows an impedance sensor 200 comprising the measuring electrode 12 and a circuit 210 connected to the measuring electrode 12. The circuit 210 can be circuit 10, 10a, 10b, 100, 100a, 100b, or 100c.

[0073] In the embodiment shown in Fig. 9, the impedance sensor 200 is used as a level sensor to continuously monitor the level of a liquid medium 220 or a bulk material 220 in a tank 230. The measuring electrode 12 can be designed as a rod. A conductive casing 13 or the wall of the tank 230 can serve as the counter electrode, provided the wall of the tank 230 is at least partially conductive. The measuring electrode 12 and the counter electrode form a capacitor. The value of the capacitor is continuously changed by the level in the tank 230 and can be measured by the circuit 210. Conductive contaminants such as biofilms or other deposits in the tank 230 can be detected via the real part, i.e., the conductive component of the measured AC resistance, by the sensor electronics of the circuit 210.However, changes in the media properties, which are reflected in the conductivity of the medium 220, can also be detected via the conductive measurement.

[0074] Another application of an impedance sensor, implemented or connected to a circuit 10, 10a, 10b, 100, 100a, 100b, 100c, can be its use as a level switch. In this case, the measuring electrode 12 can, for example, be designed as a cap. When the cap of the level switch comes into contact with a process medium, the measured capacitance changes significantly, triggering a switching action. This makes it possible to implement dry-run protection for pumps or overflow protection when filling open containers. Here, too, the conductive measurement can be used to detect contamination or changes in the process medium.

[0075] Another possible application of an impedance sensor with a circuit according to the present embodiments is its use as a proximity sensor in the automation industry. Similar to a limit switch, a switching application is triggered depending on the value of the measured AC resistance. As soon as an object approaches the measuring electrode 12, the capacitance between measuring electrode 12 and ground potential changes. If the signal change due to this capacitance is sufficiently large, a switching action can be triggered. The conductive measurement can be used to distinguish between objects. It can also be used to detect conductive deposits on the measuring electrode 12 of the proximity sensor. This makes it possible to continue detecting objects in processes where, for example, water films or puddles form on the proximity sensor.

[0076] Further applications of an impedance sensor with a circuit according to one of the embodiments described above lie in the field of human-machine interaction in the form of pushbuttons, slide or rotary encoders. Furthermore, applications as flow monitors are possible, where the properties of the process medium with regard to capacitive or conductive coupling to the impedance sensor are monitored.

Claims

Claims: Circuit for detecting a complex AC resistance, comprising a signal source (14) designed to generate an AC excitation signal, a first signal path (20), a second signal path (22), and a third signal path (24), into each of which the AC excitation signal of the signal source (14) is fed in parallel, wherein the AC excitation signal is fed into the first signal path (20) or into the second and third signal paths (22, 24) at least temporarily with a phase shift relative to the AC excitation signal generated by the signal source (14), wherein the first signal path (20) has a high-pass filter (30) and, in series therewith, the AC resistance to be detected.wherein a measurement signal tapped at a measuring point (32) between the high-pass filter (30) and the AC resistance to be measured is mixed with the AC excitation signal in the second signal path (22) and further demodulated to obtain a first demodulated measurement signal, and mixed with the AC excitation signal in the third signal path (24) and further demodulated to obtain a second demodulated measurement signal, and comprising a signal evaluation unit (56) that receives the demodulated measurement signals and determines the AC resistance to be measured from the demodulated measurement signals according to its imaginary and real parts. Circuit according to claim 1, wherein the second signal path (22) comprises a first diode (62) and a first measuring capacitor (64), the measuring capacitor (64) being connected to the measuring point (32) between the high-pass filter (30) and the AC resistance to be measured.and the third signal path (24) has a second diode (66) and a second measuring capacitor (68), wherein the second measuring capacitor (68) is connected to the measuring point (32) between the high-pass filter (30) and the AC resistance to be measured, wherein the first and the second diode (62, 66) are arranged to alternately pass a half-wave of the AC voltage excitation signal. Circuit according to claim 2, wherein the first measuring capacitor (64) and / or the second measuring capacitor (68) has / have a capacitance in the picofarad range. Circuit according to any one of claims 1 to 3, wherein an inverter is arranged in the second or third signal path (22, 24), which is connected to the signal source (14) and is connected upstream of the first diode (62) or the second diode (66), and which inverts the AC excitation signal coming from the signal source (14). Circuit according to any one of claims 1 to 4, wherein the high-pass filter (30) comprises a capacitor (60). Circuit according to any one of claims 1 to 5, wherein a frequency spectrum of the AC excitation signal generated by the signal source (14) comprises frequencies in a range from 100 kHz to 200 MHz. Circuit according to one of claims 1 to 6, wherein the signal source (14) is configured to generate the AC voltage excitation signal with variable frequency.Circuit according to any one of claims 1 to 7, wherein the phase shift is an odd integer multiple of 90°. Circuit according to any one of claims 1 to 8, further comprising a first low-pass filter (48) and a second low-pass filter (50) into which the demodulated measurement signals are fed. Circuit according to any one of claims 1 to 9, further comprising a first analog-to-digital converter (52) and / or a second analog-to-digital converter (54) for digitizing the demodulated and optionally low-pass filtered measurement signals. Circuit according to any one of claims 1 to 10, wherein the signal evaluation unit (56) subtracts the demodulated measurement signals from one another. Impedance sensor, comprising a measuring electrode (12) to which the complex AC resistance to be detected is connected relative to a reference potential, and a circuit (10; 10a; 10b; 100; 100a; 100b; 100c), according to any one of claims 1 to 11. Use of the impedance sensor (200) according to claim 12 for detecting a fill level of a medium (220) in a container (230), and / or as a level switch, and / or for detecting changes in the properties of a medium, and / or as a proximity sensor, and / or for distinguishably detecting different media and / or objects, and / or as a push button or as a slide or rotary encoder.

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